Experimental methods in RF Design [1st ed., 2nd print.] 9780872598799, 0872598799

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, EXPERIMEN~AL METHODS W in IR F DE!!I~ I :' N L=J Wes Hayward, W7Z01 Rick Campbell, KK7B Bob Larkin, W7PUA pubr shed by'

Basie '01VeSllQ"t on s in Electronics Chapters on iers, F I ers. Osc il ator Suporhetf!ro "

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EXPERIMENTAL METHODS • In ;

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Wes Hayward, W7Z01

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Rick Campb ell, KK7B Bob Larkin, W7PUA BRITISH liBRARY

DOCUMENT SUPPPLY CENTRE

12 NOV 2004 Edi tors: Jan Carman, K5MA Steve Ford , WBBIMY Dana Reed , W1LC Jim Ta Jens , N3JT Larry Wolfga ng, WR 1B

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CONTENTS

Conlents Pre fa ce

I G ... t1ing S tarted 1.1 Expe rimenting, " Homebrewing:' and the Pu rsuit of the New 1.2 Getting Sta rted - Rou tes for the Beginning Exp erim ent er 1.3 Some Guide lines for the Experimenter 1.4 Block Di agrams 1.5 An lC Based Direct Co nversion Recei ver 1.6 A Regenerati ve Rece iver 1.7 An Audio Amp lifier with Discrete Transisto rs 1.8 A Direct Conversio n Receiver Using a Di screte Co mpon ent Produ ct Detector 1.9 Po wer Supplies 1.10 RF Measurem ents 1. 11 A First T ransm itte r 1.12 A Bipo lar Transistor Po wer Amplifier 1.13 An O utput Low Pass Filter 1.14 Abo ut the Schematics in this Boo k 1 Ampli fier Des ig n Basi cs 2.1 Mod eling Simpl e So lid State Devices 2.2 Amplifier Desig n Basics 2.3 Large Signal Amplifiers 2.4 Ga in. Power. DB and Impedance Matching 2.5 Di fferential Amplifiers and the Op -Amp 2.6 Undesired Amp lifier Characteris tics 2.7 Feedback Amp lifiers 2.8 Bypassing and Decoupflng 2.9 Power Amplifier Basics 2.10 Practi ca l Power Am plifiers 2.1 1 A 30-W - 7-\ fH l Po wer Amplifier

_\ f illers a nd Im peda nce "al ch in ~ Circ uits 3.1 Filter Bas ics 3. 2 T he Lo w Pass Filter . De...ign and Exten sion 3.3 LC Ba ndp ass Filter s 3.4 Crystal Filters 3.5 Active Filter s 3.6 Impedan ce :\fat ch ing Networks ~

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Os cilla to rs a nd F re q ue ncy Synthes ts 4.1 LC-O"'cill ator Basics 4.2 Practical Han ley Ci rcuits and Oscill ator Drift Compen satio n 4.3 Th e Co lpit ts and So me Other scillarors 4,4 No ise in Osc illato rs 4.5 Crystal Oscillato rs and VXO s 4.6 Voltage Controll ed Oscillator s 4.7 Freq ue ncy Syn thesis 4.R The Ugly Week ender, MK-JI, A 7-MHz VFO T ransmitter 4.9 A Ge neral P ur pose VXO · Ex tend ing Freque ncy Sy nthesizer

S

~Ii\:l.'rs

a nd Fr eq ue ncy Mult ipli ers

5.1 Mixer Basic s 5.2 Balanced Mixer Concepts 5.3 Some Practi cal Mixers 5A Freq uenc y Multipliers 5.5 A VXO Tran smitter Using a Digital Frequency M ultiplier 6 Tra nsmitters and Receivers 6.0 Signals an d the Syste m... thai Proce ss Th em 6 .1 Recei ver Fundamenta ls 6 .2 IF Am plifiers an d AGe 6.3 Large Signals in Rece ivers and From End Design 6A Local Oscillator Sys te ms 6.5 Recei ve r" with Enhanced Dyn amic Rang e 6.6 Tra nsm itte r and Tr anscei ver Design 6. 7 Freq ue ncy Shift". Offsets a nd Incre menta l Tuning 6. 8 Transmit-Receive Antenna Switch ing 6.9 The Lichen Tr anscei ver: A Case Study 6. 10 A Monoband SS8 fC"'! Tran sceiver 6.11 A Portable DS B /CW 50 MH I Station 1 :\'ea'iu r em t'nt Eq uipme nt 7.0 Measurement Basics t . I DC f\ tesaure ments 7.2 The Osc illo...cope 7.3 RF Power Measure ment 1..1 RF Power Measurement with an Oscilloscope 7.5 Measuring Freque ncy. Ind uctance . and Ca pacita nce 1.6 Sources and Ge nerators ' 1.7 Bridge s and Impedance Measur ement 7.8 Spectrum Analysi... 7.9 Q Mea surement of LC Re sonators 7.1 () Crystal Measureme nts 7. J I Nuisc and Noise So urces 7. 12 Asso rted Circuits 8 Direct Co nversion Recetvers 8. 1 A Brief History S.2 The Basic Direct Co nver sio n Block Diag ram S.3 Pecu liarities of Direct Convers ion SA Mixe rs For Direct Co nvers ion Receivers R.5 A Mod ula r Direct Co nversio n Recei ver R.6 DC Rece iver Advan tages 9 Ph a sing H. eceivers a nd Tr ansmiU crs 9. 1 Block Diagrams 9.2 Introduct ion to the Math 9.3 fro m Mat hematics to Practice 9. ~ Sideba nd Suppresvion Design 9.5 Binaura l Rece ive rs 9.6 LO a nd RF Phase-Shift and In-Phase Sp litter-Com bine r Netw orks 9.7 Othe r Op-Am p Topologiev. Polyphase Ne tworks a nd DSP Phase Shifte rs 9.S Intellige nt Selectivity 9.9 A Next-Ge neration R2 Single-Signal Direct Con version Rece iver 9. 10 A High Perfor mance Phasing SS B Exciter 9.1 1 A Fe w Note s on Build ing Phasing Rigs 9. 1.2 Co ncl usio n

10 US.' Components 10.1 The EZ-Kit Lite 10.2 A Program Shell 10.3 DSP Compone nts IDA Signal Generation 10.5 Random Noise Ge neration 10.6 Filterin g Components 10.7 DSP IF 10.8 DSP Mixing 10.9 Other OSP Component.. 10.10 Discrete Fou rier Transform 10. 11 Automatic Noise Blankers 10.12 CW Signal Gene ration 10. 13 SSB Signal Ge neration 11 DSP Applications in Communicati ons I J .I Progra m Structure 11.2 Using a OSP Device as a Controller 11.3 An Audio Genera tor Test Box: l l A An 18-.\1Hz Transceive r 11.5 ~ S P~ 10 2-Meter Transceiver 12 Field Operation, Portable Gear a nd In tegrated Sta tions 12.1 Simp le Equipment for Portable Opera tion 12.2 The "Unfinished: ' A 7 - ~1Hl CW Transcei ver 12.3 The S7C, A Sim ple 7-MHz Super -Heterody ne Receive r 12,4 A Dual Band QRP CW Transceiver 12.5 Weak -Signal Communications Using the DSP- IO 12.6 A 28 - ~tH z QRP Module 12.7 A General Purpo se Receiver Module 12.8 Direct Conversion Transcei ver for 144-MHz SSB and CW 12.9 5 2 ~ M Hz Tunable IF for VHF and UHF Tran..ceivers 12. 10 Sleeping Bag Radio 12. 11 1 4 - ~m z CW Recei ver

Contents of CD· R0 1\l Index

.J

r PREFACE

The predece sso r for this boo k. Solid SWU' De signfo r the H" di/J Ama teur (SS D J. was first pub livhed by ARRL in earl y 197 7. T he goa l for thai rcxr was 10 prevent solid stale circuit des ig n methods to a co mmunity muc h more familiar" ith vacuu m tube met hod s. But. a no the r goal wa s inte gra ted into the text. thai of prese nting the material in 11 way that would allo w the reader to actually design his or her own circuits. Ha nd boo ks of the day pre.vented only an encyclo pedic ove rview of so lid state device s with brief qua l itative di sc ussio ns abo ut f unc tio nality. SS D described cir cu iI d eme nts in te rms of mo dels t har co uld be used for an alysis. Design consis ts of more than merely co mbining representative

circuits from a catalog or handb ook. SS!) succ eeded with design becomin g the ke y word in the title , es pecially in later yea rs as the wo rld becam e accusto med to all electro nic equi pment bei ng predo min a ntly sol id stale. Wha t surprised ma ny is tha t t he hoo k re mained po pular. eve n after ma ny of the trans istors used in the ci rcuits were no longer available. Exper imenta l Method.\ in Radio Frequency /)e_\-i~n (EMRFD ) is the seq u e l 10 SS D. with design remaining as a cen tra l the me. Our goa l i ~ 10 present moods and discu ssio n tha t will allow the use r to de sign equipm ent at bot h the circuit and the ..vsre m le vel . Our o wn i n l .: r': ~ l s are domi nated b)' rad io freque ncies . so the te n discusses problems peculiar to rad io cc mmunic ano ns eq uipmen t. A final emphasis in EMHF D is expe r-i me nt a t ln n . A vi tal pan of a n e xpe rime nt is mea sureme nt. We encou rage the reader to nOI onl y hui ld eq uipment. but 10 perform meas urement" o n that gea r a_~ it is being buil l. The word "e xperiment:' often conj ures me mor ies of sc hool exe rcises where a teacher has assem bled equipmen t and we. as st udents. go th rough a prearra nged se t of ste ps to arrive at a concl usion. also predet er min ed. Althou gh efficient. this is a poor rcp rc vcmarion of sci ence. Rather. e xperi mental scie nce be gin s with a ne w idea. An exp erim ent to te st the idea is then generat ed . the experime nt i~ built. mc usc reme nrv are made , ami the resu lts are po ndered. which ofte n result s in ne w ideas 10 test. Th is ca n all be done by o ne pe rso n work ing alone. EMRFD encou ra ges the participat ing rea der 10 build equipment with an attitu de of connnually see kin g to unde rsta nd the eq uipment and to unders tan d the p ri mitive concepts that for m the basis for Ihe equi pment and the circu its co nta ined the rei n. Our greatest hope i, that the tex t will i llustrate the potentia l of a mateu r radio. a nd ot her personal science. as a training g rou nd fo r the individual. This leu is aimed at a variety of reade rs: t he radio amateur who design s and b uilds his ow n eq uipment: college stude nts loo king fur de sign projects or wiching to ga rner practical e xperience with working hardware: young professionals wishi ng to apply the tr fresh e ngin ee ring and phy sics co urce wo rk to kitc he n tabl e projec ts: no n-e ngi nee rs want ing to dabbl e in a tec hnic al field : engineering man age rs recapt uring the fun of making t hings t inste ad o r peo ple ) wo rk: a nd technical exp lorers of alt types, The I1 N chapter of EMRFD deals wu h the problems of getting start ed with e xperi mentatio n. Xumc rcus projects arc presented, aimed at asvisti ng the e xperime nter in beginning i nvestigatio ns in electronics. Ch apters 2 thro ugh 5 then deal with spe cific circ uit functio n". Chup tcr 2 presen ts a mplif ie rs while fi lter s arc di-c uc-ed in Chapter 3, Osc i Haters e merge in Chapter 4, including

the natu ral extension of freq uency' synthesis. Mix ers, inclu di ng freq ue ncy mult ipliers, appear in the fifth chapter. Th ese chapters are laced with projects that can be co ns truc ted. but they also emphasize important basic concepu. Ch apte r 6 mo ves o n 10 prese nt cc mrmmicaticns eq uipme nt. pre do minantly us ing supe r-het e rod yne me thods. Sys tem design con ciderat rcns arc incl uded . especially with regard to distorti o n and dy namic ran ge . The ch apter cont ains se vera l proj ect s incl udi ng a high perfor mance receiver. Ch apte r 7 deal s with meas ure me nt met hod" and incl udes con sidernble test equ ipment tha t the ex perimen ter can bui ld. Chapter 8 the n moves on to a fundam ent al discussion of dir ect con version. Thiv is followed by a thoro ugh treatmen t of the phas ing method of SS E in Chapter 9. Cha pters 10 and I I prese nt fundame ntal co ncepts of digital sign al processi ng and illustra te them with projec ts. The book co nclu des wi th Chapter 12 featuring a variety of e xpe rime ntal act ivities of special Inte res t to rhe a uthors. A Co mpac t Disc is included with the boo k. Th is CD co mainv come desig n soft ware. e xte nsive listing s for DS P firmware rel ated to Chapters 10 and I I. and a sizeable collection ofj ournal a rticles relating to material pres ented in the text . The de sign so ftwar e is written for a perso nal computer using the Microsoft window s o peratin g sys tem . while the jo urnal pape rs are pres ented in Adobe Acrobat (PDF) formal. Th is boo k is a pe rso na l o ne in Ihat we have on ly writte n abo ut those thi ngs we ha ve actually ex per ie nce d. We spec ifically a void ed an e ncycl oped ic disc ussio n of materia l that we had no t actua lly ex peri e nced through ex perimen ts. Equipme nl of ime rest to the three of us do mi na te s. The amateur bands up to 2 meters arc co nside red. and are illustrated with CW and SSB gear. The book use s some math e matics where a ppropriate. It is. howe ver . kept at a basic le vel. Th e boo k cont ains numerou s proje cts th at are suitable for du plication . Pr inted circ uit boa rds arc not generally availab le for these. altho ug h boards may beco me available at a later time, Reader. should keep a n eye on the world wide web for PCB informat ion and other matter" related to the boo k. See http :// www.arrt.org/noteszxtss . We gene rally prefer tha t builders use the projects as sta rting poi nts fo r thei r o wn designs a nd e xpc rime ms rather than dup licat ing the projec ts presented.

Acknowledgments The follo wing experimenters have contributed to this book thro ug h e xpe rime nts. direct correspondence. e nco urage men t. a nd by example. We gratefull y acknowledge rhcir co ntribuuous. Bill Amidon fsk); To m Apc l. K5TRA ; Leif Asb rink, S\ f5BSZ; Kirk Baile)'. r.;7CC B; Da ve Be nso n, K ISWL: Byro n Bla nc hard. :" IEKV: De nto n Bra mwell. W7D B: Guy Brennen . K2EFB : Rod Brin k, KQ6F : Ke nt Britain . WA 5V1B: wa yne Burdick. N6 KR: Russ Carpenter, AA 7Ql; : De nnis Criss: Bob Culte r. N7F KI: George Daughters, K6GT: John Da vis. KF6 EDB: Pa ul Decker, KG7HF : Re v. Ge orge Do bbs. G3RJV ; Pete Eato n. WB9fLW : Gerry Edson. W AOKNW : Bill Ev an". W3FB:

George Fare, GJ UGQ ; Joh an Forrer, KC7WW: Dick Frey. K-lXU; Barrie G ilbert : Jack Glandon. WB-lR:\O; Joe G l a~ s , W B2PJS: Dr. Dave Oordon-Smu h. GJUUR ; ~li k e Grean ey, K3SRZ; Linley Gumm. K7H FD: Xick Hamilton. G4TXG ; Mark Hansen. KI7 N: Marku s Hansen. VE7CA: :''It:il Heckr : Ward Helms. W7S MX: Don Hilliard. woew. Fred Ho llt:r, W:!EKB: Ro bert Hughson: Pete Juliano. W6 JFR ; Hill Kelsey. N8 ET ; Ed Kessler. AA 3SJ : Paul Kiciak. .\'2PK : Don KnOlls, W7HJS: O. K , Krienke: Reb Larkin . W7 SLR : John Lawso n, K5JRK : Roy Lewallen . W7EL: John Licb cn rood. K7RO : La rry Llljcqvis t, W7SZ : R. F. Logan Jr.. \VB2NRD; Step hen Maas. W:,\ Vt lJ : Chuck .\1aeCluer. W8MQW; Jaeob Makh inson. N6NW P: Ernie Manly. W7 LHL: Dr. Skip Marsh. W6TFQ (sk I: Mi ke Michael. \\' 31 5: Jim ~I i le s . K5CX: Dave New kirk. W9VES: Ga ry Olin ·r. WA7SH I; Paul Pagel. NIFB : Dave Robert s, G8K BB : ~I i k e Reed. KLl7T S: Don Reynol ds, K7DB A (sk J: Dr. Ulrich Rohde. KA 2WEU : Dr. Dave Rutledge, KN6EK : Tom Rousseau. K7PJT : Bill Sabin. WOIYH: Tom Scott. KD7DMU: Marty Si nger . K7AY P: Derry Spittle. VE7QK: Fred Telewsk i, WA7TZY: Paul Wade . W IGHZ: AI Ward, \V5L UA : Dr. Fred Wei ss: Jim Wyckoff, K3BI : Bob Zavrcl. W7SX: Rob Zulins ki. WASM A\-1; We have certai nly missed some fol ks in our list. Please acc ep t our apo logies for ou r ove ....ight and ou r than ks for your help with

the hook and rela ted e xperi mcnrs. So me fol ks have made special contribution s and deserve spec ial thanks. Co lin Horra bin. G3SBI, Harold Johnson. W4ZCB : and Bill Carver. W7AAZ, collectiv ely fo rmed the "Triad:' a gr oup build ing the high pe rfo rmance transceiver partially descri bed in Chapter 6. We sincerely appre ciate the ir willing ness to sha re their efforts and results with us . Thanks go to Roger Hayward . KA7EX\l, for build ing so me eq uipme nt de scri bed in the book as wel l as helping with field testi ng of numero us design s. Jeff Damm, WA 7MLH. deser ves spec ial thanks for his effo rts. He built equipment describe d in SS/) and provided encou rageme nt for this version , Special thanks to Merle Cox . W7YOZ. and Jim Dave y. KRR Z, for sev eral decades or bouncing arou nd radio ideas. building the second prototypes. and manning the distant station for co untle ss experime nts . ve ry special thanks, arc exte nded to Terry White. K7TAU. Ter ry did high qu ali ry PC layouts fo r several of the de sign s presented in the text and in earlier QST artic les. He also built some equ ipment shown in th e book and pro vided meas ure me nt ass ista nce on several occasions. Special mention should be made ofthe efforts of the late Doug De'vlaw. W IFB. As co-author of SSD, he provided interest and encouragement ror th is sequel. On e of Dou g' s greatest qu alities wa s his intense. since re: interest in rad io communications. He desi gned and built rad io equipm ent. used it on the air . and then dearly wrote about the effo rts, establi shin g a stand ard for all I(l foll ow . We missed him often thro ugh the ge neration of this text. Finally. we wan t 10 thank our famil ies. and especially uur ....-i ves: Charlene (Sh on) Hayward. Sar a Rankinen. and Janet Lar kin. A book requ ires time and intense effort that often detracts fro m other activities. Our "be tter halves" have alltolerated these moments of distrac tio n.

About the Cover Ph o t o g r a p h The cover ph otograph i ~ an experimental 2.4 GHz Ie dir ect conversion receiver front-end on a gallium arseni de die . The die is a litt le more than one millimeter wide. and less than one millimeter high . Gold-bond wires con nect to the metal squares around the edg e. Th e large vpiral is a qu adrature hybrid coupled inductor. and the match ed inductors at the top are in a Wil kenson

.I

splitter. The passive ci rcuitry is simi lar to Fig 939. and the pho tograp h on page 9.43 shows thi s 1(' co nnec ted to baseband cir cuitry desc ribed in Chapter 9, Note the call signs on the die. "r-.1AL,'· wh o was not licensed in 200 1, is no w K7t\.1T1. Photogra ph hy Dean Moruhei.

Abo ut The Aut hors All thre e of the authors share a s imil ar early exposu re to rad io. obta ining an amate ur licens e as a teen or ea rlie r. Th ey all started with the novice cl ass licen se. The ir ca rly ham ex peri ences expan ded to become car eer s in science and electronics. All three are members the IEE E Micro wave Theory and Techniques Society and havc pub lishe d exten sive ly in a wide var-iety ofjournals and books. All three writers c ontrib uted to all cha pters of thi s text, but each author had a primary re sponsibili ty listed bel ow

or

Wes Ha y w a r d , W7Z0 1 Wes rece ived a BS in Physics from Washingto n Stat ", Un iver sity in 196 1 and an 1\1S EE fro m Stan ford University ill 1966. He worked on electron dev ice physics at Varian Assoc iates, The Boei ng Co., and Tektro nix. He then did Rf c ircu it design. first at Tektro nix and the n at T riQu int Se micond uctor. Wcs is no w se mi-ret ired . dividing his time betwee n writing a nd co nsulting. wcs "vas the prima ry contrib utor to C hapte rs 1 throug h 7 and large part s of 12 and c an be contacted at w7zoi@arr l.net .

Rick Ca m p bell, K K 7B Rick received a BS in Physics from Seaule Pacific Uni verviry in 197 5. aft er two years act ivc duty as a US !\' avy Rad io man. HI: worked for4 years in crys tal phys ics basic research at Bcll Labs in Murray Hill, NJ before retu rning to grad uate school at the Unive rsity of Washington He completed the MSEE deg ree in 198 1 a nd the PhD in EE in 19 H4 . He ser ved on the faculty at Mich iga n Tec h University until 1996. Since 1996 he has been with the Advanced Deve lopm ent Group at T riQuint Semicon duc tor, dexign ing microwave receiver cir cuitry , Rick had primary re sponsibility tor chaprers x. 9. and large parts of 12. He can be c ontacted at kk7b @llrrl.net.

Bob Larkin, W7 PU A Bob rece ived a BS in EE fro m the Univers ity of Wa shington and a .\ l S in EE from New York Uni versity . He work ed for 12 years at Bell Labs in New Je rsey in areas of circu it des ign and signal processing. I n 1973 he and his wife Janet started Jan el Labs where a variet y of radio freq ue ncy products we re manufactured , They moved the com pany to Co rvallis Oregon in 1975 where it operated unt il be ing acquired by Cetwave RF in 199 1. He now works as a co nsulta nt speciali zing in microwave circ uits. Bob was the prim ary contributor to Ch apters 10 and I I and wrote a sec tio n in Chapter 12. Readers can contact Bob at w7pua@ a r r 1.llct.

CHAPTER

Getting Started

1 .1 EXPERIMENTING, "HOMEBREWING," AND THE PURSUIT OF THE NEW Amateur Radio i~ a diverse a nd colorful a voc ation or h ubb y w her e the pa rtic ipants com m unicate with e ach other through the u- e o f rad io sig nals. T he co mm unic atio ns. whic h c an e ncompass and extend beyo nd the planet. arc often rout ine and predict ab le. but ca n a l times he et hereal. The romance of communica ting with the o the r sid..: of the world ble nds wit h the joy of observing a c om plicarcd pan o f nature. Fo r ..orn e of uv, the wo nder never di sap pears.

Although rad io ca n be fun, our prag matic soc iety de mand.. mo re tha n exci tement when re sou rces arc used. The virtue that most ofte n j ust ifies o ur use of the radio spect r um is the gro wth of a proficient co mm unica ti o ns vys tem tha t can be ca lled upon in limes of emergency. The e xample s of its use are numero us. But. "ha m' radio is mo re tha n this . II is a te ch nic al a voc atio n of d iverse ed ucerional pote ntial. It has values that go we ll beyo nd that o f a supple mentary co mmu nications network. Most radi o ama te urs have a n inte rest in the tec hnical details of the eq uipm ent the y usc . Historic ally . Ihis was a req uire me nt; The only way a rad io a mateur coul d assemble a n ope rat ing station war.. to person a ll y build his o r he r gea r. Co mm erc ial eq uipment was rare, a nd was often pro hib itivel y expe nsive, HUI today. high quality "ham" gea r is readi ly a vail able in Il1O.-

Ampli f i er .

Provi d es net po wer gai n .

Mixer _ Pr o vide s an out pu t f r eque ncy th at is a s urn/ di f o f i np ut fr eq uencie s. Os ci l l at or . Gene rat es an out put at a s i ng l e f r equency.

~ ~

Combin er/ Sp l i tt er . Adds t wo s i gn als or s pl i ts on e i nto t wo p ar t s whi l e i s ol ati ng t hem.

[):::

I np ut s / out p ut s . Coax ', s p e aker , microp hon e , he adp hone s.

raJ

Low Pas s Fil t er Hig h Pas s Fi l t er . Bandpas s Fil t er . All Pas s Fil t er (Pha s e Shi f t net wor k) Fig 1.5-Comm on bloc k di agram el em ents.

Aud i o LC!HPF LC!BP F

Re ce ive r~

I nput

High Gaio

RF

AUd io Amp ,

Aud i o LC! LP F TX

-.tR e y

,', ' to use and offers good per-

M dlO "",,~ l ~ fiH

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Fig 1.7-Bloc k di agram of a super-heterodyn e 5SB tran sceiver.

Re ce lv« r

raput

;-;

11) \.; I.e / BFf

.,,,

50-50 .3

Fig 1.8- Blo c k d iag ram of a phasing method SSB tra nsceive r.

0$ are not available . fou r pa rall el 200-0 I_W parts will wo rk a-, well . Th e resulting Rf vo ltage is rectified with a silicon switchi ng diode. T his sho uld be a I OO· V pan s uc h as the I :"i41-t8. 1S-t152. or simila r diode. Th e voltmeter pa rt n f the circui t is a 20-kQre"istor driving a U-} rnA met e r.

P (m illiwatts)E 10 ( V ... 0.7 )2

(1 H scale ) DC Volt.

4 Wa t t Input

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,. "" 50 mW

Input

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1. 51
u s o U( ou t )

1 _6u5

1. 1 us

1. 8u s

1_ 9 us

2 _0u s

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Fig a.ar-cc ouectcr (upper) and o utput load (lower) voltages with the p i network output circuitry.

will be the same . O utput voltage ca n. howe ver. increase as R I. grows . T o ohtain the maximum power o ut put. we wis h to pick a load that allows the co llector vol tag e to drop nearly to the ba se value (satura tio n)

whi le go ing an equal dis tan ce ab ove Vco at the op posite part ofthe cy cle . This voltage excursion should occur as the current var ie s from twice the bias value down to zero. The load resista nce that allows this i s

Eq 2.22

where If. is the de bias valu e. A mo re fam iliar fo rm e xpresses the load in terms of a desired o utput po wer ,

Eq 2.23

whe re R L is the load res istance i n O hms. V ce is the po we r su pply. VB is the DC base

desi g ned. Rather. he or she wishes to measure the amp lifi er o utp ut with 50·n invtru me ntation and perhaps driv e o ther ci rcuit s with a 50- U impeda nce. T he solution is fo und in FI~ 2.35 when' an imped ance transfo rming It-network is in...erred betwee n the 50-0 load and the co llector. Th is net work mal e" the termination -too k like" 1000 n at 10 MHz. 11 also has low pas.... filte rin g cha racteri stics. attenuating energy at 20 MHz , 30 Mil t , and higher harmonic Freq uenc ies. Fig 2.36 shows the collec tor waveform whe n the 50-U load is co nnecte d di rectly to the collec tor. T he wave fo rms a fter matching are show n in Fig 2.37 .

voltage. and Pout is the o utput i n Watts. T his form applie... to Cl ass B and C amplifiers as wel l as the class A am plifier under d iscuss io n. Applicatio n of Eq 2.22 predicts a load resis tanc e of just over 1000 n for max imum ou tpu t- C hanging the load to I 1>0 in the circ uit prod uces a ]().MHz o utpu l of 11 V pe ak-to-pea k cor respo nding to a po wer of abo ut 16 mw. Eve n larger resistance would ha ve prod uced voltage li miting, so this is close to o ptimum. More ofte n than not. 1000 n is no t the impeda nce that the desi gner wishes to use as a terminat io n fo r the a mplifier ju...t

2.4 GAIN, POWER, DB AND IMPEDANCE MATCHING Aud io and o the r lo w freq uency amplifi ers a rc easi ly ana lyzed with the low freq uency model' used fo r bia...ing. But m01>1 of o ur interes t is i n hig her freq uenc ies where meas urem e nt diffic ult ies per sist, These e nco urage us to c o nsider power instead of the vo ltages and curren ts that d ominate the vie w of Ihe c ircuit theorist. T his em phasi s is a n integra l pari of RF desig n and for ms the haf- is fo r this sect io n. The emphasis on power measurement goes back to ear ly methods. Power at radio. microwave, and even optical frequencies was measured using a Bolometer. The Bolometer i~ based upo n temperature measurements. A resistive load if- embedded in a thermally well-insulated chamber. The application of RF po wer cause s a temperature increase, \\, hich can be detected with a thermometer. But. the same increase in temperature ca n be prod uced with application of direct current. Steasurement of the direct current and related voltage then provide a very fundamental

n- sour ca

v

rv

,

•,i

p er)

,.,, ,-•

R

Fig 2.39-A vo ltage with a source resistance Rs deli vers power to a load R.

,

R-l oad

r-'\Nv'" """ . V- g e n .

R- sour ce

Cons ide r the simple circ uit of t"ig 2.39 co nsisting of a voltag e source . V, and

s tep Att e nu ator

ciency as a fu nct ion of the term inating re sistance. A similar plot is give n in Fig 2.43 wher e powe r is now plott ed against reflectio n coefficient, r. Although reflection coeffi cient, l", may seem like an esot eric impractical parameter. it is easily measu red (in magnitude) using a simple appar atus that can be built in the home lab. This circui t. show n in Fig 2.44. is call ed a rerurn loss bridge, or RLB. The three resistors in the hridg e are SO ,n when huilding a hridge for usc in a 50-11 system , The signa l generator is assumed to then have a SOon impedance as well .The transformer is ncomilion mode choke (see Chapte r 3.) Construc tion is d iscussed in Chapter 7. Th e brid ge act ion occurs becaus e a ll resistor s are 50 n , Assume that the "X" port, the unknow n, is terminated in SO n . The n half of the voltage app lied at the "RP" port ap pears at the junction of R l and R 2. But half also app e ars at the " X " port. T he voltages are eq ual on ei the r side or the common mode tran sfor mer. so no signal appear s at t he detec tor. In contrast. a larger sign al appe ars when the unknow n "X" port is eith er o pen or short circ uite d. Use of the return los s bridge is presented in Fig 2.45, where an amp lifi er inpu t will

j "-

-oet.»

:

Fig 2.44-Return Lo ss Bridg e. All resi st ors are norm all y 50 Q .

[>

50 ohm Ter mill a t i o n

/-~

Oscill oscope

50 Ohm Te r mi nat ion

Fig 2,45-Using a retu rn loss br idge wit h an amplifier.

be measured. The bridge is first ope n cir cuited at the " X" port. and the de tec tor response is not ed. T hen, a 50-12 term ina tor is pl aced un the "X " por t. A large decre ase in detector res po nse sho uld he not iced . T his respons e is a measure of how well the RLB is Iuncuo ning and is called the bridge dire ctivity. An amplifier (po wer on) is now attached to the "X" port through a coa xi al cab le, and a termi nato r is atta ch ed to the amplifier o utput. Th e detector res pons e will be lo wer than the level pre sen t with the "X" port ope n c ircui ted by a rat io ca lled the return loss , a dB value. The ste p uue nuat or in the detector can be adjusted to atten uate the re ference to better measure return loss. Return los s is related to F thro ugh R. L.

=

- 20 · Log r

Eq 2,3n

The inverse form is

- R.L.

f = 10 20

E q 2.31

While we ha ve illu stra ted the RLB with osc il loscope det ection, a 50 -n power meter or spec trum analy zer is prefe rre d. Both arc descri bed in C hapter7 . T hese are so-n instruments, so they do not require the e xternal termi nator so vital to the osc illoscop e. T he 'scope suffers from two problems th at co mprom ise this application. First. it is a wide ban d in strument. so noise li mits the se nsitivity, making it diffic ult to see the wea k sign als th at arc readily seen in a spectrum ana lyzer. Seco nd, ma ny of the ter min ations tha t we migh t mea sure are narrow ban dwi dth loads . As such. they will produce high re turn loss at one f req uency. but not at the harmonics . T he usual sig nal generato r is harmo nic rich ,T he harmo nics are reso lved and, hence , ign ored in a spectrum analyze r measurem en t.

2 .5 DIFFE R ENTIA L AMPLIFIERS AND THE OP·AM P The differenti al a mpli fier, or diff -cunp: is the fou nd ation for mo st silicon analog integ rated circuits in usc today, making it a very imp ort ant topology . Here we investigate diffe rential amplifier fu ndam entals and e xamine a major derivat ive of it. the operatio nal ampli f ier, or op -amp . fo llow ing the nam e, the differenti al amplifier is a circuit inte nde d to amplify a diffe rence . T he di ffere ntia l amplifier has two in put terminals. T he output, which c an be between two collectors or from just one,

2. 1 6

Cha pter 2

is the n proportiona l to the voltage (or c urrent) difference betwee n the input s. The basi l" differen tial amp li fier usin g NP N bi polar transistors is present ed in F ig 2.46 . We start with two ide ntic al transist ors biased at the same de has e vo ltage. The two emitters are attached and re turne d to gro un d through a co mmon re sistance . as in Fig 2.46A. Two ide nti ca l co llector resis tors are attach ed, bi ased from a corn mon s upp ly. This circuit can hav e sign als appli ed i n IWO ways. If the two bases arc

d riv e n toge ther. th e composite c ircuit wou ld beha ve as one tra nsistor. The two collector sig nals woul d the n be identi ca l. Thi s o peration is ca lled common-mode driv e or exci tation T he large e mitte r resistor beco mes a deg eneration e lem ent. cau sing the commo n-mode gain to be lo w. T he oth er d iff-am p d rive is the differentiel-mode . where one base is driven in o ne direction whil e the o ther is driven by an opposite polari ty Ass ume that Q l a nd Q 2 arc biased with a de base vo ltage of 5 The

vo ltage at the co mmo n e mitte r is then -IA. T o ta l current will be 4A rnA for an e mi tte r resi stor of 1 k r.!, If the two transistors are identical. each will be biased 10 an emitter current of 2.2 rnA . We now apply a d iffe rential si gna l causing V bt to i ncre ase by 10 mV whi le V h2 drops by a n eq ua l 10 mv. The emitte r voltage re ma ins evvenlia lly cons tant. Vel dec reases while V c2 incre ases by a n a mount rela ted to the gain. A usefu l pro per ty of this c irc uit is that tot al current docs not ch ange wit h d iffe rcmial drive. Fig 2046. pan B sho ws the cir cuit varia-

lio n fo und most often in integ rated c ircu its whe re the e mitter resistor is rep laced by a third tran sistor . Se t V b ~ 10 2 volts a nd pick the Q3 emi tter resisto r fur the sa me 4 .4 rnA. Thi s leav es bia s con d itio ns for Q ! and 02 a ~ the y we re. altho ugh the co mmo n mode gai n is e ven lo wer. Q3 is a constant current source, a cir cuit that ac ts as if the bias fo r Q I a nd Q2 ca me f ro m a ver y large negati ve pow er supply with ,HI eq uall y large resistor. Th e effect of this topology is to fo rce the sum of the cu rrents in Q I and Q 2 to remain constant. Thi s has two importan t co nseq ue nces.

V~

;." 1 1~2

01

Vsm' Vc1

Vc2

01

02

02

+Vcc

Vb.

Vbl

VOl

VOl Vb3

Fig 2,4 6-D itferential Amplifiers.

1K

First. a diffe re ntial a mplifier is very easy de couple . With co nsta nt tot a l cu rrent. sign als are not injec ted o nto the VCo powe r supply, very important when the diff-amp is on e of mall)' such ci rc uits wi thin an Ie. The other co nseq ue nc e of the constant c urre nt so urce is that drive applied to j ust o ne input will resu lt in diffe rential o utp ut s igua ts. T his is shown in the am plifier of Fig 2...7. The two co llector voltages ha ve equal a mplitu des and lire ou t of phase with each othe r. Althoug h d iffe rent ial a mplifie rs are ab undant in in teg rated ci rc uits. they a re a lso usefu l and pra ctic al in d iscrete fo rm . Fi g 2.-18 sho ws a d iff-a mp with readi ly availa ble pans that might be used to pro vide balanced local osc illator dr ive to a mixer with ou t tran sfo rmers. T his ci rcuit is 10

03

-v

1

Fig 2.49-Schematlc d ia g ram fo r an o pe ratio na l a mp lifier .

Vcc =10 1K Ve l

RFC

. -_ _ Ve 2

1K

1K 5 V

-Vee

Q3 2 V

~ ~

0 .1 ,

1K

Vc c=1 0 RFC Vel Ve 2 Iq"Q2

~

J6' 0.1 «

1K

V-bb 318

Fig 2.47--oiHere nt ial Amplifier that converts a single e nded si g na l into a d iffere ntia l one hav ing two o utputs with a d ifferenti al re la tionshi p. The 2 and S·V poi nts are fixed vo ltage, usually ge ne rated wit hin the Ie containi ng thi s diffe re ntia l pair .

Fig 2.48-0ilferential Amplifier built with discrete c o mpo ne nts. The emuter re s is tors a re a d jus ted for e q ua l c urrent in t he two tra nsislo rs. VI>tI re pre s ents a base bias po wer s u pply, which could be a simple vo ltage di vider fro m t he higher s upply.

Amplifier Design Basic s

2.17

useful becau se it pro vides a ba lanced ou tput with redu ced even order harmonics as well as power ga in. The use of two em itter resistor s cases the need to have identic al tran sistors . Having examined properties of the diffamp, we will now look at the "ultimate " diff-amp example, the operat ional ampli fier. An op-amp is shown schematic all y in Fi g 2.49 . The intern al circuitry can be ra ther complicated; famili ar exam ple s such as the 741 or 358, will include a doze n or more tra nsi stor s wh ile high performance variants will have many more . The operat ional amp lif ier (Fig 2.49) is shown with two power suppli es, altho ugh virtu ally all can be used with a single supply . The basic ope ration is, in some way s. exactly like the sim ple diff -am ps dis cussed above. The op-am p has two inputs j ust as the diff-amp has two base inputs that effect their outputs. The usual op-a mp, however, ha s just one, single en ded out put. More over, the output vo ltage can be either above or be low the inp ut voltages. The usua l up-amp has sever al ga in stages, all cascaded with the output of one feed ing the input of the next. As such, the low fre quency voltage gain is oft en very high wi th valu es ranging from 50 .000 up to over one mill ion. While op -am p gains are often expre ssed in dB (using the familiar 20 *LOG(VoutlVin) formula), this is often incorr ect. The dB form only pertains to po wer ratios . The equati on rel ating vo ltage ratio is valid only when terminat ing impedances are equ al. A typical op -amp can provide ou tpu t voltages from near the ne gative pow er supply up to wit hin a volt or two of the posit ive supply. The inp uts can also occur at a wide vari ety of volt ages . A 74 1 op-amp will work with inputs that are from about Vee +2 10Ve.,_- 2. This spa n is called the commo n mod e input range. Op -amp s using PNP bipo lar input tran sistor s can have a common mode input range that extends all the way 10 the negat ive supply. Examp le s incl ude the LM-324 and LM-35 8, wh ich arc: especially usefu l with single pow er suppli es . Ass ume that the "-" input in fig 2.4 9 is consta nt at ground with power supplies of + 15 and -IS volts. Set the "+" input scv era! volts nega tive. The output will then be very neg ati ve, as low as it can go. As the "+" input is inc reased. the output rem ains negative until the inp ut gets cl ose to ground . Th en, the outpu t will start to increase very quickly. Th e ou tpu t goes above ground as the "+" input becomes just a few millivolts pos itive. The vol tage gain may be ev alu ated from a curve of the ou tpu t vs the inp ut. Wit h eve n modes t inputs. thc output reach es the positi ve

2.18

Cha pt er 2

power supply, or "r ail." The "+" input is call ed the no n-in verting input for the output po larity follows it in direction. Cir cuit op eration is simi lar if the noninv erting ("+") inp ut is grounded and the positive going signal is applie d to the "-" or inv erting input, except that now the output moves in the oppos ite direc tion. That is, the output makes a trans ition from the pos itive power supply to the ne gative one . Repeati ng these exp eriments at refe rence voltages other than ground shows that the output depe nds onl y upon the milage difference between inputs. The input transistors for most op -am ps arc biased for low current opera tion, causing the input impedanc e to be quite high. We usually neg lect R;ndurin g the analy sis of op -amp circ uits. Op-amps are rar ely ope rat ed "open loop ," as described abo ve. Instead, they are used with negat ive feedback. This is illustrated in Fig 2.511. Powe r supp lies are omitte d in the op-a mp circu its that follow, but are assu med to be + and - 15 vo lts. Assume init ially that the "+" inpu t for Fig 2.50 is at gro und. If the outp ut was at a diffe rent vo ltage , the invert ing input would then be at a le vel other than gro und. This wou ld then produce a difference vol tage at the invert ing input that forces the input toward ground. Increas e the non-inverting input to +1 volt. Sim ilar argu me nts sho w that the output increases until the inverting input is also at + 1 vo lt. The circ uit of Fig 2,50 is a

Yi n

voltage foll ower with a gai n of +1. Th e val ue of the feedb ack res isto r is of no conseque nce for this circuit. for the inp ut current is very small . (A practical unity gai n foll ow er normally ha s the output shorted to the inverting input.) Thc mod ification in Fig 2.51 adds an equ al val ued resistor from the inverting input to gro und. Sett ing Yin In a for ces the output to grou nd. However, when we set the inpu t to +1 volt , we find that the outp ut move s to +2 volts. Our circ uit now has a non-in verting gain of 2. Thi s is co nfirmed thro ugh voltage divider act ion. The voltage at the "- " input must be half of that at the outpu t; a voltage other than + I at the "-" input would produce an inp ut diffe rence th at wo uld mo vc the outp ut. Fig 2.52 shows an invertin g amplifier. The "+" input is grou nded with an inpu t applied to a res istor attached to the inverting input. \Ve star t wi th the amplifier inpu t at gr ou nd. The ou tp ut must then be at groun d . fncreas ing the ex cit ation to + \ volt caus es the in verting input to "try" to go po sitive, an action that is inverte d with gain in the op-amp. The syst em is in equilibr ium when the output is - I volt. The amp lifier (hen has an inv ertin g gain of I. A general beh avior has emerged from this dis cussion, cas ing further anal ysis : Negative [e edhack aro und all on -amp always has the effe ct offorcing the l1-t'o inputs to have the same voltage. This can be used to derive the usua l formu las for ga in of closed loop amp lif iers. The char-

+

l OOK Fig 2.50-A unity g ain f oll o wer.

I

JcJ\A = lOOK

Fig 2.5 1 A f o llo wer with a g ain greate r t han u n ity.

~,P

~

Yin

~r/ AA

{OOK

Vi n1 V

lOvOvK

i n2

lOOK

l OOK

l OOK

Vi nl

lO OK FIg 2.52-A n m v ert m g a mp lif ier WIth un ity g ain.

FIg 2.53-A s um min g amp lif ier With th re e in puts,

/'

E

ou t

1'--"

~ ~-

R • v

1

Vi a

'\.

/ VOd R, ...

o t

R, '-~

Fig 2.54-Feed back redu ces an outpu t resistance.

ucte ris tic is mai nta ine d so lon g as all inputs and outp uts arc maint ai ned wi thi n the afluwed rang es.

The invert ing input of a dosed loop amp lifie r is o ften described as a "s ummi ng node:' ill ustrated in l'ig 2.53 with three inp uts. All thr ee ha ve the same input resisto r va lues. so the ga in fo r each input is the 2'C JK

Th~

C SER C PAR =-

tor is res ona nt at Freq uencie s where the line is 1, 2,3 . etc wavelengths long. Anot her popu lar structure for higher frcquenci es is the helical resonator. These were very popu lar for UHF FI\1 mohi le radios of j ust a few year s ago . A helica l resonator is a section (usu ally OIlC quarter wavelength) of line using a helical trans mission line. A helical line is a soleno id coil -like structure placed inside a shielded enclosure . we can think of a wave as propa gating alo ng the wire at the speed of ligh t. He nce, the propagati on veloci ty parallel to the H is is much less than that of light. This is a slow wave struc ture . Cutt ing a quarter wav elen gth sectio n, grou nding one end with the other open cir cuited. form s a resonator. The usua l helical reso nator is just under a quarter-wavelength long. The extra length required for resonance is COI11pcn satcd by adding a small adj usta ble capaci tor to the end, often nothing more t han a grounde d metal screw d ose to the "hot' end of the cen ter conductor. Nume rous re vie w art icles ha ve appeared describing the helical re son ator and filte rs using them . Equat ions are oft en gi ven for resona tor dimens ion s, an implicat ion that they must con form to a well def ined stru cture. Generally, there is much greater freedom ava ilab le to the builder. A helical filtcr may still work well if bu ilt in a volu me that is "too sm all."

Fig 3.2 3-Ba n dpass f ilte r using se ries tu ne d c irc u its. In t h is example , N=4.

[t

-

1

- 2·CJK ' C SER

---c;c--

CJK

-

-

1 t t -

Fig 3.24 -Double-tuned c ircu it w ith a sh unt cap ac itor for coupli ng between reson ators. Thi s illus tr ates o ne of nu me rous ba ndpass filte r topolog ies that are mixtures of the two methods p rese nted.

Fig 3.25-A q uarter wavelength of transmission li ne fo rms a resonant tuned circuit.

Fig 3.26-Load ing (coupling to the "outs ide world") ca n be contro lled with sma ll wire ind uct o rs.

Filters and Impedance Matching Circuits

3. 15

A casua l glance may not reveal a true identity. That is, a heli cal reso nato r with a tuning capaci tor looks like a shiel ded LC resona tor. Ho we ver , the difference becomes clear if wid ehand measurement s are done with loo sely coupled probe s like the one s that have been descr ibed for Q measure ment. Suc h measur eme nts will show a high Q at the fundamental frequency and addit iona l responses (als o ha ving high Q ) at 3. S. and other odd harmo nics of the fun damental. l n contrast , a pur e LC resonat or will not show these depa rtur es If capacitance is added to a helical resonator to decrea se tu ndam ental freq uency. t he higher freq uencies will nut move as fast. Slight cap aciti ve loading might mov e the first "sp uriou s response" to 4 Fu with greater departure as loading grows . Q rema ins high and excellent filte rs can sti ll be built. Helic al resonato rs are coupled 10 each other with a variety of meth ods. although the most popular is throug h apertures . or holes in the walls betwe en adjacent reso nators. As wi th oth er filter type s, the coupli ng can he re lated 10 th e frequ enc y spread be twee n peaks when the resonators are unloaded. End section loading is realized in a vari ety of ways with helical rc sonators A small line from a coaxial co nnector can bc tap ped onto the helix , The

3 .16

Chapter 3

Line W idth - -j

-

f----

X1

G'\

X3

X, Fig 3.27- Three

co,onator Hairpin tvp e bandpass filter.

w e hm Lin e width 1

• X; I

f-1 X5

usual tap point is very close to the gro und ed e nd, often a small fraction of one turn. Aga in. the loading may he adju sted to establish an end section loaded Q. We have on ly scr atc hed the surface with some filte r types we have built. A detailed re view of the literature will reveal num erous other filter topologies of inte rest. The bandpass filters presented here are transformed from simp le lo w-pass filters , the so-called all-pole low -pass circu its with not hing mo re than series ind uctors and shunt capa cito rs. Other low -pa ss fillers

such as the Elliptic can be transformed to hand pass form 10 ge nera te bandpa ss circuits with transmission zero s nex t to the pas sband. Another varia tion inje cts a transmissio n zero in a passband with no additional inductors. This is realized by an additional coupling ca pac ito r that co uples en ergy betw ee n no n-adj acent reso nato rs. Thi s method was use d in a 144 MHz transceiver discussed later in the boo k.!- There is a great dea l of work available to be do ne by the curious experimenter.

3 .4 CRYSTAL FILTERS No element is more intimately refuted to rad io rece ivers tha n the quartz c rystals use d in filter s. The early supe rheterodynes of the 19305 obtained singlesignal selectivity with a crystal filter using but one crystal, a practice that con tinued through the 19 70s. T he use ofhigh qua lity fillers using a multiplic ity of crystals becarne popular in the 1950s as SS B replaced cla ssic AM as the rad iotelephone method of choice.

Crystal Fundamentals A mo dern q uartz crys tal is usua lly a ro und dis c of single crystalline q uartz with mctalization o n each side . T he metal films serve to create (a nd se nse ) an electric field within the qu artz . The basic structure is sho wn in Fig 3.28. Th e basis for the interesting circ uit properties of a quartz crystal is the piezoelec tric effect. This effect is a ma terial charac teristic wh ere an electric field cau ses a mechanical displacement. The mechanical mot ion is at right ang les to the electric field in the quartz crystal. An ele ctric fiel d occurs wh en a vo ltage is placed between the two mc tal ization layers attached to the crystal. The o pposite effect also occurs; a mech anical motio n generates an e lectric field . The action of a quartz cry stal when sub j ected to an electrical impulse is analogous to striking a bell or chime with a hammer: the energy of the imp ulse causes an oscil lation to occur, a ringing t hat dies out in time. The resonant freq uenc y of the chime is re lated to mechanical d ime nsions. In the

Ta ble 3.4 shows so me measured re pre sentative val ue s fo r som e j unk- hox crystals . A cr ystal placed between a 50-0. signal ge nerator and 50-0. load shows a re sponse like tha i of F ig 3.30. If the crystal was a simp le series tuned cir cuit witho ut the para llel capacitor, Co ' the response would be a simple peak . A crystal filte r c an bc built with a single crystal wit h the sche me of Fig 3.3 1. L-netwo rks at each end transform 50 n to present 500 0. at the crystal. Transformer T l prov ide s an om -of-phase vo ltage to dri ve a phasing capaci tor . T his signal co mbi nes with the energy flo wing through the c rystal parallel capacitance to control the posi tion of the notch . Th e lO-pF capaci to r inc re ases the eff ect ive parallel C of the crystal. moving the notch closer to the pea k while the 25-pF cap acito r resonates the ferrit e transformer . Fig 3,32 and 3,33 show the result of tuning the phasing capacitor Cha nging the terminating L-nctworks c an alter the fil te r respon se , T he han dwidth will decrease if th e terminat ing impedance is dropped. A li nk cou ld be used on T1 to replace the input L netw ork whi le an output could be terminated with another wide band transformer. The modified circuit wou ld then function we ll with a wide variet y of crys tals . Bandwidth will. of co urse, vary considerably as the com -

sa me way, the resonan t freq ue ncy of a quartz crys tal is related to the crystal thickness. T he Q of a quartz crystal ca n be very high, from 10,000 to over o ne mill ion. The motions of a quartz cr ystal arc transverse wit h the crystal vib rating parall e l to the surface. Thi s allows the Q and resonant freq ue ncy to be alte red by surface effects. The reader with an Interest in the physics of quart z crystals is referred to the classic tex t by Virg il Bouom.!' Th e quartz cr ystal is modeled as the LC tuned circuit shown in Fig3.29. L m and C m are termed "mo tiona l" parameters for they relate to the mechanical motion of the cr ystal. The equivale nt seri es resistance, ESR , is an element representing losses: it is rclated to the crystal Q. The final element, Co ' is the parallel. or hol der cap aci tance . T his C is a simple consequence of the crys tal construction as a parallel-plate capacitor . This value is the sum of the parallel pla te C (the dominant element) and some stray C related to the package housing the crystal. The parallel and the motional ca pac itance are rela ted in the usual AT cut cryst al. (AT cut refers to the cr ysta llog rap hic orientation of the crystal. Many of the crystals we deal with in radio are AT cut.) The rel ation between capacitors is app roximatel y

Co == 220· C M

0

- -_~ 1 rD~~ !yv~~_-=TJr" 1 91P~ 1-

Fig 3.31-A s ing le crystal filter us ing the c rystal of Fig 3-30. T1 is 12 bifilar turns # 2 6 on a FT50-61 ferrite toroid. This filter has a 3-dB bandwidth of

4.48 uH

Yl

.i,

1 191 P F

1 0 uH

=- ;: .~K= 0.9 9 25 . 3 pF -=

Some experimente rs have mounted the po t in a pane l and swi tche d it into the circuit as needed. Th is may give inacc urate results owing to stray indu ctan ce , T he pot should be mou nted to a suitable "dummy cry stal" with short leads. A de tai led anal ysis of the method reveals errors . Thes e can be reduced subs tantially by shifting to lowe r measurement imp edance. The test se t of Fig 334 is complete. pro vid ing both motional paramete rs and Q infor mation . Howe ver , meas urements with this apparatu s become tedious. A simple crystal os cillator c an provide the mot iona l parameters . Th is c ircuit. Fig 3.35. incl ude s a se ries capacitor that may he switc hed into the ci rcuit to pro duce a Freq ue ncy shift. Rela ted equations arc included with the figure. The requi red Ou for filter applications will de pen d upon the filter bandwidth and center frequency as we ll as on the filter shape and the number of resonato rs . A reasona ble rule of thum b fo r most filters (LC and crystal ) is that the "normalized Q" must exceed twice the nu mber of resonators . Normalized q. qo. is defined as Q u

=

_

100

240

~

1 1

5- 3 0 p F

62

0

0

0

0

a [§iJ ~~

j

(" . .

- 30 1. 9 95

- co

.

:,+

Chapter 3

~: j +

.

•

""" - [

5

Fig 3.34-Simp le test set for crystal measurement. The pad is a 20-dB, 50-0 ci rcuit. The output shou ld be term inated in 50 n. A maximum input power from the ge nerator would be abou t -10 dBm , resulting in a ma ximum to the cr ysta l of - 30 dBm. The 100-n pot is substituted for t he crystal for ESR measurement. See text. App roximate equations fo r mot ional pa rameters are;

,~ + ",+

+G~~]/

.

Q

u-

1.2 10

-a 0

Fo

5. 0 0 5

Fig 3.33-Response of the s ingle crystal filter of Fig 3.31 when the phasing capac itor is at ma ximu m val ue of 30 pF . The solid line re pres e nts the case of e xact bala nce whe n t he phasing c a pa c ito r equa ls t he crystal Co'

.F

dF a

ao 1. 9 3 5-

8

b.F . R s

,+

: _ - 30 5 .0 05

Fig 3.32-Response of the s ingle crystal filte r of Fig 3.31 whe n the phas ing capacitor is at minimum va lue of 5 pF. The s olid line represents the case of exact balance when t he phasing capacitor equals t he crysta l Co'

3.18

I - 15

. 0

rt

~1 2 , the peak frequency is given by

Eq 3.19

Fig 3.48-RC active low-p ass f ilter. Th e up-a mp is assu med to be powere d fro m dual s up p lies around g ro und. Ot her biasing schemes are presented late r. The operat iona l amplifier is co nfig u red for a no n- inverting gain of 1. C2, t he feed back ca pacitor, is A x C1 whe re A is a va lue greater than 1.

Table 3 .6 A Voltage Gain 2 ,2 1.004 2.4 1.0 14

A Voltage Gain 6.8 1.4 1

3. 3 3.6 3.9 4.7

22

1.088 1 .12 1.14 1,22

10

33 47

1.67 2.4 2,9 3.46

con nec ted from the amplifier out put to ground. T he resis tor should pass a st and ing curre nt of about I rnA. Severe cros sover di stortion wil l res ult with o ut th is loading.

2.0

.

r '. 5

~

"

/ /

-

/ .

>

.=-

1.0

-- "

/

. .'

\

Figure 3.52 shows a VCVS typ e highpass filter . This circuit is the d ual of the low pa ssju st d iscu ssed . It is de signed with equal valued ca pacitors. The resistors now differ by a factor '· A'·. The usual filters have the grounded res istor as the o ne with larger value . Fig: 3.53 sho ws the re sp unse

»> ':" ,

\ ,

~.\~ . \

0 .5

.

.

,I.~"' ".~ ... .

-

0.0 0.1

High·Pa ss Filters

,

!

0,2

0.3

I

..... .:::-.r.:.........." ~ . _"=-= ...... .

, , I I ,

0.40,50,6

0.8 1.0

2.0

3.0

4.0 5.0 6.0

, , 8.0 10,0

Frequency (kHz) V(4) -

-

V(14)··· ···· ·· ··

V(24) -

.10-

-

-

."

V(34) _ . _ .

i

'" -30

Fig 3A9-Response of the filter sho wn in Fig 3-48 with A=1, 2, 5, and 10. These curves, and severa l others in this section , were generated wit h Supe r Spice from Compact Software. The solid line corresponds to A=1 while the highest peak is for A=10.

~o1~~~~E~~~t~~~~ :~~ ----l-..1.JJ.Ll.~_._U ~~

·50

"

· 60

I

IJ t = L I I I

0.Q1

0.10

dB (V(12J)

So me va lues of lo w pass voltag e gain at the response pea k are tabulated vs A. the c apacitor ra tio, in Ta ble 3.6 . The re arc nume ro us way s 10 design practical low -pas s fi lte rs with the equalions. A c ascade of sec tion s like those in

Fig 3.48 would form Butterworth or C he byshev filters of hig h order. Ea ch capacito r c orresponds to o ne pnle in the respo nse, one L or C in the tradi tiona l filter. Gene rall y, eac h two-pol e low-pass sec tion will differ from the ot hers in higher order Butte rworth o r Che hyshev f ilters . For details . see the text by Johnson. et al.!" Altern atively, se veral iden tical low-pass -ecuons c an be cascade d to form a useful c ircuit. These fil ters are easy to analy ze and design, and off er e xc ellent performance, es pecially with simple direct co nversion receivers. An example of a fi lter of

+11

this type is shown in Fig 3.50, T hree tIVOpole sections with A=2 are ca scaded to form a 6-pole filter suitable for SSH reception. The res ponse fo r this filter is shown in Fig 3.51. The dip at low frequ ency resul ts from the l -I-l F input coupling capaci tor. Cascades of peaked low -p as s filters ( A > 2) ca n be very useful. The gain c an be co nsiderable when se vera l stages are ca scaded. These fi lters lake on a bandpass like shap e. offering an attractive res pon se for direct co nversion rec eivers intended for CW use . The fi lter shown in Fig 3.50 is biased fo r sing le powe r su pply ope ration . This sc he me is especially attract ive with the low-pass fi Iter, for an en tire cha in of fi Iter se ction s may be biased with on ly one d ivider. If LM-358 or LfI-·J-3 24 op-urnps are used, a pull down resistor sho uld be

+11

20 nF

10.00

Fi g 3.51-Response fo r t he c as c ade of identical lo w-pas s sections presented in Fig 3·50 . This is a calcu lated reeu tt , although we ha ve built several sim ilar designs.

R

Co

10 o f

rl

~1V

I

Co

,[>

10 o f

R,A '.

~

Fig 3 .52-Vo ltage contro lled vo ltage sou rce high-pass filter . The operational am p lif ier is again set fer a c lo s ed loop g ain of +1.

+12

+12 20 nF

1.00

Frequency (kHz)

20 nF

10 k 4 ,7 k

0",

10 k

Fig 3.50 -Practica l lo w-pass f ilter that c an be built w it h common op -empa, such as the 741 . 1458, 358, 324, 5532.

Filters and Impedance Matc hing Circu its

3.2 5

The

vev s low -pa ss titter

wit h equ al res istors has a transfer function of

R

c

C

,

,

,

Eq 3.20

s· C · R + s - · C · R - · A

---l

Input

where s is now the complex (Lal'Iacc I frequency , sejroin the Frequency domai n. C is the sh unt capacitor while Ax e is the feed bac k capacitor. T he co rres pon ding frequency dom ain respo nse is

p"

nne

· 12

r/

!" .R

2

c' Eq 3.2 1

~+

~ nne

I

~

1 f + R+ C+ . .>\ 2 + 16

it>

I

Fig 3.54-Biasing method for high-pass filter sections. A voltage di vider crea tes a synthetic ground at half of the sing le supply.

2.0 -

2.0

>

1_ _\

I

/ .r

>

, -,

---,

I J....,....... ....•.... .. .....

1.0

.

..... ..

•

0.0 0.1

....-

,/

1./ '

.>

.==.:;,:;; ..,-,·:1 - ·11 I

I

I

08 10

2.0

3.0

0.2

03

0'

0.50.6

i

o

\ \ 10.0

Frequency (kHz)

-

Fig 3.5S-The 4x4 fi lte r, a cascade of four peaked lo w-pass sections (6.8 kQ, 10 nF, and 50 nF) fo llowed by fo ur peaked high-pass sections (20 nF, 27 kO, and 5.6 kQ)

A // .......

/

0.0 0.1 V(62) -

/41

05

/

0;

' . . -..::

\

/

10

~

15

/

1;

I I I I I I, 4.0 5.06, 0

8,0 10.0

Freq uenc y (kHz.)

V(4 )

V(14 ) · · · · ····· ··

V(24 ) -

- -

V(34 ) -

·-

··

Fig 3.53-Transfer functions for four versi ons of t he high pass section of Fig 3.52 . The resistor ratio varies, taking on values of A=1, 2, 5, and 10. The solid line corresponds to A=1 while the hig hest peak is for A=10.

for fo ur d ifferent filt ers. a ll with l fl-nf capacitors and a 20-k!:.! ungrou nded resistor . Th e gro und ed re sis tor var ies to se t ga in and peak ing . The values used are 20 kQ . 10 kn, 4 kn. and 2 kn , T he characteristics of the high -pas s section arc much lik e those of the low pass. T he c ircuit hegins to take on a pea ked response when A exceeds 2. A peaked high pa ss will have a pea k freque nc y given by

2 'II 'C. R.~

Eq 3.22

T here is no peak if A_--- f. then the rela ted an gu lar frequ en cy is (llo=211:f. Then. the L and C for a perfec t match are

L "'~ we

C=_l _ (O)c· R

Eq3A5

T he diplexe r is applied where mixer!'> (e.g., d iode rings ) mu st be ter minated in a

in FIA 3. 81. Thi s ci rcuit. using not hing mo re than a bifilar wind ing o n a ferrit e tor oid. acc ep ts e nergy from a sing le generator with a 25·{} c harac te ristic impeda nce a nd su ppli es tha t en e rgy 10 l WO outputs. each with il 50-n impe dan ce. A 50-0 in put can be transformed down to 25 n with any o f the matching sc hemes p re se nted abo ve . v ariauon, o f th is ne twork us e tran sm issio n li nes or L-Nel works. The 100-0 res is tor abs o rbs e xce v, pow er that beco mes avai lable when one of the two o utput ports h mivs-rcrminared. A co mmon applicat io n splits the o utput uf a loc al osc illator chai n to drive two mi xers . Thc ci rcuit isola tes tbe two out puts . T his circ uit is cal led a 3·dB hybrid tra nsformer. for the power in eac h output. neg lecting lo vve s. is 3 dB bel ow the input. while Hy brid re fer s to tra nsfo nner- Hke circuits th at provide isol atio n bet....·ee n IWO of three po rt!'> . Hy brids .... e re used in early te !epho nes to isolate the mic rophon e from the earphone . Fig 3.82 shows a three port circ uit where

3 .3 6

Chapter 3

widcba nd 50 n 10 minimiz e disto rtion , T he diplc xcr shown is an es pecially simple one where each arm is a one po le low pass or high pass fil ter . Nic Hamilto n. G-l-TXG. h a.~ descri bed high o rder low pas s hig h pa ~ s dip le xers. ~9 A third-order exa mple of this desig n is show n in the d iplc xc r sideba r. Diple xers ca n a lso be bu ill with co mbinatio ns of band- pas" and ba nd stop net .... o r kv. a lso su mmarized in the sidebar . An int eresti ng. ye t si mple phase sbi ft netw ork is shown in Fl~ 3_83. A gen erator dri ves two o ne po le fi lters that arc te nninared at thei r output in ope n ci rcuits. T he two c apacitors. equal in value. arc picked to have a re ac tanc e il l one freq ue ncy equ a l to R. the resis tor value used in each arm . Th e phase differe nce for this network is 90 deg rees at all freq uencies. Ho we ve r. rhc two o utput a mplitudes arc eq ua l only at thc des ig n frequ e ncy. An especiall y interesting fo ur-port ci rcu it form is the direc tio nal cou ple r. T he coupler has an input and o utput. usually with lo w loss betwee n them. A third is called the "forward" coupled port . for the

e nergy availa ble i" proportio nal to the ene rgy flowi ng fro m the " in put" 10 the "out pOI.'" A fourth is the "re flected" cou pled port wit h en ergy pro po rtion al to th at flowing from the -'OUlPUI" to the "input,' "i~ 3.XS ..ho ws a schematic rep rcscnration of a d irectiona l coupler. wh ich is a lso a practical lo po logy in microstr ip form. Part B of Fig .U\5 shows a wid eb nnd variatio n using ferrite tra ns fur mcr s.v' A pract ical ve r..ion of the widc bun d co upler using three tran sform ers wa s designed by Ro y Lewalle n'! and is incl uded on the book C D. The d irec tion al cou pler is e xtremely usefu l fo r a va riety of app lications. When used with a PO\\e r me ie r or spec trum a nalyzer. re flected e nergy is a measu re of the impedance at the oUfPU! port. le adi ng 10 popu lar in-l ine pe .... er me ters such as the W7EL desig n. BUf the co uple r can also be use d to inj ect signals on a li ne. The coupling valu e is the powe r ratio be tween she o utput and the cou pled ports and is I / N ~ fo r the ferr ite version. MoS! direction a l co uplers have co up led en erg y that is in phase with the output. The microwave liter aru re abo unds with inte resting co upl ers. A coupler is a lso characte rized by d irectivity. Assume that the thru path i 20 0

390 PFJ

;[dPF

"

;,

33 pf

~Tu e I - 0 01

both be scaled for operation at much higher frequencies . Shown in F ig 4.12 is a VHF Colpitts oscill ator. Th is circuit was originally set up as a volt age co ntrolled local oscillator in a SSB transceiver at l-l-4 MHz. It can . however, be set up for a wide frequ ency range by spread ing or co mpressing the turns on the coil , which uses an air dielectric. Numerous other oscillato r form s are available for wide frequency range app lications. Thr ee arc shown in Fig 4.13. The first bipolar circuit (Fig 4.13A ) is a pri mitive variation of the scheme used in the Motorola MC- 164X. The version shown uses NPN transistors with a negative supply. The same cir cu it will wor k with a single pos itive power supply wit h PNP transistors such as the 2N3906. The oscillator is a one-port type where two no ninverting amplifiers, an em itte r follower and a com mon-base. are cascaded. The output is returned to the input with a shunttuned circu it attached at the common point. This scheme can be built on the bench and made to function over an extremely wide frequency rang e. Low Q tank circui ts are favored. This circu it suffers fro m very low stored tank energy. the result of voltage clipping by the transi stors. The second circuit uses J-FETs in a variation of the same topology. This circuit, similar to one used in the HP-8662 synthesized generator>, does not suffer from the voltage limiting found with the simple bipolar version. The circuit shown in Fig 4.138 is one that was breadboarded from available com-

2N 4 ~ 16

~ C&

" ] 5.8 pF

lI~ ~'''l,~'CCl ~

2N3904

l

2N 5~

" ,, ~ '

"

I ~

-

lO OK

0.1

lK to 3 . 3K

I

2 00

f

m c , 25

4 . 3 uH-=Q> 200

(A)

Reg .

~ ~~

J 310

2 00

18 00

lk

(B )

+1 2 100 J 3 10

5

0 0 "K~-===---F1 r

100 1K

to

3 . 3K

' £ 00

(--t11::) E-::L

j

lk

100 uH

Fe

Fig 4.14-The Vac ka r c irc u it sho w n is identica l t o the Seiler c ircui t pre sen ted ear li er except for the c ho ice of co m po ne nt va lue s .

Osc il lator s and Freq uency Syn t hesizers

4 .9

f8 Peg. 1 00

t ! 1~ -

h !p

+1 2

J 3 10

Vc

0.1

I

b

100

f'b

1

l RFc

0.1

J 3 1:d

>-1f-:l ~

"'C

Fi g 4.15- T h is fi gure sh ow s a va ria nt of t he Vac k ar os c illator w it h a Hartley theme. T he so ur ce a nd gate are both tapped down on the resonator as a means of isolat ing t he tank fro m t he resonator.

ponents. With an inductor consisting of 20 turns on a T50-2 toroi d. the circuit operated at 5.34 MHz with 20-V peak-to-peak on the tank. Changing the resonator allowed operation up to 200 :\1Hz. Figure 4. 13C shows a third vers ion of this oscillator that was built. this time usi ng 2N3904 bipolar tran sis tors. Aga in, the signal was 20- V pea k-to -peak across

the resonato r. Fig 4.14 shows the Vackar oscillator. Part A is a J FET ada ptation of a vacuum rube design appearing in the Sth edi tion of the RSGB Rad io Communications Han dbook with co mpo nents chose n for 7-:\1Hz operation." O utput is extrac ted with a high inpu t impeda nce buffe r attach ed to the oscillator dra in Pa rt B of the Fig ure is

esse ntiall y the sa me circu it with the gro und point shifted from the source to the d rai n. T he inductance value is slig ht ly lower in B than in A, for variable capacitor C v co nnects to gro und in H. If the capacitor had been return cd to the FET so urce in B. the L val ue wou ld be the sam e as at A for 7-I\IHz resona nce. The Vac kar ci rcu it in Fig 4 ,14B is identical to the Seiler circuit pre sented earlier exce pt for the choice of componen t values. Th e unique co mpone nt in the Vac kar is the lar ge cap acitor ac ross the FET gate-sour ce. T his component is crit ical: incre asing the value will drop the star ting gain to the point that osc illation will not comme nce. A decrease in i nductor Q will have a sim ilar effect. The deeou pling bet wee n re so nator and FET is near opt imum in the Vaekar. Passive component tem perature coefficien ts will still dom inate thermal sta bility. Ftg 4.15 sho ws a variant of the Vackar osci llator with a Ha rtl ey the me. T he source and gate are both tapped down on the resonator as a mea ns of isolating the tank from the reson ator. Th is circuit is a d irect tran sfo rm ation of th at of Fig 4.14H a nd is often used at VHF for low noise osc illators.e

4.4 NOISE IN OSCILLATORS Som e me ntion has alr eady bee n made regarding oscillator noise . We don' t traditionally think of noise when d iscu ssing oxcillators. However . noi se is presen t in any practical elec tronic circui t: the oscillator is ce rtainly no exception. Indeed. exce ss LO nois e is typ ically the dominant phenomenon limiting the performance of most transceivers in the late 1990s time frame . Befor e dis cussing osc illator no ise . we should co nsider some RF measurements . A spec trum analyze r (S A) is the instrumen t normally used 10 exam ine rad io Ireq uenc y signals. Th e SA is ess entially a calibrated. swep t receiver. usually withou t audio output. Sig nal strengths arc dis played on a C RT or similar screen. Wh en a si nuso idal ca rrier is a ppli ed to a spectrum a na lyze r. a response is noted at the f req ue ncy of th at carr ie r. C ha nging the an alyzer ba nd width will have l ittle impact as we ob serve the carrier, The amp litude is unchanged. It is spe cified as a power in dBm. (Sec Chapter 2 for a discu ss ion of dBm.) Noise is differen t. If stro ng. wideband noise is applied to a spectr um analyzer, it will cause the ba sel ine to ri se. If we increase the spectrum analy zer bandwidth

4 .10

C h a pter 4

by a factor of 10. the basel ine will further increase by 10 dR. We cann ot describe the noise with a simple "dBm level ." Rather, noise is specified as a powe r density. the power that wou ld appear in a I-Hz ban dwidth. If we apply a wide band noise sourc e to a spectrum ana lyzer set to a reso luti on bandwidth of 10 kHz and the re spon se co mes up to the - 60 dfi m line . we say that the spec tral de nsity of noise is - 100 dbm/Hz; the 10-kHz ba nd wid th is ·'40 dB wider" than a I-HI. wide filter. Reca ll that 10oLog( 1O,OOOj = 40 . If a carrie r was also present in the noisy display desc ribed . we might make reference to a carrier 10 /lOi ,II' rat io (CNR.) (We USI: the term "ratio." for we an: examining the ratio of powe r. However, we calcul ate this with a simple s uhtrac rio n, for t he pow er va lues are already in a dfsm format.] If the carrier was - ]5 d Bm wit h the noise at -60 d u rn with 11 1O-kHz bandwidth, wh ich corresponded to - 100 d Bml Hz, we wo uld say the CNR was 1\5 dBc/ HI . with dfsc standing for d B wit h respect to a carrier . (We usu ally talk of CNR. car rier to noise ratio. rather than NCR, nois e to ca rrie r ratio, for the carrier is much st ronger than the noise and is the lo uder.

There is of/e n a si gn dis crepan cy in these discussions. requiring care on the part of the readcr.) Recall the ear lier discu ssion of oscillator starting. (Fig 4 ,1) Widcband noise at the a mplifier input port was amplified . but was then filt ered in a re sonator. T he " signal" withi n the bandw idth of the reso nator is transferred with lillie atten uation and is aga in app lied to the a mplif ier input. Wit h a few "trips" around the loop, the signal has grown to the poi nt tha t limit ing hegins . As li miti ng occurs. t he ne t gai n around the loop diminishes. eventually sta bilizing at unity . the level nee ded to sust ain amplitude-stable oscillation, but no mo re. Unity gain occurs at the res ona tor center freque ncy (o r very clos e to it) where the net phase shift is zero degrees. Consider the ga in charac terist ics at freque ncies close to but slig htly rem ove d from the carrier. For exa mple, suppose we build an LC oscillator operating in the amateu r lO-m eter hand with a toaded tan k Q of 100. T hl:3-dH band wid th will then be 1'7: e-

/"

/ I

.

"

v.

.

•

I tlRini VO }UI~'

t\ J310 FET wa s used wit h source resistor biasing. T he varactor d iode was a surp lus BD 104, similar to the Motorol a MV !o.I ,

The Toshib a lSV I0 3.

USl:U

in some

imported equ ip ment. mig ht be a suitable su bst itute. T wo ind ivid ual varac to r Ji -

odes ca n also he used . This osc illato r ca n be set up for a wider freq ue ncy range by pic king CI. O ver I MHz of t un ing was a vai lable with C I= IOO pF. C I was dropped to 33 pF for a redu ced range. Th e

tuning characteristics for thi s oscillator arc sh own in Fig 4.35. The c irc uit is buill in a Hammond 1590 A e nclosure w i th coaxial o utput and feed throu gh capac itor s for power and tuni ng. A 65-pF pluxtic trimmer pro vid es co arse tuning . T he usc of bac k-to- hac k varacror d iod e v is common in VCO s. for it reduces the e ffect>. of rec tification of the oscillator signal. It i~ also com mon 10 see man y d iodes operated in parallel. This tnpology shows

lo wer noi se tha n a smaller number of higher-capacitance d iodes. The phase noise of this oscilla tor was measured using a 1-l.· \ 1Hz single conversio n superhet rece ive r with exte nsive crystal filtering. The veo was bailer)" powered with the batte ry also bi:ls ing the varacto r. wh ich was filte red furth er with a lflO-!1F capacitor. The sig na l was uttcnunted to - 3 I dHm and appl ied W the rece ive r inpu t throug h a ste p att cmnuor. Audio outp ut was mon itor ed with an H P3400 A tr ue -RjdS volt mete r wit h receiver AGC set off. T he aud io noise output in the mete r was noted 5 kHz away from the ca rrier. The receiver was then tuned to the ca rrier and the step atrenuator W :t S increased unutrhe res pon se was t he same as ob served with the noise. Add itio nal attenuation of t I (J d B W,IS requ ired to reach this response. T he no ise ba nd..... idth was SOO Hz. produ ci ng a measured CNR of - 137 d RcfHz. It is nOI clear if this noise co mes from the veo or from the receiver VFO. hut thi s value is a useful "WOh t case" l imit. No phase noise co uld be det ec ted at I 0 kHz offs et. !\\l o utboard crystal filte r was used for this meas urement . plac ing us al the h rnu of what we ca n measu re wi th thi v set up . Vo lta ge tuni ng wn h d iodes ten ds to co mpromis e noise a nd sta bili ty pe rfo rmancc . How e ver, re asonable resu h~ are available i f t he tun ing range is ke pt small. An attractive sche me uvev varactor tuni ng over a small ra nge wit h PI.\' diode switching in large r frequency steps . PIN d iode ca pac itor sw itching is illu st rurcd in tr unscciv er ( The Lichen ) o ffered in Ch apte r 6. The reader wo rking o n a synt hesizer fo r a high performan ce (w ide dyna mic runge) receiver should revie w the extens ive l ite rature en volt age c ontrolled osci llators. Numerous met hods ure ava ilable to design these ci rc uits. It is ofte n the varac tor diodes that ultima te ly limit no ise performa nce. :\'u ise su pp lied to the d iode o n tun ing lines ca n also co mpromise per formancc.t-

,I

4 .7 FRE QU ENCY SYN T H ESIS Virtua ll y all of the loc al osci llator sy...te rn... u... ed in mode rn co mmunicat io ns eq ui pme nt now usc freq ue nc y synthesis in on e fo rm or another. T wo cir cuit ty pes dominate synthesis: the phase- locked loop (PLL) and di rect-d igital synthesis ( DD S ~ . T he two sche me s are ofte n used toge the r. T he Hut fn Pu ff sc heme described e arlie r i... a freq ue ncy lock method and is not usually the bars;is for synthesis. T he re aso n iv tha t frequ ency lock allows frequ e ncy errors. whic h are absen t in PLL o r DDS vynthesizcrs.

4 . 18

Chapte r 4

.-\ PLL for freque nc y s y n t hes i~ in itv si mplest form is sho wn in Fi g -1.36 . T he first c o mpo nent i.. .1 vo ftagc -c o ntrulled osc illator c haracte ri zed by a lu ning senjiitiviry in H /'/ V . T his se nsiti vity usua ll y varie s over the tu nin g range . T he ne xt co mpon e nt is th e pha se . o r phase d ille rcnc c d etector. a ci rcuit that provides a de output pro porti o na l 10 th e phase d ifference between two RF inputs. The third element is a "loop fi ller : ' In its stmplcs r form. this, is (for a second orde r loop, a ..ing!e pole RC filt er with a cou ple o t resistors and o ne

capacito r. More ofte n its an ope rational a mplifier offe ring low freq ue ncy' gain as we ll as filteri ng pro pertie s. Th e lo w pa,s f ilter ing is needed to remo ve signa l co mponents r nming fro m the phase detector. The de from the detector und loop filt er must bc of the pro pe r mag nit ude 10 drive th.. VCO tu ning li ne . Beca use th is is a ne gative feed bac k sys tem (a type of uno loop. ) the phase of the fe ed bac k signa l av it mo ves throug h the loop to eve ntually reac h th e VCO musl be tail or ed for loo p revpo nse.

A Pl .L that is "locked" forces the vc.O to be at ex actly the same freq ue ncy as the reference . If th e reference is tu ned . th e \ T O wi ll follo w. mainta ini ng not o nly the -ame Ireq uc nc v bUI a ph ase relatinn vhip th'll de pends on the ch aracteristi cs of the detector. If the loop dynami cs arc "wrong." the VCO may not respond sm oothly to a change in the reference freq ue ncy. In th e e vrrcme. the loo p ca n os cillate. We begin o ur discu ssio n ofth e PLL wi th an experiment to evaluate a Mini-Circuits SBL- I mixer opera ting a s a phase dete cto r. .\IOSl o f us ha ve no eu-y way to acc urate ly measure ph ase. but we c all do th ing s 10 inter it. In this ve in . \\'1; firs t ch arac terize a piece of coaxial cable . 11 25-foo t len gth av ailable in a ho rne lab . A "half wave" balun wa s fabri cated from the cab le. shown m Fig 4.37A. T he two halanc ed out put po ints were attached to IOO-Q resist or s with th e ju ncti o n att ached to an RF spectru m a na lyzer. The si gna l ge nerurcr was tuned unti l a nu ll was ro und at 12.SS .\1Hz. This occurs when the cable i, a ha lf wav elength long , producing 180 deg rees o f phase shift betw een the tw o ends . A half .... avele nprh in Iree space at thi s frequency I ~ 3,s,2 feet. so the veloci ty facto r of our coa x is 0.65. which is abo ut what W I; wou ld expect . The pha se del ay in the coax ial cable .... ill be di rectl y pr oportional 10 ca bl e leng th and to freq uency. We kne w the length and frequency that yi eld a phase sh ift of 1SO degrees. , 0 we ca n ca lcu late the phas e fo r .Iny arbitrary frequency. Th e cha rac te rize d coa xia l cable is now u- ed in the tes t set o f fi g 4 .37B . T he sign al gene rator o ut pu t is d ivided in a po wer -plitter con si sti ng of th ree 5 1-n res istors . This pre ser ve s a 50-0 environmen t while eq ually s plitt ing the input po we r , On e signal is app lied direc tly to the SBL - I LO po rt. The oth er is attenuat ed by 10 dR . pha se sh ift ed with th e cabl e. a nd ap plie d to the mixer RF porl. The o utp ut wa s lo w pass filtered wi th a sim pl e RC fi lter and measur ed w ith a d ig ital volt meter. T he si gnal ge ne ra tor a mpli tu de was adjus te d to prod uc e the specified + 7 dR m La drive le ve l. Th IS ov erall circ uit is fami lia r a s a delay-fin e dis cr im inator. A qu ick tuning of the sig nal gene ra tor vhnwed tha i the out p ul was ze ro at fiA .\l Hz where COil X ph ase shi rt is 90 de grees . Data w as ta ke n ov e r the :; Lo In - MHz. spectru m to generate a p lot (F ig 4 .381 showi ng o utput vo ltag e as a f unct io n of phas e. T h is is cl ose to a straight lin e over a wide phase range . wi th the d ep art ur e at lo w ang les re sulting from a vignal ge nerator output decre a se neal' 3 Ml-lz. (W e used a mode st d riv- e at the mix er RF por t: t he mixer is ap pro ximate ly line ar to RF driv e at thi . . le vel.] Ex am in at ion of th e data in Fig

~R"qm,e vc o

'------
The filter can he as simple as a -lth order Huue rworth design . However. we have bee n di sappointed with these si mple fille rs . Fil ters with 6 to 8 crystals

Transmitt ers and Receivers

6. 57

J o

o

E
.

Cohn C r ,," h

cop,,.-,gm

l F i I t ..

r ~,

n".. . H= ~ ,

H t1 ML

B=il'500

Fi g 6.105 - Two a-ele me nt c rys ta l f ilters are co mpare d. The s hape m ark ed w ith sm all squ ar es rep re s ents the Co h n f ilte r w h ile t he ot her w as designed for a 0.3 dB Cheby s hev r esp on s e. Th e two f ilters hav e s im ilar skirt res po nse, wh ich is much better t han a Bu tter wor t h s hape, bu t much wo rse t ha n a h ig her-order f ilter.

Q. often a difficult desi gn task . A better so lut io n us es an osc illator that is no t keyed . The rec ei ve r BFO usua lly fou nd in a tra n scei ver is such an osc ill ator. but it is off set, op erating at the wro ng fre q ue nc y. Th is s lig ht c hange can be compensated with a suit able offs et in the YFO . Th is is

isola te it fro m the rece iver. Oscill ator oper atio n at a harmonic is often a conv enient option. T he signa l is then divided wit h a digit al di vider during key down periods. One of o ur des igns used a 5-t-.IHz IF. but slig ht chirp wa s encountered wh en a 'i-MHz crystal oscill ator wa s keyed. T he solut ion to the pro hlem is show n in Fig 6.W6. Eve n though the f ree ru nning osci lla tor in this sch em e do e s not o per ate w ith in the rece iver IF . sh iel di ng is still req uired. A steady tone wa, heard when the 10-\fHz o sc ill ato r wa s physically ncar the 5-M Hz IF. a resul t of BFO sec o nd har mon ic energy mixing with th e higher freq uen cy si gnal. Shie lding and use of feedth roug h capacitors for power a nd c o ntro l eli mi nated the p ro blem . Th e non -integer freque ncy multip lic ation schem e de scri bed in Ch apter 4 wou ld also be we ll suited 10 generation of a CV./ ca rr ier. That schem e d iv id es a free run ning oscillnror by 2. then uses one of the robust odd har moni c s pre sent in th e square wave . In the pr ior ex amp le wi th a S MH z IF. a crystal osc illator at 3.3:rB \f Hz cou ld be used. It would b e div ided by 2 to pro d uce a 1.667 \-t H l square wav e that ha s a strong har moni c at S Ml-lz. Th is coul d be filtered in a S .\1 H l crys tal or L C filter.

often a co nve nie nt solu tion. fo r RIT cir cui try is already pre sent in the tran sceiver. Another alternat ive is a non-keyed cry stal o scill ator other than the BFO. BUI one can't normally use one within the re ceiver IF bandw id th . for it wo uld be he ard unle ss mo num ent al efforts we re taken to shiel d and

The - lU-d13 m si gnal dev elo ped by the tran smitter IF (F ig 6.1(3) is re ad y to driv e a transmit mi xer. Alt ern atively. it can be applied to an IF speech proce ss or, shown in Fig 6.H17 . The voltage rel ated to a - I O-dB m sig nal in a :'iO-D cahle is on ly 0 . 1 V peak. This is nOI eno ugh to tur n on a d iode. Howe ver , it can be increas ed with a transformer unt il diode cli ppin g occurs . Ar ter the sig na l ha s been di pped . it is amp lified an d filte red The filte ring From t he secon d crystal filler is necessary: w ith o ut the filt eri ng . intermodul at ion disto rtio n pr odu cts ge ne rated by the clipping ci rcu itr y wo uld appear outside th e IF ba nd widt h . Cl ipp ing cann ot be do ne prio r to in itia l filtering. for that cl ip ping of th e d ouble sideban d signal w ou ld cr e ate som e disto rtion pro d uc ts wi thin the eventual IF pas sband that wou ld not otherw ise occu r. Th e l F spee ch pr ocessor has the effe c t or inc reasing the avera ge p owe r within the speech sideb and without increasing the peak. Th is higher average pow er Increase s intellig ibility wit hout exce ss d istort ion out of the normal passband. This pro cess or. with the leve ls sho wn. increase s the average to peak power by about 10 or 12 d B, readily observ ed wit h an oscillos cope T he IF pro cess or has a second adv antag e : It confines the IF leve l to prev e nt

Trans mit ters and Receivers

6.59

ove rdrivm g the tra nsmit mixer . With out the procevxi ng. it would he desira ble to add A Le. or "Automatic Le vel Control." Thiv is an AGe loo p in the transmitter th ai main ta ins the lev el thro ugh the overall power chain. Inter mod ula rion divrortio n i~ rarely a

fac tor in a rransminc r IF syste m. With M l little gain required. the IF syste m can he vimp fe. B UI the huilder/des igne r should

a mixer dri ven hy a di sto rted IF signal.

Bidirectional Amplifiers

be careful to be sure that distort ion h not an iss ue. It wou ld be folly to design an ex treme ly low d is tort io n RF power cha in only 10 feed it with the output of

One view of a SSB transm itter says that it is nothing more than a superh eterodyne SSB rece iver with signals moving: in the

aa 7Bl05

+ 12

22J

0

l . .'lK

ex

l - ,"~.l

10

, , I " 22K , 74HC74 ,

4 .22

r-:

10lDfz

" •

iji

, 'K

2 .1u

aa

lK

...

Output.

2. 1u

Ql

19 K

C -S~ .l

1

"

I

12

I ""

,

1

-L

niL

82~

.

2 H]9 0,l

"

2 H3 9 0 ..l

U

41 0

1

J-J-'VV1r- + 12T

21l1 9 D4

Fi g 6 .106 - A lt ernatlve ca rrie r-osc illato r system fo r CW generation . A fr ee-running 10-MHz crys tal o s cill at or is divided w it h a di g ital divide r to gen er ate 5 MHz w hen needed. T he di vide -by- 2 c ircui t is controll ed w it h an Ie reset line. See text.

Spee ch Processor

From TX IF Arr.p

+ 12

9

4.7 I
~Mr

Fre q

-L

-L

e-eoc

C-mid

!.!fu.

MH z

pF

pF

j :-

0 22

2200

3300

375

07 04 55 0. 65 II

4 70

1000

C-twl e pF 307 14 3

1.\ 2

2\ 2 28 4

e

~ .1 3 5-T r i p l e · l u n e d

c,ni

-L

B .T,\T

., 15

C - t lUl e

L

(2,

820

1750

78

50 0

1200

34

39 0

820

10

130

39 0

II

assembled and tested in it 50-Q en viron ment prior to use in the trans mitter. A table of computer ge nerated va lues is gi ven in Fig 6.135 for several additional bands .

.i,

L

Q,

The Local Oscillator IL dB

"H 27

50

36

15

50

11

7 4

200

17

200

15

3 3

20 0 20 0

14

32

bandpass filte rs for several HF bands. The required

ed Q (vi tal) is also give n.

The IJ ) tu nes from 13.5 to 14 M l-lz wit h the he terodyne system of Fig (i.Bo, Q402 is u 2. 5 to 3-MHl Colpitts oscillator buffered with a common -base amp lifier. 0405 . Outp ut is kept lo w, for only - 10 dHm is needed by d io de ring mixe r U402. The ou tpu t is cstabfished wit h the pad driving the RF po rt. Th is lc vel, and that at the mixer LO port shou ld be meas ured during con struction . A 365-p F variable capacitor tUTII:S only ha lf of the range. The other half is tuned by switehing in a n additional cap acitor, C402 . The switching is performed with a pair of PIX diodes, 040) and 0 402 , Whe n a pos itive volt age is appli ed to H OI. 040 1 is saturated, ----i~ 1 • 101

1
"

c~

·1 ~

2N3 9 0-Q resistors in a " y " The 6-dB hybri d ~ reco mmended. Assume that the tw o generato rs have crystals to put their freq uencies at 14.03 .od 1..1.05 MH7.. T uning to either of these N!!na ls pro d uces a large met er respon se. These signals impinging on the receiver front end will inte rm odulate, gene rating distortio n produc ts above and be lo w the ",,0 desired sig nals, at 14.01 or 14.07 1fHl. These products are created wit hin the receiver, usually in the c ircuitry ahead of the main TF filter. With the two test sig nals separated by 20 kH z. the distortion sig nal will be 20 kHz above the upper desired stg141 and 20 kHz below the lower one ,

l ow p as s !i lt ~r

Rec""'", unde' tost ("9' off)

-12 ' po we r j ra rely a valid inter preta tion. furt he r j ustify ing the reactance measuring op tio ns . The bridge of Fig 7.3Kwas calibrated a t 14 .\1Hz with a handfu l of carbon resistors wit h the values then marked on the panel. While this i.. handy. it may not be necessary. Consider the variation shown in FiA7.-11). This is equiv alent to the other bridge at RF where the capacitors are virtua l ,1'1011 circu its. How. eve r, the design with capacitors can be measurcd with a d igital voltmeter attached 10 the "unknown" port . The de meas urement tells the user the status of the pot. allowi ng the RF resistance 10 be inferred.

Input

RF I nput

'" ~-

'I ~-

Li n

'"

"

H

n

" ' " U........

"".~

~~,~

' Mp , >t p .>tp

-

71 :

m

n :

l bl f l 1ar tur.. FB-1 J- I ll

. 10 o a

"non .

!t outpat

11n.

Fig 7.38--RF b ri d ge lor HF measurements. Rt is id eally a 100-0 li nea r po t, b ut all w e had was 200 O. T he Claros t at 112-inch dia meter conduct ive p lastic p art s should offer r easonable performa nc e. alt h ough we have not used t he m in this applic ation ,

7 . 22

Cha pt er 7

Fig 7.40-0ptlo nal variati on o f the re sistan ce b ri dg e.

Interi o r v iew of return loss b r idge. This one i s buill with 49.9 n , 0.1 W, 1% res istors.

RF im pedan ce bridge wit h bu ilt in mete r. A refe re nce must be att ac hed for measu rements .

Table 7.1

tor of the RF im ped anc e b r id g e. m et ry and s ho rt lead len gt hs are tairt ed du r ing co nstruction. Long .-cis are okay w it h the dc parts of the ~u i t .

Return loss bridge for HF range .

Ther e is virtue in the modified bridg e: -e a calib rated d ial is not needed, it c an built with smalltrimmer pots with much ct RF charact eristics than encoun te red ... ilh pet s wit h sha fts. Th is wi II a llow these .-adn io nal met hods to he ex tended to ;her freque nc y. (Th ese ex per im ents rcn o n our "to do" li st at this writing. ) t'iJ;: 7.41 shows a retu rn loss brid ge L B. ) a c irc uit wit h no adj usta ble ments. T he signal corning fro m thc ector port ind icatcs the qu ality of the .-pedan"' 1.•••"

•• .

' '''..1..'

Fig 7.4S- Hig her po wer versi on of a Iransm atc h wit h a r esisti ve brid ge. This u n it is raled for 40 W o r sli g htly mo r e fo r sh ort period s. T he to pology sho w n presents a 50-a load to Ihe t ra nsmitte r wh il e attenuati ng t he sig na l pu t o n th e ai r by 12 d B. If t he res iste rs spec ifi ed l or R c ann ot be pur ch as ed, paralle l co m binati o ns of 2·W res is tor s c an be us ed.

ing several. This bridge used 51-D. '/w·W resistors and a transformer consisting of JO M ilar turns {If #2 l! on a FB73-240 1. The high permea bility core ts preferred, providing an inductance of 175 IlH for each winding. A different transf ormer imp ro ved V Hr pe rformance at the CO, ! of HF d irecti vi ty. We saw 30-dB d irec tiv ity at 144 I\f Hz when the transfo rme r used 5 o f thc 6 hole, in a m ulti-hole bead. a FB4 3 ~5 1 1 1 . Th is confi guration p rodu ce s an ind uc ta nce of SA IlH per wi nding. The hybrid q ua litie s of the return los,

.r

•~ J."

UI

h

~U I

, 'c.

l-' .i-d~d 'r .I c

K._

~

••

'" . n

0. . . .. .

Fig 7.44-7 MHz lran srnalch and r es isti ve bridge 10r portable operation . Variab le ca pac itor s are s cr ewdriver ad justed, mtca compression types . All res istors are 1f2 W. S1 Is a DPDl slide o r togg le swrt c h. Th is de si g n is su itab le 10r tr an sm itt er s up to abo ut 3 W i1 the t u nin g Is done quickly .

.- I n ..,.

-=-

...-._.

.,

Cl. "

7.24

~ nun

. 111

Dl: Ho t Carrie r diode s tu::ll. as 1N5111

r.

+

i 2 .4K

15

111K

=

L_Ut U I - '

51

~f

15

=

..

L a nd C s et f or 7 MHz •

..

,r-~:.uo.

. Il.

m

~

Fig 7.46-A ud io mete r re placement scheme l o r Iransmatch tuning. See text.

bridge arc illustrated in Fig 7.-12. Generato r V1causes voltage s x and y to he equal and in phase if the hridge is bala nced. Hence. none of this power ends up in R 2. the impedance of the other source. But V 2 al so sees a balanced bridge . The power deli vered by V! force, the node with R I to he at signal ground. so none o f the V2po wer e nds up in R I. Th ese characteristics provide the isola tion that we need when combinmg signals from two generators for l~ m testing. A co nventio nal resis ta nce bridge c irc uit with buill-in det ec to r is show n in Fig 7.-13.

Th is circ uit functio ns into the L"HF ar ea. realize d by sma ll lead le ngth and careful symmetry. A photo sho ws the im ide ot the c irc uit. Th is bridge wor ks well at 144 and 4]2 MHz. as well as the HF spectrum . A simple reslstancc bridg e with incl uded detect or is ofte n used for the adjustment o f lo w powe r ante nna tu ners. This is o ften preferre d ove r an in-line direction al powe r me ter, for the trans miue r is alway, property te rminated during luning. A circuit used with po rta ble rran sce lvcr s is sho wn in Fi lf 7.44 where the co mpo-

t\ are app rop riate for the -m.m ete r band. elide or toggle sw itc h is put into the -.ne.. positi on 10 adj ust the circuit for he~ t . II i.. . then retu rned to the "operate" po• A higher po wer ho me . . tario n vercion ~\\ n in Fig 7.43. The low power varia nt ~ d germanium diod e while a . . iticon It.:hing d iode is used at higher power . Some builders han: used a lig ht emitting :-Je to replace the meter ind icating brid ge

balance. Perfor ma nce is poor. es pec ially for low power transmi tte rs. tor visual OUlput is zero unt il abo ut 1.6 V biase s the LE D . But me ters are often hea vy, difficul t to find . and expe nsive. Some refined circu its usc ferrite transforme rs fo r gre ate r se nvitivity . An alternati ve sc he me i.. "how n i n Fig 7A 6 \\ hen: an audio os cillator replaces the visu a l o utpu t. The oscill ator . a vim pl e muln- vibrator using Q2 and Q 3. is fre -

quency modulated by the brid ge sig n'll ...it h the pitch becomi ng higher with greater mismatch. Th e circuit is u..ed by sendi ng a vrring of dits into the tran . . matc h. T he pitc h become.. ide ntica l for key up a nd l c y' do wn whe n the ma tch i-, perfect . The pri mary purpose uf Q I. the JFET input. h to generate a d e o ff set from ground. ' I I JFET type is ext rem ely non-cruical . A n op-amp would also serve this function.

o f inter est a nd the tuning control is J Itac bed 10 01 moto r throug h a suitable pulley, T he mo tor also dr ives a poten tio meter that de velops 1I vol tag e prcponionnl ro the Irey' uency. Th e vol tage from the pot ind icating frequency is ro uted to the hori zont a l axis of a n oscillo sc ope w hile the s igna l fro m the receive r"s AGe. in di cari n ~ signal amplitude. is applied turhe 'scope vertical . The resu lt is o ur spectrum analy zer. Th e ucual spectru m analyzer is c a librated in freq uency. so we know the freq uency representi ng the sc reen ce nter. we also know the freque nc y \ pan. the nu mbe r of kHz or Ml-l z ussociutcd with the d ot as

it swee ps fro m left In right. T he o n-sc ree n vem cul pos ition is also calibra ted in the la borato ry spectru m ana lyzer. W hile we obtai ned a voltage fro m the receiver 10 app ly to the -scope \ en ica l axis . we calibrate with regard to the power re lated to the signal that de ve loped that voltage . The top (If the scree n i~ called the reference teve t. leaving the bottom with no spec ia l sig nific a nce , When we sec a sig nal on screen that ju,",' rea che s the rcteren ce le vel. ...e know tha t it has a strength eq ua l to that level . Th e usual spec trum analyzer displays ~ i g n a 1s logarithmicall y, sothe calibrat ion wil l bein te rms of an um -

7.8 SPECTRUM ANALYSIS Wh a t is a Spectrum Ana lyze r?

,

One o f the most useful inst ruments ' he W IO ex pe ri menter could have is the specm ana lyzer. Co mmer cia l versio n.. . are hi.. . ucated and expensive. bu t exce llent examples arc beg inning to ap pear on the owpl us market . And there are no w many .. ailable co mpo nents th at allow the en ter.." ing e xpe rime nter to build hi.. . or her n spectru m a naly zer. The firs t qu e stio n we mu st address is rno.. . t fundamen tal: What is a spectru m J.1 ) ze r·! In the ge ne ra l ma thematic a l lC11 ~e . the signals we e nco unter an: ge ner Iy co llec t io ns of si ne wa ves of the fo rm: " . . . in(2 • It • f · t) . .....'here A is an am plit ude , r i ~ frequency • Hz lind t is time in second s. We can regard I" function as eithe r one o f time . 1. or of frequency. L l n the most gene ral sen-e. any function of lime has a related spectrum or freque ncy do ma in represe ntat ion. The I WO Jo mllins or viewpoints life related through a ntthe matic al ope ration called the Fou rie r Transform. lZo 13 Also see Chapter 10 o f thiv volume.

Sell ing forma li tie s aside. we look at ckctro nic si gnals in the ti me domain with III o sci lloscope or e xamine them again st freq uency with a rad io frequency spe ctru m "'J.I F ~ r. we arc already familiar ...ith raJio freq ue ncy spectra o f se veral so rts. aliIIough they may not have been presented .b such. A rudime ntary spec trum analyz er. .a.I ~ i t un-calib rared . i~ shown in F ig 7A7 . We have extracted o ur communic ations receiver fro m no rma l se rvice and o pened it eoanac h wires to the Svmete r. a panel me ter indicatin g the slrl:ngl h of rec eived sig nals, This voltage is us ually der ived from the receiver AGe. Th e receiv er is set to a hand

Oscil l osc op e

in x-v ncee

~

COlmlunl ca t l ons Receive r

CJ

/ 0

Q 1-

~.

......,

~ .~

, ./

3~d!

~",

000 0 0 0 a I

V

I

»r

\

,

~

\

- ~ Wire Ceo. c ec: e i vec

~s:-_t

.. r "

Fig 7.47- A rudimen ta ry s pect ru m a na lyze r forme d by app lyin g mo to r dri ve 10 rece ive r t u nin g and to a pot t ha i generates a vo ltage that Indi c ate s Ihe tuned frequenc y. This vo lta ge co nt ro ls the X axis of a n osci lloscope. The vertical Y ax is is derived fro m th e receiver s -me te r circuitr y, (Tha nks 10 Bob Bales fro m Tektronix, Beaverton, OR who s ugge s te d this ex pla natI o n.)

Measurement Equipment

7 . 25

-

.....-

... .

F i ltrr

Power

Meter

1'11 t r r

I S10pLal ~,",rdtor

Fig 7.4 8--Mea su r eme nl receiver allowing rudi mentar y s pec tru m analysis. Although th is inst ru me nt Is presente .... H IS ~lIl l ocl;>

~;JI

15

'i

! OOF:!!O.l

2 F.

5000 1"':

2F.

:'0 · 12

dcp: .

•

Fig 7.60- Meas ur ement rec eiver fo r meas u re me nt of sse t ra nsmItters . T h is unit used an av ail ab le 10·mA meter mo ve me nt with a high resolution scale, but CR n be adapt ed to avai lable meters. Thi s inst rument can be adapted as a narr ow tun ing range spect rum analyzer, a refinement t hat we have yet to complete.

Measurement Equipment

7 .33

a measurement too l F ig 7.6.' show s an example where a sign al gene rator was being inves ligated fo r phase noise . The noise sho wn in the fig ure is indeed noise. fo r a cle aner oscilla tor ope rat i ng with the same ana lyzer pa ra mete rs pro d uce d a sim ilar spectru m. but witho ut the noi...e. The resol utio n ba nd widt h fo r this exa mple is 2.6 Hz ! The hard.... are and software u...ed for this e xamp le are discussed in much more dera il in Chapters IU and II . Although FFT methods often co ncern audio or "b aseband," the co ncepts are ca pab le of muc h more. So long as a sig na l ca n be sam pled in lime a nd convened 10 digital dat a. it can he transformed to the freq uency domai n, xtany modern oscill o'Co pes are built with rela tively low speed displays. But the i nco ming ana log signa l is anything but slow . The incomi ng data is a mplified and /or atten uated and preve nted to a high speed "scan converter:' essentially an A to D convener. Once the high

speed ,>ignal is re me mbered . it can be read at a lower speed and disp layed as a time sig nal. Th e dat a can also be prese nted to a FFT "en gin e:' or co mputer to generate a cor res pondin g spe ctr a. While usua lly lacking the dyn amic range of a n analog spe ctru m ana lyzer. a spectra with a dy namic range of 50 dB or better is com mo n .... ith such osci llos cop es. A block co nve rter can he used to move part of a n Rf spe ctr um dow n to audio whe re il can be exa mined with a J-'FT type spectrum ana lyzer with an example shown in f ig 7.6-& . An e xte rnal step ane nuator a nd (o ptio nal ) ba nd pass filter precede the co nvene r. A diode ring mixer then move s the signal down . The res t of the circu itry is very much like tha t found in di rect co nve rs ion recei ver s. This converter can be used ahead of the FFf analyz er implemented with the DS P hardware from Cha pte rs 10 and I I . We have als o used it .... ith a person al computer sou nd card and modest cos t so ftware. 20 One must be careful with any ofth ese schemes 10 avoid u ve nl rivin g t he Acte -D con verter; ove rdrive can turn the ent ire sc ree n to unrecognized gibberish ! Sound card solu tions seem les s robu st than the dev oted DSP tools.

Measur ement Equipment

7 .35

A block conver ter an d a bas eband FFT analy zer an: ideal fo r evaluation or SSH tran smi tter 1.\1 0 . w hat had alw ays bee n a d ifficu lt lab oratory meas ure ment is now avai lable to al most all e xper imenters. A tradi tional two- to ne au d io generator was

incl uded earli er in this ch ap ter, The narrow res o lut io n a vail ab le from an f-f-T bas ed analy ze r will also a llow the experi menter to measure in-hand tra nsm it ter d istor tion . A tone sp acing of ar ound 100 Hz the n beco mes appropriate. In -ban d

performance becomes im portant when a S SB tran scei ver is used to proce ss narro w bandwidth in forma tio n suc h as enco untere d in PSK31. Agai n, the ava ilabi lity of mea s urement roots p ro v id es the e xperi menter with great op portunity .

7.9 Q M EAS UREM ENT OF LC RESONATORS Several sc hem e s have heen used for Q measuremen ts overthe years. T hey can all wor k well when caref ully execu ted . T wo schemes ar e prese- nted here for LC tu ned c ircuits . T he rirSI met hod mea sures the bandwidth o r a tuned c ircui t co nfigured as a sy m me tri ca ll y lo aded bandp as s filter with very high insert ion loss. T he sche matic is shown in Fig 7.ciS. The t \NO coupling capacitors should be a ppro xi mat el y equal. This prevents heavy lo adin g b y the inp ut with weak output coupling which cou ld create high in scr lion loss with a wider than minimum hand width. Equal values guar an tee that the inpu t and output each contribute equally to the load ing. High insertion loss then

' 0

Utllll

Fig 7.6S-Me a s urin g a by deter mination of 3·dB ba ndw idt h. The coup ling capac ito rs , Cin and Cou t, should be ap p rox imately equal and shou ld be s ma ll e no ugh that the ins e rtio n lo s s is 30 dB o r more.

ens ures that the externalloading i s light so that ba ndw id th is dete r mi ned on ly hy resonator lo ss , The measurement is do ne with a signal gen era to r and se nsi tive detector suc h as a spe ctrum ana ly zer, a 50-n termina ted os cilloscope. or one of the power met ers descr ibed earlier. Th e gener a tor is tuned for a peak res ponse an d the ce nte r fr equency, f o, is read with a counter at tached to the generator. The oUlp ut am pli tud e response is also noted. The signal generator drive is then increased by 3 d H, ca using the output to in crease by the same amo unt. T he generator is then tuned f irst above. and then bclow the pea k until the response is identic al to the orig ina l ampli tude. T he f requ e nc ies o f the uppe r and lo wer - 3 dB points are no ted an d the dif feren ce is c alc ulated as the HW. T hen Qu = folBW where both arc meas ured in the sam e frequency units. If the in sertion lo ss is 30 d B o r mo re. the measured Q is very close to the unloaded va lue. Sec section 3.3 Th e mea surement can be don e with lo wer TL b ut correct io ns will then be rcquired to c alcu late Q u from the mea sure ment Q. Another sc heme fo r Q measuremen t use s resonato r cle me nts in a trap c ircui t, shown i n Fig 7.Mi. Again. a tunable ge nerato r and a 50-n dete cto r arc used. However. instead

of co nfiguring the reson ator as a los sy filte r. we now configure it as a trap. a circuit tha t produces hig h atten ua tion at one frc quency . T he gener ator is tuned to find the null in the output re sponse. Th e null depth. which can be very large, becomes a measure of the reso nator Q . Ei ther a paral lel co nnec ted ser ie s-tun ed circn it or a ser ies connected para llel-tuned ci rcui t can be used as trap s. There is usually little virt ue of one type over the other. we gener ally prefer the seri es-tuned c ircui t bec au se a gr ou nded and calibrated vari able ca pacitor can he used in the res onator. A photo shows a test fixture with a 14 0-pF var iable ca pac itor an d bindi ng posts. T he generator is tu ned to fi nd the null respo nse and the level i s care fully noted, A spec trum ana lyzer is ideall y used as the de tector and should be in a 1 or 2 dB per

1 0 . 99 6

Series 'rc

"-

a ~16 . 9

" Parallel TC

Fig 7.66-Measu ring a by dete rmini ng the atten uation of a trap . A 7-MHz t uned circu it is used in this exa mp le wit h L=1 IlH. The 0.176-12 resis to r in the series-tu ned circu it a nd the a lmost 11· k12 res istor in the pa rallel t uned circu it a re models rep resenting a 7-MHz a of 250. The series-t uned circuit (STC) will ha ve an atte nuation of 43.1 dB wh ile t he PTe ha s 40.9 dB .

7. 3 6

Chapte r 7

..:

~.,f-~ ;;"" '_?_~ .

:' v

,

A test fixture s imp lifies a measurement with t he pa ra llel connected series tu ned trap method. The inductor s how n was 13 t urns of #14 enamel-covered wire wo u nd on a 3.5-inch-d iameter PVC pipe fitt ing, This coil ha d a mea sure d a of 371 a t 7 MHz, The tes t fixt ure inc lude s a gro u nded post a llowing additional fixed capacitance to be added .

drvision sens itiv ity to pro vide amplit ude resolut ion. The res ona tor is the n dts con aecred and the ge nerator is connected to the detector thro ugh a step anen uator. T he ane nuatin n is adju sted until the anal yzer respo nse is exactly the same as produ ced .It the null. T he auenuator value is thc u the a ull atte nuatio n, A, in dB. Values of60d B Of more are posvihle with so me high Q tuned cir cuits. Th is sa me meas ureme nt setu p can be used to determine inductanc e if a cali brated capaci tor is use d. The unloa ded Q i5 related to att enu atio n hy

4 ·IT·f · L u

Z

ru

Z- 5 0

~1T:"

00

ss ~

,,, l

1

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eo

.1. (11.1 ss -

-

so ~

co

" '"s Eq 7.4

e

om

f. :\1Hz; A. dB ; L u ' uH; Z, Ohms if the series tuned circuit form is used,

,

0;

,

I

i I

"I R , "i"" ~~i,, ~

A(R .) _ _2° '101112'

I I

I

i

1II111

i

11

,I 0.' Serle,

, " !!O,,,.......

'"

'w

0"""

Fig 7.67-Atte n uati on vs R for t he seri es impedance. See text.

0'

Eq 7.5 f.

~1Hz;

A, dB; L u ' uH; Z. Ohms

... if the parallel tuned circu it is applied. Freq uency is mea sured in MHz, A is in dB, and induct ance is in IlH fur these equations. Z is the characteri stic impedance of the measurement e nvironm ent. usually 50 n . It is useful tu plo t series resis tan ce against atte nuat ion for the paralle l connected serie s impedance . This is shown in Fig 7.67. The ex per ime nter may wish to build a si milar curve for the scnc s connec ted parallel impedance.

It is impor ta nt that a solid 50-n load and sou rce impe dance (Z in the equ ations) be used in this measureme nt. Tfthe impedance is in que stio n, use a lO dB pad at both the generator and detector. It is also important to prevent harmonic s from confusing the resu lts. T his is guaranteed if you use a narrow band wid th detector suc h as a spectrum analyzer. A wid e hand det ec tor (a power me ter ur a 50 n ter mina ted osci lloscope) will rerespond to harm onic en ergy that is not attenuated by the tra p. The spe ctrum analyzer used for Q measurement could be ve ry sim ple , Something as simple as a sing le tuned circuit preceding an osci lloscope would work so long as a pad was used to es tab lish impeda nc e. Ahoma-

tiv ef y, a very well lo w pa ss filtered signa l genera tor co uld be used with any detector with adequate sen sitiv ity. The virtue of the trap scheme becomes apparent as soon as the two methods are compared. The traditional 3-dB bandwidth measurement de pends on precisely esta blishing the 3-dB down lever. A fract ion of one dB e rror co uld still impact accuracy, Tn contrast. the depth of a null is oft en qui te large for high Q resonators . and is eavily measured with a step ane nua ror. An accurate cap acitance measurement too l such as the AA DE or 'VI!7AAZ meters mentioned earlie r is qui te useful as a supplement to a Q measurement setup . With such a tool. accura te calibration of capacitors is ensured.

7.10 CRYSTAL M EA SU REM EN T S A quartz cry stal is modeled as a seri es RLC paralleled by a capac itance . .F ig 7.68 . Crystals are of spec ial inte rest. for they are ofte n used in construction of narrow fi lters. For this purpose. we need to know all of their parameters. G reat precision is Fig 7.6B-Mode l for a quartz crysta l. needed in knowi ng resonant freq uency, for tha t st ro ngly controls fi lter tuning. T he kno wle d ge of the o ther pa rameters is refin ed measure me nts are des ired for filneeded at an accu racy sim ilar to t hat en- ter de sign . An extremely useful, yet simple oscillator was also present ed in C hapte r 3 cou ntered in an LC filter. The re are num ero us mea sure me nt a nd is repea ted he re as Inal COlllpUlers With .4.pplicatioIlI to E/t'Tlrinl! NrI",orh. Jo hn Wiley' &. Sons .

19JH. Thl" hlM)"- cove.... the mathematical side of np rimi:mjon and is good for thuse wanti ng to spe nd svme ti me on thc suhject. Knowledge of Cale ulu, and Li near Algehra is req Uired to fully usc the mate rial. bu t BA SIC progrilm.' a nd cxa m p l e ~ are p ro\'iu~d for those who w ish to ilppwach thc , uhject exp erimentall y.

CHAPTER

DSP Applications in Communications In C hapter 10 a number of D5 P building bloc ks. such as os ci llators. fi lter" and modulato rs were explo red. In many cases the bloc ks we re a lternatives 10 tradit ional ana log func tinn v, whil e in other c ase". such a" the discrete Fo urier tran sform. we ure introducing func tionality that was nul previously prac t ical. In thi s chapter. we will explore me thod .. for co mbining seve ral bloc ks to prod uce a piece of co mmunications equipment . We will be intc -

grating three types of functions: • Tradi tional ana log co mponents. well as RF a mplifiers and Rf mixe rs. • DSP co mponents. such as were cove red in Cha pter 10.

• Comrob for both of these types of co mpone nts. Mo st often this is assoc iate d with o pera tor intera ction. invo lving both displays a nd interface contro ls.

The con trol o f the commu nic at ions eq uip ment can usua lly he improved by so me sort of co mputer. which is often a dedicated mic rop rocessor. T his may be a good a pproach. depe nd ing o n the co mplexity of th e devices. An alte rnative. howev er. is to use the same nsp device thai is pnx:e..,..,ing si g n al ~ 10 do the contro l func tions . This approach will be used severa l times in this c hapter. with the res ult of needi ng less total ha rdwa re a nd on ly a si ngle co mputer program. T he journey of an experime nter who decides to investigme thece OSP projects will begi n with the EZ-KIT Lite from Analog De vices. T he first thing .. tha t mig ht he done with this OSP board arc sim ple demo nstr auuns SIKh as aud io filters. which arc well described i n the manual s su pplied wit h the bo ard. Se vera l of the ve ca n he tied into

an existi ng rece iver and used d irec tly for on-the-a ir expe rimen ts. 'lhic c hapte r foc uses on the procevving or signals. but before gelli ng to that we nee d 10 100 1.. at so me basic control tec hniques. The flrst issue we wi ll add res s i ,~ that of computer ime rr uptc. which arc fun damen tal to having the DS P program s operate in "yn c.h romsm wi th the att a ched hardware. All the OSP progra ms neede d to bring lift: to these proj ects are included on the C D-ROM "" ith the nook and are no t repea ted in the tex t. Shown in this chapter a re a l ew tragrncms of the progra ms to illustrate a number of det a iled operations. It is rec ommended thatthe read er look at the comple te prog ram . o n oc ca ..ion. T hi-, gi ve, a "big pictu re" view of combining fragm ent , into a wor king DSP program

11.1 PROGRAM STRUCTURE All ..:omputer programs have some form of overall struc ture. rang ing from trivia l to excess ively co mple x. Ofte n ti me" the struc ture is largely de ter mined hy a gro up of programs, co llec tively referred to a ~ an operating system . For a PC . rhts co nstrains all prog rams to certain cunvemionv whi le a llo wing mult iple programs to share re so urce". such as mem ory o r processor time. To the perso n wri ting a program thiv ca n be both a co nvenience a. well as a sou rce of anxie ty. Having a set of sub rou ti ne" avail able to ha ndle standa rd ope rations com speed up prog ram writing, Howeve r, if t her e arc mu ltiple users of resou rces. there may be no g uarantee that a pa rtic ular program will finish it. 1:.1 ' 1,;

when need ed. "R eal-t ime" programming becomes proble matic under these circumstance". Fo r simpl e OS P program". it i" often poss ible to operate .....ith no rea l-time operatin g vystern. All resources arc allocated when the program is desi gned . The ove rhead of the operating vyvrem is avoi ded a nd the programs ar e guaranteed to co mplete their tasks on time. A ll the programs in th i .~ c hap ter will usc this app roac h and have sa me structure . This. consists of a bac kgro und prog ra m that processes all da ta that has no ti me deadlines. and a si ngle Interrupt S I' I"I'il"l' Routine (ISR) that incl udes all rout ines th:ll must he com ple red on a peri od ic bast s.

Interrupts .As discuss ed in Chapte r 10. data proccssing de vices requ ire so me method to change the pro gram operatio n. based o n some elect rical inp ut. Call ed interrupts. the ..e method s involve some internal dedicare d hard ware to make changes to the processor state . Normally the minimu m operat ion i-, a chan ge in the add ress at the program being executed. T he progra mmer must have placed app rop riate instrucrionv at the interrupt-alte red add ress. A complication for interrupt pro g rammi ng is the potent ia l for mult iple interrup ts. Fo r e xample , in a OSP program, these might be a n operation to output data to a D/A co nverter and a need to out put

DS? App lication s in Communicatio ns

11.1

ser ial data to a serial por t. T he first interru p t migh t come From the IJfA con verter and the second from a hardware time r that is of ten hu ilt o n the same l C as the DSP device. The progra m mer must e nsure that these t\A'Ointerrupt s wi II be processed co rrectly. regardle ss of whe n the interrupts oc c ur. includi ng the case o r one int er ru p t occ urring while a second interrupt is being proc ess ed , f or our example . the data to the Of A c onverter must be processed bet ore the next Of A request is received. If this is not done . there wi ll lar ge amou nts of sig nal dis tortion a ssoc iat ed wi th a miss ing data o utput. A simple plan t hat e nsure s a min imu m am o unt of tim e wi ll he a vailable for in terrupt prucesving is to use on ly one interr upt that occ urs on a pe riod ic has is. Al tho ugh this may requ ire some p lanni ng to acc ommod ate all pro cesse s. the simplicity of th is

sc heme opens add it iona l in te rrup t processing tim e in two way s: • The re is no po ssibili ty of two interrup ts occurr ing at t he same time and t herefore no worst-case timing con stra ints to a llow all proc esses to be finished • No commu nicatio n is req uired hetwee n processe s about tasks tha t need to bc performed . That is, the operating sy ste m is b uilt -in to the program at de sign time. If there is o nly one interr upt, all interru pt p roc e ssi ng should be co mpleted in on e pe riod , leav ing the sys tem free at the timc of the ne xt interrupt. T his is the way that com mu nicat ion betwee n tasks is minimiv ed. This p rocc sving sh ould include e ver ythi ng that needs to be c ompleted beron: the next in terru p t. Add itionally. any proce ss tha t do es not need 10 be completed befo re the next inter-

rup t shou ld he placed in to the bac kgrou nd process . E xam ples o f this are the upda tin g of data for a display or the readi ng o f a knob o r a switch. Again . the se processes can be arranged in a scqucn ria l urdcr with nu need to mon itor t he lim e incre me nt need ed. So lo ng as the inte rru pt process leaves any tim e at al l. the bad ground will be processe d. D eterminin g whe ther thi s is happen ing at a fast e nough rate can be done at de sig n time. It will on ly happen more A

Con~ener

I I I I I I I I I I I I

Frequency & Amptitude

DS'

"""""

Program

'- - - -

I I I I I I I I I I I I I I I I I I

- - J

"",. ""

Fig t t. t o-coveran b loc k d iagram 01 the tone and n o ise gene rator. The kn o b con trol s both t he frequen cy of th e sine-wa ve ge ner ators and the am plitud es of the three signats. The fu nction 01 the knob is determ ine d b y th e push butto ns. The 16-character display is al so dr iv en by th e Interfac e circuit ry een trcnee by t he OSP so ft war e.

Box 6 • DSP routine t o set phase increment f o r sine-wave generator. { Frequency in Hz in the ar register. To convert to a phase inc rement we need to multiply by 65536148000. But in the U S arithmetic. the bigge st value is 1.0. So. we mUltiply by FA2PH:O .S"6S5J&148000=0.6827 and the n shift left 1 bit. the same as multiplYIng by 2. } FR2PH=OX5762 : I Hell. for 0.6827 in 1.15 format } .const { And the code in the main body of the program: } myO",FR2PH: mr: ar "myO (55); { The tracncnar multiply . and} sreasnm mrl by 1 (hi): { the multipty by 2, which isl s res r o r Ishift mru by 1 (10); { in two pa rts 10 get LS bit }

asp Applications

in Communications

11.1 1

Fig t t.tt -cosctucecepe trace of the Aud io Genera tor out pu t. One sine wa ve is set to 150-mV RM $ and the other 10 zer o. The no ise level is 50-mV RMS makin g th e SIN 9.5 dB (20 0 10g(3» . Th e sine- wave fr eq uenc y Is 1000 Hz.

wa re and hard ware functio ns invo lved . T he ind ivid ual func tio ns. such as sinewave genera tion , knob co ntrol a nd LCD d isplay have all be cove red earlier and will not be repeated here . T he deta ils of the integration of thes e pro gram com po nen ts

can be seen in the full list ing that i-, av-a ilable i n the progra m cl t tbox.dsp o n the ('D· R O ~1 that accom pani es this book, T he more in teresting areas arc the details Ihal must be ha ndle d 10 make the sig nal ge ne rato r o perate properly. Fo r inst ance. the d isplay for freq uency il in intege r Hz. from 110:W.OOO. The vinewave generato r ha, a resol ution of about 0.73 Hz. Th e kno b co uld he used to change frequency in either in steps of 1 Hz o r O, 7) Hz. Eit her way. a co nversion must he made 10 the oth er resolution step. Thc method used was to always c hange the desired frequency by I Hz. and then to co nvert thb to a phase inc remen t co rres pond ing to the 0.73 Hz step . T his results i n the kno b alway-s prod ucing a visit-ole freq uency c hange on rhe display. hut about 1/3 of the pocvible gene rator freque ncie s are not used . The conversion from a frequency in the AR register 10 a phase increme nt in the SR 1 regicte r is as follows in 80'1: 6 . Figs 11.11 and 1 1.12 are e xampl e wavefo rm o utput s h om the Audio Ge ne rato r, Outp ut leve ls and freq uencies are shown in the captio ns.

Fig 11.12-QscWoscope tra ce of t he Au dio Generator output. The sinewaves are of equal amplit ude and the frequen cies are 700 an d 1900 Hz. The nois e is set t o zero .

If the Of A co nve n e r is. operated be low its ove rload poi nt the di stortion. ind ud ing inrc rrnod ulerion. ca n be ex pected to be very small. T he princ iple drawback to this appro ac h is the limited frequenc y range. For the hard ware used herc it is not practica l to ope rat e much above 20 kll z.

11 .4 AN 1S·MHZ TRANSCEIVER Thi.. CW/S SB tran..cet ver o pcrarc-, in the l r -mete r amateur band fro m 18.068 10 1&. 168 M H1.. Dire ct co nve rsi on. as d iscussed in C hapters 8 and 9. is used for both the receiver and trans mincr. A ll RF functio n, arc huilt with con vention al hardware. but the audio fu nctio n, are DSP based, In addi tion . co ntrol function s were delegate d 10 the DSP, to the e xtent possible . The ge nera l arrangeme nt of the tra nscciv er is cho....-n i n the bloc k d iag ra m. f ig 11.13. T he receiver begi ns with a sing le tu ned circ uit a nd an RF ampl ifie r. T he co nsid erations for sig nal-to- nois e rati o. dyn amic ra nge and LO rad iation we re disc uss ed in C hapter 8 a nd app ly her e. I n

11 .12

Chapter 11

The 18- MHz Tr ansce iver .

order to use the sa me filte r.. and mix e rs on bo th receive and tra nsmit, ther e is a PI1'\ d iode switch follo wing the RF a mplifie r. Fo r reception. this s witc h also pro vides a simple met hod for man ua lly co ntro lling the RF ga in. as the PIN d iode can also be used as an adj usta ble rcvistor. Tw o mixers arc con nected to the Rf cir c ults thro ugh a po we r di vid er. A 90de gre e powe r d ivide r supplies the conversi on osci llato r fo r the two mixe rs. In recepu on. this cre ates t he ' fn-phase ' and 'Q uad rature' o r I and Q sig nals at aud io. After Iow-pass fi ltering. a n AI D co nverter that is part of ihe DS P board . d igitizes the two vignal v.

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.

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Chapter 12

52-MHz centerfreq. 2 MHz Bandwidth 1 dB loss LO Premix filter

47.5--MHz c ente r treq . filler

.

....... ........, ... .. . . C

. ... ... .. ... ... . . . . ... ........ ..

, ,

,

,

C

C

,",

L

C

C

: Cc1

L

L

,

L

C Co

,",

, Cc2

L

C

,

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: Cc2

: Cc1

............. .... .......

,......... ......., .............. ..,... .. ..... .. ... .,........... .... ....... ..'1'... ..., \

-

t.

.

,

.

L 10T 0.50' i.d. 0 75' lo ng bare number 18 copper in and out taps 1 full tum from g round end C 50 pF air variable Cc 0.25" gimmick twisted #22 Teflon Covered

Fig 12.5 8-The 47.5-MHz p rem i x oscillator f ilter.

All L 10T 0.50 " l.d. 1.00" long bare number 18 copper in and out taps 1 full turn from ground end C 50 pF air variable Cc 1 0.32" gimmick twisted #22 Teflon Cove red Cc2 0.25" gimmick twisted #22 Tef lon Covered Fig 12. 57-Schematic of the 52-MH z pr emix. f ilt er.

when luning for UHFDX. In addit ion , the audi o dis tortion of the co mmerc ial radio con t ribute d to ope rat or fatigue over the course of a weekend cont est . The hom e-

brew S2-.\1Hz tra nscei ver has no spurious response s or birdies, and all undes ired outputs are more than 70 dB be low t he desire d o utput.

52-MHz LO Output Amplifier

Mod ular c onstr ucti on with ind ividu al shield ed modul es. and a spac io us cabi ne t. contribu tes to a very large piece of rad io equ ipm ent with fine per for manc e. T his 52-MH/. tunable IF is a "work in progrevv," wit h unfinis hed a udio gain co ntro l. metering . and mode selec tio n fun,', lio ns, It bas been in ser vice for b yca r-. J. nJ e very yea r or so a func tion wi ll be adde d. T here is ample room inside for additio nal c ircui t mod ules, a nd roo m on the rr.uu pane l fo r add itiona l cnmr ul s

+12 T

RH O

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RH O

tao

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112 W

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1

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From Premix FiRer

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~ +1 3 dBmT

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0.01 MA.V· ll

0

0,01

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HP50 62· 31&8

0,0 1

HP5 062·3 1&8

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0.01

f--------0 +13 dBm R

a ll RFC. 12T FT25·43

I

0.0 1

+12 R

Fig 12.59-The 52-MHz prem ix. LO output amplifier.

Field Operat ion, Portable Gear and Integrated Stations

12.39

52 MHzcenter freq . filter

LO Mixer

...........

52 MHz out

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. c.

•

c

: Cc 47.5 MHz in 1 6t T37· ~

151137·2

33 0

o o

4.5 MHz in +13 dBm

33 0

L

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L

+10 dBm 4 ,7k

+12 T

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l 10T 0.50' l.d, 0,75' long bare number 18 copper in andouttaps 1 full tum from ground end C 50 pF air variable Cc 0.25" gimmick twisted #22Teflon Covered

Fig 12.60-Schematic of the premix LO m ixer. Fig 12 .63-The 52-MH z filter.

52 MHz La Quadrature Hybrid with phase trim La in

61 T37 · 1O

o

o

LO O

phase trim

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