Towards THz Chipless High-Q Cooperative Radar Targets for Identification, Sensing, and Ranging (Springer Theses) 3031049756, 9783031049750

Table of contents :
Supervisor’s Foreword
Abstract
Acknowledgments
Contents
Acronyms
1 Introduction
1.1 Motivation
1.2 Why High-mm-Wave and THz Tags?
1.3 Thesis Overview
References
2 High-Q Resonators for Chipless RFID and Sensing
2.1 Passive Backscattering
2.1.1 RCS of Conventional Targets
2.1.2 Maximum Range
2.2 High-Q Resonators
2.2.1 Estimating Material Losses
2.2.2 RCS of a Resonating Tag
2.3 Clutter Suppression Methods
2.3.1 Harmonic Generation
2.3.2 Out-of-Band Channel Equalization
2.3.3 Time-Gating
2.4 Identification and Sensing with High-Q Resonators
2.4.1 Identification and Sensing
2.4.2 Hybrid Modulation, Ranging Accuracy, and Sensing Accuracy
References
3 Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves
3.1 Sensing with a Single Air-Cladded High-Q Resonator
3.1.1 Temperature Sensor for Machine Tools
3.1.2 Sensing Other Physical Parameters Through Controlled Near-Field Variations
3.2 Hybrid Modulation Enabled by a High-Q Resonator
3.2.1 Hybrid Modulation Scheme Based on a High-Q Resonator and Phase-Coded TDR
3.2.2 Implementation in Ultrawideband
3.3 Limitations and Performance Deterioration Due to Lossy Materials and Packaging
References
4 Electromagnetic Band Gap (EBG) High-Q Resonator Concepts at mm-Waves
4.1 High-Q Resonators in a Metallic Bed of Nails (BoN)
4.1.1 Single Resonator Design
4.1.2 Multi-resonator Tags
4.1.3 Discussion
4.2 High-Q Resonators in a Full-Dielectric Photonic Crystal (PhC)
4.2.1 Single Resonator Design
4.2.2 Multi-resonator Tags
4.2.3 Discussion
4.3 Potential and Limitations of EBG-Based High-Q Resonators
References
5 High-RCS Wide-Angle Retroreflective Tags Towards THz
5.1 Corner Reflector and Resonator Array Integration
5.1.1 Q-Factor and RCS of Resonator Arrays
5.1.2 Coding with a Frequency Selective Surface (FSS)
5.2 Lüneburg Lens with Integrated High-Q Coding Particles
5.2.1 80GHz Lüneburg Lens Tag
5.2.2 240GHz HR-Si Lüneburg Lens Tag
5.2.3 Fused Silica Ball Lens with FSS
5.2.4 Discussion
5.3 Tag Identification and Ranging with a FMCW Radar at 80 and 240GHz
5.4 Tag Identification and Ranging in Cluttered Environments at 80GHz
References
6 Conclusion and Outlook
Appendix A Tag Dimensions
Appendix B Measured Unloaded Q-Factors of Metal Cavities
Appendix Curriculum Vitae

Citation preview

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Alejandro Jiménez-Sáez

Towards THz Chipless High-Q Cooperative Radar Targets for Identification, Sensing, and Ranging Doctoral Thesis accepted by Technical University of Darmstadt, Darmstadt, Germany

Author Dr. Alejandro Jiménez-Sáez Institute of Microwave Engineering and Photonics Technische Universität Darmstadt Darmstadt, Germany

Supervisor Prof. Rolf Jakoby Institute of Microwave Engineering and Photonics Technische Universität Darmstadt Darmstadt, Germany

ISSN 2190-5053 ISSN 2190-5061 (electronic) Springer Theses ISBN 978-3-031-04975-0 ISBN 978-3-031-04976-7 (eBook) https://doi.org/10.1007/978-3-031-04976-7 © The Editor(s) (if applicable) and The Author(s), under exclusive license to Springer Nature Switzerland AG 2022 This work is subject to copyright. All rights are solely and exclusively licensed by the Publisher, whether the whole or part of the material is concerned, specifically the rights of translation, reprinting, reuse of illustrations, recitation, broadcasting, reproduction on microfilms or in any other physical way, and transmission or information storage and retrieval, electronic adaptation, computer software, or by similar or dissimilar methodology now known or hereafter developed. The use of general descriptive names, registered names, trademarks, service marks, etc. in this publication does not imply, even in the absence of a specific statement, that such names are exempt from the relevant protective laws and regulations and therefore free for general use. The publisher, the authors, and the editors are safe to assume that the advice and information in this book are believed to be true and accurate at the date of publication. Neither the publisher nor the authors or the editors give a warranty, expressed or implied, with respect to the material contained herein or for any errors or omissions that may have been made. The publisher remains neutral with regard to jurisdictional claims in published maps and institutional affiliations. This Springer imprint is published by the registered company Springer Nature Switzerland AG The registered company address is: Gewerbestrasse 11, 6330 Cham, Switzerland

Supervisor’s Foreword

It is a great pleasure to introduce Dr. Alejandro Jiménez Sáez’s thesis work, accepted for publication within Springer Theses, recognizing outstanding Ph.D. research. Dr. Jiménez joined my research group at the Institute of Microwave Engineering and Photonics in May 2017 after receiving a double degree, a Master in Elektrotechnik und Informationstechnikat Technische Universität Darmstadt and Máster Universitario en Ingenierìa de Telecomunicacionesat at Universitat Politécnica de Valéncia, receiving the best student awards from both universities, respectively. His research was carried out within the DFG-funded Collaborative Research Center SFB/TRR 196—Mobile Material Characterization and Localization by Electromagnetic Sensing (MARIE). He completed his doctoral study with an oral defense on 18th October 2021. In January 2022, he received the award for the best doctorate in 2021 within the Department of Electrical Engineering and Information Technology from the association Freunde der Technischen Universität Darmstadt. Ubiquitous sensor systems will shape the coming decades to engage, identify, locate, and authenticate billions of things and monitor physical and environmental parameters. A disruptive solution in critical infrastructure is battery-less, fullypassive devices, consisting of electromagnetic structures and sensing elements only, typically operating in the Radio Frequency (RF) and microwave range. This class of chipless sensor tags preserves the advantages of RFID technology in terms of both ease of use and cost-effectiveness, enabling reliable and long-time operation without any maintenance. Given the absence of any electronic circuit and battery, chipless sensor tags are potentially suitable for remote, harsh, and hazardous environments with difficult access, as well as in the presence of dust, oil, high temperatures, strong vibrations, corrosive gases, radioactivity, or ionizing radiation, where chipequipped sensor tags are not reliable, or its implementation becomes increasingly complex. However, in contrast to chip-based tags, which can employ multiple access schemes and different frequency bands for the interrogated and backscattered signal as well as amplification of the backscattered signal by using active devices, these chipless tags are inherently (1) unable to encode a high number of bits, revealing very limited coding capacity, (2) unable to overcome strong clutter particularly from

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objects around the tag, and (3) feature low Radar Cross Sections (RCS) proportional to the square of their footprint, providing limited readout ranges. All these issues are addressed in Mr. Jiménez’s thesis by systematically investigating the applicability and feasibility of resonators with high-quality (high-Q) factors from microwaves towards THz: (1) for wireless pressure and high-temperature (up to 400 °C) sensors at microwaves (2.5 GHz to 22 GHz) with long ringing times to suppress clutter by appropriate time-gating methods in the remote reader for reliable readouts in dynamic environments, and (2) for chipless RetroReflective Tags (RRTs), operating from millimeter waves (20 GHz to 40 GHz and 65 GHz to 110 GHz) towards THz (220 GHz to 330 GHz). These higher frequencies are chosen to meet the scientific objective of MARIE for precise self-localization of a mobile robot or flying drone by trilateration of fixed chipless RRT landmarks, whose positions are defined and known. An envisioned application scenario is the self-localization of a mobile robot navigating and providing real-time information about a burning building or industrial hall to autonomously localize unconscious people, inflammable materials, high-voltage cables, and other artifacts with sub-mm resolution. At higher frequencies, however, low-loss materials for high-Q dielectric resonators are difficult to find. Moreover, to realize tags with higher coding capacity, several resonators have to be embedded into a well-defined, low-loss carrier material platform for mechanical stability. To achieve this objective, Mr. Jiménez proposes and demonstrates for the very first time frequency-coded electromagnetic bandgap structures, specifically metallic Bed of Nails (BoN) and dielectric Photonic Crystals (PhC) slabs, where several high-Q cavities are embedded with low-order resonance modes. While metallic BoN tags are produced by CNC machines, dielectric PhC slabs of high-resistive silicon and alumina are processed by deep reactive ion etching, a well-established technology, and by a novel additive manufacturing technology named Lithography-based Ceramic Manufacturing (LCM), respectively. This work was in collaboration with partners within the SFB/TRR 196 MARIE. Both approaches permit the realization of compact monolithic multi-resonator tags with long ringing times and increased coding capacities between 6 and 12 bits, providing some advantages in the fabrication and handling as well as for the design of the resonators by defining their resonance mode, resonance frequency, and quality factor. Such BoN and PhC tags, however, still face very limited readout ranges, in particular at higher frequencies, since their monostatic RCS drops with the square of the frequency. Moreover, the tag’s RCS angular pattern is typically small, thus, not feasible for localization. Hence, to increase RCS and to provide a wideangle response for precise localization at higher frequencies, Mr. Jiménez proposes different concepts to realize chipless RetroReflective Tags (RRTs): (1) by combining retroreflective corner reflectors with (a) an integrated array of high-Q dielectric resonators, as well as (b) a low-Q Frequency Selective Surface (FSS) by replacing one of the metallic walls of the corner reflector, and (2) by combining retroreflective ball and Lüneburg lenses with a reflective layer of PhC slabs with embedded high-Q resonators, defined as coding particles, distributed along the focal area on the backside of the lenses. While these retroreflectors and lenses increase the overall

Supervisor’s Foreword

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RCS up to the square of their gain (neglecting losses and reflections) over a wideangular range, necessary for long-range readout and high-localization accuracy with sufficient signal-to-noise-and-interference ratio at the reader, the multiple-resonator tags as well as the FSS will provide the unique frequency signature (code) for identification. Mr. Jiménez finally characterized and tested some realized chipless RRTs in characteristic indoor environments at 80 GHz and 240 GHz, using a Vector Network Analyzer (VNA) and available FMCW radars as a reader, including various clutter suppression techniques. Especially, the concept with Lüneburg lenses and identical high-Q alumina PhC coding particles as a reflective layer along its focal area on the backside represents a scientific milestone. It proves its suitability in all indoor measurement scenarios over a wide bandwidth and superior readout ranges over 2 m with a VNA (which can be significantly increased by a high-performance radar), demonstrating its potential as energy-autarchic and robust cooperative radar targets for localization in cluttered dynamic environments at millimeter and THz waves. Darmstadt, Germany February 2022

Prof. Dr.-Ing. Rolf Jakoby Executive Director

Abstract

Radio-Frequency Identification (RFID), wireless sensing, and precise localization are key technologies driving the technological revolution of Industry 4.0 through the general deployment of the Internet of things by providing information about objects such as their state and position, as well as their surrounding environments. As a hardware solution to applications where the use of semiconductor-based technology is not possible or increasingly complex, chipless identification and sensing through cooperative radar targets has been a subject of research for now over a decade. One field of research is the realization of targets that can operate in harsh environments such as high temperatures, strong vibrations, corrosive gases, or under ionizing radiation. By being fabricated with a few materials able to withstand these conditions, their long-term behavior can be easier to predict. However, the wireless readout of chipless cooperative radar targets is unreliable in dynamic reflective environments, where their linear low-power backscattered signatures cannot be continuously distinguished from clutter by a remote reader. This work investigates the systematic use of high-Q resonators as the key coding element of cooperative radar targets to overcome clutter. Due to their high-quality factors, the backscattered signature can outlast clutter and permit reliable readouts in dynamic environments, as well as its integration in other types of cooperative radar targets for joint identification, sensing, and ranging capabilities. This is first demonstrated with temperature and pressure sensors in the microwave frequency range, which include the characterization of a novel temperature sensor for machine tool monitoring up to 400 °C, as well as inside the machine. In parallel, the progress in semiconductor technology is drastically reducing costs and pushes the performance of radar modules which can be used as a reader. An example is the deployment of the second generation of automotive radar at 77 GHz in the last years. A similar development is expected for high-mm-Wave (high-mmWave) and THz (THz) systems, where recent advances in sources and detectors have steeply increased the interest in exploiting the inherent advantages of these frequencies for wireless applications: smaller antennas, wider absolute bandwidths, and higher integration of components in small areas. These advantages increase the potential of devices operating at these frequencies, although the shorter wavelength ix

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also raises their fabrication complexity. The realization of chipless cooperative radar targets and, especially, of high-Q resonators poses additional difficulties at mm-Wave and THz frequencies such as higher metal losses, lack of high-permittivity low-loss materials, and the small size of the resonators. This work investigates and demonstrates the use of metallic as well as dielectric EBG (EBG) structures to overcome these constraints, as well as to enhance the capabilities at mm-Wave and THz frequencies compared to microwave frequencies by permitting the realization of compact monolithic multi-resonator cooperative radar targets. Nonetheless, it becomes evident that the miniaturization with the frequency of mm-Wave and THz cooperative radar targets significantly reduces their RCS (RCS), thus their backscattered power, reducing the maximum range and greatly limiting their applications. This work, relying on that same miniaturization, proposes and demonstrates the integration of resonators inside larger retroreflective structures conventionally used for ranging to provide them with a long-range frequency coding robust against clutter. Two different approaches are presented based on the integration of resonators into corner reflectors and Lüneburg lenses, and their scalability to higher frequencies is verified at around 80 GHz and 240 GHz. By demonstrating the successful readout of these cooperative radar targets in cluttered dynamic environments, as well as with readers based on FMCW (FMCW) radars, this work paves the way towards chipless high-Q radar targets for identification, sensing, and ranging, from conventional microwave, via mm-Wave, towards upcoming THz frequencies.

Acknowledgments

It’s been 6 years since I first came to Darmstadt within the double master’s program, and I am grateful for the support received from many people. I would like to start with a huge thanks to Prof. Jakoby for giving me the opportunity, professional stability, and advice for me to work on this dissertation. I greatly appreciate your support during this process. Special thanks also to Martin Schüßler, who welcomed me to the institute when looking for a student job shortly after coming to Darmstadt and whose advice escorts me and helps me set the right priorities. Without you, this journey would have been both longer and less satisfying. Finally, thanks to my colleagues in Darmstadt for the fruitful discussions, especially to Matthias Nickel, Jesús Sánchez Pastor, and Tom Burmeister. Getting my curiosity into these topics was not a given, and my Antenas professor in Valencia, Alejandro Valero-Nogueira, definitely has a big share of the blame. Having researched under your guidance is something I am deeply grateful about. From the beginning, I looked forward to being part of a large project and learn about its structure and the motivations and advances of the various subprojects. I can now say that the SFB/TRR 196 MARIE project has met all my expectations. I am genuinely grateful for the fruitful discussions and collaborations I had with Lisa Schmitt, Lukas Piotrowsky, Masoud Sakaki, and Benedikt Sievert. Special thanks also to Christian Mandel, Niels Benson, and Thomas Kaiser. Furthermore, I want to thank Petr Kadˇera for being the right person at the right moment interested in initiating a collaboration. If I am confident to say that I enjoyed these last 6 years and feel healthy after them, it is greatly supported by the outstanding people I have around me. I am thankful to the salsa community for allowing me to conceal leisure time, sport, and social contact within a single activity. Dancing with cheerful people after a tough workday is one of the best remedies I have undergone to relativize the difficulties, disconnect, and wake up the next day with renewed motivation to look for solutions. Here, I want to give special thanks to my friends in the Rueda group and the show group Agua, which have been a great support throughout all these years and to which I proudly refer to as my German family.

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Finally, thanks to my genuine family, which supports me in every decision I take and, despite the difficulties, is always willing to come to Darmstadt to celebrate each remarkable step I take in my journey. Especially to my grandmother, my parents, and my sister. I hope that my presentation in the defense helped you finally understand what I am doing here, lost in Germany. If I am not really lost, then this is significantly due to Paula, thanks to Paula, whose sustained support and company not only improves my mood, but my life as a whole.

Contents

1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.1 Motivation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.2 Why High-mm-Wave and THz Tags? . . . . . . . . . . . . . . . . . . . . . . . . . . 1.3 Thesis Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

1 1 3 5 6

2 High-Q Resonators for Chipless RFID and Sensing . . . . . . . . . . . . . . . . 2.1 Passive Backscattering . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.1.1 RCS of Conventional Targets . . . . . . . . . . . . . . . . . . . . . . . . . . 2.1.2 Maximum Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2 High-Q Resonators . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2.1 Estimating Material Losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2.2 RCS of a Resonating Tag . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.3 Clutter Suppression Methods . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.3.1 Harmonic Generation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.3.2 Out-of-Band Channel Equalization . . . . . . . . . . . . . . . . . . . . . 2.3.3 Time-Gating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.4 Identification and Sensing with High-Q Resonators . . . . . . . . . . . . . . 2.4.1 Identification and Sensing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.4.2 Hybrid Modulation, Ranging Accuracy, and Sensing Accuracy . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

9 10 12 14 15 19 20 22 24 24 24 25 25

3 Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.1 Sensing with a Single Air-Cladded High-Q Resonator . . . . . . . . . . . 3.1.1 Temperature Sensor for Machine Tools . . . . . . . . . . . . . . . . . . 3.1.2 Sensing Other Physical Parameters Through Controlled Near-Field Variations . . . . . . . . . . . . . . . . . . . . . . . 3.2 Hybrid Modulation Enabled by a High-Q Resonator . . . . . . . . . . . . . 3.2.1 Hybrid Modulation Scheme Based on a High-Q Resonator and Phase-Coded TDR . . . . . . . . . . . . . . . . . . . . . .

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3.2.2 Implementation in Ultrawideband . . . . . . . . . . . . . . . . . . . . . . 3.3 Limitations and Performance Deterioration Due to Lossy Materials and Packaging . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 Electromagnetic Band Gap (EBG) High-Q Resonator Concepts at mm-Waves . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.1 High-Q Resonators in a Metallic Bed of Nails (BoN) . . . . . . . . . . . . 4.1.1 Single Resonator Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.1.2 Multi-resonator Tags . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.1.3 Discussion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.2 High-Q Resonators in a Full-Dielectric Photonic Crystal (PhC) . . . 4.2.1 Single Resonator Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.2.2 Multi-resonator Tags . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.2.3 Discussion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.3 Potential and Limitations of EBG-Based High-Q Resonators . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 High-RCS Wide-Angle Retroreflective Tags Towards THz . . . . . . . . . . 5.1 Corner Reflector and Resonator Array Integration . . . . . . . . . . . . . . . 5.1.1 Q-Factor and RCS of Resonator Arrays . . . . . . . . . . . . . . . . . 5.1.2 Coding with a Frequency Selective Surface (FSS) . . . . . . . . . 5.2 Lüneburg Lens with Integrated High-Q Coding Particles . . . . . . . . . 5.2.1 80 GHz Lüneburg Lens Tag . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.2.2 240 GHz HR-Si Lüneburg Lens Tag . . . . . . . . . . . . . . . . . . . . 5.2.3 Fused Silica Ball Lens with FSS . . . . . . . . . . . . . . . . . . . . . . . 5.2.4 Discussion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.3 Tag Identification and Ranging with a FMCW Radar at 80 and 240 GHz . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.4 Tag Identification and Ranging in Cluttered Environments at 80 GHz . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

43 51 53 57 57 59 61 67 68 69 81 87 88 90 93 94 95 100 110 112 116 118 119 120 123 127

6 Conclusion and Outlook . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 131 Appendix A: Tag Dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 135 Appendix B: Measured Unloaded Q-Factors of Metal Cavities . . . . . . . . . 139 Curriculum Vitae . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 141

Acronyms

AMC Al2 O3 BoN BST CNC CR CRLB DR DRIE EBG EIRP EM FEM FFT FIT FMCW FSR FSS FSR GWG HPBW HR-Si IFFT ISM LCM LTCC MEMS MIMO mm-Wave PEC PhC

Artificial Magnetic Conductor Alumina Bed of Nails Barium Strontium Titanate Computer Numerical Control Corner Reflector Cramér-Rao Lower Bound Dielectric Resonator Deep Reactive Ion Etching Electro-Magnetic Bandgap Effective Isotropic Radiated Power Electro-Magnetic Finite Element Method Fast Fourier Transform Finite Integration Technique Frequency-Modulated Continuous-Wave Free Spectral Range Frequency Selective Surface Free Spectral Range Gap Waveguide Half Power Beam Width High-Resistive Silicon Inverse Fast Fourier Transform Industrial, Scientific, and Medical Lithography-based Ceramic Manufacturing Low-Temperature Cofired Ceramics Microelectromechanical systems Multiple-Input Multiple-Output Millimeter-Wave Perfect Electric Conductor Photonic Crystal xv

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PLA PVC radar RCS RF RFID SAR SEM SER SIW SINR TDR TE THz TM TRMLAT UWB VNA

Acronyms

PolyLactic Acid PolyVinyl Chloride RAdio Detection And Ranging Radar Cross Section Radio Frequency Radio-Frequency Identification Synthetic Aperture Radar Scanning Electron Microscope Symbol Error Ratio Substrate Integrated Waveguide Signal-to-Interference-plus-Noise Ratio Time-Domain Reflectometry Transversal Electric Terahertz Transversal Magnetic True-Range Multilateration Ultra-Wideband Vector Network Analyzer

Chapter 1

Introduction

1.1 Motivation The interest in wireless Radio-Frequency Identification (RFID) and sensing is growing with the appearance of applications in sectors such as retail, agriculture, security, and supply chain management. By providing additional real-time information of different processes, they assist in the optimization and automatization of tasks, improving parameters such as efficiency and reliability. The market for wireless sensors is expected to increase from US$4.4 billion to US$16.0 billion with a Compound Annual Growth Rate of 24.0 % from 2020 to 2026 [1]. Some key trends in this market are improvements in power consumption and energy harvesting, increasing the autarchic lifetime of such sensors. Furthermore, for the establishment of the Internet of Things, efficient communication is needed between multiple sensors. These developments show both ecological and economical improvements, as novel and inter-connected sensors with long lifetimes and low-energy consumption reduce their environmental footprint, while facilitating improved process efficiencies. Similarly, the market of RAdio Detection And Ranging (radar) sensors is expected to increase from US$10.3 billion to US$22.0 billion with a Compound Annual Growth Rate of 13.5 % from 2019 to 2025 [2] propelled by the appearance of highly cost-sensitive consumer market applications such as Automatic Emergency Braking in automotive radars. This growing interest has been rapidly pushing down prices for mm-Wave (mm-Wave) (30 GHz to 300 GHz) Frequency-Modulated ContinuousWave (FMCW) radar modules, which can also be used as readers for chipless identification, sensing, and ranging. The term chipless RFID is a subgroup of chipless cooperative radar targets that emerged as an alternative to the well-established chip-based backscatter RFID technology and can be divided into two groups based on their objective:

© The Author(s), under exclusive license to Springer Nature Switzerland AG 2022 A. Jiménez-Sáez, Towards THz Chipless High-Q Cooperative Radar Targets for Identification, Sensing, and Ranging, Springer Theses, https://doi.org/10.1007/978-3-031-04976-7_1

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2

1 Introduction

• to substitute chip-based RFID as a lower-cost alternative, and • to complement chip-based RFID in environments where the use of semiconductors becomes increasingly difficult. While the former group concentrates on the potential realization of chipless cooperative radar targets below the unit price of US$0.12 to US$1 imposed by chips [3], this work focuses on the latter. In the following, the widespread term tag is used to refer to any chipless cooperative radar targets. This work advances towards the utilization of chipless tags where chip-based and wired alternatives become increasingly complex and unfavorable, either due to (i) impossibility of operation, or (ii) reliability concerns. (i) comprises environments where high-temperatures above 150 ◦C are reached. Further, conditions that greatly increase the complexity of using semiconductors include the presence of ionizing radiation, corrosive gases, and strong vibrations. All these environments are referred to as harsh environments throughout this work. (ii) includes the operation in harsh environments as well as applications where frequent maintenance and replacement of the tags would considerably drive up the costs. One of such applications is structural health monitoring, where the tags are expected to reliably operate over several decades, and frequent maintenance could affect the overall system costs in a greater extent than the cost of each tag. Chipless tags can offer high reliability due to their unsophisticated structure. The use of a single or few materials permits the accurate modelling and prediction of their behavior in harsh environments and over long periods of time. Furthermore, the absence of a programmable device guarantees that no online hack is possible, and that any malevolent manipulation of the system on the tag side lacks the scalability or one-size-fits-all solution that makes most of these attacks profitable in the first place. In the case of conventional chip-based backscatter RFID technology, active, semipassive, and passive backscatter tags can be found, whereas chipless technology is inherently passive. Furthermore, the lack of a microcontroller in chipless tags results in instant responses, since no power threshold needs to be surpassed for the circuitry to activate and start modulating its backscatter response. In chipless tags, the structure itself acts as the transponder, blending concepts from microwave engineering, radar, communication technology, signal processing, and material science in the tag conception. One of the main limitations of passive chipless tags is their readout in dynamic environments, which highly limits their use and maximum range. This is mainly caused by (i) the low power reflected by the tag, and (ii) high-power reflections from the tag’s structure and the environment, that are embraced by the term clutter. For the power densities foreseen in a chipless system, the channel can be modelled linearly, i.e., as a superposition of modulated reflections of the transmitted signal at the reader without the generation of new frequencies. Consequently, an effective technique to distinguish the response of the tag from clutter is the generation of harmonics in the tag, i.e., a backscattered signal at frequencies not contained in

1.2 Why High-mm-Wave and THz Tags?

3

the interrogation pulse. Nonetheless, due to its passive nature and distance to the reader, the power available in the tag is low, and no non-linearities can be efficiently exploited to generate harmonics of sufficient power above noise. The closest alternative are quasi-chipless tags that generate harmonics through Schottky diodes for efficient non-linear tags [4]. Although they do not share all the limitations of chipbased technology, for example, by allowing operation over 150 ◦C without additional thermal-handling measures, even the use of a single diode can limit the application of these tags in harsh environments and is thus not considered in this work. A promising alternative to non-linear tags are high quality factor (high-Q) or highly resonant tags. High-power clutter reflections occur from large walls and metallic surfaces, as well as 90◦ dihedral and trihedral corners that can occur within walls and furniture. However, high-power clutter is commonly (i) received as early reflections after the interrogation pulse is sent, and (ii) barely resonant. Based on these characteristics, an alternative for effectively detecting the backscattered signal from chipless tags among clutter based on highly resonant structures by using a metal cavity resonant at 2.4 GHz was presented in [5], and for ceramic dielectric resonators resonant at 3 GHz in [6]. The principle behind is that the slowly decaying backscattered response of high-Q resonators outlasts clutter and can therefore be distinguished from it with a single measurement and no additional channel information in the reader. To better understand the concept, an acoustic parallelism can be made with the characteristic ringing of wine glasses. Imagine a door is loudly closed (pulsed interrogation). When the sound reaches the wine glasses present in the room (tags), they receive some energy and start emitting sound at their resonance frequency, which can be perceived by a person some instants after the echoes from the door have vanished. In the investigated electromagnetic system, the electromagnetic waves travel faster, and the available readers offer a wide dynamic ratio and a precise frequency resolution, so that the resonance frequencies from different resonators can be distinguished (identification), and subtle shifts of the resonance frequency monitored (sensing). The state of the art of high-Q tags below 10 GHz before this work is summarized in Bernd Kubina’s dissertation [7].

1.2 Why High-mm-Wave and THz Tags? In recent years, the interest in high-mm-Wave and Terahertz (THz)1 frequencies has been increasing, mainly driven by advances in sources and detectors which lead to higher transmit powers, uncooled detectors, as well as integrated systems [8]. The applications of THz include but are not limited to (i) high-capacity pointto-point communications, (ii) spectroscopy of gases, (iii) identification of objects 1 In this work, the frequency ranges are defined to avoid any overlap leading to ambiguity: microwave frequency range from 0.3 GHz to 30 GHz, mm-Wave frequency range from (30 GHz to 300 GHz), which is further subdivided into three frequency ranges of constant relative bandwidths: lowmm-Wave (30 GHz to 65 GHz), mid-mm-Wave (65 GHz to 140 GHz), and high-mm-Wave (140 GHz to 300 GHz), and THz frequency range (0.3 THz to 30 THz).

4

1 Introduction

behind THz transparent materials such as polymers, plastics, and papers, and (iv) localization (including self-localization) with mm-range spatial resolution. All these applications benefit from the wide absolute bandwidths available at these frequencies. Furthermore, (ii) benefits from the presence of characteristic absorption peaks due to resonances of several gases at THz, permitting their recognition by the number and position of these absorption peaks. Finally, (iii) and (iv) benefit from shorter wavelengths: (iii) due to the interaction of the waves with thin materials commonly present in objects such as paint layers in automotive and art [9], while (iv) by higher reader gains in small areas, whose lower Half Power Beam Width (HPBW) improves the spatial resolution and acts as a spatial filter that reduces clutter reflections from directions other than the reader orientation. Within the Collaborative Research Center/Transregio 196—Mobile material characterization and localization by electromagnetic sensing, the achievement of THz sub-wavelength localization accuracies is envisioned for the support of coherent material characterization methods by mobile robots. Such material characterization needs to coherently process different measurements and aggregate their information. For this to succeed, these algorithms involve the use of phase information from multiple measurements and, therefore, require accurate knowledge of the reader displacement. For example, to apply Synthetic Aperture Radar (SAR) techniques, the movement of the reader around the object to be characterized helps enhance the obtained image, but the position must be accurately known by the reader [10]. Indoor localization systems tend to be conceptualized based on existing infrastructure to permit their fast and low-cost use in most buildings. However, even with the improvements expected with new mobile standards, 5G-based indoor localization accuracies are around a meter [11], and outlooks into 6G predict cm-level accuracies [12], although such predictions tend to be optimistic until the standards are deployed. For this reason, further development of infrastructure-based systems are necessary for systems requiring high accuracy and reliability are needed. Among infrastructure-based localization systems, visible light methods usually achieve the best accuracies [13]. However, they are limited to line-of-sight where the environment and any obstructing objects and radomes are transparent to visible light. This has awakened the interest in THz-based indoor localization as a complement to other methods. Due to the use of THz, huge bandwidths are available, which help improve the accuracy of ranging methods based on time-of-arrival down to a µm [14]. Besides, THz waves can propagate through several visually opaque materials, permitting the detection of targets unrecognizable by the human eye. Furthermore, the propagation of THz waves through visually opaque gases would help such systems operate in harsh environments, like a fire in an indoor scenario, where mobile robots can explore the environment assisted by a chipless-based localization infrastructure and send real-time information to the firefighters about the presence of hazardous materials, trapped people, or critical structural damage. Although the state-of-the-art is still far from such technology, their potential motivates the research into affordable identification, sensing, and ranging tags that could enable advances through the use of THz.

1.3 Thesis Overview

5

1.3 Thesis Overview First, this work further develops the high-Q temperature sensor based on a Dielectric Resonator (DR) presented in [7] towards the integration of high-Q sensors in industrial applications by: • demonstrating the design and characterization of a high-Q resonator for machine tools, showing successful detection of the resonance peaks even in high-reflective environments such as a real machine tool with the door closed, and monitoring from the outside while the sensor rotates at angular velocities up to 10,000 rpm. • showing that wireless high-Q sensor capabilities in cluttered environments are not limited to sensing variations in their own material properties, but can also sensitively react to near-field alterations. • demonstrating the potential of hybrid coding techniques with the simultaneous integration of wideband and narrowband coding schemes. Furthermore, this work investigates the possibilities and challenges of mm-Wave and THz frequencies for the realization and packaging of high-Q resonator tags. It recognizes and demonstrates for the first time the potential of metallic and dielectric ElectroMagnetic Bandgap (EBG) surfaces to solve the main challenges faced for the realization of high-Q resonating tags able to operate in dynamic environments in the mm-Wave and THz frequency ranges. The potential of several materials and processing technologies is studied for the realization of these tags. Different suitable hardware solutions are proposed depending on the application and operation temperature, including the use of Alumina (Al2 O3 ) with a novel additive manufacturing processing technique named Lithography-based Ceramic Manufacturing (LCM). Finally, it is demonstrated that EBGs structures permit the realization of not only single, but also multiple high-Q resonators within a single monolithic tag, achieving estimated coding capacities above 9 bits at room temperature. Tags using directive antennas can be implemented to counteract the reduction in the maximum readout range with decreasing operation wavelength of single resonators. However, this reduces the angular coverage of the tags and limits its use in dynamic environments, where the relative orientations between the reader and the tag are unknown or vary over time. For this reason, approaches are investigated to increase the backscattered power by directive configurations without reducing the angular coverage. The central idea behind is the modification of ordinary wideband retroreflectors commonly used for ranging, such as corner reflectors and partially metalized Lüneburg lenses, to achieve a frequency dependent retroreflective response that adds identification and sensing capabilities via integrated high-Q resonators embedded into EBG structures, without impeding their use for ranging. An example with a Lüneburg lens is schematically represented in Fig. 1.1. Different concepts based on corner reflectors and lenses are introduced and demonstrated in the midand high-mm-Wave frequency ranges, with measured readout ranges of 4 m and 2 m, respectively.

6

1 Introduction

Fig. 1.1 Transmitted and received signals by a reader in a scenario with (i) an uncoded and (ii) a coded wideband retroreflector. (i) The uncoded reflected signal of a partially metalized Lüneburg lens, ┏ = −1, is shown in brown. (ii) By substituting the metal on the back-side of the lens with a frequency-dependent reflective structure, ┏( f ), the reflected signal is frequency-coded, as shown in orange for a single resonator. Increasing the number of resonators increases the number of backscattered resonance notches and peaks

Finally, two steps are taken to demonstrate the potential use of the mm-Wave tags fabricated in this work in real applications. First, the compatibility of the developed frequency-selective tags is demonstrated with small, cost-effective, and highly integrated FMCW radars. Second, the successful readout of the fabricated mid-mm-Wave tags is demonstrated in several cluttered environments. This work is structured as follows: Chap. 2 introduces the fundamental concepts of radar and high-Q resonators used in this scientific work. Based on the high-Q temperature resonator concept presented in [7], this work further develops the single, air-cladded high-Q resonator concept in Chap. 3. This includes the realization of a temperature sensor for machine tool monitoring as well as other sensors in Sect. 3.1. Furthermore, the simultaneous use of narrowband high-Q resonators and other wideband modulation mechanisms within the same tag and bandwidth is demonstrated in Sect. 3.2. Following, Chap. 4 investigates the realization of mm-Wave high-Q resonator tags and shows that EBG structures not only enable the realization of highQ tags at these frequencies, but also enhance their performance by using a single low-loss material. Finally, Chap. 5 investigates how the trade-off between maximum range and angular coverage of tags at higher frequencies can be overcome by using resonators for the frequency coding of wideband retroreflectors.

References 1. Mordor Intelligence, Wireless Sensors Market - Growth, Trends, COVID-19 Impact, and Forecasts. https://www.mordorintelligence.com/industry-reports/wireless-sensors-market 2. Mordor Intelligence, Radar Sensors Market—Growth, Trends, COVID- 19 Impact, and Forecasts. https://www.mordorintelligence.com/industry-reports/radar-sensors-market

References

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3. Athauda T, Karmakar N (2019) Chipped versus chipless RF identification: a comprehensive review. IEEE Microw Mag 20(9):47–57 4. Bernd K, Christian M, Martin S, Rolf J (2015) Compact quasi-chipless harmonic radar sensor with a dielectric res-onator antenna. In: IEEE MTT-S international microwave symposium. IEEE, pp 1–3 5. Thomson DJ, Card D, Bridges GE (2009) RF cavity passive wireless sensors with time-domain gating-based interrogation for SHM of civil structures. IEEE Sens J 9(11):1430–1438 6. Bernd K, Martin S, Christian M, Arshad M, Rolf J (2013) Wireless high-temperature sensing with a chipless tag based on a dielectric resonator antenna. IEEE sensor. IEEE, pp 1–4 7. Kubina B (2016) Chipless-wireless high-temperature sensing in time-variant environments. In: IMP Dissertation, TU Darmstadt 8. Hillger P, van Delden M, Sampath Miriya Thanthrige U, Mostafa Ahmed A, Wittemeier J, Arzi K, Andree M, Sievert B, Prost W, Rennings A et al (2020) Toward mobile integrated electronic systems at THz frequencies. J Infrared, Millim Terahertz Waves 41(7):846–869 9. Montserrat Gomez-Sepulveda A, Hernandez-Serrano AI, Rad-Pour R, Koch-Dandolo CL, Rojas-Landeros SC, Ascencio-Rojas LF, varo Zarate AI, Hernandez G, Gonzalez-Tirado RC, Insaurralde-Caballero M et al (2017) History of Mexican easel paintings from an altarpiece revealed by non-invasive terahertz time-domain imaging. J Infrared, Millim Terahertz Waves 38(4):403–412 10. Batra A, El-Absi M, Wiemeler M, Göhringer D, Kaiser T (2020) Indoor THz SAR trajectory deviations effects and compensation with passive sub-mm localization system. IEEE Access 8:177519–177533 11. del Peral-Rosado J, Raulefs R, López-Salcedo J, Seco-Granados G (2017) Survey of cellular mobile radio localization methods: from 1G to 5G. IEEE Commun Surv Tutorials 20(2):1124– 1148 12. De Lima C, Belot D, Berkvens R, Bourdoux A, Dardari D, Guillaud M, Isomursu M, Lohan E-S, Miao Y, Noll Barreto A et al (2021) Convergent communication, sensing and localization in 6G systems: an overview of technologies, opportunities and challenges. IEEE Access 13. Rahman ABM, Li T, Wang Y (2020) Recent advances in indoor localization via visible lights: a survey. Sensors 20(5):1382 14. Thomas S, Bredendiek C, Pohl N (2019) A SiGe-based 240-GHz FMCW radar system for high-resolution measurements. IEEE Trans Microw Theory Tech 67(11):4599–4609

Chapter 2

High-Q Resonators for Chipless RFID and Sensing

Two distinct elements are present in the scenario considered throughout this work and depicted in Fig. 2.1: a reader which transmits an interrogation signal and receives the echoes, and a chipless reflective device integrating some identification, sensing, and/or ranging capability, i.e., a cooperative radar target or tag. Additional elements in the measurement scenario such as the floor, ceiling, walls of the room, furniture, and the mechanical support of the tag are undesired, but to some extent unavoidable elements whose contributions to the radar echoes are defined as clutter. This work focuses on the realization of tags that improve the detection, identification, sensing, and ranging capabilities of chipless wireless systems. Detection in which the presence or absence of a tag in a certain location can be distinguished from the background noise and clutter, identification in which each tag can be distinguished from each other, sensing in which the tag varies its identification code or signature according to the variation of a physical measurand like temperature, pressure, or surrounding dielectric material, to name a few, and ranging in which the distance between the tag and the reader can be estimated at the latter. Notice that the attribute chipless refers to the tag and not the reader. A chipless tag dispenses with the use of batteries and semiconductors to overcome some limitations of these technologies, enabling the autarkic operation of the tag and eliminating the need for maintenance. The reader is an active element able to transmit and receive Radio Frequency (RF) signals with a certain frequency bandwidth. Some active and even passive RFID systems based on harmonic generation should be modelled as a two-way communication link to predict whether the power received at the tag overcomes a minimum threshold for the activation of the chip circuitry or the efficient harmonic generation. In chipless tags, the low power and absence of diodes translates into highly linear channel and tag responses. The reader is the only active element that generates RF power, and no new frequencies are generated once the interrogation pulse is transmitted. The link budget © The Author(s), under exclusive license to Springer Nature Switzerland AG 2022 A. Jiménez-Sáez, Towards THz Chipless High-Q Cooperative Radar Targets for Identification, Sensing, and Ranging, Springer Theses, https://doi.org/10.1007/978-3-031-04976-7_2

9

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2 High-Q Resonators for Chipless RFID and Sensing

Stx = reader

Ptx Gtx 4πd2

Ptx Prx = Srx Arx

tag σ Stx σ 4πd2

= Srx

Fig. 2.1 Schematic of the considered monostatic radar system. The terms are defined in Sect. 2.1

and tag performance of a chipless measurement scenario can therefore be modelled as a conventional radar system. Throughout this work, a monostatic reader with a single antenna is assumed, but readers with more advanced configurations such as Multiple-Input Multiple-Output (MIMO) could enhance the system capabilities. It is interpreted that, if the tags developed in this work can operate with single-antenna monostatic readers, any improvements in the reader will help further improve the performance and reliability as long as the benefits prevail over the additional cost and energy consumption of such readers. Still, all the tags developed in this work should be able to operate with single-antenna reader configurations. In this work, the term interrogation pulse is often used to specify the pulse-like behavior of the interrogation signal and highlight its time-domain characteristics, although neither of the two reader types used, a Vector Network Analyzer (VNA) throughout the work, and a FMCW radar in Sect. 5.3, actually transmit a pulse. Nevertheless, due to the linear operation of all developed tags, the superposition principle applies, and the available readout signals can be transformed from the frequency domain to the time domain and vice versa using Fourier and inverse Fourier transformations, obtaining a pulse-like response. For this reason, the terms interrogation pulse and interrogation signal are used interchangeably to ease the understanding of the system properties. In this chapter, the basic principles of radar and its application for chipless systems are introduced in Sect. 2.1. Afterwards, the concept of high-Q resonators is presented in Sect. 2.2, including the extraction of material characteristics from their quality factor and the effect of resonator losses in the backscattered power. The chapter concludes with an overview of clutter suppression techniques in Sect. 2.3, and an analysis of the employed techniques for tag identification, sensing, and ranging in Sect. 2.4.

2.1 Passive Backscattering The radar equation predicts the received power according to the measurement environment. The reader transmits (tx) an interrogation signal with power Ptx with an antenna of gain G tx . As the interrogation propagates in the direction of the tag, its

2.1 Passive Backscattering

Gtx αfs

P (dB)

Ptx

11

σ = At,rx Gt,tx αfs

Prx

Arx

0

d distance

Fig. 2.2 Monostatic radar link budget. Blue corresponds to the reader, orange to the tag, and red to the channel

power density, Stx , decays with the square of the distance αfs = 1/(4πd 2 ) before a portion of the interrogation reaches the tag and is received according to the tag’s effective aperture At,rx . Now the process is reversed, and the tag backscatters a coded variation of the interrogation towards the reader with a certain gain G t,tx . The backscattered interrogation propagates towards the reader, and the power density, Srx , decays with the same free-space propagation factor αfs = 1/(4πd 2 ) before being received by the reader antenna according to its effective antenna aperture Arx . The corresponding two-way passive communication link budget is depicted in Fig. 2.2 and described by the radar equation Prx = Ptx G tx

1 1 Arx . At,rx G t,tx 4πd 2 4πd 2

(2.1)

The equation and the figure assume a monostatic case. The previous derivation can easily be extended to the bistatic case by considering different distances in the tx and rx case, as well as different angles and, if necessary, polarization mismatch in the tag’s At,rx G t,tx . All studied cases assume a single linearly polarized reader acting as both transmitter and receiver, so only the monostatic case is considered. Similarly, absorption in the atmosphere is neglected due to the short ranges considered, but could be considered by adding a term on each propagation path. The Radar Cross Section (RCS), σ, of the tag describes the amount of power that the tag backscatters in the direction of the reader and is defined as ηt Dt,tx , σ = At,rx · G t,tx = Alossless t,rx

(2.2)

where ηt includes any power loss in the tag, Alossless the lossless effective aperture, t,rx and Dt,tx is the directivity of the tag. Furthermore, the following general relationship can be used to convert any effective aperture to gain and vice versa

12

2 High-Q Resonators for Chipless RFID and Sensing

b

r

a a

(a)

(b)

(c)

Fig. 2.3 Conventional wideband radar targets. a Metal sphere, b flat plate reflector, and c triangular trihedral corner reflector Table 2.1 Maximum RCS of conventional targets in the optical region Sphere Flat plate Triangular trihedral corner RCS, σ

πr 2

4πa 2 b2 λ2

G A = 2, λ 4π

4πa 4 3λ2

(2.3)

where λ = c/ f is the wavelength, being c the speed of light in the medium, and f the frequency. Vacuum1 is commonly assumed as the propagating medium. Rearranging Eq. 2.1 with Eq. 2.3, the following expression predicting the received power is obtained G 2 λ2 σ Prx = Ptx tx 3 4 . (2.4) (4π) d

2.1.1 RCS of Conventional Targets An overview of three conventional radar targets can be seen in Fig. 2.3 and Table 2.1. In all cases, it is assumed that they operate in the optical region, i.e., their dimensions are much larger than the wavelength to avoid frequency-dependent fluctuations. In the following, the RCS of each target is compared to its aperture and retroreflective gain. The first considered target is a metal sphere. It has the special property of being orientation insensitive, i.e., its RCS does not vary with the interrogation angle. It can be observed that the geometrical cross-section is a fundamental part of the RCS of all targets. However, in the case of the metal sphere, its RCS is proportional to the geometrical cross-section, while for the flat plate and the triangular trihedral corner reflectors (in the following, trihedral corner reflector) it is proportional to

1

The subscript 0 is used to indicate when the wavelength or the speed of light are considered in vacuum, εr = 1.

2.1 Passive Backscattering

13

its geometrical cross-section squared. This difference can be better understood by reformulating the RCS either in terms of effective aperture or gain as σ = At,rx G t,tx = At,rx At,tx

λ2 4π = G G . t,rx t,tx λ2 4π

(2.5)

In the case of the sphere, it is more intuitive to work with the first expression. When an interrogation reaches the sphere, the amount of power that interacts with the sphere is proportional to its geometrical cross-section πr 2 . However, its reflections are isotopically distributed in space, i.e., its G t,tx = 1. Multiplying these terms gives the RCS included in Table 2.1. In the case of the flat plate and the trihedral corner reflector, the same case applies to the rx case, which is proportional to their geometrical cross-section. However, the reflection is not isotropically distributed in space, but focused on a certain direction. In transmission, both targets can be modelled as a 2D antenna array fully utilizing the . Calculating geometrical cross-section of the reflector, so that its gain G t,tx = At,tx 4π λ2 the geometrical cross-section of the flat plate and the retroreflector with the defined parameters, the RCSs provided in Table 2.1 are obtained. Notice that, the higher the frequency, the higher the RCS of a metal plate and a trihedral corner reflector with f 2 . This characteristic and the need to operate in the optical region, where the dimensions are much larger than the wavelength, makes the use of reflectors specially appealing at mm-Wave and THz frequencies. For example, a trihedral corner reflector with a = 3 cm has a RCS of −6.17 dBm2 at 80 GHz, which increases to 3.37 dBm2 at 240 GHz. While the metal plate redirects the interrogation in its specular direction and therefore only shows high RCS for normal incidence, the trihedral corner reflector acts as a retroreflector for ±45◦ regarding its maximum RCS orientation. However, its RCS varies according to the geometrical cross-section from each incidence angle, as shown in Chap. 5. RCS characterization The RCS of a tag can be calculated by solving Eq. 2.4 for σ: σ=

Prx (4π)3 d 4 . Ptx G tx G rx λ2

(2.6)

When the measurement is performed by a calibrated VNA, the received and transmit powers can be substituted by the scattering parameter |S11 |2 = Prx /Ptx . The accuracy of the measurement is determined by (i) the correct estimation of the tx and rx reader gains and the distance to the tag, (ii) the correct estimation of the tx and rx reader powers, or the proper calibration of the VNA, and (iii) how accurate the radar equation describes the channel where the measurement is performed, i.e., to avoid any additional reflections in the surroundings. The characterization of the tags with this method tends to become more accurate at higher frequencies and longer distances. At higher frequencies, smaller time resolutions due to wider absolute bandwidths permit a better isolation of the tag’s response from other reflections

14

2 High-Q Resonators for Chipless RFID and Sensing

in the time domain, so that the radar equation becomes a more accurate model of the power budget. Furthermore, the longer the distance, the lesser the effect of small absolute errors in the estimation of the distance in the calculated σ(dBm2 ). An alternative method to estimate the RCS is to compare the received power between the tag under test, denoted with the subscript t, and a reference target whose RCS is known, denoted with the subscript ref. By using Eq. 2.6 for both targets and dividing their RCSs σt /σref , the following relationship is obtained: σt =

|S11,t |2 Prx σref = σref . Prx,ref |S11,ref |2

(2.7)

This method assumes that no variations between the measurement setup of the reference target and the tag under test occur. The reference target must be a target whose RCS, σref , is known or can be analytically calculated, since it directly influences the accuracy of the measurement. A metallic sphere has the advantage of showing an RCS independent of the orientation, reducing potential inaccuracies. Furthermore, due to the decay of the received power with the fourth power of the distance, it must be assured that the reference target and the tag under test are measured at the same distance from the reader. Similarly, as with the previous method, the longer the distance to the reader, the smaller the effect of small variations in the calculated σ(dBm2 ), so that increasing the distance can be an effective method to improve the accuracy. However, for long distances, the low reflected power from metal spheres might not be sufficient to be clearly distinguished in the reader. In this case, trihedral corner reflectors have the advantages of high RCS and a more stable RCS than flat metal plates against small misalignments.

2.1.2 Maximum Range To calculate the maximum distance for a certain reader and tag, we can define the difference between the transmitted power and the received power without a tag as △P = Ptx /Prx,no−tag . The minimum value for Prx,no−tag is the noise floor or reader sensitivity, which in most channels is limited by interference or clutter. Depending on the information to be detected by the reader, a minimum Signal-to-Interferenceplus-Noise Ratio (SINR), γmin = Prx,tag /Prx,no−tag is needed for successful detection. Substituting these parameters in Eq. 2.4 gives: γmin G 2tx σλ2 . = 4 △P (4π)3 dmax

(2.8)

The maximum achievable readout distance dmax can therefore be calculated as / dmax =

4

△P G 2tx σλ2 . γmin (4π)3

(2.9)

2.2 High-Q Resonators

15

(a)

(b)

Fig. 2.4 Two examples of achievable maximum ranges according to the radar equation. a Maximum range over operation frequency for G tx = 21 dBi, △P/γmin = 70 dB, b f = 240 GHz, σ = −20 dBm2 (solid), and −10 dBm2 (dashed)

△P · G tx depends mainly on the reader hardware, while γmin depends on the reader’s signal processing capabilities and coding method. For simplicity, this work models any range reduction due to interference as a reduction of △P and, with it, of the maximum range, dmax . Therefore, it is not distinguished whether such reduction is originated by noise or by unfiltered clutter reflections. Two examples of the effect of operating frequency and △P/γmin over the maximum range, dmax , as described by Eq. 2.9 are depicted in Fig. 2.4. Figure 2.4a shows that an increase in operation frequency reduces the maximum range, all other parameters staying equal. Despite this seemingly negative consequence, higher RCSs are feasible at higher frequencies by using retroreflective methods and help compensate this effect, as is investigated in Chap. 5. Furthermore, miniaturization helps achieve higher available reader gains in compact antenna structures, which in turn increase the maximum range as depicted in Fig. 2.4b. These examples show that, even with conservative reader gains and RCS, ranges over a meter at 240 GHz can be expected. The main impediment towards the widely spread use of passive radar targets in harsh environments is therefore attributed not to the low reflected power, but mainly due to their low reliability in dynamic environments, where strong clutter reflections may intermittently or permanently reduce △P, limiting the system’s reliability. This work shows that the use of high-Q resonators helps overcome this limitation in reflective environments by (i) increasing the tag’s RCS, σ, and (ii) providing means to suppress clutter and improve the △P by using high-Q resonators.

2.2 High-Q Resonators The Q-factor of a resonator is defined as Q = 2π

maximum energy storage during a cycle . average energy dissipated per cycle

(2.10)

16

2 High-Q Resonators for Chipless RFID and Sensing

As the quality factor increases, it shows an enhanced frequency selectivity of both its magnitude and phase around its resonance frequency. The following relations apply to any resonant cavity: f res , (2.11) Q= △ f −3dB Q=

ωres , 2α

(2.12)

Q=

ωres τ , 2

(2.13)

where f res is the resonance frequency of the resonator, △ f is the 3-dB bandwidth of the resonance, ωres = 2π f res is the angular resonance frequency, α is the attenuation rate, and τ is the time constant. The power decay can be described by the equation P(t) = P0 e−2·t/τ ,

(2.14)

i.e., e−2 = –8.69 dB loss every time constant, τ . The narrow-band characteristic of high-Q resonators is observed in Eq. 2.11 as an increase in the Q-factor with lower 3-dB bandwidth of the resonance peak. The equivalence of this property in the time domain is described by Eq. 2.13 as a higher τ , i.e., a slower power decay. Considering the power loss in the resonator as in Eq. 2.10, the loaded quality factor, Q l can be defined to include all loss mechanisms, and can be separated into more specific quality factors according to the contributions of different loss mechanisms, for example 1 1 1 = + , (2.15) Ql Qe Qu where the external quality factor, Q e , accounts for the resonator loss due to coupling with an external circuit that is necessary to excite (and read out) the resonator, and the unloaded quality factor, Q u , accounts for any losses that do not contribute to an increase in the received signal at the reader. We can analyze the reflection and transmission of a single resonator excited by a plane wave. The reflection and transmission coefficients describing the magnitude and phase contribution can be described by analyzing the behavior of a 2port network. Such a model is presented in [1] based on the coupled-mode-theory, which assumes linearity and time-invariance. Furthermore, weak-mode coupling is assumed, which applies as long as different resonance modes operate at sufficiently spaced resonance frequencies, so that the effect of each mode can be studied separately. In [1], the model is derived for the reflection-only case of a periodic array of resonators over a ground plane (no transmission). However, it can be easily applied to the case without a ground plane, as it is derived in the supplemental materials of this same publication. The reflection and transmission of a resonator can be obtained as:

2.2 High-Q Resonators

17

┏(w) = −

1/τe , − j · (w − wres ) + 1/τu + 1/τe

T (w) = 1 −

1/τe . − j · (w − wres ) + 1/τu + 1/τe

(2.16)

(2.17)

┏ and T are the reflection and transmission coefficient, respectively. τe and τu are the time constants that can be calculated from the external and unloaded quality factors using Eq. 2.13. Adapting the equations to use the quality factors, we obtain ┏(w) = −

−2 j ·

T (w) = 1 −

( wwres

1/Q e , − 1) + 1/Q u + 1/Q e

1/Q e . −2 j · ( wwres − 1) + 1/Q u + 1/Q e

(2.18)

(2.19)

The magnitude and phase of the reflection and transmission coefficients around the resonance frequency for different external and unloaded quality factors are shown in Fig. 2.5.

(a)

(b)

Fig. 2.5 Magnitude (top) and phase (bottom) of the a reflection, ┏(w), and b transmission, T(w), coefficients of a resonating element. Solid lines correspond to the lossless case, Q u → ∞, while dotted lines assume Q u = 500

18

2 High-Q Resonators for Chipless RFID and Sensing

Notice that the model applies to a steady state with an incident interrogation signal and no structural reflection from the resonator or the supporting structure, so that the reflection coefficient, ┏, represents the re-radiated power from the resonators and tends to 0 outside the resonance frequency. At resonance, maximum reflection is achieved. In the case of the transmission coefficient, T , it is 1 outside the resonance frequency, showing full transmission, i.e., the incident signal propagates through free space without interaction with the resonator. At resonance, the resonator has received the incident wave and constantly retransmits it with opposite phase, so that ┏ = −1, T = 0. Two important properties of high-Q resonators should be highlighted: First, the ringing time is directly proportional to the Q-factor, so that higher Q-factors are needed for a longer ringing. Second, for the same Q-factor, the time constant is reduced as the resonance frequency increases. This is an important property to consider when using time gating for the wireless readout of mm-Wave and THz high-Q resonators. Since the achievable Q-factors are similar or even lower than the ones reached in the 1–10 GHz frequency range, the time constants are often up to 3 orders of magnitude lower. Fortunately, clutter spread also tends to decrease at higher frequencies due to higher absolute bandwidths, leading to higher time resolution and an enhanced capability to separate multipath components in time-domain. The question arises as to what is understood under high-Q. In the literature, the terms high-Q, extremely high-Q, and ultra-high-Q are used in thousands of publications with quality factors ranging from below 100 in some optimized structures with lossy materials in microwave frequencies, to above a million in optical frequencies. In this work, it is only distinguished between low-Q and high-Q resonators, and the distinction is established according to their potential to be differentiated from clutter due to its slow power decay. To make the criteria independent of the channel and the operation frequency, high-Q is not defined in terms of channel properties as the clutter spread, but related to a maximum 3 dB loss of energy in the resonator for every pulse width of the interrogation pulse. The considered relative bandwidth is Br el =10 %, and the pulse width is calculated as △T = 1/B. From Eq. 2.14, it can be obtained via cross multiplication with the loss per τ that the pulse width must be equal to 3/8.69 · τ . Rearranging this expression in Eq. 2.13 gives the following threshold High-Q if Q l >

ωres 2π f res = 91. = 2 · 3/8.69 · B 2 · 3/8.69 · 0.1 f res

(2.20)

Therefore, according to the criteria that the power decay of the resonator must be slower than 3 dB per pulse width of a reader with a 10 % relative bandwidth, a resonator is considered high-Q if its loaded quality factor, Q l > 91, and low-Q if it is lower.

2.2 High-Q Resonators

19

2.2.1 Estimating Material Losses It is useful to differentiate the losses from coupling, Q coup , radiation, Q rad , dielectric, Q diel , and conductor, Q cond . They can be related by extending Eq. 2.15 as: 1 1 1 1 1 = + + + . Ql Q rad Q coup Q diel Q cond

(2.21)

It should be noted that, in the same manner as lower loss materials show higher Q diel and Q cond , also weaker coupling and less radiation are modeled as higher Q coup and Q rad , respectively. Depending on the coupling mechanism, the external quality factor corresponds to either the radiation or the coupling quality factors. For a parallel feed, where the resonator is excited from a plane wave and backscatters the signal in the form of radiation, Q e = Q rad . For a corporate feed, in which the resonator is excited through a waveguide, any radiation is considered a loss and contributes to a lower unloaded quality factor, while the external quality factor Q e = Q coup . In different parts of this work, one of the quality factors is not known, while the others can either be simulated, measured, or obtained from the datasheet of the material. When the losses of the material are unknown, either because a novel material is used, or due to novel fabrication methods that may use mixtures of materials as with the ceramic 3D printing process introduced in Sect. 4.2, high-Q resonators can be used to estimate the material losses. For this, it is useful to group the radiation and coupling quality factor in a single radiation-coupling quality factor, Q rc as 1 1 1 = + Q rc Q coup Q rad

(2.22)

since Q rc can be calculated as the loaded quality factor in lossless simulations where Q diel = Q cond = ∞. In the case of a parallel feed, Q coup = ∞, the loaded quality factor from a lossless simulation equals Q rad . In the case of tags with a corporate feed, the loaded quality factor calculated through a lossless simulation contains the effects of coupling and radiation, Q l = Q rc . By simulating the same cavity with one and two input ports and assuming the radiation quality factor does not vary due to the additional port 1 1 1 = 2−port − , Q coup Q rc Q rc 2−port

(2.23)

is the simulated two-port lossless quality factor. Once Q coup is known, where Q rc Q rad can be obtained from Eq. 2.22. There are other related methods to obtain the radiation quality factor from the maximum magnitude of the transmission S21 parameter of the 2-port simulation. However, the reliability of this method decays when

20

2 High-Q Resonators for Chipless RFID and Sensing

the feeding network of the resonator is dielectric and suffers from radiation losses, as is the case in some resonators of this work. For this reason, Eq. 2.23 is used for the estimation of Q coup . Finally, Eq. 2.21 can be used to calculate Q diel or Q cond , as long as the structure is solely metallic or dielectric and the other quality factor tends to infinite. All these calculations assumed no additional losses due to atmospheric absorption. Especially in communication systems, it is important to account for an increase in the atmospheric absorption for higher operation frequencies. For the frequency ranges included in this work up to 300 GHz, a local maximum of αair = 50 dB km−1 at 230 GHz is observed [2]. Since all presented resonators are either surrounded or filled with air, they are also affected by this increase in losses. Applying Eq. 2.12, the air quality factor, Q air , is greater than 1 × 107 for αair = 50 dB km−1 . This is more than 3 orders of magnitude higher than the quality factors achieved in this work, so its contribution is neglected in the calculations, i.e., air is considered as a lossless material with εr = 1.

2.2.2 RCS of a Resonating Tag The RCS of a tag with one or several resonators can be studied by analyzing each of the terms in Eq. 2.2. In the following, the origin and ranges of each factor are further explained. Tag directivity The effective aperture and directivity are influenced by the feed mechanism. For parallel feeds, this is mainly determined by the radiation pattern associated with the resonance mode. However, the selection of a certain resonance mode is constrained by the suitable shapes, surrounding materials, directions of the main lobe, and manufacturing methods. Furthermore, higher order modes show lower HPBWs, but a higher number of grating lobes, degrading any potential directivity increase. For these reasons, the directivity of the resonator is usually not considered when selecting the resonance mode. Still, moderate directivities around 3 dB can be obtained for single DRs. In contrast, a corporate feed approach uses an antenna to receive and transmit, adding more flexibility and potential RCS to the tag design. However, there are two main limitations to the increase in directivity: (i) higher directivities need larger antennas, and (ii) higher directivities reduce the HPBW, so that a better alignment between tag and reader is needed. (i) is usually a limitation due to tag size, but it can be also limited due to fabrication complexity. In the case of (ii), the HPBW can be estimated as 4πηD △φ3dB △θ3dB ≈ (2.24) , D

2.2 High-Q Resonators

21

where △φ3dB and △θ3dB are the HPBWs in azimuth and elevation, respectively, and D the directivity of the antenna. This approximation is valid assuming a single main beam, and an aperture efficiency ηD ≈ 0.8 can be used. Assuming △φ3dB = △θ3dB , a directivity of 5 dBi, 10 dBi, and 15 dBi gives HPBWs equal to 102◦ , 57◦ , and 32◦ , respectively. This means that increasing the tag’s RCS by increasing the directivity is limited by the coverage angles from where the tag should be readable in dynamic environments. In addition, for constant directivities, Eq. 2.5 shows that the RCS decreases with the square of the wavelength λ2 ∝ f −2 . This translates into a tendency towards either lower maximum ranges, lower coverage angles, or higher reader gains at THz frequencies. However, this tendency can be overcome by using retroreflective approaches for the realization of the tag, as it is studied in Chap. 5. An increase of the retroreflective tag’s RCS with λ−2 ∝ f 2 is achieved and helps to maintain a constant maximum range dmax for the same reader antenna gains and tag dimensions, as shown in Fig. 2.6. Interestingly, achieving a constant dmax independent of frequency with a constant reader gain has an additional benefit. Since the gain of the reader antennas is proportional to its electrical size, their dimensions become smaller with frequency and can fit into smaller devices such as mobile robots or drones. Using Eq. 2.3 and assuming a 50 % aperture efficiency, A = 0.5 · Ageom , a 21 dBi gain at 80 and 240 GHz can be achieved with a squared area of side length equal to 8.39 and 2.8 mm, respectively.

Fig. 2.6 Operation frequency over range of different tag types for constant △P/γmin = 70 dB, G tx = 21 dBi. The RCS in all solid curves is –10 dBm2 at 80 GHz, and then varies differently with frequency. The blue line resembles the behavior of a parallel and corporate feed, where the dimensions of the tag are proportional to the wavelength. The orange line stands for constant σ at all frequencies, and the green line corresponds to a retroreflector such as a trihedral corner reflector with constant dimensions, exhibiting an increasing RCS with frequency. The dashed lines show the improvement achieved by a 10 dB RCS increase at each frequency

22

2 High-Q Resonators for Chipless RFID and Sensing

Fig. 2.7 Resonator efficiency, ηQ , over the unloaded to external Q-factor ratio Q u /Q e as described by Eq. 2.25

Tag efficiency It is useful to analyze the effect of the resonator losses in the final RCS of the tag. We can define the resonator efficiency ηQ as ⎛ ηQ =

Qu Qe + Qu

⎞2 .

(2.25)

Notice that the resonator efficiency does not depend on the absolute values of Q e and Q u , but on their ratio as depicted in Fig. 2.7. It is therefore essential to have low-loss materials to (i) be able to fabricate resonators with high loaded Q-factor as shown in Eq. 2.15, and (ii) have both, a high Q l and a high resonator efficiency ηQ , which maximize the range of the chipless wireless high-Q tags according to Eq. 2.9. In the case of a parallel feed, ηt = ηQ . For a corporate feed, the propagation losses through the components integrating the feed network, usually an antenna and a waveguide, grouped as ηcorporate , need to be considered both for reception and retransmission, so that 2 · ηQ . (2.26) ηt = ηcorporate

2.3 Clutter Suppression Methods Due to the strong limitation imposed by clutter towards the widespread use of chipless wireless tags, multiple methods have been developed to reduce their effect in numerous situations. Table 2.2 shows an overview of clutter suppression methods compatible with chipless tags. Empty-room measurement Empty-room measurement is also found in literature as differential measurement and channel subtraction. In many publications, it is mentioned as a first measurement

2.3 Clutter Suppression Methods Table 2.2 Clutter suppression methods Operation in dynamic Method environments Empty-room measurement

No

Cross-polarization

Yes

Harmonic generation

Yes

Out-of-band channel estimation

Yes

Time-gating

Yes

23

Comment Less accurate the longer the distance and the higher the frequency due to decreased coherence time of the channel Assumes weak cross-polarized clutter limited improvement according to [3] Needs a non-linear material and enough power only feasible with diodes in the tag Works better combined with other methods approximate peak position needs to be known Tag’s response must outlast clutter

without the tag to remove the influence of the environment, including the measurement setup. This method is very useful to improve the accuracy of the characterization measurements on the tag, since it removes most clutter other than the generated from the tag’s structure, which receives the name of structural mode to be differentiated from the antenna mode containing the information encoded in the tag. Despite its utility for characterization and its widespread use in most chipless RFID literature, the main limitation is that the reader cannot change its position between the empty-room measurement and the readout of the tag’s response. Furthermore, the environment needs to stay substantially static in terms of wavelength, what becomes less likely at higher frequencies and longer distances between the tag and the reader. For these reasons, in this work, empty-room measurements are used to assist in the characterization of new tags, but they are neither used nor foreseen in the cluttered measurements performed, to imitate the conditions present in a dynamic environment. Cross-polarized tags Cross-polarized tags encode the backscattered response in an orthogonal polarization. The premise behind is that clutter is mainly co-polarized, so that the SINR can be improved if the tag has the property of backscattering its signature in the orthogonal polarization, mostly linear. Despite its utility, it requires the ability to send and receive with orthogonal polarizations in the reader and constrains the possible orientations between the reader and the tag. In a recent study, a considerably worse performance has been demonstrated for cross-polarized resonant tags operating in the frequency range from 2 GHz to 4 GHz (below 20% readout success in the studied scenarios) when compared to resonant-based readout methods (higher than 80%) [3].

24

2 High-Q Resonators for Chipless RFID and Sensing

2.3.1 Harmonic Generation Harmonic generation consists on creating a backscattered response with frequencies not contained in the interrogation signal to avoid interrogation-induced clutter in the system, as shown in [4]. However, an efficient generation of harmonics with the power levels available at the tag is not possible without the use of diodes. These tags are therefore named as quasi-chipless and included in the comparison for completeness, since they only need a single diode to operate and can withstand higher temperatures than other chipped tags.

2.3.2 Out-of-Band Channel Equalization This method was proposed in [5] and exploits the narrowband characteristics of highQ resonators. If the position of the resonance peak or notch can be approximately estimated, the channel response before and after that frequency region can be used to equalize or smooth the expected channel response at the resonance frequency of the sensor. As shown in [5], its potential is higher to improve the signal quality when the peak position is already known, so it can be best used in combination with other methods to increase the accuracy of the estimated resonance frequency and, therefore, improve the system’s performance.

2.3.3 Time-Gating Time-gating is a standard technique in radar and other applications to separate the desired signal or selectively remove undesired ones. It can be seen as a multiplication of the time-domain signal with one or several windows of certain shapes and length. By isolating the tag’s response, clutter is reduced and more robust readouts are achieved. Figure 2.8 shows the operating principle of time-gating on the backscattered response from high-Q resonators. Due to the time-frequency relationship, if the time-window span is too short, reductions in the resolution of the frequency-domain signal would become apparent. Throughout this work, a rectangular or boxcar window is used to get the highest resolution in the frequency domain for a given window span, and long window spans are favored to get accurate values of the measured loaded quality factors.

2.4 Identification and Sensing with High-Q Resonators

25

Fig. 2.8 Time-gating method applied to the readout of the resonance frequency of a high-Q resonator [6]

2.4 Identification and Sensing with High-Q Resonators This section describes the modulation mechanisms used in this work for identification and sensing with high-Q resonators in cluttered environments. Afterwards, the principles of hybrid modulations, ranging accuracy, and sensing accuracy are introduced.

2.4.1 Identification and Sensing To successfully distinguish different tags, each of them must have a distinct backscattered signal. In the case of tags with a single high-Q resonator, this is achieved by designing the resonator of each tag to resonate at a different resonance frequency. By integrating several resonators inside a tag, the number of distinct tags can be further increased. In the following, a model is presented to calculate the coding capacity of a system based on tags with one or several high-Q resonators. Coding capacity of high-Q resonator tags The coding capacity, C, in bits of a chipless tag is determined by the number of distinct backscatter responses or signatures, s, that can be generated and is described by the formula: (2.27) C = log2 s. A technique for coding multi-resonator tags is on-off coding, where the information is encoded by the presence or absence of a resonance peak at a certain frequency slot. If the number of resonators, n, is equal to the number of frequency slots, m,

26

2 High-Q Resonators for Chipless RFID and Sensing

Fig. 2.9 Modulation schemes used in this work for a identification, and b sensing with high-Q resonators

then the number of signatures is 2n . However, the number of resonators in a tag is often limited by size as well as efficient feeding techniques. In this case, the number of resonators, n, can be limited to a lower value than the total frequency slots, m, available, so that the number of distinguishable signatures is s=

n ⎛ ⎞ ⎲ m i=0

i

=

n ⎲ i=0

m! , i!(m − i )!

(2.28)

where the summation represents the possibility of less than n resonators being used for coding. This case is depicted in Fig. 2.9a. Since an error-free decoding can be seldom guaranteed, it is advantageous to fix the number of resonators per tag, so that the reader knows in advance how many resonance peaks or resonance notches form the signature of any tag. In this case, the number of distinct cases is reduced to s=

⎛ ⎞ m m! = . n n!(m − n)!

(2.29)

This modulation technique is appropriate for high-Q resonating tags, where the narrow resonance peaks allow for the allocation of multiple frequency slots, m, in a certain frequency bandwidth. The number of frequency slots, m, can be calculated from the available absolute bandwidth, B, and the minimum bandwidth needed for each resonator, △ f , as I m=

B △f

I

I =

B △ f 3dB · K f

I ,

(2.30)

where △ f = △ f 3dB · K f depends on the 3 dB bandwidth of the resonator as well as a margin K f to account for frequency shifts due to manufacturing tolerances and other inaccuracies. Using the Q-factor and the relative system bandwidth, Br , Eq. 2.30 becomes I I Q m = Br (2.31) = ⌊Br Q eff ⌋ . Kf

2.4 Identification and Sensing with High-Q Resonators

27

Fig. 2.10 Number of signatures that can be encoded by positioning a reduced number of resonators, n, in a larger number of frequency slots, m. Depending on the bandwidth needed per resonator, Q eff , the relative bandwidth needed can be calculated with Eq. 2.31

It can be observed that manufacturing tolerances generate an uncertainty in the exact resonance frequency of the resonator. Hence, it affects the maximum number of states similarly as a lower Q-factor. For the analysis of the maximum number of cases for different quality factors, we can therefore define an effective quality factor Q eff = Q/K f . While the maximum number of frequency slots, m, is usually limited by the available bandwidth, the number of resonators is often determined by the techniques used for the realization and excitation of the resonators as well as the available space. Figure 2.2 shows the number of signatures, s, that can be encoded depending on the number of resonators and frequency slots available, as well as the relative bandwidth needed assuming Q eff = 500. While Q eff = 500 can be considered an optimistic value for the minimum separation between resonance peaks, more conservative values can be directly extracted from Fig. 2.10 by calculating the number of frequency slots, m. For example, Q eff = 250 means that half the frequency slots are available for the same bandwidth, and five times lower if Q eff = 100. Figure 2.10 can therefore be used as a base for the number of distinct signatures possible among different tag concepts. An overview of the bit capacities achieved with the identification tags presented in this work is included in Table 2.3.

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2 High-Q Resonators for Chipless RFID and Sensing

Table 2.3 Coding capacities achievable for a 10% bandwidth with the tag technologies introduced in this work Tag type

Frequency

Time-Domain Reflectometry

Microwave

Bed of Nails

Low-mmW

n

Q ∗eff

m

s

C (bits) 512

4

200

20

4845

9 12.24

RT6010 Photonic Crystal

Mid-/high-mmW

2

72

7

21

4.39

RT6010 Photonic Crystal

Mid-/high-mmW

3

72

7

35

5.13

Al2 O3 Photonic Crystal

Mid-/high-mmW

2

121

14

66

6.04

Al2 O3 Photonic Crystal

Mid-/high-mmW

3

121

14

220

7.78

HR-Si Photonic Crystal

Mid-/high-mmW

2

174

19

136

7.09

696

9.41

7

2.81

HR-Si Photonic Crystal

Mid-/high-mmW

3

174

19

Frequency Selective Surface

Mid-/high-mmW

1

70

7

mmW: mm-Wave; * At room temperature

Sensing The ability to wirelessly detect the resonance frequency of a high-Q resonator and the sensitivity to different physical magnitudes allows for its use as a sensor as shown in Fig. 2.9b. In the case of sensing, a discretization of the frequency band into frequency slots is not needed, and a continuous shift of the resonance peak as a result of the variation of a measurand, A, can be used. In the examples introduced in this work, the measurands are temperature or pressure, and the variation of the resonance frequency of the resonator with the measurand is primarily linear. Therefore, the resonance frequency is related to A as f res (A) = f res (A0 ) · (1 + K A · (A − A0 )),

(2.32)

where K A is the sensitivity normalized at f res (A0 ), f res is the resonance frequency, and A0 is a reference value at which the measurand and the resonance frequency of the resonator are known. The measurand, A, can then be calculated as A = A0 +

f res (A) − f res (A0 ) . K A · f res (A0 )

(2.33)

2.4.2 Hybrid Modulation, Ranging Accuracy, and Sensing Accuracy A hybrid modulation is achieved when two or more modulation techniques are implemented in a tag. In this work, two examples of a hybrid modulation are presented in which one of the modulations is achieved via high-Q resonators. In Sect. 3.2, a high-Q resonator for sensing is integrated in a wideband phasecoded Time-Domain Reflectometry (TDR) identification tag. While the identification

2.4 Identification and Sensing with High-Q Resonators

29

is achieved via multiple phase-coded reflections of a wideband pulse, the sensed temperature can be distinguished as a resonance peak originated by the high-Q resonator in the frequency domain. Section 5.2 presents the integration of high-Q resonators in a Lüneburg lens. While identification or sensing can be encoded at the resonance frequency of the high-Q resonators, the wideband reflection of the lens outside these resonance frequencies can be used for ranging. In both cases, it is demonstrated that high-Q resonators can be integrated into existing wideband tags and retroreflectors, operating simultaneously, with a single interrogation pulse, and within the same frequency band. During the design of high-Q resonators for identification and sensing, the appearance of more than one resonance peak or resonance notch per resonator in the bandwidth allocated needs to be avoided. This is described by the Free Spectral Range (FSR), which is defined as the frequency bandwidth where it can be guaranteed that the reader will receive a single resonance peak or notch per resonator. If the FSR is lower than the intended bandwidth for the tag, the number of distinct signatures for identification and the range of measurands that can be distinctly sensed with certainty is reduced, limiting the performance. Low order resonance modes are preferred due to their lower mode densities, which therefore allow for larger FSRs. This needs to be considered when selecting the appropriate resonance mode for each application. Ranging and sensing accuracy The range resolution △d of a radar, i.e., the minimum distance between two objects that can be resolved by the reader, is determined by the bandwidth B in the equation △d =

c0 , 2B

(2.34)

△d is therefore equal to half the pulse or the half-power pulse width. As long as there are no multipath signals that overlap in the reader with time-separations below △d, the accuracy of the radar to resolve the maximum value of the pulse can be described statistically by the Cramér-Rao Lower Bound (CRLB) [7] as / σr (B, E s , No ) ≥ △d

1 2 π E s /N0

⎛ 1+

⎞ 1 , E s /N0

(2.35)

where E s is the average pulse energy, and N0 is the noise power density. Since the range resolution and accuracy are inversely proportional to the absolute bandwidth, an effective step towards improved resolutions and accuracies is the increase in the operation frequency of the radar. A similar relationship could be established for the readout of resonance peaks in the frequency domain, being its variance proportional to 1/Q due to the reduction of the width of the resonance peak width. This work, therefore, improves the accuracy by investigating the design, fabrication, and characterization of different tags that (i)

30

2 High-Q Resonators for Chipless RFID and Sensing

maximize the signal, (ii) minimize the interference, and (iii) reduce the resonance peak width for sensing and identification. Furthermore, the wideband reflections of the tags presented in Sect. 3.2 and Chap. 5 achieve narrow reflected pulse widths that can be used for time-domain identification and ranging. Neither an optimization of the readout method nor a detailed derivation of the CRLB are pursued in this work, which focuses on the design, fabrication, and characterization of different tag concepts based on high-Q resonators.

References 1. Qu C, Ma S, Hao J, Qiu M, Li X, Xiao S, Miao Z, Dai N, He Q, Sun S (2015) Tailor the functionalities of metasurfaces based on a complete phase diagram. Phys Rev Lett 115(23):235503 2. Sun J, Hu F, Lucyszyn S (2016) Predicting atmospheric attenuation under pristine conditions between 0.1 and 100 THz. IEEE Access 3. Lin J-A, Jhang J-Y, Lai F-P, Lin B-L, Jhang Y-M, Chen Y-S (2020) Analysis of calibration-free detection techniques for frequency-coded chipless RFID. IEEE Trans Antennas Propag 4. Kubina B, Mandel C, Schüßler M, Jakoby R (2015) Compact quasi-chipless harmonic radar sensor with a dielectric res-onator antenna. In: IEEE MTT-S international microwave symposium. IEEE, pp 1–3 5. Kubina B, Mandel C, Schüßler M, Jakoby R (2014) Dynamic interference suppression for chipless wireless sensors based on an out-of-band channel estimation method. Int J Microwave Wirel Technol 6(3–4):353–360 6. Jiménez-Sáez A, Alhaj Abbas A, Schüßler M, Abuelhaija A, El-Absi M, Sakaki M, Samfaß L, Benson N, Homann M, Jakoby R, Kaiser T, Solbach K (2020) Frequency-coded mm-wave tags for self-localization system using dielectric resonators. J Infrared, Millim, Terahertz Waves 41(8):908–925 7. Miesen R, Ebelt R, Kirsch F, Schäfer T, Li G, Wang H, Vossiek M (2011) Where is the tag? IEEE Microw Mag 12(7):S49–S63

Chapter 3

Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves

In this chapter, the potential of single high-Q resonators for the wireless readout of sensed values in highly cluttered environments is demonstrated with a temperature sensor for a machine tool in Sect. 3.1.1. In addition, the possibility to control variations in the resonator’s near field to sense other physical parameters is explored in Sect. 3.1.2. After demonstrating the potential of the concept for the realization of chipless wireless sensors robust against clutter, the use of the high-Q concept towards hybrid modulation schemes is investigated in Sect. 3.2. Finally, this chapter concludes studying the limitations of the single air-cladded high-Q resonator concept for the design of mm-Wave and THz chipless RFID and sensors.

3.1 Sensing with a Single Air-Cladded High-Q Resonator This section presents application-specific adaptations of the high-Q resonator concept. By air-cladded, it is described the fact that the main boundary around the resonator is air, so that it needs to be mounted on a carrier material, usually a metal. Similarly, as the previous work in [1], the resonators are realized with high permittivity, εr > 30, low loss, tan δ < 0.001 ceramic materials commercially available in the microwave frequency range. The exact material is specified on the description of each sensor.

3.1.1 Temperature Sensor for Machine Tools The fabrication and processing of high-end workpieces, such as aerospace and automobile parts, poses strict prerequisites on the tools and machinery. The prediction of © The Author(s), under exclusive license to Springer Nature Switzerland AG 2022 A. Jiménez-Sáez, Towards THz Chipless High-Q Cooperative Radar Targets for Identification, Sensing, and Ranging, Springer Theses, https://doi.org/10.1007/978-3-031-04976-7_3

31

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3 Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves

the tool state is currently accompanied of a considerable uncertainty, so that tools are exchanged much earlier than their expected mean life to prevent damage that could be critical for the workpiece. Despite the improvements in automatization that speed up the process, the time needed to change tools has an important weight on the final processing costs, as well as the need for additional tools. Intensive research concentrates on developing and implementing methods to improve the monitoring of tool wear and condition, which are summarized in [2]. Furthermore, recent algorithms rely on machine learning for improved predictions [3]. Despite improved algorithms, the amount and quality of the raw data from the tool operation is key for more accurate and reliable predictions. A common sign of tool failure is a sudden unexpected increase in temperature at the tip of a tool, which can be caused by additional friction with the workpiece and the chirps. For this reason, temperature measurements close to the tip can be used to predict and detect tool failure. Due to the harsh conditions inside the machine, the use of infrared thermometers is handicapped by the reflections of other temperature sources in the highly metallic environment, as well as limited line-of-sight that can be blocked by the workpiece, the chirps, or the applied cutting oil. For these reasons and due to the high-temperatures that can be achieved during operation, the availability of a wireless chipless sensor integrated in the tool that provides live temperature information would be advantageous. In the following, a proof of concept of the use of a ring-shaped DR for temperature monitoring near the tip of the tool in milling machines is presented based on [4, 5]. The sensor is designed to work in the Industrial, Scientific and Medical (ISM) band from 2.4 GHz to 2.5 GHz, so that up to 36 dBm Effective Isotropic Radiated Power (EIRP), EIRP = Pt G t is allowed without additional licensing [6]. The resonator is designed as a ring to be mounted on the tool, while its rotation symmetry keeps a centripetal force equilibrium and allows its use at the high angular velocities above 10000 rpm expected in a machine tool. Furthermore, a rotational symmetry means that the resonance mode is always oriented towards the interrogation signal transmitted by the reader antenna, therefore achieving a constant monostatic RCS independent of the interrogation angle and its radiation diagram. Finally, the ring resonator can be easily mounted and fixed to the tool with a small inset, and the contact surface permits the necessary transmission of heat from the tool to the sensor. As in [7], the ceramic material K-50 from TCI ceramics is used, which has a high relative permittivity εr = 50 and a temperature sensitivity of the dielectric constant K T = −700 ppm/◦C [8]. A high-Q resonance mode is sought via CST Studio Suite time-domain simulations. Since the sensor encodes the temperature on its resonance frequency, additional high-Q resonances could be interpreted by the reader as different sensed temperatures and must be avoided. Therefore, only low order resonance modes are considered to obtain a higher FSR without resonance peaks from other high-Q resonances. In addition to a tendency towards lower mode densities, higher-order modes would result in larger structures, i.e., bulkier ceramic resonators for a certain operation frequency, which cannot be integrated into the tool and react slower to temperature changes. For the interrogation, a linear polarization with an E-field perpendicular to the tool’s

3.1 Sensing with a Single Air-Cladded High-Q Resonator

33

steel mount

copper mounts (a)

2017 IEEE

(b)

ceramic

εr = 50

metal

10 mm

30 mm

15 mm

(c)

(d)

Fig. 3.1 a Simulated RCS and Absorption Cross Section (ACS) of the designed resonator showing different resonance modes. b Copper and steel mounts, c E-field of the selected resonance mode, EH211 , and d effect of the different insets on the resonator [4, 5]

cylinder is preferred to reduce the tool’s reflection. 2D projections of the field distribution in the ring are sketched in Fig. 3.1. Following the notation described in [9], the selected resonance mode is the EH211 . One of the four maxima of the resonance mode is always oriented towards the excitation, i.e., the reader antenna. However, it needs to be considered whether the high angular velocities of the tool may redirect this maximum towards different directions, reducing the RCS of the DR. Assuming a high angular velocity of 105 rpm and a conservatively long measurement time span of 1000 ns, a deviation of only 0.6◦ is predicted. This is more than an order of magnitude lower than typical HPBWs of low-order modes, so that no noteworthy effect in the received power is expected. In addition, the movement of the resonator is perpendicular to the communication link, minimizing Doppler effect. Nevertheless, calculating the Doppler shift with the same angular velocity of 105 rpm in the worst case of perpendicular movement towards the reader, a shift of ±520 Hz is predicted, around three orders of magnitude below

34

3 Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves

the 3-dB bandwidth of a resonator with loaded quality factor Q l =1000. Such small frequency shifts should therefore not affect the sensor’s operation. The effect of different insets in the tool to fix the position of the DR are shown in Fig. 3.1d. Increasing the amount of metal around the resonator with longer insets varies the mode distribution seen in the bottom of Fig. 3.1c, increasing the field proportion out of the dielectric and increasing the resonance frequency. However, for tool diameters from 10 mm to 15 mm, which originate insets from 0 mm to 2.5 mm, no strong perturbations of the magnitude or width of the resonance peak are observed for the selected EH211 resonance mode. This is probably due to the selected polarization parallel to the insets, i.e., the E-field components at the insets tend to zero. The designed sensor is fabricated with dummy tools emulating a real tool. The dummy tools are manufactured with copper for a low-loss reference, and with steel as a realistic tool material with higher losses. By studying both prototypes, further insights are obtained on the effect of metal losses on the performance of the resonator. One of the most sensitive parts to manufacturing inaccuracies is the dielectricmetal interface, where the resonance mode presents a maximum perpendicular Efield. Any air gap between the dielectric and the metal generates a perpendicular E-field εr /1 = 50 times higher than in the dielectric, affecting the mode distribution. For the manufactured prototype, no fillings are used to avoid or reduce the airgap. Due to the rotational symmetry of the ring resonator and the tool mounting, a self-alignment of the resonance mode to the excitation from the reader is expected. Instead of such an omnidirectional RCS pattern, Fig. 3.2 shows that four lobes are clearly differentiated in the E-plane, which cannot occur in a rotationally symmetric structure. The minima in the RCS pattern are attributed to the angles at which an E-field maximum of the EH211 resonance mode at the air-gap, whose strong influence on the quality factor and resonance frequency of dielectric resonators is mentioned in multiple publications [10–12]. This effect could be worsened by the oxidation of the metal, accelerated when exposed to high temperatures during the temperature characterization. The air-gap could and should be avoided in order to achieve an omnidirectional RCS pattern by, for example, a combination of (i) fabricating the tool and resonator to tightly fit, (ii) metalizing the inner surface of the ceramic resonator, and (iii) filling any unavoidable air-gap with conductive paste.

3.1.1.1

Temperature Characterization

The resonator is mounted on the tool and introduced in a kiln. The top metal cover of the kiln is removed, leaving 8 cm of fireclay for thermal isolation, and a horn antenna connected to a VNA Agilent Technologies PNA-X N5247A is positioned over the kiln as shown in Fig. 3.3a. With this setup, the resonance frequency of the DR can be retrieved from the S11 parameter by time-gating the received signal and therefore suppressing the high-power reflections, i.e., clutter, from the metallic kiln.

3.1 Sensing with a Single Air-Cladded High-Q Resonator

35

Fig. 3.2 Normalized RCS patterns in the a E-plane, and b H-plane. The maximum RCS is 19.5 dBm2 . A 15 cm diameter metal sphere is used as a reference for the RCS measurement

These high-power reflections correspond to approximately the first 40 ns, as shown in Fig. 3.3b. For the realization of this time-frequency representation, the S11 is first converted to the time-domain with an Inverse Fast Fourier Transform (IFFT), and multiple time-gated time-domain signals are generated, each by time-gating the original timedomain response with a time window of constant span but different start time. Finally, each of the time-gated time-domain signals is converted to the frequency domain with a Fast Fourier Transform (FFT), and represented in the time-frequency plot as a single column. The time-axis value corresponds to the start time of the time window used to generate that column. Notice that, the wider the span of the time window, the higher the resolution in the frequency domain, but the more signal is considered that came after the value specified in the x-axis. Despite the appearance of side lobes, rectangular time windows are used in all cases to maximize the frequency resolution for a given time span. These figures are used multiple times throughout this work, since they allow for a simultaneous perception of the time and frequency characteristics of the designed tags. To characterize the temperature response, and due to the high thermal energy storage, the temperature of the kiln is set above the target maximum temperature and kept constant for up to 10 min until the measured resonance frequency reaches a steady state. Afterwards, the heating is turned off and measurements are automatically taken every 25 ◦C decay by a computer script connected to the temperature sensor of the kiln and to the VNA. Due to the enhanced oxidation at high temperatures, the tools are measured up to 300 ◦C and 400 ◦C for the copper and steel tools, respectively. The measured results for both tool materials are shown in Fig. 3.3c. The higher loaded Q-factor of the resonator mounted on a copper tool is appreciated throughout the measured temperature range, attributed to the lower conductor

36

3 Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves

Fig. 3.3 Temperature characterization of the machine tool sensor. a Measurement setup, b timefrequency response with a copper tool, c extracted Q-factor and resonance frequency over temperature [4, 5]

losses of copper. Still, both tools show stable operation in the measured frequencies. The average temperature sensitivity K T = 65 ppm/◦C does not vary with the tool material. A key parameter of the temperature sensor is its response time, i.e., how fast it reacts to temperature variations in the tool. Assuming the sensor is mounted on the tool as shown in Fig. 3.1b, we define the contact surface area between the tool and the sensor as As . It is further assumed that the tool and the ceramic sensor are in vacuum, have infinite thermal conductivity within the material, and no heat radiation. Thus, the contact area between the sensor and the tool is the only heat-transfer mechanism considered in the system. If the tool is instantaneously heated up Δ T , the temperature change in the sensor is described as [13]: Tsensor (t) = T0 + Δ T (1 − e−t/τT ), with τT =

ρcp V , h c As

(3.1)

where T0 is the initial temperature and τT the time constant. ρ is the density of the ceramic, cp the specific heat, V the volume of the sensor, and h c the interfacial heat transfer conductance at the metal-ceramic interface.

3.1 Sensing with a Single Air-Cladded High-Q Resonator Table 3.1 Response times of the machine tool temperature sensor. Tool diameter (mm) τT−min (s) τT−max (s) d = 10 d = 12 d = 15 Integrated design

7.2 6.2 5.1 0.5

40.5 35.3 28.6 2.6

37

95% response time (s) 21.5...121.5 18.7...106.0 15.1...85.8 1.4...7.9

The ceramic used is proprietary, and insufficient information on it is available from the manufacturer, so polycristalline 98% dense Al2 O3 is assumed for the calculation. Its volumetric mass density is ρ = 3.9 gcm3 , and its specific heat capacity cp = 940 J kg−1 ◦C−1 at 127 ◦C [13]. According to Table 2.1 in [13], the interfacial heat transfer conductance between a metal-ceramic interface ranges from 1500 W m−2 ◦ C−1 to 8500 W m−2 ◦ C−1 , which are used as the worst (slowest) and best (fastest) case scenarios for the calculation, respectively. The volume of the ceramic resonator is V = 9.42 cm3 , and its contact surface area depends on the selected inset. Table 3.1 summarizes the heating times for the three considered tool diameters. The response time is defined as 3 · τT , i.e., when the sensor has adapted to 95% of the abrupt temperature change. Its value above 15 s in all cases shows the importance of miniaturizing the sensor. Nevertheless, these values are only a guidance to understand how different parameters may affect the sensor’ response. When operating in a tool, the sensor would react quite slow to any temperature changes, and good predictions based on the tendency of the temperature would be needed, which could compromise the reliability of the system. The design of integrated sensors would improve the response time, as is commented in the discussion.

3.1.1.2

Machine Tool Operation

The performance of the sensor in high-reflective environments and at high angular speeds has been tested in a real machine tool. In Fig. 3.4a, the copper tool with the sensor is mounted on the spindle, and the protective door closed. Outside, a VNA connected to a horn antenna is placed to measure the S11 parameter. The results for angular velocities from 10,000 rpm can be seen in Fig. 3.4b, demonstrating how the resonance can be read out at high rotational speeds. The closed-cavity with large metallic walls creates a highly reflective environment, where the interrogation pulse is reflected multiple times. It is therefore concluded that the system is limited by interference rather than noise. Furthermore, owing to the metallic inner walls of the machine tool, a standing wave is identified by moving the sensor towards the back metal wall, i.e., towards or away from the antenna in Fig. 3.4a. Depending on the distance, a constructive or destructive interference occurs, and the resonance peak is clearly visible or disappears, respectively. To counteract this effect, a different antenna position that avoids normal incidence to the machine tool’s

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3 Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves

tool

sensor

antenna (a)

(b)

Fig. 3.4 Temperature sensor integrated into a machine tool. a Measurement setup showing that the resonance frequency is monitored through the transparent window with the door of the machine closed. b Frequency response for different rotational speeds of the resonator mounted in the copper dummy tool [4]

inner metallic walls, or more complex MIMO antenna setups with propagation path diversity would be advantageous. Machine tools are often cooled with oil during their operation. Switching on the application of oil greatly distorts the resonance mode of the sensor and, in its current form, no readout is possible. 3.1.1.3

Discussion

This work has demonstrated that the generic concept of using a high-Q resonator as a chipless wireless temperature sensor introduced in [7] can be adapted to the concrete application of a machine tool. The application-specific need to withstand high rotation speeds is considered for the selection of a ring-shape resonator. We began by selecting a resonance mode with a high-Q factor, and analyzing the effects of the mounting, i.e., the tool on its performance. Besides shape and conductivity of the tool, the importance of gaining insights on its temperature transient response with a simplified model has been shown. Despite the intuitive thought that Doppler and rotation misalignment could be a problem, the calculation of worst-case values showcased that these factors can be neglected due to the short diameter and low measurement time-spans, respectively. The angle-dependent radiation pattern highlights the importance of accurate transitions between the high-εr and metal interfaces when using resonance modes with a maximum E-field at that interface. Regardless, the sensor exhibits a maximum RCS of –19.48 dBm2 . This RCS and the high-Q factors between 400 and 1300 depending on the tool material and temperature allow for the wireless temperature characterization of the sensor inside a kiln up to 400 ◦C. The sensor shows high temperature stability and linearity, with decaying Q-factors and increasing resonance frequencies for higher temperatures. The sensitivity is 65 ppm/◦C for

3.1 Sensing with a Single Air-Cladded High-Q Resonator

39

Fig. 3.5 a Smaller sensor of the same ceramic material integrated into a machine tool. b Simulated time-frequency response assuming copper as the tool material

both the copper and steel mounts. Furthermore, the operation of the sensor has been verified inside a real machine tool, where the potential of the long-ringing response is recognized. Despite its good operation, some drawbacks still need to be addressed and are discussed in the following. The high-volume and low contact area of the sensor, together with the high heat capacity of ceramics, result in a large time constant τT , i.e., slow sensor response times ≫10 s to temperature changes in the tool. The feasibility of a much smaller sensor (6 mm height, 7 mm inner and 9 mm outer radius) made from the same ceramic material, K-50, that still works well below 10 GHz and that reduces the ratio of volume to contact area V /As by a factor of 15 is studied in simulations. An additional benefit is that it entirely avoids an interfering contour. Figure 3.5 shows the sensor integrated in a dummy tool alongside time-gated S-parameter results of a full-wave analysis. The radiation quality factor for the E H211 is 1200, the simulated loaded Q-factor is 700 for a copper tool, and the thermal time constant τT can be reduced down to 0.5 s as summarized in Table 3.1. With the design of an integrated sensor, the addition of a low-permittivity coating could be considered to reduce its cross-sensitivity to external events such as (i) cutting oil applied to the tool, and (ii) material chirps that stick or dynamically move around the sensor’s position in the tool. In addition, instead of designing a sensor with omnidirectional RCS, it could be designed to be excited and re-radiate only from a few Nlobes directions. This would result in a periodic excitation of the resonator Nlobes times per rotation. A reader knowing the angular velocity of the sensor can anticipate this periodicity of the response, and use this as an additional parameter to distinguish the sensor’s response from clutter, which does not respond to the same periodic pattern. A possibility to miniaturize the size of the sensor is the use of higher permittivity materials. The ferroelectric material Barium Strontium Titanate (BST) is investigated in [14] to (i) decrease the size of the sensor without increasing the operation frequency, and (ii) achieve a high temperature sensitivity due to the strong variation of the relative permittivity with temperature.

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3 Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves

Since the high dielectric losses (Q diel < 200 for frequencies above 1 GHz) of BST notably reduce the achievable ringing time of the sensor, BST is mixed with lower loss magnesium borate (Mg3 B2 O6 , εr = 8), to get higher dielectric Q-factors with lower but still superior sensitivity. It is shown that cylindrical resonators of 0.4 cm height and 1 cm diameter resonate at 2 GHz with temperature sensitivities greater than 1250 ppm/◦C. Despite the small size and outstanding sensitivity, the measured loaded quality factors around 35 due to the high-losses of BST still limit its use as wireless high-Q resonators in high-cluttered environments. These results encourage further developments of wireless high-Q resonator-based sensors for industrial applications in harsh environments. In the following, the potential use of controlled variations of the near-field of the resonator for sensing other physical parameters is presented with the example of a pressure sensor.

3.1.2 Sensing Other Physical Parameters Through Controlled Near-Field Variations A DR is sensitive to its surrounding medium. Within the near-field region, variations outside the dielectric cavity can change its resonance frequency and quality factor. This is often a drawback, as the wireless readout of the machine tool sensor was not possible while applying cutting oil to the tool. However, such variations of the near-field can also be used to measure different physical parameters, i.e., for the realization of wireless sensors. One example is the realization of a pressure sensor by altering the near-field of a ceramic resonator, as presented in [15, 16]. Figure 3.6a shows the sketch of the potential integration of the sensor in a machine tool. Pressure is applied to a diaphragm, whose deflection affects the near-field of a resonator as shown in Fig. 3.6b. The approximation of the metallic diaphragm to the resonator changes the resonance mode distribution, shifting its resonance frequency to higher frequencies. The operation frequency of the fabricated prototype is 21 GHz, and has a lower Q-factor, varying from 140 to 200. Still, its resonance frequency can be recognized in Fig. 3.6c. Notice that, differently than for the machine tool sensor, the sensitivity of the sensor does not depend on the ceramic material, and it can be adapted to the desired application by varying the diaphragm thickness and its distance to the resonator with the spacers. Figure 3.6c, d show an example of the wireless readout at a distance of 60 cm. The membrane thickness equals 100 µm, and its distance to the resonator for a straight membrane is 1 mm. The measured sensitivity equals 266 ppm/kPa for pressures from 1 kPa to 200 kPa.

3.2 Hybrid Modulation Enabled by a High-Q Resonator In addition to increasing the readout robustness in dynamic cluttered environments, the long ringing of high-Q resonators can be used to separate information encoded in two different modulation techniques, being one of them based on the high-Q

3.2 Hybrid Modulation Enabled by a High-Q Resonator

horn antenna

high-Q resonator spacers diaphragm

rx high-Q

41

tx

oil pressure system

oil pressure system

(a)

(b)

(c)

(d)

Fig. 3.6 a Potential integration of the pressure sensor in the milling head of a machine tool, as shown in [16]. b Extruded view of the manufactured prototype. c Measured time-frequency response for 3 different pressures on the diaphragm at a distance equal to 60 cm from the reader. d Measured resonance frequency and Q-factor for different pressures

resonator concept. First, this section introduces the proposed hybrid modulation scheme. Afterwards, the potential of such a hybrid coding scheme is demonstrated by integrating a narrowband temperature sensor based on a ceramic high-Q resonator into a wideband TDR chipless RFID tag. This section, including the high-Q resonator design and hybrid operation, are based on [17, 18], while the phase-coded TDR chipless RFID tag is originally described in [19].

3.2.1 Hybrid Modulation Scheme Based on a High-Q Resonator and Phase-Coded TDR The block diagram of the tag and its modulation scheme are sketched in Fig. 3.7. A schematic representation of the proposed coding scheme is shown in Fig. 3.8. The working principle of the high-Q resonator as a temperature sensor is the same as for the machine tool sensor shown in Sect. 3.1.1, namely a resonance frequency

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3 Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves

Fig. 3.7 Block diagram of the proposed hybrid coding scheme for chipless identification and sensing based on a high-Q resonator and TDR [18]

Fig. 3.8 Proposed modulation scheme. The first four backscattered pulses (in blue) contain the identification on their phase, while the long resonating response (in orange) contains the sensed information on its resonance frequency. FT: Fourier Transform. IFT: Inverse Fourier Transform [18]

shift with temperature. In the following, the working principle of Phase-Coded TDR and the hybrid modulation are further explained. Phase-Coded Time Domain Reflectometry TDR is based on the reflection of an interrogation pulse (Reflectometry) at certain instants (Time Domain). The reader can identify the tag according to its reflected pulse pattern as shown in blue in Fig. 3.8. The interrogation pulse is received and guided through the tag and sustains dielectric and metallic losses. Furthermore, the maximum area of the tag is often limited due to fabrication or application constrains. The main technological limitation of TDR tags is therefore the maximum delay achievable within a certain maximum tag loss and area. To increase the number of bits, wide bandwidths are advantageous to reduce the pulse width, and therefore the number of reflected pulses that can be distinguished in a certain time interval. For a given number of pulses, higher-order modulation schemes increase the number of bits encoded. Likewise, for the same number of bits, the number of pulses and with it the maximum delay can be reduced, reducing the tag losses. A higher-order modulation scheme can be achieved by varying the magnitude or position of the reflected pulse, as presented in several examples in [20]. However, an amplitude modulation scheme necessarily reduces the reflected power, decreasing the maximum range. Similarly, Pulse Position Modulation (PPM) needs larger delays for each bit, resulting in higher propagating losses.

3.2 Hybrid Modulation Enabled by a High-Q Resonator

43

Phase modulation in TDR is presented in [21, 22]. In this modulation scheme, the interrogation pulse is reflected at fixed times to minimize inter-symbol interference, and the encoding of several bits is achieved by varying the phase of each reflected pulse according to the position of a reflection in the modulation section within half a wavelength. Ideal delay lines generate a fixed delay of a broadband signal with no attenuation nor dispersion, i.e., no frequency-dependent response on it. Nevertheless, due to the high propagation velocity of the guided electromagnetic signal, delay lines require long transmission lines and are the components using most of the available area in the tag. Hybrid Modulation The hybrid modulation scheme is obtained utilizing both TDR and frequency position coding within the same tag and bandwidth. Figure 3.9 shows the time domain response of the tag for different Q-factors and losses per delay section, where the following effects are observed: First, the interrogation pulse is reflected in four phase-coded TDR pulses. After the first four coding pulses, i.e., 18 ns, the identification ends, and additional pulses of decaying amplitudes are observed. These pulses are caused by the multiple reflections originated at the modulation elements, which have a symmetrical response for the incoming and outcoming pulses, i.e., S11 = S22 ; S21 = S12 . The slope of this decay is proportional to the propagation losses inside the delay sections, and is the only difference between the two plots in Fig. 3.9 with (a) 1 dBm and (b) 2 dBm loss per delay section. After (a) 50 ns and (b) 150 ns of dominant TDR signal, the power reflected by the high-Q resonator becomes more prominent in the tag’s backscattered signal due to its lower losses. The faster the decay of the multiple reflections, the sooner the characteristic linear decay of the high-Q resonator in the logarithmic scale is differentiated. The slope of this linear decay is inversely proportional to the loaded Q-factor of the resonator, i.e., higher Q-factors increase the ringing time of the signal before decaying below the noise floor and, therefore, allow for a more flexible use of time-gating to separate the resonator response from the TDR section, as well as from potential unexpected clutter reflections in a dynamic environment. Due to the narrowband response of the resonator, the distortion of the TDR pulses is low, and no difference is seen in the Symbol Error Ratio (SER) simulations in Fig. 3.10a. The addition of the S-parameter block to the simulation does reduce the performance of the TDR tag, and 2.5 dB increase in the received, i.e., transmitted power would be needed to compensate for this effect as depicted in Fig. 3.10b.

3.2.2 Implementation in Ultrawideband The presented hybrid modulation has been implemented in Rogers RT/Duroid® 6010.2LM (in the following, RT6010), using microstrip transmission lines and a high-Q resonator made out of K-50 ceramic. The material properties are shown in

44

3 Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves

100

Ql 1000 720 500 375

10−1 10−2 10−3

0

10

20 30 Es /N0

40

50

symbol error ratio (SER)

symbol error ratio (SER)

Fig. 3.9 Simulated time-domain response of a tag with 1 GHz bandwidth for high-Q resonators and five different loaded Q-factors. The resonators are simulated in CST Studio Suite and appended to the rest of the tag by S-parameter concatenation in a Matlab® script. Each simulated tag contains four phase-coded reflected pulses in the first 16 ns, and a single high-Q resonator of varying Qfactor. Substrate losses are modelled as insertion losses within each delay section and equal to a 1 dBm and b 2 dBm [18] 100

1 dB 1 dB + s 2 dB 2 dB + s

−1

10

10−2 10−3

0

10

20

30 Es /N0

40

50

(b)

(a)

Fig. 3.10 SER over symbol energy to noise power density ratio, E s /N0 , for a different resonators with 1 dBm loss per delay section, and b different losses per delay line section without and with a high-Q resonator(s). Each point in the curve is calculated with 10000 simulations using randomly generated identification codes. Three pulses are coded in 8-PSK with a 1 GHz bandwidth. An equalizer is used to compensate the deterministic influence of inter-symbol interference due to the symmetric response of the shunt stubs [18] Table 3.2 Material properties [18] εr Material RT6010 [23] K-50 ceramic [8]

10.34 ± 0.07 49.53

KT

tanδ

–700 ppm/◦C

≤ 0.0027 0.00024

Table 3.2. The tags are designed to operate in the frequency range from 7.375 GHz to 8.375 GHz, which is internationally available for Ultra Wideband (UWB) applications as shown in Fig. 3.11a. Two different tags are realized: one with an RF 3.5 mm connector, a high-Q resonator, and 4 symbols is used for the characterization of the sensor, while a tag with a monopole antenna, a high-Q resonator, and 3 symbols is used for the wireless measurements.

3.2 Hybrid Modulation Enabled by a High-Q Resonator

45

Fig. 3.11 a Regulated EIRP for UWB systems in Europe, USA, and Japan [24]. The hybrid tag’s frequency band is highlighted in black. The bandwidth could be increased if the tag is designed for one or two regions, reducing the necessary delay between pulses and therefore the size and TDR losses per delay section. b Fabricated hybrid tag [18]

Fig. 3.12 a Layout of the double-fed square monopole antenna on RT6010. The dimensions are listed in Table A.1. b Simulated reflection coefficient of the double-fed square monopole antenna. The desired operation bandwidth is highlighted in red [18]

3.2.2.1

Wireless Readout

A monopole antenna is designed to achieve wideband operation. Besides, this type of antenna presents a broad directivity pattern, which allows for its readout independently of the position of the reader. To improve the linear polarization, a double-fed square monopole antenna designed as described in [25] for εr = 1 is adapted to the higher relative permittivity substrate. The presence of two symmetrical feeds hinders the excitation of cross-polarized resonance modes. Simulation results show a return loss above 17 dBm for the whole bandwidth as shown in Fig. 3.12. The layout of the monopole is shown in Fig. 3.12a, and its dimensions on Table A.1.

3.2.2.2

Identification and Sensing

A conventional microstrip transmission line is used to guide the interrogation pulse through the tag. For the realization of the delay sections, the microstrip transmission line is meandered to reduce the maximum length of the tag, while keeping a simple design.

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3 Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves

The modulation elements generate the time-domain modulated reflection, and can be realized with different techniques such as λ/4 transformers, shunt stubs, and power dividers with one output terminated with a short or an open. The advantages and disadvantages of each technique are described in [19]. For the presented tag, the modulation sections are realized with shunt stubs. The deterministic overlap of previous symbols to the following ones due to the reciprocity of the shunt stubs can be predicted and filtered out by the reader, assuming the previous symbols have been successfully decoded, and the tag’s losses are known. By using power dividers in the modulating sections, the effect of multiple reflections could be further reduced. In the following, this work concentrates on the resonant part of the tag and its simultaneous operation with the phase-coded TDR. For more information on the implementation of the identification part, the reader is referred to [18]. Sensing In the machine tool, the high-Q resonator acts as the antenna itself, i.e., is excited by a plane wave, and re-radiates into free-space. For the current design, the temperature sensor is integrated into the tag, i.e., a corporate feed approach as introduced in Sect. 2.2.2. Thus, the resonance modes should have low radiation into free-space. The resonator is coupled to the microstrip line by progressively tapering the substrate on one side as shown in Fig. 3.13. Such progressive tapering avoids any abrupt variation of the line impedance, which would cause an undesired reflection. The resonator is fabricated from the same material as the machine tool sensor. Since the dielectric quality factor is high, Q diel = 1/tanδ = 4167, the loaded quality factor is mainly

Fig. 3.13 Integration of the DR next to the microstrip line. Ain corresponds to the input signal in the tag and Aout to the signal that reaches the TDR-based identification section of the tag. The c 2018 IEEE distance is defined from the side of the microstrip line to the beginning of the DR. ◦ [18]

3.2 Hybrid Modulation Enabled by a High-Q Resonator

47

(a) E-field, HE212

(b) H-field, HE212

(c) E-field, EH411

(d) H-field, EH411

Fig. 3.14 Simulated electric and magnetic field stengths inside the resonator. Red corresponds to maximum magnitude, while dark blue to magnitudes at least 40 dB weaker than the maximum. The resonance modes are a, b HE212 , resonating at 7.50 GHz, and c, d EH411 , resonating at 7.79 GHz. The cutting plane used for these figures can be seen in Fig. 3.13 [18]

determined by its coupling to the microstrip line as well as the conductor losses due to the presence of the ground plane. The strong influence of small air-gaps at the metal-dielectric interface on the quality factor and resonance frequency of dielectric resonators is studied with the temperature sensor integrated in the hybrid tag. Conductor losses are related to the presence of surface currents in the conductor, which are proportional to the magnetic field. To verify their influence on the sensor, two different resonance modes are considered, which are distinguished by the presence of a maximum or minimum H-field at the metal-dielectric interference and depicted in Fig. 3.14. Notice that, for the sole purpose of sensing, a single resonance mode would be sufficient, and that the presence of two is pursued to study the effect of manufacturing inaccuracies depending on the mode distribution. In both cases, the maxima of the resonance modes are contained inside the high-permittivity cylinder, indicating that both are resonances of the high-permittivity material and not from other circuit parts. While the H-field maxima of the EH411 resonance mode are present at the metal interface, the HE212 has its minima there. Both resonant modes show high loaded Q-factors > 1000 in simulations including copper as a lossy conductor. Despite similar simulations results without an air-gap, the effect of unavoidable manufacturing inaccuracies at the dielectric-metal interface should have a more pronounced effect on the EH411 resonance mode. This statement is tested by fabricating several resonators with different bottom connections to ground. The first resonator is a plain cylindrical resonator placed over the tag’s copper ground plate without any fixture. Two additional samples are metalized on the bottom surface in contact with the tag’s copper ground plate, one by sputtering 100 nm of copper, and the other with a few nm nickel adherent layer,

48

3 Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves

Fig. 3.15 Measured loaded quality factor over dr in Fig. 3.13 for the a HE212 and b EH411 resonance mode of the DR. The measurement is performed by using the tag with a RF 3.5 mm connector [18]

and electroplating 2µ 3µ of gold. Even if thick copper has a higher conductivity, i.e., lower losses than gold, 100 nm is well below the skin-depth of 735 nm, resulting in higher losses if it is not in good contact with the thicker ground plate. These three resonators are measured in two different configurations, without and with conductive glue as an adherent to the ground plane, giving a total of six configurations. The DRs can be placed at different distances from the microstrip line within the hole in the substrate, varying the coupling quality factor and, therefore, the measured loaded quality factor. The measurements are realized using time-gating with the tag connected to the VNA and, due to the TDR identification and its multiple reflections, only resonance modes with low losses and high loaded quality factors, Q l , can be distinguished from the identification. The results are shown in Fig. 3.15, where only the values are shown for which the distinction of the resonance peak was possible. For longer distances between the DRs and the microstrip line, dr , the measured loaded quality factor increases, until the resonance frequency cannot be detected anymore due to the weak coupling. A saturating effect can be observed as the loaded quality factor tends to the unloaded quality factor. A maximum quality factor of 2700 is extracted for the resonator electroplated with gold unglued to the copper plate, while the resonator with a thin 100 nm bottom ground shows lower Q l than the ceramic cylinder directly positioned on the metal plate. It is therefore concluded that the direct metallization of the sensor is advantageous, but such metallization should be thick in terms of skin depth to behave as a low-loss conductor. Furthermore, using a resonance mode with low surface currents at the metal interface greatly relaxes the manufacturing tolerances and should therefore be the preferred option, especially the higher the permittivity and operation frequency due to the shorter wavelengths. A temperature characterization of the sensor is performed by heating the glued gold DR with a hot-air gun HG 2310 LCD electronic, and reading out the temperature of the ceramic resonator with an infrared camera FLUKE© Ti10 with a resolution of ±2.5 ◦C. The resonator is heated to 80 ◦C and this temperature is kept for 1 min. Following, the hot-air gun is switched off, and the temperature is monitored by the infrared camera during cooling. The reflection response of the sensor S11 is saved, and its resonance frequency is determined after time gating. It takes around 3 min until the resonator reaches room temperature and the measurement is finished. The resulting

3.2 Hybrid Modulation Enabled by a High-Q Resonator

49

Fig. 3.16 Wireless temperature characterization of the glued gold DR for the a HE212 and b EH411 resonance mode [18]

resonance frequency shift over temperature of both resonance modes can be seen in Fig. 3.16. The resonance frequency increases with temperature, with a normalized sensitivity equal to 141 ppm/◦C for the HE212 resonance mode, and 164 ppm/◦C for the (b) EH411 resonance mode.

3.2.2.3

Hybrid Operation and Discussion

The hybrid operation is verified by obtaining both, the phase-coded TDR pulses and the resonance frequency of the high-Q resonator, with a single measurement from 7.375 GHz to 8.375 GHz. A differential measurement is performed first without and then with the tag in the setup shown in Fig. 3.17a. In Fig. 3.17b, the reflection caused by the tag’s structure, also known as structural mode, followed by the three reflected pulses can be observed. Afterwards, the response shown in Fig. 3.17c shows the long-ringing of the resonator, whose resonance frequency for three different starts of the time gating windows is depicted in Fig. 3.17d. The measured loaded Q-factors of the resonator are 1000 at 7.568 GHz, and 870 at 7.842 GHz. The presented results demonstrate that the chipless wireless high-Q resonator concept can be used to complement other wideband modulation techniques, which can share the same tag, bandwidth, and interrogation signal. The designed and characterized tag shows the integration of a high-Q temperature sensor with a 9-bit TDR tag for identification. While the results presented in this chapter prove the concept, several application-specific improvements are needed to advance towards real applications. Some generally applicable improvements on TDR tags include (i) the reduction of multiple reflections between modulating sections to achieve a faster decay of the reflected signal, as well as (ii) to reduce the area of the tag. (i) can be reduced by using power dividers instead of shunt stubs in the modulation sections [19]. In the case of (ii), the tag’s area can be halved by substituting the meandered-line delay sections with filter-based techniques as in [26], as well as using multi-layer structures

50

3 Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves

Fig. 3.17 a Measurement setup for the hybrid operation. The distance between the tag and the reader antenna is 25 cm, and a differential measurement is performed without and with the tag. b Time-domain response during the first 30 ns. After the reflections from the structural mode of the monopole antenna, the three phase-coded symbols are received from the TDR coding. Complete received signal of the hybrid modulation in the c time domain, and d frequency domain [18]

with Low Temperature Cofired Ceramics (LTCC). Alternatively, surface acoustic wave (SAW) TDR tags could be used for longer delays. Since the identification part of the tag does not rely on high-Q resonators, the advantage of clutter suppression by the long-ringing properties of the resonator only applies to the sensing. However, for certain applications, such as those where the successful identification of the sensor is not as critical as the continuous and reliable readout of the sensor’s data, these conditions could suffice. A further improvement of the readout from TDR tags via depolarizing tags as introduced in Sect. 2.3 can be used in this tag. The main drawback being the need of a more complex reader that can operate in both polarizations. More information on depolarizing tags can be found in [27, 28]. The presented hybrid modulation technique shows how the low distortion between wide- and narrowband modulation techniques can be exploited for chipless wireless systems where one of the modulations is achieved by a high-Q resonator and the identification relies on several reflected pulses. However, this principle is limited neither to sensing nor to a single resonator, as is shown in Chap. 5 with the example appli-

3.3 Limitations and Performance Deterioration Due to Lossy Materials and Packaging

51

cation of an indoor localization system with simultaneous localization (wideband), and identification (narrowband) properties. For certain applications, such as the accurate radar ranging for an indoor localization system, higher bandwidths are desired to achieve sub-mm resolutions. When transferring the presented high-Q resonator concept to mm-Wave frequencies, some limitations of the single air-cladded high-Q resonator become apparent.

3.3 Limitations and Performance Deterioration Due to Lossy Materials and Packaging For microwave frequencies, the realization of single high-Q resonators with highpermittivity materials in contact with other materials such as highly conductive polished metal surfaces is possible. However, for the realization of mm-Wave and THz tags, some additional limiting factors need to be considered: • Increasing metal losses. • Lack of low-loss materials with high relative permittivities, εr ≥ 15. • Performance deterioration due to packaging. Increasing metal losses The losses in a conductor can be described by Ohm’s law, V = I · R, where V is the voltage, and I is the current. The loss resistance of a given wire can be calculated with the formula ρl l = , (3.2) R= A σc A where ρ is the resistance of the material, σc the conductivity, l the length, and A the effective cross-section of the conductor contributing to the wave propagation. For high-frequency electromagnetic waves, the current propagates through the surface of the conductor, so that the effective cross-section, A, is lower than the physical cross-section. The current distribution along the radius of a cylindrical wire follows a decaying exponential function described by its mean value, the skin depth [29] / δ=

1 , π · f ·μ·σ

(3.3)

where f is the operation frequency, and μ is the permeability of the conductor. It can be observed that the skin depth decreases with frequency, δ ∝ f −1/2 . For a good conductor like copper, it ranges from 1300 nm at 2.5 GHz to 130 nm at 250 GHz. The decrease of the skin depth with the square root of frequency linearly reduces the cross-section of the conductor contributing to the movement of electrons, so that the material resistance, and with it the loss resistance, linearly increase with

52

3 Wireless Sensing with Single Air-Cladded High-Q Resonators at Microwaves

Metal losses ∝ R ∝ A−1/2 ∝ δ −1/2 ∝ f 1/2 ,

(3.4)

i.e., the metal losses increase with the square root of the operating frequency. The presented model assumes a flat conductor surface, which is a good approximation as long as the surface roughness remains notably below the skin depth. As they become comparable, the roughness has a noticeable impact on the distance that the wave travels between two points along the conductor. This could be included in Eq. 3.2 as an increase in the conductor’s effective length for the same start and end point, since the currents do not propagate following a straight line along the conductor. [29] shows that the losses increase up to a factor 2, while [30] determines that it can be even higher than that when the roughness is not modelled as periodic variations. Typical surface roughness values of metal cladding in RF substrates given by [31] range from 300 nm to 2400 nm, which is larger than the skin depth at high-mm-Wave and frequencies. Figure 3.18 shows the measured unloaded quality factors, Q u , of several publications centered around low-loss filter design. The expected decrease of the unloaded quality factor with the square root of frequency is observed. From the low-mm-Wave, where metal cavities are realized with very low losses, to high-mm-Wave frequencies, where the unloaded quality factor of metal cavities decays below 1000. Therefore, the use of conductors must be considered carefully when designing mm-Wave and THz tags based on high-Q resonators. More information and references about the publications consulted for the figure are included in Appendix B. Lack of low loss materials with high relative permittivities For microwave frequencies, high-permittivity low-loss ceramic materials with εr > 30, have been used in this chapter to create dielectric cavities with low-order highQ resonance modes. However, these materials present higher losses at mm-Wave frequencies and are not suitable for the realization of high-Q resonators. Other lowloss high-permittivity ceramics still show high losses at high-mm-Wave and THz frequencies, as shown in [32] for different mixtures of Al2 O3 and zirconia powders. Recently, the realization of titanium dioxide (TiO2 ) with ultra-high relative permittivity (εr = 102.3), and ultra-low-loss with Q diel = 238 was presented [33]. However, such dielectric quality factors do not permit the realization of efficient resonators with high resonator efficiency ηQ and high loaded quality factors above 200 as shown in Sect. 2.2.2. A literature review shows High-Resistive Silicon (HR-Si) and Al2 O3 as the most suitable materials for this frequency range. The relative permittivity of HR-Si is 11.68 [34], while for Al2 O3 it varies from 9 to 10, depending on its density. To achieve high-Q with single air-cladded resonators in these low-loss moderate permittivity materials, a possible solution is to rely on higher order resonant modes [9]. However, these would increase the mode density, i.e., reduce the FSR needed for successful distinct identification and sensing.

References

53

Fig. 3.18 Measured unloaded quality factors from metallic resonators pursuing high-Q, including technology and main resonator material. The source data is summarized in Table B.1. WG: WaveGuide; BoN: Bed of Nails; SIW: Substrate-Integrated Waveguide.

Performance deterioration due to packaging Even small variations in the near field of the resonators result in resonance frequency shifts and can increase the losses. In addition, for the same resonance mode, the higher the resonance frequency, the smaller these resonators become, hindering their fabrication and integration in the final devices. Therefore, the integration of the resonators in a well-defined and low-loss support structure needs to be considered in the design phase. For these three reasons, the next chapter explores the use of metallic as well as full-dielectric EBG structures to realize high-Q resonators with the limited materials available at mm-Wave, and especially at THz frequencies.

References 1. Kubina B (2016) Chipless-wireless high-temperature sensing in time-variant environments. In: IMP Dissertation, TU Darmstadt 2. Byrne G, Dornfeld D, Inasaki I, Ketteler G, König W, Teti R (1995) Tool condition monitoring (TCM)-the status of research and industrial application. CIRP Annals 44(2):541–567

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3. Wu D, Jennings C, Terpenny J, Gao RX, Kumara S (2017) A comparative study on machine learning algorithms for smart manufacturing: tool wear prediction using random forests. J Manuf Sci Eng 139(7) 4. Jiménez Sáez A, Polat F, Mandel C, Schüßler M, Kubina B, Scherer T, Lautenschläager N, Jakoby R (2016) Chipless wireless temperature sensor for machine tools based on a dielectric ring resonator. Procedia Eng 168:1231–1236 5. Mandel C, Jiménez-Sáez A, Polat E, Schüßler M, Kubina B, Scherer T, Lautenschläger N, Jakoby R (2017) Dielectric ring resonators as chipless temperature sensors for wire- less machine tool monitoring. In: 2017 11th European conference on antennas and propagation (EUCAP). IEEE, pp 3912–3916 6. AIR802. FCC 2.4 GHz band rules. https://www.air802.com/files/FCC-Rules-and-Regulations. pdf 7. Bernd K, Martin S, Christian M, Arshad M, Rolf J (2013) Wireless high-temperature sensing with a chipless tag based on a dielectric resonator antenna. In: IEEE sensors. IEEE, pp 1–4 8. Inc. National Magnetics Group. Dielectric materials. https://www.magneticsgroup.com/ material/k/.Online; accessed 12 March 2021 9. Mongia RK, Bhartia P (1994) Dielectric resonator antennas—a review and general design relations for resonant frequency and bandwidth. Int J Microw Millim-Wave Comput-Aided Eng 4(3):230–247 10. Drossos G, Wu Z, Davis L (1999) The air gap effect on a microstrip-coupled cylindrical dielectric resonator antenna. Microw Opt Technol Lett 20(1):36–40 11. Junker GP, Kishk AA, Glisson AW, Kajfez D (1995) Effect of fabrication imperfections for ground-plane-backed dielectric-resonator antennas. IEEE Antennas Propag Mag 37(1):40–47 12. Junker GP, Kishk AA, Glisson AW, Kajifez D (1994) Effect of air gap on cylindrical dielectric resonator antenna operating in TM/sub 01/mode. Electron Lett 30(2):97–98 13. Lienhard IV, John H (2020) A heat transfer textbook, 5th edn. Version 5.10. Cambridge, MA: Phlogiston Press, p 784. http://ahtt.mit.edu 14. Jiménez-Sáez A, Schumacher P, Häuser K, Schüßler M, Binder JR, Jakoby R (2019) Chipless wireless high temperature sensing based on a multilayer dielectric resonator. In: IEEE sensors. IEEE, pp 1–4 15. Schumacher P, Schuster C, Jiménez-Sáez A, Schüßler M, Jakoby R (2018) Passive chipless wireless pressure sensor for Harsh and reflective environments. In: 11th German microwave conference (GeMiC). IEEE, pp 227–230 16. Schuster C, Schumacher P, Schüßler M, Jiménez-Sáez A, Jakoby R (2017) Passive chipless wireless pressure sensor based on dielectric resonators. In: IEEE sensors. IEEE, pp 1–3 17. Alejandro J, Martin S, Matthias N, Rolf J (2017) Hybrid time-frequency modulation scheme for chipless wireless identification and sensing. In: IEEE sensors. IEEE, pp 1–3 18. Jiménez-Sáez A, Schüßler M, Nickel M, Jakoby R (2018) Hybrid time-frequency modulation scheme for chipless wireless identification and sensing. IEEE Sens J 18(19):7850–7859 19. Mandel C, Schüßler M, Nickel M, Kubina B, Jakoby R, Pöpperl M, Vossiek M (2015) Higher order pulse modulators for time domain chipless RFID tags with increased information density. In: European microwave conference (EuMC). IEEE, pp 100–103 20. Preradovic S, Karmakar NC (2010) Chipless RFID: Bar code of the future. IEEE Microw Mag 11(7):87–97 21. Mandel C, Schüßler M, Maasch M, Jakoby R (2009) A novel passive phase modulator based on LH delay lines for chipless microwave RFID applications. In: IEEE MTT-S international microwave workshop on wireless sensing, local positioning, and RFID. IEEE, pp 1–4 22. Schüßler Mn, Mandel C, Maasch M, Giere A, Jakoby R (2009) Phase modulation scheme for chipless RFID-and wireless sensor tags. In: Asia pacific microwave conference. IEEE, pp 229–232 23. Rogers Corporation (2018) RT Duroid 6006 and 6010LM laminate data sheet. https:// www.rogerscorp.com/documents/612/acs/RT-duroid-6006-6010LM-laminate-data-sheet. pdf. Online; accessed 4 June 2018

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24. Sahinoglu Z, Gezici S, Guvenc I (2008) Ultra-wideband po-sitioning systems, vol 2. Cambridge University Press, Cambridge, UK 25. Antonino-Daviu E, Cabedo-Fabres M, Ferrando-Bataller M, Valero-Nogueira A (2003) Wideband double-fed planar monopole antennas. Electron Lett 39(23):1635 26. Nickel M, Mandel C, Schüßler M, Jakoby R (2015) Filter-based slow wave structures for application in chipless microwave RFID. In: German microwave conference (GeMiC). IEEE, pp 68–71 27. Pöpperl M, Dobrev Y, Gottinger M, Mandel C, Jakoby R, Vossiek M (2016) Chipless UWB TDR RFID landmark-based positioning using polarimetric filtering. In: IEEE MTT-S international conference on microwaves for intelligent mobility (ICMIM). IEEE, pp 1–4 28. Ramos A, Perret E, Rance O, Tedjini S, Lázaro A, Girbau D (2016) Temporal separation detection for chip-less depolarizing frequency-coded RFID. IEEE Trans Microw Theory Tech 64(7): 2326–2337 29. Morgan Jr. SP (1949) Effect of surface roughness on eddy current losses at microwave frequencies. J Appl Phys 20(4):352–362 30. Horn AF, Reynolds JW, Rautio JC (2010) Conductor profile effects on the propagation constant of microstrip transmission lines. In: IEEE MTT-S international microwave symposium. IEEE, pp 868–871 31. Coonrod J (2013) Choosing circuit materials for millimeter wave applications. High Freq Electron 12(7):22–30 32. Molla J, Heidinger R, Ibarra A, Link G (1993) Dielectric properties of alumina/zirconia composites at millimeter wavelengths. J Appl Phys 73(11):7667–7671 33. Yu C, Zeng Y, Yang B, Donnan R, Huang J, Xiong Z, Mahajan A, Shi B, Ye H, Binions R, Tarakina NV et al (2017) Titanium dioxide engineered for near-dispersionless high terahertz permittivity and ultra-low-loss. Sci Rep 7(1):1–9 34. Dai J, Zhang J, Zhang W, Grischkowsky D (2004) Terahertz time-domain spectroscopy characterization of the far-infrared absorption and index of refraction of high-resistivity, float-zone silicon. JOSA B 21(7):1379–1386

Chapter 4

Electromagnetic Band Gap (EBG) High-Q Resonator Concepts at mm-Waves

This chapter investigates the use of EBG structures for the design and packaging of single and multiple high-Q resonator tags. EBG structures address the main limitations of single air-cladded high-Q resonators, providing a low-loss support of the same materials as the resonator cavity itself. This shows its benefits not only for the fabrication and handling of the tags in the measurement setup, but also in the design phase, where variations in the EBG surfaces define the resonator’s resonance mode, resonance frequency, and quality factor. This chapter begins by showing the potential of metallic EBGs for realizing single high-Q resonators as well as wireless multiresonator tags in a Bed of Nails (BoN) for low-mm-Wave applications in Sect. 4.1. Afterwards, Sect. 4.2 explores the capabilities of full-dielectric EBGs based on PhCs for mm-Wave and THz applications.

4.1 High-Q Resonators in a Metallic Bed of Nails (BoN) In 1966, Simmons and Kay [1] described that transverse grooves on the horn of a circular antenna could eliminate the E-plane edge currents in the rim of the horn by the creation of a capacitive surface reactance that prohibits the propagation of surface waves. A detailed analysis of these structures can be found in [2], where it is distinguished according to the direction of the corrugation in respect to the propagating wave between transversal corrugations (soft surfaces), and longitudinal corrugations (hard surfaces). In 2009, a novel EBG based on these concepts was presented in [3] where the corrugations are extended from one dimension to 2 dimensions, generating a periodic structure of metal pins that receives the name Bed of Nails. The top of the pins creates an Artificial Magnetic Conductor (AMC) boundary condition, and a stop© The Author(s), under exclusive license to Springer Nature Switzerland AG 2022 A. Jiménez-Sáez, Towards THz Chipless High-Q Cooperative Radar Targets for Identification, Sensing, and Ranging, Springer Theses, https://doi.org/10.1007/978-3-031-04976-7_4

57

58

4 Electromagnetic Band Gap (EBG) High-Q Resonator Concepts at mm-Waves

PEC AMC

hg< λ/4

Fig. 4.1 Cross-section of the BoN with a PEC plate on top of it. The PEC and AMC hinder any wave propagation along their opening as long as h g < λ/4 [4]

band is generated when a parallel Perfect Electric Conductor (PEC) plane is placed at a distance h g < λ/4 from the top of the pins as depicted in Fig. 4.1. Local variations in the otherwise periodic structure result in macroscopic variations of the electromagnetic properties. By realizing artifacts such as grooves or ridges, waveguides can be integrated with potentially lower losses than Substrate Integrated Waveguides (SIWs) or microstrip transmission lines due to the absence of dielectric and the lower conductor losses. Furthermore, compared to standard splitblock hollow-waveguide assemblies, these waveguides do not rely on the electrical contact between two metallic surfaces, therefore relaxing the high accuracy requirements needed at the metal junctions, and giving this group of waveguides the name of gap waveguides. The integration of waveguides in a periodic structure increases its electrical size. Nevertheless, at mm-Wave frequencies, and especially at THz, electrically large structures are often not a limiting factor due to the short wavelengths, i.e., small absolute dimensions, while losses play a major role in the performance and viability of the systems due to the more limited efficiency and maximum power available from the electronic signal generation circuitry. While the bandwidth of these waveguides depends on each topology, it is worth mentioning that stop-band bandwidths higher than an octave can be achieved [3]. For a more in-depth analysis of the stop-band generation and achievable bandwidths through different corrugation structures, the reader is referred to [5]. An example of the lack of galvanic contact between the parts is shown in [6], where the inclusion of additional electrodes for the realization of a liquid-crystal tunable phase shifter is avoided by using the two gap waveguide building blocks as the electrodes. In addition to wave guiding structures, a local defect in one or a few pins creates resonant effects that can be used for the generation of narrow-band responses. The resonance used for the realization of the high-Q resonators is presented in [7], where it is shown that the shortening of a single pin originates a T M010 resonance that can be excited by a groove gap waveguide. The potential for antenna arrays is demonstrated with a single-layer low-mm-Wave 4 × 4 slot array where each of the slots is excited by such a resonance, achieving 2.5 GHz bandwidth and 20 dBi gain [8]. As a result of the compactness of the resonator and the high isolation achieved by the BoN, the feeding network can be integrated into the same layer as the resonators.

4.1 High-Q Resonators in a Metallic Bed of Nails (BoN)

t

hg

wp

59

a

hp

Fig. 4.2 Dimensions of the BoN listed in Table 4.1 [4] Table 4.1 Dimensions of the designed BoN at 37.5 GHz [4] Parameter a wp hp Criterion Value (mm)

0.4 λ0 3.20

0.32 p 1.02

0.25 λ0 2

hg

t

0.05 λ0 0.4

0.5

In the following, the use of this resonance is presented for the realization and wireless readout of high-Q resonators in a BoN structure. Part of this work is based on the author’s Master Thesis [4]. Due to the increase in metal losses at higher frequencies as described in Sect. 3.3, the presented solution is verified in the low-mm-Wave frequency range. The dimensions of the periodic BoN structure in which the resonators are integrated is shown in Fig. 4.2 and Table 4.1. For these dimensions, a stop-band is produced from 24 GHz to 54 GHz, resulting in an 83% bandwidth.1

4.1.1 Single Resonator Design When shortening a single pin from its standard height, the capacitance between that pin and the top plate is reduced, therefore increasing the resonance frequency of one of the resonators that would otherwise contribute to the lower stop-band limit. In addition, the inductance decreases for shorter pins, intensifying the effect. The electromagnetic fields of this T M010 resonance mode are sketched in Fig. 4.3. Adapting the example of a metal cavity with a metal pin on its center from [9, p. 312] to the present case, the resonance frequency shift of the cavity can be described by the equation −2h p wp2 ω − ω0 = , 2 ω0 (h p + h g )leff

1

(4.1)

When defining the relative bandwidths of√wideband systems, the reference or center frequency is calculated by the geometrical mean f 0 = f min · f max .

60

4 Electromagnetic Band Gap (EBG) High-Q Resonator Concepts at mm-Waves

E H

E H (a)

(b)

(c)

Fig. 4.3 E- and H-fields inside the resonant cavity for an a low, and b high center nail. Stronger field intensities are represented by thicker lines. The E-field concentrates between the top plate and the top of the pins, while the H-field encircles the center pin. c Simulated resonance frequency over the pin height and Q u for a bottom aluminum part and a top copper plate. The effect of surface roughness is not considered [4]

where ω and ω0 are the angular resonance frequencies of the cavity with and without perturbation, respectively. h p is the height of the shortened center pin, wp2 is the cross-section of the pin, h p , h g are the height of the pins and the gap between the surface of the pins and the metal plate over them. leff corresponds to the size of an equivalent unperturbed rectangular waveguide cavity resonating at the same resonance frequency. Figure 4.3d shows the resonance frequency over the normalized pin height and the unloaded Q-factor, Q u , for typical materials. Varying the height of the center pin, the resonance frequency can be tuned from 25 GHz to 39.3 GHz. However, the unloaded quality factor decreases with decreasing resonance frequency, i.e., increasing pin height. Finally, a saturation in the resonance frequency shift is observed for lower pin heights, which is attributed to the lower intensity of the E-field between the center pin and the top plate as sketched in Fig. 4.3a. Due to their geometrical and electrical properties, these resonances are similar to coaxial cavities, which are used for applications such as base station filters in mobile communications [10]. Usually, a limiting factor for the realization of standard highQ coaxial cavities for mm-Wave frequencies and above is the required electrical contact between two usually Computer Numerical Control (CNC)-milled blocks, which needs to guarantee good contact over long periods of time and temperature gradients. By fabricating the cavities in a BoN, these tolerance requirements are relaxed, and a more stable operation can be expected.

4.1 High-Q Resonators in a Metallic Bed of Nails (BoN)

61

4.1.2 Multi-resonator Tags In the following, 4-resonator tags are designed with different pin heights, resulting in 4 different resonance frequencies of the same TM010 high-Q resonance mode. Two techniques are investigated for the excitation of the high-Q cavities in gap waveguides. The first one directly excites each of the cavities through a radiating slot in a parallel feed, while the second one excites the resonators through a waveguide connected to a standard horn antenna in a corporate feed.

4.1.2.1

Parallel Feed

First, simulations are realized to study the excitation of each cavity through a slot. The thickness of the copper plate is set to 500 µm or 1/16 · λ0 . Thicker plates approach a size comparable to the wavelength at 37.5 GHz and increase the attenuation due to a more self-resonating slot [11, 12]. On the other hand, thinner copper plates compromise the mechanical stability of the plate. For the bottom part, aluminum is selected since it can be easily CNC-milled, although its conductivity is slightly lower than copper σAl = 0.61σCu as shown in Table 4.2. Assuming flat surfaces, these conductivities are enough to obtain conductor quality factors, Q cond > 1000 in the low-mm-Wave frequency range. Simulations are performed with the slot at different positions. In the lossy simulation, the obtained Q-factor from the 3-dB bandwidth and the resonance frequency in Eq. 2.11 represents the loaded quality factor, Q l . The conductor quality factor, Q cond , shown in Table 4.2 includes the losses and is therefore equal to the unloaded quality factor, Q u . With these two parameters, Q l and Q u , the external quality factor, Q e is calculated by applying Eq. 2.15. The quality factors obtained from these simulations and applying the equation over the slot position can be seen in Fig. 4.4. The external and loaded quality factors tend to increase the higher the distance from the cavity due to the weaker coupling. However, this variation is not monotonous, and shows a periodicity equal to the BoNs period. Q e shows local minima when the slot is placed on top of the pins due to the local increase of the E-field, and increases in the region in-between two pins, where the AMC condition is achieved by the BoN. To create the tag, several resonators are distributed in the BoN surface as shown in Fig. 4.5. In each cavity, the height of the center pin is different and equal to 0.2, 0.15, 0.1, and 0.025 h p . Low heights are selected due to their higher Q u , while the shortest

Table 4.2 Unloaded Q factor Material Conductivity (MS/m) Copper Aluminum Cavity

58 35.6

loss (%)

Qcond

19.2 80.8 100

15,792 3,744 3,026

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4 Electromagnetic Band Gap (EBG) High-Q Resonator Concepts at mm-Waves

Q  < 100

Fig. 4.4 Quality factors over the position of the slot. Q e and Q l,s are simulated, Q u is calculated from Eq. 2.15, while Q l,m is measured [4]

pin is further reduced to compensate the lower resonance frequency sensitivity to pin height variations shown in Fig. 4.3d. On each cavity, two co-polarized slots are placed, so that each resonator can be excited at any polarization. When excited, a resonator with two cross-polarized slots open backscatters through both slots simultaneously. To generate different signatures, no variation is needed in the bottom aluminum layer, since the coding can be changed by the presence or absence of each slot. With this, the number of resonance peaks that can be used for coding are 4. By using both polarizations, up to 8 resonance peaks can be encoded, assuming that the reader is properly oriented to the tag and can transmit and receive interrogation signals with both vertical and horizontal linear polarizations. The tag is fabricated in two parts. The bottom aluminum part is CNC machined, while the slots and screw holes in the 500 µm thick copper plate are fabricated by laser cutting. The tag and measurement setup are shown in Fig. 4.6a, b, while the four resonances can be seen in Fig. 4.7. Despite a slight deviation to higher frequencies, a good agreement between simulations and measurements is observed in Fig. 4.7b. Furthermore, Fig. 4.7c demonstrates that covering one of the slots with ordinary copper tape permits the variation of the signature. The direct excitation of resonators through a slot is a straightforward solution towards tags with a high number of resonators. The use of metals allows for rel-

4.1 High-Q Resonators in a Metallic Bed of Nails (BoN)

pins

63

coupling slots

H

H

X-pol

H f1

f3

f4

f2

H

Fig. 4.5 Top view layout of the high-Q resonators and coupling slots. f 1 corresponds to the lower resonance frequency (higher center pin), and f 4 to the higher resonance frequency (shorter center pin) [4]

(a)

(b)

Fig. 4.6 a Fabricated 4-resonator tag in a BoN with radiating slots. A second top copper layer is fabricated for the characterization of Q l,m over the slot position in Fig. 4.4. b Measurement setup [4]

atively low losses in the low-mm-Wave band and, despite the use of a periodic structure with a large electrical size, the absolute dimensions of the BoN surface with 13 pins × 13 pins are 4.48 cm × 4.48 cm. However, the fabricated prototype with independent excitation of several resonators shows important issues. The main one is the strong variation of the reflected power with the readout angle, which is not equal for all resonances. Even with differential measurements removing background clutter, the structural mode of the metallic

64

4 Electromagnetic Band Gap (EBG) High-Q Resonator Concepts at mm-Waves

(a)

si e sim mulated measured as e

−60 −70

−50 |S11 | (dB)

|S11 | (dB)

−50

code 1 code 2

−60 −70 −80

−80 38

38.5 39 39.5 frequency (GHz) (b)

38

38.5 39 39.5 frequency (GHz) (c)

Fig. 4.7 a Time-frequency plot with a time span of 30 ns. b Good agreement between simulations and measurements. c Example of two different signatures by covering one of the slots with ordinary copper tape

tag greatly affects the measured results for different orientations. This, together with the low RCS of the slots because of their small aperture, causes unreliable readout in dynamic environments. For these reasons, an alternative is investigated in the following, in which all resonators are excited through a single waveguide in a corporate feed.

4.1.2.2

Corporate Feed

The resonators in the BoN can be excited through a groove gap waveguide, whose working principle and propagation mode are similar to a standard rectangular hollow waveguide, but with BoN instead of uniform metallic walls on two of its sides. The resonators are placed two pins away from the waveguide as depicted in Fig. 4.8a and the height of the pins between the waveguide and the resonator is reduced to an 80% of the original value. The waveguide is terminated by a short, so that most of the wideband interrogation pulse is reflected similarly as in the hybrid tag presented in Sect. 3.2.

4.1 High-Q Resonators in a Metallic Bed of Nails (BoN) short

65

resonators

WR-28

1 cm (a)

(b)

Fig. 4.8 a Simulated tag and b fabricated 4-resonator BoN tags in groove gap waveguide technology [4]

When integrating the resonators next to a waveguide, the distance between the resonators and the shorted-end becomes critical. This effect can be modelled considering two excitations that overlap at each resonator. One of the excitations corresponds to the interrogation pulse propagating towards the shorted-end, and the other after the pulse is reflected at the shorted-end and propagates backwards towards the antenna. These two excitations interfere at each resonator, so that they may increase the power inside it when the phase difference is lower than ±120◦ , but also reduce it when it is higher. In the worst case of a destructive interference with 180◦ difference, both excitations cancel out. In most cases, one of the excitations has a lower magnitude than the other, so that, even with a 180◦ phase shift between both excitations, the resonance is still excited with the magnitude difference between the excitations. To favor constructive interference, the distance between each resonator and the shortedend should be an odd multiple of quarter wavelengths (2n − 1)λg−res /4, where n is an integer, and λg−res is the guided wavelength in the groove gap waveguide at the resonance frequency of each resonator. However, the reference position of the short-end in gap waveguides is not as easily determined as in hollow waveguides, where it overlaps with the geometrical position of the metal wall. In addition, each resonator has a different resonance frequency. Therefore, an empirical optimization of each resonator’s position is performed through parameter sweeps, until a stable position for all 4 resonators is found. First, lossless simulations are performed to maximize the interference between the forward and backward excitation, and variations of the resonance peak heights are monitored for different positions of the short-end. After finding a stable position for all resonators, the position of the short-end is fixed, and the position of each resonator is slightly varied for further improvement. Since small variations on the manufacturing tolerances change the propagation constant, the utility of further optimization is limited, and the design effort concentrates on reaching a resonance stable against small manufacturing tolerances by analyzing the stability of the resonance peak heights for different parameter variations. The design is finished by adapting the groove gap waveguide to a WR-28 flange, so that a horn antenna can be integrated. Two different tags are fabricated, one with a nearly constant frequency separation between the resonators, and a second one with 2 groups of 2 densely packed

66

4 Electromagnetic Band Gap (EBG) High-Q Resonator Concepts at mm-Waves

(a)

(b)

Fig. 4.9 Measurement results directly connected to a WR-28 port for the tags with a 4 spaced resonance frequencies, and b 2 groups of 2 densely packed resonance peaks.

resonances to show the potential for integrating further resonances. The measured results of the tag directly connected to a WR-28 port can be seen in Fig. 4.9. In both cases, a low 10 dBi gain horn antenna is installed in the tag as a trade-off between high RCS and wide HPBW. The measurement setup is shown in Fig. 4.10, as well as the time-frequency results of both tags for frontal incidence from 30 cm distance. It can be observed that, in both cases, the 4 resonances can be clearly distinguished. The minimum relative distance between the resonances is 44 MHz or a 0.12 %, demonstrating the possibility to densely pack multiple resonances in a narrow bandwidth.

(a)

(c)

(b)

(d)

Fig. 4.10 Measurement results of the waveguide-coupled resonator approach based on a BoN from 30 cm distance. a E-, and b H-plane normalized RCS patterns with a 10 dBi antenna gain. The maximum RCS is –21 dBm2 , and the –6 dB RCS pattern is 36◦ . Time-frequency plots of c 4 separated resonances, and d 2 groups of 2 densely packed resonances measured from 90◦ incidence

4.1 High-Q Resonators in a Metallic Bed of Nails (BoN)

67

An angular characterization of the monostatic RCS of the tags is performed by rotating them in both, its E-plane and H-plane. Due to the limitation of the turntable to rotations in one plane, the reader’s antenna and the tag are rotated 90◦ to measure the orthogonal plane. The results are shown in Fig. 4.10a–d, showing a maximum RCS of –21 dBm2 and a –6 dB beamwidth of 36◦ in both the E- and H-planes.

4.1.3 Discussion The use of BoN has been demonstrated for the fabrication and packaging of radar targets with multiple high-Q resonators. Despite the 61% conductivity of aluminum compared to copper, and the increase in losses due to surface roughness, high-Q resonators can be manufactured in low-mm-Wave frequencies. Several resonators can be integrated into a single package, extending the high-Q resonator concept from sensing to identification. Little design effort is needed to enclose the high-Q resonators in a convenient prototype. An additional advantage of the use of metal is its intrinsic mechanical robustness, so that the tags are robust against human manipulation and could be used in high-vibration environments. The tags could also be used in high-temperature environments, although the decrease in the conductivity with temperature needs to be considered. While the independent radiation of each resonator through a slot permits the flexible variation of the signature either by changing the top plate or by covering slots with ordinary copper tape, undesired variations of the magnitude received from one or several resonators pose an additional challenge towards the use of such tags in dynamic environments. On the other hand, the controlled excitation of the resonators through a groove gap waveguide integrated in the BoN surface shows more stable performance, while the integration of a low gain antenna in the tag allows for stable successful readouts at longer distances with a maximum RCS of –21 dBm2 . A –6 dB RCS-beamwidth of 36◦ is wide enough to permit an unsophisticated alignment of the tag to the reader, but limits the use of these tags in dynamic environments where the relative orientation between the reader and the tag cannot be predicted. For the waveguide-coupled tag, the realization of 4-resonator tags has been demonstrated with measured loaded Q-factors Q l > 800. Assuming four resonators can be consistently separated with a slot width equal to four times the peak width, Q eff = Q l /4 = 200 along a 10% bandwidth, up to 4845 distinct signatures or 12.24 bit could be encoded according to Eq. 2.29. Several densely packed resonances could be used for a narrowband identification, while one or a few resonators are used for frequency-position coding of sensed physical parameters. Among the sensed physical parameters could be temperature by substituting the center pin of the resonator with a low-loss temperature-sensitive dielectric material, as well as pressure using the top plate as a membrane that approaches the center pin as the pressure on the other side of the plate increases. By increasing

68

4 Electromagnetic Band Gap (EBG) High-Q Resonator Concepts at mm-Waves

the height of most pins to touch the top plate, the deflection of the membrane can be constrained to the area of interest. At higher frequencies, the gradual increase in losses requires the sole use of higher conductivity materials like copper or silver, especially at the regions of the structure surrounding the high-Q resonators. The structure can be fabricated in a low-cost material, either with CNC for prototyping or with injection molding for mass production. Afterwards, good conductivity can be obtained by techniques such as vacuum deposition and electroplating of a few µm of gold [13], or, for even lower losses, silver and a few nm of an antioxidation layer. Due to the low skin depth below 0.5 µm of good conductors at mm-Wave frequencies, such metal-coating is thick enough to contain the surface currents, while the anti-oxidation layer can be very thin and have a low impact in the overall losses. However, this only suffices up to a certain frequency and, as it is shown in the following, dielectric EBGs of some very low-loss materials become increasingly convenient for high-mm-Wave frequencies.

4.2 High-Q Resonators in a Full-Dielectric Photonic Crystal (PhC) This section introduces a full-dielectric EBG structure, the PhC, and focuses on its potential for the realization of full-dielectric tags with multiple high-Q resonators. Already in 1887, Lord Rayleigh described the propagation of waves through a medium endowed with a periodic structure [14]. This nowadays well-known effect is used in devices such as dielectric mirrors, Fabry-Perot filters, and distributed feedback lasers. 100 years later, in 1987, the interest in these structures grew with a publication on structures periodic in more than one dimension [15]. Analogous to the BoN, the alternate repetition of two materials of different dielectric permittivity creates a stop-band effect. The relative width of the stop-band is proportional to the impedance contrast, which is determined by the permittivity difference between the lower and higher permittivity material. Air is therefore commonly used as the low-permittivity (εr ≈ 1) to maximize the contrast and due to its low losses and simple realization by removing parts of the high-permittivity material. Furthermore, waveguides and resonators can also be realized by local variations in the otherwise periodic structure. While the BoN is used as a 2D periodic structure, PhCs are realized in 1D, 2D, and 3D. In this work, we concentrate on 2D PhCs with a single high-permittivity material slab filled with periodic air cylinders on it as shown in Fig. 4.11. It is worth mentioning that 2D PhCs can also be realized by interchanging the high and low permittivity materials, this is, with periodic high-permittivity cylinders surrounded by air. However, this topology needs a carrier material to act as a mechanical support, increasing the fabrication complexity and potentially worsening the performance of the low-loss high-permittivity material for the realization of high-Q resonators. For a more in-depth explanation of PhCs, the reader is referred to [16].

4.2 High-Q Resonators in a Full-Dielectric Photonic Crystal (PhC)

69

Fig. 4.11 2D PhC slab used as the foundation for the design of high-Q mm-Wave and THz tags

By removing a single cylinder in the periodic PhC structure, a resonant cavity is created. An important advantage of PhCs resonators compared to standard aircladded DRs relies on the fact that the quality factor of the resonant cavity can be tuned by varying the position, size, and shape of the surrounding cylinders [17]. This enables the realization of high radiation quality factors Q rad > 10000 can be achieved with low-order resonance modes and relative permittivities from 8 to 15 solving one of the main limitations introduced in Sect. 3.3. Most PhC literature operates in the optical range. Some publications can also be found at microwave frequencies. However, the larger PhC size, and the possibility to achieve similar effects with metallic structures, limit their suitability for most applications at lower frequencies. In recent years, with the development of more efficient THz sources and detectors, different PhC structures are being proposed in the THz range. Some examples include antennas [18], couplers [19], and flexible resonators [20]. In addition, resonators as permittivity sensors have been presented for materials placed over the PhC slab [21], or through a cylinder near a resonator for microfluidic measurements [22]. This section investigates the potential of PhCs for the realization of chipless wireless tags based on high-Q resonators, demonstrating that PhCs can address all the limitations of the single air-cladded resonator approach introduced in Sect. 3.3, while also allowing for the integration of multiple high-Q resonators in a single tag. For this, first, a single-resonator PhC slab is designed and fabricated with different materials in the mid-mm-Wave and high-mm-Wave frequency ranges. Then, a temperature characterization is performed for each considered material. Finally, tags are fabricated and characterized with multiple resonators.

4.2.1 Single Resonator Design To advance towards PhC tags and sensors in the high-mm-Wave range, first it is necessary to design and fabricate a single resonator. Dielectric losses tend to increase with operating frequency, and most materials show acute losses in the high-mm-Wave

70

4 Electromagnetic Band Gap (EBG) High-Q Resonator Concepts at mm-Waves

cs2 cs1

cr

ly/2

lx rodl

E

rodw

p

y x

sups supd

(a)

(b)

Fig. 4.12 a 2D view of the PhC’s triangular lattice, where the Wigner-Seitz unit cell is highlighted in gray. b Designed resonator, including the polarization of the E-field regarding the PhC slab. The values of the parameters are listed in Table A.2 [23]

range. In the following, the design of a high-Q cavity assuming HR-Si as the dielectric (εr = 11.68) is described in more detail based on [23]. Due to its periodicity, the PhC structure can be defined by its Wigner-Seitz unit cell, highlighted in Fig. 4.12a. The slab thickness is set to 725 µm to match available wafer thicknesses while assuring that the slab is thin enough to hinder the appearance of multiple propagation modes. To excite and characterize the resonator, a reliable feeding structure is designed. As a first step towards wireless tags, a non-wireless feed is used to assure the accuracy of the measured data. A dielectric waveguide is designed in the PhC by removing a row of cylinders, and it is terminated by a triangular rod to match the PhC dielectric waveguide to a TE10 WR-10 port to perform the characterization measurements. Although an exponentially tapered rod is the best solution for a certain length, good matching is achieved with a triangular rod of length r odl = 4.75 mm, which is shorter than 1.43 λ0 at its center frequency of 89.8 GHz. Shorter rods show increased reflections due to the abrupt impedance change between the PhC and the waveguide, while longer rods do not significantly improve the response and are more difficult to handle due to their increased aspect ratio and fragility. In addition to the realization of high-Q resonators, waveguides, and transitions, two additional advantages are provided by the PhC: mechanical stability and easier packaging. Despite the presence of air cylinders in the PhC slab, enough material is left to provide mechanical stability of the PhC slabs. Furthermore, the PhC slabs can be contacted on regions where no Electro-Magnetic (EM) energy propagates with no relevant effect on their properties at the operating bandwidth, facilitating its packaging and characterization. Therefore, the low-loss dielectric material is the only lossy material which can limit the performance of the resonator, and the support structure does not need to be modelled in EM simulations, simplifying the design process.

4.2 High-Q Resonators in a Full-Dielectric Photonic Crystal (PhC)

71

The resonator is designed by first removing a cylinder in the structure, and then slightly shifting the three first neighboring cylinders on each side along the horizontal axis of the resonator as shown in Fig. 4.12b. As in other publications [24], an improved excitation of the resonance mode is observed by placing the cavity diagonally to the waveguide. To maximize the radiation quality factor, parameter sweeps are performed, varying the dimensions of the selected surrounding cylinders. An optimization could be performed in CST Studio Suite® , and further variations such as the cylinder radius and additional cylinders could further increase Q rad . However, these extra degrees of freedom are unnecessary for the design of a single PhC cavity with a radiation quality factor, Q rad ≈ 1000, and are not considered for the design. Figure 4.13a shows how the coupling quality factor, Q coup , can be set by varying the distance between the PhC dielectric waveguide and the resonator, i.e., the number of cylinders with little (< 0.2%) influence on the resonance frequency. The S-parameters of the designed resonator with 7 cylinders separation between the two PhC dielectric waveguides simulated with the time-domain Finite Integration Technique (FIT) can be seen in Fig. 4.13b. Calculating the stop-band from the S21 0 dBm2 are possible with small trihedral corner reflector dimensions in the order of centimeters. However, to distinguish between neighboring tags as required for an indoor self-localization system, a distinct response of each tag is needed. An overview of the application scenario and recent advances in the realization of retroreflective tags is presented in [3]. By replacing one of the walls of a corner reflector with a resonator array as shown in Fig. 5.1, only the wave specularly reflected by the resonator array is retroreflected towards the reader. The response of the resonator array is determined by the sum of the contributions of each element. The higher the quality factor, the more frequency dependent the behavior of the resonators around its resonance frequency, increasing the effect of small resonance frequency shifts due to deviations during manufacturing or operation on its RCS. In the following, a mathematical model is developed to study the effect of the Q-factor and resonance frequency shifts on the retroreflective properties of a resonator array.

Fig. 5.1 Sketch of a resonator array integrated into a corner reflector by replacing one of its metallic walls

5.1 Corner Reflector and Resonator Array Integration

95

5.1.1 Q-Factor and RCS of Resonator Arrays The resonator array can be modelled as an antenna array receiving a plane-wave interrogation signal, and (re)transmitting it with a certain pattern which depends on the angle of incidence of the interrogation signal, the reflected angle, and the resonance frequency and quality factor of each resonating element. Considering a linear array, its RCS can be described by adapting Eq. 2.5 σ(θi , θr ) = At,rx (θi ) · G t,tx (θi , θr )

(5.1)

where θi and θr are the incident and reflected angles, respectively. To keep the model simple and concentrate on the effect of the quality factor and manufacturing tolerances on its reflectivity, the following assumptions are made: (i) the resonating elements are omnidirectional, (ii) the separation between elements is equal to half a wavelength d = λ0 /2, and (iii) the interrogation is normal to the array, θi = 90◦ . with these assumptions, At,rx (θi = 90◦ ) can be calculated as At,rx (θi = 90◦ ) =

λ2 · N · sin(θi = 90◦ ). 4π

(5.2)

It can be seen that, by increasing the number of elements, the effective aperture of the array is increased and with it the RCS. In the case of G t,tx (θi , θr ), the analysis is more complex and should account for the quality factor and variations in the resonance frequency among elements. The rest of this section investigates these effects for the case of a linear antenna array. Random variations of the resonance frequency of each resonating element, f n , can 2 ). The resonance frequency shift be modelled with a normal distribution N ( f res , σfres of each resonating element reduces the magnitude, In , and introduces a phase shift, e jαn , of each element. To calculate the Gt,tx (θi , θr ) of such array, the magnitudes and phases of each resonating element at the mean resonance frequency, f res , are analyzed separately, and their contributions are summed up to obtain the gain as: Gt,tx (θi , θr ) =

N −1 ⎲

In · e jαn · e jknd cos(θi ) · e jknd cos(θr ) .

(5.3)

n=0

Since the interrogation angle is normal to the array, i.e., cos(θi ) = 0, the expression can be simplified to Gt,tx (90◦ , θr ) =

N −1 ⎲ n=0

In e jαn e jknd cos(θr ) .

(5.4)

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5 High-RCS Wide-Angle Retroreflective Tags Towards THz

(b)

(a)

Fig. 5.2 Magnitude and phase of a resonating element, depending on its resonance frequency f n compared to the mean resonance frequency of the array f res . Solid lines correspond to the lossless case, while dotted lines assume Q u = 500

Notice that the part of the equation that varies by having elements with different resonance frequencies is In e jαn . When applying the model to high-Q resonators with time-gating, it is considered that the interrogation pulse is finished, i.e., there is no incident wave and the resonators re-radiate the stored energy. In this case, the 1 from the transmission coefficient corresponding to the incident wave disappears, and the transmission coefficient equals the reflection coefficient as introduced in Sect. 2.2. The resonator array can be modeled with the magnitude and phase from Eq. 2.18 as In e jαn = ℾ(wn ) = −

1/Q e . −2 j · ( wwresn − 1) + 1/Q u + 1/Q e

(5.5)

If the element’s resonance frequency deviates from the mean resonance frequency by △ f , then it introduces an undesired phase shift in the retransmitted waves and a varying magnitude as shown in Fig. 5.2. The sum of all contributions becomes: Gt,tx (90◦ , θr ) =

N −1 ⎲

ℾ(wn )e jknd cos(θr ) .

(5.6)

n=0

Similar expressions to Eq. 5.6 are used for antenna arrays in array theory, as well as when calculating the radiation of reflectarrays. In this case, by using Eq. 5.5 to estimate the contribution of each element, the expression accounts for the focusing properties of the array, as well as η Q and the loss of power caused by deviations of the resonating elements from the mean resonance frequency. Notice that, if σfres = 0, ℾ(wn ) = 1 for any n and a specular reflection occurs. Furthermore, when considering a linear array, its radiation is rotationally symmetric along the array axis, which in this case is the z-axis. With this model, the effect of different resonance frequency distributions on the RCS of a linear resonator array can be analyzed by assigning each resonating element 2 ). For the following example, with a realization of the normal distribution N ( f res , σres

5.1 Corner Reflector and Resonator Array Integration

97

Fig. 5.3 G0,t,tx for two realizations of linear arrays consisting of N = 32 resonators, considering a randomly distributed resonance frequency with standard deviation σfres . a For low quality factors, standard deviations of a few percent can be tolerated with a few dB loss in the maximum value at 90◦ . b Highly resonant (Q > 500) antenna arrays are more sensitive to variations

Fig. 5.4 a Average G0,t,tx for Q = 500 for 3 different standard deviations, showing a decrease for higher standard deviations. b Average and standard deviation of the G0,t,tx reduction over the product of the standard deviation and the Q-factor σfres · Q for N = 1000 elements. For low quality factors, standard deviations of a few percent can be tolerated with a few dB loss in G0,t,tx , but the tolerances decrease for the higher the quality factor

a linear array of 32 lossless resonating elements is considered, so that Q u → ∞ and Q e = Q rad = Q. The Gt,tx (90◦ , θr ) patterns of single realizations of antenna arrays with such normally distributed variations are shown in Fig. 5.3. Since each realization shows a different pattern, and only the retroreflectivity θi = θr = 90◦ is of interest, the average Gt,tx (90◦ , 90◦ ) = G0,t,tx in the main direction and its standard deviation calculated over 10000 realizations are shown in Fig. 5.4. Figure 5.4b shows that the average retrodirectivity loss, △G0,t,tx , in dB increases the higher the product of the standard deviation and the quality factors σfres · Q. Such relationship can be approximated empirically from this analytical model by the expression

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5 High-RCS Wide-Angle Retroreflective Tags Towards THz

△G0,t,tx (dB) ≈ a · (1 − e−b·σfres ·Q ),

(5.7)

with a = 13.00, b = 0.269 for σfres · Q < 12. As long as σfres · Q < 2.3, the retrodirectivity loss △G0,t,tx is lower than 6 dB. This model fits any normally distributed variations of each resonator’s resonance frequency that could be originated by material variations, as well as inaccuracies during manufacture and assembly, as long as they can be modelled with a normal distribution. Despite high retrodirectivity losses, △G0,t,tx , with high σfres · Q, the number of elements can be increased to improve the RCS. This solution becomes more suitable the higher the operation frequency due to the miniaturization of the resonating elements and the availability of processing technologies suitable for mass production. However, as the number of elements is increased, considering progressive resonance frequency deviations between elements leads to potential retrodirectivity loss by undesired beam steering. This effect could be originated by temperature gradients, which produce a progressive variation of the resonance frequency between hotter and colder elements and, due to the correspondent progressive phase variation, deviate the main beam from the normal direction. Therefore, this effect is more pronounced the lower the HPBW, i.e., the higher the number of elements. For simplification, continuous temperature gradients are analyzed, and the low temperature sensitivity K T of the Al2 O3 PhC high-Q resonators is assumed, which is equal to –67 ppm ◦ C−1 . Figure 5.5 shows the maximum G0,t,tx depending on the number of elements for constant inter-element temperature gradients δT . As expected, the progressive phase difference originated by the inter-element temperature gradient is small, and its effect is only apparent for a large array of high-Q resonating elements. Interestingly, a saturation in the maximum G0,t,tx is reached for a certain number of elements and inter-element temperature gradients, after which additional elements do not further increase G0,t,tx .

Fig. 5.5 G0,t,tx over the number of elements, N , for different temperature gradients. For each curve, the inter-element temperature gradient, δT , is constant

5.1 Corner Reflector and Resonator Array Integration

99

Fig. 5.6 G0,t,tx over the number of elements, N , for different temperature gradients. For each curve, the total temperature gradient, △T is constant, so that the inter-element temperature gradient, δT , depends on the number of elements and is equal to △T /N . With this representation, the equal weight of Q, δT in Eq. 5.8 can be perceived

A more explanatory representation of the effect of a temperature gradient is shown in Fig. 5.6, where a constant overall temperature gradient, △T , is considered, being the inter-element temperature gradient, δT = △T /N , more pronounced for smaller arrays. In this case, the inter-element temperature gradient is reduced with the number of elements, N , while the HPBW is also reduced. A constant difference between the obtained directivity and the ideal one is observed, i.e., the higher the number of elements, the lower the divergence of the main beam, but also, in the same manner, the lower the HPBW. From this model, it can be empirically seen that △G0,t,tx ∝ △T · Q · K T .

(5.8)

As long as △T · Q · K T < 0.75, the average RCS loss △G0,t,tx is lower than 1dB. For higher 0.75 < △T · Q · K T < 26, the average RCS loss △G0,t,tx ranges from 1 dB to 12.2 dB and can be estimated empirically from this analytical model by the function (5.9) △G0,t,tx ≈ a(1 − e−b·△T ·Q·K T ), with a = 13.07, b = 0.105. It is observed that, as long as the temperature gradients are not high, and the resonators do not have a high temperature sensitivity, these effects can be neglected. However, when planning to increase the RCS of highly resonant targets by dramatically increasing the number of elements, as it is possible at THz frequencies, the limiting effect of progressive resonance frequency shifts must be considered. Based on these results and due to the high design and processing complexity of designing arrays of high-Q resonators with the available low-loss materials at

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5 High-RCS Wide-Angle Retroreflective Tags Towards THz

THz, this solution is not followed in this work to achieve high-Q, high RCS, wideangle tags. In contrast, the use of low-complexity and low-Q resonator arrays for the realization of low-Q, high-RCS, wide-angle tags is introduced in the following, while an approach that allows for the realization of high-Q, high-RCS, wide-angle tags without the limitations of large resonator arrays is presented in Sect. 5.2.

5.1.2 Coding with a Frequency Selective Surface (FSS) The resonator arrays can be realized with an array of single air-cladded DRs as the resonating elements, but the fabrication and precise positioning of each resonator can be tedious and inaccurate, accentuated the higher the operation frequency. Since no high radiation quality factors are intended with this approach, this work proposes the substitution of the array of DRs by an equivalent planar technique with Frequency Selective Surfaces (FSSs). FSS comprehends any thin, usually periodic surface, which shows a frequency-dependent reflection, transmission, or absorption of electromagnetic fields. By integrating a distinct FSS in each trihedral corner reflector, their backscattered response becomes frequency selective and can be distinguished by a reader, leading to the identification of the tag. Some concepts for the realization of frequency-coded retroreflectors include (i) the modulation of the response of dihedral corner reflectors by using PIN diodes at 10 GHz in [4], (ii) the reduction of the profile of dihedral corner reflectors via FSSs that add an extra phase shift also at 10 GHz in [5], and (iii) the patterning of a crossed slot FSS on a fused silica ball lens from 100 GHz to 350 GHz in [6]. The proposed system model can be seen in Fig. 5.7. The advantages of using a trihedral corner reflector are its 3D retroreflective properties, low cost, light weight, and favorable dimensions in the mm-Wave frequency range. Due to the low Qfactor, time-gating is not performed after the excitation of the resonators to isolate their response from the interrogation pulse, but during their excitation. By this, both the reflection and transmission coefficients introduced in Eqs. 2.18 and 2.19 apply. In the following, a stop-band FSS based on a crossed dipole unit cell is characterized and placed in front of the trihedral corner reflector to demonstrate the concept based on [7]. Afterwards, the frequency-dependent RCS is shown with the FSS in front of the triangular trihedral corner reflector, as well as substituting one of its walls.

5.1.2.1

FSS in Front of a Corner Reflector

A crossed dipole is a common FSS based on two orthogonal metal strips. Each metal strip is a shorted dipole, which resonates at its fundamental resonance frequency when its length l ≈ λeff /2, being λeff the effective wavelength. As in the case of ordinary dipole antennas, each metal strip is only resonant for E-fields parallel to the longer side of the strip, and operation in both polarizations is achieved by adding a second

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Fig. 5.7 a Block diagram of the system and b transmitted and received signals by the reader in time and frequency domain [7]

orthogonal metal strip. The measurement shown in Fig. 5.8 and Table A.4 comprises a VNA Agilent Technologies N5222A with two Anritsu 3740A W-Band extensions, two 21 dBi gain horn antennas, and a 32 × 32 FSS on a Rogers RT/Duroid® 5880 highfrequency laminate (in the following, RT5880), with relative permittivity εr = 2.2 obstructing the direct path between the ports, so that |S21 | represents the transmission through the FSS. The difference between the left and right plot in Fig. 5.8a is the polarization of the E-field regarding the rotation plane. Here, the notation Transversal Electric (TE) and Transversal Magnetic (TM) is used to describe the orientation of the fields regarding the FSS. TE polarization corresponds to an E-field normal to the rotation plane, while TM corresponds to an E-field contained in the rotation plane, i.e., an H-field normal to the rotation plane. The multiple curves on each plot correspond to different angles of incidence of the interrogation signal to the FSS, being 0◦ normal incidence to the substrate containing the FSS. It can be seen that the angular response of the FSS for incident angles depends on the polarization, being more stable for TM than for TE incidence. This is attributed

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Fig. 5.8 a Measured transmission coefficient, |S21 |, of the FSS over the angle for several angles and rotations in the TE-plane (left) and TM-plane (right). b Resonance frequency over the incident angle. The FSS is placed between both ports as shown in (c) and a 0◦ angle corresponds to normal incidence to the plane containing the FSS [7]

to the different cross-coupling between contiguous dipoles depending on the polarization of the incident field. For TM polarization, the mutual coupling between the dipoles along the rotation plane is minimum, so that the resonance of each element is less affected by phase variations in the neighboring cells according to e jknd cos(θi ) in Eq. 5.3. Extracting the frequency of the resonance notch for each incident angle in 1◦ steps, Fig. 5.8b is generated. The resonance frequency shift for TM polarization and angles below ±36◦ is 1.1 GHz, a 1.4%. Increasing the considered angular range up to ±44◦ the frequency shift is 4 GHz, a 5.2%. These values are used in the discussion to study the coding capacity achievable with these FSSs. By placing the FSS in front of the trihedral corner reflector, the mechanism leading to retroreflection can be separated into three steps:

5.1 Corner Reflector and Resonator Array Integration

(a)

(b)

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(c)

Fig. 5.9 a Manufactured FSS (left) and trihedral corner reflector (right). b Top view of the tag, the corner reflector is positioned behind the FSS. c Measurement setup for RCS measurements. A standard 21 dBi gain horn antenna is used at the reader (right) and the tag is placed on a turntable and is supported by foam (left). The distance between the reader and the tag is d = 50 cm. CC BY 4.0 [7]

(i) The wave incident in the tag is filtered by the FSS, i.e., only a frequency-coded portion of the interrogation signal propagates through it. (ii) The frequency-coded interrogation signal propagates to the corner reflector and is retroreflected back in the original direction, and therefore it (iii) propagates and is filtered again by the same FSS. If coupling mechanisms between the incoming and reflected waves at the resonator array are neglected, it can be considered that the wave is filtered twice by the FSS. For the characterization of the complete tag, a trihedral corner reflector of triangular cross-section with side length a = 3 cm and a 16 × 16 element FSS are used. A comparison between the trihedral corner reflector without and with a crossed dipole FSS in front of it is characterized in the monostatic measurement setup shown in Fig. 5.9. The result of the characterization is shown in Fig. 5.10a, including the RCS measurement at 90 GHz in Fig. 5.10b. For normal incidence, the specular direction at which the incident wave is reflected at the FSS matches the incident angle, so the FSS itself becomes retroreflective. Both the undesired FSS reflection and the desired coded corner reflector response reach the reader at slightly different times. In Fig. 5.10a, both responses overlap, and the successful decoding might be compromised unless the reflected pulses from the FSS and the trihedral corner reflector can be distinguished from each other in the time-domain signal. This requires additional post-processing steps, and its feasibility depends on the capability of the reader to distinguish them in a dynamic scenario. The performance of the tag is studied at different distances in the laboratory, with no absorbing materials nor differential measurements. To show the potential of such high-RCS retroreflectors, the reflected time-domain signal for different distances is shown in Fig. 5.11, demonstrating a range above 4 m as well as the exponential decay with the fourth power of the distance. The good agreement between the RCS with and without the FSS in front of it shows the low losses of the FSS for frequencies other than its resonance frequency. However, from the previous representations, it is difficult to understand the effects that the presence of the FSS has on the response of the trihedral corner reflector. For

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Fig. 5.10 a Measured frequency domain response for TE- (left), and TM-polarization (right) from 50 cm distance. Notice the relation between the measured reflection of the tag shown in this figure, and the measured transmission of the FSS shown in Fig. 5.8a. For TM-polarization, the notch can be observed at 77 GHz for angles other than 0◦ . b RCS over angle of the triangular trihedral corner reflector with (solid) and without (dashed) the FSS in front of it at 90 GHz [7]

Fig. 5.11 Time-domain response of a corner reflector coded with a crossed-dipole FSS in front of it at distances from 1 m to 4 m in a laboratory environment with line-of-sight and a TE incident angle equal to 12◦ . The reflected pulses by the tags can be easily distinguished from the clutter. Some late clutter reflections due to other furniture and measurement equipment can be seen after 35 ns [7]

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Fig. 5.12 RCS response over the angle for the trihedral corner reflector coded with a FSS in front of it. a Crossed dipole, b crossed slot FSS, and c uncoded trihedral corner reflector without any FSS as a reference

this reason, the reflected frequency responses over time and over angle are shown in Fig. 5.12a. For the angle-frequency plot, the filtered RCS response is obtained by time-gating the S-parameters with a predefined time window to remove undesired early and late clutter. Each resulting RCS vs frequency response is assigned a column according to the relative incident angle of the measurement. This step is repeated for each measured angle until the angle-frequency plot is obtained. Finally, due to the symmetry of the response, only one side is shown, and the other is used for the orthogonal polarization to ease the comparison. It can be seen that the wideband high-RCS reflection of the corner reflector is successfully coded by the crossed-dipole FSS. This result is compared to two other structures. The first one is the complementary FSS, where metallic crossed dipoles become slots of the same dimensions in an otherwise plain ground plane. The second one is the plain trihedral corner reflector with no coding. The results for these configurations are shown in Fig. 5.12b, c, respectively. The expected complementary responses of the crossed dipole and crossed slot FSSs are observed. First, while the resonance frequency is more stable for rotations in the TM-plane of the crossed dipole FSS, the crossed slot tag is more stable in the TE-plane. Similarly, the responses of both tags are inverted, being the highest power

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of the crossed slot where the corner reflector encoded with a resonance notch has a minimum and vice versa. Since a high-RCS over a wideband frequency range is key towards preserving a narrow pulse shape in the time domain and allowing accurate ranging, the corner reflector encoded with a crossed dipole FSS would be preferable. However, there are two main factors limiting the performance of the trihedral corner reflector with a crossed dipole FSS in front of it: 1. Normal incidence problem: performance degradation when the reader is perpendicular to the FSS plane according to Fig. 5.10a. This can be improved by integrating the FSS into the corner reflector as shown in Sect. 5.1.2.2. 2. Low coding capacity: Due to the lower Q, but especially because of the angular dependence of the resonance frequency, a wide bandwidth needs to be assigned to a single retroreflector, reducing m to less than 3 bits in a 10% bandwidth. To improve this, FSSs with orientation-insensitive resonance frequencies can be used, and the concept is introduced in Sect. 5.1.2.4.

5.1.2.2

FSS Integrated Into a Corner Reflector

To address the drawback of the reflection from the FSS reaching the reader, a straightforward alternative is to integrate the FSS into the corner reflector by substituting one of its three metal walls. By doing this, the reflection is always backscattered towards the reader, while its transmission propagates through the tag and into the environment behind it. This working principle is the opposite of the FSS in front of the trihedral corner reflector, so that the responses are complementary, i.e., the response of a tag with a crossed dipole FSS in front of the trihedral corner reflector should be similar to a tag with a crossed slot FSS integrated into it. However, there is a substantial difference in the nomenclature of the rotation planes for the characterization since, in this case, neither the E-field nor the H-field is aligned with the unit cell of the FSS during the rotation angle. For this reason, the rotation planes are not named regarding the FSS as TE and TM, but regarding the polarization of the reader. The H-plane and E-plane correspond to the H-field and the E-field of the plane wave radiated by the reader and parallel to the rotation plane of the tag, as this is the usual nomenclature when characterizing antennas. The RCS of the tags is shown in Fig. 5.13. Although the responses are complementary as expected, a strong variation depending on the polarization is observed. This is due to the incident interrogation angle in the FSS. Differently than the previous case, here the wave is incident with constant 45◦ incidence to for all angles, so that the response is strongly polarization dependent due to the mutual coupling between cells. Despite solving the normal incidence problem, the integrated configuration requires of more orientation-independent FSSs for its successful use as tags for indoor localization environments.

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Fig. 5.13 RCS response over the angle for the trihedral corner reflector coded with a FSS substituting one of its three metal walls. a Crossed dipole, and b crossed slot FSS

5.1.2.3

240 GHz Operation and Scalability

The resonance frequency of the FSS can be shifted to different frequencies by scaling the unit cell. Due to the high accuracies achieved by chemical etching and the only need of a conductor and a low-permittivity carrier substrate, the previously designed FSSs are easy to scale. To prove its scalability, all FSSs have been scaled to work at 240 GHz by reducing the unit cell size by a factor 240/80 = 3. The smallest detail of the design is the line width of the crossed dipole equal to 83 µm. Despite the increase in metal and dielectric losses, a lower Q rad = Q e of the crossed dipole resonators compared to the PhC resonators reduces the resonator losses, ηQ . To avoid substrateinduced frequency-dependent transmission-reflection properties, the thickness of the substrate is kept below λ/4. Notice that the same trihedral retroreflector can be used, since its correct operation is mainly subject to (i) a size greater, and (ii) a surface roughness much lower than the wavelength. Both conditions can be satisfied at 240 GHz with the CNC milled trihedral corner reflector, whose surface roughness is 1.6 µm according to the manufacturer. The monostatic measurement setup consists of a VNA Agilent Technologies N5222A with a Virginia Diodes WR-3 extension (220 GHz to 330 GHz) and a 21 dBi WR-3 horn antenna. The results for the FSS in front of the trihedral corner reflector and the uncoded trihedral corner reflector can be seen in Fig. 5.14. The similarity to the results at 80 GHz without any additional design steps further than scaling down the structure demonstrate the suitability of the approach for the design and fabrication of low-Q retroreflective tags in the upper mm-Wave frequency range. Furthermore, the increase of the RCS with operating frequency by keeping the same corner reflector dimensions due to the reduction of the wavelength is observed. The measured increase of the RCS between 80 GHz and 275 GHz for the trihedral corner reflector with a crossed dipole FSS is 8.5 dB, while the expected increase according to the f 2 relationship would be 9.7 dB. This is a good agreement considering the expected increase in conductor losses at higher frequencies.

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Fig. 5.14 Measured WR3 RCS response over the angle for the a trihedral corner reflector coded with a crossed-dipole FSS in front of it, and b uncoded trihedral corner reflector

5.1.2.4

Orientation-Insensitive Retroreflective Tag Concepts

The resonance frequency shift for TE polarization increases the bandwidth allocation needed per resonance peak. A solution would be to operate solely with TMpolarized interrogations. However, this condition can only be met when the reader interrogates with an E-field parallel to one of the crossed dipoles. Outside these two planes, a purely TM-polarized interrogation is not possible This condition cannot be guaranteed in 3D dynamic environments such as for indoor localization, so that retroreflectors are required with enhanced stability of the frequency coding to a broad angle of interrogation angles. Recently, an approach based on a Master Thesis supervised during the realization of this work has been extended and published at a conference [8]. It aims to reduce the frequency shift with the interrogation angle of FSSs by miniaturizing the unit cell at mid-mm-Wave frequencies, and shows a maximum resonance frequency shift of 2.56%. This reduces the resonance frequency shift, but not the variation of the RCS with the interrogation angle. While the FSS generates the frequency coding by reducing the RCS at certain frequencies, it is the trihedral corner reflector that accounts for the retroreflective response leading to the high-RCS. For this reason, assuming the FSS response is lossless transmission, i.e., transparent, at a certain frequency, the RCS angle dependence is solely defined by the corner reflector. Reduced transmission by losses and resonant effects decreases the RCS, so that the original value can be seen as an upper limit. This effect is observed in the comparison of the trihedral corner reflector with and without FSS in Fig. 5.10b. Figure 5.15 shows the angular variation of the retroreflected power comparing a trihedral corner reflector and a partially metalized Lüneburg lens. A single trihedral corner reflector presents a (i) 10 dB to (ii) 15 dB variation in RCS depending on the angle. Therefore, the maximum readout range achievable for a certain tag at 40◦ is reduced to (i) 56% to (ii) 42% of its value at frontal incidence. This means that, if a certain minimum readout range is required for all angles, the corner reflector side length needs to be (i) 33% to (ii) 54% longer to compensate for these RCS variation.

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Fig. 5.15 Comparison of the angular variation of the RCS over the angle of a Lüneburg lens and a triangular trihedral corner reflector. The contributions from normal incidence at the metal walls are not considered

To reduce the decrease of the RCS with the interrogation angle, other retroreflectors such as lenses can be used. However, the use of lenses as retroreflectors poses new challenges, while also permitting the realization of high-Q high-RCS retroreflectors and is therefore considered in Sect. 5.2.

5.1.2.5

Discussion

Despite limiting the study to a single triangular trihedral corner reflector, up to 8 of them can be mounted to cover the full space around the tag. Therefore, the main limitation of the presented structures is the angular shift for the TE polarization, so that a wider bandwidth needs to be assigned to each resonance peak or resonance notch. An alternative FSS design improving the angular dependency of the resonance frequency is presented in [8]. Recently, this approach has been proposed and studied in different designs. In [9], a dihedral corner reflector and a 2D dielectric resonator array are used to generate a resonance notch in the frequency-domain response. However, using DRs instead of resonator array as the base element or the array poses additional fabrication complexity and increases the potential deviations between resonators, which would reduce the RCS as studied in Sect. 5.1.1. Due to the lower Q-factors, special attention should be paid to the nearby clutter surrounding the tag. Since the identification is encoded by a resonance notch, not only the maximum RCS must be higher than any nearby clutter, but also the SINR, γ, between the trihedral corner reflector response and any clutter must be sufficiently high to permit successful detection of the resonance notch. Despite this seemingly difficult task, the high temporal resolution of wideband mm-Wave systems provides sufficient capabilities to filter any cluttered response separated fractions of a ns, i.e., some centimeters, from the trihedral corner reflector response, as long as the reader can successfully differentiate the resonance peak corresponding to the position of the tag. This is further studied in Sect. 5.4. It is with this assumption where the greatest challenges may lay towards the practical implementation of this approach into clutter-rich environments. The reader needs to find the tag in the room by extracting and analyzing the frequency response of all pulsed responses it finds, and successfully distinguish any cluttered response

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from room or furniture corners from the coded corner reflector response. As an illustrative example, a high-RCS notch-like clutter response can be expected from a corner reflector occurring at the corner between walls and furniture made of wood, εr from 1.5 to 2.5 depending on the density [10], or polymer, εr from 2 to 4, with a quarter wavelength thickness. Finally, the maximum number of bits that can be coded is limited by two properties. First, in their fundamental form shown in this work, each FSS presents a single resonance, i.e., a single resonance peak or resonance notch is used for coding. Secondly, a relatively broad bandwidth needs to be allocated for each resonance frequency. Even without frequency shifts, the low-Q factor creates a broader resonance, which needs a broader bandwidth to be successfully distinguished at the reader, all other variables (distance, transmit power, corner reflector size...) being equal as seen in Sects. 2.4.1 and 2.4.2. Assuming TM polarization and ±36◦ , the resonance frequency shifts around a 1.4%, this translates into Q eff = 71 and m = 7. With a single resonance notch per tag n = 1, this leads to 7 signatures or C = 2.8 bit.

5.2 Lüneburg Lens with Integrated High-Q Coding Particles To enable the realization of high-Q retroreflectors able to outlast nearby clutter responses, the integration of high-Q coding particles into a lens structure is proposed. In contrast to reflectarrays, a lens alters the behavior of the electromagnetic radiation that propagates through it, instead of reflecting it. In this work, we are interested in lenses that concentrate an incident interrogation signal propagating as a plane wave into a focal point, and vice versa. Two different types of lenses are shown in Fig. 5.16. In static environments, the biconvex lens is a valid solution, since it concentrates the incident plane wave in a focal point. By placing a tag as designed in Sect. 4.2 in its focal point, it is excited and re-radiates a frequency-dependent response, which is collimated by the lens towards the reader. Therefore, the RCS of the single PhC slab is enhanced by the lens through (i) a higher received power, and (ii) a more directive retransmission of its frequency-dependent backscattered response. For consistency, in the following the term tag refers to the whole radar target, including the lens. The high-Q PhC slabs, when integrated into a lens, are the component of the tag responsible for the frequency-coding and are referred to as coding particles. Despite the high tag’s RCS, the use of a biconvex lens is not a valid solution when the relative position of the reader in respect to the tag is not known or varies dynamically, since the performance is optimum only for a single tag-reader orientation. This issue can be solved by using spherical or ball lenses. If a material with a relative permittivity equal or slightly lower than 4 such as fused silica is selected for the lens material, the focal point appears at or slightly outside the opposite side of the lens as depicted in Fig. 5.16b. Due to the geometrical symmetry of this type of lenses, ball lenses concentrate a plane-wave interrogation into a focal

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Fig. 5.16 Two conventional lens types: a biconvex lens, focusing a plane wave into a focal point for normal incidence. b Homogeneous lens with εr = 4, having a focal point at its opposite side

point for any interrogation angle. Since the position of the focal point wanders with the interrogation angle, multiple coding particles need to be placed around the lens to enhance the angular coverage of the tag. Recent publications in parallel to this work within Collaborative Research Center/Transregio 196—MARIE have addressed the excitation of DRs by lenses. Alhaj Abbas et al. [11] shows at 5.5 GHz that placing DRs with different resonance frequencies around a homogeneous lens helps create an angle-of-arrival sensor. However, the appearance of self-resonances for a big lens limits the maximum size of the lens possible to distinguish the frequency coding accomplished by the DRs. This is because the impedance mismatch between free space and the lens generates external, but also internal reflections, converting the lens into an electrically large resonator. Higher order resonance modes are excited in the desired frequency bandwidth, which also tend to have higher radiation quality factors the higher the order and, therefore, mask any response from the coding particles. To reduce this, [11] uses a ball lens with a low relative permittivity of εr = 2.22 that presents a focal point at a certain distance of it. Furthermore, the lens diameter used is kept low at 2.16λ0 to avoid the appearance of high-order, high-Q self-resonances. A lens with a similar optical behavior as the homogeneous lens, but matched to free space to avoid reflection at its surface, is the Lüneburg lens. The use of a 2D Lüneburg lens for the excitation of spherical DRs is presented in [12, 13]. Despite these recent publications, two important concerns have not been solved: • Operating frequency: The mentioned publications demonstrate the coupling between a DR and a lens at frequencies below 10 GHz. On the resonator size, no scalable solution is provided for the accurate placement of the resonators around the lens. In addition, the fabrication of the Lüneburg lens relies on drilling a periodic hole pattern of diameter much smaller than the wavelength in a substrate, whose

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high-Q coding tags

Lüneburg lens

(a)

(b)

Fig. 5.17 Lüneburg lens encoded with high-Q coding particles under a normal incidence, and b oblique 50◦ incidence. Depending on the angle, one or a few coding particles are excited

increasing number and maximum tolerable diameter hinder its direct transfer to higher frequencies. • Q-factor: The mentioned publications use DRs with low loaded Q-factors below 60. Despite their increased fabrication complexity, such resonators are not suitable for efficient clutter suppression in dynamic environments through time gating. Therefore, this work investigates and demonstrates for the first time the realization of mm-Wave high-Q, high-RCS, and wide-angle retroreflectors. The design and fabrication of a Lüneburg lens at 80 GHz [14] combined with several 2-resonator PhC slabs PhC slabs as sketched in Fig. 5.17. Finally, a 240 GHz functioning prototype is demonstrated with a 2D DRIE HR-Si tag [15]. Both publications are the result of an ongoing collaboration with the Department of Radio Electronics at Brno University of Technology. The concept of the integration of high-Q coding particles in a Lüneburg lens is part of this work, as well as the high-Q coding particles, which are based on the 2-resonator PhC slabs introduced in Sect. 4.2.2.2. The lens designs were realized by a visiting scientist and are therefore only shortly introduced in this work.

5.2.1 80 GHz Lüneburg Lens Tag The gradient permittivity profile of an ideal Lüneburg lens varies radially with its distance from the center, r , following the expression [16] εr (r ) = 2 −

( r )2 R

,

(5.10)

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113

where εr is the effective relative permittivity and R the lens’ radius. A convenient technique to achieve such complex relative permittivity profiles is the design of subwavelength patterns into one or a few materials to gradually change their permittivity at the desired frequency band. These structures show homogeneous permittivity at the desired frequency band despite their heterogeneity at higher frequencies. If one end of the Lüneburg lens is metalized, its RCS at boresight is described by the equation 4π(π R 2 )2 . (5.11) σLun = λ20 As an example, a Lüneburg lens with a diameter of 6.7 λ0 or 25 mm at 80 GHz can achieve a maximum RCS equal to –6.67 dBm2 , assuming perfect fabrication and no material losses. In this work, the continuous permittivity profile is modelled by 31 perforated RT5880 layers (thickness = 0.787 mm, εr = 2.2, tanδ = 0.0037 characterized from 65 GHz to 110 GHz) stacked on top of each other. Each layer is composed of squared unit-cells of 0.9 mm × 0.9 mm with a cylindrical hole of varying diameter in the center. The diameters of the holes vary along 15 layers from 0.4 mm to 0.85 mm. Applying the effective medium theory to calculate the effective permittivity of an etched hole, effective permittivities from 2 to 1.3 are achieved for the smaller and larger holes, respectively. The relative permittivity of 1.3 differs from 1 as described by the original Lüneburg lens, originating a weak mismatch between the plane wave propagating through air and the contour of the lens. Assuming normal incidence of a plane wave in a dielectric, the reflection coefficient can be calculated as (1 − √ √ εr )/(1 + εr ). For a relative permittivity εr equal to 4 (fused silica), 2.2 (RT5880), and 1.3 (designed edge of the Lüneburg lens), the reflection coefficient equals –9, –14, and –24 dB, respectively. The holes are drilled by a circuit board plotter LPKF ProtoMat S100, and the copper cladding of the substrate is removed with a FeCl3 solution after drilling. Afterwards, all 31 layers are stacked and fixed with 30 µm thick double-sided tape. Since this complex fabrication process might lead to unexpected inaccuracies, the fabricated lens is characterized in an anechoic chamber at the Department of Radio Electronics, Brno University of Technology, with an antenna scanner NSI 700S-30 and a VNA Rohde and Schwarz ZVA67 with Rohde and Schwarz ZVA-Z110E WBand extensions. A standard WR-10 open waveguide flange is used for the excitation of the Lüneburg lens, while a standard 21 dBi gain horn antenna is used as a receiver. A 0.5 mm gap is left between the open waveguide and the Lüneburg lens, and both are electronically rotated in 1◦ steps, obtaining a 24.5 dBi gain and the radiation pattern shown in Fig. 5.18. The measured HPBW is 8◦ in E-plane and 9◦ in H-plane. Furthermore, a 150◦ aluminum cup is placed behind the lens so that the incident plane wave is retroreflected. The maximum measured RCS is equal to –9.3 dBm2 . A more detailed description of the design and characterization of the Lüneburg lens

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(b)

(a)

Fig. 5.18 a Designed Lüneburg lens and b measured radiation diagram when fed with a WR-10 flange [14]

normalized E-field (dB) 0 -6 -12 -18

-∞ (a) CC BY 4.0

(b) CC BY 4.0

Fig. 5.19 Simulated E-field of the Lüneburg lens and the coding particle when excited through a plane wave a at resonance, which can be read out by the reader for identification, and b outside of resonance. Due to the high Q-factor of the resonators, most power is reflected back outside of resonance and can be used for ranging [14]

can be found in [14]. In the following, this work concentrates on the encoding of the lens with high-Q resonators. A 5-element array of the 2-resonator high-Q coding particle in Sect. 4.2.2.2 is designed. The coding particles are repeated in one plane every 18◦ to be as close as possible while assuring the functionality of the resonators. The tag layout and its simulated E-fields can be seen in Fig. 5.19. By placing the rod antennas around the Lüneburg lens, the interrogation signal is focused into one or few of the coding particles, increasing their RCS and maximum reading range, while defects in single coding particles only affect the performance of the tag for certain interrogation angles. The coding particles are manufactured in two materials, drilled RT6010 and LCM alumina, and the monostatic RCS measurement equipment of the frequencycoded Lüneburg lens is performed by placing the tag on the turntable and adjusting its position with foam as shown in Fig. 5.20a. The electric turntable and VNA are controlled with a script to save measurements with 1◦ resolution at 60 cm distance. An empty-room measurement is performed without the lens and subtracted from the measurement results at each angle. A triangular trihedral corner reflector with 3 cm

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Fig. 5.20 Results of the Lüneburg lens tag with RT6010 coding particles. a Time-frequency plot for 0◦ incidence showing the longer ringing of the high-Q resonators, and b angular response with a time-gating start of 5 ns

reader lens

PhC (a)

(b)

(c)

Fig. 5.21 Results of the Lüneburg lens tag with Al2 O3 coding particles. a Measurement setup, b time-frequency plot showing the longer ringing of the high-Q resonators, and c angular response with a time-gating start of 5.2 ns

side length is used as the reference for the RCS calibration. The results for RT6010 and Al2 O3 can be seen in Figs. 5.20 and 5.21, respectively. Compared to the original 2-resonator PhC tags without the lens, the measured RCS is increased by 26 dB, resulting in a potential range increase of up to 4.5 times. However, the angular separation of 18◦ is considerably higher than the 8◦ to 9◦ HPBW of the lens, so that there are angles from which the interrogation signal is not efficiently coupled into any high-Q coding particle, i.e., blind spots. Finally, a characterization of a Lüneburg lens with a single 2-resonator Al2 O3 coding particle is performed at different distances as shown in Fig. 5.22. Timefrequency plots at distances from 1 m to 4 m are performed. The two resonance peaks around 72 GHz are clearly distinguishable up to 2 m. Furthermore, subtle

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Fig. 5.22 Received time-domain signal for four different measurements at distances from 1 to 4 m

higher received power can be perceived at the resonance frequencies up to 4 m, so it can be expected that the resonance frequencies could be read out with an adapted post-processing algorithm. Still, the lower receive power indicates that readout errors would be more probable in a dynamic channel.

5.2.2 240 GHz HR-Si Lüneburg Lens Tag Similarly to the 80 GHz example, a tag design based on a Lüneburg lens and highQ PhC is demonstrated at 240 GHz with HR-Si. Due to the high accuracy of the DRIE process, frequencies in the high-mm-Wave and THz frequency ranges can be targeted. In this case, a single-layer lens is designed similarly as in [17], since a 3D design would increase the manufacturing cost and assembly complexity. The following results have been recently published in [15]. The effective medium theory is applied to calculate the effective permittivity of an etched hole as described in [15]. 19 different hole diameters from 95 µm to 110 µm discretize the theoretically continuous gradient index of the Lüneburg lens, resulting in relative permittivities from 2.2 to 1.4, respectively. This planar version of the Lüneburg lens only collimates the beam in one plane as observed in Fig. 5.23. Nonetheless, a simulated gain of 20.9 dBi is achieved at 240 GHz, with a HPBW of 3.3◦ in the E-plane and 46.7◦ in the H-Plane. The diameter of the lens is 20.4 mm, this is, 16.3λ0 at 240 GHz. With this gain, the maximum RCS according to Eq. 2.5 is –21.23 dBm2 . An array of nine PhC-based 2-resonator high-Q coding particles is designed to be integrated behind the Lüneburg lens every 17.5◦ . The 2-resonator coding particles are designed with the same method described in Sect. 4.2.2.2 and resonate at 237.6 GHz and 243.6 GHz with loaded Q-factors of 480 and 435, respectively. The tags are fabricated with a DRIE process as described in Sect. 4.2.1.1. The lens and the PhC slab are manufactured in two different pieces to ease their fabrication and handling due to the different hole diameters of the lens. Rohacell® HF is used to adjust the height of the tag, which is placed in the electric turntable. For the characterization of the RCS, a metal sphere of 20 mm diameter is used as a reference target. Figure 5.24 shows the time-frequency and angular stability responses of the frequency-coded retroreflector.

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(b)

(a)

Fig. 5.23 a Planar HR-Si Lüneburg lens and b simulated radiation diagram showing the narrow beam of the lens in the E-plane [15]

(a)

2021 IEEE

(b)

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(c)

(d)

(e)

(f)

Fig. 5.24 a Fabricated HR-Si Lüneburg lens with 2-resonator PhC high-Q coding particles as a reflective layer. b Monostatic RCS measurement setup. c Time- and d RCS response at a distance of 30 cm. The e time-frequency and f angle-frequency characterizations are measured at 10 cm to improve the SINR. The angle-frequency plot is generated with a time-gating start of 1.3 ns [15]

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The maximum measured RCS of the lens is –30 dBm2 at 240 GHz. The discrepancy from simulations is expected to come from (i) misalignment between the Lüneburg lens and the coding particles, and (ii) inaccuracies caused by the support structure. Due to the low permittivity of the lens and its flat topology, it is very sensitive to any material in contact with it. Both drawbacks could be solved by fabricating the lens and the coding particle in a single piece and holding it from parts of the PhC structure not contributing to the tag’s operation.

5.2.3 Fused Silica Ball Lens with FSS Recently, an alternative for the realization of high-Q tags by combining a fused silica ball lens and FSSs has been studied in a Master Thesis supervised during the realization of this work and published at a conference [18]. The working principle is sketched in Fig. 5.25. Fused silica shows moderate dielectric quality factors, Q diel of up to 1800 at 80 GHz. At 240 GHz, the losses increase and dielectric quality factors between 600 and 1400 [19] are possible depending on the purity of the material. Due to its relative permittivity εr ≈ 4, a fused silica ball lens concentrates an interrogating signal into a focal point on its opposite side. By placing a conductor at the focal point, the ball lens acts as a retroreflector. Furthermore, due to its large electrical size, the ball lens has a high mode density, whose high-order resonances also have a high Q rad . Therefore, time gating can be used to remove clutter and detect the presence or absence of the tag. However, the high mode density hinders the successful identification of different tags based solely on its self-resonances. Therefore, a distinct frequency coding is achieved by patterning a FSS around the ball lens as shown in Fig. 5.25, so that some frequencies can be filtered before they reach the ball lens, selectively reducing the number of resonances and adding identification capabilities to the tag. The advantages of the tag are (i) enhanced detection due to the clutter suppression enabled by the self-resonating ball lens, (ii) that the spherical patterning of the FSS

Fig. 5.25 Working principle of the fused silica ball lens with a FSS at the front-side facing the reader, and a metallic reflector on the back-side [18]

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reduces the frequency shifts for different interrogation angles, and (iii) a slower decay of the RCS with the interrogation angle than a trihedral corner reflector. However, the additional metallic structures around the ball lens, namely the FSS and the metallic reflector, increase the material losses. Moreover, the number of signatures is still limited by the FSS encoding to around 7 for a 10% bandwidth, as shown in Table 2.3. Furthermore, the patterning of the FSSs on the lens surface is complex. In [18], the fabrication of a crossed-slot FSS has been accomplished using metal additive manufacturing, but its limited resolution currently restricts its use up to the midmm-Wave frequency range. A similar design can be found in [6], where the FSS is metallized with vacuum deposition and chemical etching, but no study is performed of the self-resonances of the lens nor the wideband frequency domain response.

5.2.4 Discussion It has been shown that the range can be greatly improved by placing a high-Q coding particle at the focal point of a Lüneburg lens, while the angles from which it can be read out are increased by repeating high-Q coding particles around the lens. The examples shown in this work demonstrate this concept, but also show that the spacing between the coding particles should be reduced to avoid having blind spots, i.e., angles from which the tag cannot be read out. A method to avoid blind spots would be to blur the focal point of the lens, which can be achieved either by changing the lens, or simply by placing the high-Q coding particles at a different distance from the lens. In both cases, the maximum RCS of the retroreflector is reduced due to a decreasing directivity of the lens. The closer the repetitions of the high-Q coding particles around the lens, the smaller the region for each coding particle, i.e., the higher the maximum directivity of the lens without blind spots. Therefore, the maximum directivity of the lens achievable without blind spots is proportional to the number of high-Q coding particles around it, L, so that the achievable blind-spot-free RCS increases with L 2 . Despite the presented additively manufactured PhC tags at 80 GHz and 240 GHz, none of the lens could be additively manufactured yet. The main technological challenge is the x y resolution of conventional 3D printers like fused filament fabrication, which depends on the print nozzle diameter, and stereolithography, which depends on the laser spot size. Furthermore, with stereolithography, the requirement for support structures and the extraction of the uncured resin from the inner holes of the 3D lens hindered the successful fabrication of a lens. Finally, the available materials often show higher dielectric losses (tanδ > 0.01) in the mm-Wave [20] and THz [21] frequency ranges. The use of LCM for the fabrication of a Lüneburg lens faces several challenges. The first one is the high permittivity of Al2 O3 and the low effective permittivities needed for the lens, which decreases the percentage of high-permittivity material per unit cell and could compromise the mechanical stability of the structure before sintering. This challenge could be softened by using Silica, εr ≈ 4 [19], instead of

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Al2 O3 , εr ≈ 9.5. In addition, the actual laser spot size of 25 µm × 25 µm and the material behavior during debindering and sintering need to be better understood and controlled to enable the processing of finer permittivity gradients.

5.3 Tag Identification and Ranging with a FMCW Radar at 80 and 240 GHz An important barrier towards the use of the presented mm-Wave chipless tags in real-world applications is the availability of suitable, affordable readers. The VNAs and extensions used in this work not only have a very high cost, but they are also bulky and unnecessarily complex for their use as a reader. In the last decade, the cost of mm-Wave components has been reduced by advances in technology and the appearance of consumer-market applications. One that can be highlighted are

Fig. 5.26 Measurement setups at a 80 GHz and b 240 GHz © 2022 IEEE [22]. The distances between the tag and the radar are a 109 cm and b 64 cm for all measurements. Figures c and d compare the response from an uncoded trihedral corner reflector with two different readers, a VNA (left) and these FMCW radars (right)

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automotive radars, which, in their second generation, shifted from 24 GHz to 77 GHz and added complex I/Q baseband architectures to reduce noise level. In the following, the suitability of two FMCW radars that are cheaper, smaller, and more energy efficient than a VNA is demonstrated with some of the frequency-coded high-RCS retroreflectors previously presented. The measurements with FMCW radars have been performed in the Institute for Integrated Systems at Ruhr-University Bochum within the Collaborative Research Center/Transregio 196—MARIE. The 80 GHz radar is presented in [23], while more information on the 240 GHz radar can be found in [24]. The main advantages of these radars when compared to conventional FMCW radars are (i) their higher operating frequency, and (ii) their possibility to obtain both magnitude and phase through I/Q baseband demodulation. The achievable ranging accuracies shown in the publications are ±4.5 µm for the 80 GHz [25] and ±0.5 µm for the 240 GHz [24] FMCW radar. More information on the use of these radars as a cost-effective VNAs is shown in [26], while detailed information on their ranging capabilities can be obtained from [27]. The measurement setup for the characterization of the tags is shown in Fig. 5.26a, b. In these measurements, the FMCW radars are controlled from a computer and timegating can be performed in the same manner as with the measurements from the VNA. Both FMCW radars are operated without calibration, as can be perceived from the measurement results of an uncoded corner reflector in Fig. 5.26c, d. The RCS of the uncoded corner reflector is flat with frequency, as can be seen in the calibrated VNA measurements. The VNA results are performed with WR3 extensions and therefore start at 220 GHz. The strong variations in the received power for the 240 GHz FMCW radar are mainly caused by a transmitted power around 20 dB lower below 210 GHz than at its maximum from 230 GHz to 245 GHz [24]. The corner reflectors are then encoded with a crossed dipole FSS in front of it. The frequency-domain results for different angles are shown in Fig. 5.27, demonstrating the high similarity of these FMCW radar measurements with VNA results.

Fig. 5.27 Measured results of the corner reflector encoded with a crossed dipole FSS in front of it at a 80 GHz, TM polarization and b 240 GHz, TE polarization

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(a)

(b)

Fig. 5.28 Measured time-gated response of the Lüneburg lens with a single Al2 O3 coding particle in a time, and b frequency domain

For all figures, a single measurement is performed, and a △P of around 70 dB is estimated for both radars. Due to the high antenna gain of 35 dBi of the 240 GHz FMCW radar, high maximum ranges of dmax > 6 m can be expected according to Eq. 2.9, assuming γmin = 20 dB and an RCS equal to 3 dBm2 as estimated for the trihedral corner reflector. Furthermore, the SINR could be improved by performing several measurements. Doubling the number of measurements increases the SINR up to 3 dB, assuming a noise-limited response with a coherent oscillator and no vibrations or any other time-dependent variations in the measurement setup. Each measurement takes around 4 ms [23] and 15 ms [24] for the 80 GHz and 240 GHz FMCW radars, respectively. These measurements are possible not only with corner reflectors, but also with the high-Q Lüneburg lens tags. Figure 5.28 shows the time and frequency domain results of the 80 GHz radar at 60 cm with the RT5880 Lüneburg lens described in Sect. 5.2.1 and five 2-resonator Al2 O3 PhC coding particles around it as depicted in Fig. 5.21. Despite the appearance of some additional peaks at other frequencies, the lack of calibration, and the lower SINR than the VNA, both resonance peaks can be distinguished. The possibility to obtain the time-gated frequency response from the FMCW radar bridges the gap between the laboratory measurements previously performed and the potential use of resonating tags in real environments. In the following, the suitability of these tags is studied in cluttered environments.

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5.4 Tag Identification and Ranging in Cluttered Environments at 80 GHz To conclude this work, this section presents measurements with two representative high-RCS tags in five different cluttered environments. The measurements are performed with the VNA at 80 GHz. Two tags are compared: The low-Q tag consists of the corner reflector encoded by a crossed-dipole FSS in front of it, and the high-Q tag consists of the Lüneburg lens encoded by a 2-resonator Al2 O3 PhC slab. The following scenarios are considered for each tag: (i) (ii) (iii) (iv)

An uncoded trihedral corner reflector (clutter) placed next to the tag. An uncoded trihedral corner reflector (clutter) placed 10 cm behind the tag. Tag close to a flat wall under normal incidence. Tag close to a commonly occurring dihedral corner reflector between two walls of the room. (v) Tag close to a commonly occurring trihedral corner reflector between two walls and the ceiling.

In all cases, the distance between the tag and the reader is 1 m. For each scenario, two signals are shown for each of the tags. The first one is a coarse time gating from 4 ns to 12 ns that includes the tag backscattered response as well as clutter. The second one is a fine time gated response that intends to separate the desired response of each tag from clutter. In the case of the corner reflector tag, this response is centered at the main pulse, and the desired response is a resonance notch in the frequency domain response. For the Lüneburg lens tag, the desired readout are the resonance frequencies of the resonators, so that the gated signal can be any section behind the resonators as long as the resonance peaks can be differentiated in the frequency domain. The time gating windows are summarized in Table 5.1. Scenario (i) is shown in Fig. 5.29. Similar clutter reflections could occur in real scenarios in case of bulky support structures where the tag is mounted, so that the response from the support or structural mode overlaps with the tag’s response. Due to the nearly zero range difference between the uncoded corner reflector and the low-Q tag, the resonance notch cannot be differentiated. However, the two resonance peaks of the high-Q tag are clearly differentiated among the 53.22% relative bandwidth of the reader. In scenario (ii), depicted in Fig. 5.30, the 10 cm separation from the main clutter contribution generates a main clutter contribution around 0.67 ns after the contribution of the tag. This separation is sufficient to distinguish the low-Q tag response before clutter as well as the high-Q contribution after it.

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Table 5.1 Time-gating parameters for each scenario Corner reflector and FSS Coarse (ns) Fine (ns) (i) (ii) (iii) (iv) (v)

4.0 to 12.0 4.0 to 12.0 4.0 to 12.0 4.0 to 12.0 4.0 to 12.0

6.8 to 7.2 7.0 to 7.4 7.1 to 7.4 7.0 to 7.4 6.9 to 7.3

(a)

Lüneburg lens Coarse (ns)

Fine (ns)

4.0 to 12.0 4.0 to 12.0 4.0 to 12.0 4.0 to 12.0 4.0 to 12.0

7.7 to 10.7 8.5 to 11.5 8.3 to 11.3 8.5 to 11.5 8.0 to 11.0

(b)

(c) Fig. 5.29 Cluttered measurement scenario (i). An uncoded trihedral corner reflector is placed next to the tag. a Measurement setup, b time-domain response, and c frequency-domain response

Scenario (iii) in Fig. 5.31 shows the highest clutter contribution among all scenarios, even higher than the retroreflective responses in (iv) and (v). In addition, the distance between the tag and the wall is lower than in the previous case and equal to 0.24 ns between the coded response and the clutter contribution from the

5.4 Tag Identification and Ranging in Cluttered Environments at 80 GHz

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Fig. 5.30 Cluttered measurement scenario (ii). An uncoded trihedral corner reflector is placed 10 cm behind the tag. a Measurement setup, b time-domain response, and c frequency-domain response

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(c)

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(d)

Fig. 5.31 Cluttered measurement scenario (iii). a Measurement setup, the tag is placed at a wall as shown in (b). c time-domain response, and d frequency-domain response

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(b)

(c) Fig. 5.32 Cluttered measurement scenario (iv). The tag is placed in front of a commonly occurring dihedral corner reflector between two walls. a Measurement setup, b time-domain response, and c frequency-domain response

wall, corresponding to 0.24 ns·c0 /2 = 1.2 cm distance between the tag and the wall. Nevertheless, facilitated by the wide bandwidth of the VNA, the time time-gating window can be precisely set to be able to distinguish the response of the low-Q tag from clutter. In the case of the high-Q tag, no fine time resolution is needed. Similar results are obtained for the scenarios (iv) and (v) shown in Figs. 5.32 and 5.33, respectively. In both cases, the characteristic resonance notches or resonance peaks from the tag can be differentiated in the time-gated reflected signal. These measurements do not serve as a systematic evaluation of the performance of each tag. For such an evaluation, measurements are needed not only in different environments, but also at several distances, from different orientations, and with varying reader transmit powers and antenna gains. For these evaluations, an automatized readout process is needed, which could be based in a matched filter technique with the expected tag response. However, the realization and verification of these readout techniques is beyond the scope of this project. Nevertheless, the presence or absence

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(c)

Fig. 5.33 Cluttered measurement scenario (v). The tag is placed near a commonly occurring trihedral corner reflector between two walls and the ceiling. Several metallic pipelines further contribute to clutter. a Measurement setup, b time-domain response, and c frequency-domain response

of a resonance notch or a resonance peak in the presented results should be taken as a demonstration of the potential of these tags. Especially for the high-Q tag, the two peaks of the Al2 O3 coding particle can be clearly differentiated in all scenarios and over a wide bandwidth, demonstrating the potential of high-Q resonators for the realization of cooperative radar targets robust against clutter in dynamic environments.

References 1. Freeman A (1992) SAR calibration: an overview. IEEE Trans Geosci Remote Sens 30(6):1107– 1121 2. Yongsheng Z, Chuanrong L, Lingling M, Michael Ying Y, Qi L (2014) Improved trihedral corner reflector for high-precision SAR calibration and validation. In: 2014 IEEE geoscience and remote sensing symposium. IEEE, pp 454–457 3. Jiménez-Sáez A, Alhaj Abbas A, Schüßler M, Abuelhaija A, El-Absi M, Sakaki M, Samfaß L, Benson N, Hoffmann M, Jakoby R, Kaiser T, Solbach K (2020) Frequency-coded mm-wave tags for self-localization system using dielectric resonators. J Infrared, Millim, Terahertz Waves 41(8):908–925 4. Lazaro A, Lorenzo J, Villarino R, Girbau D (2015) Modulated corner reflector using frequency selective surfaces for FMCW radar applications. In: 2015 European microwave conference (EuMC). IEEE, pp 111–114 5. Morrow IL, Morrison K, Finnis M, Whittow W (2015) A low profile retrodirective frequency selective surface for radar earth ob-servation. In: 2015 loughborough antennas and propagation conference (LAPC). IEEE, pp 1–4

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6. Williams RJ, Gatesman AJ, Goyette TM, Giles RH (2014) Radar cross section measurements of frequency selective terahertz retroreflectors. In: Terahertz physics, devices, and systems VIII: advanced applications in industry and defense, vol 9102. International society for optics and photonics, p 91020R 7. Jiménez-Sáez A, Schüßler M, El-Absi M, Al-haj Abbas A, Solbach K, Kaiser T, Jakoby R (2020) Frequency selective surface coded retroreflectors for chipless indoor localization tag landmarks. IEEE Antennas Wirel Propag Lett 8. Sánchez-Pastor J, Jiménez-Sáez A, Schüßler M, Jakoby R (2021) Gridded square-ring frequency selective surface for angular-stable response on chipless indoor localization tag landmarks. In: 2021 15th European conference on antennas and propagation (EuCAP). IEEE 9. Alhaj Abbas A, El-Absi M, Abuelhaija A, Solbach K, Kaiser T (2020) Metallic reflectors with notched RCS spectral signature using dielectric resonators. Electron Lett 56(6):273–276 10. Tanaka S, Shiraga K, Ogawa Y, Fujii Y, Okumura S (2014) Applicability of eflective medium theory to wood density measurements using terahertz time-domain spectroscopy. J Wood Sci 60(2):111–116 11. Alhaj Abbas A, El-Absi M, Abualhijaa A, Solbach K, Kaiser T (2019) Dielectric resonatorbased passive chipless tag with angle-of-arrival sensing. IEEE Trans Microw Theory Tech 67(5):2010–2017 12. Alhaj Abbas A, El-Absi M, Abuelhaija A, Solbach K, Kaiser T (2019) RCS enhancement of dielectric resonator tag using spherical lens. Frequenz 73(5–6):161–170 13. Zhao Y, Weidenmueller J, vom Bögel G, Grab-maier A, Abbas Ali A, Solbach K, JiménezSáez A, Schüßler M, Jakoby R (2019) 2d metamaterial luneburg lens for enhancing the RCS of chipless dielectric resonator tags. In: 2019 Second international workshop on mobile terahertz systems (IWMTS). IEEE, pp 1–6 14. Kadera P, Jiménez-Sáez A, Burmeister T, Lacik J, Schuessler M, Jakoby R (2020) Gradientindex-based frequency-coded retroreflective lenses for mm-wave indoor localization. IEEE Access 15. Kadera P, Jiménez-Sáez A, Schmitt L, Schüßler M, Hoffmann M, Lacik J, Jakoby R (2021) Frequency coded retroreflective landmark for 230 GHz indoor self-localization systems. In: 2021 15th European conference on antennas and propagation (EuCAP). IEEE 16. Larimore Z, Jensen S, Good A, Lu A, Suarez J, Mirotznik M (2018) Additive manufacturing of Luneburg lens anten- nas using space-filling curves and fused filament fabrication. IEEE Trans Antennas Propag 66(6):2818–2827 17. Headland D, Withayachumnankul W, Yamada R, Fujita M, Nagatsuma T (2018) Terahertz multi-beam an-tenna using photonic crystal waveguide and Luneburg lens. APL Photon 3(12):126105 18. Sánchez-Pastor J, Jiménez-Sáez A, Schüßler M, Jakoby R (2021) Frequency-coded spherical retroreflector for wide-angle indoor localization tag landmarks. In: 2021 fourth international workshop on mobile terahertz systems (IWMTS), accepted. IEEE, pp 1–5 19. Afsar MN, Button KJ (1983) Precise millimeter-wave measurements of complex refractive index, complex dielectric permittivity and loss tangent of GaAs, Si, SiO/sub 2/, A1/sub 2/O/sub 3/, BeO, Macor, and Glass. IEEE Trans Microw Theory Tech 31(2):217–223. https://doi.org/ 10.1109/TMTT.1983.1131460 20. Paolella AC, Corey C, Foster D, Desjardins J, Smith C, Walters L (2018) Broadband millimeter wave characterization of 3-D printed materials. In: 2018 IEEE/MTT-S international microwave symposium-IMS. IEEE, pp 1565–1568 21. Duangrit N, Hong B, Burnett AD, Akkaraek-thalin P, Robertson ID, Somjit N (2019) Terahertz dielectric property characterization of photopolymers for additive manufacturing. IEEE Access 7:12339–12347 22. Sánchez-Pastor J, Piotrowsky L, Jiménez-Sáez A, Schüßler M, Pohl N, Jakoby R (2022) Evaluation of chipless RFID indoor landmarks at 80 GHz and 240 GHz using FMCW radars. In: 2022 16th European conference on antennas and propagation (EuCAP), accepted. IEEE, pp 1–5

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Chapter 6

Conclusion and Outlook

This work demonstrates the steps from concept to realization of chipless wireless tags for reliable identification, sensing, and ranging in highly cluttered environments for operation frequencies in three frequency decades, from the low GHz range up to 330 GHz. This work focused on four key aspects of high-Q cooperative radar targets: (1) Investigating high-Q resonator approaches in the microwave range towards real highly cluttered environments by designing and wirelessly characterizing a machine tool temperature sensor at up to 400 ◦ C in a metallic oven. Its operation has been demonstrated in a machine tool, where the resonance frequency as a measure of the tool’s temperature could be read out despite the strong clutter and high angular velocities of up to 10,000 rpm. Furthermore, the extension of the concept to other physical quantities such as pressure has been demonstrated. Last, the integration of the sensor with wideband chipless tags to achieve hybrid modulation has been proved with the realization of a TDR-high-Q resonator tag where identification and sensing can simultaneously operate within the same tag and bandwidth. (2) Transferring the high-Q resonator approaches to higher frequencies, via mmWave and towards THz frequencies, by introducing EBG structures for the realization and packaging of high-Q resonators. While metallic EBGs are suitable for robust high-Q tags in the low-mm-Wave frequency range, the increase in conduction losses makes them unsuitable at higher frequencies, where the lack of high-permittivity, ε > 15, materials also impedes the realization of air-cladded high-Q factors with single low-order DRs. However, using all-dielectric PhCs enables the realization and packaging of low order high-Q resonators within a single material, solving limitations such as the lack of high-permittivity materials, the higher metal losses with frequency, and the additional losses introduced by supporting structures. Furthermore, wide bandwidths free of undesired resonances are obtained for unambiguous identification and sensing. The use of a novel ceramic additive manufacturing process, LCM, is demonstrated for the fabrication of low-loss mm-Wave components by realizing high-Q PhC slabs © The Author(s), under exclusive license to Springer Nature Switzerland AG 2022 A. Jiménez-Sáez, Towards THz Chipless High-Q Cooperative Radar Targets for Identification, Sensing, and Ranging, Springer Theses, https://doi.org/10.1007/978-3-031-04976-7_6

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with Al2 O3 at 80 and 240 GHz with loaded Q-factors over 450. These PhC slabs have been also realized with HR-Si, showing loaded quality factors above 700 at both frequencies, enabled by the extremely low losses of the material at room temperature. Furthermore, Rogers RT/Duroid® 6010.2LM is shown as an alternative for the rapid prototyping of 2D PhC slabs in the mid-mm-Wave frequency range, since it can be processed with more accessible 2D CNC drilling machines. Despite good performance with loaded Q-factors around 300 at 90 GHz and room temperature, its dielectric losses are considerably higher than Al2 O3 and HR-Si. While the Q-factor of Al2 O3 remains constant over a wide temperature range up to at least 250 ◦ C, the one for HR-Si decreases significantly the higher the temperature. Hence, aiming for applications in harsh environments, LCM Al2 O3 is the first choice. Finally, the integration of multiple resonators within a PhC tag is demonstrated with two and three resonator tags on LCM Al2 O3 . To the best of the author’s knowledge, this is the first work that recognizes, studies, and demonstrates the potential of metallic and dielectric EBG surfaces to solve the main challenges faced for the realization of high-Q resonating tags able to operate in dynamic and harsh environments in the mm-Wave and THz frequency ranges. (3) Demonstrating the potential of hybrid modulations for simultaneous identification, sensing, and ranging by showing the possibility to integrate high-Q resonators in other chipless tags. This has been demonstrated for phase-coded TDR tags and Lüneburg lens retroreflectors, pointing that hybrid modulation mechanisms are possible and allow for simultaneous identification, sensing, and ranging within the same tag and bandwidth. (4) Investigating techniques to overcome the trade-off between coverage angle and maximum range with two different methods. The first one is the realization of resonator arrays. A theoretical model shows that, the higher the Q-factor, the tighter the tolerances for the fabrication of the resonator arrays. Furthermore, the realization of arrays of high-Q resonating elements with sub-wavelength element spacing requires air-cladded resonating elements and suffers from the limitations listed in (2). A low-Q, low-cost alternative is presented through the use of FSS for the coloring, i.e., frequency coding of triangular trihedral corner reflectors. Despite not obeying the high-Q principle, i.e., no clutter suppression is possible through time-gating, it is shown that they might represent a low-cost alternative to the realization of tags with two limitations: (i) no clutter reflections at the same time as the tag’s response, and (ii) low coding capacity, i.e., wide bandwidth is required for a small number of distinct signatures. Ranges up to 4 m are verified with a moderate 21 dBi gain monostatic reader. The second solution that overcomes the trade-off between coverage angle and range involves the use of lenses. By using spherical lenses, high-RCS is achieved with single high-Q PhC slabs by placing them at the focal point of the lens. A 3D Lüneburg lens has been processed at 80 GHz with a maximum RCS above –10 dBm2 , demonstrating that ranges above 2 m are possible with a single measurement and a modest reader gain of 21 dBi. Furthermore, a tag has been fabricated based on a 2D Lüneburg lens at 240 GHz with a maximum RCS of –30 dBm2 .

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After the successful realization of resonating tags at 80 and 240 GHz, two further steps have been taken towards the verification of the convenience of these tag concepts for mm-Wave and THz frequencies. The first one are measurements with a FMCW radar, demonstrating that the developed tags are compatible with small, costeffective and highly integrated monostatic readers. The second step is the measurement of the tags under clutter conditions. Here, the robustness of high-Q resonators against different cluttered environments has been proven in the mid-mm-Wave frequency range. Furthermore, it is seen that the wider bandwidths available at these frequencies permit enhanced separation of clutter, so that the response from FSS coded retroreflectors can be distinguished unless the clutter reflection reaches the reader at a close instant. These promising results shown demonstrate the potential of chipless mm-Wave and THz tags for robust identification and sensing, and ranging. The present work has clarified and solved several issues hindering the realization of such tags, but important steps need to be taken towards their practical use. The first is the study of automatized readout methods. In this work, the successful readout is interpreted by the presence of a peak or notch at a certain frequency, which is a certain but suboptimal decoding of the backscattered signal. The standard optimum decoding approach for noise-limited channels based on a matched filter should be developed and compared to other techniques in terms of reliability, energy consumption, and real-time operation in interference-limited scenarios expected in a dynamic system. Furthermore, after the development of one or several automatized readout methods, studies of the reliable detection, identification, and ranging of different tags in several scenarios and under varying conditions must be performed, which are fundamental towards the verification of any wireless system. These two steps would greatly benefit from multi-disciplinary work, which will be possible by collaborations with the Institute of Digital Signal Processing at the University of Duisburg-Essen, as well as the Institute of Digital Communication Systems at the Ruhr University Bochum within the second phase of the Collaborative Research Center/Transregio 196—MARIE. Currently, the materials used for processing the Lüneburg lenses compromise the high-temperature performance of the high-RCS, high-Q tags. To exploit these tags at high-temperatures and long ranges above 0.5 m at high-mm-Wave and THz frequencies without substantially increasing the reader gain, transmit power, or measurement time, either (i) rod antenna arrays to increase the RCS of single PhC tags, or (ii) ceramic lenses need to be processed. (i) suffers from the wide-angle, high-RCS trade-off, while (ii) is currently limited by the accuracy of the manufacturing process. Therefore, collaborations with material scientists in the Institute of Technology for Nanostructures at the University of Duisburg-Essen need to be continued to improve the accuracy and extend the available materials of LCM. Furthermore, the trade-off between the number of coding particles around the lens and RCS has been introduced. Methods to reduce the distance between coding particles around the lens need to be further studied to achieve high-RCS without blind spots. For this, 3D PhCs, whose fabrication is compatible with the LCM process,

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6 Conclusion and Outlook

will help to increase the number of coding particles around the lens, as well as to integrate more high-Q resonators per coding particle. The main advantages of encoding corner reflectors with FSSs are its simplicity and low cost. However, improvements in the coding capacity are needed, which should be approached primarily by reducing the frequency shift in dependence of the incident angle. Furthermore, developing automatized readout methods will help to quantify in what extent the normal-incidence reflection from the FSS in front of the corner reflector disturbs the readout in different situations foreseen in dynamic environments. The realization of multi-resonator PhC slabs has demonstrated its potential for identification and sensing. In the future, the design of identification and sensing tags should be pursued for concrete applications. Beyond the temperature and pressure sensor concepts presented in this work, the miniaturization of the resonators at higher frequencies allows for novel approaches such as the realization of chipless wireless sensors with memory capability through their integration with MEMS. This will be investigated in collaboration with the Institute of Microsystems Technology at the Ruhr University of Bochum. Besides, small resonators could be integrated with different functional surfaces to selectively react to the presence of certain chemical or biological compounds. In short, the use of high-Q resonators can be further investigated to achieve wireless reliable operation of sensors commonly limited to (i) wired-only, or (ii) controlled calibrated wireless operation, specially for its operation in cluttered dynamic environments, and when it is crucial to avoid any hazardous contact between the reader and the sensed phenomena. In summary, this work has addressed the issues related to the feasibility of mmWave and THz chipless high-Q cooperative radar targets for identification, sensing, and ranging. By demonstrating their manufacturability and wireless operation with a superior robustness against cluttered dynamic environments, this work paves the way towards tailored solutions for applications such as indoor localization and wireless sensing.

Appendix A

Tag Dimensions

Hybrid modulation enabled by a high-Q resonator See Table A.1.

Table A.1 Hybrid tag dimensions [1] Description Parameter DRh DRd st wl lm df h lp lf wf ls1 ls2

Dielectric resonator height Dielectric resonator diameter Substrate thickness Microstrip line width, 50 Ohms Monopole length Monopole feed distance Distance monopole—ground plane Monopole feed parallel line length Monopole feed line length Monopole feed line width Substrate width of the tags Distance monopole—end of substrate

© The Editor(s) (if applicable) and The Author(s), under exclusive license to Springer Nature Switzerland AG 2022 A. Jiménez-Sáez, Towards THz Chipless High-Q Cooperative Radar Targets for Identification, Sensing, and Ranging, Springer Theses, https://doi.org/10.1007/978-3-031-04976-7

Value (mm) 4 11.7 0.635 0.567 10.8 5.08 1.66 5 8.95 0.227 80 19.60

135

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Appendix A: Tag Dimensions

High-Q resonators in a full-dielectric Photonic Crystal (PhC) See Tables A.2 and A.3. Table A.2 Single resonator 2-port PhC tag dimensions [2] Parameter Description p cr lx ly cs1 cs2 t rodl rodw

Cylinder period Cylinder radius, 0.3p Length of the PhC in x Length of the PhC in y Shift of the first cylinder (towards) Shift of the second cylinder (away) Wafer thickness Rod length Rod width

Value (µm) 900 255 10522 11700 90 90 725 4750 1048

Table A.3 Cylinder shifts of the 2-resonator PhC tags to achieve different resonance frequencies in 6010.2LM [3] 75 GHz 76 GHz 77 GHz Horizontal shift h s 204.01 (µm) Vertical shift vs1 (µm) 100 Vertical shift vs2 (µm) 100

220

248.65

−10 −10

−48.53 −65.47

Crossed dipole FSS in front of a corner reflector See Table A.4.

Table A.4 Unit cell dimensions [4] Parameter Metal thickness Unit cell width Substrate thickness Dipole length Dipole width

Value (mm) 0.035 2.26 0.127 1.48 0.25

Appendix A: Tag Dimensions

137

References 1. Jiménez-Sáez A, Schüßler M, Nickel M, Jakoby R (2018) Hybrid time-frequency modulation scheme for chipless wireless identification and sensing. IEEE Sens J 18(19):7850–7859. 2. Jiménez-Sáez A, Schüßler M, Jakoby R, Krause C, Meyer F, Vom Bögel G (2018) Photonic crystal THz high-Q resonator for chipless wireless identification. In: 2018 first international workshop on mobile terahertz systems (IWMTS). IEEE, pp 1–5. 3. Burmeister T, Jiménez-Sáez A, Sakaki M, Schüßler M, Sánchez-Pastor J, Benson N, Jakoby R (2021) Chipless frequency-coded RFID tags integrating high-Q resonators and dielectric rod antennas. In: 2021 15th European conference on antennas and propagation (EuCAP). IEEE. 4. Jiménez-Sáez A, Schüßler M, El-Absi M, Al-haj Abbas A, Solbach K, Kaiser T, Jakoby R (2020) Frequency selective surface coded retroreflectors for chipless indoor localization tag landmarks. IEEE Antennas Wirel Propag Lett

Appendix B

Measured Unloaded Q-Factors of Metal Cavities

The following table contains the publications used to prepare Fig. 3.18 shown in Sect. 3.3 (Table B.1).

Table B.1 Measured unloaded Q-factors in literature Cavity type f (GHz) Qu References Pucci et al. [1] Pucci et al. [1] Cassivi et al. [2] Cassivi et al. [2] del Olmo-Olmeda et al. [3] Rahiminejad et al. [4] Rahiminejad et al. [4] Brown et al. [5] Brown et al. [5] Jguirim et al. [6]

Material

Comment

13.4 13.5 21.4

5,883 5,400 542

Copper Copper RT5880

Milling Milling CE

32.2

544

RT5880

CE

40.0

960

Aluminum

Milling

BoN

234.0

336

Gold

MM + gold EP

BoN

284.0

527

Gold

MM + gold EP

BoN Hollow WG Substrateintegrated Substrateintegrated BoN

Hollow WG

24.0

1,117

Gold

MM + gold EP

Hollow WG

38.0

1,163

Gold

MM + gold EP

Hollow WG

89.9

370

Gold

MM + gold EP

WG: WaveGuide; CE: Chemical Etching; MM: Micromachining; EP: Electroplating

© The Editor(s) (if applicable) and The Author(s), under exclusive license to Springer Nature Switzerland AG 2022 A. Jiménez-Sáez, Towards THz Chipless High-Q Cooperative Radar Targets for Identification, Sensing, and Ranging, Springer Theses, https://doi.org/10.1007/978-3-031-04976-7

139

140

Appendix B: Measured Unloaded Q-Factors of Metal Cavities

References 1. Pucci E, Uz Zaman A, Rajo-Iglesias E, Kildal P-S, Kishk A (2013) Study of Q-factors of ridge and groove gap waveguide resonators. IET MicrowS, Antennas Propag 7(11):900–908 2. Cassivi Y, Perregrini L, Wu K, Conciauro G (2002) Low-cost and high-Q millimeter-wave resonator using substrate integrated waveguide technique. In: 2002 32nd European microwave conference. IEEE, pp 1–4 3. del Olmo-Olmeda A, Baquero-Escudero M, Boria-Esbert VE, Valero-Nogueira A, BerenguerVerdú AJ (2013) A novel band-pass filter topology for millimeter-wave applications based on the groove gap waveguide. In: 2013 IEEE MTT-S international microwave symposium digest (MTT). IEEE, pp 1–4 4. Rahiminejad S, Uz Zaman A, Pucci E, Raza H, Vassilev V, Haasl S, Lundgren P, Kildal P-S, Enoksson P (2012) Micromachined ridge gap waveguide and resonator for millimeter-wave applications. Sens Actuators A: Phys 186:264–269

5. Brown AR, Blondy P, Rebeiz GM (1998) Microwave and millimeter-wave highQ micromachined resonators. Int J RF Microw Comput-Aided Eng: Co-Spons Cent Adv Manuf Packag Microw, Opt, Digit Electron (CAMPmode) Univ Color Boulder 9(4):326–337 6. Jguirim N, Dalmay C, Passerieux D, Blondy P (2020) W-band micro-fabricated waveguide band-pass filters. In: 2020 IEEE/MTT-S international microwave symposium (IMS). IEEE, pp 135–138

Curriculum Vitae

Personal Details Name Date of birth Place of birth

Alejandro Jiménez Sáez 29.09.1992 Valencia, Spain

Education 2014–2017

Technische Universität Darmstadt and Universidad Politécnica de Valencia Double Master in – M.Sc., Elektrotechnik und Informationstechnik Best student award, summer term 2017 – M.Eng., Máster Universitario en Ingeniería de Telecomunicaciones Best student award, year 2017–2018

2013–2014

Student exchange, Karlsruher Institut für Technologie (60 ECTS) Elektrotechnik und Informationstechnik

2010–2014

Universidad Politécnica de Valencia Grado en Ingeniería de Tecnologías y Servicios de Telecomunicación

© The Editor(s) (if applicable) and The Author(s), under exclusive license to Springer Nature Switzerland AG 2022 A. Jiménez-Sáez, Towards THz Chipless High-Q Cooperative Radar Targets for Identification, Sensing, and Ranging, Springer Theses, https://doi.org/10.1007/978-3-031-04976-7

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Curriculum Vitae

Employment 2017–2021

Technische Universität Darmstadt Research assistant at the Institute of Microwave Engineering and Photonics

Teaching • Antennas and Adaptive Beamforming—Exercise: Assistant from WS 17/18 to WS 18/19. Responsible from WS 19/20—ongoing. • Project Seminar Advanced µWave Components & Antennas: – WS 17/18: Junaid Arshad and Fouad Tajjiou. Investigation on concepts for the detuning of dielectric resonators in chipless RFID applications – WS 19/20: Fahd Tayanne and Mehdi Ben Moussa. Frequency Selective Surface Coded Retroreflectors for Chipless Indoor Localization Tag Landmarks.

Supervised Student Theses • Prannoy Agrawal: Investigations on a Liquid Crystal Based Ridge Gap Waveguide Phase Shifter, Master Thesis, May 2018 • Junaid Arshad: Investigation of planar retroreflective structures to maximize the radar cross section of chipless passive RFID-tags, Master Thesis, November 2018 • Peter Schumacher: Characterization of Cylindrical Multilayer Dielectric Resonators for Chipless Wireless High-Temperature Sensing, Master Thesis, December 2018 • Nazia Jahan: Investigation on Frequency Selective Surfaces for the Frequency Coding of Retroreflective Structures for Indoor Localization, Master Thesis, July 2019 • Tom Burmeister: Investigation of Photonic Crystal-Integrated Antennas for Indoor Localization Tag Landmarks, Bachelor Thesis, March 2020 • Jesús Sánchez Pastor: Investigation on Frequency Selective Surfaces for the Realization of Orientation Insensitive Indoor Localization Tag Landmarks, Master Thesis, April 2020

Curriculum Vitae

143

References Own Publications and Patents First and Shared First Author 1. Jiménez Sáez A, Valero-Nogueira A, Herranz JI, Rodrigo VM (2015) Ring resonances in groove gap waveguides with application to slot array antennas. In: 2015 IEEE international symposium on antennas and propagation and USNC/URSI national radio science meeting. IEEE, pp 260–261 2. Jiménez Sáez A, Polat E, Mandel C, Schüßler M, Kubina B, Scherer T, Lautenschläger N, Jakoby R (2016) Chipless wireless temperature sensor for machine tools based on a dielectric ring resonator. In: Procedia Eng 168:1231–1236 3. Jiménez Sáez A, Valero-Nogueira A, Her-Ranz JI, Bernardo B (2016) Single-layer cavitybacked slot array fed by groove gap waveguide. In: IEEE Antennas Wirel Propag Lett 15:1402– 1405 4. Jiménez-Sáez A, Schüßler M, Nickel M, Jakoby R (2017) Hybrid time-frequency modulation scheme for chipless wireless identification and sensing. In: 2017 IEEE sensors. IEEE, pp 1–3 5. Jiménez-Sáez A, Schüßler M, Jakoby R, Krause C, Meyer F, Vom Bögel G (2018) Photonic crystal THz high-Q resonator for chipless wireless identification. In: 2018 First international workshop on mobile terahertz systems (IWMTS). IEEE, pp 1–5 6. Jiménez-Sáez A, Schüßler M, Nickel M, Jakoby R (2018) Hybrid time-frequency modulation scheme for chipless wireless identification and sensing. In: IEEE Sens J 18(19):7850–7859 7. Jiménez-Sáez A, Schumacher P, Häuser K, Schüßler M, Binder JR, Jakoby R (2019) Chipless wireless high temperature sensing based on a multilayer dielectric resonator. In: 2019 IEEE Sensors. IEEE, pp 1–4 8. Jiménez-Sáez A, Schüßler M, Krause C, Pandel D, Rezer K, Vom Bögel G, Benson N, Jakoby R (2019) 3D printed alumina for low-loss millimeter wave components. In: IEEE Access 7:40719–40724 9. Jiménez-Sáez A, Schüßler M, Pandel D, Ben-Son N, Jakoby R (2019) 3D printed 90 GHz frequency-coded chipless wireless RFID tag. In: 2019 IEEE MTT-S international microwave workshop series on advanced materials and processes for RF and THz applications (IMWSAMP). IEEE, pp 4–6 10. Jiménez-Sáez A, Alhaj Abbas A, Schüßler M, Abuelhaija A, El-Absi M, Sakaki M, Samfaß L, Benson N, Hoffmann M, Jakoby R, Kaiser T, Solbach K (2020) Frequency-coded mmwave tags for self-localization system using dielectric resonators. J Infrared, Millim, Terahertz Waves 41(8):908–925 11. Jiménez-Sáez A, Schüßler M, El-Absi M, Al-haj Abbas A, Solbach K, Kaiser T, Jakoby R (2020) Frequency selective surface coded retroreflectors for chipless indoor localization tag landmarks. In: IEEE Antennas Wirel Propag Lett 12. Jiménez-Sáez A, Schüßler M, Pandel D, Krause C, Zhao Y, vom Bögel G, Benson N, Jakoby R (2020) Temperature characterization of high-Q resonators of different mate-rials for mmwave indoor localization tag landmarks. In: 2020 14th European conference on antennas and propagation (EuCAP). IEEE, pp 1–5 13. Kadera P, Jiménez-Sáez A, Burmeister T, Lacik J, Schuessler M, Jakoby R (2020) Gradientindex-based frequency-coded retroreflective lenses for mm-wave indoor localization. IEEE Access 14. Nickel M, Jiménez-Sáez A, Agrawal P, Gadallah A, Malignaggi A, Schuster C, Reese R, Tesmer H, Polat E, Wang D et al (2020) Ridge gap waveg-uide based liquid crystal phase shifter. IEEE Access 8:77833–77842

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Curriculum Vitae

Coauthor 1. Burmeister T, Jiménez-Sáez A, Sakaki M, Schüßler M, Sánchez-Pastor J, Benson N, Jakoby R (2021) Chip-less frequency-coded RFID tags integrating High-Q resonators and dielectric rod antennas. In: 2021 15th European conference on antennas and propagation (EuCAP). IEEE 2. Haj Hassan M, Alhaj Abbas A, Jiménez-Sáez A, Mostafa Ahmad A, Sievert B, Schüßler M, Rennings A, Solbach K, Kaiser T, Jakoby R, Sezgin A, Erni D (2020) Passive orbital angular momentum RFID tag based on dielectric resonator arrays. In: Third international workshop on mobile terahertz systems (IWMTS). IEEE, pp 1–4 3. Hassan MH, Sievert B, Svejda JT, Alhaj Abbas A, Jiménez-Sáez A, Mostafa Ahmad A, Schüßler M, Rennings A, Solbach K, Kaiser T et al (2020) OAM mode order conversion and clutter rejection with OAM-coded RFID tags. IEEE Access 8:218729–218738 4. Kadera P, Jiménez-Sáez A, Schmitt L, Schüßler M, Hoffmann M, Lacik J, Jakoby R (2021) Frequency coded retroreflective landmark for 230 GHz indoor self-localization systems. In: 2021 15th European conference on antennas and propagation (EuCAP). IEEE 5. Mandel C, Jiménez-Sáez A, Polat E, Schüßler M, Kubina B, Scherer T, Lautenschläger N, Jakoby R (2017) Dielectric ring resonators as chipless temperature sensors for wireless machine tool monitoring. In: 2017 11th European conference on antennas and propagation (EUCAP). IEEE, pp 3912–3916 6. Sánchez-Pastor J, Jiménez-Sáez A, Schüßler M, Jakoby R (2021) Frequency-coded spherical retroreflector for wide-angle indoor localization tag landmarks. In: 2021 fourth international workshop on mobile terahertz systems (IWMTS), accepted. IEEE, pp 1–5 7. Sánchez-Pastor J, Jiménez-Sáez A, Schüßler M, Jakoby R (2021) Gridded square-ring frequency selective surface for angular-stable response on chipless indoor localization tag landmarks. In: 2021 15th European conference on antennas and propagation (EuCAP). IEEE 8. Schuster C, Schumacher P, Schüßler M, Jiménez-Sáez A, Jakoby R (2017) Passive chipless wireless pressure sensor based on dielectric resonators. In: 2017 IEEE Sensors. IEEE, pp 1–3 9. Schumacher P, Schuster C, Jiménez-Sáez A, Schüßler M, Jakoby R (2018) Passive chipless wireless pressure sensor for Harsh and reflective environments. In: 2018 11th German microwave conference (GeMiC). IEEE, pp 227–230 10. Zhao Y, Weidenmueller J, vom Bögel G, Grabmaier A, Alhaj Abbas A, Solbach K, JiménezSáez A, Schüßler M, Jakoby R (2019) 2D metamaterial luneburg lens for enhancing the RCS of chipless dielectric resonator tags. In: Second international workshop on mobile terahertz systems (IWMTS). IEEE, pp 1–6

Patents 1. Jiménez-Sáez A, Schüßler M, Jakoby R (2019) Identification element and a method for identifying associated objects.DE102017122196A1 2. Mandel C, Schüßler M, Kubina B, Jiménez-Sáez A, Maune H, Jakoby R, Landfried K-C, Hofmann K, Keil F (2018) Bauteil miteiner antenne. DE102016117092A1