Satellite Communications [2 ed.] 047137007x, 9780471370079

Table of contents :
Cover
Half Title Page
Title Page
Copyright
Dedication
About the Authors
Preface
Contents
Chapter 1: Introduction
1.1 Background
1.2 A Brief History of Satellite Communications
1.3 Satellite Communications in 2000
1.4 Overview of Satellite Communications
1.5 Summary
References
Chapter 2: Orbital Mechanics and Launchers
2.1 Orbital Mechanics
Developing the Equations of the Orbit
Kepler’s Three Laws of Planetary Motion
Describing the Orbit of a Satellite
Locating the Satellite in the Orbit
Locating the Satellite with Respect to the Earth
Orbital Elements
Example 2.1.1 Geostationary Satellite Orbit Radius
Example 2.1.2 Low Earth Orbit
Example 2.1.3 Elliptical Orbit
2.2 Look Angle Determination
The Subsatellite Point
Elevation Angle Calculation
Azimuth Angle Calculation
Specialization to Geostationary Satellites
Visibility Test
Example 2.2.1 Geostationary Satellite Look Angles
2.3 Orbital Perturbations
Longitudinal Changes: Effects of the Earth’s Oblateness
Inclination Changes: Effects of the Sun and the Moon
Example 2.3.1 Drift with a Geostationary Satellite
2.4 Orbit Determination
2.5 Launches and Launch Vehicles
Expendable Launch Vehicles (ELVs)
Placing Satellites into Geostationary Orbit
2.6 Orbital Effects in Communications Systems Performance
Doppler Shift
Example 2.6.1 Doppler Shift for a LEO Satellite
Range Variations
Solar Eclipse
Sun Transit Outage
2.7 Summary
References
Problems
Chapter 3: Satellites
3.1 Satellite Subsystems
Attitude and Orbit Control System (AOCS)
Telemetry, Tracking, Command and Monitoring (TTC&M)
Power System
Communications Subsystems
Satellite Antennas
3.2 Attitude and Orbit Control System (AOCS)
Attitude Control System
Orbit Control System
3.3 Telemetry, Tracking, Command, and Monitoring
Telemetry and Monitoring System
Tracking
Command
3.4 Power Systems
3.5 Communications Subsystems
Description of the Communications System
Transponders
3.6 Satellite Antennas
Basic Antenna Types and Relationships
Example 3.6.1 Global Beam Antenna
Example 3.6.2 Regional Coverage Antenna
Satellite Antennas in Practice
3.7 Equipment Reliability and Space Qualification
Space Qualification
Reliability
Redundancy
3.8 Summary
References
Problems
Chapter 4: Satellite Link Design
4.1 Introduction
4.2 Basic Transmission Theory
Example 4.2.1
Example 4.2.2
4.3 System Noise Temperature and G/T Ratio
Noise Temperature
Calculation of System Noise Temperature
Example 4.3.1
Example 4.3.2
Noise Figure and Noise Temperature
Example 4.3.3
G/T Ratio for Earth Stations
Example 4.3.4
4.4 Design of Downlinks
Link Budgets
Link Budget Example: C-Band Downlink for Earth Coverage Beam
4.5 Satellite Systems Using Small Earth Stations
Direct Broadcast TV
Example 4.5.1
4.6 Uplink Design
Example 4.6.1
4.7 Design for Specified C/N: Combining C/N and C/I Values in Satellite Links
Example 4.7.1
Overall (C/N)0 with Uplink and Downlink Attenuation
Uplink and Downlink Attenuation in Rain
Uplink Attenuation and (C/N)up
Downlink Attenuation and (C/N)dn
System Design for Specific Performance
Satellite Communication Link Design Procedure
4.8 System Design Examples
System Design Example 4.8.1
Ku-Band Uplink Design
Ku-Band Downlink Design
Rain Effects at Ku Band
Summary of Ku-Band Link Performance
System Design Example 4.8.2 Personal Communication System Using Low Earth Orbit Satellites
Inbound Link: Mobile Terminal to Gateway Station
Mobile Terminal to Satellite Link
Satellite to Gateway Station Link
Outbound Link
Downlink C/N Budget
Optimizing System Performance
Link Margins with FEC
Rain Attenuation at Ku Band
Path Blockage at L-Band
Summary of L-band Mobile PCS System Performance
4.9 Summary
References
Problems
Chapter 5: Modulation and Multiplexing Techniques for Satellite Links
5.1 Frequency Modulation
Waveform Equation for FM
Bandwidth of FM Signals: Carson’s Rule
Baseband S/N Ratio for FM Signals
Pre-emphasis and de-emphasis
Pre-emphasis
5.2 Analog FM Transmission by Satellite
Television Signals
S/N Ratios for FM Video Transmission
Example 5.2.1
FM Threshold
SCPC FM Links
Example 5.2.2
Data Transmission Using Analog FM Channels
Example 5.2.3
5.3 Digital Transmission
Baseband Digital Signals
Baseband Transmission of Digital Data
Band-pass Transmission of Digital Data
Example 5.3.1
Example 5.3.2
Transmission of QPSK Signals through a Bandlimited Channel
Example 5.3.3
Example 5.3.3
5.4 Digital Modulation and Demodulation
Terminology
Modulation and Coding
Bit and Symbol Error Rates
Binary Phase Shift Keying (BPSK)
Probability of a Symbol Error
BPSK Bit Error Rate
QPSK Bit Error Rate
Example 5.4.1
Example 5.4.2
Generation of Quadrature Phase Shift Keying (QPSK) Signals
QPSK Variants
5.5 Digital Transmission of Analog Signals
Sampling and Quantizing
Nonuniform Quantization: Compression and Expansion
Signal-to-Noise Ratio in Digital Voice Systems
Digital Television
5.6 Time Division Multiplexing
TDM Terminology: The U.S. T1 24-Channel System
Other TDM Systems
Channel Synchronization in TDM
5.7 Summary
References
Problems
Chapter 6: Multiple Access
6.1 Introduction
6.2 Frequency Division Multiple Access (FDMA)
Intermodulation
Intermodulation Example
Calculation of C/N with Intermodulation
Example 6.2.1 Power Sharing in FDMA
Example 6.2.2 Channel Capacity with Demand Access FDMA
6.3 Time Division Multiple Access (TDMA)
Bits, Symbols, and Channels
TDMA Frame Structure
Example 6.3.1 TDMA in a Fixed Station Network
Reference Burst and Preamble
Unique Word
Guard Times
Synchronization in TDMA Networks
Transmitter Power in TDMA Networks
Example 6.3.2 TDMA in a VSAT Network
Example 6.3.3 TDMA in a Fixed Earth Station Network
Satellite Switched TDMA
6.4 Onboard Processing
Baseband Processing Transponders
Satellite Switched TDMA with Onboard Processing
6.5 Demand Access Multiple Access (DAMA)
Example 6.5.1 FDMA-SCPC-DA
6.6 Random Access (RA)
6.7 Packet Radio Systems and Protocols
6.8 Code Division Multiple Access (CDMA)
Spread Spectrum Transmission and Reception
DS-SS CDMA Capacity
Example 6.8.1 CDMA in a Fixed Earth Station Network
Example 6.8.2 CDMA in an LEO Satellite Network
Example 6.8.3 GPS
6.9 Summary
References
Problems
Chapter 7: Error Control for Digital Satellite Links
7.1 Error Detection and Correction
7.2 Channel Capacity
7.3 Error Control Coding
Example 7.3.1
Linear and Cyclic Block Codes
Golay Codes
7.4 Performance of Block Error Correction Codes
7.5 Convolutional Codes
7.6 Implementation of Error Detection on Satellite Links
Example 7.6.1
7.7 Concatenated Coding and Interleaving
7.8 Turbo Codes
7.9 Summary
References
Problems
Chapter 8: Propagation Effects and their Impact on Satellite–Earth Links
8.1 Introduction
8.2 Quantifying Attenuation and Depolarization
Example 8.2.1
8.3 Propagation Effects that Are Not Associated with Hydrometeors
Atmospheric Absorption
Cloud Attenuation
Tropospheric Scintillation and Low Angle Fading
Faraday Rotation in the Atmosphere
Ionospheric Scintillations
8.4 Rain and Ice Effects
Characterizing Rain
Rain Climate Maps
Rainfall Rate Exceedance Contour Maps
Raindrop Distributions
8.5 Prediction of Rain Attenuation
Example 8.5.1
Example 8.5.2
Calculation of Long-Term Statistics for NGSO Systems
Scaling Attenuation with Elevation Angle and Frequency
Cosecant Law
Example 8.5.3
Squared Frequency Scaling Law
Example 8.5.4
ITU-R Long-Term Frequency Scaling of Rain Attenuation
8.6 Prediction of XPD
Canting Angle
Tilt Angle
Example 8.6.1
Example 8.6.2
Ice Crystal Depolarization
Rain Effects on Antenna Noise
Example 8.6.3
8.7 Propagation Impairment Countermeasures
Attenuation
Power Control
Signal Processing
Diversity
Depolarization
8.8 Summary
References
Problems
Chapter 9: VSAT Systems
9.1 Introduction
9.2 Overview of VSAT Systems
9.3 Network Architectures
One-Way Implementation
Split-Two-Way (Split IP) Implementation
Two-Way Implementation
9.4 Access Control Protocols
Delay Considerations
9.5 Basic Techniques
Multiple Access Selection
Signal Formats
Modulation, Coding, and Interference Issues
9.6 VSAT Earth Station Engineering
Antennas
Transmitters and Receivers
9.7 Calculation of Link Margins for a VSAT Star Network
9.8 System Design Procedure: Example 9.1
Description of System
System Parameters
Preliminary Calculations
Link C/N Ratios
Inbound Links
Inbound Links with 270 Channels
Outbound Links
System Analysis
9.9 Some New Developments
9.10 Summary
References
Problems
Chapter 10: Low Earth Orbit and Non-Geostationary Satellite Systems
10.1 Introduction
10.2 Orbit Considerations
Equatorial Orbits
Inclined Orbits
Elliptical Orbits
Molniya Orbit
Radiation Effects
Sun Synchronous Orbit
10.3 Coverage and Frequency Considerations
General Aspects
Frequency band
Elevation Angle Considerations
Number of Beams per Coverage
Off-Axis Scanning
Determination of Optimum Orbital Altitude
Radiation Safety and Satellite Telephones
Projected NGSO System Customer Service Base
10.4 Delay and Throughput Considerations
10.5 System Considerations
Incremental Growth
Interim Operations
Replenishment Options
End-to-End System Implementation
10.6 Operational NGSO Constellation Designs
Ellipso
Globalstar
New ICO
Iridium
Orbcomm
Skybridge
Teledesic
Example 10.6.1 System Design
10.7 Summary
References
Problems
Chapter 11: Direct Broadcast Satellite Television and Radio
11.1 C-Band and Ku-Band Home Satellite TV
11.2 Digital DBS TV
11.3 DBS-TV System Design
11.4 DBS-TV Link Budget
11.5 Error Control in Digital DBS-TV
11.6 Master Control Station and Uplink
11.7 Installation of DBS-TV Antennas
11.8 Satellite Radio Broadcasting
11.9 Summary
References
Chapter 12: Satellite Navigation and the Global Positioning System
12.1 Introduction
12.2 Radio and Satellite Navigation
12.3 GPS Position Location Principles
Position Location in GPS
GPS Time
12.4 GPS Receivers and Codes
The C/A Code
12.5 Satellite Signal Acquisition
12.6 GPS Navigation Message
12.7 GPS Signal Levels
12.8 Timing Accuracy
12.9 GPS Receiver Operation
12.10 GPS C/A Code Accuracy
Dilution of Precision: HDOP, VDOP, and GDOP
12.11 Differential GPS
12.12 Summary
References
Problems
Appendix A: Decibels in Communications Engineering
Appendix B: FDM/FM/FDMA Analog Telephone Transmission
Baseband Voice Signal
Voice Signal Multiplexing
Frequency Modulation with Multiplexed Telephone Signals
Bandwidth Calculation for FDM/FM Telephone Signals
Telephone Performance Specifications
Practical Examples
Example B.1
Example B.2
References
Appendix C: Complementary Error Function erfc(x) and Q Function Q(z)
Equivalence Formulas and Tables of Values
References
Appendix D: The Simple Attenuation Model
Example D.1
References
Glossary on Terms and Acronyms
Index
Back Cover

Citation preview

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Pratt Bostian Allnutt

SATELLITE COMMUNICATIONS

FOR SALE ONLY IN :

Second Edition

SECOND EDITION

SATELLITE

COMMUNICATIONS Timothy Pratt Charles Bostian Jeremy Allnutt

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Satellite Communications

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Satellite Communications Second Edition

Timothy Pratt Charles W. Bostian Department of Electrical & Computer Engineering Virginia Polytechnic Institute and State University

Jeremy E. Allnutt Department of Electrical & Computer Engineering George Mason University

John Wiley & Sons

Satellite Communications Second Edition Authorized reprint by Wiley India Pvt. Ltd., 4436/7, Ansari Road, Daryaganj, New Delhi – 110002. Copyright © 2003 by John Wiley & Sons, Inc. All rights reserved. All rights reserved. AUTHORIZED REPRINT OF THE EDITION PUBLISHED BY JOHN WILEY & SONS, INC. No part of this book may be reproduced in any form without the written permission of the publisher. Limits of Liability/ Disclaimer of Warranty: The publisher and the author make no representations or warranties with respect to the accuracy or completeness of the contents of this work and specifically disclaim all warranties, including without limitation warranties of fitness for a particular purpose. No warranty may be created or extended by sales or promotional materials. The advice and strategies contained herein may not be suitable for every situation. This work is sold with the understanding that the publisher is not engaged in rendering legal, accounting, or other professional services. If professional assistance is required, the services of a competent professional person should be sought. Neither the publisher nor the author shall be liable for damages arising herefrom. The fact that an organization or website is referred to in this work as a citation and/or a potential source of further information does not mean that the author or the publisher endorses the information the organization or website may provide or recommendations it may make. Further, readers should be aware that Internet websites listed in this work may have changed or disappeared between when this work was written and when it is read. Wiley also publishes its books in a variety of electronic formats. Some content that appears in print may not be available in electronic books. For more information about Wiley products, visit our website at: www.wiley.com. Authorized India Edition ISBN: 978-81-265-0833-4 ISBN: 978-93-88991-29-2 (ebk)

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Dedication To our wives: Maggie, Frieda, and Norma

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ABOUT THE AUTHORS

Timothy Pratt is a professor in the Bradley Department of Electrical and Computer Engineering at Virginia Tech, where he has been a faculty member since 1981. He received his B.Sc. and Ph.D. degrees in electrical engineering from the University of Birmingham, United Kingdom, and has taught communications subjects in the United Kingdom and the United States. His research interests are in satellite communications, position location, and avionics. Dr. Pratt is a senior member of the IEEE and a member of the IEE (London). Charles W. Bostian is Clayton Ayre Professor of Electrical and Computer Engineering at Virginia Tech where he has been a faculty member since 1969. His primary research interests are in the areas of wireless communications and radiowave propagation. He is coauthor of the Wiley Text, Solid State Radio Engineering, published in 1980. Professor Bostian received his degrees in electrical engineering from North Carolina State University and is a fellow of the IEEE. Jeremy E. Allnutt is a professor in the Electrical and Computer Engineering Department of George Mason University and Director of the MS in Telecommunications Program. His primary interest is radiowave propagation effects on satellite links, which he pursued at research establishments in England and Canada, before working at INTELSAT in the United States from 1979 to 1994. Prior to joining George Mason University in 2000, he was a professor at the University of York, England, and at Virginia Tech. Dr. Allnutt obtained his B.Sc. and Ph.D. in Electrical Engineering from Salford University, England, and is a Fellow of IEE and Senior member IEEE.

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PREFACE

There have been many new developments in satellite communications technology since the first edition of this text was published in 1986. However, the underlying principles of the transmission of radio signals via satellites remain the same. Thus the basic material in the textbook relating to satellite orbits, look angles, transponders on communications satellites, link budget calculations, SN and BER for analog and digital links, multiple access techniques, error control, and the propagation of radio waves through the earth’s atmosphere remains as important now as in 1986. What has changed is that new applications have been developed for satellite communication systems, and new satellites and terminals have been built to implement the new systems. The second edition of Satellite Communications makes no attempt to describe all of the satellite systems now in operation. That would require an encyclopedia. The text concentrates on the principles of satellite communication systems with the aim of providing the reader with a sound understanding of how a satellite communication system successfully transfers information from one earth station to another. The first edition of this text was written by Charles Bostian and Tim Pratt to support the courses we taught on satellite communication. The book found wide popularity, both as a text for students in senior year or beginning graduate courses at universities, and as a basic reference for practicing engineers. In the second edition, we are honored to be joined by our friend and colleague Jeremy Allnutt, with whom we have worked on satellite systems for over 25 years. He contributed the chapters on orbital mechanics, propagation, nongeostationary satellite systems and VSAT networks. Much material that was included in the first edition has been omitted in the second to make way for chapters covering VSAT systems, LEO and NGSO systems, direct broadcast television, and satellite navigation. The advent of personal communications via low earth orbit (LEO) satellites was not anticipated when the text was written in 1984, nor the development of direct broadcasting from satellites using digital transmission. The growth of very small aperture terminal (VSAT) systems has also occurred since 1986, and has led to application of many of the techniques discussed in the first edition. The Global Positioning System (GPS) has become the dominant radio navigation aid, using a constellation of 24 satellites to provide accurate position location everywhere on earth. Perhaps the greatest change in technology over the past fifteen years has been the transition from analog to digital transmission techniques. The transition is almost complete in the United States, with only the distribution of video signals to cable TV head ends remaining as a last bastion of analog transmission. The section in Chapter 5 of the first edition that covered FDM/FM/FDMA systems has been retained as an appendix because such systems continue in operation in some parts of the world. The emphasis throughout the text is on digital transmission techniques; Chapter 5 reviews the basic theory of digital radio transmission, which is fundamental to all digital satellite systems. In parallel with the transition to digital satellite transmission, great changes have occurred in terrestrial communication systems. Optical fibers were just starting to come into use in 1986, and the Internet was still in its infancy. Cellular telephones were barely in use. Many of the developments in terrestrial communication systems have carried over to satellite systems, and much of the technology that was new in 1986 has now matured and has been well described elsewhere. ix

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PREFACE

Within the United States, satellites are not used for telephone (voice) links. The development of terrestrial optical fiber links has made satellite transmission of telephone traffic uneconomic, and the delay associated with GEO satellite link is a nuisance. Domestic satellites serving the United States now carry video signals for distribution to cable TV companies or direct to homes and serve networks of VSAT stations linked to central hubs in major cities. The development of direct to home satellite broadcast television (DBS-TV) has had a major impact on the marketplace. In the United States, digital DBS-TV transmissions are now received in 15 million homes (2001 figure), and in Europe a similar number of homes receive satellite television programming. Video distribution, to cable companies and direct to home, accounts for more than half of all the worldwide earnings from satellite communication systems. The authors would like to thank their colleagues and students who, over the years, have made many valuable suggestions to improve this text. Their advice has been heeded, and the second edition of Satellite Communication is the better for it. Many more worked examples have been added to the second edition to illustrate how calculations are carried out for each topic.

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CONTENTS

Preface 1.

ix Introduction

1

1.1 Background 1 1.2 A Brief History of Satellite Communications 1.3 Satellite Communications in 2000 6 1.4 Overview of Satellite Communications 15 1.5 Summary 16 References 16

3

2.

Orbital Mechanics and Launchers 17

2.1

Orbital Mechanics 17 Developing the Equations of the Orbit 17 Kepler’s Three Laws of Planetary Motion 22 Describing the Orbit of a Satellite 23 Locating the Satellite in the Orbit 25 Locating the Satellite with Respect to the Earth 27 Orbital Elements 29 Example 2.1.1 Geostationary Satellite Orbit Radius 29 Example 2.1.2 Low Earth Orbit 29 Example 2.1.3 Elliptical Orbit 30 Look Angle Determination 30 The Subsatellite Point 31 Elevation Angle Calculation 32 Azimuth Angle Calculation 34 Specialization to Geostationary Satellites 35 Visibility Test 36 Example 2.2.1 Geostationary Satellite Look Angles 36 Orbital Perturbations 38 Longitudinal Changes: Effects of the Earth’s Oblateness 39 Inclination Changes: Effects of the Sun and the Moon 40 Example 2.3.1 Drift with a Geostationary Satellite 42 Orbit Determination 42 Launches and Launch Vehicles 43 Expendable Launch Vehicles (ELVs) 44 Placing Satellites into Geostationary Orbit 48 Orbital Effects in Communications Systems Performance 49 Doppler Shift 49 Example 2.6.1 Doppler Shift for a LEO Satellite 50 Range Variations 51 Solar Eclipse 51 Sun Transit Outage 53

2.2

2.3

2.4 2.5

2.6

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2.7 Summary References 54 Problems 55

54

3.

57

Satellites

3.1

Satellite Subsystems 57 Attitude and Orbit Control System (AOCS) 57 Telemetry, Tracking, Command and Monitoring (TTC&M) Power System 59 Communications Subsystems 59 Satellite Antennas 59 3.2 Attitude and Orbit Control System (AOCS) 60 Attitude Control System 60 Orbit Control System 66 3.3 Telemetry, Tracking, Command, and Monitoring 68 Telemetry and Monitoring System 68 Tracking 68 Command 70 3.4 Power Systems 71 3.5 Communications Subsystems 72 Description of the Communications System 72 Transponders 75 3.6 Satellite Antennas 80 Basic Antenna Types and Relationships 80 Example 3.6.1 Global Beam Antenna 82 Example 3.6.2 Regional Coverage Antenna 83 Satellite Antennas in Practice 83 3.7 Equipment Reliability and Space Qualification 87 Space Qualification 87 Reliability 88 Redundancy 90 3.8 Summary 92 References 93 Problems 93 4.

Satellite Link Design

4.1 4.2

Introduction 96 Basic Transmission Theory 100 Example 4.2.1 104 Example 4.2.2 104 System Noise Temperature and G/T Ratio 105 Noise Temperature 105 Calculation of System Noise Temperature 107 Example 4.3.1 110 Example 4.3.2 110 Noise Figure and Noise Temperature 111 Example 4.3.3 112 G/T Ratio for Earth Stations 112 Example 4.3.4 112

4.3

96

59

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CONTENTS

4.4

Design of Downlinks 112 Link Budgets 113 Link Budget Example: C-Band Downlink for Earth Coverage Beam 115 4.5 Satellite Systems Using Small Earth Stations 117 Direct Broadcast TV 118 Example 4.5.1 123 4.6 Uplink Design 124 Example 4.6.1 127 4.7 Design for Specified CN: Combining CN and CI Values in Satellite Links Example 4.7.1 129 Overall (CN)0 with Uplink and Downlink Attenuation 129 Uplink and Downlink Attenuation in Rain 130 Uplink Attenuation and (CN)up 130 Downlink Attenuation and (CN)dn 131 System Design for Specific Performance 131 Satellite Communication Link Design Procedure 131 4.8 System Design Examples 132 System Design Example 4.8.1 133 Ku Band Uplink Design 133 Ku Band Downlink Design 134 Rain Effects at Ku Band 135 Summary of Ku Band Link Performance 137 System Design Example 4.8.2 Personal Communication System Using Low Earth Orbit Satellites 137 Inbound Link: Mobile Terminal to Gateway Station 141 Mobile Terminal to Satellite Link 142 Satellite to Gateway Station Link 143 Outbound Link 144 Downlink CN Budget 145 Optimizing System Performance 146 Link Margins with FEC 147 Rain Attenuation at Ku Band 147 Path Blockage at L-Band 149 Summary of L-band Mobile PCS System Performance 149 4.9 Summary 150 References 150 Problems 151 5. 5.1

5.2

Modulation and Multiplexing Techniques for Satellite Links Frequency Modulation 157 Waveform Equation for FM 158 Bandwidth of FM Signals: Carson’s Rule 159 Baseband SN Ratio for FM Signals 159 Pre-emphasis and de-emphasis 161 Pre-emphasis 162 Analog FM Transmission by Satellite 164 Television Signals 165 SN Ratios for FM Video Transmission 167 Example 5.2.1 168

156

127

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FM Threshold 168 SCPC FM Links 169 Example 5.2.2 170 Data Transmission Using Analog FM Channels 170 Example 5.2.3 171 5.3 Digital Transmission 172 Baseband Digital Signals 172 Baseband Transmission of Digital Data 172 Band-pass Transmission of Digital Data 179 Example 5.3.1 181 Example 5.3.2 181 Transmission of QPSK Signals through a Bandlimited Channel 182 Example 5.3.3 185 Example 5.3.3 185 5.4 Digital Modulation and Demodulation 187 Terminology 187 Modulation and Coding 187 Bit and Symbol Error Rates 188 Binary Phase Shift Keying (BPSK) 189 Probability of a Symbol Error 191 BPSK Bit Error Rate 194 QPSK Bit Error Rate 194 Example 5.4.1 195 Example 5.4.2 197 Generation of Quadrature Phase Shift Keying (QPSK) Signals 198 QPSK Variants 199 5.5 Digital Transmission of Analog Signals 201 Sampling and Quantizing 201 Nonuniform Quantization: Compression and Expansion 204 Signal-to-Noise Ratio in Digital Voice Systems 206 Digital Television 208 5.6 Time Division Multiplexing 209 TDM Terminology: The U.S. T1 24-Channel System 209 Other TDM Systems 211 Channel Synchronization in TDM 212 5.7 Summary 212 References 213 Problems 214 6.

Multiple Access

6.1 6.2

Introduction 221 Frequency Division Multiple Access (FDMA) 223 Intermodulation 226 Intermodulation Example 228 Calculation of CN with Intermodulation 230 Example 6.2.1 Power Sharing in FDMA 231 Example 6.2.2 Channel Capacity with Demand Access FDMA Time Division Multiple Access (TDMA) 233 Bits, Symbols, and Channels 234 TDMA Frame Structure 235

6.3

221

232

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Example 6.3.1 TDMA in a Fixed Station Network 237 Reference Burst and Preamble 238 Unique Word 239 Guard Times 241 Synchronization in TDMA Networks 242 Transmitter Power in TDMA Networks 243 Example 6.3.2 TDMA in a VSAT Network 244 Example 6.3.2 TDMA in a Fixed Earth Station Network 244 Satellite Switched TDMA 246 6.4 Onboard Processing 246 Baseband Processing Transponders 247 Satellite Switched TDMA with Onboard Processing 248 6.5 Demand Access Multiple Access (DAMA) 249 Example 6.5.1 FDMA-SCPC-DA 252 6.6 Random Access 254 6.7 Packet Radio Systems and Protocols 254 6.8 Code Division Multiple Access (CDMA) 257 Spread Spectrum Transmission and Reception 258 DS-SS CDMA Capacity 262 Example 6.8.1 CDMA in a Fixed Earth Station Network 263 Example 6.8.2 CDMA in an LEO Satellite Network 263 Example 6.8.3 GPS 264 6.9 Summary 266 References 267 Problems 267 7.

Error Control for Digital Satellite Links

273

7.1 7.2 7.3

Error Detection and Correction 273 Channel Capacity 275 Error Control Coding 277 Example 7.3.1 278 Linear and Cyclic Block Codes 279 Golay Codes 280 7.4 Performance of Block Error Correction Codes 281 7.5 Convolutional Codes 282 7.6 Implementation of Error Detection on Satellite Links Example 7.6.1 287 7.7 Concatenated Coding and Interleaving 288 7.8 Turbo Codes 290 7.9 Summary 292 References 292 Problems 293

284

8.

Propagation Effects and their Impact on Satellite–Earth Links

8.1 8.2

Introduction 297 Quantifying Attenuation and Depolarization 298 Example 8.2.1 301 Propagation Effects that Are Not Associated with Hydrometeors 306 Atmospheric Absorption 307 Cloud Attenuation 308

8.3

295

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Tropospheric Scintillation and Low Angle Fading 308 Faraday Rotation in the Atmosphere 310 Ionospheric Scintillations 312 8.4 Rain and Ice Effects 312 Characterizing Rain 312 Rain Climate Maps 314 Rainfall Rate Exceedance Contour Maps 315 Raindrop Distributions 315 8.5 Prediction of Rain Attenuation 317 Example 8.5.1 319 Example 8.5.2 323 Calculation of Long-Term Statistics for NGSO Systems 324 Scaling Attenuation with Elevation Angle and Frequency 325 Cosecant Law 325 Example 8.5.3 325 Squared Frequency Scaling Law 326 Example 8.5.4 326 ITU-R Long-Term Frequency Scaling of Rain Attenuation 326 8.6 Prediction of XPD 326 Canting Angle 328 Tilt Angle 328 Example 8.6.1 330 Example 8.6.2 331 Ice Crystal Depolarization 332 Rain Effects on Antenna Noise 332 Example 8.6.3 333 8.7 Propagation Impairment Countermeasures 333 Attenuation 333 Power Control 334 Signal Processing 335 Diversity 335 Depolarization 337 8.8 Summary 338 References 339 Problems 340 9.

VSAT SYSTEMS

9.1 9.2 9.3

Introduction 343 Overview of VSAT Systems 345 Network Architectures 347 One-Way Implementation 347 Split-Two-Way (Split IP) Implementation 347 Two-Way Implementation 348 Access Control Protocols 349 Delay Considerations 351 Basic Techniques 354 Multiple Access Selection 355 Signal Formats 362 Modulation, Coding, and Interference Issues 362

9.4 9.5

343

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CONTENTS

9.6

VSAT Earth Station Engineering 366 Antennas 366 Transmitters and Receivers 367 9.7 Calculation of Link Margins for a VSAT Star Network 9.8 System Design Procedure: Example 9.1 372 Description of System 373 System Parameters 373 Preliminary Calculations 374 Link CN Ratios 375 Inbound Links 376 Inbound Links with 270 Channels 378 Outbound Links 379 System Analysis 380 9.9 Some New Developments 383 9.10 Summary 384 References 385 Problems 385

370

10.

Low Earth Orbit and Non-Geostationary Satellite Systems

10.1 10.2

Introduction 389 Orbit Considerations 391 Equatorial Orbits 391 Inclined Orbits 392 Elliptical Orbits 394 Molniya Orbit 396 Radiation Effects 398 Sun Synchronous Orbit 403 Coverage and Frequency Considerations 406 General Aspects 406 Frequency band 406 Elevation Angle Considerations 408 Number of Beams per Coverage 411 Off-Axis Scanning 412 Determination of Optimum Orbital Altitude 418 Radiation Safety and Satellite Telephones 420 Projected NGSO System Customer Service Base 420 Delay and Throughput Considerations 421 System Considerations 423 Incremental Growth 424 Interim Operations 424 Replenishment Options 424 End-to-End System Implementation 425 Operational NGSO Constellation Designs 425 Ellipso 425 Globalstar 426 New ICO 428 Iridium 428 Orbcomm 429 Skybridge 429

10.3

10.4 10.5

10.6

388

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CONTENTS

Teledesic 430 Example 10.6.1 System Design 10.7 Summary 434 References 434 Problems 435 11.

432

Direct Broadcast Satellite Television and Radio

11.1 C-Band and Ku-Band Home Satellite TV 11.2 Digital DBS TV 441 11.3 DBS-TV System Design 447 11.4 DBS-TV Link Budget 449 11.5 Error Control in Digital DBS-TV 450 11.6 Master Control Station and Uplink 452 11.7 Installation of DBS-TV Antennas 454 11.8 Satellite Radio Broadcasting 455 11.9 Summary 456 References 457 12.

439

440

Satellite Navigation and the Global Positioning System

Introduction 458 Radio and Satellite Navigation 461 GPS Position Location Principles 463 Position Location in GPS 464 GPS Time 466 12.4 GPS Receivers and Codes 467 The CA Code 468 12.5 Satellite Signal Acquisition 470 12.6 GPS Navigation Message 472 12.7 GPS Signal Levels 473 12.8 Timing Accuracy 475 12.9 GPS Receiver Operation 476 12.10 GPS CA Code Accuracy 480 Dilution of Precision: HDOP, VDOP, and GDOP 12.11 Differential GPS 482 12.12 Summary 484 References 485 Problems 485

458

12.1 12.2 12.3

Appendix A Decibels in Communications Engineering Appendix B

481

487

FDM/FM/FDMA Analog Telephone Transmission

491

Baseband Voice Signal 491 Voice Signal Multiplexing 493 Frequency Modulation with Multiplexed Telephone Signals 496 Bandwidth Calculation for FDM/FM Telephone Signals 497 Telephone Performance Specifications 498 Practical Examples 499 Example B.1 499

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Example B.2 503 References 503 Appendix C

Complementary Error Function erfc(x) and Q Function Q(z) 504 Equivalence Formulas and Tables of Values References 504

Appendix D

The Simple Attenuation Model Example D.1 511 References 512

Glossary Index

513

522

507

504

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CHAPTER

INTRODUCTION 1.1

BACKGROUND Two developments in the twentieth century changed the way people lived: the automobile and telecommunications. Prior to the widespread availability of personal automobiles, individuals had to travel on foot, by bicycle, or on horseback. Trains provided faster travel between cities, but the lives of most people were centered on their hometowns and immediate surroundings. A journey of 100 miles was a major expedition for most people, and the easy mobility that we all take for granted in the twenty-first century was unknown. Before the telegraph and telephone came into widespread use, all communication was face-to-face, or in writing. If you wanted to talk to someone, you had to travel to meet with that person, and travel was slow and arduous. If you wanted to send information, it had to be written down and the papers hand carried to their destination. Telecommunication systems have now made it possible to communicate with virtually anyone at any time. Early telegraph and telephone systems used copper wire to carry signals over the earth’s surface and across oceans, and high frequency (HF) radio, also commonly called short wave radio, made possible intercontinental telephone links. Artificial earth satellites have been used in communications systems for more than 35 years and have become an essential part of the world’s telecommunications infrastructure. Satellites allow people to talk by telephone and exchange electronic mail from anywhere in the world and to receive hundreds of TV channels in their homes. The origins of satellite communications can be traced to an article written by Arthur C. Clarke in the British radio magazine Wireless World in 19451. At the time, Clarke was serving in the British Royal Air Force, and was interested in long-distance radio communication. He later became famous as the author of 2001: A Space Odyssey, and other science fiction books2. In 1945, HF radio was the only available method for radio communication over transcontinental distances, and it was not at all reliable. Sunspots and ionospheric disturbances could disrupt HF radio links for days at a time. Telegraph cables had been laid across the oceans as early as the mid-1800s, but cables capable of carrying voice signals across the Atlantic did not begin service until 1953. Clarke suggested that a radio relay satellite in an equatorial orbit with a period of 24 h would remain stationary with respect to the earth’s surface and make possible longdistance radio links. At the time Clarke wrote, there were no satellites in orbit nor rockets powerful enough to launch them. But his ideas for what we now know as a geostationary satellite system were not science fiction, as the launch of the Russian satellite Sputnik in 1957 was to prove. In 1965 the first geostationary satellite, Early Bird, began to provide telephone service across the Atlantic Ocean, fulfilling Clarke’s vision of 20 years earlier. Satellite communication systems were originally developed to provide long-distance telephone service. In the late 1960s, launch vehicles had been developed that could place a 500 kg satellite in geostationary earth orbit (GEO), with a capacity of 5000 telephone circuits, marking the start of an era of expansion for telecommunication satellites. Geostationary satellites were soon carrying transoceanic and transcontinental telephone calls. 1

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INTRODUCTION

For the first time, live television links could be established across the Atlantic and Pacific oceans to carry news and sporting events. The geostationary orbit is preferred for all high capacity communication satellite systems because a satellite in GEO appears to be stationary over a fixed point on the ground. It can establish links to one-third of the earth’s surface using fixed antennas at the earth stations. This is particularly valuable for broadcasting, as a single satellite can serve an entire continent. Direct broadcast satellite television (DBS-TV) and the distribution of video signals for cable television networks are the largest single revenue source for geostationary satellites, accounting for $17 B in revenues in 1998. By year 2001, nearly 200 GEO communication satellites were in orbit, serving every part of the globe. Although television accounts for much of the traffic carried by these satellites, international and regional telephony, data transmission, and Internet access are also important. In the populated parts of the world, the geostationary orbit is filled with satellites every 2° or 3° operating in almost every available frequency band. GEO satellites have grown steadily in weight, size, lifetime, and cost over the years. Some of the largest satellites launched to date are the KH and Lacrosse surveillance satellites of the U.S. National Reconnaissance Office weighing an estimated 13,600 kg (30,000 lb)3. By 2000, commercial telecommunications satellites weighing 6000 kg with lifetimes of 15 years were being launched into geostationary orbit at a typical cost around $125 M for the satellite and launch. The revenue earning capacity of these satellites must exceed $20 M per year for the venture to be profitable, and they must compete with optical fibers in carrying voice, data, and video signals. A single optical fiber can carry 4.5 Gbps, a capacity similar to that of the largest GEO satellites, and optical fibers are never laid singly but always in bundles. But GEO satellites can compete effectively on flexibility of delivery point. Any place within the satellite coverage can be served by simply installing an earth terminal. To do the same with a fiber-optic link requires fiber to be laid. Fiber-optic transmission systems compete effectively with satellites where there is a requirement for high capacity or, equivalently, when the user density exceeds the required economic threshold. GEO satellites have been supplemented by low and medium earth orbit satellites for special applications. Low earth orbit (LEO) satellites can provide satellite telephone and data services over continents or over the entire world, and by 2000 three systems were in orbit or nearing completion, with a total of 138 LEO satellites. LEO satellites are also used for earth imaging and surveillance. Although not strictly a satellite communications system, the Global Positioning System (GPS), which uses 24 medium earth orbit (MEO)

SIDEBAR The high capacity of both optical fibers and satellites, and the steady move of telecommunications traffic from analog signals to digital has lowered the cost of long-distance telephone calls and increased enormously the number of circuits available. In 1960, prior to the advent of satellite communications, the United States had 550 overseas telephone circuits. Calls to Europe cost more than $1 per minute at 1960 prices, and had to be placed through an operator, with delays of many hours being common. In 2000, virtually all international calls could be dialed by the end user, and rates to Europe had dropped to below $0.10 per minute.

To put the reduction in the cost of an international telephone call in perspective, we must remember that incomes have risen significantly over this time period. In the 1950s, a typical blue-collar wage was $1.50 per hour, so a blue-collar worker had to work for 40 min to pay for a call to Europe, ignoring any tax deductions. In 2000, the average worker in the United States earned $11.00 per hour, and had to work less than 1 min to pay for the international call. The United States now has hundreds of thousands of overseas telephone circuits, and video links daily carry live news reports from all over the globe.

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3

satellites, has revolutionized navigation. GPS receivers have become a consumer product. Eventually every car and cellular telephone will have a GPS receiver built into it so that drivers will not get lost and emergency calls from cellular phones will automatically carry information about the phone’s location.

1.2 A BRIEF HISTORY OF SATELLITE COMMUNICATIONS Satellite communications began in October 1957 with the launch by the USSR of a small satellite called Sputnik I. This was the first artificial earth satellite, and it sparked the space race between the United States and the USSR. Sputnik I carried only a beacon transmitter and did not have communications capability, but demonstrated that satellites could be placed in orbit by powerful rockets. The first satellite successfully launched by the United States was Explorer I, lofted from Cape Canaveral on January 31, 1958 on a Juno I rocket. The first voice heard from space was that of President Eisenhower, who recorded a brief Christmas message that was transmitted back to earth from the Project Score satellite in December 1958. The Score satellite was essentially the core of the Atlas ICBM (intercontinental ballistic missile) booster with a small payload in the nose. A tape recorder on Score had a storage capacity that allowed a 4 min message received from an earth station to be retransmitted. The batteries on Score failed after 35 days in orbit. After some early attempts to use large balloons (Echo I and II) as passive reflectors for communication signals, and some small experimental satellite launches, the first true communications satellites, Telstar I and II, were launched in July 1962 and May 1963. The Telstar satellites were built by Bell Telephone Laboratories and used C-band transponders adapted from terrestrial microwave link equipment. The uplink was at 6389 MHz and the downlink was at 4169 MHz, with 50-MHz bandwidth. The satellites carried solar cells and batteries that allowed continuous use of the single transponder, and demonstrations of live television links and multiplexed telephone circuits were made across the Atlantic Ocean, emphatically demonstrating the feasibility of satellite communications. The Telstar satellites were launched into what is now called a medium earth orbit, with periods of 158 and 225 min. This allowed transatlantic links to operate for about 20 min while the satellite was mutually visible. The orbits chosen for the Telstar satellites took them through several bands of high energy radiation which caused early failure of the electronics on board. However, the value of communication satellites had been demonstrated and work was begun to develop launch vehicles that could deliver a payload to geostationary orbit, and to develop satellites that could provide useful communication capacity. On July 24, 1961, U.S. President John F. Kennedy defined the general guidelines of U.S. policy in regard to satellite communications and made the first unambiguous references to a single worldwide system. On December 20, 1961, the U.S. Congress recommended that the International Telecommunications Union (ITU) should examine the aspects of space communications for which international cooperation would be necessary. The most critical step was in August 1962, when the U.S. Congress passed the Communications Satellite Act. This set the stage for commercial investment in an international satellite organization and, on July 19, 1964, representatives of the first 12 countries to invest in what became Intelsat (the International Telecommunications Satellite Organization) signed an initial agreement. The company that represented the United States at this initial signing ceremony was Comsat, an entity specifically created to act for the United States within Intelsat. It should be remembered that, at this point, the Bell System had a complete monopoly of all long-distance telephone communications within the United

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CHAPTER 1

INTRODUCTION

States. When Congress passed the Communications Satellite Act, the Bell System was specifically barred from directly participating in satellite communications, although it was permitted to invest in Comsat. Comsat essentially managed Intelsat in the formative years and should be credited with the remarkable success of the international venture. The first five Intelsat series of satellites (INTELSAT I through V) were selected, and their procurement managed, by teams put in place under Comsat leadership. Over this same phase, though, large portions of the Comsat engineering and operations groups transferred over to Intelsat so that, when the Permanent Management Arrangements came into force in 1979, many former Comsat groups were now part of Intelsat. In mid-1963, 99% of all satellites had been launched into LEO. LEO, and the slightly higher medium earth orbit (MEO), were much easier to reach than GEO with the small launchers available at that time. The intense debate was eventually settled on launcher reliability issues rather than on payload capabilities. The first 6 years of the so-called space age was a period of both payload and launcher development. The new frontier was very risky, with about one launch in four being fully successful. The system architecture of the first proposed commercial communications satellite system employed 12 satellites in an equatorial MEO constellation. Thus, with the launch failure rate at the time, 48 launches were envisioned to guarantee 12 operational satellites in orbit. Without 12 satellites in orbit, continuous 24-h coverage could not be offered. Twenty-four hours a day, seven days a week—referred to as 247 operation—is a requirement for any successful communications service. A GEO systems architecture requires only one satellite to provide 247 operation over essentially one-third of the inhabited world. On this basis, four launches would be required to achieve coverage of one third of the earth; 12 for the entire inhabited world. Despite its unproven technological approach, the geostationary orbit was selected by the entities that became Intelsat. The first Intelsat satellite, INTELSAT I (formerly Early Bird) was launched on April 16, 1965. The satellite weighed a mere 36 kg (80 lb) and incorporated two 64 GHz transponders, each with 25-MHz bandwidth. Commercial operations commenced between Europe and the United States on June 28, 1965. Thus, about 2 decades after Clarke’s landmark article in Wireless World, GEO satellite communications began. Intelsat was highly successful and grew rapidly as many countries saw the value of improved telecommunications, not just internationally but for national systems that provided high quality satellite communications within the borders of large countries. Canada was the first country to build a national telecommunication system using GEO satellites. Anik 1A was launched in May 1974, just 2 months before the first U.S. domestic satellite, WESTAR 1. The honor of the first regional satellite system, however, goes to the USSR Molniya system of highly elliptic orbit (HEO) satellites, the first of which was launched in April 1965 (the same month as INTELSAT I). Countries that are geographically spread like the USSR, which covers 11 time zones, have used regional satellite systems very effectively. Another country that benefited greatly from a GEO regional system was Indonesia, which consists of more than 3000 islands spread out over more than a thousand miles. A terrestrially based telecommunication system was not economically feasible for these countries, while a single GEO satellite allowed instant communications region wide. Such ease of communications via GEO satellites proved to be very profitable. Within less than 10 years, Intelsat was self-supporting and, since it was not allowed to make a profit, it began returning substantial revenues to what were known as its Signatories. Within 25 years, Intelsat had more than 100 Signatories4 and, in early 2000, there were 143 member countries and Signatories that formed part of the international Intelsat community.

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5

The astonishing commercial success of Intelsat led many nations to invest in their satellite systems. This was particularly true in the United States. By the end of 1983, telephone traffic carried by the U.S. domestic satellite systems earned more revenue than the Intelsat system. Many of the original Intelsat Signatories had been privatized by the early 1990s and were, in effect, competing not only with each other in space communications, but with Intelsat. It was clear that some mechanism had to be found whereby Intelsat could be turned into a for-profit, private entity, which could then compete with other commercial organizations while still safeguarding the interests of the smaller nations that had come to depend on the remarkably low communications cost that Intelsat offered. The first step in the move to privatizing Intelsat was the establishment of a commercial company called New Skies and the transfer of a number of Intelsat satellites to New Skies. In the 1970s and 1980s there was rapid development of GEO satellite systems for international, regional, and domestic telephone traffic and video distribution. In the United States, the expansion of fiber-optic links with very high capacity and low delay caused virtually all telephone traffic to move to terrestrial circuits by 1985. However, the demand for satellite systems grew steadily through this period, and the available spectrum in C band was quickly occupied, leading to expansion into Ku band. In the United States, most of the expansion after 1985 was in the areas of video distribution and VSAT (very small aperture terminal) networks. By 1995 it was clear that the GEO orbit capacity at Ku band would soon be filled, and Ka-band satellite systems would be needed to handle the expansion of digital traffic, especially wide band delivery of high-speed Internet data. SES, based in Luxemburg, began two-way multimedia and Internet access service in western and central Europe at Ka band using the Astra 1H satellite in 20016. Several Ka-band satellite systems are expected to be operational in the United States by 20037,8. The ability of satellite systems to provide communication with mobile users had long been recognized, and the International Maritime Satellite Organization (Inmarsat) has provided service to ships and aircraft for several decades, although at a high price. LEO satellites were seen as one way to create a satellite telephone system with worldwide coverage; numerous proposals were floated in the 1990s, with three LEO systems eventually reaching completion by 2000 (Iridium, Globalstar, and Orbcomm). The implementation of a LEO and MEO satellite system for mobile communication has proved much more costly than anticipated, and the capacity of the systems is relatively small compared to

SIDEBAR The first step in the move to privatizing Intelsat was the establishment of a commercial company called New Skies. New Skies is based in the Netherlands and, on 30 November 1998, six satellites were transferred from Intelsat ownership to New Skies. There was one INTELSAT V series satellite (IS-513 at 183° E), one INTELSAT VII series satellite (IS-703 at 57° E), two INTELSAT VIII series satellites (IS-803 at 338.5° E and IS-806 at 319.5° E), the INTELSAT-K satellite (in inclined orbit at 338.5° E), and a new satellite designed for direct broadcast services (K-TV at 95° E). New Skies has as their prime businesses plan the provisioning of TV services, both distribution and direct to home.

Intelsat is currently (2000) in the process of renewing its major assets through the purchase of up to seven INTELSAT IX satellites from SS-Loral to replace the current fleet of INTELSAT VI, and some of the INTELSAT VII, satellites. Each of these satellites carries the equivalent of 96 units of 36 MHz bandwidth. The satellites will be located at 62° E, 60° E, 335.5° E, 325.5° E, 332.5° E, 342° E, and 328.5° E. More details on the Intelsat fleet of satellites can be found at http://www.intelsat.int. Intelsat is moving forward with plans to privatize the remainder of the organization in the 2002/2003 time frame. Any reorganization will contain strong safeguards for smaller users to the system.

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CHAPTER 1

INTRODUCTION

GEO satellite systems, leading to a higher cost per transmitted bit. Satellite telephone systems were unable to compete with cellular telephone systems because of the high cost and relatively low capacity of the space segment. The Iridium system, for example, cost over $5 B to implement, but provided a total capacity for the United States of less than 10,000 telephone circuits. Iridium Inc. declared bankruptcy in early 2000, having failed to establish a sufficiently large customer base to make the venture viable. The entire Iridium system was sold to Iridium Satellite LLC for a reported $25 M, approximately 0.5% of the system’s construction cost. The future of the other LEO and MEO satellite telephone systems also seemed uncertain at the time this book was written. Satellite navigation systems, notably the Global Positioning System, have revolutionized navigation and surveying. The Global Positioning System took almost 20 years to design and fully implement, at a cost of $12 B. By 2000, GPS receivers could be built in Original Equipment Manufacturer (OEM) form for less than $25, and the worldwide GPS industry was earning billions of dollars from equipment sales and services. In the United States, aircraft navigation will depend almost entirely on GPS by 2010, and blind landing systems using GPS will also be available. Accurate navigation of ships, especially in coastal waters and bad weather, is also heavily reliant on GPS. Europe is building a comparable satellite navigation system called Gallileo.

1.3

SATELLITE COMMUNICATIONS IN 2000 Tables 1.1, 1.2, and 1.3 list the majority of the GEO, MEO, and LEO communication satellites in orbit in 2000. The list is not exhaustive, and excludes satellites used solely for military communications and surveillance, and those used primarily for weather forecasting and earth imaging. Not all the communications satellites are included, and experimental and scientific satellites are omitted. In all, Tables 1.1 and 1.2 list a total of 172 geostationary communication satellites. When other satellites in geostationary orbit are considered, there were close to 200 GEO satellites in operation in 2000 (Table 1.4). GEO satellites have always been the backbone of the commercial satellite communications industry. Large GEO satellites can serve one-third of the earth’s surface, and can carry up to 4 Gbps of data, or transmit up to 16 high power direct broadcast satellite television (DBS-TV) signals, each of which can deliver several video channels. The weight and power of GEO satellites have also increased. In 2000 a large GEO satellite could weigh 10,000 kg (10 tons), might generate 12 kW of power, and carry 60 transponders, with a trend toward even higher powers but lower weight. For example, in 2001 Space System/Loral contracted with APT Satellite Company Ltd. in Hong Kong to build the Apstar-V satellite, a GEO satellite serving Asia with a mass of 4845 kg when injected into geostationary orbit and an expected lifetime of 13 years. Apstar-V will generate an initial power of 10.6 kW, and carry 38 C-band transponders with 60-W output power and 16 Ku-band transponders at 141 W each5. Satellites generating 25 kW and carrying antennas with hundreds of beams are planned for the time frame 2005–2010. Television program distribution and DBS-TV have become the major source of revenue for commercial satellite system operators, earning more than half of the industry’s $30 B revenues for 1998. By the end of 2000 there were over 14 million DBS-TV customers in the United States. The high capacity of GEO satellites results from the use of high-power terrestrial transmitters and relatively high gain earth station antennas. Earth station antenna gain translates directly into communication capacity, and therefore into revenue. Increased capacity lowers the delivery cost per bit for a customer. Systems with fixed directional antennas can deliver bits at a significantly lower cost than systems using

Telecommunications

Inmarsat 3F, 1–5

Intelsat

Echostar Communications Corp., Littleton, CO http://www.dishnetwork.com GE Americom, Princeton, NJ http://www.geamericom.com

Telecommunications, video distribution, broadcasting, VSAT networks

GE-1, 1A, 2–8

GE Gstar 4

GE-1E

Direct to home digital television broadcasting

Echostar 1–5

Tempo 2

Direct to home digital television broadcasting

Data relay Mobile ship and aircraft communications

TDRSS-5, 6 Inmarsat 2F, 1–4

DBS-1, 2, 3, Directv-3R

Telecommunications

Columbia 515 (formerly Intelsat 515)

Directv Inc., El Segundo, CA http://www.directv.com

Mobile land communications

AMSC-1

American Mobile Satellite Corp., Reston, VA http://www.AmMobile.com Columbia Communications Corp., Bethesda, MD http://www.tdrss.com Comsat Corp., Bethesda, MD http://www.comsat.com Comsat is the U.S. provider of Inmarsat and Intelsat services

Type

Satellites

GEO Satellite Systems: U.S. Operators (after 3, 5)

Organization

TABLE 1.1

16 Ku band

24 C band Up to 28 Ku band 16 Ku band

Ku band BSS band DBS-1, 2, 3, 3R: 16 HP transponders Tempo 2: 11 transponders Ku band BSS band 16 transponders per satellite

19 GEO satellites C band and Ku band See Intelsat entry

4, 12 C band 9 GEO satellites L band See Inmarsat entry

12 C band 12 Ka band

16 L band

Transponders

105° W

5° E

119° W 61.5° W 148° W 110° W 139° W

(continued )

119° W Echostar 1, 2: Echostar 3: Echostar 4: Echostar 5: 79° W through

101° W

110° W

Many locations

174.3° E, 47° W Many locations

37.7° W

101° W

Orbit location

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7

8

(continued )

Telecommunications Telecommunications DBS-TV broadcasting to South America Telecommunications

PAS 1, 3R, 4, 5, 6B, 7 PAS 2, 8 PAS 6

Broadcasting Broadcasting

AmeriStar AsiaStar

6 HP Ku band

14, 14, 19 Ku band

Up to 24 C band Up to 36 Ku band 16 C band, 16 Ku band 24 C band, 24 Ku band 36 Ku band

24 C band, 8 Ku band 8 C band, 32 Ku band

95° W 105° E

77° W, 123° W, 74° W 21° E

169° W, 166° W 43° W

43° W through 68.5° W

89° W, 97° W, 93° W, 129° W 37.5° W, 34° W 133° W, 125° W, 74° W, 123° W 95° W 91° W 91° W

85° W 85° W, 83° W

131° W through 139° W

Orbit location

Telecommunications means any form of signal that can be sent through a satellite transponder, including analog and digital voice, data, and video.

Audio broadcasting

AfriStar

SBS-4, 5, 6

Telecommunications Telecommunications DBS-TV broadcasting to Latin America

Broadcasting, video distribution, telecommunications Telecommunications

Galaxy 3R Galaxy 7 Galaxy 8L

Telstar 11, 12 Galaxy 1RR, 5, 6, 9

Telstar 4, 5, 6, 7

16 Ku band 18 C band 6 Ku band 24 C band 16 to 28 Ku band 34, 38 Ku band 24 C band

24 C band

GE-1A and GE-5 are designated for broadcasting only

Satcom C-1, 3, 4, 5 Satcom K-2 Spacenet 3, 4

Transponders

Type

Satellitesa

There are 71 GEO satellites listed in the above table.

For more complete information about these satellite systems consult reference 6.

a

WorldSpace Corp., Washington, DC http://www.worldspace.com

Loral Skynet, Bedminster, NJ http://www.loralskynet.com PanAmSat Corp., Greenwich, CT http://www.panamsat.com

Organization

TABLE 1.1

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Broadcasting Broadcasting, telecommunications Broadcasting Broadcasting Telecommunications Broadcasting Telecommunications

DBS-TV Broadcasting Broadcasting, telecommunications

DBS-TV broadcasting

Arabsat 2A Arabsat 2B Arabsat 3A L-Star 1 L-Star 2 Asiasat 1 Asiasat 2 Asiasat 3S

BSat-1A, BSat-1B BS-3N DFS Kopernikus (1, 2)

Brasilsat A2 Brasilsat B1 Brasilsat B2 Brasilsat B3

Mobile communications

Garuda 1

ACeS Asia Cellular Satellite, Indonesia www.acesinternational.com Arab Satellite Communications Organization, Riyadh, Saudi Arabia www.arabsat.com Asia Broadcasting and Communications Network, Ltd., Bangkok, Thailand Asia Satellite Telecommunications Co. Ltd., Hong Kong, PRC www.asiasat.com Broadcasting Satellite System Corp., Tokyo, Japan Deutsche Telekom Geschaftsbereich Rundfunk, Bon-Bad Godesburg, Germany www.dtag.de Embratel, Rio De Janiero, Brazil

Type

Satellites

Organization

28 C band, 1 X band 28 C band, 1 X band 28 C band

24 C band

3 Ku band 10 Ku band, 1 Ka band

4 Ku band

24 C band 24 C band, 9 Ku band 28 C band, 16 Ku band

32 Ku band

20 Ku band 32 Ku band

22 C band, 12 Ku band

22 C band, 12 Ku band

140 with spot beams

Transponders

TABLE 1.2 GEO Satellite Systems: Non-U.S. and International Operators (after 3, 5)

70° W 65° W 84° W

92° W

(continued )

109.85° E 23.5 ° E, 28.5° E

110° E

122° E 100.5° E 105.5° E

126° E

26° E 126° E

30.5° E

26° E

123° E

Orbit location

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9

10

Japan Satellite Systems Inc., Tokyo, Japan www.jcsat.co.jp

Intelsat, Washington, DC, USA www.intelsat.int

Telecommunications Broadcasting Telecommunications

JCSat-3 JCSat-4A

Broadcasting, telecommunications

Intelsat 801, 802, 804 Intelsat 805 JCSat-1B, JCSat-2

Intelsat 601, 602, 603, 604, 605 Intelsat 701, 702, 704, 705, 709 Intelsat 706, 707

Intelsat 510, 511

Inmarsat 3F-1, 3F-2, 3F-3, 3F-4, 3F-5 Intelsat 505 Broadcasting, telecommunications

Broadcasting, telecommunications DBS-TV DBS-TV Mobile telecommunications

Eutelsat W2, W3

Inmarsat Ltd., London, UK www.inmarsat.org

Broadcasting, telecommunications

Eutelsat 2 F-1, F-2, F-3, F-4

Hot Bird Hot Bird 2, 3, 4 Inmarsat 2F-1, 2F-2, 2F-3, 2F-4

Broadcasting, telecommunications

Eutelsat 1 F-4, F-5

Eutelsat, Paris, France www.eutelsat.com

Type

Satellites

(continued )

Organization

TABLE 1.2

32 Ku band

12 C band, 28 Ku band

64 C band, 12 Ku band 36 C band, 6 Ku band 32 Ku band

42 C band, 28 Ku band

42 C band, 20 Ku band

26 C band, 6 Ku band, 1 L band 64 C band, 24 Ku band

21 C band, 6 Ku band

16 Ku band 20 Ku band L band, demand assigned

24 Ku band

16 Ku band (8 spare)

10 Ku band (2 spare)

Transponders

124° E

128° E

328.5° E, 174° E, 64° E 304.5° E 150° E, 154° E

325.5° E, 62° E, 335.5° E 60° E, 332.5° E 180° E, 177° E, 66° E 342° E, 310° E 307° E, 359° E

33° E, 330.5° E

64° E, 15.5° E, 178° E 54° W, 25° E 72° E

13° E All at 13° E 179° E, 98° W, 65° E 109° E

48° E, 12.5° E, 36° E, 10° E 16° E, 7° E

25.5° E, 21.5° E

Orbit location

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PT Pasifik Satelit Nusantara, Bekasi, Indonesia www.psn.co.id Sino Satellite Communications Co., Ltd., Beijing, PRC www.sinosat.com Societe Europenne des Satellites, SA (SES), Betzdorf, Luxembourg www.astra.lu

Korea Telecom, Korea www.kt.co.kr Mabuhay Philippines Satellite Corp., Makati City, Philippines NahuelSat, SA, Buenos Aires, Argentina www.nahuelsat.com.ar New Skies Satellites, N.V., The Hague, Netherlands www.newskiessat.com Telecommunications DBS-TV, broadcasting, telecommunications Broadcasting Telecommunications Telecommunications Broadcasting, telecommunications Broadcasting, telecommunications Broadcasting, telecommunications Broadcasting, telecommunications Mobile communications

Koreasat 3 Agila 2

Nahuel 2 NS 513 NS5 703

multimedia multimedia multimedia multimedia multimedia multimedia

DBS-TV, DBS-TV, DBS-TV, DBS-TV, DBS-TV, DBS-TV,

Astra Astra Astra Astra Astra Astra

1A, 1B 1C, 1D 1E, 1F 1G 2A, 2B 2G

Broadcasting, telecommunications Broadcasting, telecommunications

Palapa C1 SinoSat 1

Garuda 1

NSS K

NSS 806

NS5 803

Nahuel 1

Broadcasting

Koreasat 1, 2

16 Ku band 20 Ku band 20, 22 Ku band 30 Ku band 32, 30 Ku band 16 Ku band

24 C band, 14 Ku band

4 Ku band

140 spot beams

16 Ku band

36 C band, 6 Ku band

64 C band, 12 Ku band

46 C band, 20 Ku band

46 C band, 36 Ku band 42 C band, 12 Ku band

18 Ku band

30 Ku band, 3 Ka band 30 C band, 24 Ku band

15 Ku band

19.2° 19.2° 19.2° 19.2° 28.2° 28.2°

E E E E E E

123° E

113° E

123° E

21.5° W

40.5° W

21.5° W

57° E

81° W 183° E

71.8° W

116° E 146° E

116° E, 113° E

(continued )

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11

12 Turksat 1B, 1C

Broadcasting, telecommunications

Mobile communications

14 C band, 12 Ku band 16 L band, 1 Ku band

14 Ku band 14 C band, 12 Ku band

Thor 3 Anik E1 Anik E2 MSat 1

15 Ku band

15 Ku band 5 Ku band

32 Ku band

5 Ku band

Thor 2

Broadcasting, telecommunications

DBS-TV, VSAT networks DBS-TV Broadcasting, telecommunications

Sirius 2 Sirius 3 Thor 1

DBS-TV

Sirius 1

23 Ku band, 2 Ka band 7 Ku band

DBS-TV, telecommunications

23 Ku band, 2 Ka band

Superbird C Amos 1

There are 101 satellites listed in Table1.2.

Telesat Canada, Gloucester, ON, Canada www.telesat.ca TMI Communications, Ottawa, Canada www.tmisolutions.com Turk Telekom, Ankara, Turkey

Telenor Satellite Services AS, Oslo, Norway www.telenor.com

Spacecom Satellite Communication Services, Ramat-Gan, Israel www.spacecom.co.il Swedish Space Corp., Solna, Sweden www.ssc.se

Broadcasting, telecommunications

Transponders

23 Ku band, 2 Ka band

Superbird A

Space Communications Corp., Tokyo, Japan www.superbird.co.jp

Type

Superbird B

Satellites

(continued )

Organization

TABLE 1.2

31.3° E, 42° E

107.3° W 106.5° W

1° E 111.1° W

1° E

1° E

5° E

5° E

144° E 4° W

162° E

158° E

Orbit location

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Satellites 48 LEO satellites with four spares in orbit

66 LEO satellites with seven spares in orbit

28 LEO satellites

Globalstar, San Jose, CA, USA www.globalstar.com

Iridium LLC, Washington, DC, USA www.iridium.com

Orbcomm Global L.P., Dulles, VA, USA www.orbcomm.com

Data transmission to handheld and mobile terminals

Mobile communications, satellite telephones, all digital

Mobile communications, satellite telephones, all digital

Type

LEO and MEO Satellite Systems (after 3, 5)

Organization

TABLE 1.3

16 spot beams within footprint. Each beam has multiple 1.25-MHz channels with 1 to 13 channels per beam. Multiple access through CDMA. L- and S-band links to mobiles 48 spot beams with seven RF channels in 8 MHz. L-band links to mobiles. Ka-band links to Gateways. 22 GHz satellite cross links. Multiple access through FDMA/TDMA Bent pipe transponder with earth coverage beams. Data rate up to 2400 bps in 0.1-s bursts vhf links to mobiles (uplink 148 MHz, downlink 137 MHz)

Transponders

24 satellites in 45° inclined orbits. Two in 70° inclined orbits, two inclined 108°

Six orbital planes inclined at 84.6°, eleven satellites per plane 898 km altitude (485 nm)

Six orbital planes inclined at 52°, eight satellites per plane 1413 km altitude (763 nm)

Constellation orbit

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TABLE 1.4 Other Satellite Systems System

Satellites

Type and lifetime

Application

Orbits

Global positioning System (GPS), operated by U.S. Air Force Useful web sites: www.navcen.uscg.mil www.laafb.af.mil/SMC/ CZ/homepage/ http://gps.faa.gov/ http://www.spacecom. af.mil

Navstar GPS 13 through 21 22 through 40

Design lifetime 7.5 years

Navigation, early warning

43, 44, 45

Design lifetime 10 years

Six orbital planes with four satellites per plane at 20,200 km altitude. Inclination of orbital plane is 55°

Design lifetime 10 years

All satellites broadcast CDMA signals on two L-band frequencies

low gain antennas, such as those designed for use by mobile users. Consequently, GEO satellites look set to be the largest revenue earners in space for the foreseeable future. Figure 1.1 shows the estimated growth in revenue from all satellite communication services, projected to 2010. All radio systems require frequency spectrum, and the delivery of high-speed data requires a wide bandwidth. Satellite communication systems started in C band, with an allocation of 500 MHz, shared with terrestrial microwave links. As the GEO orbit filled up with satellites operating at C band, satellites were built for the next available frequency band, Ku band. There is a continuing demand for ever more spectrum to allow satellites to provide new services, with high speed access to the Internet forcing a move to Ka-band and even higher frequencies. Access to the Internet from small transmitting Ka-band earth stations located at the home offers a way to bypass the terrestrial telephone network and achieve much higher bit rates. SES began two-way Ka-band Internet access in Europe in 1998 with the Astra-K satellite, and the next generation of Ka-band satellites in the United States will offer similar services.

Worldwide revenue in billions of $U.S.

200

100

0 2000 2010 Year FIGURE 1.1 Growth of worldwide revenues from satellite communications 1980 through 2010. Beyond 2000, the curve is a projection. 1980

1990

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1.4 OVERVIEW OF SATELLITE COMMUNICATIONS

15

Successive World Radio Conferences have allocated new frequency bands for commercial satellite services that now include L, S, C, Ku, K, Ka, V, and Q bands. Mobile satellite systems use vhf, uhf, L, and S bands with carrier frequencies from 137 to 2500 MHz, and GEO satellites use frequency bands extending from 3.2 to 50 GHz. Despite the growth of fiber-optic links with very high capacity, the demand for satellite systems continues to increase. Satellites have also become integrated into complex communications architectures that use each element of the network to its best advantage. Examples are VSAT/WLL (very small aperture terminals/wireless local loop) in countries where the communications infrastructure is not yet mature and GEO/LMDS (local multipoint distribution systems) for the urban fringes of developed nations where the build-out of fiber has yet to be an economic proposition.

1.4 OVERVIEW OF SATELLITE COMMUNICATIONS Satellite communication systems exist because the earth is a sphere. Radio waves travel in straight lines at the microwave frequencies used for wideband communications, so a repeater is needed to convey signals over long distances. Satellites, because they can link places on the earth that are thousands of miles apart, are a good place to locate a repeater, and a GEO satellite is the best place of all. A repeater is simply a receiver linked to a transmitter, always using different radio frequencies, that can receive a signal from one earth station, amplify it, and retransmit it to another earth station. The repeater derives its name from nineteenth century telegraph links, which had a maximum length of about 50 miles. Telegraph repeater stations were required every 50 miles in a long-distance link so that the Morse code signals could be re-sent before they became too weak to read. The majority of communication satellites are in geostationary earth orbit, at an altitude of 35,786 km. Typical path length from an earth station to a GEO satellite is 38,500 km. Radio signals get weaker in proportion to the square of the distance traveled, so signals reaching a satellite are always very weak. Similarly, signals received on earth from a satellite 38,500 km away are also very weak, because of limits on the weight of GEO satellites and the electrical power they can generate using solar cells. It costs roughly $25,000 per kilogram to get a geostationary satellite in orbit. This obviously places severe restrictions on the size and weight of GEO satellites, since the high cost of building and launching a satellite must be recovered over a 10 to 15 year lifetime by selling communications capacity. Satellite communication systems are dominated by the need to receive very weak signals. In the early days, very large receiving antennas, with diameters up to 30 m, were needed to collect sufficient signal power to drive video signals or multiplexed telephone channels. As satellites have become larger, heavier, and more powerful, smaller earth station antennas have become feasible, and Direct Broadcast Satellite TV (DBS-TV) receiving systems can use dish antennas as small as 0.5 m in diameter. Satellite systems operate in the microwave and millimeter wave frequency bands, using frequencies between 1 and 50 GHz. Above 10 GHz, rain causes significant attenuation of the signal and the probability that rain will occur in the path between the satellite and an earth station must be factored into the system design. Above 20 GHz, attenuation in heavy rain (usually associated with thunderstorms) can cause sufficient attenuation that the link will fail. For the first 20 years of satellite communications, analog signals were widely used, with most links employing frequency modulation (FM). Wideband FM can operate at low carrier-to-noise ratios (C/N), in the 5 to 15 dB range, but adds a signal-to-noise

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improvement so that video and telephone signals can be delivered with signal-to-noise ratios (S/N) of 50 dB. The penalty for the improvement is that the radio frequency (RF) signal occupies a much larger bandwidth than the baseband signal. In satellite links, that penalty results because signals are always weak and the improvement in signal-to-noise ratio is essential. The move toward digital communications in terrestrial telephone and data transmission has been mirrored by a similar move toward digital transmission over satellite links. In the United States, only TV distribution at C band remains as the major analog satellite transmission system. Even this last bastion of analog signaling seems destined to disappear as cable TV stations switch over to digital receivers that allow six TV signals to be sent though a single Ku-band transponder. More importantly, dual standards permitting the transmission of not only digital TV but also high definition TV (HDTV), will eventually remove analog TV from consideration. Almost all other signals are digital—telephony, data, DBS-TV, radio broadcasting, and navigation with GPS all use digital signaling techniques. All of the LEO and MEO mobile communication systems are digital, taking advantage of voice compression techniques that allow a digital voice signal to be compressed into a bit stream at 4.8 kbps. Similarly, MPEG 2 (Moving Picture Coding Expert Group) and other video compression techniques allow video signals to be transmitted in full fidelity at rates less then 6.2 Mbps.

1.5

SUMMARY

Satellite communication systems have become an essential part of the world’s telecommunications infrastructure, serving billions of people with telephone, data, and video services. Despite the growth of fiber-optic links, which have much greater capacity than satellite systems and a lower cost per bit, satellite systems continue to thrive and investment in new systems continues. Satellite services have shifted away from telephony toward video and data delivery, with television broadcasting directly to the home emerging as one of the most powerful applications. GEO satellites carry the majority of

services, because the use of high gain fixed antennas at earth stations maximizes the capacity of the satellite. Over the years, there has been a trend away from trunk communications using very large earth station antennas toward delivery from more powerful satellites to individual users using much smaller antennas. LEO and MEO satellites are used for mobile communications and navigation systems and, as the need for Geographic Information Systems grows with a variety of applications, LEO earth imaging satellites have the potential to provide strong revenue streams.

REFERENCES 1. A. C. CLARKE, “Extra-terrestrial Relays,” Wireless World, pp. 305–308, 1945. 2. A. C. CLARKE, 2001: A Space Odyssey, New American Library, New York. 3. Aviation Week and Space Technology, Aerospace Source Book, McGraw-Hill, New York, Vol. 153, No. 3, January 17, 2000. 4. D. W. E. REES, “Satellite Communications: The First Quarter Century of Service,” John Wiley & Sons, New York, 1989.

5. Aviation Week and Space Technology, Aerospace Source Book, McGraw-Hill, New York, Vol. 154, No. 3, pp. 161–179 and pp. 249–266, Jan. 15, 2001. 6. http://www.astra.lu 7. www.astrolink.com 8. www.hns.com.spaceway

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2

ORBITAL MECHANICS AND LAUNCHERS 2.1

ORBITAL MECHANICS Developing the Equations of the Orbit This chapter is about how earth orbit is achieved, the laws that describe the motion of an object orbiting another body, how satellites maneuver in space, and the determination of the look angle to a satellite from the earth using ephemeris data that describe the orbital trajectory of the satellite. To achieve a stable orbit around the earth, a spacecraft must first be beyond the bulk of the earth’s atmosphere, i.e., in what is popularly called space. There are many definitions of space. U.S. astronauts are awarded their “space wings” if they fly at an altitude that exceeds 50 miles (80 km); some international treaties hold that the space frontier above a given country begins at a height of 100 miles (160 km). Below 100 miles, permission must be sought to over-fly any portion of the country in question. On reentry, atmospheric drag starts to be felt at a height of about 400,000 ft (76 miles  122 km). Most satellites, for any mission of more than a few months, are placed into orbits of at least 250 miles (400 km) above the earth. Even at this height, atmospheric drag is significant. As an example, the initial payload elements of the International Space Station (ISS) were injected into orbit at an altitude of 397 km when the shuttle mission left those modules on 9 June 1999. By the end of 1999, the orbital height had decayed to about 360 km, necessitating a maneuver to raise the orbit. Without onboard thrusters and sufficient orbital maneuvering fuel, the ISS would not last more than a few years at most in such a low orbit. To appreciate the basic laws that govern celestial mechanics, we will begin first with the fundamental Newtonian equations that describe the motion of a body. We will then give some coordinate axes within which the orbit of the satellite can be set and determine the various forces on the earth satellite. Newton’s laws of motion can be encapsulated into four equations: s  ut  1 12 2at 2 v2  u2  2at v  u  at P  ma

(2.1a) (2.1b) (2.1c) (2.1d)

where s is the distance traveled from time t  0; u is the initial velocity of the object at time t  0 and v the final velocity of the object at time t; a is the acceleration of the object; P is the force acting on the object; and m is the mass of the object. Note that the acceleration can be positive or negative, depending on the direction it is acting with respect to the velocity vector. Of these four equations, it is the last one that helps us understand the motion of a satellite in a stable orbit (neglecting any drag or other perturbing forces). Put into words, Eq. (2.1d) states that the force acting on a body is equal to the mass of 17

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the body multiplied by the resulting acceleration of the body. Alternatively, the resulting acceleration is the ratio of the force acting on the body to the mass of the body. Thus, for a given force, the lighter the mass of the body, the higher the acceleration will be. When in a stable orbit, there are two main forces acting on a satellite: a centrifugal force due to the kinetic energy of the satellite, which attempts to fling the satellite into a higher orbit, and a centripetal force due to the gravitational attraction of the planet about which the satellite is orbiting, which attempts to pull the satellite down toward the planet. If these two forces are equal, the satellite will remain in a stable orbit. It will continually fall toward the planet’s surface as it moves forward in its orbit but, by virtue of its orbital velocity, it will have moved forward just far enough to compensate for the “fall” toward the planet and so it will remain at the same orbital height. This is why an object in a stable orbit is sometimes described as being in “free fall.” Figure 2.1 shows the two opposing forces on a satellite in a stable orbit1. Force  mass  acceleration and the unit of force is a Newton, with the notation N. A Newton is the force required to accelerate a mass of 1 kg with an acceleration of 1 m/s2. The underlying units of a Newton are therefore (kg)  m/s2. In Imperial Units, one Newton  0.2248 ft lb. The standard acceleration due to gravity at the earth’s surface is 9.80665  103 km/s2, which is often quoted as 981 cm/s2. This value decreases The satellite has a mass, m, and is traveling with velocity, v, in the plane of the orbit

2 FOUT = mv r

FIN = GM2Em r ME

FIGURE 2.1 Forces acting on a satellite in a stable orbit around the earth (from Fig. 3.4 of reference 1). Gravitational force is inversely proportional to the square of the distance between the centers of gravity of the satellite and the planet the satellite is orbiting, in this case the earth. The gravitational force inward (FIN, the centripetal force) is directed toward the center of gravity of the earth. The kinetic energy of the satellite (FOUT, the centrifugal force) is directed diametrically opposite to the gravitational force. Kinetic energy is proportional to the square of the velocity of the satellite. When these inward and outward forces are balanced, the satellite moves around the earth in a “free fall” trajectory: the satellite’s orbit. For a description of the units, please see the text.

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19

with height above the earth’s surface. The acceleration, a, due to gravity at a distance r from the center of the earth is1 a  mr 2 km/s2

(2.1)

where the constant  is the product of the universal gravitational constant G and the mass of the earth ME. The product GME is called Kepler’s constant and has the value 3.986004418  105 km3/s2. The universal gravitational constant is G  6.672  1011 Nm2/kg2 or 6.672  1020 km3/kg s2 in the older units. Since force  mass  acceleration, the centripetal force acting on the satellite, FIN, is given by FIN  m  1mr 2 2  m  1GME r 2 2

(2.2a) (2.2b)

In a similar fashion, the centrifugal acceleration is given by1 a  v2 r

(2.3)

which will give the centrifugal force, FOUT, as FOUT  m  1v2r2

(2.4)

If the forces on the satellite are balanced, FIN  FOUT and, using Eqs. (2.2a) and (2.4), m  mr 2  m  v2 r hence the velocity v of a satellite in a circular orbit is given by v  1mr2 12

(2.5)

If the orbit is circular, the distance traveled by a satellite in one orbit around a planet is 2r, where r is the radius of the orbit from the satellite to the center of the planet. Since distance divided by velocity equals time to travel that distance, the period of the satellite’s orbit, T, will be Giving

T  12pr2 v  12pr2  3 1mr2 12 4 T  12pr 32 2  1m12 2

(2.6)

Table 2.1 gives the velocity, v, and orbital period, T, for four satellite systems that occupy typical LEO, MEO, and GEO orbits around the earth. In each case, the orbits are

TABLE 2.1 Orbital Velocity, Height, and Period of Four Satellite Systems

Satellite system

Orbital height (km)

Orbital velocity (km/s)

Orbital period (h min s)

Intelsat (GEO) New-ICO (MEO) Skybridge (LEO) Iridium (LEO)

35,786.03 10,255 1,469 780

3.0747 4.8954 7.1272 7.4624

23 56 4.1 5 55 48.4 1 55 17.8 1 40 27.0

Mean earth radius is 6378.137 km and GEO radius from the center of the earth is 42,164.17 km.

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z

Earth rotation Satellite r

c

Equatorial plane

y

x

FIGURE 2.2 The initial coordinate system that could be used to describe the relationship between the earth and a satellite. A Cartesian coordinate system with the geographical axes of the earth as the principal axes is the simplest coordinate system to set up. The rotational axis of the earth is about the axis cz, where c is the center of the earth and cz passes through the geographic north pole. Axes cx, cy, and cz are mutually orthogonal axes, with cx and cy passing through the earth’s geographic equator. The vector r locates the moving satellite with respect to the center of the earth.

circular and the average radius of the earth is taken as 6378.137 km1. A number of coordinate systems and reference planes can be used to describe the orbit of a satellite around a planet. Figure 2.2 illustrates one of these using a Cartesian coordinate system with the earth at the center and the reference planes coinciding with the equator and the polar axis. This is referred to as a geocentric coordinate system. With the coordinate system set up as in Figure 2.2, and with the satellite mass m located at a vector distance r from the center of the earth, the gravitational force F on the satellite is given by F

GME m r r3

(2.7)

Where ME is the mass of the earth and G  6.672  1011 Nm2/kg2. But force  mass  acceleration and Eq. (2.7) can be written as Fm

d2r dt 2

(2.8)

From Eqs. (2.7) and (2.8) we have 

r

d2r

r

dt 2

3m 

(2.9)

Which yields d2r dt

2



r r3

m0

(2.10)

This is a second-order linear differential equation and its solution will involve six undetermined constants called the orbital elements. The orbit described by these orbital elements can be shown to lie in a plane and to have a constant angular momentum. The solution to Eq. (2.10) is difficult since the second derivative of r involves the second derivative of the unit vector r. To remove this dependence, a different set of coordinates can

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21

z z0

y0

c

x0

FIGURE 2.3 The orbital plane coordinate system. In this coordinate system, the orbital plane of the satellite is used as the reference plane. The orthogonal axes x0 and y0 lie in the orbital plane. The third axis, z0, is perpendicular to the orbital plane. The geographical z-axis of the earth (which passes through the true North Pole and the center of the earth, c) does not lie in the same direction as the z0 axis except for satellite orbits that are exactly in the plane of the geographical equator.

be chosen to describe the location of the satellite such that the unit vectors in the three axes are constant. This coordinate system uses the plane of the satellite’s orbit as the reference plane. This is shown in Figure 2.3. Expressing Eq. (2.10) in terms of the new coordinate axes x0, y0, and z0 gives xˆ0 a

m1x0 xˆ0  y0 yˆ0 2 d 2x0 d 2y0 ˆ b  y a 0 0 2 2 b  dt dt 1x20  y20 2 32

(2.11)

Equation (2.11) is easier to solve if it is expressed in a polar coordinate system rather than a Cartesian coordinate system. The polar coordinate system is shown in Figure 2.4. With the polar coordinate system shown in Figure 2.4 and using the transformations x0 y0 xˆ0 yˆ0

 r0 cos f0  r0 sin f0  rˆ0 cos f0  fˆ 0 sin f0  fˆ 0 cos f0  rˆ0 sin f0

(2.12a) (2.12b) (2.12c) (2.12d)

and equating the vector components of r0 and 0 in turn in Eq. (2.11) yields m df0 d 2r0 b 2 2  r0 a dt dt r0

(2.13)

y0

r0 φ0

z0

x0

FIGURE 2.4 Polar coordinate system in the plane of the satellite’s orbit. The plane of the orbit coincides with the plane of the paper. The axis z0 is straight out of the paper from the center of the earth, and is normal to the plane of the satellite’s orbit. The satellite’s position is described in terms of the radius from the center of the earth r0 and the angle this radius makes with the x0 axis, 0.

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and r0 a

d 2f0 dr0 df0 ba b0 2 b  2a dt dt dt

(2.14)

Using standard mathematical procedures, we can develop an equation for the radius of the satellite’s orbit, r0, namely p (2.15) r0  1  e cos 1f0  u0 2

Where 0 is a constant and e is the eccentricity of an ellipse whose semilatus rectum p is given by p  1h2 2 m

(2.16)

and h is magnitude of the orbital angular momentum of the satellite. That the equation of the orbit is an ellipse is Kepler’s first law of planetary motion.

Kepler’s Three Laws of Planetary Motion Johannes Kepler (1571–1630) was a German astronomer and scientist who developed his three laws of planetary motion by careful observations of the behavior of the planets in the solar system over many years, with help from some detailed planetary observations by the Hungarian astronomer Tycho Brahe. Kepler’s three laws are 1. The orbit of any smaller body about a larger body is always an ellipse, with the center of mass of the larger body as one of the two foci. 2. The orbit of the smaller body sweeps out equal areas in equal time (see Figure 2.5).

t2

A 12

t3

A 34

t4

E t1

FIGURE 2.5 Illustration of Kepler’s second law of planetary motion. A satellite is in orbit about the planet earth, E. The orbit is an ellipse with a relatively high eccentricity, that is, it is far from being circular. The figure shows two shaded portions of the elliptical plane in which the orbit moves, one is close to the earth and encloses the perigee while the other is far from the earth and encloses the apogee. The perigee is the point of closest approach to the earth while the apogee is the point in the orbit that is furthest from the earth. While close to perigee, the satellite moves in the orbit between times t1 and t2 and sweeps out an area denoted by A12. While close to apogee, the satellite moves in the orbit between times t3 and t4 and sweeps out an area denoted by A34. If t1  t2  t3  t4 then A12  A34.

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23

SIDEBAR Kepler’s laws were subsequently confirmed, about 50 years later, by Isaac Newton, who developed a mathematical model for the motion of the planets. Newton was one of the first people to make use of differential calculus, and with his understanding of gravity, was able to describe the motion of planets from a mathematical model based on his laws of motion and

the concept of gravitational attraction. The work was published in the Philosophiae Naturalis Principia Mathematica in 1687. At that time, Latin was the international language of formally educated people, much in the way English has become the international language of e-mail and business today, so Newton’s Principia was written in Latin.

3. The square of the period of revolution of the smaller body about the larger body equals a constant multiplied by the third power of the semimajor axis of the orbital ellipse. That is, T 2  (4 2a3) where T is the orbital period, a is the semimajor axis of the orbital ellipse, and  is Kepler’s constant. If the orbit is circular, then a becomes distance r, defined as before, and we have Eq. (2.6). Describing the orbit of a satellite enables us to develop Kepler’s second two laws.

Describing the Orbit of a Satellite The quantity 0 in Eq. (2.15) serves to orient the ellipse with respect to the orbital plane axes x0 and y0. Now that we know that the orbit is an ellipse, we can always choose x0 and y0 so that 0 is zero. We will assume that this has been done for the rest of this discussion. This now gives the equation of the orbit as r0 

p 1  e cos f0

(2.17)

The path of the satellite in the orbital plane is shown in Figure 2.6. The lengths a and b of the semimajor and semiminor axes are given by a  p 11  e2 2 b  a11  e2 2 12

(2.18) (2.19)

The point in the orbit where the satellite is closest to the earth is called the perigee and the point where the satellite is farthest from the earth is called the apogee. The perigee and apogee are always exactly opposite each other. To make 0 equal to zero, we have chosen the x0 axis so that both the apogee and the perigee lie along it and the x0 axis is therefore the major axis of the ellipse. The differential area swept out by the vector r0 from the origin to the satellite in time dt is given by dA  0.5r 02 a

df0 b dt  0.5hdt dt

(2.20)

Remembering that h is the magnitude of the orbital angular momentum of the satellite, the radius vector of the satellite can be seen to sweep out equal areas in equal times. This is Kepler’s second law of planetary motion. By equating the area of the ellipse (ab) to the area swept out in one orbital revolution, we can derive an expression for the orbital period T as T 2  14p2a3 2 m

(2.21)

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yo

b ro φo Apogee

a

xo

O

C

Perigee

ae

a (1 + e)

a (1 − e)

FIGURE 2.6 The orbit as it appears in the orbital plane. The point O is the center of the earth and the point C is the center of the ellipse. The two centers do not coincide unless the eccentricity, e, of the ellipse is zero (i.e., the ellipse becomes a circle and a  b). The dimensions a and b are the semimajor and semiminor axes of the orbital ellipse, respectively.

This equation is the mathematical expression of Kepler’s third law of planetary motion: the square of the period of revolution is proportional to the cube of the semimajor axis. (Note that this is the square of Eq. (2.6) and that in Eq. (2.6) the orbit was assumed to be circular such that semimajor axis a  semiminor axis b  circular orbit radius from the center of the earth r.) Kepler’s third law extends the result from Eq. (2.6), which was derived for a circular orbit, to the more general case of an elliptical orbit. Equation (2.21) is extremely important in satellite communications systems. This equation determines the period of the orbit of any satellite, and it is used in every GPS receiver in the calculation of the positions of GPS satellites. Equation (2.21) is also used to find the orbital radius of a GEO satellite, for which the period T must be made exactly equal to the period of one revolution of the earth for the satellite to remain stationary over a point on the equator. An important point to remember is that the period of revolution, T, is referenced to inertial space, namely, to the galactic background. The orbital period is the time the orbiting body takes to return to the same reference point in space with respect to the galactic background. Nearly always, the primary body will also be rotating and so the period of revolution of the satellite may be different from that perceived by an observer who is standing still on the surface of the primary body. This is most obvious with a geostationary earth orbit (GEO) satellite (see Table 2.1). The orbital period of a GEO satellite is exactly equal to the period of rotation of the earth, 23 h 56 min 4.1 s, but, to an observer on the ground, the satellite appears to have an infinite orbital period: it always stays in the same place in the sky.

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To be perfectly geostationary, the orbit of a satellite needs to have three features: (a) it must be exactly circular (i.e., have an eccentricity of zero); (b) it must be at the correct altitude (i.e., have the correct period); and (c) it must be in the plane of the equator (i.e., have a zero inclination with respect to the equator). If the inclination of the satellite is not zero and/or if the eccentricity is not zero, but the orbital period is correct, then the satellite will be in a geosynchronous orbit. The position of a geosynchronous satellite will appear to oscillate about a mean look angle in the sky with respect to a stationary observer on the earth’s surface. The orbital period of a GEO satellite, 23 h 56 min 4.1 s, is one sidereal day. A sidereal day is the time between consecutive crossings of any particular longitude on the earth by any star, other than the sun1. The mean solar day of 24 h is the time between any consecutive crossings of any particular longitude by the sun, and is the time between successive sunrises (or sunsets) observed at one location on earth, averaged over an entire year. Because the earth moves round the sun once per 365 1⁄4 days, the solar day is 1440365.25  3.94 min longer than a sidereal day.

Locating the Satellite in the Orbit Consider now the problem of locating the satellite in its orbit. The equation of the orbit may be rewritten by combining Eqs. (2.15) and (2.18) to obtain r0 

a11  e2 2 1  e cos f0

(2.22)

The angle 0 (see Figure 2.6) is measured from the x0 axis and is called the true anomaly. [Anomaly was a measure used by astronomers to mean a planet’s angular distance from its perihelion (closest approach to the sun), measured as if viewed from the sun. The term was adopted in celestial mechanics for all orbiting bodies.] Since we defined the positive x0 axis so that it passes through the perigee, 0 measures the angle from the perigee to the instantaneous position of the satellite. The rectangular coordinates of the satellite are given by x0  r0 cos f0 y0  r0 sin f0

(2.23) (2.24)

As noted earlier, the orbital period T is the time for the satellite to complete a revolution in inertial space, traveling a total of 2 radians. The average angular velocity  is thus h  12p2 T  1m12 2  1a32 2

(2.25)

If the orbit is an ellipse, the instantaneous angular velocity will vary with the position of the satellite around the orbit. If we enclose the elliptical orbit with a circumscribed circle of radius a (see Figure 2.7), then an object going around the circumscribed circle with a constant angular velocity  would complete one revolution in exactly the same period T as the satellite requires to complete one (elliptical) orbital revolution. Consider the geometry of the circumscribed circle as shown in Figure 2.7. Locate the point (indicated as A) where a vertical line drawn through the position of the satellite intersects the circumscribed circle. A line from the center of the ellipse (C) to this point (A) makes an angle E with the x0 axis; E is called the eccentric anomaly of the satellite.

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yo axis

A

a

yo E C

O

xo

xo axis

Orbit

Circumscribed Circle FIGURE 2.7 The circumscribed circle and the eccentric anomaly E. Point O is the center of the earth and point C is both the center of the orbital ellipse and the center of the circumscribed circle. The satellite location in the orbital plane coordinate system is specified by (x0, y0). A vertical line through the satellite intersects the circumscribed circle at point A. The eccentric anomaly E is the angle from the x0 axis to the line joining C and A.

It is related to the radius r0 by r0  a11  e cos E2

(2.26)

a  r0  ae cos E

(2.27)

Thus We can also develop an expression that relates eccentric anomaly E to the average angular velocity , which yields h dt  11  e cos E2 dE

(2.28)

Let tp be the time of perigee. This is simultaneously the time of closest approach to the earth; the time when the satellite is crossing the x0 axis; and the time when E is zero. If we integrate both sides of Eq. (2.28), we obtain h1t  tp 2  E  e sin E

(2.29)

The left side of Eq. (2.29) is called the mean anomaly, M. Thus M  h1t  tp 2  E  e sin E

(2.30)

The mean anomaly M is the arc length (in radians) that the satellite would have traversed since the perigee passage if it were moving on the circumscribed circle at the mean angular velocity . If we know the time of perigee, tp, the eccentricity, e, and the length of the semimajor axis, a, we now have the necessary equations to determine the coordinates (r0, 0)

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and (x0, y0) of the satellite in the orbital plane. The process is as follows 1. 2. 3. 4. 5. 6.

Calculate  using Eq. (2.25). Calculate M using Eq. (2.30). Solve Eq. (2.30) for E. Find r0 from E using Eq. (2.27). Solve Eq. (2.22) for 0. Use Eqs. (2.23) and (2.24) to calculate x0 and y0.

Now we must locate the orbital plane with respect to the earth.

Locating the Satellite with Respect to the Earth At the end of the last section, we summarized the process for locating the satellite at the point (x0, y0, z0) in the rectangular coordinate system of the orbital plane. The location was with respect to the center of the earth. In most cases, we need to know where the satellite is from an observation point that is not at the center of the earth. We will therefore develop the transformations that permit the satellite to be located from a point on the rotating surface of the earth. We will begin with a geocentric equatorial coordinate system as shown in Figure 2.8. The rotational axis of the earth is the zi axis, which is through the geographic North Pole. The xi axis is from the center of the earth toward a fixed location in space called the first point of Aries (see Figure 2.8). This coordinate system moves through space; it translates as the earth moves in its orbit around the sun, but it does not rotate as the earth rotates. The xi direction is always the same, whatever the earth’s position around the sun and is in the direction of the first point of Aries. The (xi, yi) plane contains the earth’s equator and is called the equatorial plane. Angular distance measured eastward in the equatorial plane from the xi axis is called right ascension and given the symbol RA. The two points at which the orbit

zi

δ

RA

xi

yi

FIGURE 2.8 The geocentric equatorial system. This geocentric system differs from that shown in Figure 2.1 only in that the xi axis points to the first point of Aries. The first point of Aries is the direction of a line from the center of the earth through the center of the sun at the vernal equinox (about March 21 in the Northern Hemisphere), the instant when the subsolar point crosses the equator from south to north. In the above system, an object may be located by its right ascension RA and its declination .

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penetrates the equatorial plane are called nodes; the satellite moves upward through the equatorial plane at the ascending node and downward through the equatorial plane at the descending node, given the conventional picture of the earth, with north at the top, which is in the direction of the positive z axis for the earth centered coordinate set. Remember that in space there is no up or down; that is a concept we are familiar with because of gravity at the earth’s surface. For a weightless body in space, such as an orbiting spacecraft, up and down have no meaning unless they are defined with respect to a reference point. The right ascension of the ascending node is called . The angle that the orbital plane makes with the equatorial plane (the planes intersect at the line joining the nodes) is called the inclination, i. Figure 2.9 illustrates these quantities. The variables  and i together locate the orbital plane with respect to the equatorial plane. To locate the orbital coordinate system with respect to the equatorial coordinate system we need , the argument of perigee west. This is the angle measured along the orbit from the ascending node to the perigee. Standard time for space operations and most other scientific and engineering purposes is universal time (UT), also known as zulu time (z). This is essentially the mean solar time at the Greenwich Observatory near London, England. Universal time is measured in hours, minutes, and seconds or in fractions of a day. It is 5 h later than Eastern Standard Time, so that 07:00 EST is 12:00:00 h UT. The civil or calendar day begins at 00:00:00 hours UT, frequently written as 0 h. This is, of course, midnight (24:00:00) on the previous day. Astronomers employ a second dating system involving Julian days and Julian dates. Julian days start at noon UT in a counting system whereby noon on December 31, 1899, was the beginning of Julian day 2415020, usually written 241 5020. These are extensively tabulated in reference 2 and additional information is in reference 14. As an example, noon on December 31, 2000, the eve of the twenty-first century, is the start of Julian day 245 1909. Julian dates can be used to indicate time by appending a decimal fraction; 00:00:00 h UT on January 1, 2001—zero hour, minute, and

zi Satellite Perigee

yi

ω

i Ω

Ascending node

xi FIGURE 2.9 Locating the orbit in the geocentric equatorial system. The satellite penetrates the equatorial plane (while moving in the positive z direction) at the ascending node. The right ascension of the ascending node is  and the inclination i is the angle between the equatorial plane and the orbital plane. Angle , measured in the orbital plane, locates the perigee with respect to the equatorial plane.

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second for the third millenium A.D.—is given by Julian date 245 1909.5. To find the exact position of an orbiting satellite at a given instant in time requires knowledge of the orbital elements.

Orbital Elements To specify the absolute (i.e., the inertial) coordinates of a satellite at time t, we need to know six quantities. (This was evident earlier when we determined that a satellite’s equation of motion was a second order vector linear differential equation.) These quantities are called the orbital elements. More than six quantities can be used to describe a unique orbital path and there is some arbitrariness in exactly which six quantities are used. We have chosen to adopt a set that is commonly used in satellite communications: eccentricity (e), semimajor axis (a), time of perigee (tp), right ascension of ascending node (), inclination (i), and argument of perigee (). Frequently, the mean anomaly (M) at a given time is substituted for tp. EXAMPLE 2.1.1 Geostationary Satellite Orbit Radius The earth rotates once per sidereal day of 23 h 56 min 4.09 s. Use Eq. (2.21) to show that the radius of the GEO is 42,164.17 km as given in Table 2.1. Answer

Equation (2.21) gives the square of the orbital period in seconds T 2  14p2a3 2 m

Rearranging the equation, the orbital radius a is given by a3  T 2m 14p 2 2 For one sidereal day, T  86,164.09 s. Hence a3  186,164.12 2  3.986004418  105 14p2 2  7.496020251  1013 km3 a  42,164.17 km

This is the orbital radius for a geostationary satellite, as given in Table 2.1.



EXAMPLE 2.1.2 Low Earth Orbit The Space Shuttle is an example of a low earth orbit satellite. Sometimes, it orbits at an altitude of 250 km above the earth’s surface, where there is still a finite number of molecules from the atmosphere. The mean earth’s radius is approximately 6378.14 km. Using these figures, calculate the period of the shuttle orbit when the altitude is 250 km and the orbit is circular. Find also the linear velocity of the shuttle along its orbit. Answer The radius of the 250-km altitude Space Shuttle orbit is (re  h)  6378.14  250.0  6628.14 km From Eq. 2.21, the period of the orbit is T where T 2  14p2a3 2 m  4p2  16628.142 33.986004418  105 s2  2.88401145  107 s2 Hence the period of the orbit is T  5370.30 s  89 min 30.3 s.

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This orbit period is about as small as possible. At a lower altitude, friction with the earth’s atmosphere will quickly slow the Shuttle down and it will return to earth. Thus, all spacecraft in stable earth orbit have orbital periods exceeding 89 min 30 s. The circumference of the orbit is 2a  41,645.83 km. Hence the velocity of the Shuttle in orbit is 2paT  41,645.835370.13  7.755 km/s Alternatively, you could use Eq. (2.5): v  (r)12. The term   3.986004418  105 km3/s2 and the term r  (6378.14  250.0) km, yielding v  7.755 km/s. Note: If  and r had been quoted in units of m3/s2 and m, respectively, the answer would have been in meters/second. Be sure to keep the units the same during a calculation procedure. A velocity of about 7.8 km/s is a typical velocity for a low earth orbit satellite. As the altitude of a satellite increases, its velocity becomes smaller. 

EXAMPLE 2.1.3 Elliptical orbit A satellite is in an elliptical orbit with a perigee of 1000 km and an apogee of 4000 km. Using a mean earth radius of 6378.14 km, find the period of the orbit in hours, minutes, and seconds, and the eccentricity of the orbit. Answer The major axis of the elliptical orbit is a straight line between the apogee and perigee, as seen in Figure 2.7. Hence, for a semimajor axis length a, earth radius re, perigee height hp, and apogee height ha, 2a  2re  hp  ha  2  6378.14  1000.0  4000.0  17,756.28 km Thus the semimajor axis of the orbit has a length a  8878.14 km. Using this value of a in Eq. (2.21) gives an orbital period T seconds where T 2  14p 2a 3 2 m  4p2  18878.072 33.986004418  105 s2  6.930872802  107 s2 T  8325.1864 s  138 min 45.19 s  2 h 18 min 45.19 s The eccentricity of the orbit is given by e, which can be found from Eq. (2.27) by considering the instant at which the satellite is at perigee. Referring to Figure 2.7, when the satellite is at perigee, the eccentric anomaly E  0 and r0  re  hp. From Eq. (2.27), at perigee r0  a11  e cos E 2

and cos E  1

Hence re  hp  a11  e2

e  1  1re  hp 2 a  1  7,378.148878.14  0.169

2.2



LOOK ANGLE DETERMINATION Navigation around the earth’s oceans became more precise when the surface of the globe was divided up into a gridlike structure of orthogonal lines: latitude and longitude. Latitude is the angular distance, measured in degrees, north or south of the equator and longitude is the angular distance, measured in degrees, from a given reference longitudinal line. At the time that this grid reference became popular, there were two major seafaring nations vying for dominance: England and France. England drew its reference zero longitude through Greenwich, a town close to London, England, and France,

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SIDEBAR Frequencies and orbital slots for new satellites are registered with the International Frequency Registration Board (IFRB), part of the ITU located in Geneva. The initial application by an organization or company that wants to orbit a new satellite is made to the national body that controls the allocation and use of radio frequencies—the FCC in the United States, for example—which must first approve the application and then forward it to the IFRB. The first organization to file with the IFRB

for a particular service is deemed to have protection from newcomers. Any other organization filing to carry the same service at, or close to, that orbital location (within 2°) must coordinate their use of the frequency bands with the first organization. The first user may cause interference into subsequent filer’s satellite systems, since they were the first to be awarded the orbital slot and frequencies, but the later filers’ satellites must not cause interference with the first user’s system.

not surprisingly, drew its reference longitude through Paris, France. Since the British Admiralty chose to give away their maps and the French decided to charge a fee for theirs, it was not surprising that the use of Greenwich as the zero reference longitude became dominant within a few years. [It was the start of .com market dominance through giveaways three centuries before E-commerce!] Geometry was a much older science than navigation and so 90° per quadrant on the map was an obvious selection to make. Thus, there are 360° of longitude (measured from 0° at the Greenwich Meridian, the line drawn from the North Pole to the South Pole through Greenwich, England) and 90° of latitude, plus being measured north of the equator and minus south of the equator. Latitude 90° N (or 90°) is the North Pole and latitude 90° S (or 90°) is the South Pole. When GEO satellite systems are registered in Geneva, their (subsatellite) location over the equator is given in degrees east to avoid confusion. Thus, the INTELSAT primary location in the Indian Ocean is registered at 60° E and the primary location in the Atlantic Ocean is at 335.5° E (not 24.5° W). Earth stations that communicate with satellites are described in terms of their geographic latitude and longitude when developing the pointing coordinates that the earth station must use to track the apparent motion of the satellite. The coordinates to which an earth station antenna must be pointed to communicate with a satellite are called the look angles. These are most commonly expressed as azimuth (Az) and elevation (El), although other pairs exist. For example, right ascension and declination are standard for radio astronomy antennas. Azimuth is measured eastward (clockwise) from geographic north to the projection of the satellite path on a (locally) horizontal plane at the earth station. Elevation is the angle measured upward from the local horizontal plane at the earth station to the satellite path. Figure 2.10 illustrates these look angles. In all look angle determinations, the precise location of the satellite is critical. A key location in many instances is the subsatellite point.

The Subsatellite Point The subsatellite point is the location on the surface of the earth that lies directly between the satellite and the center of the earth. It is the nadir pointing direction from the satellite and, for a satellite in an equatorial orbit, it will always be located on the equator. Since geostationary satellites are in equatorial orbits and are designed to stay “stationary” over

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Local vertical

Path to satellite

El North

Az Projection of path onto local horizontal plane

East FIGURE 2.10 The definition of elevation (El) and azimuth (Az). The elevation angle is measured upward from the local horizontal at the earth station and the azimuth angle is measured from true north in an eastward direction to the projection of the satellite path onto the local horizontal plane.

the earth, it is usual to give their orbital location in terms of their subsatellite point. As noted in the example given earlier, the Intelsat primary satellite in the Atlantic Ocean Region (AOR) is at 335.5° E longitude. Operators of international geostationary satellite systems that have satellites in all three ocean regions (Atlantic, Indian, and Pacific) tend to use longitude east to describe the subsatellite points to avoid confusion between using both east and west longitude descriptors. For U.S. geostationary satellite operators, all of the satellites are located west of the Greenwich meridian and so it has become accepted practice for regional systems over the United States to describe their geostationary satellite locations in terms of degrees W. To an observer of a satellite standing at the subsatellite point, the satellite will appear to be directly overhead, in the zenith direction from the observing location. The zenith and nadir paths are therefore in opposite directions along the same path (see Figure 2.11). Designers of satellite antennas reference the pointing direction of the satellite’s antenna beams to the nadir direction. The communications coverage region on the earth from a satellite is defined by angles measured from nadir at the satellite to the edges of the coverage. Earth station antenna designers, however, do not reference their pointing direction to zenith. As noted earlier, they use the local horizontal plane at the earth station to define elevation angle and geographical compass points to define azimuth angle, thus giving the two look angles for the earth station antenna toward the satellite (Az, El).

Elevation Angle Calculation Figure 2.12 shows the geometry of the elevation angle calculation. In Figure 2.12, rs is the vector from the center of the earth to the satellite; re is the vector from the center of the earth to the earth station; and d is the vector from the earth station to the satellite. These three vectors lie in the same plane and form a triangle. The central angle measured between re and rs is the angle between the earth station and the

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Nadir direction

Zenith direction

Sub

C

FIGURE 2.11 Zenith and nadir pointing directions. The line joining the satellite and the center of the earth, C, passes through the surface of the earth at point Sub, the subsatellite point. The satellite is directly overhead at this point and so an observer at the subsatellite point would see the satellite at zenith (i.e., at an elevation angle of 90°). The pointing direction from the satellite to the subsatellite point is the nadir direction from the satellite. If the beam from the satellite antenna is to be pointed at a location on the earth that is not at the subsatellite point, the pointing direction is defined by the angle away from nadir. In general, two off-nadir angles are given: the number of degrees north (or south) from nadir; and the number of degrees east (or west) from nadir. East, west, north, and south directions are those defined by the geography of the earth.

Satellite

Local horizontal

d

rs

El ψ Earth station

re

γ Center of earth

FIGURE 2.12 The geometry of elevation angle calculation. The plane of the paper is the plane defined by the center of the earth, the satellite, and the earth station. The central angle is . The elevation angle El is measured upward from the local horizontal at the earth station.

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satellite, and is the angle (within the triangle) measured from re to d. Defined so that it is nonnegative, is related to the earth station north latitude Le (i.e., Le is the number of degrees in latitude that the earth station is north from the equator) and west longitude le (i.e., le is the number of degrees in longitude that the earth station is west from the Greenwich meridian) and the subsatellite point at north latitude Ls and west longitude ls by cos 1g2  cos 1Le 2 cos 1Ls 2 cos 1ls  le 2  sin 1Le 2 sin 1Ls 2

(2.31)

The law of cosines allows us to relate the magnitudes of the vectors joining the center of the earth, the satellite, and the earth station. Thus 1 2 re 2 re d  rs c 1  a b  2 a b cos 1g2 d rs rs

(2.32)

Since the local horizontal plane at the earth station is perpendicular to re, the elevation angle El is related to the central angle by El  c  90°

(2.33)

rs d  sin 1c2 sin 1g2

(2.34)

By the law of sines we have

Combining the last three equations yields cos 1El2  

rs sin 1g2 d

sin 1g2 1 2 re 2 re c 1  a b  2 a b cos 1g2 d rs rs

(2.35)

Equations (2.35) and (2.31) permit the elevation angle El to be calculated from knowledge of the subsatellite point and the earth station coordinates, the orbital radius rs, and the earth’s radius re. An accurate value for the average earth radius is 6378.137 km1 but a common value used in approximate determinations is 6370 km.

Azimuth Angle Calculation Because the earth station, the center of the earth, the satellite, and the subsatellite point all lie in the same plane, the azimuth angle Az from the earth station to the satellite is the same as the azimuth from the earth station to the subsatellite point. This is more difficult to compute than the elevation angle because the exact geometry involved depends on whether the subsatellite point is east or west of the earth station, and in which of the hemispheres the earth station and the subsatellite point are located. The problem simplifies somewhat for geosynchronous satellites, which will be treated in the next section. For the general case, in particular for constellations of LEO satellites, the tedium of calculating the individual look angles on a second-by-second basis has been considerably eased by a range of commercial software packages that exist for predicting a variety of orbital

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SIDEBAR A popular suite of software employed by many launch service contractors is that developed by Analytical Graphics: the Satellite Tool Kit3. The core program in early 2001, STK 4.0, and the subsequent subseries, was used by Hughes to rescue AsiaSat3 when that satellite was stranded in a highly elliptical orbit following the failure of an upper stage in

the launch vehicle. Hughes used two lunar flybys to provide the necessary additional velocity to circularize the orbit at geostationary altitude. A number of organizations offer web sites that provide orbital plots in a three-dimensional graphical format with rapid updates for a variety of satellites (e.g., the NASA site4).

dynamics and intercept solutions (see reference 13 for a brief review of 10 software packages available in early 2001).

Specialization to Geostationary Satellites For most geostationary satellites, the subsatellite point is on the equator at longitude ls, and the latitude Ls is 0. The geosynchronous radius rs is 42,164.17 km1. Since Ls is zero, Eq. (2.31) simplifies to cos 1g2  cos 1Le 2 cos 1ls  le 2

(2.36)

Substituting rs  42,164.17 km and re  6,378.137 km in Eqs. (2.32) and (2.35) gives the following expressions for the distance d from the earth station to the satellite and the elevation angle El at the earth station d  42,164.1731.02288235  0.30253825 cos 1g2 4 1 2 km sin 1g2 cos 1El2  31.02288235  0.30253825 cos 1g2 4 12

(2.37) (2.38)

For a geostationary satellite with an orbital radius of 42,164.17 km and a mean earth radius of 6378.137 km, the ratio rsre  6.6107345 giving El  tan1 3 16.6107345  cos g2 sin g4  g

(2.39)

To find the azimuth angle, an intermediate angle  must first be found. The intermediate angle  permits the correct 90° quadrant to be found for the azimuth since the azimuthal angle can lie anywhere between 0° (true north) and clockwise through 360° (back to true north again). The intermediate angle is found from a  tan1 c

tan 1ls  le 2 sin 1Le 2

d

(2.40)

Having found the intermediate angle , the azimuth look angle Az can be found from: Case 1:

Case 2:

Earth station in the Northern Hemisphere with (a) Satellite to the SE of the earth station: Az  180°   (b) Satellite to the SW of the earth station: Az  180°   Earth station in the Southern Hemisphere with (c) Satellite to the NE of the earth station: Az   (d) Satellite to the NW of the earth station: Az  360°  

(2.41a) (2.41b) (2.41c) (2.41d)

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rs

El re cos γ

Subsatellite point Earth station

re

γ FIGURE 2.13 The geometry of the visibility calculation. The satellite is said to be visible from the earth station if the elevation angle El is positive. This requires that the orbital radius rs be greater than the ratio re /cos( ) where re is the radius of the earth and is the central angle.

Center of earth

Visibility Test For a satellite to be visible from an earth station, its elevation angle El must be above some minimum value, which is at least 0°. A positive or zero elevation angle requires that (see Figure 2.13) rs

re cos 1g2

(2.42)

This means that the maximum central angular separation between the earth station and the subsatellite point is limited by re g cos1 a b rs

(2.43)

For a nominal geostationary orbit, the last equation reduces to 81.3° for the satellite to be visible. EXAMPLE 2.2.1 Geostationary Satellite Look Angles An earth station situated in the Docklands of London, England, needs to calculate the look angle to a geostationary satellite in the Indian Ocean operated by Intelsat. The details of the earth station site and the satellite are as follows: Earth station latitude and longitude are 52.0° N and 0°. Satellite longitude (subsatellite point) is 66.0° E.

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Step 1: Find the central angle

cos 1g2  cos 1Le 2 cos 1ls  le 2

 cos 152.02 cos 166.02  0.2504

yielding  75.4981° The central angle is less than 81.3° so the satellite is visible from the earth station. Step 2: Find the elevation angle El El  tan1 3 16.6107345  cos g2 sin g4  g

 tan1 3 16.6107345  0.25042 sin 175.49812 4  75.4981  5.847°

Step 3: Find the intermediate angle  a  tan1 c

tan 1ls  le 2 sin 1Le 2

d

 tan1 3 1tan 166.0  022 sin 152.02 4  70.667°

Step 4: Find the azimuth angle The earth station is in the Northern Hemisphere and the satellite is to the southeast of the earth station. From Eq. (2.41a), this gives Az  180°  a  180  70.667  109.333°1clockwise from true north2



Note that, in the example above, the elevation angle is relatively low (5.85°). Refractive effects in the atmosphere will cause the mean ray path to the satellite to bend in the elevation plane (making the satellite appear to be higher in the sky than it actually is) and to cause the amplitude of the signal to fluctuate with time. These aspects are discussed more fully in the propagation effects chapter. While it is unusual to operate to a satellite below established elevation angle minima (typically 5° at C band, 10° at Ku band, and in most cases, 20° at Ka band and above), many times it is not possible to do this. Such cases exist for high latitude regions and for satellites attempting to reach extreme east and west coverages from their given geostationary equatorial location. To establish whether a particular satellite location can provide service into a given region, a simple visibility test can be carried out, as shown earlier in Eqs. (2.42) and (2.43). A number of geosynchronous orbit satellites have inclinations that are much larger than the nominal 0.05° inclination maximum for current geosynchronous satellites. (In general, a geosynchronous satellite with an inclination of 0.1° may be considered to be geostationary.) In extreme cases, the inclination can be several degrees, particularly if the orbit maneuvering fuel of the satellite is almost exhausted and the satellite’s position in the nominal location is only controlled in longitude and not in inclination. This happens with most geostationary communications satellites toward the end of their operational lifetime since the reliability of the payload, or a large part of the payload, generally exceeds that of the lifetime of the maneuvering fuel. Those satellites that can no longer be maintained in a fully geostationary orbit, but are still used for communications services, are referred to as inclined orbit satellites. While they now need to have tracking antennas at the earth terminals once the inclination becomes too large to allow the satellite to remain within the 1-dB beamwidth of the earth station antennas, substantial additional revenue can be earned beyond the normal lifetime of the satellite. Those satellites that eventually reach significantly inclined orbits can also be used to communicate to parts of the high latitude regions that were once beyond reach, but only

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for a limited part of the day. The exceptional reliability of electronic components in space, once they have survived the launch and deployment sequences, has led spacecraft designers to manufacture satellites with two end-of-life criteria. These are: end of design life (EODL), which refers to the lifetime expectancy of the payload components and end of maneuvering life (EOML), which refers to the spacecraft bus capabilities, in particular the anticipated lifetime of the spacecraft with full maneuver capabilities in longitude and inclination. Current spacecraft are designed with fuel tanks that have a capacity that usually significantly exceeds the requirement for EODL. Once the final mass of the spacecraft (without fuel) is known, a decision can be made as to how much additional fuel to load so that the economics of the launch and the anticipated additional return on investment can be balanced. Having additional fuel on board the spacecraft can be advantageous for many reasons, in addition to adding on-orbit lifetime. In many cases, satellites are moved to new locations during their operational lifetime. Examples for this are opening up service at a new location with an older satellite or replacing a satellite that has had catastrophic failure with a satellite from a location that has fewer customers. Each maneuver, however, consumes fuel. A rule of thumb is that any change in orbital location for a geostationary satellite reduces the maneuvering lifetime by about 1 month. Moving the satellite’s location by 1° in longitude takes as much additional fuel as moving the location by 180°: both changes require an acceleration burn, a drift phase, and a deceleration burn. The 180° location change will clearly take longer, since the drift rates are the same in both cases. Another use for additional fuel is to allow for orbital perturbations at any location.

2.3

ORBITAL PERTURBATIONS The orbital equations developed in Section 2.1 modeled the earth and the satellite as point masses influenced only by gravitational attraction. Under these ideal conditions, a “Keplerian” orbit results, which is an ellipse whose properties are constant with time. In practice, the satellite and the earth respond to many other influences including asymmetry of the earth’s gravitational field, the gravitational fields of the sun and the moon, and solar radiation pressure. For low earth orbit satellites, atmospheric drag can also be important. All of these interfering forces cause the true orbit to be different from a simple Keplerian ellipse; if unchecked, they would cause the subsatellite point of a nominally geosynchronous satellite to move with time. Historically, much attention has been given to techniques for incorporating additional perturbing forces into orbit descriptions. The approach normally adopted for communications satellites is first to derive an osculating orbit for some instant in time (the Keplerian orbit the spacecraft would follow if all perturbing forces were removed at that time) with orbital elements (a, e, tp, , i, ). The perturbations are assumed to cause the orbital elements to vary with time and the orbit and satellite location at any instant are taken from the osculating orbit calculated with orbital elements corresponding to that time. To visualize the process, assume that the osculating orbital elements at time t0 are (a0, e0, tp, 0, i0, 0). Then assume that the orbital elements vary linearly with time at constant rates given by (dadt, dedt, etc.). The satellite’s position at any time t1 is then calculated from a Keplerian orbit with elements a0 

da de 1t  t0 2, e0  1t1  t0 2, etc. dt 1 dt

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39

This approach is particularly useful in practice because it permits the use of either theoretically calculated derivatives or empirical values based on satellite observations. As the perturbed orbit is not an ellipse, some care must be taken in defining the orbital period. Since the satellite does not return to the same point in space once per revolution, the quantity most frequently specified is the so-called anomalistic period: the elapsed time between successive perigee passages. In addition to the orbit not being a perfect Keplerian ellipse, there will be other influences that will cause the apparent position of a geostationary satellite to change with time. These can be viewed as those causing mainly longitudinal changes and those that principally affect the orbital inclination.

Longitudinal Changes: Effects of the Earth’s Oblateness The earth is neither a perfect sphere nor a perfect ellipse; it can be better described as a triaxial ellipsoid1. The earth is flattened at the poles; the equatorial diameter is about 20 km more than the average polar diameter. The equatorial radius is not constant, although the noncircularity is small: the radius does not vary by more than about 100 m around the equator1. In addition to these nonregular features of the earth, there are regions where the average density of the earth appears to be higher. These are referred to as regions of mass concentration or Mascons. The nonsphericity of the earth, the noncircularity of the equatorial radius, and the Mascons lead to a nonuniform gravitational field around the earth. The force on an orbiting satellite will therefore vary with position. For a low earth orbit satellite, the rapid change in position of the satellite with respect to the earth’s surface will lead to an averaging out of the perturbing forces in line with the orbital velocity vector. The same is not true for a geostationary (or geosynchronous) satellite. A geostationary satellite is weightless when in orbit. The smallest force on the satellite will cause it to accelerate and then drift away from its nominal location. The satellite is required to maintain a constant longitudinal position over the equator, but there will generally be an additional force toward the nearest equatorial bulge in either an eastward or a westward direction along the orbit plane. Since this will rarely be in line with the main gravitational force toward the earth’s center, there will be a resultant component of force acting in the same direction as the satellite’s velocity vector or against it, depending on the precise position of the satellite in the GEO orbit. This will lead to a resultant acceleration or deceleration component that varies with longitudinal location of the satellite. Due to the position of the Mascons and equatorial bulges, there are four equilibrium points in the geostationary orbit: two of them stable and two unstable. The stable points are analogous to the bottom of a valley, and the unstable points to the top of a hill. If a ball is perched on top of a hill, a small push will cause it to roll down the slope into a valley, where it will roll backwards and forwards until it gradually comes to a final stop at the lowest point. The satellite at an unstable orbital location is at the top of a gravity hill. Given a small force, it will drift down the gravity slope into the gravity well (valley) and finally stay there, at the stable position. The stable points are at about 75° E and 252° E and the unstable points are at around 162° E and 348° E1. If a satellite is perturbed slightly from one of the stable points, it will tend to drift back to the stable point without any thruster firings required. A satellite that is perturbed slightly from one of the unstable points will immediately begin to accelerate its drift toward the nearer stable point and, once it reaches this point, it will oscillate in longitudinal position about this point until (centuries later) it stabilizes at that point. These stable points are sometimes called the graveyard geosynchronous orbit locations (not to be confused with the graveyard orbit for a geosynchronous satellite, which is the orbit to which the satellite is raised once the

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satellite ceases to be useful). Note that, due to the nonsphericity of the earth, etc., the stable points are neither exactly 180° apart, nor are the stable and unstable points precisely 90° apart.

Inclination Changes: Effects of the Sun and the Moon The plane of the earth’s orbit around the sun—the ecliptic—is at an inclination of 7.3° to the equatorial plane of the sun (Figure 2.14). The earth is titled about 23° away from the normal to the ecliptic. The moon circles the earth with an inclination of around 5° to the equatorial plane of the earth. Due to the fact that the various planes—the sun’s equator, the ecliptic, the earth’s equator (a plane normal to the earth’s rotational axis), and the moon’s orbital plane around the earth—are all different, a satellite in orbit around the earth will be subjected to a variety of out-of-plane forces. That is, there will generally be a net acceleration force that is not in the plane of the satellite’s orbit, and this will tend to try to change the inclination of the satellite’s orbit from its initial inclination. Under these conditions, the orbit will precess and its inclination will change. The mass of the sun is significantly larger than that of the moon but the moon is considerably closer to the earth than the sun (see Table 2.2). For this reason, the acceleration force induced by the moon on a geostationary satellite is about twice as large as that of the sun. The net effect of the acceleration forces induced by the moon and the sun on a

Moon

Earth’s Eq. plane

5° Earth 23°

Satellite 7.3° Sun’s Eq. plane Sun

FIGURE 2.14 Relationship between the orbital planes of the sun, moon, and earth. The plane of the earth’s orbit around the sun is the ecliptic. The geostationary orbit plane (the earth’s equatorial plane) is about 23° out of the ecliptic, and leads to maximum out-ofgeostationary-orbit-plane forces at the solstice periods (approximately June 21 and December 21). The orbit of the moon is inclined about 5° to the earth’s equatorial plane. The moon revolves around the earth in 27.3 days, the earth (and the geostationary satellite) rotates once about 24 h, and the earth revolves around the sun every 365.25 days. In addition, the sun—which has a greater girth at the equator than at the poles—has its equator inclined about 7.3° to the ecliptic. All of these various angular differences and orbital periods lead to conditions where all of the out-of-plane gravitational forces are in one direction with respect to the equatorial (geostationary orbital) plane at a given time as well as to conditions where the various gravitational out-of-plane forces partially cancel each other out. The precessional forces that cause the inclination of the geostationary satellite’s orbit to move away from the equatorial plane therefore vary with time.

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TABLE 2.2

Comparative Data for the Sun, Moon, and Earth

Mean radius Sun Moon Earth

41

696,000 km 3,476 km 6,378.14 km

Mass

Mean orbit radius

Spin period

333,432 units 0.012 units 1.0 units

30,000 light years 384,500 km 149,597,870 km

25.04 earth days 27.3 earth days 1 earth day

The orbit radius refers to the center of the home galaxy (Milky Way) for the sun, center of earth for the moon, and center of the sun for the earth, respectively.

geostationary satellite is to change the plane of the orbit at an initial average rate of change of 0.85°/year from the equatorial plane1. When both the sun and moon are acting on the same side of the satellite’s orbit, the rate of change of the plane of the geostationary satellite’s orbit will be higher than average. When they are on opposite sides of the orbit, the rate of change of the plane of the satellite’s orbit will be less than average. Examples of maximum years are 1988 and 2006 (0.94°/year) and examples of minimum years are 1997 and 2015 (0.75°/year)1. These rates of change are neither constant with time nor with inclination. They are at a maximum when the inclination is zero and they are zero when the inclination is 14.67°. From an initial zero inclination, the plane of the geostationary orbit will change to a maximum inclination of 14.67° over 26.6 years. The acceleration forces will then change direction at this maximum inclination and the orbit inclination will move back to zero in another 26.6 years and out to 14.67° over a further 26.6 years, and so on. In some cases, to increase the orbital maneuver lifetime of a satellite for a given fuel load, mission planners deliberately place a satellite planned for geostationary orbit into an initial orbit with an inclination that is substantially larger than the nominal 0.05° for a geostationary satellite. The launch is specifically timed, however, so as to set up the necessary precessional forces that will automatically reduce the inclination “error” to close to zero over the required period without the use of any thruster firings on the spacecraft. This will increase the maneuvering lifetime of the satellite at the expense of requiring greater tracking by the larger earth terminals accessing the satellite for the first year or so of the satellite’s operational life. Under normal operations, ground controllers command spacecraft maneuvers to correct for both the in-plane changes (longitudinal drifts) and out-of-plane changes (inclination changes) of a satellite so that it remains in the correct orbit. For a geostationary satellite, this means that the inclination, ellipticity, and longitudinal position are controlled so that the satellite appears to stay within a “box” in the sky that is bounded by 0.05° in latitude and longitude over the subsatellite point. Some maneuvers are designed to correct for both inclination and longitude drifts simultaneously in the one burn of the maneuvering rockets on the satellite. In others, the two maneuvers are kept separate: one burn will correct for ellipticity and longitude drift; another will correct for inclination changes. The latter situation of separated maneuvers is becoming more common for two reasons. The first is due to the much larger velocity increment needed to change the plane of an orbit (the so-called north–south maneuver) as compared with the longitude/ellipticity of an orbit (the so-called east–west maneuver). The difference in energy requirement is about 10 :1. By alternately correcting for inclination changes and in-plane changes, the attitude of the satellite can be held constant and different sets of thrusters exercised for the required maneuver. The second reason is the increasing use of two completely different types of thrusters to control N–S maneuvers on the one hand and E–W maneuvers on the other. In the

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mid-1990s, one of the heaviest items that was carried into orbit on a large satellite was the fuel to raise and control the orbit. About 90% of this fuel load, once on orbit, was to control the inclination of the satellite. Newer rocket motors, particularly arc jets and ion thrusters, offer increased efficiency with lighter mass. In general, these low thrust, high efficiency rocket motors are used for N–S maneuvers leaving the liquid propellant thrusters, with their inherently higher thrust (but lower efficiency) for orbit raising and in-plane changes. In order to be able to calculate the required orbit maneuver for a given satellite, the controllers must have an accurate knowledge of the satellite’s orbit. Orbit determination is a major aspect of satellite control. EXAMPLE 2.3.1 Drift with a Geostationary Satellite A quasi-GEO satellite is in a circular equatorial orbit close to geosynchronous altitude. The quasiGEO satellite, however, does not have a period of one sidereal day: its orbital period is exactly 24 h—one solar day. Calculate (i) the radius of the orbit (ii) the rate of drift around the equator of the subsatellite point in degrees per (solar) day. An observer on the earth sees that the satellite is drifting across the sky. (iii) Is the satellite moving toward the east or toward the west? Answer Part (i) The orbital radius is found from Eq. (2.21), as in worked Example 2.2.1. Equation (2.21) gives the square of the orbital period in seconds (remembering that T here is one solar day) T 2  14p 2a3 2 m Rearranging the equation, the orbital radius a is given by a3  T 2m 14p2 2  186,4002 2  3.986004418  1054p2  7.5371216  1013 km3 a  42,241.095 km Part (ii) The orbital period of the satellite (one solar day) is longer than a sidereal day by 3 min 55.9 s  235.9 s. This will cause the subsatellite point to drift at a rate of 360°  235.986400 per day or 0.983° per day. Part (iii) The earth moves toward the east at a faster rate than the satellite, so the drift will appear to an observer on the earth to be toward the west. 

2.4

ORBIT DETERMINATION Orbit determination requires that sufficient measurements be made to determine uniquely the six orbital elements needed to calculate the future orbit of the satellite, and hence calculate the required changes that need to be made to the orbit to keep it within the nominal orbital location. Three angular position measurements are needed because there are six unknowns and each measurement will provide two equations. Conceptually, these can be thought of as one equation giving the azimuth and the other the elevation as a function of the six (as yet unknown) orbital elements. The control earth stations used to measure the angular position of the satellites also carry out range measurements using unique time stamps in the telemetry stream or communications carrier. These earth stations are generally referred to as the TTC&M (telemetry tracking command and monitoring) stations of the satellite network. Major satellite networks maintain their own TTC&M stations around the world. Smaller satellite systems generally contract for such TTC&M functions from the spacecraft manufacturer or from

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the larger satellite system operators, as it is generally uneconomic to build advanced TTC&M stations with fewer than three satellites to control. Chapter 3 discusses TTC&M systems.

2.5

LAUNCHES AND LAUNCH VEHICLES A satellite cannot be placed into a stable orbit unless two parameters that are uniquely coupled together—the velocity vector and the orbital height—are simultaneously correct. There is little point in obtaining the correct height and not having the appropriate velocity component in the correct direction to achieve the desired orbit. A geostationary satellite, for example, must be in an orbit at a height of 35,786.03 km above the surface of the earth (42,164.17-km radius from the center of the earth) with an inclination of zero degrees, an ellipticity of zero, and a velocity of 3074.7 m/s tangential to the earth in the plane of the orbit, which is the earth’s equatorial plane. The further out from the earth the orbit is, the greater the energy required from the launch vehicle to reach that orbit. In any earth satellite launch, the largest fraction of the energy expended by the rocket is used to accelerate the vehicle from rest until it is about 20 miles (32 km) above the earth. To make the most efficient use of the fuel, it is common to shed excess mass from the launcher as it moves upward on launch: this is called staging. Figure 2.15 gives a schematic of a Proton launch from the Russian Baikonur complex at Kazakhstan, near Tyuratam. Most launch vehicles have multiple stages and, as each stage is completed, that portion of the launcher is expended until the final stage places the satellite into the desired trajectory. Hence the term: expendable launch vehicle (ELV). The Space Shuttle, called the Space Transportation System (STS) by NASA, is partially reusable. The solid rocket boosters are recovered and refurbished for future missions and the shuttle vehicle itself is flown back to earth for refurbishment and reuse. Hence the term: reusable launch vehicle (RLV) for such launchers. More advanced launch vehicles are being developed that would provide both single stage to orbit (SSTO) and RLV capabilities. The NASA series of X-33 and X-34 test vehicles form the public portion of this quest (see the NASA home page4).

25:00 4th 10:00 3rd stage stage roll/allign separation

05:34 Payload fairing jettison

00:21 Roll 02:07 1st stage Separation/ 2nd stage ignition

05:41 2nd stage separation 06:10 3rd stage ignition

Major Events from GTO to final User Handoff 1:27:00 3:59:10 6:58:00 7:09:20 7:09:50 7:10+

Reach GTO Completion of programed turns Completion of the compensation turn Second 4th stage ignition (2 sec) Spacecraft separates from 4th stage, GEO Handoff to User

Lift-off

FIGURE 2.15 Schematic of a Proton launch (after reference 5).

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There are also a number of private ventures that aim to achieve RLV capabilities in the first decade of the twenty-first century. Two excellent web sites to keep abreast of these, and related space issues, are those maintained by Spaceviews6 and Orbreport7. Of equal importance to the orbital height the satellite is intended for is the inclination of the orbit that the spacecraft needs to be launched into. The earth spins toward the east. At the equator, the rotational velocity of a sea level site in the plane of the equator is (2  radius of the earth)(one sidereal day)  0.4651 km/s. This velocity increment is approximately 1000 mph (1610 km/h). An easterly launch from the equator has a velocity increment of 0.465 km/s imparted by the rotation of the earth. A satellite in a circular, equatorial orbit at an altitude of 900 km requires an orbital velocity of about 7.4 km/s tangential to the surface of the earth. A rocket launched from the equator needs to impart an additional velocity of (7.4  0.47) km/s  6.93 km/s: in other words, the equatorial launch has reduced the energy required by about 6%. This equatorial launch “bonus” led to the concept of a sea launch by Hughes and Boeing8. If the launch is not to be into an equatorial orbit, the payload capabilities of any given rocket will reduce as the inclination increases. A satellite launched into a prograde orbit from a latitude of degrees will enter an orbit with an inclination of degrees to the equator. If the satellite is intended for geostationary orbit, the satellite must be given a significant velocity increment to reorient the orbit into the earth’s equatorial plane. For example, a satellite launched from Cape Canaveral at 28.5° N latitude requires a velocity increment of 366 m/s to attain an equatorial orbit from a geosynchronous orbit plane of 28.5°. Ariane is launched from the Guiana Space Center in French Guiana, located at latitude 5° S in South America, and SeaLaunch can launch from the equator. The lower latitude of these launch sites results in significant savings in the fuel used by the apogee kick motor (AKM).

Expendable Launch Vehicles (ELVs) 1998 was an important year for ELVs: it was the year when the number of commercial launches in the United States surpassed the number of government launches for the first time9. The gap between commercial and government launches will continue to grow. The Teal Group estimated in mid-1999 that 1447 satellites would be launched worldwide between 2000 and 2009 on 850 to 900 launch vehicles10. At an average cost of $100 M per launch, this represents a business worth about $ 90 B over 10 years. Of these 1447 satellites, 893 were considered commercial ventures with the remainder split between military and civilian government spacecraft. There is therefore a healthy market for ELVs and a number of companies, consortia, and national entities are seeking to enter this expanding field. Reference 15 contains a good survey of the ELVs being developed for the twentyfirst century. Figure 2.16 presents a rough comparison between the main launch vehicles

SIDEBAR The STS can launch approximately 65,000 lb. (29,478 kg) into a standard 28.5° orbital inclination at an orbital height of about 200 km from the Kennedy Space Flight Center in Cape Canaveral. If the Vandenburg Air Force Base launch site in California still had the capability of launching the Shuttle, the payload capability of the Shuttle for a polar launch (inclination 90°) would be reduced to 32,000 lb (14,512 kg). Since the Challenger accident in January 1996, the

shuttle is rarely used to launch civilian payloads, its mission being confined to military payloads [e.g., TDRSS (Tracking and Data Relay Satellite System) satellites], joint ventures with other agencies [e.g., ESA (European Space Agency) Spacelab facility], “big science” missions (e.g., the X-ray telescope Chandra), and International Space Station flights. The vast majority of the satellite launches are therefore conducted by expendable launch vehicles.

$95

$105

$85

$95

$110

–

$120

ARIANE 44P

ATLAS IIAS

ARIANE 42L ARIANE 44LP ARIANE 44L

TITAN III

ARIANE 5

2,050 (4,520) 3.65 (12) KOUROU

2.54 (8.3)

CCAS (VAFB)

Payload Max Diameter m (ft)

Launch Site (Projected)

1990

1,820 (1) (4,010)

CCAS (VAFB)

3.65 (12)

2,810 (2) (6,200)

1991

CCAS (VAFB)

3.65 (12)

3,050 (2) (6,710)

1992

KOUROU

3.65 (12)

2,840 (6,260)

1991

KOUROU

3.65 (12)

3,320 (7,320)

1991

CCAS (VAFB)

3.65 (12)

3,700 (2) (8,150)

1993

KOUROU

3.65 (12)

3,380 (7,450)

1993

KOUROU

3.65 (12)

4,060 (8,950)

1988

KOUROU

3.65 (12)

4,520 (9,965)

1989

CCAS

3.65 (12)

5,000 (1) (11,000)

1989

FIGURE 2.16 Representative ELVs (after reference 5). CCAS, Cape Canaveral Air Station; VAFB, Vandenburg Air Force Base.

(1) DELTA & TITAN @ i = 28 deg (2) ATLAS GTO REFERENCE ORBIT IS 167 × 35,788 KM (90 × 19,324 NM), i = 27 deg

200 × 35,786 Km (108 × 19,323 NM)

GTO, i = 7 deg

Performance Kg (lb)

1990

KOUROU

4.57 (15)

5,000-6,700 (11-13,900)

1996

2 ARIANE 42P

2 ATLAS IIA

4

4

ATLAS II

6

6

ARIANE 40

8

8

DELTA II-7925

10

10

First Flight

PERFORMANCE × 1000 Lb to GTO (i = 7°)

12

$75

12

$90

14

$80

LAUNCH 14 VEHICLES

$60

16

$50

16

1996 Market Prices, $M

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$M 200 H-2

180 160 140

Atlas IIAR

120

13.9

Atlas IIAS

100

Atlas II

Atlas IIA

Ariane 44L

Ariane 5

Delta III

80

Zenit 3

60

Long March 3B

40

Proton D-1-e

Delta II

20 0 4

6

8 10 Pounds into Geostationary Transfer Orbit

12

FIGURE 2.17 Launch vehicle market price vs performance, 1996 prices (after reference 5). The launch vehicles have been normalized to a launch into geostationary transfer orbit at an inclination of 28°. The trend line for launchers is shown as $12,000 per pound. Note that Long March, Zenit, and Proton are well below this trend line, mainly due to aggressive pricing objectives to break into a market long dominated by U.S. and European launchers.

used for Geostationary Transfer Orbit (GTO) injection during the 1990s, plus the Ariane 5 launcher. The 1996 pricing of these vehicles is shown in Figure 2.17. Not included in these data are the advanced Chinese launch vehicles being developed for both unmanned and manned missions in the twenty-first century. The largest of these Chinese launch vehicles rivals the Ariane 5 vehicle with a geostationary transfer orbit capability of 26,000 lb. TABLE 2.3 Some Next Generation Launchers Compared with Ariane 44 and Atlas IIAS Baseline Vehicles (1999 Prices)

Launcher Ariane 44 Ariane 5 Atlas IIAS Atlas IIIA Atlas IIIB Atlas V Delta III Delta IV(small) Delta IV(med.) Delta IV(heavy) Titan III Titan IV Proton M

Weight to orbit (kg) 4000 6800 3700 4120 4500 6500 3800 2177 4173 13200 4500 5700 4800

Total cost ($M) 130 120 100 125 135 150* 130 60* 120* 400* 260 435 80

Lead time (months)

Max. payload diameter (m)

Launch latitude (°)

36 36 36 36 48 48 36 36* 36* 48* 36 48 24

3.65 4.57 3.45 4.19 4.19 5.40 4.00 3.00 4.00 5.00 3.65 4.57 3.68

5.2 5.2 28.5 28.5 28.5 28.5 28.7 28.7 28.7 28.7 28.6 28.6 51.6

* These data are estimated values. The Atlas V and Proton M vehicles are planned for operational flights beginning in 2002 or 2003. The Delta IV family of launch vehicles will become operational from 2002 to 2004.

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TABLE 2.4

47

Some Launch Vehicle Selection Factors

Price/cost Reliability Recent launch success/failure history Dependable launch schedule Urgency of your launch requirement Performance Spacecraft fit to launcher (size, acoustic, and vibration environment) Flight proven (see recent launch history) Safety issues Launch site location Availability What is the launcher backlog of orders? What is the launch site backlog of launches? Market issues What will the market bear at this particular time?

It can be seen from Figure 2.17 that there was a well-established trend line of about $25,000 per kg into GTO prior to the introduction of the Chinese Long March and the Russian Zenit and Proton vehicles. The pricing of the Chinese and Russian launchers reflected an aggressive marketing strategy to break into the launch services field. Ariane 5 was the first of the next-generation launchers aimed at both large, single payloads into GTO and multiple payload injection into LEO and MEO. Some more next-generation launchers are shown in Table 2.3 on the previous page. It is anticipated that the bulk of the large satellite launches will be conducted with Atlas V, Delta IV, and Ariane vehicles and their Russian and Chinese equivalents over the first 2 decades of the twenty-first century. The decision on which particular rocket to use in a given situation will depend on a variety of factors. Some of these are set out in Table 2.4. The decision-making routine using the above criteria is shown in Figure 2.18.

Launch Vehicle Selection Factors

• Cost to manufacturer • “Performance”, or throw-weight to orbit • Reliability • Schedule dependability • Market forces • Insurance

• Price/cost • Reliability – Recent failures • Dependable launch schedule – Urgency of the customer • Performance • Spacecraft fit • Flight proven • Safety • Launch site location • Availability—Launch site; vehicle; schedule; • Market conditions—What the market will bear

FIGURE 2.18 Schematic of the decision making process to select a rocket for a given satellite requirement (after reference 5).

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Apogee: AKM fires at this point

GEO LEO

GTO

Perigee: GTO insertion starts here

FIGURE 2.19 Illustration of the GTO/AKM approach to geostationary orbit (not to scale). The combined spacecraft and final rocket stage are placed into low earth orbit (LEO) around the earth. After careful orbit determination measurements, the final stage is ignited in LEO and the spacecraft inserted into a transfer orbit that lies between the LEO and the geostationary orbit altitude: the so-called geostationary transfer orbit or GTO. Again, after more careful orbit determination, the apogee kick motor (AKM) is fired on the satellite and the orbit is both circularized at geostationary altitude and the inclination reduced to close to zero. The satellite is then in GEO.

Some of the launch vehicles deliver the spacecraft directly to geostationary orbit (called a direct-insertion launch) while others inject the spacecraft into a geostationary transfer orbit (GTO). Spacecraft launched into GTO must carry additional rocket motors and/or propellant to enable the vehicle to reach the geostationary orbit. There are three basic ways to achieve geostationary orbit.

Placing Satellites into Geostationary Orbit Geostationary Transfer Orbit and AKM The initial approach to launching geostationary satellites was to place the spacecraft, with the final rocket stage still attached, into low earth orbit. After a couple of orbits, during which the orbital elements are measured, the final stage is reignited and the spacecraft is launched into a geostationary transfer orbit. The GTO has a perigee that is the original LEO orbit altitude and an apogee that is the GEO altitude. Figure 2.19 illustrates the process. The position of the apogee point is close to the orbital longitude that would be the in-orbit test location of the satellite prior to it being moved to its operational position. Again, after a few orbits in the GTO while the orbital elements are measured, a rocket motor (usually contained within the satellite itself) is ignited at apogee and the GTO is raised until it is a circular, geostationary orbit. Since the rocket motor fires at apogee, it is commonly referred to as the apogee kick motor

SIDEBAR The first successful GEO satellite was Syncom, launched in 1963. Hughes Corporation built the satellite and the spacecraft was spin-stabilized while it was in geostationary transfer orbit. In this way, the satellite was correctly aligned for the apogee motor firing. The apogee motor was fairly powerful and the apogee burn was only for a few minutes. During this apogee burn, all of the satel-

lite’s deployable elements (e.g., solar panels, antennas) were stowed and locked in place to avoid damage while the AKM accelerated the satellite to GEO. Hughes patented the technique of spin stabilizing the spacecraft in GTO. To avoid infringing this patent, other satellite manufacturers developed a new way to achieve GEO, known as a slow orbit raising technique.

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GTO LEO

Successive orbit raisings from GTO

GEO

49

FIGURE 2.20 Illustration of slow orbit raising to geostationary orbit (not to scale). The combined spacecraft and final rocket stage are placed into low earth orbit (LEO) around the earth. As before (see Figure 2.19), the spacecraft is injected into GTO but, in this case, once the satellite is ejected from the final rocket stage, it deploys many of the elements that it will later use in GEO (solar panels, etc.) and stabilizes its attitude using thrusters and momentum wheels, rather than being spin-stabilized. The higher power thrusters are then used around the apogee to raise the perigee of the orbit until the orbit is circular at the GEO altitude. At the same time as the orbit is being raised, the thruster firings will be designed gradually to reduce the inclination to close to zero.

(AKM). The AKM is used both to circularize the orbit at GEO and to remove any inclination error so that the final orbit of the satellite is very close to geostationary. Geostationary Transfer Orbit with Slow Orbit Raising In this procedure, rather than employ an apogee kick motor that imparts a vigorous acceleration over a few minutes, the spacecraft thrusters are used to raise the orbit from GTO to GEO over a number of burns. Since the spacecraft cannot be spin-stabilized during the GTO (so as not to infringe the Hughes patent), many of the satellite elements are deployed while in GTO, including the solar panels. The satellite has two power levels of thrusters: one for more powerful orbit raising maneuvers and one for on-orbit (low thrust) maneuvers. Since the thrusters take many hours of operation to achieve the geostationary orbit, the perigee of the orbit is gradually raised over successive thruster firings. The thruster firings occur symmetrically about the apogee although they could occur at the perigee as well. The burns are typically 60 to 80 min long on successive orbits and up to six orbits can be used. Figure 2.20 illustrates the process. In the first two cases, AKM and slow orbit raising, the GTO may be a modified orbit with the apogee well above the required altitude for GEO. The excess energy of the orbit due to the higher-than-necessary altitude at apogee can be traded for energy required to raise the perigee. The net energy to circularize the orbit at GEO is therefore less and the satellite can retain more fuel for on-orbit operations. Direct Insertion to GEO This is similar to the GTO technique but, in this case, the launch service provider contracts to place the satellite into GEO. The final stages of the rocket are used to place the satellite directly into GEO rather than the satellite using its own propulsion system to go from GTO to GEO.

2.6 ORBITAL EFFECTS IN COMMUNICATIONS SYSTEMS PERFORMANCE Doppler Shift To a stationary observer, the frequency of a moving radio transmitter varies with the transmitter’s velocity relative to the observer. If the true transmitter frequency (i.e., the

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frequency that the transmitter would send when at rest) is fT, the received frequency fR is higher than fT when the transmitter is moving toward the receiver and lower than fT when the transmitter is moving away from the receiver. Mathematically, the relationship [Eq. (2.44a)] between the transmitted and received frequencies is fR  fT ¢f VT   vp fT fT

(2.44a)

¢f  VT fTc  VTl

(2.44b)

or where VT is the component of the transmitter velocity directed toward the receiver, vp  c the phase velocity of light (2.9979  108  3  108 m/s in free space), and is the wavelength of the transmitted signal. If the transmitter is moving away from the receiver, then VT is negative. This change in frequency is called the Doppler shift, the Doppler effect, or more commonly just “Doppler” after the German physicist who first studied the phenomenon in sound waves. For LEO satellites, Doppler shift can be quite pronounced, requiring the use of frequency-tracking receivers. For geostationary satellites, the effect is negligible. EXAMPLE 2.6.1 Doppler Shift for a LEO Satellite A low earth orbit satellite is in a circular polar orbit with an altitude, h, of 1000 km. A transmitter on the satellite has a frequency of 2.65 GHz. Find (i) The velocity of the satellite in orbit (ii) The component of velocity toward an observer at an earth station as the satellite appears over the horizon, for an observer who is in the plane of the satellite orbit. (iii) Hence, find the Doppler shift of the received signal at the earth station. Use a mean earth radius value, re, of 6378 km. The satellite also carries a Ka-band transmitter at 20.0 GHz. (iv) Find the Doppler shift for this signal when it is received by the same observer. Answer

Part (i) The period of the satellite is found from Eq. (2.21): T 2  14p2a3 2 m

T 2  4p 2  16378  10002 33.986004418  10 5  3.977754  107 s2

T  6306.94 s The circumference of the orbit is 2a  46,357.3 km so the velocity of the satellite in orbit is vs where vs  46,357.36306.94  7.350 km/s Part (ii) The component of velocity toward an observer in the plane of the orbit as the satellite appears over the horizon is given by vr  vs cos , where  is the angle between the satellite velocity vector and the direction of the observer at the satellite. The angle  can be found from simple geometry to be cos u  re  1re  h2  63787378  0.8645 Hence the component of satellite velocity toward the observer is vr  vs cos u  6.354 km/s  6354 m/s Part (iii) The Doppler shift of the received signal is given by Eq. (2.44b). Hence, for this satellite and observer, with a transmitter frequency of 2.65 GHz,  0.1132 m, and the Doppler shift

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in the received signal is ¢f  VTl  63540.1132  56,130 Hz  56.130 kHz Part (iv) A Ka-band transmitter with frequency 20.0 GHz has a wavelength of 0.015 m. The corresponding Doppler shift at the receiver is ¢f  VTl  63540.015  423.60 kHz Doppler shift at Ka band with a LEO satellite can be very large and requires a fast frequencytracking receiver. Ka-band LEO satellites are better suited to wideband signals than narrowband voice communications. 

Range Variations Even with the best station-keeping systems available for geostationary satellites, the position of a satellite with respect to the earth exhibits a cyclic daily variation. The variation in position will lead to a variation in range between the satellite and user terminals. If time division multiple access (TDMA) is being used, careful attention must be paid to the timing of the frames within the TDMA bursts (see Chapter 6) so that the individual user frames arrive at the satellite in the correct sequence and at the correct time. Range variations on LEO satellites can be significant, as can path loss variations. While guard times between bursts can be increased to help in any range and/or timing inaccuracies, this reduces the capacity of the transponder. The on-board capabilities of some satellites permit both timing control of the burst sequence and power level control of individual user streams.

Solar Eclipse A satellite is said to be in eclipse when the earth prevents sunlight from reaching it, that is, when the satellite is in the shadow of the earth. For geostationary satellites, eclipses occur during two periods that begin 23 days before the equinoxes (about March 21 and about September 23) and end 23 days after the equinox periods. Figure 2.21 from reference 11 and Figure 2.22 from reference 12 illustrate the geometry and duration of the eclipses. Eclipses occur close to the equinoxes, as these are the times when the sun, the earth, and the satellite are all nearly in the same plane. During full eclipse, a satellite receives no power from its solar array and it must operate entirely from its batteries. Batteries are designed to operate with a maximum depth of discharge; the better the battery, the lower the percentage depth of discharge can be. If the battery is discharged below its maximum depth of discharge, the battery may not recover to full operational capacity once recharged. The depth of discharge therefore sets the power drain limit during eclipse operations. Nickel–Hydrogen batteries, long the mainstay of communications satellites, can operate at about a 70% depth of discharge and recover fully once recharged. Ground controllers perform battery-conditioning routines prior to eclipse operations to ensure the best battery performance during the eclipse. The routines consist of deliberately discharging the batteries until they are close to their maximum depth of discharge, and then fully recharging the batteries just before eclipse season begins. The eclipse season is a design challenge for spacecraft builders. Not only is the main power source withdrawn (the sun) but also the rapidity with which the satellite enters and exits the shadow can cause extreme changes in both power and heating effects over relatively short periods. Just like a common light bulb is more likely to fail when the current is switched on as opposed to when it is under steady state conditions, satellites can suffer many of their component failures under sudden stress situations. Eclipse periods are

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Satellite in sun transit outage Satellite in eclipse N Sun

Earth shadow R

Earth

S

C

A

Sun

-ray at e direct quin ion ox

D

Geostationary orbit FIGURE 2.21 Eclipse geometry (Source: J. J. Spilker, Jr., Digital Communications by Satellite, Prentice Hall, p. 144, copyright © 1977, Pearson Education, Upper Saddle River, NJ, reprinted with permission). During the equinox periods around the March 21 and September 23, the geostationary plane is in the shadow of the earth on the far side of the earth from the sun. As the satellite moves around the geostationary orbit, it will pass through the shadow and undergo an eclipse period. The length of the eclipse period will vary from a few minutes to over an hour (see Figure 2.22), depending on how close the plane of the geostationary orbit is with respect to the center of the shadow thrown by the earth.

80

60

70

Day of the year 80 90 100

110

Full shadow Half shadow

70

240 80

50 40 30

Day of the year 260 270 280

290

70

50 40 30

20

20

10

10

0

250

60 Eclipse time (min.)

60 Eclipse time (min.)

52

1

11 March

21

31

10 April

20

0 28 7 August

17 27 7 17 September October Date Date FIGURE 2.22 Dates and duration of eclipses. (Source: James Martin, Communications Satellite Systems, Prentice Hall, p. 37, copyright © 1978 Pearson Education, Upper Saddle River, NJ. Reprinted with permission.)

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therefore monitored carefully by ground controllers, as this is when most of the equipment failures are likely to occur.

Sun Transit Outage During the equinox periods, not only does the satellite pass through the earth’s shadow on the “dark” side of the earth, but the orbit of the satellite will also pass directly in front of the sun on the sunlit side of the earth (Figure 2.23). The sun is a “hot” microwave source with an equivalent temperature of about 6000 to 10,000 K, depending on the time within the 11-year sunspot cycle, at the frequencies used by communications satellites (4 to 50 GHz). The earth station antenna will therefore receive not only the signal from the satellite but also the noise temperature transmitted by the sun. The added noise temperature will cause the fade margin of the receiver to be exceeded and an outage will occur. These outages may be precisely predicted. For satellite system operators with more

The sun

Geostationary orbit

Communications signal

Thermal noise from the sun

Earth station FIGURE 2.23 Schematic of sun outage conditions. During the equinox periods, not only does the earth’s shadow cause eclipse periods to occur for geostationary satellites, during the sunlit portion of the orbit, there will be periods when the sun appears to be directly behind the satellite. At the frequencies used by communications satellites (4 to 50 GHz), the sun appears as a hot noise source. The effective temperature of the sun at these frequencies is on the order of 10,000 K. The precise temperature observed by the earth station antenna will depend on whether the beamwidth partially, or completely, encloses the sun.

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than one satellite at their disposal, traffic can be off-loaded to satellites that are just out of, or are yet to enter, a sun outage. The outage in this situation can therefore be limited as far as an individual user is concerned. However, the outages can be detrimental to operators committed to operations during daylight hours.

2.7

SUMMARY

Newton’s laws of motion explain the forces on a satellite in orbit. The balance between the force pulling a satellite inward to the earth—i.e. gravity—and that trying to fling a satellite away from the earth—kinetic energy—is a fine one. To achieve stable orbit, a satellite must have the correct velocity, be traveling in the right direction, and be at the right height for its velocity. As the orbital height increases, the gravitational acceleration decreases, the orbital velocity decreases, and the period of the satellite increases. Calculation procedures for obtaining the period of a satellite and its velocity are set out. It is seen that Kepler’s constant, the product of the universal gravitational constant, G, and the mass of the earth ME, is fundamental to many of the equations that give the forces on the satellite and the velocity of the satellite in its orbit. Kepler’s three laws describing the motion of one body orbiting another are given and the terminology employed in satellite ephemeris data is explained. The relationship between the astronomers’ use of Julian dates and Julian days and the Universal Time Constant

(UTC), otherwise referred to as GMT, is given. The use of Julian days, which begin at noon, was introduced by astronomers to allow them to make observations overnight without having the day change on them (as normal UTC days do at midnight). Locating the satellite in its orbit is a complex process, with a number of possible frames of reference. Different approaches are discussed. Procedures for calculating the look angles from the earth to a geostationary satellite are given. The natural forces that act on a satellite to cause orbital perturbations are set out and the need for orbital maneuvers explained. The important difference between orbital maneuver life and orbital design life is explained. Details on launch procedures and launch vehicles are provided, with typical launch campaign information set out. The two basic methods of launching geostationary satellites are described, one using an apogee kick motor and the other a slow orbit raising technique. Finally, Doppler shift, range variations, solar eclipse, and sun transit outage are reviewed.

REFERENCES 1. GARY D. GORDON and WALTER L. MORGAN, Principles of Communications Satellites, John Wiley & Sons, ISBN 0-471-55796-X, 1993. 2. The American Ephemeris and Nautical Almanac, U.S. Government Printing Office, Washington, DC (published annually). 3. http://www.stk.com 4. The NASA liftoff home page is http://liftoff.msfc.nasa. gov/realtime/JTrack/Spacecraft.html The home page allows you to see the International Space Station, weather and research satellites, and the Shuttle track if it is in orbit. The page specializing in three-dimensional graphical views of satellites is http://liftoff.msfc.nasa. gov/realtime/jtrack/3d/Jtrack3d.html 5. Private communication, EE 4644 Spring 1997, DAVID WALSH and CLIF GROVES. 6. http://www.spaceviews.com 7. http://www.orbreport.com is a site dedicated to the space transportation industry and is an element of ISIR, International Space Industry Report. 8. K. ROUNDTREE, “Launching Payloads by Sea,” Launchspace, pp. 38–39, May/June 1999; see also in the Hughes web site at http://www.hcisat.com

9. R. DEKOK, “Spacelift in and beyond the next millennium,” Launchspace, p. 6, May/June 1999. 10. As reported in [email protected] in July 1999, the biweekly update from the web site given in reference 6. 11. J. J. SPILKER, Jr., Digital Communications by Satellite, Prentice-Hall, Englewood Cliffs, NJ, 1977. 12. JAMES MARTIN, Communications Satellite Systems, Prentice-Hall, Englewood Cliffs, NJ, 1978. 13. D. M. RUSSELL, “Browsing orbital analysis tools,” Launchspace, Vol. 3, No. 6, p. 24, December 1, 1998. 14. JAMES R. WERTZ and WILEY J. LARSON, eds., Space Mission Analysis and Design, 3rd Ed., Kluwer Academic Publishers, 2001. ISBN 0-7923-5901-1. 15. Aviation Week and Space Technology, Vol. 151, No. 24, December 13, 1999. Special issue on 21st century launch vehicles. 16. “Chinese Rockets and R&D Advances.” Aviation Week and Space Technology, Vol. 155, No. 20, pp. 54 – 55, November 12, 2001.

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PROBLEMS 1. Explain what the terms centrifugal and centripetal mean with regard to a satellite in orbit around the earth. A satellite is in a circular orbit around the earth. The altitude of the satellite’s orbit above the surface of the earth is 1400 km. (i) What are the centripetal and centrifugal accelerations acting on the satellite in its orbit? Give your answer in m/s. (ii) What is the velocity of the satellite in this orbit? Give your answer in km/s. (iii) What is the orbital period of the satellite in this orbit? Give your answer in hours, minutes, and seconds. Note: Assume the average radius of the earth is 6378.137 km and Kepler’s constant has the value 3.986004418  105 km3/s2. 2. A satellite is in a 322-km high circular orbit. Determine: a. The orbital angular velocity in radians per second; b. The orbital period in minutes; and c. The orbital velocity in meters per second. Note: Assume the average radius of the earth is 6378.137 km and Kepler’s constant has the value 3.986004418  105 km3/s2. 3. The same satellite in Problem 2 above (322-km circular orbit) carries a 300-MHz transmitter. a. Determine the maximum frequency range over which the received signal would shift due to Doppler effects if received by a stationary observer suitably located in space. Note: The frequency can be shifted both up and down, depending on whether the satellite is moving toward or away from the observer. You need to determine the maximum possible change in frequency due to Doppler (i.e., 2 f). b. If an earth station on the surface of the earth at mean sea level, 6370 km from the center of the earth, can receive the 300-MHz transmissions down to an elevation angle of 0°, calculate the maximum Doppler shift that this station will observe. Note: Include the earth’s rotation and be sure you consider the maximum possible Doppler shift for a 322-km circular orbit. 4. What are Kepler’s three laws of planetary motion? Give the mathematical formulation of Kepler’s third law of planetary motion. What do the terms perigee and apogee mean when used to describe the orbit of a satellite orbiting the earth? A satellite in an elliptical orbit around the earth has an apogee of 39,152 km and a perigee of 500 km. What is the orbital period of this satellite? Give your answer in hours. Note: Assume the average radius of the earth is 6378.137 km and Kepler’s constant has the value 3.986004418  10 5 km3/s2.

5. An observation satellite is to be placed into a circular equatorial orbit so that it moves in the same direction as the earth’s rotation. Using a synthetic aperture radar system, the satellite will store data on surface barometric pressure, and other weather related parameters, as it flies overhead. These data will later be played back to a controlling earth station after each trip around the world. The orbit is to be designed so that the satellite is directly above the controlling earth station, which is located on the equator, once every 4 h. The controlling earth station’s antenna is unable to operate below an elevation angle of 10° to the horizontal in any direction. Taking the earth’s rotational period to be exactly 24 h, find the following quantities: a. The satellite’s angular velocity in radians per second. b. The orbital period in hours. c. The orbital radius in kilometers. d. The orbital height in kilometers. e. The satellite’s linear velocity in meters per second. f. The time interval in minutes for which the controlling earth station can communicate with the satellite on each pass. 6. What is the difference, or are the differences, between a geosynchronous satellite and a geostationary satellite orbit? What is the period of a geostationary satellite? What is the name given to this orbital period? What is the velocity of a geostationary satellite in its orbit? Give your answer in km/s. A particular shuttle mission released a TDRSS satellite into a circular low orbit, with an orbital height of 270 km. The shuttle orbit was inclined to the earth’s equator by approximately 28°. The TDRSS satellite needed to be placed into a geostationary transfer orbit (GTO) once released from the shuttle cargo bay, with the apogee of the GTO at geostationary altitude and the perigee at the height of the shuttle’s orbit. (i) What was the eccentricity of the GTO? (ii) What was the period of the GTO? (iii) What was the difference in velocity of the satellite in GTO between when it was at apogee and when it was at perigee? Note: Assume the average radius of the earth is 6378.137 km and Kepler’s constant has the value 3.986004418  105 km3/s2. 7. For a variety of reasons, typical minimum elevation angles used by earth stations operating in the commercial fixed services using satellites (FSS) communications bands are as follows: C band 5°; Ku band 10°; and Ka band 20°. (i) Determine the maximum and minimum range in kilometers from an earth station to a geostationary satellite in the three bands. (ii) To what round-trip signal propagation times do these ranges correspond?

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You may assume the signal propagates with the velocity of light in a vacuum even when in the earth’s lower atmosphere. 8. Most commercial geostationary communications satellites must maintain their orbital positions to within 0.05° of arc. If a geostationary satellite meets this condition (i.e., it has an apparent motion 0.05° of arc N–S and 0.05° of arc E–W, as measured from the center of the earth), calculate the maximum range variation to this satellite from an earth station with a mean elevation angle to the center of the satellite’s apparent motion of 5°. You may assume that the equatorial and polar diameters of the earth are the same. 9. An interactive experiment is being set up between the University of York, England (approximately 359.5° E, 53.5° N) and the Technical University of Graz, Austria (approximately 15° E, 47.5° N) that will make use of a geostationary satellite. The earth stations at both universities are constrained to work only above elevation angles of 20° due to buildings, etc., near their locations. The groups at the two universities need to find a geostationary satellite that will be visible to both universities simultaneously, with both earth stations operating at, or above, an elevation angle of 20°. What is the range of sub-satellite points between which the selected geostationary satellite must lie? 10. The state of Virginia may be represented roughly as a rectangle bounded by 39.5° N latitude on the north, 36.5° N latitude on the south, 76.0° W longitude on the east, and 86.3° W longitude on the west. If a geostationary satellite must be visible throughout Virginia at an elevation angle no lower than 20°, what is the range of longitudes within which the subsatellite point of the satellite must lie? 11. A geostationary satellite system is being built which incorporates intersatellite links (ISLs) between the satellites. This permits the transfer of information

between two earth stations on the surface of the earth, which are not simultaneously visible to any single satellite in the system, by using the ISL equipment to link up the satellites. In this question, the effects of ray bending in the atmosphere may be ignored, processing delays on the satellites may initially be assumed to be zero, the earth may be assumed to be perfectly circular with a flat (i.e., not hilly) surface, and the velocity of the signals in free space (whether in the earth’s lower atmosphere or essentially in a vacuum) may be assumed to be the velocity of light in a vacuum. (i) What is the furthest apart two geostationary satellites may be so that they can still communicate with each other without the path between the two satellites being interrupted by the surface of the earth? Give your answer in degrees longitude between the subsatellite points. (ii) If the longest, one-way delay permitted by the ITU between two earth stations communicating via a space system is 400 ms, what is the furthest apart two geostationary satellites may be before the transmission delay of the signal from one earth station to the other, when connected through the ISL system of the two satellites, equals 400 ms? The slant path distance between each earth station and the geostationary satellite it is communicating with may be assumed to be 40,000 km. (iii) If the satellites in part (ii) employ onboard processing, which adds an additional delay of 35 ms in each satellite, what is the maximum distance between the ISLlinked geostationary satellites now? (iv) If both of the two earth stations used in parts (ii) and (iii) must additionally now send the signals over a 2500-km optical fiber line to the end user on the ground, with an associated transmission delay in the fiber at each end of the link, what is the maximum distance between the ISL-linked geostationary satellites now? You may assume a refractive index of 1.5 for the optical fiber and zero processing delay in the earth station equipment and end user equipment.

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CHAPTER

3

SATELLITES Maintaining a microwave communication system in orbit in space is not a simple problem, so communications satellites are very complex, extremely expensive to purchase, and also expensive to launch. A typical large geostationary satellite, for example, is estimated to cost around $125 M, on station (see Chapter 2). The cost of the satellite and launch are increased by the need to dedicate an earth station to the monitoring and control of the satellite, at a cost of several million dollars per year. The revenue to pay these costs is obtained by selling the communication capacity of the satellite to users, either by way of leasing circuits or transponders, or by charging for circuit use, as in the international telephone and data transmission service. Communications satellites are usually designed to have a typical operating lifetime of 10 to 15 years. The operator of the system hopes to recover the initial and operating costs well within the expected lifetime of the satellite, and the designer must provide a satellite that can survive the hostile environment of outer space for that long. In order to support the communications system, the satellite must provide a stable platform on which to mount the antennas, be capable of station keeping, provide the required electrical power for the communication system, and also provide a controlled temperature environment for the communications electronics. In this chapter we discuss the subsystems needed on a satellite to support its primary mission of communications. We also discuss the communications subsystem itself in some detail, and other problems such as reliability. The emphasis throughout this chapter is on satellites in geostationary orbit. Communications satellites for low earth orbit are in most cases quite similar to small GEO satellites and have similar requirements. The discussion of satellites in this chapter is necessarily brief. For more details of the many subsystems used on satellites and their construction and operation the reader should refer to reference 1. Much information about individual satellites can be found on the web sites of satellite manufacturers and operators. See Table 1.1–1.4 in Chapter 1 for an extensive listing, which includes Web addresses.

3.1

SATELLITE SUBSYSTEMS The major subsystems required on the satellite are given below. Figure 3.1 shows an exploded view of a typical geostationary (GEO) satellite with several of the subsystems indicated.

Attitude and Orbit Control System (AOCS) This subsystem consists of rocket motors that are used to move the satellite back to the correct orbit when external forces cause it to drift off station and gas jets or inertial devices that control the attitude of the satellite. 57

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Diameter: 238 cm (93 in.) Overall Height: 701 cm (275 in.) Weight: 785 kg in orbit (1732 lb)

Telemetry & command bicones

Receiver reflector Nutation damper Receive feed horns Transmit odd channel reflector

Transmit even channel reflector

Odd channel feed horns transmit

Even channel feed horns transmit

Global transmit horn Global receive horn

Spot beam output multiplexer Telemetry horns

Global output multiplexer

Input filters

Beacon transmitter

TWTA EPCs

Bearing and power transfer assembly (BAPTA) Inboard sunshield

Looking forward

Earth sensors Conical sunshield

Despun shelf Communication receivers Position and orientation propellant tanks (4)

Radial jet Sun sensor

Encoder/decoder Command receivers

Battery controller Axial jet

Spinup jet

Radial jet Axial jet Earth sensor Booster adapter Spun electronics stack

FIGURE 3.1 Exploded view of a spinner satellite based on the Boeing (Hughes) HS 376 design. INTELSAT IVA (courtesy of Intelsat)

58

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Telemetry, Tracking, Command, and Monitoring (TTC&M) These systems are partly on the satellite and partly at the controlling earth station1. The telemetry system sends data derived from many sensors on the satellite, which monitor the satellite’s health, via a telemetry link to the controlling earth station. The tracking system is located at this earth station and provides information on the range and the elevation and azimuth angles of the satellite. Repeated measurement of these three parameters permits computation of orbital elements, from which changes in the orbit of the satellite can be detected. Based on telemetry data received from the satellite and orbital data obtained from the tracking system, the control system is used to correct the position and attitude of the satellite. It is also used to control the antenna pointing and communication system configuration to suit current traffic requirements, and to operate switches on the satellite.

Power System All communications satellites derive their electrical power from solar cells. The power is used by the communications system, mainly in its transmitters, and also by all other electrical systems on the satellite. The latter use is termed housekeeping, since these subsystems serve to support the communications system.

Communications Subsystems The communications subsystem is the major component of a communications satellite, and the remainder of the satellite is there solely to support it. Frequently, the communications equipment is only a small part of the weight and volume of the whole satellite. It is usually composed of one or more antennas, which receive and transmit over wide bandwidths at microwave frequencies, and a set of receivers and transmitters that amplify and retransmit the incoming signals. The receiver–transmitter units are known as transponders. There are two types of transponder in use on satellites: the linear or bent pipe transponder that amplifies the received signal and retransmits it at a different, usually lower, frequency, and the baseband processing transponder which is used only with digital signals, that converts the received signal to baseband, processes it, and then retransmits a digital signal.

Satellite Antennas Although these form part of the complete communication system, they can be considered separately from the transponders. On large GEO satellites the antenna systems are very complex and produce beams with shapes carefully tailored to match the areas on the earth’s surface served by the satellite. Most satellite antennas are designed to operate in a single frequency band, for example, C band or Ku band. A satellite which uses multiple frequency bands usually has four or more antennas. The subsystems listed above are discussed in more detail in this chapter. There are other subsystems that are not discussed here, but which are essential to the operation of the satellite—the thermal control system that regulates the temperature inside a satellite, for example. The reader who is interested in spacecraft design should refer to the literature of that field, particularly the IEEE Transactions on Aerospace and Electronic Systems2 and the American Institute of Aeronautics and Astronautics Transactions and annual Conference Proceedings 3,4. Only a brief review of the subsystems that support the communication mission is included here.

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3.2 ATTITUDE AND ORBIT CONTROL SYSTEM (AOCS) The attitude and orbit of a satellite must be controlled so that the satellite’s antennas point toward the earth and so that the user knows where in the sky to look for the satellite. This is particularly important for GEO satellites since the earth station antennas that are used with GEO satellites are normally fixed and movement of the satellite away from its appointed position in the sky will cause a loss of signal. There are several forces acting on an orbiting satellite that tend to change its attitude and orbit, as discussed in Chapter 2. The most important are the gravitational fields of the sun and the moon, irregularities in the earth’s gravitational field, solar pressure from the sun, and variations in the earth’s magnetic field. Solar pressure acting on a satellite’s solar sails and antennas, and the earth’s magnetic field generating eddy currents in the satellite’s metallic structure as it travels through the magnetic field, tend to cause rotation of the satellite body. Careful design of the structure can minimize these effects, but the orbital period of the satellite makes many of the effects cyclic, which can cause nutation (a wobble) of the satellite. The attitude control system must damp out nutation and counter any rotational torque or movement. The presence of gravitational fields from the sun and the moon cause the orbit of a GEO satellite to change with time. At GEO orbit altitude, the moon’s gravitational force is about twice as strong as the sun’s. The moon’s orbit is inclined to the equatorial plane by approximately 5°, which creates a force on the satellite with a component that is normal to the satellite’s orbit. The plane of the earth’s rotation around the sun is inclined by 23° to the earth’s equatorial plane. As discussed in Chapter 2, there is a net gravitational pull on the satellite that tends to change the inclination of the satellite’s orbit, pulling it away from the earth’s equatorial plane at an initial rate of approximately 0.86° per year. The orbital control system of the satellite must be able to move the satellite back into the equatorial plane before the orbital inclination becomes excessive. LEO satellites are less affected by gravitational fields of the sun and moon. Since they are much closer to the earth than GEO satellites, the earth’s gravity is much stronger, and the pull from the sun and moon are proportionately weaker. The earth is not quite a perfect sphere. At the equator, there are bulges of about 65 m at longitudes 162° E and 348° E, with the result that a satellite is accelerated toward one of two stable points in the GEO orbit at longitude 75° E and 252° E, as shown in Figure 3.2. To maintain accurate station keeping, the satellite must be periodically accelerated in the opposite direction to the forces acting on it. This is done as a sequence of station-keeping maneuvers, using small rocket motors (sometimes called gas jets or thrusters) that can be controlled from the earth via the TTC&M system.

Attitude Control System There are two ways to make a satellite stable in orbit, when it is weightless. The body of the satellite can be rotated, typically at a rate between 30 and 100 rpm, to create a gyroscopic force that provides stability of the spin axis and keeps it pointing in the same direction. Such satellites are known as spinners. The popular Hughes 376 (now Boeing 376) satellite is an example of a spinner design. Alternatively, the satellite can be stabilized by one or more momentum wheels. This is called a three-axis stabilized satellite, of which the Hughes (Boeing) 701 series is an example. The momentum wheel is usually a solid metal disk driven by an electric motor. Either there must be one momentum wheel for each of the three axes of the satellite, or a single momentum wheel can be mounted on

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Drift

61

Drift Satellite stable point 75° E E

W

EEN

Satellite semistable point 165° E

N

DIA

ERI

M ICH

0°

GR

Satellite semistable point 15° W

W

Equator 65-m bulge

Synchronous orbit

Satellite stablepoint 105° W Drift Drift

FIGURE 3.2

Forces on a synchronous satellite.

gimbals and rotated to provide a rotational force about any of the three axes5,6. Increasing the speed of the momentum wheel causes the satellite to precess in the opposite direction, according the principle of conservation of angular momentum. Figure 3.3 shows examples of both the spinner and three-axis design of satellite. The spinner design of satellite is typified by many satellites built by the Hughes Aircraft Corporation for domestic satellite communication systems. As shown in Figures 3.1 and 3.3a, the satellite consists of a cylindrical drum covered in solar cells that contains the power systems and the rocket motors. The communications system is mounted at the top of the drum and is driven by an electric motor in the opposite direction to the rotation of the satellite body to keep the antennas pointing toward the earth. Such satellites are called despun.

SIDEBAR In the early days of satellite communication despun antennas were not used, so antennas with a circular symmetric pattern were employed. These antennas have low gain and are now used only for basic TTC&M systems that must operate regardless of the satellite’s orientation. By despinning the antennas and transponders,

no RF rotating joints are needed in the communication system. Electrical power and some control signals must be brought through the despin bearing assembly with slip rings. The design of the bearing and slip rings to guarantee friction-free operation for 10 to 15 years in the total vacuum of outer space is a challenge.

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(a )

(b)

FIGURE 3.3 (a) A spinner satellite, INTELSAT IV A (b) A three-axis stabilized satellite, INTELSAT V (courtesy of Intelsat).

The satellite is spun up by operating small radial gas jets mounted on the periphery of the drum, at an appropriate point in the launch phase. The despin system is then brought into operation so that the main TTC&M antennas point toward the earth. The main TTC&M system operates at 64 GHz on the Intelsat satellite, with a 2-GHz backup system for use during the launch phase. A variety of liquid propulsion mixes have been used for the gas jets, the most common being a variant of hydrazine (N2H4), which is easily liquefied under

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pressure, but readily decomposes when passed over a catalyst7. Increased power can be obtained from the hydrazine gas jets by electrically heating the catalyst and the gas. Satellites that use liquid fuel thrusters have standardized on bipropellant fuels, that is fuels that mix together to form the thruster fuel. The most common bipropellents used for thruster operations are mono-methyl hydrazine and nitrogen tetroxide, although standard hydrazine is still used in place of mono-methyl hydrazine by some satellite manufacturers. The bipropellants are hypogolic: that is they ignite spontaneously on contact, and so do not need either a catalyst or a heater. By adjusting the flow of the bipropellants, pulses of thrust can be generated at the correct time and in the correct direction. There are two types of rocket motors used on satellites. The traditional bipropellant thruster described above, and arc jets or ion thrusters. The fuel that is stored on a GEO satellite is used for two purposes: to fire the apogee kick motor that injects the satellite into its final orbit, and to maintain the satellite in that orbit over its lifetime. If the launch is highly accurate, a minimum amount of fuel is used to attain the final orbit. If the launch is less accurate, more fuel must be used up in maneuvering the satellite into position, and that reduces the amount left for station keeping. A new development in thrusters uses a high voltage source to accelerate ions to a very high velocity, thus producing thrust. The ion engine thrust is not large, but because the engine can be driven by power from the solar cells it saves on expendable fuel. Ion engines can also be used to slowly raise a GEO satellite from a transfer orbit to GEO orbit as described in Chapter 2, although the process takes months rather than hours as with a conventional rocket engine. Arc jets or ion thrusters are mainly used for north–south station keeping, which is where the greatest use of fuel is required for station-keeping maneuvers, and became operational on the Hughes (Boeing) 600 series of satellite buses. Arc jets or ion thrusters lack the total thrust required to move satellites quickly (e.g., for major longitudinal changes in position) but a small, continuous thrust is adequate to maintain N–S and E–W position keeping. In a three-axis stabilized satellite, one pair of gas jets is needed for each axis to provide for rotation in both directions of pitch, roll, and yaw. An additional set of controls, allowing only one jet on a given axis to be operated, provides for velocity increments in the X, Y, and Z directions. When motion is required along a given axis, the appropriate gas jet is operated for a specified period of time to achieve the desired velocity. The opposing gas jet must be operated for the same length of time to stop the motion when the satellite reaches its new position. Fuel is saved if the velocity of the satellite is kept small, but progress toward the destination is slow. Let us define a set of reference Cartesian axes (XR, YR, ZR) with the satellite at the origin, as shown in Figure 3.4. The ZR axis is directed toward the center of the earth and is in the plane of the satellite orbit. It is aligned along the local vertical at the satellite’s subsatellite point. The XR axis is tangent to the orbital plane and lies in the orbital plane. The YR axis is perpendicular to the orbital plane. For a satellite serving the Northern Hemisphere, the directions of the XR and YR axes are nominally east and south. Rotation about the XR, YR, and ZR axes is defined as roll about the XR axis, pitch about the YR axis, and yaw about the ZR axis, in exactly the same way as for an aircraft or ship traveling in the X direction. The satellite must be stabilized with respect to the reference axes to maintain accurate pointing of its antenna beams. The axes XR, YR, and ZR are defined with respect to the location of the satellite; a second set of Cartesian axes, X, Y, Z, as shown in Figure 3.4, define the orientation of the satellite. Changes in a satellite’s attitude cause the angles , , and  in Figure 3.4 to vary as the X, Y, and Z axes move relative to the fixed reference axes XR, YR, and ZR. The Z axis is usually directed toward a reference point on the earth, called the Z-axis intercept. The location of the Z-axis intercept defines the pointing of the satellite antennas; the Z-axis intercept point may be

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Earth

O Equator

Z

S

E

N ZR

XR Roll X

Yaw

W

Pitch

Y

S

Spacecraft

YR

(a )

Z ZR

φ

Orbital path

θ

θ ψ

Y

XR

X

YR (b )

FIGURE 3.4 (a) Forces on a satellite. (b) Relationship between axes of a satellite.

moved to repoint all the antenna beams by changing the attitude of the satellite with the attitude control system. In a spinner-type satellite, the axis of rotation is usually the Y axis, which is maintained close to the YR axis, perpendicular to the orbital plane. Pitch correction is required only on the despun antenna system and can be obtained by varying the speed of the despin motor. Yaw and roll are controlled by pulsing radially mounted jets at the appropriate instant as the body of the satellite rotates. Attitude control of a three-axis stabilized satellite requires an increase or a decrease in the speed of the inertia wheel. If a constant torque exists about one axis of the satellite, a continual increase or decrease in momentum wheel speed is necessary to maintain the correct attitude. When the upper or lower speed limit of the wheel is reached, it must be unloaded by operating a pair of gas jets and simultaneously reducing or increasing the wheel speed. Closed-loop control of attitude is employed on the satellite to maintain

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North Earth

Fields of view Upper sensor Lower sensor

Voltage Upper sensor output

South

Lower sensor output Time

Error to N

No error

Error to S

FIGURE 3.5 Principle of N–S control of a spinner satellite using infrared Earth sensors.

the correct attitude. When large, narrow beam antennas are used, the whole satellite may have to be stabilized within 0.1° on each axis. The references for the attitude control system may be the outer edge of the earth’s disk, as observed with infrared sensors, the sun, or one or more stars. Figure 3.5 illustrates how an infrared sensor on the spinning body of a satellite can be used to control pointing toward the earth. Figure 3.6 shows a typical control system loop using the technique illustrated in Figure 3.5. The control system will be more complex for a three-axis stabilized satellite and may employ an onboard computer to process the sensor data and command the gas jets and momentum wheels. Despin control system (E–W) Upper sensor Lower sensor

Earth pulse input

E–W control unit

Antenna drive servo

Antenna position sensor

Attitude control system (N–S) Earth pulse comparator

Attitude control system

Gas jet control unit

N–S jet controls

6 GHz horn

4 GHz horn Command receiver

Control system

Telemetry transmitter

Telemetry formatter

Command and telemetry system FIGURE 3.6

Typical onboard control system for a spinner satellite.

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Correct position for spacecraft in equatorial plane

N Earth

N W Equatorial plane E

S

Inclined orbital plane

Spacecraft position in inclined orbital plane

FIGURE 3.7 Satellite in inclined orbit.

Orbit Control System As discussed in Chapter 2, a geostationary satellite is subjected to several forces that tend to accelerate it away from its required orbit. The most important, for the geostationary satellite, are the gravitation forces of the moon and the sun, which cause inclination of the orbital plane, and the nonspherical shape of the earth around the equator, which causes drift of the subsatellite point. There are many other smaller forces that act on the satellite causing the orbit to change. Accurate prediction of the satellite position a week or 2 weeks ahead requires a computer program with up to 20 force parameters; we shall restrict our discussion here to the two major effects. Figure 3.7 shows a diagram of an inclined orbital plane close to the geostationary orbit. For the orbit to be truly geostationary, it must lie in the equatorial plane, be circular, and have the correct altitude. The various forces acting on the satellite will steadily pull it out of the correct orbit; it is the function of the orbit control system to return it to the correct orbit. This cannot be done with momentum wheels since linear accelerations are required. Gas jets that can impart velocity changes along the three references axes of the satellite are required. If the orbit is not circular, a velocity increase or decrease will have to be made along the orbit, in the X-axis direction in Figure 3.4. On a spinning satellite, this is achieved by pulsing the radial jets when they point along the X axis. On a three-axis stabilized satellite, there will usually be two pairs of X-axis jets acting in opposite directions, one pair of which will be operated for a predetermined length of time to provide the required velocity change. The orbit of a geostationary satellite remains approximately circular for long periods of time and does not need frequent velocity corrections to maintain circularity. Altitude corrections are made by operating the Z-axis gas jets. The inclination of the orbit of a satellite that starts out in a geostationary orbit increases at an average rate of about 0.85° per year, with an initial rate of change of

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SIDEBAR The precessional forces that cause inclination changes can be used to increase satellite station-keeping lifetime by deliberately launching the satellite so that the inclination is not zero, but the precessional forces will act to reduce the inclination to close to zero. During this period, the E–W station-keeping tolerance is closely maintained. Once the inclination is close to zero, normal station keeping is started to maintain a tight orbit control in both axes. In this way, approximately a year of additional maneuvering lifetime may be obtained for each degree the inclination is nonzero

at the start of operations. Some communications systems do not need very tight station-keeping tolerance, either because the earth segment can track the satellite accurately or because no tracking is required (as would be the case for omnidirectional antennas). GEO satellites in such systems may relax their inclination (N–S) tolerance but may never relax the E–W station keeping tolerance as this would lead to unacceptable interference into other systems. GEO satellites in such relaxed orbits are sometimes called inclined-orbit satellites.

inclination for a satellite in an equatorial orbit between 0.75° to 0.94° per year (see Chapter 2). Most GEO satellites are specified to remain within a box of 0.05° and so, in practice, corrections, called a north–south station-keeping maneuver are made every 2 to 4 weeks to keep the error small. It has become normal to split the E–W and N–S maneuvers so that at intervals of 2 weeks the E–W corrections are made first and then after 2 more weeks, the N–S corrections are made. If arc jets or ion thrusters are used for N–S station-keeping maneuvers, these tend to operate almost continuously since their thrust levels are low when compared with traditional liquid fueled engines. Correcting the inclination of a satellite orbit requires more fuel to be expended than for any other orbital correction. This places a weight penalty on those satellites that must maintain very accurate station keeping, and reduces the communications payload they can carry. As much as half the total satellite weight at launch may be station keeping fuel when the satellite’s expected lifetime on orbit is 15 years. East–west station keeping is effected by use of the X-axis jets of the satellite. For a satellite located away from the stable points at 75° E and 252° E, a slow drift toward these points will occur. Typically, the X-axis jets are pulsed every 2 or 3 weeks to counter the drift and add a small velocity increment in the opposite direction. The satellite then drifts through its nominal position, stops at a point a fraction of a degree beyond it, and then drifts back again. East–west station keeping requires only a modest amount of fuel and is necessary on all geostationary communications satellites to maintain the spacing between adjacent satellites. With orbital locations separated by 2° or 3°, east–west drifts in excess of a fraction of a degree cannot be tolerated, and most GEO satellites are held within 0.05° of their allotted longitude. Some communications satellites such as the Russian Molniya series are not in geostationary orbit. (Molniya means lightning in Russian.) Early Molniya satellites were launched into a highly elliptical 12-h orbit with a large (65°) inclination angle to provide communication to northerly latitudes like Siberia. The Russian satellite gave its name to any satellite in a highly elliptical inclined orbit. Low earth orbit (LEO) and medium earth orbit (MEO) satellites also need AOC systems to maintain the correct orbit and attitude for continuous communication. Because of the much stronger gravitational force of the earth in LEO orbit, attitude stabilization is often accomplished with a rigid gravity gradient boom. This is a long pole that points toward the center of the earth, providing damping of oscillations about the satellite’s z axis by virtue of the difference in gravitational field at the top of the pole and at the bottom.

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3.3 TELEMETRY, TRACKING, COMMAND, AND MONITORING The TTC&M system is essential to the successful operation of a communications satellite. It is part of the satellite management task, which also involves an earth station, usually dedicated to that task, and a group of personnel. The main functions of satellite management are to control the orbit and attitude of the satellite, monitor the status of all sensors and subsystems on the satellite, and switch on or off sections of the communication system. The TTC&M earth station may be owned and operated by the satellite owner, or it may be owned by a third party and provide TTC&M services under contract. On large geostationary satellites, some repointing of individual antennas may be possible, under the command of the TTC&M system. Tracking is performed primarily by the earth station. Figure 3.8 illustrates the functions of a controlling earth station.

Telemetry and Monitoring System The monitoring system collects data from many sensors within the satellite and sends these data to the controlling earth station. There may be several hundred sensors located on the satellite to monitor pressure in the fuel tanks, voltage and current in the power conditioning unit, current drawn by each subsystem, and critical voltages and currents in the communications electronics. The temperature of many of the subsystems is important and must be kept within predetermined limits, so many temperature sensors are fitted. The sensor data, the status of each subsystem, and the positions of switches in the communication system are reported back to the earth by the telemetry system. The sighting devices used to maintain attitude are also monitored via the telemetry link: this is essential in case one should fail and cause the satellite to point in the wrong direction. The faulty unit must then be disconnected and a spare brought in, via the command system, or some other means of controlling attitude devised. Telemetry data are usually digitized and transmitted as phase shift keying (PSK) of a low-power telemetry carrier using time division techniques. A low data rate is normally used to allow the receiver at the earth station to have a narrow bandwidth and thus maintain a high carrier to noise ratio. The entire TDM frame may contain thousands of bits of data and take several seconds to transmit. At the controlling earth station a computer can be used to monitor, store, and decode the telemetry data so that the status of any system or sensor on the satellite can be determined immediately by the controller on the earth. Alarms can also be sounded if any vital parameter goes outside allowable limits.

Tracking A number of techniques can be used to determine the current orbit of a satellite. Velocity and acceleration sensors on the satellite can be used to establish the change in orbit from the last known position, by integration of the data. The earth station controlling the satellite can observe the Doppler shift of the telemetry carrier or beacon transmitter carrier to determine the rate at which range is changing. Together with accurate angular measurements from the earth station antenna, range is used to determine the orbital elements. Active determination of range can be achieved by transmitting a pulse, or sequence of pulses, to the satellite and observing the time delay before the pulse is received again. The propagation delay in the satellite transponder must be accurately known, and more than one earth station may make range measurements. If a sufficient number of earth stations with an adequate separation are observing the satellite, its position can be established

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Satellite

Satellite TTC&M antenna

Receive antenna

Transmit antenna

Telecommand transmitter

Telemetry receiver Tracking system

Data processor

Computer for attitude and orbital control Ephemeris data

Controller

FIGURE 3.8 Typical tracking, telemetry, command and monitoring system.

by triangulation from the earth station by simultaneous range measurements. With precision equipment at the earth stations, the position of the satellite can be determined within 10 m. Ranging tones are also used for range measurement. A carrier generated on board the satellite is modulated with a series of sine waves at increasing frequency, usually harmonically related. The phase of the sine wave modulation components is compared at an earth station, and the number of wavelengths of each frequency is calculated. Ambiguities in the numbers are resolved by reference to lower frequencies, and prior knowledge of the approximate range of the satellite. If sufficiently high frequencies are used, perhaps even the carrier frequency, range can be measured to millimeter accuracy. The technique is similar to that used in the terrestrial telurometer and in aircraft radar altimeters.

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Command A secure and effective command structure is vital to the successful launch and operation of any communications satellite. The command system is used to make changes in attitude and corrections to the orbit and to control the communication system. During launch, it is used to control the firing of the apogee kick motor and to spin up a spinner or extend the solar sails and antennas of a three-axis stabilized satellite. The command structure must possess safeguards against unauthorized attempts to make changes to the satellite’s operation, and also against inadvertent operation of a control due to error in a received command. Encryption of commands and responses is used to provide security in the command system. A typical system of the type shown in Figure 3.8 will originate commands at the control terminal of the computer. The control code is converted into a command word, which is sent in a TDM frame to the satellite. After checking for validity in the satellite, the word is sent back to the control station via the telemetry link where it is checked again in the computer. If it is found to have been received correctly, an execute instruction will be sent to the satellite so that the command is executed. The entire process may take 5 or 10 s, but minimizes the risk of erroneous commands causing a satellite malfunction. The command and telemetry links are usually separate from the communication system, although they may operate in the same frequency band (6 and 4 GHz). Two levels of command system are used in the Intelsat satellite: the main system operates in the 6GHz band, in a gap between the communication channel frequencies; the main telemetry system uses a similar gap in the 4-GHz band. The TTC&M antennas for the 64 GHz system can be seen in Figure 3.1 on the satellite. These are earth-coverage horns, so the main system can be used only after correct attitude of the satellite is achieved. During the launch phase and injection into geostationary orbit, the main TTC&M system may be inoperable because the satellite does not have the correct attitude or has not extended its solar sails. A backup system is used at this time, which controls only the most important sections of the satellite. A great deal of redundancy is built into this system, since its failure will jeopardize the entire mission. Near omnidirectional antennas are used at either UHF or S band (2–4 GHz), and sufficient margin is allowed in the signalto-noise ratio (SN) at the satellite receiver to guarantee control under the most adverse conditions. The backup system provides control of the apogee kick motor, the attitude control system and orbit control thrusters, the solar sail deployment mechanism (if fitted), and the power conditioning unit. With these controls, the satellite can be injected into

SIDEBAR Controlling a satellite in orbit is a complex process which requires considerable care. In one case, an incorrect sequence of command instructions caused loss of control of the Olympus satellite, a European large GEO satellite used for experiments in the 3020 GHz band. The E–W thrusters fired for a lengthy period, causing the satellite to drift toward the east at 5° per day, and the satellite also began rotating with a period of 90 s. All communication with the satellite was lost, and the batteries discharged fully because the solar sails no longer pointed at the sun.

The satellite drifted round the earth over a period of 21⁄2 months, and was eventually recovered by a team of experts using large antennas in Australia and the United States to send telemetry commands to the satellite. The solar cells provided short bursts of power as they rotated past the sun’s direction, allowing commands to be sent for a few seconds every 90 s. The rotation of the satellite was eventually stopped and Olympus returned to its correct location, but with shortened life expectancy due to the loss of station-keeping fuel.

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geostationary orbit, turned to face the earth, and switched to full electrical power so that handover to the main TTC&M system is possible. In the event of failure of the main TTC&M system, the backup system can be used to keep the satellite on station. It is also used to eject the satellite from geostationary orbit and to switch off all transmitters when the satellite eventually reaches the end of its useful life.

3.4

POWER SYSTEMS All communications satellites obtain their electrical power from solar cells, which convert incident sunlight into electrical energy. Some deep space planetary research satellites have used thermonuclear generators to supply electrical power, but because of the danger to people on the earth if the launch should fail and the nuclear fuel be spread over an inhabited area, communications satellites have not used nuclear generators. The sun is a powerful source of energy. In the total vacuum of outer space, at geostationary altitude, the radiation falling on a satellite has an intensity of 1.39 kW/m2. Solar cells do not convert all this incident energy into electrical power; their efficiency is typically 20 to 25% at beginning of life (BOL) but falls with time because of aging of the cells and etching of the surface by micrometeor impacts. Since sufficient power must be available at the end of life (EOL) of the satellite to supply all the systems on board, about 15% extra area of solar cells is usually provided as an allowance for aging. A spin-stabilized satellite usually has a cylindrical body covered in solar cells. Because the solar cells are on a cylindrical surface, half of the cells are not illuminated at all, and at the edges of the illuminated half, the low angle of incidence results in little electrical power being generated. The output from the solar cells is slightly higher than would be obtained with normal incidence on a flat panel equal in area to the projected area of the cylinder, that is, its width times its height. The cells that are not illuminated by sunlight face cold space, which causes them to cool down. The solar cells on a spinner satellite have a lower temperate than those on solar sails, which increases their efficiency somewhat. Early satellites were of small dimensions and had relatively small areas of solar cells. More recently, large communications satellites for direct broadcast operation generate up to 6 kW from solar power. A three-axis stabilized satellite can make better use of its solar cell area, since the cells can be arranged on flat panels that can be rotated to maintain normal incidence of the sunlight. Only one-third of the total area of solar cells is needed relative to a spinner, with some saving in weight. A primary advantage, however, is that by unfurling a folded solar array when the satellite reaches geostationary orbit, power in excess of 10 kW can be generated with large arrays. To obtain 10 kW from a spinner requires a very large body on which to place the solar cells, which may then exceed the maximum payload dimensions of the launch vehicle. Solar sails must be rotated by an electric motor once per 24 h to keep the cells in full sunlight. This causes the cells to heat up, typically to 50° to 80°C, which causes a drop in output voltage. In the spinner design, the cells cool down when in shadow and run at 20° to 30°C, with somewhat higher efficiency. The bombardment of the sails by protons and electrons is also more severe, and a thicker layer of glass may be needed to slow down deterioration of the cells, with a consequent weight penalty. A rotary joint must be used with each solar sail to transfer current from the rotating sail to the body of the satellite. The satellite must carry batteries to power the subsystems during launch and during eclipses. Eclipses occur twice per year, around the spring and fall equinoxes, when

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the earth’s shadow passes across the satellite, as illustrated in Figures 2.21 and 2.22. The longest duration of eclipse is 70 min, occurring around March 21 and September 21 each year. To avoid the need for large, heavy batteries, part or all of the communications system load may be shut down during eclipse, but this technique is rarely used when telephony or data traffic is carried. TV broadcast satellites may not carry sufficient battery capacity to supply their high-power transmitters during eclipse, and may shut down. By locating the satellite 20° W of the longitude of the service area, the eclipse will occur after 1 A.M. local time for the service area, when shutdown is more acceptable. Batteries are usually of the nickel–hydrogen type which do not gas when charging and have good reliability and long life, and can be safely discharged to 70% of their capacity. A powerconditioning unit controls the charging current and dumps excess current from the solar cells into heaters or load resistors on the cold side of the satellite. Sensors on the batteries, power regulator, and solar cells monitor temperature, voltage, and current and supply these data to both the onboard control system and the controlling earth station via the telemetry downlink. Typical battery voltages are 20 to 50 V with capacities of 20 to 100 ampere-hours.

3.5

COMMUNICATIONS SUBSYSTEMS Description of the Communications System A communications satellite exists to provide a platform in geostationary orbit for the relaying of voice, video, and data communications. All other subsystems on the satellite exist solely to support the communications system, although this may represent only a small part of the volume, weight, and cost of the satellite in orbit. Since it is the communications system that earns the revenue for the system operator, communications satellites are designed to provide the largest traffic capacity possible. The growth in capacity is well illustrated in Figure 3.9 for the Intelsat system. Successive satellites have become larger, heavier, and more costly, but the rate at which traffic capacity has increased has been much greater, resulting in a lower cost per telephone circuit or transmitted bit with each succeeding generation of satellite. The satellite transponders have limited output power and the earth stations are at least 36,000 km away from a GEO satellite, so the received power level, even with large aperture earth station antennas, is very small and rarely exceeds 1010 W. For the system to perform satisfactorily, the signal power must exceed the power of the noise generated in the receiver by between 5 and 25 dB, depending on the bandwidth of the transmitted signal and the modulation scheme used. With low power transmitters, narrow receiver bandwidths have to be used to maintain the required signal-to-noise ratios. Early communications satellites were fitted with transponders of 250 or 500 MHz bandwidth, but had low gain antennas and transmitters of 1 or 2 W output power. The earth station receiver could not achieve an adequate signal-to-noise ratio when the full bandwidth was used with the result that the system was power limited. Later generations of communications satellites have transponders with greatly increased output power—up to 200 W for DBS-TV satellites—and have steadily improved in bandwidth utilization efficiency, as seen in Figure 3.9. The total channel capacity of a satellite that uses a 500-MHz band at 64 GHz can be increased only if the bandwidth can be increased or reused. The trend in high-capacity satellites has been to reuse the available bands by employing several directional beams at the same frequency (spatial frequency reuse) and orthogonal polarizations at the same frequency (polarization

50 MHz

Total bandwidth

Hughes

$11,000

$4.6 M

$3.5 M

3 years

240

130 MHz

85 W

76 kg

TRW

$1,600

$6 M

$4.5 M

5 years

1500

360 MHz

125 W

152 kg

1.42 m dia × 1.98 m high

1968

INTELSAT III

Hughes

$810

$20 M

$14 M

7 years

5000

450 MHz

569 W

595 kg

2.38 m dia × 7.01 m high

1971

INTELSAT IV

Hughes

$494

$20 M

$18 M

7 years

11,000 plus 2 TV channels

720 MHz

708 W

786 kg

2.38 m dia × 7.01 m high

1975

INTELSAT IV-A

Illustration of the growth in size and weight of Intelsat satellites over 3 decades.

Hughes

Contractor

FIGURE 3.9

$4.6 M

$23,000

Cost per telephone circuit year

$3.6 M

Spacecraft cost

Launch cost

1.5 years

Design lifetime

240

46 W

Notional capacity two-way telephone circuits

34 kg

End of life primary power

1.42 m dia × 0.67 m high

0.71 m dia × 0.59 m high

Dimensions

On orbit weight

1967

INTELSAT II

1965

INTELSAT I

Year of first launch

Spacecraft

Ford Aerospace

$200

$23 M

$25 M

10 years

24,000 plus 2 TV channels

2250 MHz

1220 W

1020 kg

15.27 m across solar sails × 6.71 m high

1980

INTELSAT V

Hughes

?

?

$140 M (first five satellites)

10 years

33,000 plus 2 TV channels

3360 MHz

2100 W

1800 kg

3.6 m dia × 11.7 m high

1986 (planned)

INTELSAT VI

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frequency reuse). Large GEO satellites also use both the 64 GHz and 1411 GHz bands to obtain more bandwidth; for example, some GEO satellites have achieved an effective bandwidth of 2250 MHz within a 500-MHz band at 64 GHz and a 250-MHz band at 1411 GHz by a combination of spatial and polarization frequency reuse, and later generations of Intelsat satellites have achieved up to sevenfold reuse of their frequency bands. The designer of a satellite communication system is not free to select any frequency and bandwidth he or she chooses. International agreements restrict the frequencies that may be used for particular services, and the regulations are administered by the appropriate agency in each country—the Federal Communication Commission (FCC) in the United States, for example. Frequencies allocated to satellite services are listed in Tables 4.1 and 4.2 in Chapter 4. The bands currently used for the majority of services are 64 GHz and 1411 GHz, with 3020 GHz coming into service. The 64 GHz band was expanded from 500-MHz bandwidth in each direction to 1000-MHz by a World Administrative Radio Conference in 1979, but other services share the new part of the band and may cause interference to the satellite communication link. Similar bandwidth is available at 1411 GHz, and Ka-band satellites will be launched in the first decade of 2000 to exploit the wider bandwidths available in the 3020 GHz bands. A different frequency is required for the transmit path (normally the higher frequency) at the satellite because the high-power transmit signal would overload the receiver if they both operated at the same frequency. The 500-MHz bands originally allocated for 64 and 1411 GHz satellite communications have become very congested and are now completely filled for some segments of the geostationary orbit, such as that serving North America. Extension of the bands to 1000 MHz will eventually provide greater capacity as the new frequencies come into use. Many systems now use 1411 GHz for TV broadcast and distribution, and 3020 GHz systems are introducing Internet-like services from GEO. The standard spacing between GEO satellites was originally set at 3°, but under regulations covering North America and much of the rest of the world, the spacing has been reduced to 2°. The move to 2° spacing opened up extra slots for new satellites in the 64 and 1411 GHz bands. Satellite systems designed for Ku band (1411 GHz) and Ka band (3020 GHz), have narrower antenna beams, and better control of coverage patterns than satellites using C band (64 GHz). As the available orbital slots for GEO satellites have filled up with satellites using the 64 and 1411 GHz bands, attention has focused on the use of

SIDEBAR Ka-band satellite links must accept larger outage times due to rain than 1411 GHz links, and are therefore better suited to data transmission than voice. The transmission of data can be delayed when a rain fade affects the link, but a telephone conversation will be ended if the link fails for more than a few seconds. In the time frame 2001–2010 several Ka-band satellite systems will come into service. The major market appears to be data relay, particularly Internet access. Satellites support asymmetric data links such as Internet service to indi-

vidual users very well. The user wants high-speed, high volume data delivery from the Internet to send video clips and high resolution photographs (known generically as multimedia), but does not send equivalent volumes of data into the Internet. A lower capacity link from the user is adequate, and this suits Ka-band links with large hub antennas and small user terminals. The link will suffer occasional outages, perhaps for a total of 25 h in a year, when heavy rain is falling through the path between the satellite and the user’s small terminal.

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75

the 3020 GHz band. Originally, this band had 3 GHz bandwidth allocated to satellite services, but part of the band was reallocated to the Land Multipoint Distribution Service (LMDS). Approximately 2 GHz of bandwidth is still available for satellite systems at Ka band on an exclusive or shared basis, which is equal to the combined allocations of C and Ku bands. However, propagation in rain becomes a major factor at frequencies above 10 GHz. Attenuation (in dB) in rain increases at roughly the square of the frequency, so at 20 GHz rain attenuation is four times larger, in dB, than at 10 GHz. It is almost impossible to provide rain attenuation margins larger than 20 dB in any system, and many Ka-band systems will have margins of only 4 or 5 dB when operating to small earth station antennas.

Transponders Signals (known as carriers) transmitted by an earth station are received at the satellite by either a zone beam or a spot beam antenna. Zone beams can receive from transmitters anywhere within the coverage zone, whereas spot beams have limited coverage. The received signal is often taken to two low noise amplifiers and is recombined at their output to provide redundancy. If either amplifier fails, the other one can still carry all the traffic. Since all carriers from one antenna must pass through a low noise amplifier, a failure at that point is catastrophic. Redundancy is provided wherever failure of one component will cause the loss of a significant part of the satellite’s communication capacity. Figure 3.10 shows a simplified block diagram of a satellite communication subsystem for the 64 GHz band. The 500-MHz bandwidth is divided up into channels, often 36 MHz wide, which are each handled by a separate transponder. A transponder consists of a band-pass filter to select the particular channel’s band of frequencies, a downconverter to change the frequency from 6 GHz at the input to 4 GHz at the output, and an output amplifier. The communication system has many transponders, some of which may be spares; typically 12 to 44 active transponders are carried by a high-capacity satellite. The transponders are supplied with signals from one or more receive antennas and send their outputs to a switch matrix that directs each transponder band of frequencies to the appropriate antenna or antenna beam. In a large satellite there may be four or five beams to which any transponder can be connected. The switch setting can be controlled from the earth to allow reallocation of the transponders between the downlink beams as traffic patterns change. In the early satellites such as INTELSAT I and II, one or two 250-MHz bandwidth transponders were employed. This proved unsatisfactory because of the nonlinearity of the traveling wave tube transmitter used at the output of the transponder, and later GEO satellites have used up to 44 transponders each with 36, 54, or 72 MHz bandwidth. The reason for using narrower bandwidth transponders is to avoid excessive intermodulation problems when transmitting several carriers simultaneously with a nonlinear transmitter, as discussed in Chapter 6. Intermodulation distortion is likely to occur whenever a high power amplifier is driven close to saturation. Since we generally want to have more than one earth station transmitter sending signals via a satellite, one solution would be to provide one transponder for each earth station’s signal. In the case of the Intelsat global system, this could result in a requirement for as many as 100 transponders per satellite. As a compromise, 36 MHz has been widely used for transponder bandwidth, with 54 and 72 MHz adopted for some satellites. Many domestic satellites operating in the 64 GHz band carry 24 active transponders. The center frequencies of the transponders are spaced 40 MHz apart, to allow

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3720 1

3760 3

3880 5

3740 3701 2 T/M

3880 9

3820 6

3860 8

Transmit 3920 3960 11 13

3900 10

3940 12

4000 15

3980 14

4040 17

4020 16

4080 19

4060 18

4120 21

4100 20

4160 23

4199 T/M 4180 24

4140 22

Frequency MHz

3700 5945 1

5985 3

5965 2

6025 5

6005 4

6065 7

6045 6

6105 9

6085 8

Receive 6145 6185 11 13

6125 10

6165 12

4200 6225 15

6205 14

6265 17

6245 16

6305 19

6285 18

6345 21

6325 20

6385 23

6424 CMD 6405 24

6365 22

Frequency MHz

5925

6425

Frequency plan. Note: Number below channel center frequency refers to transponder identity. V

TWTA

Input filter

Input multiplexer

1 5 9 13 17 TWTA 21

Output multiplexer

Output filter

Input multiplexer

3 7 11 15 19 TWTA 23

Output multiplexer

Output filter

Receiver/driver (redundant)

V

TWTA

Input multiplexer Receiver/driver (redundant)

2 6 10 14 18 TWTA 22

D

Output multiplexer

Output filter

Output multiplexer

Output filter

D

H

TWTA

Input multiplexer

4 8 12 16 20 TWTA 24

V H

TWTA

Input filter

D

H

FIGURE 3.10 Transponder arrangement of RCA’s SATCOM satellites and frequency plan. The translation frequency is 2225 MHz. [Reproduced with permission from W. H. Braun and J. E. Keigler, “RCA Satellite Networks: High Technology and Low User Cost,” Proceedings of the IEEE, 72, 1483–1505 (November 1984). Copyright © 1984 IEEE.]

guard bands for the 36 MHz filter skirts. With a total of 500 MHz available, a single polarization satellite can accommodate 12 transponders across the band. When frequency reuse by orthogonal polarizations is adopted, 24 transponders can be accommodated in the same 500 MHz bandwidth. Traditional linear transponder-type satellites now have sixfold reuse (INTELSAT VI and IX) or even sevenfold reuse (INTELSAT VIII) at C band. The reuse is achieved through microwave switch interconnections between subbeams. Internet-like satellites need a plethora of beam interconnections—more than 50 in most cases. The only way to achieve this level of beam /path interconnections is via on board processing (OBP).

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14-GHz Dual polarized

11-GHz Dual polarized Ch. 1-2, 5-6, 7-12

Uplink

Upconverters

R×1 R×2

Spot beam antennas

77

R×3

Downlink TWTA’s and O/P MUX

I/P MUX

R×4

6-GHz Dual polarized Receivers 6-GHz

Uplink zone beam antenna

O/P MUX

TWTA’s

O/P MUX

I/P MUX Ch. 1-2, 3-4, 5-6 7, 8, 9

Receivers 6-GHz

TWTA’s

Switch matrix

I/P MUX Ch. 1-2, 3-4, 5-6, 7-8

TWTA’s

O/P MUX

TWTA’s

O/P MUX

Spot beam antennas

4-GHz Dual polarized

Downlink zone beam antenna

6-GHz Receivers 6-GHz Global beam horn

I/P MUX Ch. 9, 10, 11, 12, 7-8

4-GHz

Global beam horn

FIGURE 3.11 Simplified block diagram of an INTELSAT V communication system. Note that the switch matrix allows many possible interconnections between uplink beams and downlink transmitters. (Courtesy C. F. Hoeber, Ford Aerospace and Communications Corp.)

Figure 3.11 shows a simplified diagram of the communication system carried by INTELSAT V satellites. The later series of Intelsat satellites use a similar arrangement. The bulk of the traffic is carried by the 64 GHz section, with a total bandwidth of 2000 MHz available by frequency reuse. The switch matrix allows a very large number of variations in connecting the 6-GHz receivers to the 4-GHz transmitters, and also interconnects the 64 and 1411-GHz sections. This provides Intelsat with a great deal of flexibility in setting up links through the satellite. When more than one signal shares a transponder (using frequency division multiple access, FDMA) the power amplifier must be run below its maximum output power to maintain linearity and reduce intermodulation products. The degree to which the transmitter output power is reduced below its peak output is known as output backoff: in FDMA systems, 2 to 7 dB of output backoff is typically used, depending on the number of accesses to the transponder and the extent to which the characteristics of the HPA have been linearized. Backoff results in a lower downlink carrier-to-noise ratio at the earth station with FDMA when multiple accesses to each transponder are required. Time division multiple access (TDMA) can theoretically be used to increase the output power of transponders by limiting the transponder to a single access. However, most TDMA systems are hybrid FDMA–TDMA schemes known as multifrequency TDMA (MF-TDMA), in which several TDMA signals share the transponder bandwidth using FDMA. Linearity of the HPA remain an issue for MF-TDMA systems.

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LNA

6 GHz uplink antenna

6 GHz BPF

Downconverter

Local oscillator 2225 MHz

4 GHz BPF

4 GHz LPA

4 GHz HPA

4 GHz downlink antenna

FIGURE 3.12 Simplified single conversion transponder (bent pipe) for 64 GHz band.

Figure 3.12 shows a typical single conversion bent pipe transponder of the type used on many satellites for the 64 GHz band. The output power amplifier is usually a solid state power amplifier (SSPA) unless a very high output power (50 W) is required, when a traveling wave tube amplifier (TWTA) would be used9. The local oscillator is at 2225 MHz to provide the appropriate shift in frequency from the 6-GHz uplink frequency to the 4-GHz downlink frequency, and the band-pass filter after the mixer removes unwanted frequencies resulting from the down-conversion operation. The attenuator can be controlled via the uplink command system to set the gain of the transponder. Redundancy is provided for the high-power amplifiers (HPA) in each transponder by including a spare TWT or solid-state amplifier (SSPA) that can be switched into circuit if the primary power amplifier fails. The lifetime of HPAs is limited, and they represent the least reliable component in most transponders. Providing a spare HPA in each transponder greatly increases the probability that the satellite will reach the end of its working life with all its transponders still operational. Transponders can also be arranged so that there are spare transponders available in the event of a total failure. The arrangement is known as M for N redundancy. For example, it is common to have 16 for 10 redundancy or even 14 for 10. That is, 16 (or 14) output amplifiers are connected in a ring such that any of the 10 signals can pass through them. Thus, 6 (or 4) amplifiers are acting as back-up amplifiers while 10 are on line. Most HPAs have bandwidths much larger than the allocated frequency band and so it matters little which signals are passing through them. At Ku band, ring redundancy is still used, but it is much more like 2 for 1, that is, one spare for every active unit. Transponders for use in the 1411-GHz bands normally employ a double frequency conversion scheme as illustrated in Figure 3.13. It is easier to make filters, amplifiers, and equalizers at an intermediate frequency (IF) such as 1100-MHz than at 14 or 11 GHz, so the incoming 14-GHz carrier is translated to an IF of around 1 GHz. The amplification and filtering are performed at 1 GHz and a relatively high-level carrier is translated back to 11 GHz for amplification by the HPA. Stringent requirements are placed on the filters used in transponders, since they must provide good rejection of unwanted frequencies, such as intermodulation products, and also have very low amplitude and phase ripple in their pass bands. Frequently a filter will be followed by an equalizer that smoothes out amplitude and phase variations in the pass band. Phase variation across the pass band produces group delay distortion, which is particularly troublesome with wideband FM signals and high-speed phase shift keyed data transmissions. A considerable increase in the communications capacity of a satellite can be achieved by combining onboard processing with switched-beam technology. A switched-beam satellite generates a narrow transmit beam for each earth station with which it communicates, and then transmits sequentially to each one using time division multiplexing of the signals. The narrow beam has to cover only one earth station, allowing the satellite transmit antenna to have a very high gain compared to a zone-coverage antenna. A narrow scanning beam can also be used, or a combination of fixed and scanning beams. Unless the satellite has a

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LNA

14 GHz BPF

Downconverter

1 GHz BPF

79

1 GHz IF amplifier

13 GHz Local oscillator

14 GHz uplink antenna

10 GHz Local oscillator

1 GHz BPF

11 GHz HPA

11 GHz downlink antenna

Upconverter

FIGURE 3.13 Simplified double conversion transponder (bent pipe) for 1411 GHz band.

zone-coverage receiver antenna, data storage is required at the satellite since it communicates with only one earth station at a time. The high gain antennas used in switched-beam systems raise the EIRP (effective isotropically radiated power) of the satellite transmitter and thus increase the capacity of the downlink. Switched beam systems on GEO satellites work best at Ka band where the wavelength is short enough that the limited dimensions of antennas on the satellite still allow beams of less than 0.4° beamwidth to be generated. Multiple beam antennas with baseband processing transponders are used on GEO and LEO satellites providing service to mobile terminals and handheld telephones10. The low gain of the near omnidirectional antenna of a mobile earth station must be compensated by a high gain antenna on the satellite, necessitating the use of multiple beam antennas. It is possible to conserve uplink bandwidth by using different modulation techniques on the uplink and downlink and by providing a baseband processor on the satellite. A high level modulation such as 16-QAM with four bits per symbol can be used on the link between the satellite and a large earth station to improve bandwidth efficiency. This approach has been adopted in the Astrolink and Spaceway 3020 GHz satellites16,17. Onboard processing may also be used to advantage to switch between the uplink access technique (e.g., MF-TDMA) and the downlink access technique (e.g., TDM) so that small earth stations may access each other directly via the satellite. The processor can provide the data storage needed for a switched-beam system and also can perform error correction independently on the uplink and downlink. A typical arrangement of the communication system for a satellite employing onboard processing is shown in Figure 3.14. Multiple beam uplink antenna

Multiple beam downlink antenna

Receiver Demodulator Rx

Demod

FEC decoder

Baseband Processor

FEC encoder

Modulator

Transmitter

FEC

OBP

FEC

Mod

Tx

C On-board controller

FIGURE 3.14 Onboard processing transponder.

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SATELLITE ANTENNAS Basic Antenna Types and Relationships Four main types of antennas are used on satellites. These are 1. 2. 3. 4.

Wire antennas: monopoles and dipoles. Horn antennas. Reflector antennas. Array antennas.

Wire antennas are used primarily at VHF and UHF to provide communications for the TTC&M systems. They are positioned with great care on the body of the satellite in an attempt to provide omnidirectional coverage. Most satellites measure only a few wavelengths at VHF frequencies, which makes it difficult to get the required antenna patterns, and there tend to be some orientations of the satellite in which the sensitivity of the TTC&M system is reduced by nulls in the antenna pattern. An antenna pattern is a plot of the field strength in the far field of the antenna when the antenna is driven by a transmitter. It is usually measured in decibels (dB) below the maximum field strength. The gain of an antenna is a measure of the antenna’s capability to direct energy in one direction, rather than all around. Antenna gain is defined in Chapter 4, Section 4.2. At this point, it will be used with the simple definition given above. A useful principle in antenna theory is reciprocity. Reciprocity means that an antenna has the same gain and pattern at any given frequency whether it transmits or receives. An antenna pattern measured when receiving is identical to the pattern when transmitting. Figure 3.15 shows typical satellite antenna coverage zones. The pattern is frequently specified by its 3-dB beamwidth, the angle between the directions in which the radiated

17°

Global beam

Phased array antenna

Spot beams

Vertical polarization

Horizontal polarization Multiple spot beams and scanning beams

Orthogonally polarized beams

FIGURE 3.15 Typical satellite antenna patterns and coverage zones. The antenna for the global beam is usually a waveguide horn. Scanning beams and shaped beams require phased array antennas or reflector antennas with phased array feeds.

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81

70

Latitude (degrees)

60

50

−7−6 −5 −4 −3 −2

−1

40

10

0

10

20

30

Longitude (degrees) FIGURE 3.16 Typical coverage patterns for Intelsat satellites over the Atlantic Ocean.

(or received) field falls to half the power in the direction of maximum field strength. However, a satellite antenna is used to provide coverage of a certain area, or zone on the earth’s surface, and it is more useful to have contours of antenna gain as shown in Figure 3.16. When computing the signal power received by an earth station from the satellite, it is important to know where the station lies relative to the satellite transmit antenna contour pattern, so that the exact EIRP can be calculated. If the pattern is not known, it may be possible to estimate the antenna gain in a given direction if the antenna boresight or beam axis direction and its beamwidth are known. Horn antennas are used at microwave frequencies when relatively wide beams are required, as for global coverage. A horn is a flared section of waveguide that provides an aperture several wavelengths wide and a good match between the waveguide impedance and free space. Horns are also used as feeds for reflectors, either singly or in clusters. Horns and reflectors are examples of aperture antennas that launch a wave into free space from a waveguide. It is difficult to obtain gains much greater than 23 dB or beamwidths narrower than about 10° with horn antennas. For higher gains or narrow beamwidths a reflector antenna or array must be used. Reflector antennas are usually illuminated by one or more horns and provide a larger aperture than can be achieved with a horn alone. For maximum gain, it is necessary to generate a plane wave in the aperture of the reflector. This is achieved by choosing a reflector profile that has equal path lengths from the feed to the aperture, so that all the energy radiated by the feed and reflected by the reflector reaches the aperture with the same phase angle and creates a uniform phase front. One reflector shape that achieves this with a point source of radiation is the paraboloid, with a feed placed at its focus. The

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paraboloid is the basic shape for most reflector antennas, and is commonly used for earth station antennas. Satellite antennas often use modified paraboloidal reflector profiles to tailor the beam pattern to a particular coverage zone. Phased array antennas are also used on satellites to create multiple beams from a single aperture, and have been used by Iridium and Globalstar to generate up to 16 beams from a single aperture for their LEO mobile telephone systems11. Some basic relationships in aperture antennas can be used to determine the approximate size of a satellite antenna for a particular application, as well as the antenna gain. More accurate calculations are needed to determine the exact gain, efficiency, and pattern of a satellite antenna, and the interested reader should refer to one of the many excellent texts in this field for details.12–14 The following approximate relationships will be used here to guide the selection of antennas for a communications satellite. An aperture antenna has a gain G given by G  hA 4pAl2

(3.1)

where A is the area of the antenna aperture in meters,  is the operating wavelength in meters, and A is the aperture efficiency of the antenna. The aperture efficiency A is not easily determined, but is typically in the range 55 to 68% for reflector antennas with single feeds, lower for antennas with shaped beams. Horn antennas tend to have higher efficiencies than reflector antennas, typically in the range 65 to 80%. If the aperture is circular, as is often the case, Eq. (3.1) can be written as G  hA 1pDl2 2

(3.2)

where D is the diameter of the circular aperture in meters. The beamwidth of an antenna is related to the aperture dimension in the plane in which the pattern is measured. A useful rule of thumb is that the 3 dB beamwidth in a given plane for an antenna with dimension D in that plane is u3 dB  75lD degrees

(3.3)

where 3 dB is the beamwidth between half power points of the antenna pattern and D is the aperture dimension in the same units as the wavelength . The beamwidth of a horn antenna may depart from Eq. (3.2) quite radically. For example, a small rectangular horn will produce a narrower beam than suggested by Eq. (3.2) in its E plane and a wider beamwidth in the H plane. Since both Eqs. (3.2) and (3.3) contain antenna dimension parameters, the gain and beamwidth of an aperture antenna are related. For antennas with A  60%, the gain is approximately G  33,000  1u3 dB 2 2

(3.4)

where 3 dB is in degrees and G is not in decibels. If the beam has different beamwidths in orthogonal planes, 3 dB should be replaced by the product of the two 3 dB beamwidths. Values of the constant in Eq. (3.3) vary between different sources, with a range 28,000 to 35,000. The value 33,000 is typical for reflector antennas used in satellite communication systems. EXAMPLE 3.6.1 Global Beam Antenna The earth subtends an angle of 17° when viewed from geostationary orbit. What are the dimensions and gain of a horn antenna that will provide global coverage at 4 GHz? If we design our horn to give a circularly symmetric beam with a 3-dB beamwidth of 17° using Eq. (3.2) Dl  75 1u3 dB 2  4.4

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83

At 4 GHz,   0.075 m, so D  0.33 m (just over 1 ft). If we use a circular horn excited in the TE11 mode, the beamwidths in the E and H planes will not be equal and we may be forced to make the aperture slightly smaller to guarantee coverage in the E plane. A corrugated horn designed to support the HE hybrid mode has a circularly symmetric beam and could be used in this application. Waveguide horns are generally used for global beam coverage. Reflector antennas are not efficient when the aperture diameter is less than 8. Using Eq. (3.4), the gain of the horn is approximately 100, or 20 dB, at the center of the beam. However, in designing our communication system we will have to use the edge of beam gain figure of 17 dB, since those earth stations close to the earth’s horizon, as viewed from the satellite, are close to the 3 dB contour of the transmitted beam. 

EXAMPLE 3.6.2 Regional Coverage Antenna The continental United States (48 contiguous states) subtends an angle of approximately 6°  3° when viewed from geostationary orbit. What dimension must a reflector antenna have to illuminate half this area with a circular beam 3° in diameter at 11 GHz? Can a reflector be used to produce a 6°  3° beam? What gain would the antenna have? Using Eq. (3.2), we have for a 3° circular beam Dl  753  25 and with   0.0272 m, D  0.68 m (just over 2 ft). The gain of this antenna, from Eq. (3.3) is approximately 35 dB. To generate a beam with different beamwidths in orthogonal planes we need an aperture with different dimensions in the two planes. In this case, a rectangular aperture 25  12.5 would generate a beam 6°  3°, and would have a gain of 32 dB, approximately. In order to illuminate such a reflector, a horn with unequal beamwidths is required, since the reflector must intercept most of the radiation from the feed if it is to have an acceptable efficiency. Rectangular, or more commonly elliptical, outline reflectors are used to generate unequal beamwidths. When orthogonal polarizations are to be transmitted or received, it is better to use a circular reflector with a distorted profile to broaden the beam in one plane, or a feed cluster to provide the appropriate amplitude and phase distribution across the reflector. 

Satellite Antennas in Practice The antennas of a communications satellite are often a limiting element in the complete system. In an ideal satellite, there would be one antenna beam for each earth station, completely isolated from all other beams, for transmit and receive. However, if two earth stations are 300 km apart on the earth’s surface and the satellite is in geostationary orbit, their angular separation at the satellite is 0.5°. For 3 dB to be 0.5°, D must be 150, which requires an aperture diameter of 11.3 m at 4 GHz. Antennas this large have been flown on satellites (ATS-6 deployed a 2.5 GHz, 10-m diameter antenna, for example), and large unfurled antennas are used to create multiple spot beams from GEO satellites serving mobile users. However, at 20 GHz, an antenna with D  150 is only 1.5 m wide, and such an antenna can readily be flown on a 3020 GHz satellite. A phased array feed is used to create many 0.5° beams which can be clustered to serve the coverage zone of the satellite. To provide a separate beam for each earth station would also require one antenna feed per earth station if a multiple-feed antenna with a single reflector were used. A compromise between one beam per station and one beam for all stations has been used in many satellites by using zone-coverage beams and orthogonal polarizations within the same beam to provide more channels per satellite. Figure 3.3b shows a GEO satellite that

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West spot beam West zone beam

West hemi beam

East spot beam East zone beam

East hemi beam

Subsatellite point FIGURE 3.17 Contour plot of the spot beam of ESA’s OTS satellite projected onto the earth. The contours are in 1 dB steps, normalized to 0 dB at the center of the beam. (Courtesy of ESA.)

has four reflector antennas. Each reflector is illuminated by a complex feed that provides the required beam shape to permit communication between earth stations within a given coverage zone. Figure 3.17 shows the coverage zones provided by a typical Intelsat satellite. The largest reflector on the satellite transmits at 4 GHz and produces the “peanut” shaped patterns for the zone beams, which are designed to concentrate the transmitted energy onto densely populated areas such as North America and western Europe where much telecommunications traffic is generated. The smaller antennas are used to provide hemisphere transmit and receive beams, and the 1411 GHz spot beams. In addition, there are horn antennas providing global beam coverage. Countries such as the United States create an enormous demand for communication services, and a number of domestic satellite communication systems have been established to meet that demand. In 2000 the geostationary orbit had domestic satellites spaced every 2°, operating at 64 GHz and 1411 GHz from longitude 60° W to 140° W. This encompasses all orbital locations that can be simultaneously viewed by earth stations in the United States and Canada, and each operator has been given a limited number of orbital slots in which to place a satellite. As a result, there is a great deal of pressure on the operating companies to obtain the maximum number of channels per satellite in order to give the operator the greatest possible revenue-earning capacity. This has encouraged the development of frequency reuse antennas by means of orthogonal polarizations and multiple beams, the combination of 64 and 1411 GHz communication systems on one satellite, and the use of multilevel digital modulation and TDMA to increase capacity.

SIDEBAR Some of the largest commercial GEO satellites proposed to date are the Inmarsat I-4 series19. These satellites are based on the Matra Marconi Space Eurostar 3000 platform, a 5 metric ton three-axis stabilized satellite with 9 kW onboard power. The satellites have

200 spot beams for mobile services and Internet access. Three Inmarsat I-4 satellites are to be built by the European consortium Astrium at a total cost of $700 M, with two scheduled for launch in 2003–2004. The third satellite is a spare.

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FIGURE 3.18 Intelsat VI satellite on station.

The requirements of narrow antenna beams with high gain over a small coverage zone leads to large antenna structures on the satellite. Frequently, the antennas in their operating configuration are too large to fit within the shroud dimensions of the launch vehicle, and must be folded down during the launch phase. Once in orbit, the antennas then can be deployed. In many larger satellites, the antennas use offset paraboloidal reflectors with clusters of feeds to provide carefully controlled beam shapes. The feeds mount on the body of the satellite, close to the communications subsystem, and the reflector is mounted on a hinged arm. Figure 3.18 shows an example of this design of antenna for the INTELSAT VI satellite. For launch, the solid reflectors fold down to provide a compact structure; in orbit, the hinged arms are swung out and locked in place to hold the reflectors in the correct position. When the satellite is in geostationary orbit it is weightless, so very little energy is required to move the large reflector.

SIDEBAR One interesting idea is the inflatable antenna, several examples of which have been flown experimentally18. The antenna can be squeezed into a small space for launch and inflated from a pressurized gas bottle when the satellite is in orbit. Once inflated, a foam material emitted along with the inflation gas hardens to make a rigid structure. Plastic materials

can be sprayed with a metallic coating made up of very small particles of aluminum to create a reflecting surface for electromagnetic (EM) waves. Inflatable antennas can be made very large without a significant weight penalty, and are therefore attractive for any satellite requiring multiple narrow beams.

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ATS-6 in orbit

1. After separation

2. Solar array booms extended

4. 30-ft reflector deploys

3. Solar array panels extended

5. Fully deployed configuration

FIGURE 3.19 Deployment sequence of ATS-6 10-m antenna. (Courtesy of NASA.)

Figure 3.19 shows the deployment sequence used for the 30-ft antenna carried by ATS-6: the antenna was built as a series of petals that folded over each other to make a compact unit during launch, which then unfurled in orbit. The solar sails folded down over the antenna, and were deployed first. Springs or pyrotechnic devices can be used to provide the energy for deployment of antennas or solar sails, with a locking device to ensure correct positioning after deployment. Similar unfurlable antennas are used on GEO satellites that provide satellite telephone service at L band using multiple narrow beams.

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3.7 EQUIPMENT RELIABILITY AND SPACE QUALIFICATION Communications satellites built in the 1980s and 1990s have provided operational lifetimes of up to 15 years. Once a satellite is in geostationary orbit, there is little possibility of repairing components that fail or adding more fuel for station keeping. The components that make up the satellite must therefore have very high reliability in the hostile environment of outer space, and a strategy must be devised that allows some components to fail without causing the entire communication capacity of the satellite to be lost. Two separate approaches are used: space qualification of every part of the satellite to ensure that it has a long life expectancy in orbit and redundancy of the most critical components to provide continued operation when one component fails.

Space Qualification Outer space, at geostationary orbit distances, is a harsh environment. There is a total vacuum and the sun irradiates the satellite with 1.4 kW of heat and light on each square meter of exposed surface. Where surfaces are in shadow, heat is lost to the infinite sink of space, and surface temperature will fall toward absolute zero. Electronic equipment cannot operate at such extremes of temperature and must be housed within the satellite and heated or cooled so that its temperature stays within the range 0° to 75°C. This requires a thermal control system that manages heat flow throughout a GEO satellite as the sun moves around once every 24 h. Thermal problems are equally severe for a LEO satellite that moves from sunlight to shadow every 100 min. The first stage in ensuring high reliability in a satellite is by selection and screening of every component used. Past operational and test experience of components indicates which components can be expected to have good reliability. Only components that have been shown to have high reliability under outer space conditions will be selected. Each component is then tested individually (or as a subsystem) to ensure that it meets its specification. This process is known as quality control or quality assurance and is vital in building any equipment that is to be reliable. Once individual components and subsystems have been space qualified, the complete satellite must be tested as a system to ensure that its many systems are reliable. When a satellite is designed, three prototype models are often built and tested. The mechanical model contains all the structural and mechanical parts that will be included in the satellite and is tested to ensure that all moving parts operate correctly in a vacuum, over a wide temperature range. It is also subjected to vibration and shock testing to simulate vibration levels and G forces likely to be encountered on launch. The thermal model contains all the electronics packages and other components that must be maintained at the correct temperature. Often, the thermal, vacuum, and vibration tests of the entire satellite will be combined in a thermal vacuum chamber for what is known in the industry as a shake and bake test. The antennas are usually included on the thermal model to check for distortion of reflectors and displacement or bending of support structures. In orbit, an antenna may cycle in temperature from above 100°C to below 100°C as the sun moves around the satellite. The electrical model contains all the electronic parts of the satellite and is tested for correct electrical performance under total vacuum and a wide range of temperatures. The antennas of the electrical model must provide the correct beamwidth, gain, and polarization properties. Testing carried out on the prototype models is designed to overstress the system and induce failure in any weak components: temperature cycling will be carried out to 10% beyond expected extremes; structural loads and G forces 50% above those expected in

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flight may be applied. Electrical equipment will be subjected to excess voltage and current drain to test for good electronic and thermal reliability. The prototype models used in these tests will not usually be flown. A separate flight model (or several models) will be built and subjected to the same tests as the prototype, but without the extremes of temperature, stress, or voltage. Preflight testing of flight models, while exhaustive, is designed more to cause failure of parts, rather than to check that they will operate under worst-case conditions. Space qualification is an expensive process, and one of the factors that makes large GEO satellites expensive. Some low earth orbit satellites have been built successfully using less expensive techniques and relying on lower performance in orbit. LEO satellite systems require large numbers of satellites that are generally less expensive than large GEO satellites. The Iridium system, for example, was designed with 66 operational satellites in its constellation to provide continuous worldwide coverage, with at least eight spare satellites in orbit at any time. If one operational satellite fails, a spare is moved in to take its place. This allowed Iridium satellites to be built with a higher probability of failure than a GEO satellite. Experimental satellites have also been built using low cost techniques. The University of Surrey, U.K., for example, built a series of digital store and forward satellites that were used by radio amateurs and others which each cost less than $1 M15. Most of the components on the Surrey satellites were not space qualified, but were selected carefully to ensure best possible lifetimes at reasonable cost and then the entire satellite was subjected to shake and bake tests. Many of the electronic and mechanical components that are used in satellites are known to have limited lifetimes, or a finite probability of failure. If failure of one of these components will jeopardize the mission or reduce the communication capacity of the satellite, a backup, or redundant, unit will be provided. The design of the system must be such that when one unit fails, the backup can automatically take over or be switched into operation by command from the ground. For example, redundancy is always provided for traveling wave tube amplifiers used in the transponders of a communications satellite, as these are known to have a limited lifetime. The success of the testing and space qualification procedures used by NASA has been well illustrated by the lifetime achieved by many of its scientific satellites. Satellites designed for a specific mission lasting 1 or 2 years have frequently operated successfully for up to 25 years. Sufficient reliability was designed into the satellite to guarantee the mission lifetime such that the actual lifetime has been much greater. In the next section we will look at how reliability can be quantified.

Reliability We need to be able to calculate the reliability of a satellite subsystem for two reasons: we want to know what the probability is that the subsystem will still be working after a given time period, and we need to provide redundant components or subsystems where the probability of a failure is too great to be accepted. The owner of a satellite used for communications expects to be able to use a predetermined percentage of its communications capacity for a given length of time. Amortization of purchase and launch costs will be calculated on the basis of an expected lifetime. The manufacturers of satellites must provide their customers with predictions (or guarantees) of the reliability of the satellite and subsystems: to do this requires the use of reliability theory. Reliability theory is a mathematical attempt to predict the future and is therefore less certain than other mathematical techniques that operate in absolute terms. The application of reliability theory has enabled satellite engineers to build satellites that perform as expected,

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End of life Burn-in Probability of failure

0 Time FIGURE 3.20 Bathtub curve for probability of failure.

at acceptable construction costs. It should be noted, however, that the cost of a satellite is very high compared to other equipment with a comparable number of components: a large GEO satellite costs around $100 M to build, close to the cost of a Boeing 747 jet airliner. The cost is acceptable because of the high revenue-earning capability of the satellite. The reliability of a component can be expressed in terms of the probability of failure after time t, PF(t). For most electronic equipment, probability of failure is higher at the beginning of life—the burn-in period—than at some later time. As the component ages, failure becomes more likely, leading to the bathtub curve shown in Figure 3.20. Components for satellites are selected only after extensive testing. The aim of the testing is to determine reliability, causes of failure, and expected lifetime. The result is a plot similar to Figure 3.20. Testing is carried out under rigorous conditions, representing the worst operating conditions likely to be encountered in space, and may be designed to accelerate failure in order to shorten the testing duration needed to determine reliability. Units that are exposed to the vacuum of space are tested in a vacuum chamber, and components subjected to sunlight are tested under equivalent radiant heat conditions. The initial period of reduced reliability can be eliminated by a burn-in period before a component is installed in the satellite. Semiconductors and integrated circuits that are required to have high reliability are subjected to burn-in periods from 100 to 1000 h, often at a high temperature and excess voltage to induce failures in any suspect devices and to get beyond the initial low reliability part of the bathtub curve.

SIDEBAR The bathtub curve is familiar to owners of automobiles. A new car may have defects when it is delivered, and errors in manufacturing may lead to components failing soon after purchase. This is one of the reasons that manufacturers offer warranties for the initial life of their products. Once these defects have been overcome, by repair or replacement, reliability improves for a number of years until mechanical parts start to wear and failures occur. In an automobile, preventive maintenance can be carried

out to replace parts that are known to wear most quickly. For example, spark plugs and drive belts may be replaced every 40,000 miles. The skill in owning an automobile is to judge the time at which the vehicle is starting up the end-of-life portion of the bathtub curve. That is the optimum time to dispose of the vehicle, and the worst possible time to buy one. Preventive maintenance is not possible with a geostationary satellite, at present, so other strategies must be adopted.

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The reliability of a device or subsystem is defined as R1t2 

Ns 1t2 Number of surviving components at time t  N0 Number of components at start of test period

(3.5)

The numbers of components that failed in time t is Nf (t) where Nf 1t2  N0  Ns 1t2

(3.6)

From the engineering viewpoint, what we need to know is the probability of any one of the N0 components failing: this is related to the mean time before failure (MTBF). Suppose we continue testing devices until all of them fail. The ith device fails after time ti where MTBF  m 

1 N0 a ti N0 i1

(3.7)

The average failure rate , is the reciprocal of the MTBF, m. If we assume that  is a constant, then l

Number of failures in a given time Number of surviving components

l

1 ¢Nf 1 dNf   1 MTBF Ns ¢t Ns dt

(3.8)

Failure rate  is often given as the average failure rate per 109 h. The rate of failure, dNfdt, is the negative of the rate of survival dNsdt, so we can redefine  as l

1 dNs Ns dt

(3.9)

By definition from Eq. (3.5), the reliability R is NsN0, so l

1 d 1 dR 1N0 R2  N0 R dt R dt

(3.10)

A solution of Eq. (3.10) is R  elt

(3.11)

Thus the reliability of a device decreases exponentially with time, with zero reliability after infinite time, that is, certain failure. However, end of useful life is usually taken to be the time tl, at which R falls to 0.37(1e), which is when tl  1 l  m

(3.12)

The probability of a device failing, therefore, has an exponential relationship to the MTBF and is represented by the right-hand end of the bathtub curve.

Redundancy The equations in the preceding section allow us to calculate the reliability of a given device when we know its MTBF. In a satellite, many devices are used, each with a different MTBF, and failure of one device may cause catastrophic failure of a complete subsystem. If we incorporate redundant devices, the subsystem can continue to function correctly. We can define three different situations for which we want to compute subsystem reliability:

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R1

R2

R3

91

RN

(a)

R1

R2 (b )

R1

R2

R3

R4

(c)

R1 S1

R2

R3 (d )

S2

FIGURE 3.21 Redundancy connections. (a) Series connection. (b) Parallel connection. (c) Series/parallel connection. (d) Switched connection.

series connection, used in solar cells arrays, parallel connection, used to provide redundancy of the high power amplifiers in satellite transponders, and a switched connection, often used to provide parallel paths with multiple transponders. These are illustrated in Figure 3.21; also shown is a hybrid arrangement, a series/parallel connection, widely used in electronic equipment. The switched connection arrangement shown in Figure 3.21d is also referred to as ring redundancy since any component can be switched in for any other. Switches S1 and S2 are a little more complicated than as shown, affording the choice of multiple paths in an M for N ring redundancy configuration. The important point to note is that the active devices (R1, R2, p , Rn) have sufficient bandwidth, power output range, etc., to be able to handle any of the channels that might be switched through to them. Most TWTAs and SSPAs are such wideband, large power range devices. An example of parallel redundancy for the HPA of a 64 GHz bent pipe transponder is shown in Figure 3.22. The transponder translates incoming signals in the 6-GHz uplink band by 2225 MHz and retransmits them in the 4-GHz band. The high-power output stage of the transponder has two parallel TWT amplifiers. One TWTA will be switched off, but must present a matched load when both off and on. If one TWTA fails, the other is switched on either automatically, or by command from earth. The TWT is a thermionic device with a heated cathode and a high voltage power supply. In common with other

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TWTA #1

6 GHz 4 GHz BPF Downconverter BPF

LNA

Combiner

4 GHz LPA

S

6 GHz uplink antenna

Local oscillator 2225 MHz

C

4 GHz downlink antenna

Splitter

Redundant 4 GHz HPAs

TWTA #2

FIGURE 3.22 Redundant TWTA configuration in HPA of a 64 GHz bent pipe transponder.

thermionic devices such as cathode ray tubes and magnetrons, they have a relatively short MTBF. Although the MTBF may be 50,000 h, this is the period after which 50% of such devices will have failed, on average. The parallel connection of two TWTs, as shown in Figure 3.22 raises the reliability of the amplifier stage to 0.60 at the MTBF period, assuming zero probability of a short circuit. A lifetime of 50,000 h is approximately 6 years of continuous operation, which is close to the typical design lifetime of a satellite. To further improve the reliability of the transponders, a second redundant transponder may be provided with switching between the two systems. Note that a combination of parallel and switched redundancy is used to combat failures that are catastrophic to one transponder channel and to the complete communication system.

3.8

SUMMARY

Satellites that carry communications relays must provide a stable platform in orbit. Large GEO satellites have payload design lives that exceed 10 years and sufficient fuel to provide a maneuvering lifetime that typically exceeds 15 years. The satellite must carry a number of subsystems to support its communications mission. The attitude and orbital control system keeps the satellite in the correct orbit and on station, and pointing in the correct direction. The telemetry, tracking, command, and monitoring system allows an earth station to control the subsystems in the satellite and to monitor their health. The power system provides the electrical energy needed to run the satellite (housekeeping) and the communications system. Solar cells generate the electrical power, a power conditioning unit controls its distribution, and batteries provide power during launch and eclipses.

Satellites often employ frequency reuse, either by using the same frequencies again in spatially separated beams or by using the same frequencies in orthogonal polarizations within the same beam. Sometimes, both reuse techniques are used simultaneously. Frequency reuse allows the same RF spectrum to be used more than once to increase the satellite’s capacity. Antennas are a limiting factor in all radio communication systems. Very complex antennas have been developed for satellites to provide multiple beams and orthogonal polarizations from a single antenna. Reflector antennas with clustered feeds and phased array antennas are used to generate shaped and multiple beams. Reliability is an important issue in satellites. Redundancy can be used to provide additional receivers and high-power amplifiers that can take over when a unit fails.

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REFERENCES 1. WALTER L. MORGAN, GARY D. GORDON, Communication Satellites Handbook, Wiley Interscience, New York, 1989. 2. IEEE Transactions on Aerospace and Electronic Systems, AS-1 through AS-37, 1963–2000, Institute of Electronic and Electrical Engineers, New York. 3. “Astronautics and Aeronautics,” Journal of the American Institute of Aeronautics and Astronautics, Vol. 1 through 37, 1963–2000, AIAA, 1633 Broadway, New York. 4. J. ALPER and J. N. PELTON, eds., “The Intelsat Global Satellite System,” Progress in Astronautics and Aeronautics, Vol. 93, The American Institute of Aeronautics and Astronautics, New York, 1984. 5. KEIGLER, W. LINDORTER, and L. MUHLFELDER, “Stable Attitude Control for Synchronous Communications Satellites,” in Progress in Astronautics and Aeronautics, 33, MIT Press, Cambridge, MA, 1974. 6. G. E. SCHMIDT, Jr., “Magnetic Attitude Control for Geosynchronous Satellite,” Proceedings of the AIAA Communications Satellite Systems Conference, San Diego, CA, April 1978, pp. 110–112. 7. E. W. SCHMIDT, Hydrazine and Its Derivatives, Wiley, New York, 1984. 8. W. H. BRAUN and J. E. KEIGLER, “RCA Satellite Networks: High Technology and Low User Cost,” Proceedings of the IEEE, 72, 1483–1505, November 1984.

9. J. WOLKENSTEIN and J. N. LAPADRE, “Solid State Power Amplifiers Replacing TWTs in C-Band Satellites,” RCA Engineer, 27, 7, October/November 1982. 10. ROY MAUGER and CATHERINE ROSENBERG, “QoS Guarantees for Multimedia Services on a TDMA-based Satellite Network,” IEEE Communications Society Magazine Vol. 35, No. 7, July 1997. 11. www.globalstar.com 12. W. L. STUTZMAN and G. A. THIELE, Antenna Theory and Design, John Wiley & Sons, New York, 1981. 13. A. W. RUDGE, K. MILNE, A. D. OLVER, and P. KNIGHT, Handbook of Antenna Design, Volume 1, IEE Electromagnetic Wave Series No. 15, Peter Perigrinus Ltd., Stevenage, Herts, UK, 1983. 14. S. SILVER, ed., Microwave Antenna Theory and Design, originally published as Vol. 12 of the MIT Radiation Laboratory Series, 1949. (Republished by Peter Perigrinus Ltd., Stevenage, Herts, UK, 1984.) 15. www.ee.surrey.ac.uk/SSC/SSHP/sshp.html 16. www.astrolink.com 17. www.hns.com.spaceway 18. “Inflatable Structures Taking to Flight,” Aviation Week, pp. 60–62, January 25, 1999. 19. “Astrium Signs Inmarsat Deal,” Aviation Week, p. 39, May 22, 2000.

PROBLEMS 1. The telemetry system of a geostationary communications satellite samples 100 sensors on the spacecraft in sequence. Each sample is transmitted to earth as an 8-bit word in a TDM frame. An additional 200 bits are added to the frame for synchronization and status information. The data are then transmitted at a rate of 1 kilobit per second using BPSK modulation of a low-power carrier. a. How long does it take to send a complete set of samples to earth from the satellite? b. Including the propagation delay, what is the longest time the earth station operator must wait between a change in a parameter occurring at the spacecraft and the new value of that parameter being received via the telemetry link? (Assume a path length of 40,000 km.) 2. A spinner satellite has solar cells wrapped round a cylindrical drum 3.00 m in diameter, with a height of 5.0 m on station. The drum is rotated at 60 rpm to spin-stabilize the satellite. At the end of life, the solar cells are required to deliver 4.0 kW of electrical power. a. Calculate the efficiency of the solar cells at end of life. Assume an incident solar power of 1.39 kW/m2,

and that the effective solar radiation absorbing area of the solar cells is equal to the cross-sectional area of the drum. b. If the solar cells degrade by 15% over the lifetime of the satellite, so that the end-of-life output power is 85% of the beginning-of-life output power, what is the output of the solar cells immediately after launch? c. If the drum covered in solar cells of the spinner design had been replaced by solar sails that rotated to face the sun at all times, what area of solar sails would have been needed? Assume that cells on solar sails generate only 90% of the power of cells on a spinner due to their higher operating temperature. 3. A Direct Broadcast Television (DBS-TV) satellite is in geostationary orbit. The electrical power required to operate the satellite and its transmitters is 4 kW. Two designs of satellite can be used: three axis stabilized with solar cells and a spinner. a. A three-axis stabilized satellite has two solar sails of equal area that rotate to face the sun at all times. The efficiency of the solar cells at end of life is predicted to be 15%.

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Calculate the area of cells required by the GEO satellite, and the length of each sail if its width is 2.0 m. b. A spinner design of DBS-TV satellite is made up from a drum coated in solar cells. The drum has a diameter of 3.5 m. The efficiency of the solar cells is predicted to be 18% at end of life. Since half the solar cells are in darkness, and some are weakly illuminated by the sun, the effective area of solar cells on a spinner is equal to the diameter of the satellite multiplied by the height of the solar cells on the drum. Calculate the height of the drum to provide the required 4 kW of electrical power. 4. Batteries make up a significant part of the in-orbit weight of a communications satellite but are needed to keep the communications system operating during eclipses. A direct broadcast TV satellite requires 500 W of electrical power to operate the housekeeping functions of the satellite and 5 kW to operate its 16 high-power transponders. The longest duration of an eclipse is 70 min, during which time the batteries must provide power to keep the satellite operating, but the batteries must not discharge below 70% of their capacity. The satellite bus operates at 48 V. a. What is the current that must be supplied by the power conditioning unit to keep the satellite operating normally? b. Battery capacity is rated in ampere hours, the product of the current (in amps) that the battery can supply multiplied by the length of time that this current can be supplied before the battery is fully discharged. The satellite batteries must not discharge beyond 70% of their rated capacity during eclipse. Find the battery capacity required for this DBS-TV satellite. c. If batteries weigh 1.25 kg per ampere-hour of capacity, how much weight on this satellite is devoted to batteries? d. If half of the transponders are shut down during eclipse, what saving in battery weight is achieved? 5. A geostationary satellite provides service to a region which can be covered by the beam of an antenna on the satellite with a beamwidth of 1.8°. The satellite carries transponders for Ku band and Ka band, with separate antennas for transmit and receive. For center frequencies of 14.011.5 GHz and 30.020.0 GHz, determine the diameters of the four antennas on the satellite. a. Find the diameters of the two transmitting antennas. Specify the diameter and calculate the gain at each frequency. b. Find the diameters of the two receiving antennas. Specify the diameter and calculate the gain at each frequency.

6. A geostationary satellite provides communications within the United States at Ku band. The antennas on the satellite have beamwidths of 6° in the E–W direction and 3° in the N–S direction. A separate antenna is used for transmitting in the 11 GHz band and receiving in the 14 GHz band. a. Find the dimensions and estimate the gain of the transmitting antenna in the N–S and E–W directions. b. Find the dimensions and estimate the gain of the receiving antenna in the N–S and E–W directions. 6. The state of Virginia can be represented approximately on a map as an area bounded by 39.5° N latitude, 36.5° N latitude, 76.0° W longitude, and 83.0° W longitude. A geostationary satellite located at 79.5° W longitude has an antenna with a spot beam that covers all of Virginia at a downlink center frequency of 11.155 MHz. In this problem you will estimate the antenna dimensions subject to two different assumptions. In both cases use an aperture efficiency of 65%. a. The antenna is a circular parabolic reflector generating a circular beam with a 3-dB beamwidth equal to the diagonal of the area bounding the state of Virginia. Estimate the length of the diagonal by measuring the distance on a map of the United States, and calculate the beamwidth of the antenna from simple geometry. Hence determine the diameter of the antenna on the satellite in meters and its approximate gain in decibels. b. The antenna is an elliptical parabolic reflector with 3-dB beamwidths in the N–S and E–W directions that are equal to the height and the width of the area bounding the state of Virginia. Find the N–S and E–W dimensions from a map of the United States, and use geometry to calculate the required 3-dB beamwidths of the satellite antenna. Calculate the approximate gain of the antenna. 7. The state of Pennsylvania is approximately 1° wide (E–W) by one-half degree high (N–S) when viewed from geostationary orbit at a longitude of 75° W. Calculate: a. The dimensions of a downlink Ku-band antenna on a geostationary satellite with 3-dB beamwidths equal to the width and height of Pennsylvania. Use a frequency of 11.0 GHz. Identify the dimensions as E–W and N–S. b. The dimensions of an uplink Ka-band antenna on a geostationary satellite with 3-dB beamwidths equal to the width and height of Pennsylvania. Use a frequency of 30.0 GHz. Identify the dimensions as E–W and N–S. c. Suppose that the maximum dimension of the satellite at launch is 3 m wide, determined by the shroud of the ELV. Describe in a paragraph how you would

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launch the satellites in (a) and (b) above carrying: (a) the Ku-band antenna, and (b) the Ka-band antenna. 8. A constellation of low earth orbit satellites has an altitude of 1000 km. Each satellite has two multiple beam antennas that generate 16 beams. One antenna is used to transmit at 2.4 GHz and the other antenna receives at 1.6 GHz. a. Using simple geometry, find the coverage angle of the satellite antenna when the lowest elevation angle for an earth station is 10°. (Hint: Draw a diagram of the earth and the satellite and use the law of sines to solve the angles in a triangle.) b. Estimate the coverage area over the surface of the earth, in km.

95

c. Assuming that all 16 beams from the satellite antennas have equal beamwidths, determine the beamwidth of one beam. (Hint: Draw a circle representing the coverage area and fit 16 circles representing the 3-dB beamwidths of the beams inside the first circle.) d. Find the gain and the dimensions of each antenna on the satellite. 9. Calculate the total power radiated by the sun in watts and in dBW. Hint: The sun is 93 million miles (about 150 million kilometers) from the earth. At that distance, the sun produces a flux density of 1.39 kW/m2. This power density is present over all of a sphere with a radius of 150 million km.

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4

SATELLITE LINK DESIGN The design of a satellite communication system is a complex process requiring compromises between many factors to achieve the best performance at an acceptable cost. We will first consider geostationary satellite systems, since GEO satellites carry the vast majority of the world’s satellite traffic.

4.1

INTRODUCTION Figure 2.17 of Chapter 2 shows that the cost to build and launch a GEO satellite is about $25,000 per kg. Weight is the most critical factor in the design of any satellite, since the heavier the satellite the higher the cost, and the capital cost of the satellite must be recovered over its lifetime by selling communication services. The overall dimensions of the satellite are critical because the spacecraft must fit within the confines of the launch vehicle. When stowed for launch, the diameter of the spacecraft typically must be less than 3.5 m. Most large GEO satellites use deployable solar panels and antennas, but the antenna reflectors require accurate surfaces and are not folded for launch. This limits the maximum aperture dimension to about 3.5 m. As in most radio systems, antennas are a limiting factor in the capacity and performance of the communication system. The weight of a satellite is driven by two factors: the number and output power of the transponders on the satellite and the weight of station-keeping fuel. As much as half the total weight of satellites intended to remain in service for 15 years may be fuel. High power transponders require lots of electrical power, which can only be generated by solar cells. Increasing the total output power of the transponders raises the demand for electrical power and the dimensions of the solar cells, adding more weight to the satellite. Three other factors influence system design: the choice of frequency band, atmospheric propagation effects, and multiple access technique. These factors are all related, with the frequency band often being determined by what is available. Tables 4.1 and 4.2 tabulate

SIDEBAR The information carrying capacity of any radio communication link is determined by the RF power at the receiver input. Large antennas are needed to receive weak signals, and the signals from satellites in geostationary orbit are invariably weak. Early satellites were small and light, carrying small antennas and transponders with low output powers, which resulted in very weak signals at the earth’s surface. The earth stations required for communication with these satellites were large and expensive, with 25-m diameter antennas and high-power transmitters. The trend over the 35 years that GEO satellites have been in

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operation has been toward larger satellites with high output powers and large antennas and smaller earth stations, exemplified by VSAT networks and DBSTV. Direct to home satellite television broadcasting requires millions of receiving terminals, so these are made small and low cost, with the result that the DBSTV satellites are large and expensive. The designer of a satellite communication system must work to minimize the capital cost of the entire system, and must also ensure that sufficient revenue can be earned from the system to recover the large capital cost of building and launching satellites.

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TABLE 4.1 Major Frequency Allocations for Fixed Satellite Service and Broadcasting Satellites Frequency

Fixed satellite service

2320–2345 MHz 2500–2535 2500–2655 2655–2690

Down Down

3400–3700 3700–4200 4500–4800 5725–5850 5850–5925 5925–7075 7250–7450 7450–7550 7550–7750 8215–8400

Down Down Down, up Up region I Up region I Up Down, government Down, government Down, government Up, government

10.7–11.7 GHz 11.7–12.2 12.2–12.7

Down Down region II

12.50–12.75 12.75–13.25 14.00–14.25 14.25–14.50 14.5–14.8 17.3–17.7 17.7–18.6 18.1–18.6 18.6–18.8 18.8–19.7 27.0–27.5 27.5–29.5 30.0–31.0 37.5–39.5 39.5–40.5 40.5–42.5 42.5–43.5 47.2–50.2 50.4–51.4 71.0–75.5 81–84 84–86 92–95 102–105 149–164 202–217 231–241 265–275

Up region I and II, down region I

region II region II

Broadcasting satellites Radio broadcasting Down region III Down region II Down region II Up region II, III

Up region I Down regions I and III Down regions I and II, U.S. Direct Broadcast Satellite TV Up

Up Up Up Up Down Down Down regions I and III Down Up regions II and III Up Up Down Down Down Up Up Up Up Down Down Up Down Down Up Down Up

Regions I, II, and III are regions of the earth’s surface defined in the International Telecommunication Union’s Radio Regulations. Region I covers Europe, Africa, and northern Asia. Region II covers North and South America, and Region III covers the remainder of Asia.

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TABLE 4.2

Major Frequency Allocations for Mobile Satellite Services

Frequency

Aeronautical mobile

Maritime mobile

137–138 MHz 148–149.9 149.9–150.05 399.9–400.05 400.15–401 406–406.1 890–896 1559–1610 1530–1535 1535–1544 1544–1545 1545–1555 1555–1559 1559–1610 1610–1625.5 1625.5–1631.5 1631.5–1634.5 1634.5–1645.5 1645.5–1646.5 1646.5–1656.5 1656.5–1660 1660.0–1660.5 2483.5–2500 5.00–5.25 GHz 7.30–7.75 15.4–15.7 20.2–21.2 29.5–31.0 39.5–40.5 43.5–45.5 45.5–47.0 66.0–71.0 71.0–74.0 81.0–84.0 95.0–100 134–142 190–200 252–265

Down Down Down

Land mobile and other services Down, shared Up, shared Up, shared Up Down, shared Emergency beacons Region II (limited use) Shared with cellular radio Navigation satellite, down Down region I only

Down Down Navigation satellite, down Navigation satellite, up Up Up Up

Up Shared Up

Down

Down

Up Up Down

Up Up, government Down Down Up Down Up, government Up Down Up Down

Up Up, government Down Down Up Down Up, government Up Down Up Down

Up Up, government Down Down Up Down Up, government Up Down Down Down

Up Up

Regions I, II, and III are regions of the earth’s surface defined by the International Telecommunications Union. (See Table 4.1 for an explanation of their geographic locations.)

the most important frequencies allocated for satellite communications. The major bands are the 64 GHz, 1411 GHz, and 3020 GHz bands. (The uplink frequency is quoted first, by convention.) However, over much of the geostationary orbit there is already a satellite using both 64 GHz and 1411 GHz every 2°. This is the minimum spacing used for satellites in

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GEO to avoid interference from uplink earth stations. Additional satellites can only be accommodated if they use another frequency band, such as 3020 GHz. Rain in the atmosphere attenuates radio signals. The effect is more severe as the frequency increases, with little attenuation at 4 and 6 GHz, but significant attenuation above 10 GHz. Attenuation through rain (in decibels) increases roughly as the square of frequency, so a satellite uplink operating at 30 GHz suffers four times as much attenuation as an uplink at 14 GHz. Low earth orbit (LEO) and medium earth orbit (MEO) satellite systems have similar constraints to GEO satellite systems, but require more satellites which each serve a smaller area of the earth’s surface. Although the satellites are much closer to the earth than GEO satellites and therefore produce stronger signals, this advantage is usually lost since the earth terminals need low gain omnidirectional antennas because the position of the satellite is continually changing. LEO and MEO satellites use multiple beam antennas to increase the gain of the satellite antenna beams, and also to provide frequency reuse. Mobile satellite terminals must operate with low gain antennas at the mobile unit, and at as low a RF frequency as can be obtained. The link between the satellite and the major earth station (often called a hub station) is usually in a different frequency band as it is a fixed link. Figure 4.1 shows an illustration of a maritime satellite communication Maritime satellite

6-GHz uplink 1.6-GHz uplinks from ships

4-GHz downlink Interconnection to terrestrial network

Shore station 90-ft dish 1.5-GHz downlinks to ships

2-m dish

FIGURE 4.1

A maritime satellite communication system.

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system using a GEO satellite and L-band links to mobiles, with C-band links to a fixed hub station. All communication links are designed to meet certain performance objectives, usually a bit error rate (BER) in a digital link or a signal-to-noise ratio (SN) in an analog link, measured in the baseband channel. The baseband channel is where an information carrying signal is generated or received; for example, a TV camera generates a baseband video signal, and a TV receiver delivers a baseband video signal to the picture tube to form the images that the viewer watches. Digital data are generated by computers at baseband, and BER is measured at baseband. The baseband channel BER or SN ratio is determined by the carrier-to-noise ratio (CN) at the input to the demodulator in the receiver. In most satellite communications applications, the CN ratio at the demodulator input must be greater than 6 dB for the BER or SN objective to be achieved. Digital links operating at CN ratios below 10 dB must use error correction techniques to improve the BER delivered to the user. Analog links using frequency modulation (FM) require wideband FM to achieve a large improvement in SN ratio relative to CN ratio. The CN ratio is calculated at the input of the receiver, at the output terminals (or port) of the receiving antenna. RF noise received along with the signal and noise generated by the receiver are combined into an equivalent noise power at the input to the receiver, and a noiseless receiver model is used. In a noiseless receiver, the CN ratio is constant at all points in the RF and IF chain, so the CN ratio at the demodulator is equal to the CN ratio at the receiver input. In a satellite link there are two signal paths: an uplink from the earth station to the satellite, and a downlink from the satellite to the earth station. The overall CN at the earth station receiver depends on both links, and both, therefore must achieve the required performance for a specified percentage of time. Path attenuation in the earth’s atmosphere may become excessive in heavy rain, causing the CN ratio to fall below the minimum permitted value, especially when the 3020 GHz band is used, leading to a link outage. Designing a satellite system therefore requires knowledge of the required performance of the uplink and downlink, the propagation characteristics and rain attenuation for the frequency band being used at the earth station locations, and the parameters of the satellite and the earth stations. Additional constraints may be imposed by the need to conserve RF bandwidth and to avoid interference with other users. Sometimes, all of this information is not available and the designer must estimate values and produce tables of system performance based on assumed scenarios. It is usually impossible to design a complete satellite communication system at the first attempt. A trial design must first be tried, and then refined until a workable compromise is achieved. This chapter sets out the basic procedures for the design of satellite communication links, and includes design examples for a digital TV link using a GEO satellite and quadrature phase shift keying (QPSK) modulation, and a LEO satellite system for personal communication.

4.2

BASIC TRANSMISSION THEORY The calculation of the power received by an earth station from a satellite transmitter is fundamental to the understanding of satellite communications. In this section, we discuss two approaches to this calculation: the use of flux density and the link equation.

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Area A m2 Isotropic source EIRP = P t watts

Distance R m

Flux density F watts/m2 FIGURE 4.2

Flux density produced by an isotropic source.

Consider a transmitting source, in free space, radiating a total power Pt watts uniformly in all directions as shown in Figure 4.2. Such a source is called isotropic; it is an idealization that cannot be realized physically because it could not create transverse electromagnetic waves. At a distance R meters from the hypothetical isotropic source transmitting RF power Pt watts, the flux density crossing the surface of a sphere with radius R is given by F

Pt W/m2 4pR 2

(4.1)

All real antennas are directional and radiate more power in some directions than in others. Any real antenna has a gain G(), defined as the ratio of power per unit solid angle radiated in a direction  to the average power radiated per unit solid angle G1u2 

P1u2 P0 4p

(4.2)

where P() is the power radiated per unit solid angle by the antenna P0 is the total power radiated by the antenna G() is the gain of the antenna at an angle  1 The reference for the angle  is usually taken to be the direction in which maximum power is radiated, often called the boresight direction of the antenna. The gain of the antenna is then the value of G() at angle   0°, and is a measure of the increase in flux density radiated by the antenna over that from an ideal isotropic antenna radiating the same total power. For a transmitter with output Pt watts driving a lossless antenna with gain Gt, the flux density in the direction of the antenna boresight at distance R meters is F

PtGt W/m2 4pR 2

(4.3)

The product PtGt is often called the effective isotropically radiated power or EIRP, and it describes the combination of transmitter power and antenna gain in terms of an equivalent isotropic source with power PtGt watts, radiating uniformly in all directions.

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Isotropic source EIRP = P t watts

Incident flux density F watts/m2

Receiver

Pr

Receiving antenna with Area A m2, gain G r FIGURE 4.3 Power received by an ideal antenna with area A m2. Incident flux density is F  Pt /4R 2 W/m2. Received power is Pr  F  A  Pt A/4R 2 W.

If we had an ideal receiving antenna with an aperture area of A m2, as shown in Figure 4.3, we would collect power Pr watts given by Pr  F  A watts

(4.4)

A practical antenna with a physical aperture area of Ar m2 will not deliver the power given in Eq. (4.4). Some of the energy incident on the aperture is reflected away from the antenna, and some is absorbed by lossy components. This reduction in efficiency is described by using an effective aperture Ae where Ae  hA Ar

(4.5)

and A is the aperture efficiency of the antenna2. The aperture efficiency A accounts for all the losses between the incident wavefront and the antenna output port: these include illumination efficiency or aperture taper efficiency of the antenna, which is related to the energy distribution produced by the feed across the aperture, and also other losses due to spillover, blockage, phase errors, diffraction effects, polarization, and mismatch losses. For parabolodial reflector antennas, A is typically in the range 50 to 75%, lower for small antennas and higher for large Cassegrain antennas. Horn antennas can have efficiencies approaching 90%. Thus the power received by a real antenna with a physical receiving area Ar and effective aperture area Ae m2 is Pr 

PtGt Ae watts 4pR 2

(4.6)

Note that this equation is essentially independent of frequency if Gt and Ae are constant within a given band; the power received at an earth station depends only on the EIRP of the satellite, the effective area of the earth station antenna, and the distance R. A fundamental relationship in antenna theory2 is that the gain and area of an antenna are related by G  4pA e l2

(4.7)

where  is the wavelength (in meters) at the frequency of operation. Substituting for Ae in Eq. (4.6) gives Pr 

PtGtGr watts 14pRl2 2

(4.8)

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This expression is known as the link equation, and it is essential in the calculation of power received in any radio link. The frequency (as wavelength, ) appears in this equation for received power because we have used the receiving antenna gain, instead of effective area. The term [4R]2 is known as the path loss, Lp. It is not a loss in the sense of power being absorbed; it accounts for the way energy spreads out as an electromagnetic wave travels away from a transmitting source in three-dimensional space. Collecting the various factors, we can write Power received 

EIRP  Receiving antenna gain watts Path loss

(4.9)

In communication systems, decibel quantities are commonly used to simplify equations like Eq. (4.9). In decibel terms, we have Pr  EIRP  Gr  Lp dBW

(4.10)

where EIRP  10 log10 1Pt Gt 2 dBW

Gr  10 log10 14pAe l2 2 dB Path loss Lp  10 log10 3 14pRl2 2 4  20 log10 14pRl2 dB Equation (4.10) represents an idealized case, in which there are no additional losses in the link. It describes transmission between two ideal antennas in otherwise empty space. In practice, we will need to take account of a more complex situation in which we have losses in the atmosphere due to attenuation by oxygen, water vapor, and rain, losses in the antennas at each end of the link, and possible reduction in antenna gain due to mispointing. All of these factors are taken into account by the system margin but need to be calculated to ensure that the margin allowed is adequate. More generally, Eq. (4.10) can be written Pr  EIRP  Gr  Lp  La  L ta  L ra dBW

(4.11)

where La  attenuation in atmosphere Lta  losses associated with transmitting antenna Lra  losses associated with receiving antenna The conditions in Eq. (4.11) are illustrated in Figure 4.4. The expression dBW means decibels greater or less than 1 W (0 dBW). The units dBW and dBm (dB greater or less than 1 W and 1 mW) are widely used in communications engineering. EIRP, being the product of transmitter power and antenna gain is often quoted in dBW. Note that once a value has been calculated in decibels, it can readily be scaled if one parameter is changed. For example, if we calculated Gr for an antenna to be 48 dB, at a frequency of 4 GHz, and wanted to know the gain at 6 GHz, we can multiply Gr by (64)2. Using decibels, we simply add 20 log(64) or 20 log(3)  20 log(2)  9.5  6  3.5 dB. Thus the gain of our antenna at 6 GHz is 51.3 dB. Appendix A gives more information on the use of decibels in communications engineering.

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Satellite EIRP = Pt G t Loss L ta

Path loss L p

Atmospheric loss L a

Earth station antenna gain G r Loss L ra

LNA Received power Pr FIGURE 4.4

A satellite link. LNA, low noise amplifier.

EXAMPLE 4.2.1 A satellite at a distance of 40,000 km from a point on the earth’s surface radiates a power of 10 W from an antenna with a gain of 17 dB in the direction of the observer. Find the flux density at the receiving point, and the power received by an antenna at this point with an effective area of 10 m2. Using Eq. (4.3) F  PtGt 14pR 2 2  10  50 14p  14  107 2 2 2  2.49  1014 W/m2 The power received with an effective collecting area of 10 m2 is therefore Pr  2.49  1013 W The calculation is more easily handled using decibels. Noting that 10 log10 4  11.0 dB

Then

F in dB units  10 log10 1PtGt 2  20 log10 1R2  11.0  27.0  152.0  11.0  136.0 dB1W/m2 2 Pr  136.0  10.0  126 dBW.

Here we have put the antenna effective area into decibels greater than 1 m2 (10 m2  10 dB greater than 1 m2). 

EXAMPLE 4.2.2 The satellite in Example 4.2.1 operates at a frequency of 11 GHz. The receiving antenna has a gain of 52.3 dB. Find the received power.

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Using Eq. (4.10) and working in decibels Pr  EIRP  Gr  path loss 1dBW2 EIRP  27.0 dBW Gr  52.3 dB Path loss  14pRl2 2  20 log10 14pRl2 dB  20 log10 3 14p  4  107 2  12.727  102 2 4 dB  205.3 dB Pr  27.0  52.3  205.3  126.0 dBW We have the same answer as in Example 4.2.1 because the figure of 52.3 dB is the gain of a 10 m2 aperture at a frequency of 11 GHz. 

Equation (4.10), with other parameters for antenna and propagation losses, is commonly used for calculation of received power in a microwave link and is set out as a link power budget in tabular form using decibels. This allows the system designer to adjust parameters such as transmitter power or antenna gain and quickly recalculate the received power. The received power, Pr, calculated by Eqs. (4.6) and (4.8) is commonly referred to as carrier power, C. This is because most satellite links use either frequency modulation for analog transmission or phase modulation for digital transmission. In both of these modulation systems, the amplitude of the carrier is not changed when the data are modulated onto the carrier, so received carrier power C is always equal to received power Pr.

4.3 SYSTEM NOISE TEMPERATURE AND G/T RATIO Noise Temperature Noise temperature is a useful concept in communications receivers, since it provides a way of determining how much thermal noise is generated by active and passive devices in the receiving system. At microwave frequencies, a black body with a physical temperature, Tp degrees kelvin, generates electrical noise over a wide bandwidth. The noise power is given by3 Pn  kTpBn

(4.12)

where k  Boltzmann’s constant  1.39  1023 J/K  228.6 dBW/K/Hz Tp  physical temperature of source in kelvin degrees Bn  noise bandwidth in which the noise power is measured, in hertz Pn is the available noise power (in watts) and will be delivered only to a load that is impedance matched to the noise source. The term kTp is a noise power spectral density, in watts per hertz. The density is constant for all radio frequencies up to 300 GHz. We need a way to describe the noise produced by the components of a low noise receiver. This can conveniently be done by equating the component to a black body radiator with an equivalent noise temperature, Tn kelvins. A device with a noise temperature of Tn kelvins (symbol K, not K) produces at its output the same noise power as a black body at a temperature Tn degrees kelvin followed by a noiseless amplifier with the same

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gain as the actual device. The description of a low noise component by an equivalent noise source at the input of a noiseless amplifier is very useful because we can add noise temperatures to determine the total noise power in a receiver, as shown in the following analysis. Note that the unit of noise temperature is kelvins, not degrees kelvin. This distinction is often lost on the suppliers of consumer satellite broadcast receiving equipment. In satellite communication systems we are always working with weak signals (because of the large distances involved) and must make the noise level as low as possible to meet the CN ratio requirements. This is done by making the bandwidth in the receiver, usually set by the IF amplifier stages, to be just large enough to allow the signal (carrier and sidebands) to pass unrestricted, while keeping the noise power to the lowest value possible. The bandwidth used in Eq. (4.12) should be the equivalent noise bandwidth. Frequently we do not know the equivalent noise bandwidth and use the 3-dB bandwidth of our receiving system instead. The error introduced by using the 3-dB bandwidth is small when the filter characteristic of the receiver has steep sides. Noise temperatures from 30 K to 200 K can be achieved without physical cooling if GaAsFET (gallium arsenide field effect transistor) amplifiers are employed. GaAsFET amplifiers can be built to operate at room temperature with noise temperatures of 30 K at 4 GHz and 100 K at 11 GHz. Typically, noise temperatures increase with frequency, and an LNA for a 20 GHz receiver might have a noise temperature of 150 K. One might ask how an amplifier can have a noise temperature that is lower than its physical temperature. Noise temperature simply relates the noise produced by an amplifier to the thermal noise from a matched load at the same physical temperature placed at the input to the amplifier. If the amplifier produced no noise at all, its noise temperature would be 0 K. If the amplifier produces less noise than a matched load at the same physical temperature, its noise temperature will be lower than its physical temperature. To determine the performance of a receiving system we need to be able to find the total thermal noise power against which the signal must be demodulated. We do this by determining the system noise temperature, Ts. Ts is the noise temperature of a noise source, located at the input of a noiseless receiver, which gives the same noise power as the original receiver, measured at the output of the receiver and usually includes noise from the antenna. If the overall end-to-end gain of the receiver is Grx (Grx is a ratio, not in decibels) and its narrowest bandwidth is Bn Hz, the noise power at the demodulator input is Pno  kTs BnGrx watts

(4.13a)

where Grx is the gain of the receiver from RF input to demodulator input. The noise power referred to the input of the receiver is Pn where Pn  kTs Bn watts

(4.13b)

Let the antenna deliver a signal power Pr watts to the receiver RF input. The signal power at the demodulator input is PrGrx watts, representing the power contained in the carrier and sidebands after amplification and frequency conversion within the receiver. Hence, the carrier-to-noise ratio at the demodulator is given by Pr Grx Pr C   N kTs BnGrx kTs Bn

(4.14)

The gain of the receiver cancels out in Eq. (4.14), so we can calculate CN ratios for our receiving terminals at the antenna output port. This is convenient, because a link budget will find Pr at this point. Using a single parameter to encompass all of the sources of noise in a receiving terminal is very useful because it replaces several sources of noise in the receiver by a single system noise temperature, Ts.

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Signal from satellite Receiving antenna

LNA

BPF

Mixer

BPF

IF amplifier

Pr G rf

Gm

IF output

G IF

Local oscillator FIGURE 4.5

Simplified earth station receiver. BPF, bandpass filter.

Calculation of System Noise Temperature Figure 4.5 shows a simplified communications receiver with an RF amplifier and single frequency conversion, from its RF input to the IF output. This is the form used for all radio receivers, with few exceptions, known as the superhet (short for superheterodyne). The superhet receiver has three main subsystems: a front end (RF amplifier, mixer and local oscillator) an IF amplifier (IF amplifiers and filters), and a demodulator and baseband section. The RF amplifier in a satellite communications receiver must generate as little noise as possible, so it is called a low noise amplifier or LNA. The mixer and local oscillator form a frequency conversion stage that downconverts the RF signal to a fixed intermediate frequency (IF), where the signal can be amplified and filtered accurately.

SIDEBAR In general, it is difficult to make good narrowband filters with a ratio of bandwidth to center frequency less than 1%. If we want a 10-MHz bandwidth filter for a signal received from a satellite at 11 GHz, it is very difficult to implement the filter at the RF frequency of 11 GHz, where the filter bandwidth is less than 0.1% of the center frequency. We would instead down convert the 11-GHz signal to a frequency around 1 GHz, where the 10-MHz bandwidth is 1% of the IF frequency. This is the advantage of the superhet receiver design: very accurate filters can be used by converting the signal to a convenient frequency. The downconversion in the front end is achieved by multiplying the received signal and the local oscillator frequency in a nonlinear device—a mixer. Multiplication of two RF signals creates products at their sum and difference frequencies; the frequency of the local oscillator (LO) is usually set to fsignal  fif. This is called low side injection. The receiver

could also receive another RF signal at a frequency fsignal  flo which would produce an output from the mixer at fif. This is called an image frequency. It is blocked by a band-pass filter in the RF amplifier that is wide enough to pass the wanted range of signal frequencies but has high attenuation for the image frequencies. One further advantage of the superhet receiver design is that tuning of the receiver can be done with the local oscillator. The IF is at a fixed frequency, and the local oscillator frequency is varied to select the wanted signal. The front end is followed by an IF amplifier stage which contains bandpass filters that exactly match the spectrum of the received signal. In many small earth stations, known as very small aperture terminals (VSATs), the LNA, LO, and first IF amplifier and filters are all included in a single package called a low noise block converter (LNB) located immediately behind the antenna feed.

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Signal from satellite Receiving antenna LNA

BPF

Mixer

BPF

First IF amplifier

Second IF amplifier

Demodulator

Pr G rf First local oscillator

BPF

Mixer

BPF

D Tunable channel select filter

Second local oscillator (tunable)

Baseband output

FIGURE 4.6 Double conversion earth station receiver. The first downconversion shifts signals in a 500-MHz band to the first IF range 900–1400 MHz. The second downconverter has a tunable local oscillator and channel selection filter to select the wanted transponder signal in the second IF centered at 70 MHz.

Many earth station receivers use the double superhet configuration shown in Figure 4.6 which has two stages of frequency conversion. The front end of the receiver is mounted behind the antenna feed and converts the incoming RF signals to a first IF in the range 900 to 1400 MHz. This allows the receiver to accept all the signals transmitted from a satellite in a 500-MHz bandwidth at C band or Ku band, for example. The RF amplifier has a high gain and the mixer is followed by a stage of IF amplification. This section of the receiver is called a low noise block converter (LNB). The 900–1400 MHz signal is sent over a coaxial cable to a set-top receiver that contains another down-converter and a tunable local oscillator. The local oscillator is tuned to convert the incoming signal from a selected transponder to a second IF frequency. The second IF amplifier has a bandwidth matched to the spectrum of the transponder signal. Direct broadcast satellite TV receivers at Ku band use this approach, with a second IF filter bandwidth of 20 MHz. The equivalent circuits in Figure 4.7a can be used to represent a receiver for the purpose of noise analysis. The noisy devices in the receiver are replaced by equivalent noiseless blocks with the same gain and noise generators at the input to each block such that the block produces the same noise at its output as the device it replaces. The entire receiver is then reduced to a single equivalent noiseless block with the same end-to-end gain as the actual receiver and a single noise source at its input with temperature Tn. The total noise power at the output of the IF amplifier of the receiver in Figure 4.7a is given by Pn  GIF kTIF Bn  GIFGmkTmBn  GIFGmGRF kBn 1TRF  Tin 2

(4.15)

where GRF, Gm, and GIF are the gains of the RF amplifier, mixer, and IF amplifier, and TRF, Tm, and TIF are their equivalent noise temperatures. Tin is the noise temperature of the antenna, measured at its output port.

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Noiseless RF amplifier

Tin

Gain G RF

+

TRF

Noiseless IF amplifier

Noiseless mixer Gain Gm

+

Tm

109

Pn

Gain GIF

+

TIF

Equivalent noise sources (a ) Noiseless lossy device

Noiseless receiver

Tin

Gain G RF .Gm .G IF

+

Pn

Tin

Gain Gl

+

Pn

Noise source Tno

Noise source Ts (c )

(b )

FIGURE 4.7 (a) Noise model of receiver. The noisy amplifiers and downconverter have been replaced by noiseless units, with equivalent noise generators at their inputs. (b) Noise model of receiver. All noisy units have been replaced by one noiseless amplifier, with a single noise source Ts as its input. (c) Noise model for a lossy device. The lossy device has been replaced by a lossless device, with a single noise source Tno at its output.

Equation (4.15) can be rewritten as Pn  GIFGmGRF 3 1kTIFBn 2  1GRFGm 2  1kTmBn 2 GRF  1TRF  Tin 2 4  GIFGmGRF kBn 3TRF  Tin  TmGRF  TIF 1GRFGm 2 4

(4.16)

The single source of noise shown in Figure 4.7b with noise temperature Ts generates the same noise power Pn at its output if Pn  GIFGmGRF kTsBn

(4.17)

The noise power at the output of the noise model in Figure 4.7b will be the same as the noise power at the output of the noise model in Figure 4.7a if kTs Bn  kBn 3 1Tin  TRF  TmGIF  TIFGmGRF 2 4 Hence the equivalent noise source in Figure 4.7b has a system noise temperature Ts where Ts  3Tin  TRF  Tm GRF  TIF 1GmGRF 2 4

(4.18)

Succeeding stages of the receiver contribute less and less noise to the total system noise temperature. Frequently, when the RF amplifier in the receiver front end has a high gain, the noise contributed by the IF amplifier and later stages can be ignored and the system noise temperature is simply the sum of the antenna noise temperature and the LNA

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noise temperature, so Ts  Tantenna  TLNA. Note that the values for component gains in Eq. (4.18) must be linear ratios, not in decibels. The noise model shown in Figure 4.7b replaces all the individual sources of noise in the receiver by a single noise source at the receiver input. This assumes that all the noise comes in from the antenna or is internally generated in the receiver. In some circumstances, we need to use a different model to deal with noise that reaches the receiver after passing through a lossy medium. Waveguide and rain losses are two examples. When raindrops cause attenuation, they radiate additional noise whose level depends on the attenuation. We can model the noise emission as a noise source placed at the “output” of the atmosphere, which is the antenna aperture. The noise model for an equivalent output noise source is shown in Figure 4.7c, and produces a noise temperature Tno given by Tno  Tp 11  Gl 2

(4.19)

where Gl is the linear gain (less than unity, not in decibels) of the attenuating device or medium, and Tp is the physical temperature in degrees kelvin of the device or medium. For an attenuation of A dB, the value of Gl is given by Gl  10 A10

(4.20)

EXAMPLE 4.3.1 Suppose we have a 4-GHz receiver with the following gains and noise temperatures: Tin  25 K TRF  50 K TIF  1000 K Tm  500 K

GRF  23 dB GIF  30 dB

Calculate the system noise temperature assuming that the mixer has a gain Gm  0 dB. Recalculate the system noise temperature when the mixer has a 10-dB loss. How can the noise temperature of the receiver be minimized when the mixer has a loss of 10 dB? The system noise temperature is given by Eq. (4.18) Ts  3 25  50  15002002  110002002 4  87.5 K If the mixer had a loss, as is usually the case, the effect of the IF amplifier would be greater. For Gm  10 dB, the linear value is Gm  0.1 as a ratio. Then Ts  3 25  50  15002002  11000202 4  137.5 K

The lowest system noise temperatures are obtained by using a high gain LNA. Suppose we increase the LNA gain in this example to GRF  50 dB, giving ratio GRF  105. Ts  3 25  50  1500105 2  11000104 2 4  75.1 K

The high gain of the RF LNA amplifier has made the system noise temperature almost as low as it can go: Ts  Tin  TRF  75 K in this example. LNAs for use in satellite receivers usually have gains in the range 40–55 dB. 

EXAMPLE 4.3.2 The system illustrated in Example 4.3.1 has an LNA with a gain of 50 dB. A section of lossy waveguide with an attenuation of 2 dB is inserted between the antenna and the RF amplifier. Find the new system noise temperature for a waveguide temperature of 300°K. The waveguide loss of 2 dB (ratio 1.58) can be treated as a gain, Gl, that is less than unity: (Gl  11.58  0.631). The lossy waveguide attenuates the incoming noise and adds noise

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111

generated by its own ohmic loss. The equivalent noise generator placed at the output of the section of waveguide that represents the noise generated by the waveguide has a noise temperature Twg, where Twg  Tp 11  Gl 2  30011  0.6312  110.7 K The waveguide attenuates the noise from the antenna, so Tin  0.63  25  15.8 K The new system noise temperature, referred to the input of the LNA, is Ts  315.8  110.7  50  1500105 2  11000104 2 4  176.6 K We can refer the system noise temperature to the antenna output port by dividing the above result by Gl. This transfers the noise source from the LNA input to the waveguide input. Ts  176.60.631  279.9 K The new system noise temperature is 5.7 dB higher than the system noise temperature without the lossy waveguide. 

Note that when the system noise temperature is low, each 0.1 dB of attenuation ahead of the RF amplifier will add approximately 6.6 K to the system noise temperature. (Using the formula in Example 4.3.2 with Tp  290°K, Gl  0.1 dB  0.977 and Tno  290  0.023  6.6 K.) This is the reason for placing the front end of the receiver at the output of the antenna feed. Waveguide losses ahead of the LNA can have a disastrous effect on the system noise temperature of low noise receiving systems.

Noise Figure and Noise Temperature Noise figure is frequently used to specify the noise generated within a device. The operational noise figure (NF) is defined by the following formula4: NF 

1SN2 in 1S N2 out

(4.21)

Because noise temperature is more useful in satellite communication systems, it is best to convert noise figure to noise temperature, Td. The relationship is Td  T0 1NF  12

(4.22)

where the noise figure is a linear ratio, not in decibels, and where T0 is the reference temperature used to calculate the standard noise figure—usually 290 K. NF is frequently given in decibels and must be converted to a ratio before being used in Eq. (4.18). Table 4.3 gives a comparison between noise figure and noise temperature over the range encountered in typical systems. TABLE 4.3

Comparison of Noise Temperature and Noise Figure

Noise temperature (K) Noise figure (dB)

0 0

20 0.29

40 0.56

60 0.82

80 1.06

100 1.29

120 1.50

150 1.81

200 2.28

Noise temperature (K) Noise figure (dB)

400 3.8

600 4.9

800 5.8

1,000 6.5

1,500 7.9

2,000 9.0

3,000 10.5

5,000 12.6

10,000 15.5

290 3.0

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EXAMPLE 4.3.3 An amplifier has a quoted noise figure of 2.5 dB. What is its equivalent noise temperature? Using Eq. (4.22) Td  29011.78  12  226 K This value of noise temperature could then be used in Eq. (4.17), with other appropriate data, to calculate system noise temperature. 

G/T Ratio for Earth Stations The link equation can be rewritten in terms of (CN) at the earth station PtGtGr PtGr l 2 l 2 Gr C  c dc d  c dc d c d N kTsBn 4pR kBn 4pR Ts

(4.23)

Thus CN r Gr Ts, and the terms in the square brackets are all constants for a given satellite system. The ratio Gr Ts, which is usually quoted as simply GT in decibels, with units dB/K, can be used to specify the quality of a receiving earth station or a satellite receiving system, since increasing Gr Ts increases the received CN ratio. Satellite terminals may be quoted as having a negative GT which is below 0 dB/K. This simply means that the numerical value of Gr is smaller than the numerical value of Ts. EXAMPLE 4.3.4 An earth station antenna has a diameter of 30 m, has an overall efficiency of 68%, and is used to receive a signal at 4150 MHz. At this frequency, the system noise temperature is 79 K when the antenna points at the satellite at an elevation angle of 28°. What is the earth station GT ratio under these conditions? If heavy rain causes the sky temperature to increase so that the system noise temperature rises to 88 K, what is the new GT value? First calculate the antenna gain. For a circular aperture: Gr  hA4pAl2  hA 1pDl2 2 At 4150 MHz,   0.0723 m. Then

G  0.68  1p 300.07232 2  1.16  106

or 60.6 dB

Converting Ts into dBK Ts  10 log 79  19.0 dBK GT  60.6  19.0  41.6 dB/K If Ts  88 K in heavy rain, GT  60.6  19.4  41.2 dB/K

4.4



DESIGN OF DOWNLINKS The design of any satellite communication is based on two objectives: meeting a minimum CN ratio for a specified percentage of time, and carrying the maximum revenue earning traffic at minimum cost. There is an old saying that “an engineer is a person who can do for a dollar what any fool can do for one hundred dollars.” This applies to satellite communication systems. Any satellite link can be designed with very large antennas to achieve high CN ratios under all conditions, but the cost will be high. The art of good

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system design is to reach the best compromise of system parameters that meets the specification at the lowest cost. For example, if a satellite link is designed with sufficient margin to overcome a 20-dB rain fade rather than a 3-dB fade, earth station antennas with seven times the diameter are required. All satellite communications links are affected by rain attenuation. In the 64 GHz band the effect of rain on the link is small. In the 1411 GHz (Ku) band, and even more so in the 3020 GHz (Ka) band, rain attenuation becomes all important. Satellite links are designed to achieve reliabilities of 99.5 to 99.99%, averaged over a long period of time, typically a year. That means the CN ratio in the receiver will fall below the minimum permissible value for proper operation of the link for between 0.5 and 0.01% of the specified time; the link is then said to suffer an outage. The time period over which the percentage of time is measured can be a month, sometimes the “worst month” in attenuation terms, or a year. Rain attenuation is a very variable phenomenon, both with time and place. Chapter 8 discusses the prediction of path attenuation and provides ways to estimate the likely occurrence of outages on a given link. In this chapter we will simply assume certain rain attenuation statistics to use in examples of link design. C-band links can be designed to achieve 99.99% reliability because the rain attenuation rarely exceeds 1 or 2 dB. The time corresponding to 0.01% of a year is 52 min; at this level of probability the rain attenuation statistics are usually not stable and wide fluctuations occur from year to year. Outages occur in heavy rain, usually in thunderstorms, and thunderstorm occurrence varies widely. A link designed to have outages totaling 52 min each year may well have outages of several hours one year and none the next. Most Ka-band links cannot be designed to achieve 99.99% reliability because rain attenuation generally exceeds 10 dB, and often 20 dB, for 0.01% of the time. Outage times of 0.1 to 0.5% of a year (8 to 40 h) are usually tolerated in Kaband links. The allowable outage time for a link depends in part on the traffic carried. Telephone traffic needs real-time channels that are maintained for the duration of a call, so C band or Ku band is used for voice channels with sufficient link margin that outage times are small. Internet transmissions are less affected by short outages and generally do not require a real-time channel, making Ka band better suited for Internet access.

Link Budgets CN ratio calculation is simplified by the use of link budgets. A link budget is a tabular method for evaluating the received power and noise power in a radio link. Link budgets invariably use decibel units for all quantities so that signal and noise powers can be calculated by addition and subtraction. Since it is usually impossible to design a satellite link at the first attempt, link budgets make the task much easier because, once a link budget has been established, it is easy to change any of the parameters and recalculate the result. Tables 4.4a and 4.4b show a typical link budget for a C-band downlink using a global beam on a GEO satellite and a 9-m earth station antenna. The link budget must be calculated for an individual transponder, and must be repeated for each of the individual links. In a two-way satellite communication link there will be four separate links, each requiring a calculation of CN ratio. When a bent pipe transponder is used the uplink and downlink CN ratios must be combined to give an overall CN. In this section we will calculate the CN ratio for a single link. Later examples in this chapter demonstrate the evaluation of a complete satellite communication system.

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TABLE 4.4a C-Band GEO Satellite Link Budget in Clear Air C-band satellite parameters Transponder saturated output power Antenna gain, on axis Transponder bandwidth Downlink frequency band Signal FM-TV analog signal FM-TV signal bandwidth Minimum permitted overall C/N in receiver Receiving C-band earth station Downlink frequency Antenna gain, on axis, 4 GHz Receiver IF bandwidth Receiving system noise temperature Downlink power budget Pt  Satellite transponder output power, 20 W Bo  Transponder output backoff Gt  Satellite antenna gain, on axis Gr  Earth station antenna gain Lp  Free space path loss at 4 GHz Lant  Edge of beam loss for satellite antenna La  Clear air atmospheric loss Lm  Other losses Pr  Received power at earth station Downlink noise power budget in clear air k  Boltzmann’s constant Ts  System noise temperature, 75 K Bn  Noise bandwidth, 27 MHz N  Receiver noise power C/N ratio in receiver in clear air

20 W 20 dB 36 MHz 3.7–4.2 GHz 30 MHz 9.5 dB 4.00 49.7 27 75

GHz dB MHz K

13.0 2.0 20.0 49.7 196.5 3.0 0.2 0.5 119.5

dBW dB dB dB dB dB dB dB dBW

228.6 18.8 74.3 135.5

dBW/K/Hz dBK dBHz dBW

C/N  Pr  N  119.5 dBW  (135.5 dBW)  16.0 dB

Link budgets are usually calculated for a worst case, the one in which the link will have the lowest CN ratio. Factors which contribute to a worst case scenario include: an earth station located at the edge of the satellite coverage zone where the received signal is typically 3 dB lower than in the center of the zone because of the satellite antenna pattern, maximum path length from the satellite to the earth station, a low elevation angle at TABLE 4.4b

C-Band Downlink Budget in Rain

Prca  Received power at earth station in clear air 119.5 A  Rain attenuation 1.0 Prain  Received power at earth station in rain 120.5 Nca  Receiver noise power in clear air 135.5 Nrain  Increase in noise temperature due to rain 2.3 Nrain  Receiver noise power in rain 133.2 C/N ratio in receiver in rain C/N  Prain  Nrain  120.5 dBW  (133.2 dBW)  12.7

dBW dB dBW dBW dB dBW dB

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the earth station giving the highest atmospheric path attenuation in clear air, and maximum rain attenuation on the link causing loss of received signal power and an increase in receiving system noise temperature. The edge of the coverage pattern of the satellite antenna and the longest path usually go together. However, when a satellite has a multiple beam antenna, this may not always be the case. Earth station antennas are assumed to be pointed directly at the satellite, and therefore operate at their on-axis gain. If the antenna is mispointed, a loss factor is included in the link budget to account for the reduction in antenna gain. The calculation of carrier to noise ratio in a satellite link is based on the two equations for received signal power and receiver noise power that were presented in Sections 4.1 and 4.2. Equation 4.11 gives the received carrier power in dB watts as Pr  EIRP  Gr  Lp  La  Lr  Lt dBW

(4.24)

A receiving terminal with a system noise temperature TsK and a noise bandwidth Bn Hz has a noise power Pn referred to the output terminals of the antenna where Pn  kTsBn watts

(4.25)

The receiving system noise power is usually written in decibel units as N  k  Ts  Bn dBW

(4.26)

where k is Boltzmann’s constant (228.6 dBW/K/Hz), Ts is the system noise temperature in dBK, and Bn is the noise bandwidth of the receiver in dBHz. Note that because we are working in units of power, all decibel conversions are made as 10 log10(Ts) or 10 log10(Bn). The 20 log10 factor used in the calculation of path loss results from the (4R)2 term in the path loss equation.

Link Budget Example: C-Band Downlink for Earth Coverage Beam The satellite used in this example (see Tables 4.4a and 4.4b) is in geostationary earth orbit and carries 24 C-band transponders, each with a bandwidth of 36 MHz. The downlink band is 3.7–4.2 GHz and the satellite uses orthogonal circular polarizations to provide an effective RF bandwidth of 864 MHz. The satellite provides coverage of the visible earth, which subtends an angle of approximately 17 from a satellite in a geostationary orbit, by using a global beam antenna. Since antenna beamwidth and gain are linked together [3 dB beamwidth  1133,000 G2 where G is a ratio, not in decibels], the on-axis gain of the global beam antenna is approximately 20 dB. However, we must make the link budget calculation for an earth station at the edge of the coverage zone of the satellite where the effective gain of the antenna is 3 dB lower, at 17 dB. The CN ratio for the downlink is calculated in clear air conditions and also in heavy rain. An antenna with a gain of 20 dB has an effective aperture diameter of 5.6 wavelengths [G  (D)2], which gives D  0.42 m at a frequency of 4 GHz. The calculation of CN ratio is made at a mid-band frequency of 4 GHz. The saturated output power of the transponder is 20 W  13 dBW. We will assume an output back-off of 2 dB, so that the power transmitted by the transponder is 11 dBW. Hence the on-axis EIRP of the transponder and antenna is PtGt  11  20  31 dBW. The transmitted signal is a single 30-MHz bandwidth analog FM-TV channel in this example. Following common practice for analog TV transmission, the receiver noise bandwidth is set to 27 MHz, slightly less than the 30-MHz bandwidth of the FM-TV signal.

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The receiving earth station has an antenna with an aperture diameter of 9 m and a gain of 49.7 dB at 4 GHz, and a receiving system noise temperature of 75 K in clear air conditions. The GT ratio for this earth station is GT  49.7  10 log10 75  30.9 dBK1. The maximum path length for a GEO satellite link is 40,000 km, which gives a path loss of 196.5 dB at 4 GHz (  0.075 m). We must make an allowance in the link budget for some losses that will inevitably occur on the link. At C band, propagation losses are small, but the slant path through the atmosphere will suffer a typical attenuation of 0.2 dB in clear air. We will allow an additional 0.5-dB margin in the link design to account for miscellaneous losses, such as antenna mispointing, polarization mismatch, and antenna degradation, to ensure that the link budget is realistic. The earth station receiver CN ratio is first calculated for clear air conditions, with no rain in the slant path. The CN ratio is then recalculated taking account of the effects of rain. The minimum permitted overall CN ratio for this link is 9.5 dB, corresponding to the FM threshold of an analog satellite TV receiver. Table 4.4a shows that we have a downlink CN of 16.0 dB in clear air, giving a link margin of 6.5 dB. This link margin is available in clear air conditions, but will be reduced when there is rain in the slant path. Heavy rain in the slant path can cause up to 1 dB of attenuation at 4 GHz, which reduces the received power by 1 dB and increases the noise temperature of the receiving system. Using the output noise model discussed in the previous section with a medium temperature of 273 K, and a total path loss for clear air plus rain of 1.2 dB (ratio of 1.32), the sky noise temperature in rain is Tsky  273  11  1 1.322  66 K In clear air the sky noise temperature is about 13 K, the result of 0.2 dB of clear air attenuation. The noise temperature of the receiving system has therefore increased by (66  13) K  53 K to 75  53 K  128 K with 1 dB rain attenuation in the slant path, from a clear air value of 75 K. This is an increase in system noise temperature of 2.3 dB. We can now adjust the link budget very easily to account for heavy rain in the slant path without having to recalculate the CN ratio from the beginning. The received carrier power is reduced by 1 dB because of the rain attenuation and the system noise temperature is increased by 2.3 dB. Table 4.4b shows the new downlink budget in rain. The CN ratio in rain has a margin of 3.2 dB over the minimum permissible CN ratio of 9.5 dB for an analog FM-TV transmission. The CN margin will translate into a higher than needed SN ratio in the TV baseband signal, and can be traded off against earth station antenna gain to allow the use of smaller (and therefore lower cost) antenna. We should always leave a small margin for unexpected losses if we want to guarantee a particular level of reliability in the link. In this case, we will use a 2-dB margin and examine how the remaining 1.2 dB of link margin can be traded against other parameters in the system. A reduction in earth station antenna gain of 1.2 dB is a reduction in the gain value, as a ratio, of 1.32. Antenna gain is proportional to diameter squared, so the diameter of the earth station antenna can be reduced by a factor of 11.32  1.15, from 9 m to 7.8 m. We could transmit a QPSK signal from the satellite instead of an analog FM signal. Using the 27-MHz noise bandwidth receiver, we could transmit a digital signal at 54 Mbps using QPSK, but would require a minimum CN ratio in the receiver of 14.6 dB, allowing a 1-dB implementation margin and a minimum BER of 106. The link margin would be 1.9 dB under heavy rain conditions, so we would need to increase the earth station antenna diameter by a factor of 1.55 to 13.9 m to provide a CN ratio of 14.6 dB under heavy rain conditions. A 54-Mbps digital signal could carry seven digital TV signals

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using MPEG-2 compression, a much more attractive proposition than carrying a single analog FM-TV signal, although at the cost of a larger earth station antenna. Global beam antennas are not widely used, although most Intelsat satellites carry them. Regional TV signal distribution is much more common, so the C-band link in Tables 4.4a and 4.4b is more likely to use a regional antenna, serving the United States, for example, with a 6 by 3 beam. The gain of a typical satellite antenna providing coverage of the 48 contiguous states is 32.0 dB on axis (G  33,0001  2), which is 12.0 dB higher than the on-axis gain of a global beam. Using the link budget in Tables 4.4a and 4.4b, we can trade the extra 12-dB gain of a regional coverage satellite antenna for a reduction in earth station antenna dimensions. For the example of a 9.0-m antenna receiving analog FM-TV, we could reduce the antenna diameter by a factor of 4 to 2.25 m (approximately 7 ft 6 inch diameter). This is the smallest size of antenna used by home satellite TV systems operating in C band. The above examples show how the link budget can be used to study different combinations of system parameters. Most satellite link analyses do not yield the wanted result at the first try, and the designer or analyst must use the link budget to adjust system parameters until an acceptable result is achieved. More examples of link budgets are included later in this chapter.

4.5 SATELLITE SYSTEMS USING SMALL EARTH STATIONS There are many applications in which satellites carry only one or two telephone or data channels, or a direct-broadcast TV signal and use small, low-cost earth stations. In these cases, earth stations costing a few hundred or a few thousand dollars are needed. There are only two parameters in the equation for received power that we can adjust at the satellite to allow us to use a small receiving antenna: satellite transmitted power and satellite antenna beamwidth. In domestic satellite systems, narrow beams can be used for transmitting from the spacecraft to provide coverage over only the region that the system is designed to serve. Because the dimensions of the antennas that can be mounted on most spacecraft are limited, the coverage zone cannot be made arbitrarily small. The earth’s disk subtends an angle of about 17 when viewed from geostationary orbit, and can be illuminated with a microwave horn having an aperture a few wavelengths in diameter. At 4 GHz, to obtain a 4 spot beam, a dish 1.4 m in diameter is needed. As the frequency is increased, the diameter of the spacecraft antenna in wavelengths is increased for a given dish diameter, making it feasible to use more directive beams. However, unless a switched or multiple beam system is used, the single transmit (or receive) beam must cover the whole region that the domestic satellite serves. The topic of spacecraft antennas is explored in more detail in Chapter 3. As an example, consider the problem of providing service to the 48 contiguous states of the United States, known as Conus, as illustrated in Figure 4.8; the constraints then become apparent. Viewed from a geostationary orbit, at a longitude of around 100 W, the continental United States subtends an angle of about 3 in the latitude plane (N–S) and 6 in the longitude plane (E–W). Regardless of the frequency used, an aperture antenna to produce a single beam with 3-dB beamwidths of 6 by 3° has dimensions approximately 13 by 26, and a gain of 32 dB. For a receiving earth station at the edge of the coverage zone, the gain of the satellite antenna in that direction is typically 3 dB lower, or

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Equator

FIGURE 4.8 Domestic satellite link to the Continental United States viewed from geostationary orbit, a beam approximately 6°  3° is required for coverage of the 48 contiguous states. Additional beams may be provided for Alaska and Hawaii.

29 dB. For the GEO satellite system shown in Figure 4.8, the received carrier level and CN ratio can readily be calculated for an earth station with a 3-m antenna at the edge of the coverage zone, using a transponder with an output power of 5 W at 4 GHz, transmitting a single carrier. Ignoring all losses, and using a receiving antenna gain of 40.0 dB (63% efficiency) Pr  7.0  29.0  196.5  40.0  120.5 dBW

(4.27)

The noise power at the input to a low noise receiver with noise bandwidth of 30 MHz and system noise temperature of 100 K is N  kTsBn  228.6  20.0  74.8  133.8 dBW

(4.28)

Thus for this system the CN is 13.3 dB. This is some 3.8 dB above an FM threshold of 9.5 dB and provides an adequate margin for an operational system. These figures are typical of those used in U.S. domestic satellite systems designed for distribution of television programs to cable TV networks and broadcast TV stations using a single C-band transponder for each video signal and FM with analog video signals.

Direct Broadcast TV Direct broadcast satellite television (DBS-TV) originally started in Europe in the 1980s using analog FM transmission in Ku band. It achieved a reasonable measure of success, due in part to the much slower introduction of cable TV systems in Europe than

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SIDEBAR The threshold is at a CN of 9 dB for a nominal 30MHz RF bandwidth, where threshold is defined as an extra loss of 1 dB in SN ratio as the CN ratio is reduced.

S/N at demodulator output (dB)

FM demodulators exhibit a threshold effect which causes impulsive noise to appear at the output of the demodulator at low CN ratios. When noise in the receiver approaches the carrier magnitude, sudden rapid changes in phase of the FM signal can occur. The FM demodulator produces an output voltage proportional to the rate of change of phase (i.e., the instantaneous frequency) and interprets the rapid phase changes caused by noise as large voltage changes. The voltage spikes at the FM demodulator output occur at random time intervals, with increasing numbers of spikes as the CN ratio falls toward the threshold. In a telephone circuit, the noise pulses sound like clicks or crackles; they can easily be seen on an oscilloscope or counted by a pulse counter. In FM-TV systems, noise spikes that appear close to threshold cause black and white or colored dots across an otherwise good quality TV picture7. (In satellite TV jargon, these noise spikes are called sparklies.) The interested reader should refer to a text on communication system theory for an analysis of threshold effects in FM demodulators5,6. Figure 4.9 shows the characteristics of a typical FM threshold extension demodulator of a satellite FM-TV receiver.

50

40 FM threshold 30 0

10 20 30 C/N ratio at demodulator (dB)

FIGURE 4.9 Characteristics of a typical FM threshold extension demodulator for TV reception. The FM threshold is at CN  9 dB, where the SN at the demodulator output has fallen 1 dB below the straight line.

occurred in the United States. In the 1990s, digital transmission became possible, and several systems were developed in the United States in the 12.2 to 12.7 GHz band allocated to DBS-TV services. In the United States, DIRECTV, a system built by a consortium led by Hughes, has been very successful and had over 10 million customers by year-end 2000, offering two hundred television and audio channels. Another DBS-TV provider in the United States, Echostar, offered similar services and had 4 million customers in year 2000. TV and audio channels are available from DBS-TV providers in a mixture of subscription packages, much like cable TV companies offer, and as pay per view for individual movies and special events. In rural areas of the United States, DBS-TV offers hundreds of television channels in place of the three or four terrestrial broadcasting stations that are typically available. In city areas, DBS-TV offers an alternative to cable television at a similar cost. The development of low cost Ku-band antennas and receivers, and high speed digital integrated circuits specifically for DBS television, has made DBS-TV practical. Chapter 11 gives more detail on the design of DBS-TV systems. The 12.2- to 12.7-GHz band was set aside for exclusive use by DBS-TV satellites in geostationary orbit so that high-power transponders could be used on specially designed DBS-TV satellites. Typical transponder output levels are 100 to 200 W with a flux density at the earth’s surface of up to 100 dBW/m2. The satellites can carry 16 transponders, with a typical total transmitted RF power of 2.6 kW, higher than for other commercial satellites. DBS-TV satellites are large and heavy, generally use a threeaxis stabilized design, and have a large area of solar cells to generate the power required

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by the transponders. Typical mass for a DBS-TV satellite is 6800 kg at launch, among the largest commercial GEO satellites. The flux density at the earth surface produced by a DBS-TV 160-W transponder is typically in the range 105 to 115 dBW/m2, which allows small receiving antennas (dishes) to be used for DBS-TV reception, with diameters in the range 0.45–0.75 m. The small dish required for DBS-TV reception played a critical part in the acceptance and success of DBS-TV in the United States. Previously, DBS-TV reception of cable television signals was only possible at C band and Ku band with 2.0- to 3.5-m dishes. The local governments of many cities and towns refused to permit these large dishes in residential areas, although they became popular in rural areas and an estimated 4 M systems were sold in the 1980s. Congress passed laws in the 1990s that prevented local governments from restricting the use of antennas less than 1 m in diameter, opening up a large market for Ku-band DBS-TV services. The high flux density created by powerful transponders makes sharing of the DBSTV frequency bands impossible, so the 12-GHz DBS band, known as the Broadcast Satellite Service (BSS) band, is allocated exclusively for television broadcasting. The small home receiving antenna has a wide beam, typically 4 for a 0.45 m (18 inch) dish, which forces wide spacing of DBS-TV satellites to avoid interference by the signals from adjacent DBS-TV satellites. A 9 spacing in the GEO arc has been adopted in the United States, which restricts the number of DBS-TV satellites that can be placed in geostationary orbit to serve the United States. In the 1990s the U.S. FCC successfully auctioned spectrum and orbital locations for DBS-TV satellites, raising hundreds of millions of dollars from companies that foresaw a profitable commercial venture. The DBS-TV system must provide a received signal power at the small receiving antenna that has an adequate CN margin in clear sky conditions. Heavy rain will cause attenuation that exceeds the link margin, so occasional outages will be experienced, especially during the summer months when thunderstorms and heavy rain are more frequent. The CN margins used in DBS-TV systems are usually quite small to avoid the need for a large receiving antenna. The selection of a CN margin is a design trade-off between the outage level that customers can be expected to tolerate, the maximum allowable diameter of the receiving dish antenna, and the power output from the satellite transponders. Typical designs with receiving antennas in the 0.5 to 0.75 m range and 100–200 W satellite transponders yield rain attenuation margins of 3 to 8 dB depending on the location of the receiving terminal within the satellite antenna coverage, and outage times totaling 10 to 40 h per year. These link margins are much lower than those found in high capacity communication systems. Availability of the link has been exchanged for a smaller earth terminal antenna and lower equipment costs to the user. In discussing Ku-band rain attenuation, we will use statistics that are representative of many locations in the central and eastern parts of the United States, where typical path attenuation in rain exceeds 3 dB for 0.2% (15 h) and 6 dB for 0.01% (52 min) of an average year. Such attenuation levels at the given time percentage are typical of many temperate latitude locations such as the west coast of the United States, central United States, Virginia, and other states north of Virginia on the east coast of the United States, Europe, Chile, Uruguay, and New Zealand. (See Chapter 8 for details of rain attenuation probabilities and distributions.) DIRECTV claims that receiving systems designed for their DBSTV transmissions have an average annual availability exceeding 99.7%, which corresponds to an outage time of 0.3% of the year, or about 25 h. For much of the United States, this corresponds to rain attenuation in the slant path of 3 dB and requires a link margin of 5.7 dB when allowance is made for the increase in antenna noise temperature that accompanies 3 dB of rain attenuation.

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TABLE 4.5

121

Link Budget for Ku-Band DBS-TV Receiver

DBS-TV terminal received signal power Transponder output power, 160 W Antenna beam on-axis gain Path loss at 12.2 GHz, 38,000-m path Receiving antenna gain, on axis Edge of beam loss Clear sky atmospheric loss Miscellaneous losses Received power, C DBS-TV terminal receiver noise power Boltzmann’s constant, k System noise temperature, clear sky, 145 K Receiver noise bandwidth, 20 MHz Noise power, N DBS-TV terminal C/N in clear sky Clear sky overall C/N Link margin over 8.6-dB threshold Link availability throughout United States

22.0 34.3 205.7 33.5 3.0 0.4 0.4 119.7

dBW dB dB dB dB dB dB dBW

228.6 21.6 73.0 134.0

dBW/K/Hz dBK dBHz dBW

14.3 dB 5.7 dB Better than 99.7%

A representative link budget for a GEO DBS-TV system serving the United States is shown in Table 4.5. The threshold CN value is set at 8.6 dB, corresponding to a system using QPSK with an implementation margin of 0.8 dB, half rate forward error correction that produces 6 dB of coding gain, and a maximum BER of 105. This requires a clear sky CN ratio in the DBS-TV receiver of 8.6  5.7  14.3 dB. The link budget in Table 4.5 shows how the required clear sky CN is achieved for a receiver located on the 3 dB contour of the satellite antenna beam. A receiver located in the center of this beam would have a clear sky CN 3 dB higher, and a corresponding fade margin of 8.7 dB, sufficient to ensure only a few outages each year. In Table 4.5 a transponder output power of 160 W is used, with no backoff because a single QPSK signal is transmitted. The satellite antenna gain is 34.3 dB on axis, corresponding to a high efficiency antenna with a beam that is shaped to cover the land mass of the United States. The beam is approximately 5.5 wide in the E–W direction and 2.5 in the N–S direction. The resulting coverage zone, taking account of the earth’s curvature, is approximately 4000 km E–W and 2000 km N–S. A maximum path length of 38,000 km is used in this example. The receiving antenna is a high efficiency design with a frontfed offset parabolic reflector 0.45 m in diameter and a circularly polarized feed. The offset design ensures that the feed system does not block the aperture of the antenna, which increases its efficiency. The gain of this antenna is 33.4 dB at 12.0 GHz with an aperture efficiency of 67%. The receiver is located at the 3dB contour of the transmitting antenna, and miscellaneous losses of 0.4dB for clear sky attenuation at 12 GHz and 0.4 dB for receive antenna mispointing and other losses are allowed. The result is a received carrier power of 119.7 dBW in clear sky conditions. The noise power budget of the link is based on a receiver noise bandwidth of 20 MHz. The IF filters in the receiver must be designed to match the symbol rate of the transmitted signal, and to approximate a root raised cosine (RRC) transfer function. (See Chapter 5 for details of digital transmission techniques.) The noise bandwidth of all RRC filters is always equal to the symbol rate of the digital transmission. In the DBS-TV system

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described in Table 4.5, a QPSK signal with a symbol rate of 20 Msps is assumed, which results in a receiver noise bandwidth of 20 MHz. The 20-Msps QPSK transmission delivers a bit rate of 40 Mbps, but the half rate FEC coding reduces the data rate to 20 Mbps. A 20-Mbps data stream can carry three live compressed digital video signals using MPEG 2 encoding, or up to 10 prerecorded and processed video signals. The antenna noise temperature is set at 35 K in clear sky conditions, and a 12-GHz LNA with 110 K noise temperature is used. The result is a noise power of 134.0 dBW referred to the input of the LNA, and a clear sky CN ratio of 14.3 dB. This is a worst case result for clear sky conditions, since most of the DBS-TV receivers will lie inside the 3 dB contour of the satellite beam. DBS-TV receivers can be used outside the 3 dB contour of the satellite beam, but will have a lower link margin and consequently more outages per year, if heavy rain occurs frequently. For example, a receiver on the 5 dB contour of the satellite beam will have a link margin of 3.7 dB, which would allow about 2 dB of rain attenuation before the CN reaches threshold. If the user is in a relatively dry area, for example, central Canada, the performance of the receiving system may be quite acceptable. The CN ratio in the home receiver will fall when rain is in the path between the satellite and the receiving antenna. Much of the reduction in CN ratio is caused by an increase in the sky noise temperature. The following calculations show how the system noise temperature and (CN)dn are determined when rain attenuation is present. The first step is to determine the total path attenuation, A in dB, which is the sum of the clear sky path attenuation due to atmospheric gaseous absorption, Aca and attenuation due to rain, Arain. A  Aca  Arain dB

(4.29)

The sky noise temperature resulting from a path attenuation Atotal dB is found from the output noise model of Section 4.3 using an assumed medium temperature of 270 K for the rain. Tsky  270  11  10A10 2 K

(4.30)

The antenna noise temperature may be assumed to be equal to the sky noise temperature, although in practice not all of the incident noise energy from the sky is output by the antenna, and a coupling coefficient, c, of 90 to 95% is often used when calculating antenna noise temperature in rain. Thus antenna noise temperature may be calculated as TA  hc  Tsky K

(4.31)

Almost all satellite receivers use a high gain LNA as the first element in the receiver front end. This makes the contribution of all later parts of the receiver to the system noise temperature negligible. System noise temperature is then given by Ts rain  TLNA  TA K

(4.32)

In Eq. (4.32), the LNA is assumed to be placed right at the feed horn so that there is no waveguide or coaxial run between the feed horn of the antenna and the LNA. We will assume that there are no feed losses. The increase in noise power, Nrain dB, caused by the increase in sky noise temperature is given by ¢Nrain  10 log10 c

kTs rain Bn Ts rain d  10 log10 c d dB kTsca Bn Tsca

where Tsca is the system noise temperature in clear sky conditions

(4.33)

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The received power is reduced by the attenuation caused by the rain in the slant path, so in rain the value of carrier power is Crain where Crain  Cca  Arain dB

(4.34)

The resulting (CN)dn rain value when rain intersects the downlink is given by 1CN2 dn rain  1CN2 dn ca  Arain  ¢Nrain dB

(4.35)

where (CN)dn ca is the downlink CN ratio in clear sky conditions. If a linear (bent pipe) transponder is used, the (CN)up ratio must be combined with (CN)dn rain to yield the overall (CN)0 ratio for the link. Many digital systems use regenerative transponders that provide constant output power regardless of uplink attenuation provided that the received CN ratio at the satellite is above the threshold of the onboard processing demodulator. In this case the value of (CN)dn rain will be used as the overall (CN)0 value in rain for the link. EXAMPLE 4.5.1 In the example of a DBS-TV system in Table 4.5, a link margin of 5.7 dB is available before the (CN)0 threshold of 8.6 dB is reached. We must calculate the increase in system noise temperature that results from 3-dB rain attenuation in the downlink path to determine the increase in noise power and thus the value of (CN)dn rain. This example shows how the link margin can be distributed between rain attenuation and an increase in receiver system noise power caused by an increase in sky noise. The clear sky attenuation is given in Table 4.5 as 0.4 dB. Thus total excess attenuation is 3.4 dB, and the sky noise temperature in rain will be, from Eq. (4.30) Tsky rain  270  11  103.410 2  147 K In clear sky conditions the sky noise temperature is Tca  [0.95  (270  (1  100.04))]  23 K. The sky temperature has increased from 23 K in clear sky conditions to 147 K when 3 dB rain attenuation occurs in the downlink. We must calculate the new system noise temperature when rain is present in the slant path, remembering that the antenna noise temperature has two parts: a fixed part due to noise from the ground entering the antenna’s sidelobes, and a variable part due to sky temperature. The LNA of the system in Table 4.5 has a noise temperature of 110 K and the clear sky system noise temperature is Tca  145 K. The antenna temperature in clear sky conditions is made up of a fixed portion of 12 K and a sky noise portion of 23 K. If we assume 95% coupling of sky noise temperature to antenna noise temperature, the system noise temperature, given by Eq. (4.32), increases to Ts rain  TLNA  TA  110  12  147  269 K The increase in receiver noise power referred to the receiver input is given by Eq. 4.33 ¢Nrain  10  log10 3Ts rainTs ca 4  10  log10 3 269145 4  2.7 dB From Eq. (4.35) 1CN2 dn rain  14.3  3.0  2.7  8.6 dB Thus the link margin required to withstand 3 dB of rain attenuation in the downlink path is 5.7 dB, and this will guarantee a link availability exceeding 99.7% of an average year for the majority of the DBS-TV system customers in the United States. In the south eastern United States, in states such as Florida and Louisiana where very heavy rain occurs more often than in other parts of the United States, link availability may be slightly less than 99.7%. Shaping of the satellite beam to direct more power to these parts of the United States helps to reduce the number of outages experienced in that region. Receivers located within the 1 dB contour of the satellite antenna beam have shorter path lengths giving 2.3 dB higher CN than the receiver used in the example shown in Table 4.5, so they have a link margin of 8 dB.

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The calculation of the availability of these receivers requires some care, because we cannot just add the extra 2.3 dB of link margin to the rain attenuation. Antenna noise increases with every decibel of extra rain attenuation, reducing the received power level, C, and increasing the system noise power N. An iterative (trial and error) procedure must be used to find the combination of reduction in C and increase in N that leads to an additional 2.3-dB degradation in the CN value. We will guess that increasing the rain attenuation from 3 to 4.5 dB leads to an increase in noise power of 0.8 dB. With 4.5-dB rain attenuation, the system noise temperature is Ts rain  110  12  0.95  270 11  0.3552  287 K The increase in noise power from the clear sky condition is ¢N  10 log10 12871452  3.0 dB Hence the decrease in CN for 4.5 dB of rain attenuation is 4.5  3.0  7.5 dB, and we are a little below our new margin of 8.0 dB. Increasing the rain attenuation by a further 0.4 dB to 4.9 dB and repeating the calculations gives a new system noise temperature in rain of 295 K. The corresponding increase in noise power is 3.1 dB giving a reduction in CN of 8.0 dB, equal to the link margin. A rain attenuation margin of 4.9 dB at Ku band would give an availability of 99.88% or better over the central region of the United States. This example demonstrates that the increase in noise temperature of a low noise DBS-TV Ku-band receiving system is a significant factor when rain attenuation is present in the downlink path. Rain attenuation alone cannot be equated to link margin. 

4.6

UPLINK DESIGN The uplink design is easier than the downlink in many cases, since an accurately specified carrier power must be presented at the satellite transponder and it is often feasible to use much higher power transmitters at earth stations than can be used on a satellite. However, VSAT systems use earth stations with small antennas and transmitter powers below 5 W, giving low uplink EIRP. Satellite telephone handsets are restricted to transmitting at power levels below 1 W because of the risk of EM radiation hazards. In mobile systems the uplink from the satellite telephone is usually the link with the lowest CN ratio. The cost of transmitters tends to be high compared with the cost of receiving equipment in satellite communication systems. The major growth in satellite communications has been in point-to-multipoint transmission, as in cable TV distribution and direct broadcast satellite television. One high-power transmit earth station provides service via a DBS satellite to many low-cost receive-only stations, and the high cost of the transmitting station is only a small part of the total network cost. The satellite transponder is a quasilinear amplifier and the received carrier level determines the output level. Where a traveling wave tube is used as the output high-power amplifier (HPA) in the transponder, as is often the case, and FDMA is employed, the HPA must be run with a predetermined backoff to avoid intermodulation products appearing at the output. The output backoff is typically 1 to 3 dB when more than one signal is present in the transponder, and is determined by the uplink carrier power level received at the spacecraft. Accurate control of the power transmitted by the earth station is therefore essential, which is easily achieved in a fixed network of earth stations. Where a very large number of earth stations access a single transponder using FDMA, such as in some VSAT networks and Intelsat satellites, transponder output backoff of 5 to 7 dB may be required to maintain intermodulation products at a sufficiently low level10. Even with a single access to the transponder (i.e., only one carrier present) some backoff is normally applied to avoid the PM-AM conversion that occurs when modulated signals are transmitted through nonlinear devices9.

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Earth station transmitter power is set by the power level required at the input to the transponder. This can be done in one of two ways. Either a specific flux density is required at the satellite, or a specific power level is required at the input to the transponder. Early Intelsat C-band satellites required high flux densities to saturate their transponders, in the range 73.7 to 67.5 dBW/m2, depending on the transponder gain setting8. This is a high flux density which requires a large earth station and a powerful transmitter generating up to 3 kW. Domestic GEO satellites operating into North America generally require lower flux densities allowing the use of smaller earth station antennas. At C band, a typical uplink earth station transmits 100 W with a 9-meter antenna, giving a flux density at the satellite of 100 W/m2. Although flux density at the satellite is a convenient way to determine earth station transmit EIRP requirements, analysis of the uplink requires calculation of the power level at the input to the transponder so that the uplink CN ratio can be found. The link equation is used to make this calculation, using either a specified transponder CN ratio or a required transponder output power level. When a CN ratio is specified for the transponder, the calculation of required transmit power is straightforward. Let (CN)up be the specified CN ratio in the transponder, measured in a noise bandwidth Bn Hz. The bandwidth Bn Hz is the bandwidth of the band-pass filter in the IF stage of the earth station receiver for which the uplink signal is intended. Even if Bn is much less than the transponder bandwidth, it is important that the uplink CN ratio be calculated in the bandwidth of the receiver, not the bandwidth of the transponder. The noise power referred to the transponder input is Nxp W where Nxp  k  Txp  Bn dBW

(4.36)

where Txp is the system noise temperature of the transponder in dBK and Bn is in units of dBHz. The power received at the input to the transponder is Prxpwhere Prxp  Pt  Gt  Gr  Lp  Lup dBW

(4.37)

where PtGt is the uplink earth station EIRP in dBW, Gr is the satellite antenna gain in dB in the direction of the uplink earth station and Lp is the path loss in dB. The factor Lup dB accounts for all uplink losses other than path loss. The value of (CN)up at the LNA input of the satellite receiver is given by CN  10 log10 3Pr 1k Ts Bn 2 4  Prxp  Nxp dB

(4.38)

The earth station transmitter output power Pt is calculated from Eq. (4.23) using the given value of CN in Eq. (4.38) and the noise power Nxp calculated from Eq. (4.36). Note that the received power at the transponder input is also given by Prxp  N  CN dBW

(4.39)

The earth station transmitter output power Pt can also be calculated from the output power of the transponder and transponder gain when these parameters are known and a bent pipe transponder is used. In general Prxp  Psat  BO0  Gxp dBW

(4.40)

where Psat is the saturated power output of the transponder in dBW, BO0 is the output backoff in dB, and Gxp is the gain of the transponder in dB. With small-diameter earth stations, a higher power earth station transmitter is required to achieve a similar satellite EIRP. This has the disadvantage that the interference level at adjacent satellites rises, since the small earth station antenna inevitably has a wider beam. Thus it is not always possible to trade off transmitter power against uplink antenna size.

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G(θ ) dB 40

G(θ ) = 29 − 25 log θ 20

G(θ ) = 32 − 25 log

FIGURE 4.10 ITU-R specifications on the sidelobe envelope of transmit antenna patterns for 2° GEO satellite spacing.

0 −20

−10

0

10

20

Angle off axis, θ (degrees)

There is a specification for transmit station antenna patterns, designed to minimize interference from adjacent uplinks. It is the uplink interference problem that determines satellite spacing and limits the capacity of the geostationary orbit in any frequency band. Figure 4.10 shows the ITU-R (formerly CCIR) specification, G()  (32  25 log10 ) dB, where  is in degrees, for  1 degree off axis, for satellites spaced by 3. To increase the capacity of the crowded geostationary orbit arc south of the United States, the FCC introduced new regulations in 1983 requiring better control of 6-GHz earth station antenna transmit patterns so that intersatellite spacing could be reduced to 2. The same specification has now been adopted by the ITU-R for the entire geostationary arc. The requirement is for the transmit antenna pattern to lie below G()  29  25 log10  dB in the range 1  7 from the antenna boresight and G()  32  25 log10  dB beyond 711. The required antenna pattern envelope is shown in Figure 4.10. At frequencies above 10 GHz, for example, 14.6 GHz and 30 GHz, propagation disturbances in the form of fading in rain cause the received power level at the satellite to fall. This lowers the uplink CN ratio in the transponder, which lowers the overall (CN)0 ratio in the earth station receiver when a linear (bent pipe) transponder is used on the satellite. Uplink power control (UPC) can be used to combat uplink rain attenuation. The transmitting earth station monitors a beacon signal from the satellite, and watches for reductions in power indicating rain fading on the downlink. Automatic monitoring and control of transmitted uplink power is used in 14-GHz uplink earth stations to maintain the uplink CN ratio in the satellite transponder during periods of rain attenuation. New generations of Ka-band satellites employ uplink power level detection at the satellite. A control link to each uplink earth station closes the loop. Since the downlink is always at a different frequency from the uplink, a downlink attenuation of A dB must be scaled to estimate uplink attenuation. The scaling factor used is typically ( fup fdown)a where a is typically between 2.0 and 2.4. For example, an uplink station transmitting at 14.0 GHz to a Ku-band satellite monitors the satellite beacon at 11.45 GHz. The uplink attenuation is therefore given by Aup  Adown  1 fup fdown 2 a

dB

(4.41)

where Aup is the estimated uplink rain attenuation and Adown is the measured downlink rain attenuation. For a value of a  2.2 and ( fup fdown)  1.222 the factor ( fup fdown)a is 1.56.

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127

Hence a downlink rain attenuation of 3 dB would give an estimated uplink attenuation of 4.7 dB. This uplink attenuation value applies only to rain and does not include gaseous attenuation or scintillation, which require different scaling ratios. Uplink power control cannot be applied until a certain amount of attenuation has built up in the link. This is typically around 2 dB for the downlink due to measurement inaccuracies, corresponding to about 3 dB for a Ka-band uplink. As rain begins to affect the link between the earth station and satellite, the uplink CN ratio in the transponder will fall until UPC starts to operate in the earth station transmitter. The transponder CN ratio will then remain relatively constant until the UPC system reaches the maximum available transmit power. Further attenuation on the uplink will cause the CN ratio in the transponder to fall. EXAMPLE 4.6.1 A transponder of a Ku-band satellite has a linear gain of 127 dB and a nominal output power at saturation of 5 W. The satellite’s 14-GHz receiving antenna has a gain of 26 dB on axis, and the beam covers western Europe. Calculate the power output of an uplink transmitter that gives an output power of 1 W from the satellite transponder at a frequency of 14.45 GHz when the earth station antenna has a gain of 50 dB and there is a 1.5-dB loss in the waveguide run between the transmitter and antenna. Assume that the atmosphere introduces a loss of 0.5 dB under clear sky conditions and that the earth station is located on the 2 dB contour of the satellite’s receiving antenna. If rain in the path causes attenuation of 7 dB for 0.01% of the year, what output power rating is required for the transmitter to guarantee that a 1-W output can be obtained from the satellite transponder for 99.99% of the year if uplink power control is used? The input power required by the transponder is simply the output power minus the transponder gain, so Pin  0 dBW  127 dB  127 dBW The uplink power budget is given by Eq. (4.11) Pr  EIRP  Gr  Lp  Lat  Lta  Lra dBW Rearranging and putting in the appropriate losses Pt  Pr  Gt  Gr  Lp  Lta  Lat  Lpt dBW where Lta is the waveguide loss, Lat is the atmospheric loss, and Lpt is the pointing loss (antenna pattern loss). Then assuming a path length of 38,500 km Pt  127.0  50  26  207.2  1.5  0.5  2.0 dBW That is Pt  7.2 dBW or 5.2 W If we provide an extra 7 dB of output power to compensate for fading on the path due to rain, the transmitter output power will be Pt rain  7.2  7  14.2 dBW or 26.3 W



4.7 DESIGN FOR SPECIFIED C/N: COMBINING C/N AND C/I VALUES IN SATELLITE LINKS The BER or SN ratio in the baseband channel of an earth station receiver is determined by the ratio of the carrier power to the noise power in the IF amplifier at the input to the demodulator. The noise present in the IF amplifier comes from many sources. So far in

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our analysis of uplinks and downlinks we have considered only the receiver thermal noise and noise radiated by atmospheric gases and rain in the slant path. When a complete satellite link is engineered, the noise in the earth station IF amplifier will have contributions from the receiver itself, the receiving antenna, sky noise, the satellite transponder from which it receives the signal, and adjacent satellites and terrestrial transmitters which share the same frequency band. In the first edition of this text, a method was presented for adding these noise contributions together in a hypothetical reference circuit. This was the standard technique used in CCIR (now ITU-R) Recommendations at the time the first edition of this book was written. More recently, addition of CN and CI (carrier-to-interference) ratios has been widely used, and is easier to apply than the reference circuit approach, although both methods will lead to the same result. The latter method using CN values is presented here. When more than one CN ratio is present in the link, we can add the individual CN ratios reciprocally to obtain an overall CN ratio, which we will denote here as (CN)0. The overall (CN)0 ratio is what would be measured in the earth station at the output of the IF amplifier 1C N2 0  1 31  1CN2 1  1  1CN2 2  1  1CN2 3  p 4

(4.42)

This is sometimes referred to as the reciprocal CN formula. The CN values must be linear ratios, NOT decibel values. Since the noise power in the individual CN ratios is referenced to the carrier power at that point, all the C values in Eq. (4.42) are the same. Expanding the formula by cross multiplying gives the overall (CN)0 as a power ratio, not in decibels 1C N2 0  1 1N1C  N2 C  N3 C  p 2  C 1N1  N2  N3  p 2

(4.43)

In decibel units: 1C N2 0  C dBW  10 log10 1N1  N2  N3  p W2 dB

(4.44)

Note that (CN)dn cannot be measured at the receiving earth station. The satellite always transmits noise as well as signal, so a CN ratio measurement at the receiver will always yield (CN)0, the combination of transponder and earth station CN ratios. To calculate the performance of a satellite link we must therefore determine the uplink (CN)up ratio in the transponder and the downlink (CN)dn in the earth station receiver. We must also consider whether there is any interference present, either in the satellite receiver or the earth station receiver. One case of importance is where the transponder is operated in a FDMA mode and intermodulation products (IM) are generated by the transponder’s nonlinear input–output characteristic. If the IM power level in the transponder is known, a CI value can be found and included in the calculation of (CN)0 ratio. Interference from adjacent satellites is likely whenever small receiving antennas are used, as with VSATs (very small aperture terminals) and DBS-TV receivers. Since CN values are usually calculated from power and noise budgets, their values are typically in decibels. There are some useful rules of thumb for estimating (CN)0 from two CN values: • If the CN values are equal, as in the example above, (CN)0 is 3 dB lower than either value. • If one CN value is 10 dB smaller than the other value, (CN)0 is 0.4 dB lower than the smaller of the CN values. • If one CN value is 20 dB or more greater than the other CN value, the overall (CN)0 is equal to the smaller of the two CN values within the accuracy of decibel calculations (0.1 dB).

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EXAMPLE 4.7.1 Thermal noise in an earth station receiver results in a (CN)dn ratio of 20.0 dB. A signal is received from a bent pipe transponder with a carrier to noise ratio (CN)up  20.0 dB. What is the value of overall (CN)0 at the earth station? If the transponder introduces intermodulation products with (CI) ratio  24 dB, what is the overall (CN)0 ratio at the receiving earth station? Using Eq. (4.42) and noting that (CN)  20.0 dB corresponds to a (CN) ratio of 100 1CN2 0  c

1 1 d  c d  50 or 17.0 dB 1 1CN2 up  1 1CN2 dn 0.01  0.01

The intermodulation (CI) value of 24.0 dB corresponds to a ratio of 0.004. The overall (CN)0 value is then 1CN2 0  c

1 d  41.7 or 16.2 dB 0.01  0.01  0.004



Overall (C/N)0 with Uplink and Downlink Attenuation Most satellite links are designed with link margins to allow for attenuation that may occur in the link or increases in noise power caused by interference. (Interference is almost always treated as though it were white noise, regardless of whether the interfering signal actually has a uniform spectral power distribution or Gaussian statistics. When the interference has known characteristics, such as a depolarized cochannel or a jamming signal, cancellation techniques can be used to reduce the level of interference.) The effect of a change in the uplink CN ratio has a different impact on overall (CN)0 depending on the operating mode and gain of the transponder. There are three different transponder types or operating modes: Linear transponder: Pout  Pin  Gxp Nonlinear transponder: Pout  Pin  Gxp  ¢G Regenerative transponder: Pout  constant

dBW dBW dBW

where Pin is the power delivered by the satellite’s receiving antenna to the input of the transponder, Pout is the power delivered by the transponder HPA to the input of the satellite’s transmitting antenna, Gxp is the gain of the transponder, and all parameters are in decibel units. The parameter G is dependent on Pin and accounts for the loss of gain caused by the nonlinear saturation characteristics of a transponder which is driven hard to obtain close to its maximum power output—the gain is effectively falling as the input power level increases. (See Chapter 6 for a detailed discussion of intermodulation effects and nonlinearity in transponders.) The maximum output power from a transponder is called the saturated output power and is the nominal transponder power output rating that is usually quoted. The transponder input–output characteristic is highly nonlinear when operated at this output power level. When a transponder is operated close to its saturated output power level, digital waveforms are changed, resulting in intersymbol interference (ISI), and FDMA operation results in the generation of intermodulation products by multiplication of the individual signals. Transponders are usually operated with output backoff, to make the characteristic more nearly linear. The exact amount of output backoff required in any given application depends on the specific characteristics of the transponder and the signals it carries. Typical values of output backoff are 1 dB for a single FM or PSK carrier to 3 dB for FDMA operation with several

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carriers. The corresponding input backoff values might be 3 dB and 5 dB, but the individual transponder characteristic must be known to make an accurate assessment. For convenience in this text, we will frequently assume linear transponder operation when calculating the overall (CN)0 ratio, even if this may not, in fact, be the case.

Uplink and Downlink Attenuation in Rain Rain attenuation affects uplinks and downlinks differently. We usually assume that rain attenuation is occurring on either the uplink or the downlink, but not on both at the same time. This is usually true for earth stations that are well separated geographically, but not if they are close together ( 20 km). Heavy rain occurs with a somewhat random geographic distribution for less than 1% of the time, so the probability of significant attenuation occurring on both the uplink and downlink simultaneously is small. In the following analysis of uplink and downlink attenuation effects, it will be assumed that one link is attenuated and the other is operating in clear air.

Uplink Attenuation and (C/N)up The transponder receiver noise temperature does not change significantly when rain is present in the uplink path to the satellite. The satellite receiving antenna beam is always sufficiently wide that it “sees” a large area of the (warm) earth’s surface and local noise temperature variations are insignificant. The noise temperature of the earth seen by a GEO satellite varies from a maximum of 270 K for a satellite antenna beam over Africa and northern Europe, to a minimum of 250 K over the Pacific Ocean. The corresponding system noise temperature for the transponders on a GEO satellite is in the range 400 to 500 K. There is effectively no increase in uplink noise power when heavy rain is present in the link between an earth station and a satellite because the satellite antenna beam sees the tops of cumulonimbus clouds above the rain, which are always colder than 270 K, instead of the earth’s surface. Rain attenuation on the uplink path to the satellite reduces the power at the satellite receiver input, and thus reduces (CN)up in direct proportion to the attenuation on the slant path. If the transponder is operating in a linear mode, the output power will be reduced by the same amount, which will cause (CN)dn to fall by an amount equal to the attenuation on the uplink. When both (CN)up and (CN)dn are reduced by Aup dB, the value of (CN)0 is reduced by exactly the same amount, Aup dB. Hence for the case of a linear transponder and rain attenuation in the uplink of Aup dB 1C N2 0 uplink rain  1CN2 0 clear air  Aup dB

Linear transponder

(4.45)

If the transponder is nonlinear, the reduction in input power caused by uplink attenuation of Aup dB results in a smaller reduction in output power, by an amount G.

1C N2 0 uplink rain  1C N2 0 clear air  Aup  ¢G dB Nonlinear transponder (4.46)

If the transponder is digital and regenerative, or incorporates an Automatic Gain Control (AGC) system to maintain a constant output power level 1CN2 0 uplink rain  1CN2 0 clear air dB

Regenerative transponder or AGC

(4.47)

The above equation will hold only if the received signal is above threshold and the BER of the recovered digital signal in the transponder is small. If the signal falls below threshold, the uplink will contribute significantly to the BER of the digital signal at the receiving earth station.

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Downlink Attenuation and (C/N)dn The earth station receiver noise temperature can change very significantly when rain is present in the downlink path from the satellite. The sky noise temperature can increase to close to the physical temperature of the individual raindrops, particularly in very heavy rain. A reasonable temperature to assume for temperate latitudes in a variety of rainfall rates is 270 K, although values above 290 K have been observed in the tropics. An increase in sky noise temperature to 270 K will increase the receiving antenna temperature markedly above its clear air value. See Section 4.5 for an illustration of this effect. The result is that the received power level, C, is reduced and the noise power, N, in the receiver increases. The result for downlink CN is given by Eq. (4.48) 1CN2 dn rain  1CN2 dn clear air  Arain  ¢Nrain dB

(4.48)

The overall CN is then given by 1CN2 0  1  31 1CN2 dn rain  1  1C N2 up 4 dB

(4.49)

As noted earlier, unless we are making a loop-back test, we will assume that the value of (CN)up is for clear air, and remains constant regardless of the attenuation on the downlink.

System Design for Specific Performance A typical two-way satellite communication link consists of four separate paths: an outbound uplink path from one terminal to the satellite and an outbound downlink to the second terminal; and an inbound uplink from the second terminal to the satellite and an inbound downlink to the first terminal. The links in the two directions are independent and can be designed separately, unless they share a single transponder using FDMA. A broadcast link, like the DBS-TV system described earlier in this chapter, is a one-way system, with just one uplink and one downlink.

Satellite Communication Link Design Procedure The design procedure for a one-way satellite communication link can be summarized by the following 10 steps. The return link design follows the same procedure. 1. Determine the frequency band in which the system must operate. Comparative designs may be required to help make the selection. 2. Determine the communications parameters of the satellite. Estimate any values that are not known. 3. Determine the parameters of the transmitting and receiving earth stations. 4. Start at the transmitting earth station. Establish an uplink budget and a transponder noise power budget to find (CN)up in the transponder. 5. Find the output power of the transponder based on transponder gain or output backoff. 6. Establish a downlink power and noise budget for the receiving earth station. Calculate (CN)dn and (CN)0 for a station at the edge of the coverage zone (worst case). 7. Calculate SN or BER in the baseband channel. Find the link margins.

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8. Evaluate the result and compare with the specification requirements. Change parameters of the system as required to obtain acceptable (CN)0 or SN or BER values. This may require several trial designs. 9. Determine the propagation conditions under which the link must operate. Calculate outage times for the uplinks and downlinks. 10. Redesign the system by changing some parameters if the link margins are inadequate. Check that all parameters are reasonable, and that the design can be implemented within the expected budget.

4.8

SYSTEM DESIGN EXAMPLES The following sample system designs demonstrate how the ideas developed in this chapter can be applied to the design of satellite communication systems.

TABLE 4.6

System and Satellite Specification

Ku-band satellite parameters Geostationary at 73° W longitude, 28 Ku-band transponders Total RF output power Antenna gain, on axis (transmit and receive) Receive system noise temperature Transponder saturated output power: Ku band Transponder bandwidth: Ku band Signal Compressed digital video signals with transmitted symbol rate of 43.2 Msps Minimum permitted overall (CN)0 in receiver Transmitting Ku-band earth station Antenna diameter Aperture efficiency Uplink frequency Required CN in Ku-band transponder Transponder HPA output backoff Miscellaneous uplink losses Location: 2 dB contour of satellite receiving antenna Receiving Ku-band earth station Downlink frequency Receiver IF noise bandwidth Antenna noise temperature LNA noise temperature Required overall (CN)0 in clear air Miscellaneous downlink losses Location: 3 dB contour of satellite transmitting antenna Rain attenuation and propagation factors Ku-band clear air attenuation Uplink 14.15 GHz Downlink 11.45 GHz Rain attenuation Uplink Downlink

0.01% of year 0.01% of year

2.24 31 500 80 54

kW dB K W MHz

9.5 dB 5m 68% 14.15 GHz 30 dB 1 dB 0.3 dB

11.45 43.2 30 110 17 0.2

GHz MHz K K dB dB

0.7 dB 0.5 dB 6.0 dB 5.0 dB

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System Design Example 4.8.1 This example examines the design of a satellite communication link using a Ku-band geostationary satellite with bent pipe transponders to distribute digital TV signals from an earth station to many receiving stations throughout the United States. The design requires that an overall CN ratio of 9.5 dB be met in the TV receiver to ensure that the video signal on the TV screen is held to an acceptable level. The uplink transmitter power and the receiving antenna gain and diameter are determined for each system. The available link margins for each of the systems are found and the performance of the systems is analyzed when rain attenuation occurs in the satellite–earth paths. The advantages and disadvantages of implementing uplink power control are considered. In this example, the satellite is located at 73 W. However, for international registration of this satellite location, the location would be denoted as 287 E. The link budgets developed in the examples below use decibel notation throughout. The satellite and earth stations are specified in Table 4.6, and Figure 4.11 shows an illustration of the satellite television distribution system.

Ku-Band Uplink Design We must find the uplink transmitter power required to achieve (CN)up  30 dB in clear air atmospheric conditions. We will first find the noise power in the transponder for 43.2 MHz bandwidth, and then add 30 dB to find the transponder input power level. Uplink Noise Power Budget k Ts B N

   

Boltzmann’s constant 500 K 43.2 MHz transponder noise power

228.6 27.0 76.4 125.2

dBW/K/Hz dBK dBHz dBW

The received power level at the transponder input must be 30 dB greater than the noise power. Pr  power at transponder input  95.2 dBW

Ku-band satellite

Uplink Earth station

Ku-band uplink Ku-band downlinks

Program material from studios To cable TV network Cable TV receive stations FIGURE 4.11 Satellite television distribution system.

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The uplink antenna has a diameter of 5 m and an aperture efficiency of 68%. At 14.15 GHz the wavelength is 2.120 cm  0.0212 m. The antenna gain is Gt  10 log 30.68  1pDl2 2 4  55.7 dB

The free space path loss is Lp  10 log [(4R)2]  207.2 dB Uplink Power Budget Pt Gt Gr Lp Lant Lm Pr

      

Earth station transmitter power Earth station antenna gain Satellite antenna gain Free space path loss ES on 2 dB contour Other losses Received power at transponder

Pt 55.7 31.0 207.2 2.0 1.0 Pt  123.5

dBW dB dB dB dB dB dB

The required power at the transponder input to meet the (CN)up  30 dB objective is 95.2 dBW. Hence Pt  123.5 dB  95.2 dBW Pt  28.3 dBW or 675 W This is a relatively high transmit power so we would probably want to increase the transmitting antenna diameter to increase its gain, allowing a reduction in transmit power.

Ku-Band Downlink Design The first step is to calculate the downlink (CN)dn that will provide (CN)0  17 dB when (CN)up  30 dB. From Eq. (4.43) 1 1C N2 dn  1  1C N2 0  1  1CN2 up

Thus

1not in dB2

1  1CN2 dn  1 50  1 1000  0.019 1C N2 dn  52.6 1 17.2 dB

We must find the required receiver input power to give (CN)dn  17.2 dB and then find the receiving antenna gain, Gr. Downlink Noise Power Budget k Ts Bn N

   

Boltzmann’s constant 30  110 K  140 K 43.2 MHz transponder noise power

228.6 dBW/K/Hz 21.5 dBK 76.4 dBHz 130.7 dBW

The power level at the earth station receiver input must be 17.2 dB greater than the noise power in clear air. Pr  power at earth station receiver input  130.7 dBW  17.2 dB  113.5 dBW We need to calculate the path loss at 11.45 GHz. At 14.15 GHz path loss was 207.2 dB. At 11.45 GHz path loss is Lp  207.2  20 log10 114.1511.452  205.4 dB

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The transponder is operated with 1 dB output backoff, so the output power is 1 dB below 80 W (80 W 1 19.0 dBW) Pt  19 dBW  1 dB  18 dBW Downlink Power Budget Pt Gt Gr Lp La Lm Pr

      

Satellite transponder output power Satellite antenna gain Earth station antenna gain Free space path loss ES on 3 dB contour of satellite antenna Other losses Received power at transponder

18.0 31.0 Gr 205.4 3.0 0.8 Gr  160.2

dBW dB dB dB dB dB dB

The required power into the earth station receiver to meet the (CN)dn  17.2 dB objective is Pr  120.1 dBW. Hence the receiving antenna must have a gain Gr where Gr  160.2 dB  113.5 dBW Gr  46.7 dB or 46,774 as a ratio The earth station antenna diameter, D, is calculated from the formula for antenna gain, G, with a circular aperture Gr  0.65  1pDl2 2  46,774

At 11.45 GHz, the wavelength is 2.62 cm  0.0262 m. Evaluating the above equation to find D gives the required receiving antenna diameter as D  2.14 m.

Rain Effects at Ku Band Uplink Under conditions of heavy rain, the Ku-band path to the satellite station suffers an attenuation of 6 dB for 0.01% of the year. We must find the uplink attenuation margin and decide whether uplink power control would improve system performance at Ku band. The uplink CN was 30 dB in clear air. With 6 dB uplink path attenuation, the CN in the transponder falls to 24 dB, and assuming a linear transponder characteristic and no uplink power control, the transponder output power falls to 18  8  12 dBW. The downlink CN falls by 6 dB from 17.2 dB to 11.2 dB, and the overall (CN)0 falls by 6 dB to 11 dB. With the minimum overall CN set at 9.5 dB, the additional margin for uplink attenuation is 1.5 dB. Hence the link margin available on the uplink is 7.5 dB without uplink power control. This is an adequate uplink rain attenuation margin for many parts of the United States, and would typically lead to rain outages of less than 1 h total time per year. Uplink power control (UPC) could be implemented so that the earth station transmitter output power is increased when the uplink attenuation is estimated to have reached 3 dB. This would hold the value of overall (CN)0 in the receiver at 14 dB. If the UPC system has a dynamic range of 6 dB, the uplink rain attenuation margin is increased to 12 dB and the maximum Ku-band transmitter power is increased to 34.3 dBW (2690 W). Rain attenuation can exceed 12 dB at 14 GHz for a few minutes at a time in very heavy thunderstorms, but there would only be a handful of such occurrences in an average year. UPC definitely improves the ability of the uplink to resist rain attenuation, but at the expense of a considerably more powerful, and expensive, uplink transmitter. The extra expense can be justified in a television distribution system with many receiving stations. There is also an increased risk that the additional power radiated by the uplink station

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when UPC is active will cause interference at an unacceptable level into other satellite links using the same frequencies. It would be advisable to increase the earth station antenna diameter to increase its gain, and thus reduce the maximum transmit power required. Downlink Attenuation and Sky Noise Increase The 11.45-GHz path between the satellite and the receive station suffers rain attenuation exceeding 5 dB for 0.01% of the year. Assuming 100% coupling of sky noise into antenna noise, and 0.5-dB clear air gaseous attenuation, calculate the overall CN under these conditions. Assume that the uplink station is operating in clear air. We must calculate the available downlink fade margin. We need to find the sky noise temperature that results from a total excess path attenuation of 5.5 dB (clear air attenuation plus rain attenuation); this is the new antenna temperature in rain, because we assumed 100% coupling between sky noise temperature and antenna temperature. We must evaluate the change in received power and increase in system noise temperature in order to calculate the change in CN ratio for the downlink. In clear air, the atmospheric attenuation on the downlink is 0.5 dB. The corresponding sky noise temperature is 270(1  100.05)  29 K, which leads to the antenna temperature of 30 K given in the Ku-band system specification. When the rain causes 5-dB attenuation, the total path attenuation from the atmosphere and the rain is 5.5 dB. The corresponding sky noise temperature is given by Tsky rain  T0 11  G2 where G  10A10  0.282 Tsky rain  270 11  0.2822  194 K Thus the antenna temperature has increased from 30 K in clear air to 194 K in rain. The system noise temperature in rain, Ts rain, is increased from the clear air value of 140 K (30 K sky noise temperature plus 110 K LNA temperature) Ts rain  194  110  304 K or 24.8 dBK The increase in noise power is ¢N  10 log 1304 1402  3.4 dB The signal is attenuated by 5 dB in the rain, so the total reduction in downlink CN ratio is 8.4 dB, which yields a new value 1C N2 dn rain  17.2  8.4  8.8 dB The overall CN is then found by combining the clear air uplink (CN)up of 30 dB with the rain faded downlink (CN)dn rain of 8.8 dB, giving 1CN2 0 rain  8.8 dB

The overall (CN)0 is below the minimum acceptable value of 9.5 dB. The downlink link margin is Downlink fade margin  1CN2 dn  1C N2 min  17.2  9.5  7.7 dB Since downlink rain attenuation of 5 dB causes the overall (CN)0 to go below the minimum permitted value of 9.5 dB, we should recalculate the maximum attenuation that the downlink can sustain. This involves an iterative process, since changing the attenuation changes both C and N values in (CN)dn. At an attenuation level of 5 dB, the increase in noise power is 3.4 dB, so a starting guess would be that decreasing the attenuation by 0.3 dB will decrease the noise power by 0.2 dB. The rain attenuation will then be a little less than 5 dB.

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Recalculating (CN)dn for a rain attenuation value of 4.7 dB gives

Tsky rain  T0 11  G2 where G  10A10  0.339 Tsky rain  27011  0.3392  178 K ¢N  10 log 12881402  3.1 dB 1C N2 dn rain  17.2  4.7  3.1  9.4 dB 1CN2 0 rain  9.36  9.4 dB

The result is close enough to the required value of (CN)0 min  9.5 dB to conclude that we can tolerate about 4.7 dB of rain attenuation on the downlink. If better availability is required—less outage time—the diameter of the receiving antenna can be increased. For example, if the receiving antenna diameter is increased to 2.4 m, (about 8 ft) the increase in antenna gain is 20 log10(2.402.14)  1.0 dB, which increases the downlink margin to 8.7 dB. Repeating the iterative calculation outlined above, the corresponding rain attenuation on the downlink is 5.5 dB with a noise power increase of 3.2 dB. The downlink CN with 5.5-dB rain attenuation is 17.2  8.7  9.5 dB, and the overall (CN)0  9.5 dB. The extra antenna gain now ensures that the link meets the required specification, which will keep outages to a total of about 50 min in an average year in the eastern United States. However, an increase in antenna diameter will reduce the beamwidth of the antenna and may require an upgrade in the tracking requirements. With a fixed pointing antenna, diurnal motion of the satellite may cause a variation in received signal strength as the satellite moves through the antenna beam.

Summary of Ku-Band Link Performance The Ku-band link with a 2.4 m earth station antenna will suffer rain outages because attenuation exceeding 5.5 dB will occur occasionally on the downlinks, affecting individual customers. Outages will rarely occur on the uplink. With uplink power control (UPC) and a more powerful transmitter, uplink outages can be restricted to a few minutes per year. A 2.4-m receiving antenna is needed to ensure that the Ku-band downlink will be out for no more than 0.01% of an average year with the given attenuation statistics. The threshold value for overall CN was set at 9.5 dB because we can use QPSK and half rate error correction coding to obtain an equivalent (CN) ratio of about 15.5 dB without coding. Allowing a 1 dB implementation margin (see Chapter 5), the BER on the downlink will remain below 107 except when very heavy rain affects the downlink. In clear sky conditions there will be no errors on the link. The 43.2 Msps QPSK signal with half rate FEC can deliver a data rate of 43.2 Mbps, which can support seven MPEG-2 video channels. The video distribution system described here is designed to deliver multiple video channels to cable TV stations with low risk of outages. Direct broadcast satellite television delivers video signals directly to the customer’s location using a much smaller 0.5-m receiving antenna. The smaller antenna can be used because the DBS-TV satellites transmit at a higher power level (160 W), the symbol rate is lower (20 Mbps) and availability of the signal at the receiving antenna is guaranteed for only 99.7% of the year.

System Design Example 4.8.2 Personal Communication System Using Low Earth Orbit Satellites Low earth orbit (LEO) satellite systems are designed to provide personal communication service similar to a cellular telephone, but over a much wider area. LEO satellite systems

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SIDEBAR Low earth orbit satellite communication systems use a large number of satellites in multiple orbital planes at altitudes between 700 and 1400 km. The satellites can communicate with a limited portion of the earth’s surface because of the low orbit altitude, and appear to an observer on the earth to fly across the sky in a few minutes. Communication is maintained between the user and a gateway station by switching the channels from one satellite to the

next as one satellite goes below the horizon and another comes into view. The satellites are all identical, so the link design is based on a single link between one user and the gateway station via one satellite. Examples of such LEO satellite systems are Iridium and Globalstar. An example of a medium earth orbit (MEO) satellite system is ICO Global, which uses a smaller number of satellites at an altitude of about 10,000 km.

can cover sparsely populated regions of a country, or the world, where there are no terrestrial cellular telephone systems. The user has a handset similar to a cellular telephone handset that provides two-way voice communications through a gateway station, usually to a conventional telephone in a home or office connected to the public switched telephone network (PTSN). Satellite telephones can equally well connect to another satellite handset, or to a terrestrial cellular telephone. Most LEO satellite systems operate in L band, in the 1500- and 1600-MHz bands, and in the lower part of S band around 2460 MHz, frequency bands that are allocated for mobile satellite communications. Some LEO systems use intersatellite links so a user can connect to any point in the world without an intermediate return to earth. However, the signals invariably pass through a gateway station at each end of the link to facilitate control of the call and to ensure that users can be charged for using the system. Connections between the gateway earth stations and the satellites use S-band, C-band, Ku-band, or Ka-band frequencies, depending on the system requirements. Only a small portion of the radio spectrum at L band is allocated to LEO and MEO satellite systems, so L-band frequencies are reserved for the critical links between the user and the satellite. A handoff process is required for LEO satellites similar to that used in cellular telephone networks, but the handoff between satellites should not be apparent to the user. Most LEO satellites have multiple beam antennas, and the beam pattern moves across the earth’s surface at the speed of the satellite—typically about 7.7 km/s or 17,200 mph. A single beam is typically 500 km in diameter, so an individual user is in any one beam for less than a minute. The system provides automatic switching from beam to beam within the same satellite antenna coverage, much like a cellular telephone system switches users from cell to cell, which nearly always requires a change in the link frequencies, but as with satellite to satellite handoffs, the process must be transparent to the user. The example below analyzes the links between a user and a gateway station. LEO satellite systems employ digital transmission so that advantage can be taken of forward error correction coding (FEC) and speech compression techniques. The bit rate of digital voice in an LEO satellite link is typically 4800 bps, requiring powerful compression algorithms. The low bit rate allows more signals to be sent in the available transponder bandwidth and also helps maintain the CN ratio in the receivers. When FEC is applied to a digital bit stream, carrier to noise ratios down to 5 dB can be used. The low bit rate and operation of the receivers at low CN ratios are essential to make personal communication via an LEO satellite possible. The link between the gateway station and the mobile terminal is defined as the outbound link, and the link from the mobile terminal to the gateway is the inbound link. Note that there are four satellite paths, just as in all other two-way satellite communication systems:

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LEO satellite

Ku-band uplink and downlink

Gateway Earth station To PSTN

Multiple L-band uplink and downlink beams

Users Satellite telephone

FIGURE 4.12 Two-way personal communication system using L-band LEO satellite links.

outbound uplink, outbound downlink, inbound uplink, inbound downlink. Each has its own unique frequency, and in most LEO satellite systems, one of the links will be weaker than the other three links and will thus limit the system performance. One objective in the example that follows is to identify the weakest path and to then attempt to improve that part of the system. Figure 4.12 illustrates the two-way link between the gateway station and the handset. Note that separate transponders are used for the inbound and outbound paths. In this example, the mobile terminals transmit to a transponder on the satellite using frequency division multiple access (FDMA) and single channel per carrier (SCPC) techniques. FDMA and SPSC are discussed in Chapter 6. However, the principle is simple: each transmitter is allocated its own frequency, just like broadcast stations. The available frequencies are shared among active users on demand, as in a cellular telephone system, so a call begins with a start-up sequence that establishes communication between the mobile terminal and the local gateway station via the nearest LEO satellite. The gateway station then allocates frequencies for the call. At the end of the call, the frequencies are released and become available for another user. This is called demand assignment (DA), and the multiple access technique is identified by the acronym SCPC-FDMA-DA, or, alternatively, SCPC-FDMA-DAMA where DAMA stands for demand assignment multiple access. A common set of control channels at preassigned frequencies enables call setup and teardown. The link from the gateway station via the satellite to the mobile terminal uses time division multiplexing (TDM). A TDM signal consists of a sequence of packets with addresses that repeat every 20 to 100 ms. The addresses identify which terminal should receive each packet. The TDM bit stream rate must exceed the total bit rate of all active terminals in a two-way telephone system so that there is sufficient capacity available for each terminal within the TDM bit stream. In this example we begin by assuming that 50 active users share one common TDM channel. We will also assume that the gateway earth station operates at Ku band to and from the satellite, and that the satellite employs a linear transponder (bent pipe) rather than having onboard processing. The parameters of the satellite transponder, the mobile

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TABLE 4.7 LEO Satellite Personal Communication System Parameters Satellite parameters Saturated output power Transponder bandwidth Uplink frequency for mobile terminal Downlink frequency for mobile terminal Antenna gain 1650 MHz uplink (one beam) Antenna gain 1550 MHz downlink (one beam) Uplink frequency for gateway station Downlink frequency for gateway station Antenna gain 14 GHz uplink Antenna gain 11.5 GHz downlink Satellite receiver system noise temperature Maximum range to edge of coverage zone Mobile terminal parameters Transmitter output power Antenna gain (transmit and receive) Receiver system noise temperature Transmit bit rate Receive bit rate Required maximum bit error rate

0.5 0 300 4800 96 104

W dB K bps kbps

Gateway station parameters Transmitter output power (maximum per transponder) Antenna gain (transmit, 14.0 GHz) Antenna gain (receive, 11.5 GHz) Receive system noise temperature (clear air) Transmit bit rate (before FEC encoder) Receive bit rate (after FEC decoder) Required maximum bit error rate

10 55 53.5 140 300 4800 104

W dB dB K kbps bps

10 W 1 MHz 1650 MHz 1550 MHz 23 dB 23 dB 14 GHz 11.5 GHz 3 dB 3 dB 500 K 2200 km

terminal, and the gateway station are given in Table 4.7. The table gives the maximum path length for any satellite–earth link. It is left as an exercise for the reader to determine a suitable combination of orbital altitude and minimum elevation angle for the LEO system. The user’s transmitter and receiver is called a mobile terminal in this example. It could be a handheld device like a cellular telephone, sometimes called a satellite telephone or handset, or the terminal could be mounted in a vehicle. The satellite has multiple L band beams serving different parts of its instantaneous coverage zone because a single beam from an LEO satellite serving different parts of its instantaneous coverage zone would have both low gain and limited capacity. For an antenna with a gain of 23 dB, G  200 and the beamwidth is 3 dB where u3 dB  133,000 2002 12  12.8°

The use of a multiple beam antenna on the satellite increases the antenna gain toward the mobile terminal, which increases the CN ratio of the signals in the mobile terminal and gateway station receivers. The Ku-band antennas that link the satellite to the gateway station have broad beams and low gain. The CN on these links is high through the use of a relatively large antenna and a high transmitter power at the gateway earth

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station, allowing the use of small and simple Ku-band antennas on the satellite. Figures 10.14, 10.15, and 10.16 in Chapter 10 illustrate satellite systems of this type. The antenna gain at the mobile terminal is low, with a value of 0 dB used for calculation, because the antenna coverage of the terminal must be very broad. If the terminal is a satellite telephone, an omnidirectional antenna allows the user to move around freely. If the mobile terminal antenna gain were to be increased, its beam would be correspondingly narrower, and the user would have to point the handset at the satellite. In an LEO satellite system, the user does not know which satellite is being used nor where it is in the sky, so requiring the user to point the handset antenna at the satellite is not a feasible option. When the mobile terminal is mounted in a vehicle with the antenna on the roof, pointing the antenna at the satellite is not possible unless a sophisticated (and expensive) steered antenna is used. In this example, we will begin by assuming that there are 50 users sharing a single transponder on the satellite, and that one transponder serves one of the L-band beams within the LEO satellite coverage, operating within a given set of frequencies. A large number of users can share an LEO satellite through the provision of many transponders, each of which is connected to one of the individual beams in the multiple L-band antenna coverage of the satellite. The signal received by a mobile terminal from the gateway is a TDM sequence of packets carrying 50 digital voice channels, each at 4800 bps. The bit rate of the TDM signal would be 240 kbps if it carried only the voice signals, but will be higher in practice because additional bits must be sent with each packet; a TDM bit rate of 300 kbps is used in this example. Individual mobile terminals pull off their assigned packets from within the TDM stream and ignore the rest. Initially, the links will be analyzed without forward error correction. All digital links are designed with ideal (Nyquist) filters which have noise bandwidth, Bn Hz, equal to the symbol rate of the digital signal in symbols per second. In this example, handheld transceivers send and receive binary phase shift keyed (BPSK) modulation. The maximum permitted bit error rate of the digital signal of 104 leads to a SN ratio in the speech channel of 34 dB. (SN  14Pe, where Pe is the BER—see Chapter 5.)

Inbound Link: Mobile Terminal to Gateway Station Each terminal transmits a BPSK signal at 4800 bps at an allocated frequency. The satellite transponder shifts all received L-band signals in frequency before retransmission at Ku band to the gateway station, and also amplifies the signals with a linear transponder. At the gateway station, the antenna and RF receiver are connected to many identical IF receivers tuned to the individual frequencies of the handheld transmitters. Each IF receiver has a noise bandwidth of 4800 Hz, set by a square root raised cosine filter with   0.5, giving an occupied channel bandwidth of 7.2 kHz (see Chapter 5 for details of the design of digital links). At the receiving end of the link, the CN at the input to the BPSK demodulator must be high enough to provide an acceptable bit error rate. Here, we require a maximum BER of 104 which provides a minimum SN of 34 dB in the speech channel. In Chapter 5 it is shown that the theoretical CN required to achieve a bit error rate of 104 with BPSK modulation is 8.4 dB. In a practical digital communication system, we always need a higher CN than theory suggests because we do not have ideal Nyquist filters, and other parts of the system are also not ideal, so we must add an implementation margin to account for the nonideal nature of the system. In this example, the implementation margin is set at 0.6 dB; we need a minimum CN  9.0 dB to meet the BER and SN specifications. We can now design the satellite link to achieve the minimum CN.

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Mobile Terminal to Satellite Link We will establish power and noise link budgets for each of the four paths, beginning with the uplink from the mobile terminal to the satellite. The received power at the output of the uplink antenna on the satellite from Eq. (4.11) is Pr  EIRP  Gr  Lp  Lm dBW where EIRP is the product of transmitter output power and transmitting antenna gain, PtGt in dBW, Gr is the satellite receive antenna gain, Lp is the path loss of the link, and Lm accounts for all other losses. The noise power, N, at the input to the satellite receiving system from Eq. (4.13) is N  Pn  kTsBn  k  Ts  Bn Path loss Lp is found from Eq. (4.12)

Lp  34pRl4 2

watts dBW

or 20 log10 14pR/l2 dB

where R is the distance in meters between the transmitting and receiving antennas in the link and  is the wavelength in meters. The uplink frequency is 1650 MHz, giving   0.1818 m. The maximum range is 2200 km so maximum path loss is Lp  20 log10 14p  2.2  106 0.18182  163.6 dB We will assume that there are miscellaneous losses in the 1550 MHz link of 0.5 dB, caused by polarization misalignments, gaseous absorption in the atmosphere, etc. The calculation of the CN ratio is made for the worst case of an earth station located on the 3 dB contour of the satellite antenna beam, so a 3 dB reduction in satellite antenna gain is applied, making the value of Lm  3.5 dB. We can now set out the link power and noise budgets for clear line of sight conditions, when there is no attenuation caused by obstructions in the path. Uplink Power Budget Parameter EIRP of handheld unit Gain of receiving antenna Path loss at 1650 MHz Miscellaneous losses Received power at satellite

Symbol PtGt Gr Lp Lm Pr

Value Units 3 23 163.6 3.5 147.1

dBW dB dB dB dBW

Transponder Noise Power Budget Parameter Boltzmann’s constant System noise temperature Noise bandwidth Noise power

Symbol k Ts Bn N

Value Units 228.6 27.0 36.8 164.8

dBW/K/Hz dBK dBHz dBW

The inbound uplink CN ratio in the transponder can now be calculated from the power and noise budgets: 1C N2 up  Pr N  147.1 dBW  1164.8 dBW2  17.7 dB

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Note that this is the lowest CN ratio that should occur in the transponder in clear air conditions, since the calculation was made for a mobile terminal at the longest range from the satellite and at the edge of a satellite antenna beam. The mobile terminal antenna gain has also been set to its minimum value of 0 dB. If the satellite were directly overhead the range would be 1000 km instead of 2200 km, making the path loss lower by 6.8 dB, and the miscellaneous losses would be 3 dB lower at the center of the satellite antenna beam, making the power received at the transponder 10.8 dB greater, and then (CN)up  28.5 dB. However, we cannot use this figure for the system design, otherwise there would be only one user who could make calls, and then only for a brief moment as the satellite passes directly overhead. We must ensure that all users within the satellite’s coverage zone have adequate CN ratios in their links for successful communication.

Satellite to Gateway Station Link The next step in calculating the CN ratio for the inbound link is to calculate (CN)dn in the gateway receiver. We are operating the transponder in FDMA, so the individual mobile terminal signals must share the output power of the transponder. We will assume that 50 active terminal signals share the 1 MHz transponder bandwidth and that 3 dB backoff is used at the transponder output to obtain quasi-linear operation of the transponder HPA (remembering that we have assumed linear transponder operation in this example). The transponder output power is therefore 10 dBW  3 dB  7 dBW (5 W). The 5-W transponder output power must be shared equally between the 50 signals in the transponder, giving 0.1 W  10 dBW per signal at the transponder output for the downlink to the gateway station. We can now establish a link budget for a single channel downlink from the satellite to the gateway station. We will use the same worst-case conditions as for the uplink— maximum path length and minimum satellite antenna gain, with miscellaneous losses of 3.5 dB, including the edge of satellite beam effect. Downlink Power Budget Parameter EIRP per channel Gain of receiving antenna Path loss at 11.5 GHz Miscellaneous losses Received power at satellite

Symbol PtGt Gr Lp Lm Pr

Value Units 10.0 53.5 180.5 3.5 140.5

dBW dB dB dB dBW

Gateway Station Noise Power Budget Parameter Boltzmann’s constant System noise temperature Noise bandwidth Noise power

Symbol k Ts Bn N

Value Units 228.6 21.5 36.8 170.3

dBW/K/Hz dBK dBHz dBW

The CN ratio in the 4.8-kHz noise bandwidth of a gateway station IF receiver is given by: 1CN2 dn  Pr N  140.5  1170.32  29.8 dB

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(CN)dn for the inbound downlink is higher than (CN)up for the inbound uplink because of the high gain of the gateway station antenna. Because the gain of the antenna is high, 53.5 dB, which corresponds to an antenna diameter of 5 m and an aperture efficiency of 60%, its beamwidth is narrow, about 0.4, and the gateway station must track the satellite as it crosses the sky. The overall (CN)0 at the gateway is calculated by combining the uplink CN and downlink CN values using Eq. (4.43), since both the transponder and the gateway station receiver add noise to the signal. The values used in the formula are ratios, that is, CN values are not in decibels. 1 1CN2 0  1  1C N2 up  1  1CN2 dn

For the inbound uplink, (CN)up  17.7 dB 1 58.9 as a ratio. For the inbound downlink, (CN)dn  29.8 dB 1 955.0 as a ratio. 1CN2 0  1 11 58.9  1955.02  55.5

or 17.4 dB

The overall CN ratio of 17.4 dB at the gateway station receiver guarantees that with BPSK and a bit rate of 4800 bps there will be extremely few bit errors and the SN of the speech channel will be set by quantization noise in the analog to digital converters. The maximum permitted BER is 104, which occurs with (CN)0  9.0 dB. We therefore have an inbound link margin of (17.4  9.0)  8.4 dB. However, we must calculate the individual link margins for the uplink and downlink in order to be able to use the margins for fading analysis. This will be done at the end of the example.

Outbound Link The outbound link from the gateway station to the mobile terminal sends a continuous 300 kbps TDM bit stream using BPSK modulation and a separate transponder with 1 MHz bandwidth. The bit stream is a series of packets addressed to all 50 active terminals. The noise bandwidth of the terminal receiver is 300 kHz, assuming ideal Nyquist filters. The outbound uplink and downlink CN values are calculated in exactly the same way as for the inbound link, and the power and noise budgets are combined to give CN ratios directly from a single table. At the uplink frequency of 14 GHz, clear air atmospheric attenuation of 1.0 dB is included in the miscellaneous losses, together with the usual 3 dB loss for the user at the edge of the satellite antenna beam. Uplink CN Budget Parameter Gateway station EIRP Gain of receiving antenna Path loss at 14.0 GHz Miscellaneous losses Received power at satellite Boltzmann’s constant System noise temperature Noise bandwidth Noise power Uplink CN

Symbol PtGt Gr Lp Lm Pr k Ts Bn N (CN)up 

Value Units 65.0 dBW 3.0 dB 182.2 dB 4.0 dB 118.2 dBW 228.6 dBW/K/Hz 27.0 dBK 54.8 dBHz 146.8 dBW 118.2  (146.8)  28.6 dB

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Downlink C/N Budget The satellite transponder carrying the single 300 kbps TDM outbound signal can be operated close to saturation because there is only one signal in the transponder, thus eliminating intermodulation problems. We will allow 1.0 dB backoff at the transponder output to avoid saturating the transponder, giving a transmitted power Pt  9.0 dBW. Miscellaneous losses on the downlink are 0.5 dB atmospheric loss and 3 dB for the edge of the antenna beam. Parameter EIRP of satellite Gain of receiving antenna Path loss at 1550 MHz Miscellaneous losses Received power at mobile Boltzmann’s constant System noise temperature Noise bandwidth Noise power Downlink CN

Symbol PtGt Gr Lp Lm Pr k Ts Bn N (CN)dn 

Value Units 32.0 dBW 0 dB 163.1 dB 3.5 dB 134.6 dBW 228.6 dBW/K/Hz 24.8 dBK 54.8 dBHz 149.0 dBW 134.6  (149.0)  14.4 dB

Combining the CN values for the uplink and downlink gives the overall (CN)0 ratio at the mobile terminal receiver. Converting the CN values from decibels gives (CN)up  28.6 dB  724.4, (CN)dn  14.4 dB  27.5 Hence, the overall (CN)0 for the outbound link is 1C N2 0  1 31  1C N2 up  1  1CN2 dn 4  1 30.00139  0.03644  26.5

or 14.2 dB

Note that the downlink CN ratio is so much lower than the uplink CN ratio that the overall CN ratio is almost equal to the downlink CN ratio. The clear air (CN)0 value is 5.2 dB above the minimum allowed for BER  104 on the outbound link, leaving a 5.2 dB margin for blockage by buildings, the user’s head, multipath effects, the ionosphere, or vegetative shadowing on the downlink. The link margins for the outbound link are much lower than for the inbound link, and it is therefore the weakest part of the system. Attenuation exceeding 5.2 dB in the downlink from the satellite to the mobile terminal will cause the BER to exceed 104 and the SN in the speech channel will fall below 30 dB. A SN ratio of 30 dB in a speech channel is regarded as the minimum acceptable value for intelligible communication. Because of the very steep characteristics of the BER vs CN ratio curve for BPSK, the speech channel will be unusable if downlink attenuation exceeds 5.2 dB. The link margins are quite small for a mobile system in which the line of sight between the satellite and the user can easily be blocked by trees, or by the user’s body. It is the link between the mobile terminal and the satellite that sets the overall CN value for both the inbound and the outbound links, but there is little room to change the system parameters to yield higher margins. When the mobile terminal is a satellite telephone handset, the transmitter power is limited by FCC regulations to ensure that there is no short-term biological hazard to the user when the handset is transmitting. (See Chapter 10 for further details on radiation limits for portable equipment.) The power from the satellite is limited by the transponder HPA output power and the low gain of the handset antenna. However, a higher gain antenna would have a narrower beam and would have to track the satellite automatically—a smart antenna could be built to do this, but

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the small size of most mobile telephone handsets limits the available improvement to no more than 3 or 4 dB. In the next section, we will see that the performance of the link can be improved by using forward error correction coding to lower the CN value at which the BER is 104, thus increasing the margins available on the links.

Optimizing System Performance The preceding calculations show that the LEO satellite system can support two-way digital speech with 50 active users per transponder, and provides a link margin of 8.4 dB in the inbound link and a margin of 5.3 dB in the outbound link. The RF bandwidth used by the inbound and outbound links is found from the symbol rates and the  values of the filters (see Chapter 5). For the outbound link using   0.5, the symbol rate is 300 kbaud, giving Boutbound  300  11  a2  450 kHz

For the inbound link using   0.5, the symbol rate for one speech channel is 4800 baud, giving Binbound  4.8  11  a2  7.2 kHz The inbound channels access the satellite transponder using SCPC-FDMA, so the RF signals are distributed across the transponder bandwidth. We must space the channels more than 7.2 kHz apart in the transponder so that the narrow band-pass filters in the gateway station receiver can extract each speech channel without interference from the adjacent channels. If we use a 10 kHz channel spacing, there will be a frequency gap, called a guard band of 2.8 kHz between each channel, which will ensure minimal interference from adjacent channels. With 50 channels sharing one transponder, the total bandwidth occupied in the inbound link transponder will be 500 kHz. Neither the inbound nor the outbound transponder bandwidth is fully utilized; in fact only half of the available 1 MHz is used in each case. However, we cannot add additional speech channels to the system because the CN values are already low, indicating that the system is power limited with the given link margins. We can incorporate FEC coding, however, which lowers the CN threshold for the minimum BER. Half rate convolutional coding would be a good choice in this system because the threshold CN value can be much lower. This allows a wider noise bandwidth to be used and thus better utilization of the available transponder bandwidth. Using constraint length eight and soft decision decoding, the CN ratio for BER  104 can be lowered to 3.5 dB; alternatively Turbo coding could be used. (See Chapter 7 for details of FEC techniques.) However, the bit rate of the signal is now doubled, since a half rate code adds as many coding bits as there are data bits in the bit stream. The new outbound bit rate with FEC is 600 kbps, and each inbound speech channel has a bit rate of 9600 bps. The corresponding RF bandwidths for   0.5 Nyquist filters are 900 kHz outbound and 14.4 kHz per channel inbound. With 50 active users, the RF signal bandwidths are within the available 1 MHz bandwidth of the satellite transponders. Lowering the threshold value of CN for the maximum permitted BER of 104 improves the link margins by a factor called coding gain. Coding gain is typically quoted as the difference between the CN value required for a given BER without coding and the CN required to obtain the same BER with coding. In this example, the coding gain is 8.4  3.5  4.9 dB. However, the coding gain cannot simply be added to the system margin, because the half rate FEC code doubles the bit rate of the signals and also doubles the noise bandwidth of the filters in the receivers. Thus noise power increases by 3 dB in every link receiver when FEC is added, and the CN values all fall by 3 dB. This results

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in the overall values of (CN)0 for both the inbound and the outbound links falling by 3 dB. There is no need to recalculate all the link noise budgets and CN values, since all of the values change by the same amount. (This is one advantage of using decibels for link calculations.) With half rate FEC added to the system, the new CN values are all 3 dB lower than for the system without FEC. Inbound Link (CN)up  14.7 dB (CN)dn  26.8 dB (CN)0  14.4 dB

Outbound Link (CN)up  25.6 dB (CN)dn  11.4 dB (CN)0  11.2 dB

The new link margins with a threshold overall (CN)0 of 3.5 dB are: inbound 10.9 dB, outbound 7.7 dB. Although the improvement over the earlier values without FEC is only 2 dB, the increased link margins are valuable, so FEC is invariably used in satellite personal communication systems, as it is in almost all digital wireless applications, whether satellite or terrestrial. FEC can be implemented by inserting a coding IC in the terminal, and identical ICs in the gateway station, in the baseband bit streams. The 2 dB advantage that FEC brings to the system’s link margins cannot easily be obtained any other way.

Link Margins with FEC Rain attenuation affects the Ku-band links between the gateway station and the satellite, and blockage affects the link between the mobile terminal and the satellite. Individual link margins must be calculated to determine the amount of fading or blockage that can be tolerated in each link. Excessive rain attenuation in the Ku-band links could cause the links to fail, which affects all 50 users. We must therefore ensure that the Ku-band link margins are sufficiently large to make a rain outage unlikely. Blockage of the line of sight to a mobile terminal may cause that one terminal to lose its link, but this is less serious than losing all 50 links simultaneously. The margin available for overcoming blockage should be as large as possible, but is set by the system design and cannot be improved beyond the values given above without a reduction in the number of users in the system. It should be remembered that all the calculations are for a worst case: the user is at the edge of the satellite coverage zone where the satellite is at maximum range, and also at the edge of one of the satellite’s multiple L-band beams. Most of the users have higher CN ratios on their links than the calculated worst case values most of the time, allowing greater margins for blockage of the path to the handset. In commercial satellite system design, the fact that most users are not at the edge of the coverage zone most of the time is used in developing a “coverage advantage” factor that increases the average link margin available per user and thus optimizes traffic capacity.

Rain Attenuation at Ku Band Rain causes attenuation at Ku band, as discussed in Chapter 8. We must calculate the rain attenuation margins for the inbound downlink and the outbound uplink and determine the probability of an outage. The link margin is the number of decibels by which

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the CN ratio on an uplink or a downlink can be reduced before the overall (CN)0 for that link falls to the threshold value. We will use 3.5 dB as the threshold value for overall CN in each case, assuming that half rate FEC is used. We will also assume that clear sky conditions prevail on the uplink when extreme attenuation occurs on the downlink, and vice versa. For the inbound Ku-band downlink, using half rate FEC, the clear air CN ratio is 26.8 dB (ratio 478.6) and the L-band clear uplink CN ratio is 14.7 dB (ratio 29.5). With a threshold at 3.5 dB (ratio 2.24), the minimum downlink CN will be given by (using ratios, not dB) 1 1C N2 dn min  1 1CN2 0  1  1CN2 up  12.24  1 478.6  0.444

Hence the minimum permitted value for (CN)dn  2.25 1 3.5 dB. The downlink margin is 26.8  3.5  23.3 dB. Rain attenuation at 11.5 GHz very rarely exceeds this value in the United States, so for a U.S. system, the Ku-band downlink margin is adequate. The Ku-band uplink has a clear sky (CN)up ratio of 25.6 dB, but attenuation of the uplink signal causes a reduction in received power at the transponder input. If the satellite transponders are linear (bent pipe), the output power will fall when the input power is reduced by uplink rain attenuation. Because the transponder is operated close to saturation, there will not be a one-to-one correspondence in the changes in power level at the input and output, but exact analysis is beyond the scope of this example; a linear relationship will be assumed here. The transponder nonlinearity actually increases the uplink rain attenuation margin, because the output signal from the satellite will fall less than the input signal to the satellite, so the results that follow represent a pessimistic estimate of the margin available. A regenerative repeater always transmits at constant output power and is very desirable in a digital system. It avoids the difficulty of attenuation on the uplink causing a reduction in transponder output power. Applying the same analysis as used for the Ku-band downlink, with (CN)dn  11.4 dB (ratio 13.8) in clear sky conditions and (CN)0 min  3.5 dB (ratio 2.24) 1 1C N2 up min  1/1CN2 0  1  1CN2 dn  12.24  1 13.8  0.374

Thus the minimum (CN)up ratio is 10 log(10.374)  4.3 dB, ignoring the effects of coupling between input and output power in the transponder. When the latter effect is considered with a linear transponder characteristic, the limit is set by the (CN)dn ratio falling to 3.5 dB. This will occur with 11.4  3.5  7.9 dB uplink attenuation, which is the limiting value. Uplink power control (UPC) can be used to prevent the input power level of the transponder from falling when rain affects the uplink. It would be straightforward to use uplink power control in this case. Attenuation on the downlink at 11.5 GHz is measured using the satellite beacon, scaled to 14.4 GHz, and used to set the gateway station transmit power level. If a 10 dB dynamic range of UPC is available, and attenuation is allowed to reach 2 dB at 14.4 GHz before the UPC comes in, the downlink CN ratio will not fall below 9.4 dB in an uplink fade, leaving a downlink margin of 5.9 dB during uplink rain fades. This gives the 14.4 GHz uplink a rain attenuation margin of 12 dB, which would maintain the link for better than 99.99% of a year throughout the United States. The openloop UPC system discussed here would probably have a margin of error of at least 1 dB in estimating uplink attenuation under low fading conditions. Uncertainties in identifying the propagation mechanism that is causing the fading and the difficulty of accurately setting the clear sky baseline for the signal make greater accuracy unlikely.

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If two mobile terminals are located within the same satellite beam coverage, and are therefore operating through the same gateway earth station, the assumption of nonsimultaneous outage of the two links would not be valid. Such situations are expected to be rare occurrences. The gateway station would typically be sited in a dry region, such as Wyoming or Idaho in the United States, to minimize the number and severity of rain attenuation events. Thus rain attenuation at Ku band can be overcome by a large link margin for the downlink and implementation of uplink power control in the uplink, and by intelligent siting of the station. All 50 channels can be guaranteed to be unaffected by rain at the gateway station.

Path Blockage at L-Band Trees, buildings, and people are the most likely causes of blockage that affect the performance of the mobile terminal at L band. Blockage by buildings is too severe to allow the L-band link to operate, and most LEO satellite telephones will not work indoors. Some systems like Iridium incorporate a cellular telephone into the handset. The cellular telephone is used in preference to the satellite phone to reduce loading on the LEO satellite system, and also whenever the satellite signal is unavailable, such as indoors. Paging options have been designed into some mobile satellite systems which permit users to be alerted that there is an incoming call. The user still has to run outdoors to be able to receive the call, and this has evidently been a factor deterring the use of satellite telephone by business people. The link margins for the L-band links are calculated in the same way as the Kuband margins. Repeating the calculations with a minimum overall CN ratio of 3.5 dB and no rain attenuation in the Ku-band links gives L-band uplink margin  11.4 dB L-band downlink margin  7.8 dB The downlink from the satellite to the mobile terminal is therefore the most vulnerable of the links, and cannot be made robust without reducing the number of users per transponder. However, the value of 7.8 dB for the downlink margin is a worst case value and most of the users will have a margin several decibels higher. A margin of 7.8 dB can be exceeded by attenuation through a stand of trees. For example, if the user is in a vehicle traveling along a road cut through a forest, and the satellite has a low elevation angle, the 7.8 dB attenuation margin may be exceeded from time to time, causing repeated break up of the downlink signal. Transmission protocols and signal buffering can be designed to reduce the impact of this type of intermittent loss of signal. Multipath effects when the satellite is at a low elevation angle can also cause variations in signal level leading to lower performance and occasional outages in extreme cases, such as an overwater path.

Summary of L-Band Mobile PCS System Performance The personal communication system in this example uses a network of low earth orbit satellites to link a user anywhere in the system’s coverage zone to a gateway station, and then to the public switched telephone network or another mobile terminal. The user’s terminal operates in L band and is similar to a cellular telephone, with a low gain, omnidirectional antenna. The transmissions are digital, with BPSK modulation, and use speech

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compression to achieve a bit rate of 4.8 kbps per speech channel. There are a maximum of 50 users in each of the satellite’s 50 L-band beams, giving a nominal satellite capacity of 2500 users. The inbound link from the user to the gateway station has a margin of 11.4 dB for tree shadowing on the uplink to the satellite. The downlink from the satellite has a blockage margin of 7.8 dB. The Ku-band links between the satellite and the gateway station have large margins, and uplink power control is used to prevent uplink rain attenuation at 14 GHz from adversely affecting the downlinks to the mobile terminals. TDM is used on the outbound link with half rate FEC coding and a bit rate of 600 kbps. SCPC-FDMADAMA is used on the inbound links, with a channel bit rate of 9.6 kbps after FEC coding is applied.

4.9

SUMMARY

This chapter has set out the procedures for calculation of received power from a satellite and noise power in a receiver. Together, these figures give the CN ratio for the receiving system. The specification of a system will always require a minimum CN in the receiver, below which the link is considered inoperable. The design of a link to achieve that minimum CN requires repeated application of the link and noise power equations to give CN for clear air conditions with acceptable bandwidth and antenna dimensions. When a linear (bent pipe) transponder is used, the clear air value of CN ratio for the uplink and downlink must be combined to give the overall (CN)0 ratio in the earth station receiver. Once clear air performance has been calculated, the effect of rain on the slant paths must be determined and the propagation path statistics need to be studied to determine how much margin is required to meet worst-case conditions. Examples are presented throughout this chapter showing how link power and noise budgets are

used to find the overall (CN)0 ratio for different systems. Fading of both uplink and downlink simultaneously is unlikely for 64 and 1411 GHz systems and can safely be ignored when computing link statistics. At 3020 GHz the possibility cannot be ignored and the joint effect has to be calculated. No attempt has been made to derive an optimization procedure for the design of the “best” system within a frequency band and CN ratio specification. There are too many variables in the system, including the cost of antennas, receivers, and other components, to produce a single optimization procedure. Iterative techniques must be used to find a set of parameters for the earth stations and satellite that provide the performance required from the satellite communication system. The designer of a satellite communication system may have to go through several trial design procedures and compare the resulting systems to determine which one best suits the particular application.

REFERENCES 1. S. SILVER, ed., Microwave Antenna Theory and Design, Vol. 12, MIT Radiation Lab Series, 1947. (Republished by Peter Perigrinus, Stevenage, Herts, UK, 1984.) 2. W. L. STUTZMAN and G. A. THIELE, Antenna Theory and Design, John Wiley & Sons, New York, 1981. 3. J. D. KRAUS, Radio Astronomy, Cygnus-Quasar Books, Powell, OH, 1982. (Originally published by McGrawHill, New York, 1966.) 4. H. L. KRAUSS, C. W. BOSTIAN, and F. H. RAAB, Solid State Radio Engineering, John Wiley & Sons, New York, 1980. 5. K. S. SHANMUGAM, Digital and Analog Communication Systems, John Wiley & Sons, New York, 1979, pp. 356–360.

6. H. TAUB and D. L. SCHILLING, Principles of Communication Systems, McGraw-Hill, New York, 1971. 7. M. L. GUSTAFSON, “The KLM Sky Eye 10 Receiver,” Satellite Television, 2, 66–70, November 1984. 8. J. DICKS and M. BROWN, JR., “INTELSAT IV-A Transmission System Design,” Comsat Technical Review. 5, 73–103, 1975. 9. I. A. GLOVER and P. M. GRANT, Digital Communications, Prentice-Hall, Europe, 1998. 10. WALTER L. MORGAN and GARY D. GORDON, Communications Satellite Handbook, John Wiley & Sons, New York, 1989. 11. G. MARAL and M. BOUSQUET, Satellite Communication Systems, John Wiley & Sons, New York, 1986.

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PROBLEMS 1. A C-band earth station has an antenna with a transmit gain of 54 dB. The transmitter output power is set to 100 W at a frequency of 6.100 GHz. The signal is received by a satellite at a distance of 37,500 km by an antenna with a gain of 26 dB. The signal is then routed to a transponder with a noise temperature of 500 K, a bandwidth of 36 MHz, and a gain of 110 dB. a. Calculate the path loss at 6.1 GHz. b. Calculate the power at the output port (sometimes called the output waveguide flange) of the satellite antenna, in dBW. c. Calculate the noise power at the transponder input, in dBW, in a bandwidth of 36 MHz. d. Calculate the CN ratio, in dB, in the transponder. e. Calculate the carrier power, in dBW and in W, at the transponder output. 2. The satellite in Problem 1 above serves the 48 contiguous states of the United States. The antenna on the satellite transmits at a frequency of 3875 GHz to an earth station at a distance of 39,000 km. The antenna has a 6 E–W beamwidth and a 3 N–S beamwidth. The receiving earth station has an antenna with a gain of 53 dB and a system noise temperature of 100 K and is located at the edge of the coverage zone of the satellite antenna. (Assume antenna gain is 3 dB lower than in the center of the beam.) Ignore your result for transponder output power in Problem 1 above. Assume the transponder carrier power is 10 W at the input port of the transmit antenna on the satellite. a. Calculate the gain of the satellite antenna in the direction of the receiving earth station. [Use the approximate formula G  33,000(product of beamwidths).] b. Calculate the carrier power received by the earth station, in dBW. c. Calculate the noise power of the earth station in 36 MHz bandwidth. d. Hence find the CN in dB for the earth station. 3. A 1411 GHz satellite communication link has a transponder with a bandwidth of 52 MHz which is operated at an output power level of 20 W. The satellite transmit antenna gain at 11 GHz is 30 dB toward a particular earth station. Path loss to this station is 206 dB, including clear air atmospheric loss. The transponder is used in FDMA mode to send 500 BPSK voice channels with half rate FEC coding. Each coded BPSK signal has a symbol rate

of 50 kbps and requires a receiver with a noise bandwidth of 50 kHz per channel. The earth stations used to receive the voice signals have antennas with a gain of 40 dB (1 m diameter) and a receiver with Tsystem  150 K in clear air, and IF noise bandwidth of 50 kHz. a. Calculate the power transmitted by the satellite in one voice channel. b. Calculate the CN in clear air for an earth station receiving one BPSK voice signal. c. What is the margin over a coded BPSK threshold of 6 dB? 4. Geostationary satellites use L, C, Ku, and Ka bands. The path length from an earth station to the GEO satellite is 38,500 km. For this range, calculate the path loss in decibels for the following frequencies: a. 1.6 GHz, 1.5 GHz b. 6.2 GHz, 4.0 GHz c. 14.2 GHz, 12.0 GHz d. 30.0 GHz, 20.0 GHz 5. Low earth orbit satellites use mainly L band, with ranges varying from 1000 to 2500 km. Calculate the maximum and minimum path loss from earth to a satellite, in dB, for the uplink frequency of 1.6 GHz, and the downlink frequency of 1.5 GHz. 6. A geostationary satellite carries a transponder with a 20 W transmitter at 4 GHz. The transmitter is operated at an output power of 10 W and drives an antenna with a gain of 30 dB. An earth station is at the center of the coverage zone of the satellite, at a range of 38,500 km. Using decibels for all calculations, find: a. The flux density at the earth station in dBWm2. b. The power received by an antenna with a gain of 39 dB, in dBW. c. The EIRP of the transponder in dBW. 7. An LEO satellite has a multibeam antenna with a gain of 18 dB in each beam. A transponder with transmitter output power of 0.5 W at 2.5 GHz is connected to one antenna beam. An earth station is located at the edge of the coverage zone of this beam, where the received power is 3 dB below that at the center of the beam, and at a range of 2000 km from the satellite. Using decibels for all calculations, find: a. The power received by an antenna with a gain of 1 dB, in dBW. b. The noise power of the earth station receiver for a noise temperature of 260 K and an RF channel bandwidth of 20 kHz.

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c. The CN ratio in dB for the LEO signal at the receiver output. 8. A satellite in GEO orbit is at a distance of 39,000 km from an earth station. The required flux density at the satellite to saturate one transponder at a frequency of 14.3 GHz is 90.0 dBW/m2. The earth station has a transmitting antenna with a gain of 52 dB at 14.3 GHz. Find: a. The EIRP of the earth station. b. The output power of the earth station transmitter. 9. A 12-GHz earth station receiving system has an antenna with a noise temperature of 50 K, a LNA with a noise temperature of 100 K and a gain of 40 dB, and a mixer with a noise temperature of 1000 K. Find the system noise temperature. 10. A geostationary satellite carries a C-band transponder which transmits 20 W into an antenna with an onaxis gain of 30 dB. An earth station is in the center of the antenna beam from the satellite, at a distance of 38,000 km. For a frequency of 4.0 GHz: a. Calculate the incident flux density at the earth station in watts per square meter and in dBW/m2. b. The earth station has an antenna with a circular aperture 2 m in diameter and an aperture efficiency of 65%. Calculate the received power level in W and in dBW at the antenna output port. c. Calculate the on-axis gain of the antenna in dB. d. Calculate the free space path loss between the satellite and the earth station. Calculate the power received, Pr, at the earth station using the link equation: Pr  PtGtGrLp where PtGt is the EIRP of the satellite transponder and Lp is the path loss. Make your calculation in dB units and give your answer in dBW. 11. Repeat parts (a) through (d) of Problem 10 for a Ka-band transponder transmitting at a frequency of 20.0 GHz. 12. This sequence of questions requires you to design a communication link through a geostationary satellite to meet a CN and link margin specification. Use these constants: Boltzmann’s constant k  228.6 dBW/K/Hz Path length to satellite  38,500 km SATELLITE Geostationary at 73 W longitude. 24 C-band transponders, 28 Ku-band transponders 3.2 kW RF power output

Antenna gain, on axis, C band and Ku band (transmit and receive) Receive system noise temperature (C band and Ku band) Transponder saturated output power: C band Transponder bandwidth: C band Transponder saturated output power: Ku band Transponder bandwidth: Ku band

 31 dB  500 K  40 W  36 MHz  80 W  54 MHz

SIGNALS FM-TV analog signal to be received in a bandwidth of 27 MHz. Multiplexed digital TV signals transmitted as QPSK with symbol rate 27 Msps using half rate FEC with coding gain 5.5 dB Minimum permitted CN overall  9.5 dB 12.1 Design a transmitting earth station to provide a clear air CN of 26 dB in a C-band transponder at a frequency of 6.285 GHz. Use an uplink antenna with a diameter of 9 m and an aperture efficiency of 68%, and find the uplink transmitter power to achieve the required CN. The uplink station is located on the 2 dB contour of the satellite footprint. Allow 0.5 dB for clear air atmospheric attenuation and other losses. 12.2 Design a C-band receiving earth station to provide an overall clear air CN of 13 dB in a 27 MHz IF noise bandwidth at a carrier frequency of 4.06 GHz. The antenna noise temperature is 20 K and the LNA noise temperature is 55 K. You may assume a high gain LNA and ignore the noise generated in other parts of the receiver. The C-band satellite transponder is operated with 1 dB output backoff. Clear air atmospheric attenuation on the downlink and other losses total 0.5 dB. Determine the diameter of the receiving antenna, assuming an aperture efficiency of 65%. The receiving terminal is located on the 3 dB contour of the satellite footprint. Reminder: Overall CN includes the effect of noise radiated by the satellite transponder. 12.3 a. Under conditions of heavy rain, the C-band path from the transmitting station suffers an attenuation of 2 dB. Calculate the overall CN at the earth station in a bandwidth of 27 MHz under these conditions, and find the uplink link margin. Reminder: The uplink margin is the number of dB of attenuation that can occur on the uplink before the receiver overall CN reaches the limit of 9.5 dB. b. Under conditions of heavy rain, the C-band path to the receive station suffers an attenuation of 1.5 dB. Assuming 100% coupling of sky noise into antenna

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noise, and 0.3 dB clear air gaseous attenuation, calculate the overall CN under these conditions, and find the downlink margin. Hint: You need to find the sky noise temperature that results from a total excess path attenuation of 1.8 dB (clear air attenuation plus rain attenuation); this is the antenna temperature. Then compute the new CN in rain, using the new Tsystem and received power values. 12.4 Design a transmitting earth station to provide a clear air CN of 30 dB in a Ku-band transponder at a frequency of 14.15 GHz. Use an uplink antenna with a diameter of 5 m and an aperture efficiency of 68%, and find the uplink transmitter power required to achieve the required CN. The uplink station is located on the 2 dB contour of the satellite footprint. Allow 1.0 dB on the uplink for miscellaneous and clear air losses. 12.5 Design a Ku-band receiving earth station to provide an overall clear air CN of 17 dB in a 27 MHz IF noise bandwidth at a carrier frequency of 11.45 GHz. The antenna noise temperature is 30 K and the LNA noise temperature is 110 K. You may assume a high gain LNA and ignore the noise generated in other parts of the receiver. Determine the diameter of the receiving antenna. The receiving terminal is located on the 3 dB contour of the satellite footprint, and clear air attenuation on the path and other losses total 0.8 dB. 12.6 a. Under conditions of heavy rain, the Kuband path to the satellite station suffers an attenuation of 6 dB. Calculate the overall CN at the earth station in a bandwidth of 27 MHz under these conditions, and find the uplink link margin. b. Under conditions of heavy rain, the Ku-band path to the receive station suffers an attenuation of 5 dB. Assuming 100% coupling of sky noise into antenna noise, and 0.3 dB clear air attenuation, calculate the overall CN under these conditions, and find the downlink margin. 13. A direct broadcast television (DBS-TV) satellite is in geostationary orbit at 100 west longitude. It carries 16 transponders, each with a saturated output power of 200 W and a bandwidth of 25 MHz. The antenna on the satellite has a gain (on axis) of 34 dB. The receiving terminals all use antennas with a circular aperture with a diameter of 18 inches and an aperture efficiency of 65%. The noise bandwidth of the digital TV receiver is 20 MHz. Use a distance to the GEO satellite of 38,500 km in your calculations. a. Calculate the free space path loss and the receiving terminal antenna gain at 12.2 GHz.

153

b. Draw up a link budget for the downlink from the satellite to an earth station on the 3 dB contour of the satellite antenna beam. Assume that the satellite transmits at a power level of 180 W. Include a clear air atmospheric loss of 0.5 dB and miscellaneous losses of 0.2 dB in your downlink power budget. c. The receiving terminal has a system noise temperature of 110 K in clear air. Draw up a noise power budget for the receiver using the receiver’s noise bandwidth. d. Calculate the clear air CN ratio for the receiver with a noise bandwidth of 20 MHz. The minimum permissible CN ratio is 10.0 dB. What is the clear air link margin? e. For 0.3% of the time at the receiving location, heavy rain causes 2 dB excess path attenuation and the system noise temperature of the receiver increases to 210 K. Calculate the CN under these rain conditions, and the link margin above the CN threshold of 10.0 dB. f. Many of the DBS-TV system customers live inside the 2 dB contour of the satellite beam. Calculate the clear air link margin and 0.3% time link margin for a receiver located on the 2 dB contour of the satellite footprint. g. An uplink station for the DBS-TV satellite described in Question 1 is located in Utah, and transmits digital TV signals to 16 transponders on the satellite using QPSK with three-quarter rate forward error correction. The transmit earth station has a circular aperture antenna with diameter of 6 m and an aperture efficiency of 65%. Each transponder operates at a different carrier frequency in the 17 GHz band, and the RF channel noise bandwidth is 20 MHz. The noise temperature of the satellite receiver is 500 K (the satellite always looks toward the “hot” earth). Use these values in the remaining parts of this question. Calculate the uplink path loss and the uplink antenna gain at 17.5 GHz. h. The gain of the receiving antenna on the satellite in the direction of Utah is 31 dB. Draw up a clear air uplink budget for the link from the earth station to a single transponder on the satellite using a transmit power of Pt watts, and atmospheric and other losses of 1.0 dB. i. Calculate the noise power at the input to the satellite receiver in a noise bandwidth of 20 MHz. Hence, find the uplink transmitter power required to achieve a CN of 28 dB in the satellite transponder. j. The gain of the satellite transponder must be set to amplify the received signal at the transponder input to an output level of 180 W. Calculate the gain of the

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transponder in decibels. (Ignore the change in frequency in the transponder.) When designing RF equipment, a common rule to avoid oscillation is to make the amplification at any given frequency no higher than 60 dB. How would you design a bent pipe DBS-TV transponder to provide the end to end gain that you calculated? k. The minimum permissible CN in the transponder is 16.0 dB. Calculate the clear air link margin for the uplink. l. Ignore the result you calculated for the downlink CN in Problem 1, and use a value of 15 dB in this question. Convert the clear air uplink and downlink CN values to power ratios, and then find the overall CN, in dB, in the earth station receiving terminal. Use the following formula (where CN values are ratios, not in dB) and give your answer in decibels: 1 1CN2 overall  1 1C N2 up  1 1CN2 down 14. This is a multipart question. All the questions are about the satellite communications system described below. DESCRIPTION OF SYSTEM A satellite communication system consists of 50 LEO satellites in 750 km orbits, several hubs stations operating in Ka band, and many handheld transceivers operating in L band. The handheld units transmit to transponders at 1600 MHz and receive from transponders at 2500 MHz. The system uses digital speech compressed into a transmission channel (RF) bandwidth of 16 kHz. Channels are spaced 20 kHz apart to allow a guard band between channels. The parameters of the system are given below. (You may not need all of these.) SYSTEM VALUES Uplink frequency for handheld transceiver

1600 MHz

Downlink frequency for handheld transceiver

2500 MHz

Uplink frequency for hub station

29 GHz

Downlink frequency for hub station

19 GHz

Maximum range to edge of coverage zone

2000 km

SATELLITE TRANSPONDER Maximum output power Pt Transponder bandwidth Transponder input noise temperature Ts

20 W 2 MHz 500 K

HANDHELD TRANSCEIVER PARAMETERS Transmitter output power Antenna gain (transmit and receive) G

1.0 W 0 dB

Receiver system noise temperature Ts

300 K

Receiver system noise bandwidth Bn

100 kHz

HUB STATION PARAMETERS Maximum transmit power Pt

100 W

Receiver system noise temperature (clear air) Ts

250 K

Antenna gain at 29 GHz (transmit) Gt

54 dB

Antenna gain at 19 GHz (receive) Gr

52 dB

CONSTANTS Boltzmann’s constant k  1.38  1023 J/K  228.6 dBW/K/Hz 14.1. Preliminary calculations a. Calculate the path loss, in dB, for a 2000 km path at 1.6, 2.5, 19, and 29 GHz. b. Calculate the noise power, in dBW, for the receiver in the transponder and for the receivers at the hub station and the handheld unit, in a single voice channel bandwidth of 10 kHz. (Note: Use the bandwidth of one speech channel, 10 kHz for all the calculations, not 2 MHz.) c. The satellite has broad coverage antennas at L band and Ka band with half-power beamwidths of 120. Estimate the gain, in dB, of the antennas at each frequency. 14.2. C/N Ratios Use the values you obtained in Problem 1 above, for path loss, antenna gain, and noise power in this question. Calculate CN values for stations located at the edge of the coverage zone of the satellite, where the satellite antenna gain is 3 dB below its maximum value, and the range to the satellite is 2000 km. Take care to use the correct path loss and receiver noise power values for each frequency. Give your answers in decibels. a. Calculate the CN in the satellite transponder for the signal transmitted by one handheld transceiver located at the edge of the coverage zone (satellite antenna gain 3 dB below maximum) and at maximum range from the satellite (2000 km). b. Calculate the CN in the satellite transponder for the signal transmitted by a hub station, using its full output power. c. Calculate the CN in the hub station receiver for the signal transmitted by a satellite transponder using its full output power.

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d. Calculate the CN in the receiver of the handheld unit for the signal transmitted by a satellite transponder using its full output power. e. Calculate the overall CN ratios at the hub station and at the handheld receiver. 14.3. Trade-off studies The link between the hub station and the satellite operating at Ka band uses a high gain antenna at the hub station and achieves a high CN. The transceiver operating in L band uses a low gain, omnidirectional antenna with low gain, which results in low CN. For satisfactory operation under all weather conditions, the Ka-band links should have a minimum CN of 20 dB in clear air, and the L-band links should have a minimum CN of 10 dB. The CN of the handheld transceivers can be improved by using a multiple beam L-band antenna on the satellite, with higher gain and narrower beamwidth per beam. The high CN of the hub station links can be traded for increased capacity. The hub station and transponder transmitter power can be shared among a group of voice channels. a. Determine the minimum gain required by the Lband antennas on the spacecraft to achieve a CN of 10 dB at each L-band frequency. Using the higher of the two values, find the 3 dB beamwidth of one of the multiple beams. Estimate the number of beams that will be needed to serve the coverage zone of a single 120 beamwidth antenna. b. Find the excess CN available on the Ka-band links between the hub station and the satellite. By trading carrier power for capacity, find the number of channels that the Ka-band links can carry with CN  20 dB. If the channel spacing is 20 kHz, can all of these channels fit into a 2-MHz bandwidth transponder? c. Determine whether the transponders are power limited or bandwidth limited. Give reasons for your answer.

155

d. The communication system described needs two transponders to permit two-way voice communication between the hub station and the many transceivers. Based on your answers in Problem 3, find the gain of each transponder from input port to output port. (Note: The transponder gain does not include antenna gain.) 14.4. Costs For any satellite system to be viable, the communication capacity must be sold at a price that is attractive to customers. This question looks at the cost of the system over its lifetime and calculates a minimum cost per voice circuit. a. Each LEO satellite carries 20 transponders. What is the total number of speech channels that the satellite can support when fully loaded? How many telephone circuits (it takes two channels to make a telephone circuit)? b. Each LEO satellite costs $40 M in orbit and the LEO system costs $100 M per year to run. The expected lifetime of the satellites is 10 years, and the system requires a total of 10 spare satellites to be launched over the 10-year period. Calculate the cost of operating the system for a 10-year period. Add a 27% factor to cover interest payments and dividends, and calculate the 10-year cost of the entire system. c. Calculate the cost per minute per voice circuit assuming that each satellite can be loaded to an average of 20% of its capacity over its lifetime. d. Write two paragraphs discussing the cost of the system and the cost of a voice circuit. What price per minute would you set for a satellite voice circuit? Would you expect customers to be willing to pay this amount for a satellite telephone connection? How does the cost compare to terrestrial cellular telephone charges?

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5

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Communications satellites are used to carry telephone, video, and data signals, and can use both analog and digital modulation techniques. When GEO satellites were first used for communications in the 1960s and 1970s, the signals were almost exclusively analog. The advent of satellite communications made possible the transmission of wide bandwidth signals between continents. For the first time, video signals could be sent between North America, Europe, and Asia. Thousands of telephone channels could be multiplexed through one transponder and sent across the United States or across the Atlantic or Pacific oceans. The modulation and multiplexing techniques that were used at this time were analog, adapted from the technology developed for microwave links in the previous two decades. Frequency modulation (FM) was the modulation of choice and frequency division multiplexing (FDM) was used to combine hundreds or thousands of telephone channels onto a single microwave carrier. Regional domestic and international satellite systems were developed to exploit the high capacity and bandwidth that satellites offered. In the 1980s, optical fibers came into widespread use, and GEO satellites were no longer used for telephony within the United States. Long-distance telephone links using optical fibers are digital, so all telephone signals sent via optical fiber have to be in digital form. At the same time, telephone exchanges became large digital computers instead of large banks of mechanical switches. The change to digital voice signals made it easier for long-distance communication carriers to mix digital data and telephone traffic and send it through the same optical fibers and telephone exchanges. This forced telephone signals to be converted to digital form at the telephone exchange, and rendered all the analog multiplexing methods obsolete. FDM has all but disappeared as a way to combine analog telephone signals, replaced by time division multiplexing of digital voice signals. In the first edition of Satellite Communications, the FM-FDM multiplexing techniques were covered in detail. This material is included in the second edition in Appendix B, because there are parts of the world where the older analog technology is still in use, and some readers may want to refer to that material. The distribution of television program material in North America and much of the rest of the world is by satellite. Satellites are particularly useful for distributing the same signal to hundreds or thousand of receivers (point to multipoint) and many of the world’s GEO satellite transponders are used for video distribution to cable television systems. Television signals continue to be transmitted via GEO satellites using frequency modulation and analog video signals in 2000, but it seems probable that eventually all distribution of video signals will be digital. Digital television has emerged in the form of direct broadcast satellite television (DBS-TV) and high definition television (HDTV) is also digital. Frequency modulation for the transmission of video signals, and also for single channel per carrier voice signals, is included in this chapter. FM is the last remaining form of analog modulation used in satellite systems. 156

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Analog multiplexing in the form of FDM has virtually disappeared, but frequency division multiple access, FDMA, remains one of the major ways in which transponder capacity is shared among users. FDMA divides up the frequency band in the transponder into channels, which are allocated to different signals on a fixed or on-demand basis. Most SCPC systems employ FDMA, for example, and FDMA is widely used in VSAT systems. Multiple access techniques are discussed in Chapter 6: this chapter concentrates on the modulation methods and the digital multiplexing techniques that are used for voice signals. The T-system and ITU digital hierarchies are described briefly in this chapter, and since most signals are now transmitted digitally, analog signals must first be converted to digital form and that process is also described. Once an analog signal is in digital form, it can be transmitted over any digital communication link, multiplexed with other digital signals, and sent very long distances without degradation. One major advantage of digital transmission systems over analog is that error free transmission is possible. In a digital telephone system, error free transmission means that no noise is injected into the baseband channel, regardless of the transmission distance, so a telephone call over a distance of 10,000 km has the same quality as a call over a distance of 10 km. The design of digital communication links requires a different approach from the design of analog links, although achieving optimum performance in a digital communication system requires a link with linear characteristics—an analog circuit rather than a digital circuit. The techniques for the design of digital communication systems are covered in some detail because a good understanding of these methods is essential in setting CN ratios correctly for a digital satellite link, and for estimating the bit error rate that can be expected. The topic of bit error rates for BPSK and QPSK is covered briefly in this chapter, and there is also a description of the compression techniques used for digital voice and video signals. For a more extensive treatment, the reader is referred to any of several excellent texts on communication theory and communication systems1–4.

5.1

FREQUENCY MODULATION Frequency modulation is used widely in analog radio communication systems. FM broadcasting is the most familiar, and FM is also used in analog radio communication systems such as AMPS cellular telephones. Video signal distribution for the cable TV industry is now the only significant remaining analog FM satellite service. Digital signals may be transmitted using a frequency modulation system called frequency shift keying (FSK). FSK is rarely used in satellite links; phase shift keying (PSK) is preferred. In general PSK gives lower BER for a given CN ratio than FSK. Frequency modulation is used in analog satellite systems because it can provide an improvement in the baseband SN ratio relative to the CN ratio in the IF part of the receiver. We saw in Chapter 4 that satellite links typically have receiver (CN)0 ratios between 5 and 25 dB. Most analog systems, including TV, try to maintain a SN ratio of 50 dB in the baseband channel. The use of FM can provide the improvement that can deliver baseband SN ratios of 50 dB when the CN is 15 or 20 dB. In general, we have a baseband SN ratio given by SN  1CN2 0  FM improvement dB

(5.1)

where (CN)0 is the overall carrier-to-noise ratio in the earth station receiver at the input to the FM demodulator. The main disadvantage of using FM in this way is that we need wideband FM (WBFM) to achieve the improvement factor, and wideband FM, as its name implies, uses up a lot of precious RF bandwidth. There is a general principle in communication theory

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that you can trade bandwidth for power: in this case trading occupied RF bandwidth for carrier power at the earth station input. As a result, a NTSC (National Television Standards Committee) video signal that occupies 4.2 MHz bandwidth at baseband becomes an RF signal with a bandwidth of 30 MHz in a satellite transponder. In an FM signal the information is carried by changes in the carrier frequency—called the frequency deviation, or just deviation. The amplitude of the FM wave is constant, so an FM signal is sometimes called a constant power waveform since the power in the waveform doesn’t change with the modulation. The frequency of an FM wave varies directly in proportion to the voltage of the baseband modulation signal. Thus the instantaneous frequency, fi, of an FM wave is given by the linear relationship fi  fc  ¢f  fc  kf m1t2

(5.2)

where fc is the carrier frequency, f is the frequency deviation, m(t) is the modulating voltage, and kf is the modulator constant (modulator sensitivity) in hertz/volt. The modulator constant can have any value, although values around 10 MHz/V are common in analog satellite communication systems used for video signal distribution. Recovering the information signal from an FM wave is conceptually very simple— we need a frequency to voltage converter with a constant K  1kf volts/hertz. Then the recovered baseband signal is v(t) where v1t2  ¢f  1 kf  kf m1t2  1 kf  m1t2

(5.3)

Thus an FM radio link can have unity gain between the transmitter baseband and the receiver baseband. An FM link can also transmit a DC (direct current) level, since voltage is represented by frequency deviation of the carrier. Both of these features are valuable for telemetry links.

Waveform Equation for FM We must find the phase angle of the FM waveform because it is conventional to write the equation of the wave in the form VFM(t)  A cos(t   (t))  A cos(2 fc  (t)). This expresses the frequency deviation in terms of phase variations. (The units of t  (t) in the term cos(t  (t)) must be radians or degrees, since the argument of a cosine must be an angle.) After a time t, the phase angle, (t), of the FM wave modulated by an information signal m(t) is t

£1t2 

 32pf  2pk m 1t2 4 dt  f c

f

0

radians

(5.4)

0

where 0 is the starting phase angle at t  0. Hence the equation of the FM signal is given by VFM 1t2  A cos vct 

t

 1 32pk m1t2 4dt  f 2 f

0

volts

(5.5)

0

In general, we cannot evaluate the spectrum of the FM wave from Eq. (5.5) unless we know an analytical expression for m(t). The spectrum that is generated with any modulating waveform in an FM system is much more complex than for amplitude modulation (AM) because FM is inherently a nonlinear modulation technique. A sine wave test tone at frequency fm applied to an ideal FM modulator generates many sidebands at all possible integer multiples of the test tone

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frequency. Thus instead of the familiar pair of upper and lower sidebands in AM, frequency modulation with a sine wave modulating signal generates an infinite number of sidebands at frequencies fc  fm, fc  2fm, fc  3fm … In practice, real modulators and transmitters have finite bandwidths, so a finite range of sidebands is actually present in any transmitted FM signal. The magnitudes of all the sidebands depend on both the amplitude and the frequency of the modulating signal, and to make matters worse, if the modulating signal is a sinusoid, the magnitudes of the sidebands can be found only from Tables of Bessel functions. If two sine waves at angular frequencies 1 and 2 are simultaneously applied to a wideband FM modulator, the spectrum of the resulting FM wave contains not only all the sidebands for each sine wave, but also all the sidebands from every possible crossmodulation product between the waves—frequency terms of the form c  m1  n2. Clearly, any real modulating signal which contains many frequencies will generate an FM spectrum that is continuous, rather than having identifiable sidebands.

Bandwidth of FM Signals: Carson’s Rule Most texts on communication theory expand the FM waveform equation using a sinusoidal test tone as the baseband modulating signal. The expansion of the FM waveform equation leads to an infinite number of harmonic terms of the form Jn() cos(c  nm)t where m is the frequency of the test tone. The magnitudes of the terms are given by Bessel function coefficients Jn(). There is, fortunately, a much more simple and useful way to determine the bandwidth required for an FM signal: Carson’s rule1. Carson’s rule is empirical, but it works well and is the way that almost every communications engineer working with an FM system actually determines the bandwidth of FM signals. Carson’s rule states that the bandwidth required to transmit an FM signal is given by B  21 ¢fpk  fmax 2 Hz

(5.6)

where fpk is the peak frequency deviation and fmax is the highest frequency present in the modulating signal. Students often have difficulty in understanding frequency modulation, particularly when Bessel functions are involved. In practice, Bessel functions are not needed to understand FM. Frequency modulation is very straightforward if you remember two facts: 1. The frequency deviation of the carrier is directly proportional to the modulating signal voltage. 2. The bandwidth required to transmit an FM signal is found from Carson’s rule.

Baseband SN Ratio for FM Signals Chapter 4 described how to calculate the overall (CN)0 ratio for an earth station receiver. The calculation was made using the noise bandwidth of the narrow band-pass filter in the IF amplifier of the receiver, at the output of the IF amplifier and immediately before the FM demodulator input. For an FM receiver, the value of overall (CN)0 ratio calculated for the earth station receiver is the CN ratio used here to calculate baseband SN values. The carrier power is constant in any FM wave, which is simply a sine wave of varying frequency. Assuming a 1 ohm impedance level, a sine wave with amplitude A volts has a

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power C  A22 watts. For a receiver with IF noise bandwidth Bn Hz, the noise power at the filter output, assuming white noise at the input, is N0 B watts, where N0 is the single sided noise power spectral density. Hence, the CN ratio at the output of the IF amplifier, measured in a noise bandwidth Bn Hz is C N  1A222  1N0 B2  A2 12N0 Bn 2

(5.7)

S  12 1K¢fpeak 2 2  1K¢frms 2 2 watts

(5.8)

Information is carried in the deviation of the carrier frequency, f, by making f proportional to the modulating signal voltage with a modulator constant 1K. If we assume sine wave modulation at its maximum permissible magnitude, which is the usual way to calculate SN ratios, we will create a peak frequency deviation fpeak. The correct demodulator frequency to voltage conversion constant is K  1kf to provide unity end-to-end gain. Achieving unity gain is useful in any communication system because it helps to establish defined signal levels at the input and output of the communication link. The signal voltage at the output of the FM demodulator is proportional to the frequency deviation. However, to analyze signal to noise ratios we must work in terms of signal power. For analysis purposes, we always assume an impedance of 1 ohm, and convert a sine wave signal V cost volts to an equivalent power of P watts using P  21 V 21  12V 2 watts. Let us assume that the signal is a test tone sine wave of maximum permitted amplitude. The baseband signal power, S, is given by

One characteristic of all FM demodulators operating above threshold is that they suppress the noise at the demodulator output in proportion to the signal strength. Another important characteristic of FM demodulators is that the output noise power spectral density (NPSD) is proportional to the square of the baseband frequency, as illustrated in Figure 5.1. Thus the baseband noise power at the demodulator output must be obtained by integrating the baseband noise power spectral density, N0, over the baseband bandwidth—from the lower edge frequency f1 to the upper edge frequency f2, as shown in Figure 5.1. Hence the noise power present in the baseband at the output of the demodulator is given by4 Nout  2N0 c

K 2 d A



f2

f1

f 2d f  2N0 c

K 2 3 d 1 f 2 f 13 2 3 A

NPSD

NPSD ∝ f 2

0

f max

Frequency, f

FIGURE 5.1 Noise power spectral density at the output of an FM demodulator.

(5.9)

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We will consider only the case where the baseband extends almost to DC, giving f1  0 Hz, so ( f 32 f 31)3  f 323  ( fmax)33, where fmax is the upper end of the baseband frequency range. Combining the expressions for signal power and noise power from Eqs. (5.8) and (5.9) 1SN2 out  1CN2  32  1Bfmax 2  1¢fpeak fmax 2 2

(5.10)

In decibel form, the baseband SN ratio for an FM receiver with test tone modulation is 1SN2 out  C N  10 log10 1BRFfmax 2  20 log10 1 ¢fpeak fmax 2  1.8 dB

(5.11)

where BRF  IF bandwidth of receiver  RF BW of FM signal from Carson’s rule ¢fpeak  peak frequency deviation at transmitter fmax  maximum frequency of baseband signal  receiver baseband bandwidth and the factor of 1.8 dB is equivalent to the numerical ratio 32. The part of the Eq. (5.11) that follows CN is the SN improvement factor of Eq. (5.1). Equation (5.11) gives an important result for FM transmission. Although Eq. (5.11) was derived for a test tone signal, it is applied to any FM link to determine the baseband SN ratio for the particular parameters of the link. The SN ratio calculated from Eq. (5.11) therefore serves as a reference value for the quality of the FM link. The baseband signal-to-noise ratio can be increased well above the CN ratio of the received FM carrier. In satellite communication systems the CN ratio is always low; too low to be converted directly to a baseband signal-to-noise ratio of an output signal. The SN improvement factor can be made large by using a large deviation ratio, D, i.e., D W 1, where D is defined as D  ¢fpeak fmax

(5.12)

However, a large peak frequency deviation ratio results in a wide RF bandwidth. Carson’s rule (Eq. 5.6) shows that BRF  2(fpeak  fmax)  2fmax (D  1). So using a large deviation ratio to secure a large SN improvement causes the occupied RF bandwidth to become much greater than the baseband bandwidth. This is spectrally inefficient, but necessary in almost all analog FM satellite links.

Pre-emphasis and de-emphasis Pre-emphasis and de-emphasis are used with all FM transmissions because it is possible to reduce the noise power at the output of the de-emphasis circuit in the receiver and thus to improve the baseband SN ratio. To understand pre-emphasis and de-emphasis, we must start with the de-emphasis process in the baseband channel of the FM receiver. Equation (5.9) shows that the noise power spectral density at the output of an FM demodulator is proportional to the square of baseband frequency. This is illustrated in Figure 5.1, which shows baseband NPSD at the output of an FM demodulator plotted against baseband frequency. At the high frequency end of the baseband there is much more noise power than at lower frequencies; the noise power is concentrated into the higher frequencies of the baseband. De-emphasis flattens the noise power curve and therefore reduces baseband noise, improving the baseband SN ratio. White noise across the bandwidth of the IF section of the FM receiver (ahead of the demodulator) has equal power at all frequencies—a flat noise power spectral density (NPSD). At the output of the FM demodulator noise power is proportional to the square of the baseband

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frequency. Thus the rms noise voltage Vn, which is equal to the square root of the noise power, is proportional to the baseband frequency fb, as illustrated in Figure 5.2a Vn  aVn rms fb

(5.13)

where Vn rms is the rms noise voltage at the input to the FM demodulator and a is a constant across the IF bandwidth. The function of a de-emphasis circuit is to flatten the NPSD by adding a circuit with a transfer function Gde( f ), proportional to 1fb after the demodulator Gde 1 f 2  b fb

(5.14)

where b is another constant. When the FM demodulator noise output is processed by the de-emphasis circuit the output of the circuit is a voltage Vno where Vno  aVn rms fb  b fb  abVn rms

(5.15)

The value ab is a constant. If it is less than one, we can add amplification of the signal at baseband to compensate. The NPSD at the output of the FM demodulator followed by a de-emphasis circuit is now independent of baseband frequency instead of following a square law relationship. The required gain characteristic of the de-emphasis circuit can be implemented quite easily with a simple R–C circuit consisting of a series resistor–capacitor circuit wired as a low pass filter. The circuit has constant gain up to its corner frequency, fd, and then has the required gain proportional to 1fb, as shown in Figure 5.2b. The de-emphasis circuit is placed immediately after the FM demodulator as illustrated in Figure 5.2c.

Pre-emphasis The de-emphasis circuit in the FM receiver has the characteristics of a low pass filter, which would cut all the high frequency content of the signal. We must add a complementary circuit in the baseband of the transmitter to counteract the effect of the deemphasis circuit in the receiver. This is called a pre-emphasis circuit, and it has a transfer function, Gpe( f ), that is proportional to frequency Gpe 1 f 2 

1 f b b

(5.16)

When a signal is transmitted through the link, the signal is first pre-emphasized at the transmitter and then de-emphasized in the receiver. The transfer function of the system, G( f ), is therefore G1 f 2  Gpe 1 f 2  Gde 1 f 2 

1 b  fb   1 b fb

(5.17)

Thus the signal is unaffected by the process of pre- and de-emphasis, while the noise power in the receiver baseband is significantly reduced. A pre-emphasis circuit consists of two resistors and one capacitor in a high pass configuration, with the same corner frequency fd that is used in the de-emphasis circuit. The pre-emphasis circuit provides a voltage gain that is proportional to baseband frequency above fd, up to the top of the baseband, as illustrated in Figure 5.2d. Although the transfer functions of the pre- and de-emphasis circuits do not exactly match Eqs. (5.14) and (5.16), the low frequency regions where the circuit transfer functions have constant gain is where noise power from the FM demodulator is smallest, so the departure of the circuit characteristics from the ideal has little effect on the improvement in baseband SN ratio achieved with de-emphasis.

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Noise voltage at detector output

Characteristic desired after filtering

Detector alone

Modulating frequency, fm

fd (a)

Transfer function (dB)

fm

fd (b)

FM demodulator

De-emphasis filter

(c)

Transfer function (dB)

fm

fd (d)

FIGURE 5.2 Preemphasis and deemphasis. (a) Noise voltage at FM detector output. (b) De-emphasis filter characteristic. (c) Location of deemphasis filter. (d) Pre-emphasis filter characteristic.

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The baseband SN ratio is increased by a factor P dB with de-emphasis in the receiver. The value of P depends on the nature of the signal, the baseband bandwidth, and the circuits used. P is different for voice, and video signals; typical values are 9 dB for NTSC video and 5–10 dB for analog FM voice transmission. With de-emphasis in the receiver, Eq. (5.11) becomes 1SN2 U  C N  10 log10 1BRFfmax 2  20 log10 1¢fpk fmax 2  1.8  P dB (5.18) The suffix U appended to the (SN) ratio indicates that this value of SN ratio is unweighted. Weighting factors are discussed in the following paragraphs. Pre-emphasis amplifies the high frequencies in the signal more than the low frequencies, thus adding extra power to the baseband signal. With most voice and video signals, the energy in the baseband spectrum at the higher frequencies is low, and little extra power is added to the baseband signal by the pre-emphasis circuit. This is fortunate, because an increase in the overall power in the baseband signal would require a reduction in the frequency deviation to keep within the allotted RF bandwidth. That would reduce the FM demodulator SN improvement and negate part of the overall SN ratio improvement obtained by adding de-emphasis.

5.2 ANALOG FM TRANSMISSION BY SATELLITE For the first 20 years of satellite communications, most signals were in analog form, and were transmitted using frequency modulation. The traffic sent over satellite channels was primarily voice (telephony) with some video signal distribution. Voice channels were multiplexed using frequency division multiplexing (FDM), a technique that had been employed in terrestrial telephone circuits since the 1920s. FDM has largely disappeared from both terrestrial and satellite circuits as digital transmission over optical fibers has replaced analog transmission over microwave and coaxial cable links. FDM remains in use in some countries however, so information relating to FDM telephony and satellite transmissions using FDM/FM is included in Appendix B. The interested reader will find a description of FDM/FM transmission techniques used in Intelsat satellites, and a method for calculation of transponder capacity using FDM/FM, in Appendix B. Prior to the availability of satellites for video signal distribution, hundreds of microwave links were needed to deliver network television signals throughout the United States.

SIDEBAR The distribution of video signals to cable TV companies via geostationary satellites gained importance in the 1970s, and led to the expansion of television networks in the United States. Television program material is generated at many different locations by major networks, and also by smaller companies, at sporting events and from electronic newsgathering vehicles. These signals are distributed very effectively via satellite, because a single uplink station

can send the same signal to thousands of cable service providers across the entirety of North America. There are more than 10,000 individual cable television systems in the United States, each serving a large or small geographic area. Apart from local TV networks which are received over the air with VHF and UHF antennas, virtually all television program material is sent to the cable stations by satellite.

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A substantial investment was needed to create a national TV network, and only three national networks had been created prior to 1970: ABC, CBS, and NBC. The lower cost and wide distribution of video signals by satellite led to the creation of many new networks and TV program channels, which initially were available through cable television systems. The development of direct broadcast satellite television and digital cable systems has encouraged the creation of more television program channels, with over 200 available in the United States alone. Video distribution of signals for cable TV providers used analog frequency modulation from the mid-1970s through the mid-1990s, at C band and Ku band, using a full transponder per video signal. The typical cost of leasing a C-band or Ku-band transponder in the mid-1990s was $1 M per year. Digital transmission was introduced to take advantage of video signal compression techniques, notably MPEG 2, which can reduce the bit rate of a full motion video signal to 6.2 Mbps. Several such digital compressed video signals can fit into one transponder, saving the distribution companies millions of dollars each year. As a result, there has been rapid conversion to digital video transmission, and analog FM transmissions continue (in 2000, in the United States) largely because some cable TV receiving stations are not yet equipped to handle digital signals. Digital transmission techniques are discussed in the second half of this chapter, and digital satellite systems are reviewed in Chapter 10. Television signals have traditionally been generated and distributed in analog form. High definition television (HDTV) is digital, and the signals transmitted by direct broadcast satellites are also digital. It is clear that digital transmission will quickly overtake the remaining analog services, but the distribution of analog video signals via geostationary satellites using frequency modulation remains in place. For that reason, a discussion of analog television by satellite is included here.

Television Signals While a number of television transmission standards exist worldwide, the two in most common use are the North American and Japanese 525 line60 Hz NTSC system and the European 625 line50 Hz PAL system, respectively. In this text we will emphasize the NTSC system. The video signal of a monochrome (black and white) TV transmission carries an analog representation of the brightness (i.e., the amount of white light) in the picture along a series of horizontal scanning lines. This is called the luminance signal. Along with the luminance signal, synchronization pulses are transmitted so that the TV receiver can recreate the scanning process of the camera. Historically, monochrome TV developed before color, and in the United States color TV was designed so that the color information could be added to monochrome transmissions without degrading the performance of existing black-and-white receivers. Any color may be created by an appropriate combination of red, green, and blue light. Color TV could be transmitted by transmitting the color components of each picture separately, but this scheme would require excessive bandwidth. Instead, three linear combinations of the three components are transmitted and the component values themselves are recovered at the receiver. The TV camera generates voltage levels corresponding to the red, green, and blue light at each point in the picture. We will identify these voltage levels by the letters R, G, and B. A monochrome receiver responds to the amount of white light at a point in the picture; this is the luminance, Y, and is related to the color voltage levels by Y  0.30R  0.59G  0.11B

(5.19)

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The luminance signal is transmitted so that monochrome receivers can receive a color image in black and white. For color reconstruction, two other independent linear combinations of R, G, and B must be transmitted along with Y so that all of the color components can be recovered. These are called the I and Q signals, given by I  0.60R 0.29G 0.32B Q  0.21R 0.52G  0.31B

(5.20) (5.21)

The letters I and Q stand for in-phase and quadrature, and together the I and Q signals (when decoded with the luminance signal) carry the chrominance information about the color at each point in the picture. The I and Q signals modulate a color (or chrominance) subcarrier in such a way that the amplitude of the resulting chrominance signal determines the saturation (degree of purity) of the color at a point and the phase of the chrominance signal determines the hue (perceived shade) of the color. From the amplitude and phase of the chrominance signal, a TV receiver determines the shade of the color and the amount of white light to add. From the luminance signal it determines how bright the color should be. In terrestrial broadcasting the luminance (Y) signal, filtered to occupy the band from 0 to 4.2 MHz, modulates the video carrier with a vestigial sideband (VSB) modulator. The upper sideband is transmitted in full; the lower sideband is partially removed. The resulting VSB signal is all that needs to be transmitted for the video portion of monochrome television. The chrominance information is transmitted by a color subcarrier at 3.579545 MHz (hereafter abbreviated as 3.58 MHz). This value was chosen because it places the chrominance signal at a relatively empty part of the luminance spectrum and minimizes color interference with black-and-white reception3. Both the I and Q signals modulate the color subcarrier through double-balanced mixers to generate double sideband suppressed carrier (DSBSC) signals. The subcarrier is phase shifted by 90 before it enters the Q modulator. Thus, both I and Q components may be recovered at the receiver. Figure 5.3 shows the spectrum of the baseband video signal. The baseband audio signal extends from 50 Hz to 15 kHz. It frequency modulates an audio subcarrier and the resulting FM waveform is added to the video baseband signal. This leads to the composite TV signal of Figure 5.3b; it consists of the baseband video signal below an FM modulated audio subcarrier. In U.S. domestic systems transmitting analog television signals using FM, an audio subcarrier frequency of 6.8 MHz is standard; 6.2 MHz is also used. In terrestrial broadcasting, the audio and video signals are combined and shifted in frequency to an appropriate part of the VHF or UHF band for transmission. The radiated signal is a complex combination of FM (the sound), VSB (the luminance), and quadrature DSBSC (the chrominance) signals occupying a 6-MHz bandwidth. For satellite transmission the baseband video signal (luminance and chrominance) frequency modulates a video carrier and two stereo audio signals frequency modulate two audio carriers. The details of the video modulation depend on the transponder bandwidth available. Typical values for network TV are a peak deviation fpk of 10.75 MHz and a maximum video modulating frequency, fmax, of 4.2 MHz. By Carson’s rule (Eq. 5.6) this requires a transponder bandwidth of 29.9 MHz. FM television signals are often overdeviated, trading the larger improvement in video (SN) that results for the smaller degradation in picture quality associated with truncating some of the sidebands5.

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Luminance signal

167

Color subcarrier at 3.579545 MHz

Chrominance signal

0

2

4

f

6

Baseband frequency, MHz (a)

Audio subcarrier at 6.8 MHz

Video

Audio

0

2

4

6

f

Baseband frequency, MHz (b) FIGURE 5.3 Spectra of baseband TV signals (a) Baseband video signal. (b) The composite (video plus audio) TV signal as transmitted by U.S. domestic satellites.

A TV signal from a satellite is quite different from a broadcast TV signal. Converters (sometimes called set-top boxes) that allow reception of satellite television transmission on conventional home TV receivers must demodulate the incoming FM signals, recover the baseband video and audio channels, and send separate video and audio signals to the TV receiver. Alternatively, the baseband TV signal can be remodulated into the VSB format of broadcast TV transmissions using a locally generated carrier (usually VHF TV channel 3 or 4).

SN Ratios for FM Video Transmission The SN ratio in the baseband channel of an analog FM satellite TV receiver is given by Eq. (5.18) with one additional factor, Q. 1S  N2 W  C  N  10 log10 1BRFfmax 2  20 log10 1¢fpk fmax 2  1.8  P  Q dB (5.22) The factor Q, called a subjective improvement or weighting factor, combines two effects that improve the apparent quality of a television picture when transmitted by FM. The suffix W appended to the SN ratio indicates that a subjective weighting has been applied. This reflects the nature of the video signal and the eye’s response to the on-screen display. Video signals usually vary between 0 V for maximum brightness of the TV screen, and 1.0 V for full black. The calculation of SN ratio is based on sine wave signals, which

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gives a pessimistic result for the video signal. The second effect relates to the nature of noise from a FM receiver when it appears on a TV screen. Because the FM demodulator has low baseband noise power at low frequencies, the appearance of FM noise on a TV screen is less annoying than white noise. The SN ratio given by Eq. (5.18) has to be increased by the subjective factor Q to account for these effects. A typical value for Q in a NTSC TV receiver is 8 dB. EXAMPLE 5.2.1 A standard NTSC signal has a baseband video bandwidth of 4.2 MHz and is transmitted over the satellite link in an RF bandwidth of 30 MHz using frequency modulation and standard preemphasis and de-emphasis. At the receiving earth station the CN ratio in clear air conditions is 15 dB. Calculate the baseband SN ratio for the video signal. Assume a de-emphasis improvement of 9 dB and a subjective improvement factor of 8 dB in the baseband signal to noise ratio. The first step in this calculation is to use Carson’s rule (Eq. 5.6) to find the peak frequency deviation. BRF  21¢fpk  fmax 2 Hz Hence ¢fpk  BRF2 fmax  15 4.2  10.8 MHz Now we can evaluate Eq. (5.22) to find the baseband video SN ratio SN  CN  10 log10 1BRFfmax 2  20 log10 1¢fpkfmax 2  1.8  P  Q dB  15  10 log10 1304.22  20 log10 110.84.22  1.8  9  8 dB  15  8.5  8.2  18.8 dB  50.5 dB The FM improvement here is 35.5 dB including the 8 dB subjective weighting. The unweighted improvement is 27.5 dB. Figure 4.9 in Chapter 4 shows the unweighted (SN )out versus (CN )in of an FM demodulator designed for FM satellite TV using the NTSC standard. The unweighted SN improvement is the difference between (SN )out and (CN )in when the input signal is above the 8-dB threshold. Reference to Figure 4.9 shows an unweighted improvement value of 28 dB. These values are typical of cable TV industry systems using analog FM transmission of video signals. A baseband video SN ratio of 50 dB is considered to produce a very good quality signal. The ratings for video signals generally follows these standards: SN  55 dB No perceptible noise, studio quality signal SN  50 dB Very good quality signal, noise is just perceptible in background SN  45 dB Good quality signal, some noise is visible but not annoying SN  40 dB Poor quality signal, with a lot of noise visible SN  35 dB Bottom limit for picture quality, high level of noise, very annoying The cable TV industry uses video SN  45 dB as the lower limit for video signals delivered by cable to customers’ homes. Satellite links delivering video signals to cable head-ends need to provide a baseband SN ratio that is well above 45 dB to allow for degradation in the distribution network. 

FM Threshold All FM demodulators exhibit a threshold input CN value below which the SN improvement degrades. The degradation is caused by noise spikes that start to appear at the

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output of a conventional FM demodulator when the input CN ratio is below about 13 dB. Analog FM satellite TV became economically feasible only with the development of threshold extension demodulators that lowered the FM threshold to 9 dB, or lower. On a TV screen, the noise spikes appear as sparklies, which are bright white dots and bars. A very simplified explanation of FM threshold follows. When the magnitude of the rms noise voltage in the IF amplifier approaches onequarter of the magnitude of the signal, occasional peaks of the noise will be equal to or larger than the signal because of the random fluctuations of the noise voltage. As the carrier power is reduced and noise becomes relatively stronger, the probability that an instantaneous noise voltage exceeds the signal magnitude becomes more likely. If the phase of the noise, which is also random, is 180° from the phase of the signal, small changes in phase angle of either the signal or the noise around 180° cause the sum of the signal and noise to execute very rapid phase changes. The FM demodulator responds to changes in frequency, which are rates of change of phase. The larger the frequency deviation, and therefore the larger the rate at which the signal phase angle is changing, the bigger the noise output from the FM demodulator. Very rapid changes of phase of the signal at the FM demodulator input cause very large noise spikes at the demodulator output. The noise spikes are heard as clicks in an FM radio, and seen as sparklies when viewing FM video. Threshold for an FM demodulator is defined as the value of (CN)in at which there is a loss of 1 dB in SN improvement. This point is at (CN)in  8.5 dB in Figure 4.9. The rate at which noise spikes appear at the demodulator output increases rapidly as the CN ratio at the demodulator input is reduced. For a conventional FM demodulator, threshold occurs around CN  13 dB. One of the key developments in the early 1980s that made home satellite TV systems possible was the threshold extension FM demodulator. These demodulators have a threshold 4 to 5 dB lower than conventional FM demodulators, which translates directly to the possibility of operating at a CN ratio 4 or 5 dB lower than could be achieved with a conventional demodulator. For information on threshold extension techniques in FM demodulators, the reader is referred to reference 6.

SCPC FM Links FM is not widely used for voice links now, having been superseded by digital modulations. However, analog single channel per carrier (SCPC) links have been quite widely used for voice and have also been used for data transmission by the amateur satellite community. In an SCPC system, each earth station transmits at an assigned RF frequency. In demand access (DA) systems, the frequency is allocated to the station at the time it requests a channel. Single channel voice systems work well with wideband FM, and data transmission can be achieved with transmitters designed for voice applications. SCPC-FM voice systems obtain improvement in SN ratio in exactly the same way as FM-TV. There is no subjective or weighting improvement though, unless companding is used (Companding is a contraction of compressing and expanding, a process that decreases the dynamic range of the input signal with a nonlinear amplifier (compression) at the transmitter and compensates with a complimentary nonlinear amplifier (expander) at the receiver. The reduced dynamic range of the signal allows it to be transmitted at a higher average level, which improves the signal-to-noise ratio at the receiver. The -law and A-law compression circuits used in digital telephony are typical companding operations.)

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For a linear system without companding, baseband SN ratio is found from Eq. (5.18) 1SN2 FM voice  C N  10 log10 1BRFfmax 2  20 log10 1 ¢fpk fmax 2  1.8  P dB Pre-emphasis and de-emphasis can be applied to FM voice signals with a typical subjective improvement of 7 or 8 dB. Threshold for narrow band threshold extension FM demodulators can be as low as 5 dB. Example 5.2.2 illustrates the performance of a typical FM satellite voice link. The baseband bandwidth of 3.4 kHz corresponds to a telephone quality voice channel. The RF channel bandwidth of 45 kHz is also typical.

EXAMPLE 5.2.2 A SCPC-FM satellite link has an RF channel bandwidth of 45 kHz and a baseband maximum frequency of 3.4 kHz. De-emphasis provides a subjective improvement in baseband SN ratio of 7 dB. Calculate the baseband SN ratio for the voice channel for a receiver CN ratio of 13 dB. If the FM demodulator has an FM threshold at 6 dB, what is the link margin for this system? We must first find the peak frequency deviation for the FM wave. Using the same approach as in Example 5.2.1 ¢fpk  BRF2 fmax  22.5 3.4  19.1 kHz The baseband voice channel SN is given by Eq. (5.18) 1SN2 FM voice  CN  10 log10 1BRFfmax 2  20 log10 1¢fpk fmax 2  1.8  P dB  13  10 log10 145 3.42  20 log10 119.23.42  1.8  P dB  13  11.2  15.0  8.8  48 dB

This is an acceptable voice channel SN ratio. If the satellite link suffers attenuation in the slant path, for example from heavy rain, the system is considered operational until the CN value falls to the FM threshold value, at 6 dB. The link margin is therefore 13 6  7 dB. With CN  6 dB, the FM demodulator is operating at threshold, so the voice channel SN ratio is reduced by 7  1  8 dB, to 40 dB. This is an acceptable lower value for a voice channel. Noise would be heard, but not at an annoying level. The 1 dB additional loss in SN ratio is because the FM demodulator is at threshold. 

Data Transmission Using Analog FM Channels Data signals can be sent over an analog FM channel in much the same way as voice signals. Section 5.3 discusses the transmission of digital signals over radio links. It is shown that the key to successful transmission of data signals is to create the correct baseband waveform at the output of the radio receiver, which also means creating the correct waveform at the input to the transmitter. The received waveform is sampled and thresholded to yield a digital signal identical to that transmitted, so there are no bit errors. The baseband waveforms used in digital radio transmissions are called zero ISI waveforms (see Section 5.3 for a description), so the waveform at the transmitter input is an analog signal, and must be sent through a linear transmission system to avoid distortion of the waveform, which could lead to bit errors. The theoretical baseband CN ratio of a binary digital waveform must be at least 10 dB to keep bit errors at an acceptable level. An implementation margin of between 0.5 and 2 dB has to be added to the theoretical figure to account for waveform distortion and nonideal filters, etc., giving a typical minimum required SN ratio of about 12 dB. The RF channel bandwidth for FM voice transmissions in the VHF and UHF amateur bands

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SIDEBAR Transmitters designed to carry FM voice signals can be used to carry digital waveforms. Since an FM transmitter–receiver link is essentially linear, it is ideal for carrying zero ISI waveforms, and can operate at relatively low CN ratios. This technique has been developed very successfully by the amateur satellite radio community for the relay of data signals over LEO amateur satellites8. Data signals are generated by a personal computer, and then sent to a terminal node controller (TNC). The TNC packetizes the data and creates the required baseband

waveform to drive the FM modulator of a voice channel transmitter. Bit rates of 4.8 and 9.6 kbps can be sent over an RF channel with 15 kHz bandwidth. Some of the amateur satellites operate in the VHF and UHF amateur bands and employ store-and-forward techniques. Data received at the satellite are stored on board in memory until another station requests a download. All amateur radio transmissions are required by law to be sent in the clear, not encrypted, and can be downloaded by any suitably equipped earth station.

is only 15 kHz, so narrow band FM must be used. This results in the CN ratio at the output of the FM receiver being numerically similar to the IF CN ratio—there is little FM improvement. However, if the received signal from the satellite is above the FM threshold of the receiver, the baseband SN will be high enough to ensure a small number of errors in the recovered data stream. The RF signal created by the FM transmitter is classified as FSK, but the waveform is an example of continuous phase FSK (CPFSK). CPFSK changes the phase of the RF waveform continuously, rather than abruptly as in classic FSK. The choice of the peak frequency deviation in a CPFSK signal is related to the bit rate, and some choices lead to another form of digital modulation, minimum shift keying (MSK). CPFSK and MSK are both digital modulations that have good spectral efficiency and low error rates for a given CN ratio. Example 5.2.3 shows how a low data rate digital satellite link using CPFSK might be implemented.

EXAMPLE 5.2.3 Let’s examine a system that sends a digital data signal at a bit rate of 9.6 kbps in a nominal RF channel bandwidth of 15 kHz. With near-ideal Nyquist filters, which can be implemented with digital signal processing (DSP) techniques in the TNC, the bandwidth of the baseband digital waveform can be restricted to 0.5  bit rate. For a 9.6 kbps data stream, the baseband bandwidth is 4.8 kHz. If we use a peak frequency deviation of 3.6 kHz, the resulting RF signal bandwidth can be found from Carson’s rule (Eq. 5.06) BRF  2  1¢fpk  fmax 2

 2  13.6  4.82  16.8 kHz

Sending this RF signal through band-pass filters designed for 15 kHz FM analog transmissions results in little distortion of the recovered waveform. We can find the FM improvement for the baseband channel waveform by applying Eq. (5.11), assuming no de-emphasis for the moment 1SN2 out  CN  10 log10 1BRFfmax 2  20 log10 1¢fpeakfmax 2  1.8 dB  CN  6.2 2.5  1.8  CN  5.5 dB Thus, if the CN ratio for the signal from the satellite is 10 dB, the baseband waveform has SN  15.5 dB, which is sufficient to ensure no bit errors in the recovered digital signal, even allowing for an implementation margin to cover distortion of the waveform and nonideal filters. 

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MODULATION AND MULTIPLEXING TECHNIQUES FOR SATELLITE LINKS

Digital modulation is the obvious choice for satellite transmission of signals that originate in digital form and that are used by digital devices. Virtually all signals sent via satellites are now digital. Familiar examples are data transmissions to and from the Internet, communications between remote terminals and computers, digital telephony, and TV signals in digital form, such as HDTV and DBS-TV. Digital transmission lends itself naturally to time division multiplexing (TDM) and time division multiple access (TDMA). Analog signals that are transmitted digitally can share channels with digital data, since all digital signals are handled in the same way, and their content is immaterial. Thus a digital satellite link can carry a mix of telephone and data signals that varies with traffic demand. This section contains a review of digital transmission techniques. All digital links are designed in much the same way, using a specific symbol rate, and specific filters that minimize intersymbol interference (ISI). A symbol in a baseband link is a pulse of current or voltage. In a satellite link, a symbol is almost always a phase state (BPSK and QPSK) or a phase and amplitude state (QAM). Digital links are designed for a specific symbol rate, but one symbol can carry more than one bit. It is important to distinguish between symbols and bits. They are easily confused because with binary modulations, such as BPSK, the symbol rate and bit rate are the same. Symbol rates are given in baud or in symbols per second, abbreviated to sps. We will use symbols per second in the analysis that follows because the difference between symbols per second and bits per second is then more obvious, with the comment that the baud as the unit of transmission rate equal to symbols per second is still in widespread use. The name of the unit, the baud, is derived from the name Baudot, who was an early French pioneer of the telegraph.

Baseband Digital Signals We will represent baseband digital signals as serially transmitted logical ones and zeroes. While in computer circuitry a logical zero may be represented by a low voltage (nominally zero) and a logical one may be represented by a high voltage (e.g., 5 V), this arrangement is inconvenient for transmission over any significant distance and is not used. To understand why, imagine a transmission line carrying a bit stream encoded this way and containing approximately equal numbers of ones and zeroes. About half the time the line voltage will be 5 V and about half the time it will be 0 V; hence the line signal will have a 2.5 V DC component. All circuits that carry this signal must have a frequency response that extends to DC, and this is difficult to achieve since many communication circuits contain transformers. To avoid this problem, digital modulators usually accept their input in a polar non-return-to-zero (NRZ) format: logical ones and zeroes are transmitted as plus or minus a stated value. Thus a logical one might be transmitted as 1 V and a logical zero might be transmitted as 1 V. Zero volts is not transmitted except as a transient value. Throughout this text we will assume a polar NRZ format for data signals unless we explicitly state otherwise.

Baseband Transmission of Digital Data Satellite links always carry RF signals, which requires that data be modulated onto a radio frequency carrier for transmission. However, to provide a better understanding of the way in which digital transmission systems are designed, we will begin by examining the case of a

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baseband data link. In a baseband link, the frequency response of the link is assumed to extend from DC to an upper limit fmax, where fmax is equal to the bandwidth of the link, B Hz. Data is transmitted in the form of polar pulses; in a binary system, the pulses have amplitudes V and V volts, where V can take any value. As mentioned earlier, the average number of V and V pulses is made equal so that the average DC voltage on the transmission line is zero. We begin the analysis of baseband links by determining the conditions under which intersymbol interference can be minimized. The numerical results for bandwidth and symbol rate in this section do not correspond to the requirements for a satellite link; they refer only to a baseband link using a transmission line. A random sequence of rectangular binary pulses has a power spectral density G1 f 2  Ts c

sin1pf Ts 2 2 d pf Ts

(5.23)

where Ts is the duration of the pulse7. This spectrum is illustrated in Figure 5.4d. The familiar sin xx (also called sinc x) spectrum shows that energy exists at all frequencies; to retain the rectangular pulse shape would require an infinite transmission bandwidth. Practical communication systems are always bandwidth limited. Not only is infinite bandwidth not available, interference considerations in radio links dictate that a communication system should use the smallest possible bandwidth, and this is usually one of the design criteria of a communication system. In any digital communication system, a symbol is defined by the rate at which information is sent over the link, in the form of pulses at baseband, or changes in phase angle of a carrier, for example, in a PSK system. Our discussion of digital transmission systems will be based on symbols, rather than bits, because we often want to send more than one bit per symbol in an RF system to conserve bandwidth. Popular modulations that transmit more than one bit per symbol are QPSK (two bits/symbol) and QAM (up to 10 bits/symbol). QPSK is widely used on satellite links, including direct broadcast satellite television. High-speed modems designed for telephone lines use QAM to send a high bit rate in a small bandwidth (e.g., 28.8 kbps in 4 kHz bandwidth). Nyquist’s criterion for zero ISI, which forms the design basis for every digital transmission system, is based on the use of square root raised cosine (RRC) filters and a specified symbol rate on the link. If the transmission is binary, the symbol is a bit, and the symbol has two states. When two bits are sent per symbol, the symbols have four possible states and the system is denoted as quaternary, hence the Q in QPSK. If a symbol represents more than one bit, the system is known generically as m-ary, with one symbol having m states. QAM is a modulation that combines the four phase states of QPSK with multiple pulse amplitudes as in m-ASK. For example, 256-QAM is a modulation in which each symbol represents 8 bits and has 256 possible states. There are four phase states and pulses can have 26  64 possible amplitude levels. If we take the random pulse train shown in Figure 5.4a and band limit it by passing the pulses through a low pass filter, the pulse shape will be altered. As an example, consider the effect of passing the rectangular pulse train through a single RC section, representing a very simple low pass filter. The resulting waveform, shown in Figure 5.4b, has been delayed and pulses are smeared in time—the decaying pulse from one transition extends into the next pulse interval. Where the pulse pattern is 10 or 01, the amplitude of the second pulse at the sampling instant shown in Figure 5.4 has been reduced by the presence of a delayed portion of the preceding pulse. This is called intersymbol interference (ISI) and is likely to occur whenever a digital signal is passed through a band-limiting filter. When noise is added to the waveform, ISI increases the likelihood that the receiver will detect a bit incorrectly, causing a bit error. In a baseband system, ISI can be avoided

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v1(t )

t

0

(a )

v2(t )

t

0

sampling

Ts

2Ts

3Ts

4Ts

instants

5Ts

(b ) log |H (f )|

R

v1(t )

C v2(t )

Ts = RC f0 =

f

f0

1 2πRC (c ) 1 Before filtering After filter

−4/Ts −3/Ts −2/Ts −1/Ts

0

1/Ts

2/Ts

3/Ts

4/Ts

f

(d ) FIGURE 5.4 Illustration of the effect of low pass filtering on an NRZ signal. (a) Random NRZ polar pulse train. (b) Waveform output from an RC filter with Ts  RC. (c) RC filter and its transfer function 0H(f )0. (d) Spectrum of bandlimited NRZ pulse train.

by an appropriate choice of low pass filter. Nyquist9 proposed a technique that can theoretically produce zero ISI, now known as the Nyquist criterion. The objective is to create in the receiver a pulse that resembles the sin xx shape, crossing the axis at intervals of Ts, where Ts, is the symbol period. The receiver samples the incoming wave at intervals of Ts, as shown in Figure 5.5, so that at the instant one pulse is sampled, the “tails” from all preceding pulses have zero value. Thus previous pulses cause zero intersymbol interference (zero ISI) at each sampling instant. Filters that produce the required zero ISI waveform in the receiver can be realized in several ways. The baseband transfer function proposed by Nyquist was the raised cosine

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Data Impulse data stream

1

1

1

175

0

t

Ts

Individual pulse waveforms

t

Vt (t ) Baseband signal transmitted

t Vr (t ) Noisy received signal

t

Sampling instants

t

Ts Recovered NRZ pulses

t Data

1

FIGURE 5.5

1

1

0

Transmission and reception of baseband zero-ISI pulses.

function, VNQ( f ), which has a normalized two-sided frequency characteristic given by VNQ 1 f 2  1

for 0 f 0 6

Rs 11 a2 2 Rs p VNQ 1 f 2  cos2 e c 0 f 0 11 a2 d f 2aRs 2 Rs Rs for 11 a2  f  11  a2 2 2 Rs VNQ 1 f 2  0 11  a2 for 0 f 0 7 2

(5.24)

where 0  1 and Rs  1Ts is the symbol rate in symbols per second. (Some texts use the symbol r instead of .) The entire communication link must have this transfer

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function to ensure zero ISI. The pulse shape generated at the output of the link is vNQ(t), the required zero ISI waveform, when the filter input is driven by an impulse, (t). The waveform vNQ (t) is obtained as the inverse Fourier transform of the output from the Nyquist raised cosine transfer function, which is simply the spectrum of the input pulse multiplied by the frequency response of the system. vNQ 1t2  F 1 3VNQ 1 f 2  S1 f 2 4

(5.25)

where F 1[ ] indicates the inverse Fourier transform and S( f ) is the spectrum of the input pulse. If we use an impulse (t) as the input signal, the input signal spectrum is S( f )  1, which is referred to as a flat spectrum, and then vNQ 1t2  F 1 3VNQ 1 f 2 4

(5.26)

The raised cosine transfer function with VNQ( f ) given by Eq. (5.24) has an impulse response vNQ(t) given by vNQ 1t2  c

cos 1paRst2 sin 1pRs t2 d  c d 11 2aRs t2 pRs t

(5.27)

Figure 5.6 shows the shape of several raised cosine function characteristics for values of  between 0 and 1, and the corresponding waveforms generated by the impulse response of these filters. Note that the raised cosine function shown in Figure 5.6 is a voltage transfer function, and that all functions have a value VNQ( f )  0.5 at a baseband frequency f  Rs2. The case of   0 in Eq. (5.24) yields a rectangular function with a bandwidth of Rs2. This is the minimum bandwidth through which a signal with a symbol rate Rs can be transmitted while still satisfying the zero ISI condition. Such a function is not realizable in practice, since we cannot have an infinitely rapid attenuation slope at one frequency. In fact, none of the Nyquist raised cosine functions can be created in practice. The requirement in Eq. (5.24) that there be zero transmission above a frequency Rs2  (1  ) cannot be met with any real circuit. Consequently, all digital transmission systems can only approximate the ideal Nyquist transfer function, and will therefore always generate some intersymbol interference. However, the basis for the design of all digital links is the ideal zero ISI Nyquist criterion. Real filters that give the link a transfer function that approximates the ideal Nyquist zero ISI transfer function will minimize ISI and maximize symbol rate. The reader is reminded again that these results apply to a baseband link, not to a satellite link. Implementation of a link with a Nyquist transfer function requires specific parts of the link to have different transfer functions. There are three separate parts to any communication system: the transmitter, the transmission link, and the receiver. These three parts are in series, so the overall system transfer function is the product of the three individual transfer functions. We will denote their transfer functions as Ht ( f ) for the transmitter, L( f ) for the transmission link, and Hr( f ) for the receiver. We want the output of the receiver to be a zero ISI waveform, which we achieve by creating a zero ISI spectrum VNQ( f ) at the receiver output. The spectrum of the waveform at the output of the receiver is given by Vr 1 f 2  S1 f 2  Ht 1 f 2  L1 f 2  Hr 1 f 2

(5.28)

where S( f ) is the spectrum of the signal at the input of the transmitter. We will specify that S( f )  1, corresponding to an input consisting of delta functions (t) or (t) representing logical ones and zeroes. We will also specify that the transfer function of the

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Vr (f ) 1

α = 0.5 α=0

α=1

0

Rs /2

0

3Rs /4

f

R

Frequency (a)

vr (t )

Voltage

α=0

α = 0.5

α=1

0

Ts

2Ts

3Ts

t

Time (b) FIGURE 5.6 Raised cosine filter frequency characteristic and impulse response. (a) Raised cosine function transfer characteristics. (b) Corresponding impulse responses.

link must be flat, such that 0L1 f 2 0  1 and that the phase response of the link is linear with frequency. With these conditions in place, we want the end-to-end transfer function of the link to be a Nyquist zero ISI raised cosine transfer function. Hence or

Vr 1 f 2  S1 f 2  Ht 1 f 2  L1 f 2  Hr 1 f 2  1  Ht 1 f 2  1  Hr 1 f 2  VNQ 1 f 2 (5.29) VNQ 1 f 2  Ht 1 f 2  Hr 1 f 2

(5.30)

Equation (5.30) is an important result in the design of digital communication links. It states that the transfer functions of the transmitter and receiver when multiplied together must equal the Nyquist raised cosine transfer function. One obvious way to

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achieve this result is to make the transfer functions of the transmitter and receiver identical, so that Ht 1 f 2  Hr 1 f 2  1VNQ 1 f 2

(5.31)

A filter with a transfer function equal to the square root of a raised cosine function is called a square root raised cosine filter or often just a root raised cosine (RRC) filter. RRC filters are used as the basis for the design of most digital communication links, even though no such filters actually exist. Real filters (such as Butterworth, Chebychev, Elliptic function) can only approximate the RRC filter’s transfer function. Using identical filters in the transmitter and receiver satisfies another desired criterion in communication systems: the matched filter criterion1,4. A communication link achieves the best possible SN ratio at the receiver when the spectrum of the signal at the receiver input is a replica of the transfer function of the receiver. The easiest way to achieve this condition is with identical (matched) filters in the transmitter and receiver. The resulting structure for a baseband digital communication link is shown in Figure 5.7, together with the corresponding waveforms and spectra. Because the RRC filters in the transmitter and receiver have zero transmission above the frequency fmax  Rs2  (1  ),

Transmitter

v 1( t )

Receiver

v 2( t )

v 3( t )

v 4( t ) Link

NRZ data Impulse generator

Square root raised cosine filter

Zero ISI waveform Square root raised cosine filter

(a )

v 1( t )

v1(f )

NRZ pulse

0

t

Ts

v 2( t )

2/Ts

Td Td + 2Ts

Td + 4Ts

f v3(f )

Receiver O/P zero ISI

t

f

Impulse spectrum

t

0

v 4( t )

1/Ts v2(f )

Impulse

NRZ spectrum

Transmitter O/P spectrum

f

1/T

s

Time

Frequency

(b )

(c )

FIGURE 5.7 Waveforms and spectra in a baseband data system with raised cosine filters. (a) System block diagram. (b) Waveforms. (c) Spectra.

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the requirement that the link transfer function have a flat magnitude and linear phase extends only up to fmax. Similarly, the delta functions at the input can be narrow rectangular pulses with a sin xx spectrum that has a first null well above fmax. A useful rule of thumb is that a pulse of duration Ts  110 fmax is sufficiently short that its spectrum approximates to a delta function.

Band-pass Transmission of Digital Data In a radio frequency communication system that transmits digital data, a parameter of the RF wave must be varied, or modulated, to carry the baseband information. The most popular choice of modulation for a digital satellite communication system is phase shift keying (PSK), as described in the following section. Band-pass (or radio frequency) transmission of digital data differs from baseband transmission only because modulation of an RF wave is required: the receiver demodulates the modulated RF wave to recover the baseband data stream. Thus intersymbol interference will occur at the receiver due to band limiting of the modulated waveform, unless filters that satisfy the Nyquist criterion are used. An additional constraint usually exists with radio communication systems. The bandwidth occupied by any radio transmission is specified to avoid interference with other transmissions at adjacent frequencies. The output of a transmitter must have a carefully controlled spectrum that reduces out-of-band signals to a low level. Figure 5.8 shows the

Modulator

v 1( t )

Receiver

Transmitter

v 2( t )

v 3( t )

v 4( t )

v 5( t )

v 6( t )

v 7( t )

Eq. Link BPF

Equalizer x sin x

fc

LPF

fc (a )

v 1( t )

v2(f )

NRZ pulse

0

t

Ts

v 2( t )

fc − Rs

t v4(f )

Receiver O/P zero ISI

Td

Td + 2Ts (b )

t Td + 4Ts

fc

v3(f )

PSK wave

v 7( t )

PSK spectrum

fc + Rs

f

BPF O/P spectrum Rs = 1/Ts

fc − Rs

fc

fc − Rs

fc

f fc + Rs Transmitted PSK spectrum

fc + Rs

f

(c )

FIGURE 5.8 Waveforms and spectra in a PSK data system with raised cosine filters and xsin x equalization. (a) Block diagram of one channel of a QPSK system. (Equivalent to a BPSK system.) (b) Waveforms. (c) Spectra.

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spectrum of a binary PSK (BPSK) signal generated from a random train of binary digits. The slow decay of the spectrum beyond fc  Rs results from the sudden phase reversals of the PSK waveform. The Nyquist filters used in a radio link must be band-pass filters, centered at the carrier frequency of the RF signal. The single-sided transfer function of a band-pass RRC filter is identical to the two sided baseband frequency response of the equivalent baseband filter, with its center frequency shifted from 0 Hz to the carrier frequency fc Hz. Thus an important difference between baseband RRC filters and band-pass (RF) RRC filters is that the RF version has a bandwidth twice that of the baseband filter. In radio transmitters and receivers, RRC band-pass filters are always implemented in the intermediate frequency (IF) section of the transmitter or receiver, rather than the much higher radio frequency section. At the transmitter, the RRC filter limits the bandwidth of the transmitted baseband signal to Bocc  Rs(1  ), where Bocc is called the occupied bandwidth of the transmitted signal, and Rs is the symbol rate. (Some authors call this the absolute bandwidth of the signal1.) At the receiver, the RRC filter limits the noise that can reach the receiver output to a noise bandwidth of Bn  Rs. Note the very important distinction here: the signal occupies a bandwidth Bocc Hz, but the noise bandwidth of every band-pass RRC filter is Rs Hz. It is common in electronic circuits to think in terms of 3 dB bandwidth, but the analysis of radio communication systems requires knowledge of two different bandwidths: the total bandwidth required to carry the radio signal, and the noise bandwidth of the receiver. The total bandwidth, which is the occupied bandwidth, Bocc, when Nyquist filters are used in the transmitter, defines the RF spectrum required to transmit all of the signal energy. It is often referred to as channel bandwidth, because we can think in terms of establishing a radio channel with bandwidth Bocc. Hence for the case of a satellite communication system with a symbol rate Rs sps, the required RRC filters are identical band-pass filters which have the following bandwidths: At the transmitter, the RRC filter creates a signal with an occupied bandwidth Bocc  Rs 11  a2 Hz

(5.32)

At the receiver, the noise bandwidth of the RRC filter is Bn  Rs Hz

(5.33)

Every RRC band-pass filter, regardless of the value of the roll-off factor  (also given the symbol r in some texts), has a noise bandwidth equal to the symbol rate of the link. This is the most important design features of digital radio links, and must be observed whenever noise power is calculated in a digital radio receiver. Every RRC band-pass filter in a digital radio transmitter generates a radio signal with a bandwidth Rs(1  ) Hz, which is always greater than the numerical value of the symbol rate since   0. As stated above, the band-pass filters in the transmitter and receiver are usually implemented at an intermediate frequency fIF. The frequency response of the band-pass RRC filters therefore extends from a lower limit fIF  R2s 11  a2 to an upper limit fIF  Rs 2 11  a2 . The corresponding frequency range of the RF signal is the occupied bandwidth Bocc, centered on the RF carrier frequency fc Bocc  c fc 

Rs Rs 11  a2 d  c fc  11  a2 d 2 2

(5.34)

Throughout the radio link, every RF and IF component must have a flat frequency response (and linear phase characteristics) over the bandwidth occupied by the signal. If

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any part of the link does not meet the frequency response requirement, an equalizer must be added in series with that part to force the required frequency response. EXAMPLE 5.3.1 A satellite link has an RF channel with a bandwidth 1.0 MHz. The transmitter and receiver have RRC filters with   0.5. What is correct symbol rate (pulse rate) for this link? The relationship between symbol rate and bandwidth is given by Eq. (5.32) Bocc  Rs 11  a2 Hz 106  Rs 11  0.52  1.5 Rs Rs  1061.5  666.7 ksps



EXAMPLE 5.3.2 A Ku-band satellite uplink has a carrier frequency of 14.125 MHz and carries a symbol stream at Rs  16 Msps. The transmitter and receiver have RRC filters with   0.25. What is bandwidth occupied by RF signal, and what is the frequency range of the transmitted RF signal? From Eq. (5.32) Bocc  Rs 11  0.252  1.25 Rs

 1.25  16  106  20 MHz

The RF signal occupies the frequency range given by Eq. (5.34) fc 

Rs Rs 11  a2 fRF fc  11  a2 2 2

Hence, the frequency band occupied by the uplink signal extends from 14.125  0.01  14.115 GHz to 14.125  0.01  14.135 GHz



In many data transmission systems the baseband waveform at the transmitter input has an NRZ format. A link with a Nyquist transfer function will produce a zero ISI waveform at the receiver output only when driven by an impulse, as shown by Eqs. (5.28) and (5.29). If the transmitter is driven by an NRZ waveform with a symbol rate Rs, the spectrum of the driving pulse has a shape S1 f 2 

sin1pTs f 2 pTs f

(5.35)

and the spectrum of the output of the RRC filter will be Vt 1 f 2  1VNQ 1 f 2 c

sin1pTs f 2 d pTs f

To obtain zero ISI at the receiver, we must force the spectrum of the signal from the transmitter to be 1VNQ 1 f 2, which can be achieved by using an equalizer in the pTs f transmitter with a transfer function given by Et 1 f 2  c d . Then the output sin1pTs f 2 spectrum from the transmitter is given by Vt 1 f 2  S1 f 2  Et 1 f 2  Ht 1 f 2 sin1pTs f 2 pTs f  c d  c d  1VNQ 1 f 2  1VNQ 1 f 2 pTs f sin1pTs f 2

(5.36)

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SIDEBAR Adaptive equalizers are used in all high-speed telephone line modems because the frequency characteristics change each time a new number is dialed. The equalizer is implemented as a transversal equalizer, which operates in the time domain rather than the frequency domain1. A transversal equalizer works to improve the pulse shape at the output of the receiver so that the pulses more closely resemble the ideal zero ISI shape. The received pulses are sampled repeatedly within the pulse period, and then weighted and delayed samples are added to the original pulse to im-

prove its shape. A training sequence is required in the signal to allow the transversal equalizer to adjust to the received signal. In a dial-up modem, the training sequence is sent at the time the connection is first made, with the assumption that the line characteristics will not change during the subsequent data transfer session. In a mobile radio system the equalizer must continuously adapt to the changing propagation path. For more detail on transversal equalizers, the reader should refer to a text on communications system theory10.

The arrangement is illustrated in Figure 5.8a. The raised cosine filters have zero transmission beyond frequencies defined by f  fc  fmax, where fmax  Rs2  (1  ). The xsin x equalizer operates only within the central lobe of the sin xx function. At f  1Ts the xsin x function goes to infinity, so the RRC filter parameter  must be less than 1 for this system to work. In practice, RF filters with raised cosine shaping use  values between 0.2 and 0.5. The maximum gain of the equalizer for   0.5 is given by G  20 log10 [0.75sin(0.75)]  10.5 dB, at the edge of the equalizer band. Transfer functions are given in voltage terms, hence we must use 20 log ( ) to obtain the gain of the equalizer in decibels. The bandwidth of the link must at least equal the bandwidth occupied by the signal, otherwise the spectrum of the received signal will be altered by transmission over the link and we will not have a zero ISI raised cosine waveform at the output of the receiver. Thus the transfer function of the link, L( f ), must be such that 0 L( f )0  1 and £( f )  kf, where £( f ) is the phase characteristic of the link with frequency, and k is a constant, over the bandwidth occupied by the signal, Rs(1  ) Hz. If this condition is not met, either in magnitude or phase, we can insert an equalizer in series with the link to force the required condition. The equalizer can be at the transmitter or the receiver. In systems where the characteristics of the link vary with time, as in a mobile link which suffers from multipath interference, for example, the link equalizer can be adaptive. Multipath is less of a problem for mobile satellite links than for cellular telephone systems, where adaptive equalizers are commonly used, because the satellites are used only when their elevation angle is at least 5. The discussion of filter characteristics and signal spectra thus far has ignored the phase response of the filters and the resulting phase spectra of the waveforms. It can readily be shown7 that the phase response of all filters and equalizers must be linear with frequency for the zero ISI condition to be met. Achieving a linear phase response throughout a communication system can be difficult in practice, and may require the use of phase equalizers.

Transmission of QPSK Signals through a Bandlimited Channel Figure 5.8 shows a block diagram of a typical BPSK link. The same transmitter and receiver system, and the same satellite link, can be used to send QPSK signals, since QPSK is nothing more than two BPSK signals generated from carriers that are in phase quadrature, which can share a common link bandwidth. (Any pair of signals that is orthogonal

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can be separated by suitable signal processing in a radio receiver. Signals in frequency bands that do not overlap are orthogonal, and signals in phase quadrature are orthogonal. CDMA signals using ideal codes are also orthogonal.) The input equalizer with a xsin x transfer function Et( f ) is placed after the PSK modulator because typically PSK modulators are binary state devices which need to be driven by NRZ digital signals with voltage levels V. The input to the PSK modulator must therefore be a V signal, with V  data l and V  data 0 (or vice versa), created from the logic levels of the NRZ waveform that carries data bits (usually 5 V and 0 V, or 3.3 V and 0 V). The waveform at the output of the receiver, v7(t), must be a zero ISI waveform. This can be achieved only if we can make the transfer function of the entire link, from input terminal to output terminal, equal to VNQ( f ), the Nyquist raised cosine transfer function. The condition for zero ISI in the link is therefore Vi 1 f 2  Ht 1 f 2  Et 1 f 2  L1 f 2  Hr 1 f 2  GLPF 1 f 2  VNQ 1 f 2

(5.37)

where V i( f )  sin xx, the spectrum of the NRZ data pulses at the system input Ht( f )  Hr( f ), the transfer function of the band-pass RRC filters in the transmitter and receiver E t( f )  xsin x, the equalizer for the input NRZ waveform L( f ) is the transfer function of the link, equalized if necessary to make it linear GLPF( f ) is the transfer function of the baseband low pass filter following the demodulator. In order to satisfy the conditions for Eq. (5.37) to be true within the limits of the bandwidth occupied by the signals in the link, Rs(1  ), we must ensure that Vi 1 f 2  Et 1 f 2 L1 f 2 GLPF 1 f 2 Ht 1 f 2  Hr 1 f 2

1 1 1  VNQ 1 f 2

(5.38)

QPSK (quadrature phase shift keying) is the most popular choice of modulation technique for use in satellite communication links carrying digital data. It will be described in more detail in Section 5.4, but basically a digital data stream is taken two bits at a time and used to generate one of four possible phase states of the transmitted carrier. If the data rate is Rb, bits/s, the symbol rate for the QPSK carrier is Rs  Rb2 sps. In order to recover the symbol stream with zero ISI, we must shape the transmitted spectrum such that after demodulation a single symbol creates a zero ISI waveform at the output of the demodulator. Then, sampling of the symbol stream can be achieved with zero intersymbol interference. In practice, a QPSK system has two demodulators, one for each pair of symbols (phase states) in the QPSK carrier. We shall consider only one channel in looking at ISI. Figure 5.8a shows a typical arrangement for one-half of a QPSK transmit–receive link. The other half is identical except that the carrier used for modulation and demodulation is shifted in phase by 90°. Since the carriers in the two channels have a 90° phase difference, the channels are identified as in-phase (I) and quadrature (Q). The data presented to the QPSK modulators is in NRZ format and causes a jump in carrier phase at each symbol transition, as seen in Figure 5.8b. The input data rate to the demodulator is Rs  Rb2 sps, giving the QPSK spectrum shown in Figure 5.9a.

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0

−10

−20

−30 −1.0

+1.0

0 Normalized frequency [f − fc ] Rs (a)

Relative gain (dB)

0 α = 0.45 −10

−20

−0.8

−0.4 0 0.4 Normalized frequency [f − fc ] Rs

0.8

(b )

0 Relative gain (dB)

184

−10

α = 0.45 equalized by x sin x

−20 −0.8

0.4 −0.4 0 Normalized frequency [f − fc ] Rs (c )

0.8

FIGURE 5.9 (a) The frequency spectrum of an unfiltered QPSK signal with carrier frequency fc and symbol rate Rs. Only the central lobe of the spectrum is shown. (b) Transfer function of RRC band-pass filter with   0.45. (c) The frequency spectrum of a QPSK signal with carrier frequency fc and symbol rate Rs after equalization by xsin x and processing by an RRC filter with   0.45.

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The central lobe of the unfiltered QPSK spectrum extends from ( fc Rs) to ( fc  Rs), giving a band occupancy of 2Rs. The spectrum must be narrowed for transmission via a radio channel, and this is achieved by use of a band-pass filter meeting the zero ISI criterion, for example, a square root raised cosine filter. The band-pass square root raised cosine filter is a transformation of the low pass square root raised cosine filter and has a response 0 H( f )0  12 at frequencies ( fc Rs2) and ( fc  Rs2). The frequencies at which H( f ) falls to zero are determined by the rolloff factor  in Eq. (5.24). Matched filter operation of the link requires that the raised cosine filter response be split between the transmit end and the receive end of the link. Thus a square root raised cosine response filter is required after the modulator in the transmitter and before the demodulator in the receiver. Finally, because we are using NRZ pulses rather than impulses, we need an xsin x equalizer with x  [ fc  f Rs2] to equalize the spectrum from the modulator.

EXAMPLE 5.3.3 A satellite transponder has a bandwidth of 36 MHz. Earth stations use RRC filters with   0.4. What is the maximum bit rate that can be sent through this transponder with (i) BPSK (ii) QPSK? The maximum symbol rate for an RF link is given by Eq. (5.32) Rs  B 11  a2  36 M1.4  25.7 Msps The corresponding bit rates for BPSK and QPSK are (i) BPSK Rb  Rs  25.7 Mbps (ii) QPSK Rb  2  Rs  51.4 Mbps



EXAMPLE 5.3.3 A data stream at 240 Mbps is to be sent via a satellite using QPSK. The receiver IF frequency is 240 MHz. Find the RF bandwidth needed to transmit the QPSK signal when raised cosine filters with   0.45 are used. The 240 Mbps signal is divided into two 120-Msps symbol streams and applied to I and Q channel modulators fed by an IF carrier, with a 90° phase difference. The resulting spectrum from each modulator has a width of 240 MHz between zeros of the central lobe of the PSK spectrum. The I and Q signals are added and applied to an xsin x equalizer with x  [ fc fRs] extending to 87 MHz from the carrier. The maximum gain of the function xsin x at 87 MHz from the carrier is 9.5 dB. Figure 5.10a shows a block diagram of one-half of the transmit end of the QPSK link. The equalized spectrum is applied to the square root raised cosine filter. The response of this filter is 3 dB down at fc  Rs2, that is, at fc  60 MHz. In practice, one filter combining the square root raised cosine and xsin x responses is used. The ideal combined response of this single filter is shown in Figure 5.9c. Thus in the IF amplifier of the receiver, the signal spectrum is 6 dB down at 180 and 300 MHz. The band-pass filter will have zero response for f fc (Rs2)(1  ) and f  fc  (Rs2)(1  ), that is, below 153 MHz and above 327 MHz. Figure 5.10b shows the transmitted QPSK spectrum centered of the IF carrier. If we examine the spectrum of the QPSK signal at the receiver, we find that the 3-dB bandwidth is 120 MHz and the total frequency band containing all of the signal energy is 174 MHz. A typical satellite transponder for such a signal would have a 3 dB bandwidth of 140 MHz. Beyond

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PSK modulator

X sin X Equalizer

Square root raised cosine filter

QPSK signal at RF

Eq. NRZ symbols 120 Mbps 240 MHz IF carrier

Upconverter

Orthogonal channel (a)

Normalized Power Spectral Density (dBW/Hz)

186

0

−10

−20 140 MHz transponder channel bandwidth −30 120

180

240 Frequency (MHz)

300

360

f

(b) FIGURE 5.10 A QPSK data system with an NRZ format and symbol rate Rs  120 Msps symbol rate. IF frequency is 240 MHz. (a) Transmit end block diagram for model of 120 Msps symbol rate QPSK link. Only one channel is shown. (b) Transmitted QPSK spectrum after processing by square root raised cosine filter and xsin x equalization. Filter roll-off factor  is 0.45.

140 MHz the spectrum of the QPSK signal would be attenuated by the transponder filter, leading to some spectral distortion of the receiver signal and consequent ISI in the demodulated waveform. However, the energy contained in the QPSK spectrum beyond 70 MHz from the carrier is small, and the ISI caused by the transponder filter is minimal. 

Practical filters invariably cause some ISI because it is impossible to realize the square root raised cosine characteristic exactly. Typically, with a high-speed digital link operating at 120 Msps, up to 2 dB of additional carrier power must be provided to achieve a 10 8 BER, compared to the theoretical power needed for this error rate, in a carefully filtered QPSK link. The extra power is called an implementation margin. Implementation margin accounts for the nonideal nature of real communication systems. Timing jitter leading to incorrect sampling instants, real filters rather than Nyquist RRC filters, and phase and amplitude distortions all contribute to ISI levels that are higher than theory predicts. Implementation margin covers all of these effects.

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5.4 DIGITAL MODULATION AND DEMODULATION In this section we will review methods for digital transmission used on satellite links. We will not attempt to summarize the extensive literature of digital communications in general; the interested reader should refer to the references that deal specifically with digital communication systems (e.g., [1, 10–13, 17]).

Terminology While any parameter of a carrier waveform—amplitude, frequency, or phase—may be digitally modulated, phase modulation is almost universally used for satellites. For historical reasons, digital phase modulation is called phase shift keying, abbreviated PSK. An m-phase PSK modulator puts the phase of a carrier into one of m states according to the value of a modulating voltage. Two-state or biphase PSK is usually called BPSK, and four-state or quadriphase PSK is termed QPSK. Other numbers of states and some combinations of amplitude and phase modulation are possible and are employed in terrestrial links, but historically, satellite users have been reluctant to adopt anything besides BPSK or QPSK. An important reason for this is the higher values of CN ratios required for acceptable bit error rates. Several of the next generation of Ka-band satellites designed for Internet access will use 16-QAM on the link between the satellite and hub station, where higher values of CN can be maintained18,19. Baseband processing transponders are used to convert the 16-QAM signals on the hub–satellite link to QPSK for the links to the customer earth stations. The 16-QAM signal does not pass through a nonlinear transponder and intermodulation problems are avoided. Any type of PSK can be direct or differential, depending on whether it is the state of the modulating voltage or the change in state of the modulating voltage that determines the transmitted phase. Whether direct or differential, a PSK modulator causes the phase of a carrier waveform to go to one of a finite set of values. The transition time plus the time spent at the desired phase constitute a fixed time interval called the symbol period; the transmitted waveform during the interval is called a symbol. The set of all symbols for a particular modulation type is called its alphabet. Thus BPSK has a two-symbol alphabet and QPSK has a four-symbol alphabet. In the digital modulation process, a stream of incoming bits determines which symbol of the m available in the alphabet will be transmitted. Mathematically, Nb bits are required to specify which of m possible symbols is being transmitted where Nb and m are related by Nb  log2 m

(5.39)

As defined by this equation, Nb is the number of bits per symbol for the m-PSK modulation scheme. Standard practice is to make m a power of 2 so that Nb will be an integer.

Modulation and Coding The boundary between digital modulation and digital encoding is not well defined. In encoding for forward error correction (FEC), redundant bits are added to an incoming bit stream so that errors in transmission may be detected and corrected at the other end of the link. When the redundant bits are added at baseband and the composite (information bits plus redundant bits) bit stream is used to phase modulate a carrier and produce the transmitted symbols, then the division between modulation and encoding is obvious. But the modulator itself may be designed to add redundant bits during the modulation

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process, as in trellis coding, for example, making encoding and modulation inseparable15,16. In this section we will ignore encoding for FEC and concentrate strictly on the modulation process for turning an incoming bit stream into RF symbols. We will assume that any FEC encoding is done ahead of the modulator by methods to be presented later. It is unfortunate that differential phase modulation is frequently called differential encoding, since it is a characteristic of the modulation and demodulation equipment and plays no role in coding as it is usually understood. Differential encoding would more properly be called differential modulation, and we will discuss it after we have presented direct modulation.

Bit and Symbol Error Rates The figure of merit for a digital radio link is its bit error rate (BER), which should strictly be called the bit error probability (Pb). Mathematically, this is the probability that a bit sent over the link will be received incorrectly (i.e., that a 1 will be read as a 0 or vice versa) or, alternatively, the fraction of a large number of transmitted bits that will be received incorrectly. Like a probability, it is usually stated as a single number, for example, 1  10 4 or 0.0001. The BER plays the same role as an indicator of quality in a digital communication system that the signal-to-noise ratio plays in an analog link. It is important to remember that, despite its name, bit error rate does not depend on the speed of a digital transmission. It is simply the likelihood that a single bit error will occur within N received bits, independent of the rate of transmission. Physically a bit error occurs because a symbol error has occurred. At some point in the link noise has corrupted the transmitted symbol so that the decision circuitry at the receiver cannot identify it correctly. For example, the carrier phase may have been transmitted as 90° but additive noise may have changed the received carrier phase to 90°. If one symbol carries Nb bits and if differential modulation is not used, then a single symbol error may cause 1, 2, … , Nb bit errors. With differential modulation, an error in one symbol will cause the symbol that follows to be misinterpreted, and the number of bit errors per symbol error may exceed Nb, the number of bits per symbol. The mapping of pairs of data bits to the phase angles of a QPSK signal is usually done using Gray coding. Gray coding ensures that adjacent phase states differ by only a single bit, rather than two bits. Noise must cause a phase error in excess of 135° to cause a two bit error in a QPSK signal. Noise that causes a phase error of more than 45° but less than 135° will cause a single bit error. It is therefore much more likely that a single bit error will occur than a two bit error when Gray coding is used. As a result, it is usual to assume that for QPSK signals the probability of a symbol error, Pe is equal to the probability of a bit error, Pb. The curve relating BER to CN is so steep that the coefficient applied to Pb has very little influence on the bit error rate calculation, so it is usually set to unity in practice. (See Figure 5.13.) Symbol errors arise from thermal noise, from external interference, and from intersymbol interference. If only thermal noise is considered, then the symbol error rate or symbol error probability (Pe) may be calculated unambiguously from EsN0, the energy per symbol in joules divided by the noise power density in W/Hz, measured in the IF bandwidth at the demodulator input. The higher the value of EsN0 the lower will be the BER. The value of EsN0 may be determined from the input value of CN ratio, expressed as a ratio (i.e., not in decibels). Assume that C watts of carrier power are transmitted during one symbol interval Ts seconds, where Ts  1Rs and Rs is the symbol rate on the link. The energy received during that symbol period is Es, where Es  C  Ts  C Rs

(5.40)

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The noise power spectral density N0 is the received noise power N (in watts) divided by the noise bandwidth Bn (in hertz) at the demodulator input N0  NBn

(5.41)

Combining the last two equations we have EsN0  CBn N0 Rs

(5.42)

The square root raised cosine (RRC) filter discussed in the preceding section has a noise bandwidth Bn Hz numerically equal to the symbol rate Rs in symbols per second. Thus a receiver designed with filters of this type to achieve zero ISI also has BnTs  1 and EsN0  CN. Practical filters such as Butterworth or Chebychev come close to the shape of the square root raised cosine filter, giving BnTs products close to unity.

Binary Phase Shift Keying (BPSK) In binary phase shift keying, an incoming bipolar bit stream u(t) sets the phase of a carrier to 90° (2 rad). Thus, if ui is the ith bit, then the transmitted carrier v(t), is given by v1t2  Vc cos1vct  uip22

(5.43)

where Vc is an arbitrary amplitude frequently set to 1, and c is the carrier frequency in rad/s. Since ui must have a value of 1, a logical one (baseband voltage V) is transmitted by setting the phase to 2 rad and a logical zero (baseband voltage V) is transmitted with a 2 phase. Using trigonometric identities, we may rewrite Eq. (5.43) as v1t2  uiVc sin1vct2

(5.44)

and we see that BPSK resembles amplitude modulation in which the modulating signal has a value 1 or 1 only. This causes the BPSK waveform to have a constant amplitude and an envelope AM detector cannot demodulate it. To recover the data bits ui the receiver uses a correlation detector, which is the equivalent of a matched filter. A correlation detector multiplies the received signal by a replica of the transmitted signal, integrates the result over the symbol period, and samples the output of the integrator at the end of the period1. The practical implementation of a correlation detector for BPSK signals is shown in Figure 5.11. The replica of the transmitted signal is a locally generated carrier that creates a coherent (in phase) sine wave at the carrier frequency. The BPSK detector of Figure 5.11 multiplies the received signal by the local carrier in a mixer, and then uses a low pass filter, rather than an integrator to recover the detected waveform. Ideally, the recovered time waveform at the low pass filter IF BPSK signal

Zero ISI waveform

multiplier

Sample gate

LPF

Symbol recovery Output symbols

sync Carrier recovery

sync IF oscillator

Bit clock

FIGURE 5.11 A coherent BPSK detector and symbol recovery circuit.

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output is a zero ISI waveform, which has a maximum value in the center of the received symbol period. Multiplying the BPSK signal by a coherent carrier, followed by low pass filtering is the key to recovering the baseband signal. The BPSK signal is defined in Eq. (5.44), without shaping by the Nyquist filters of the transmitter and receiver. When multiplied by a locally generated, in phase carrier sin t, the multiplier output is v0(t) where v0 1t2  uiVc sin vct  sin vt  uiVc sin2 vct  uiVc  12 11 cos 2vct2

(5.45)

The low pass filter removes the double frequency term cos2 t leaving an output signal vr(t) vr 1t2  12 uiVc

(5.46)

Thus the BPSK demodulator has recovered the modulating signal ui. The magnitude factor Vc is removed in the IF stages of the receiver by limiting the magnitude of the received signal, exactly in the same way that a broadcast FM receiver limits a received FM signal prior to demodulation. At the center of each symbol interval, the output of the demodulator is sampled and a decision circuit decides whether vr(t) is greater or less than zero volts (i.e., whether the sample is a positive or a negative voltage, and thus determines whether ui was a 1 and represented a data one or whether it was a 1 and represented a data zero. The reference carrier that drives the multiplier (mixer) of the BPSK demodulator shown in Figure 5.11 must be derived from the received signal. This is accomplished with a carrier recovery circuit. One such circuit applies the BPSK signal to both inputs of a multiplier (mixer) to obtain the square of the BPSK signal. This has the effect of stripping off the modulation, but returns a double frequency signal component. The BPSK signal given by Eq. (5.44) is squared to give 3v1t2 4 2  1ui 2 2V 2c sin2 vct

(5.47)

Since ui is either 1 or 1, (ui) is always 1. We will ignore the term since it is easy to limit an AC (alternating current) waveform to a predetermined magnitude. Expanding the sin2 ct term gives an output from the squarer circuit of 2

v0 1t2  12 31 cos 2 vct4  12 12 cos 2 vct

V 2c

(5.48)

We can extract the double frequency term with a high pass filter and then divide it by two, which is easily accomplished with a phase locked loop (PLL), to give a reference carrier at the correct frequency, and with the correct phase angle. (The PLL provides a 90° phase shift that converts cos t to sin t.) Other techniques for carrier recovery, such as the Costas loop, are also used in coherent PSK receivers10,17. Most carrier recovery loops for BPSK have a 180° phase ambiguity; that is, when the loop is locked the phase of the recovered carrier may differ by 180° from the correct value. This has the effect of interchanging logical ones and zeroes and causes the demodulated bit stream to be the complement of what was transmitted. With QPSK, the loop can lock up in four different states, with offsets of 90°, 180°, and 90°. There are several ways to eliminate the ambiguity; one is to use differential encoding in which adjacent symbols have the same phase if the modulating voltage is a logical 1 and are 180° out of phase if the modulating voltage is a logical 0. This may be realized by a binary phase shifter that toggles between 0° phase shift and 180° phase shift each time the modulating bit is a 0. Incoming logical 1 values have no effect. Differential modulation is more error prone than direct modulation, since an error on a single bit in a differential system will cause one or more subsequent bits to be

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interpreted incorrectly. See Section 5.6 of reference 11 for a detailed analysis of errors in differential PSK. Most practical satellite systems avoid differential encoding and check the status of the recovered carrier phase periodically by transmitting a known (unique) word. Logic in the receiver looks for the unique word. If the unique word is received correctly, then the recovered carrier phase is correct. If the unique word is the complement of the known word (1s and 0s interchanged), than the recovered carrier phase is off by 180° (in a BPSK receiver) and the demodulated data stream should be complemented before it is sent to the end user.

Probability of a Symbol Error The received signal will always be accompanied by noise. The noise power in the receiver is calculated using the techniques discussed in Chapter 4, which provide a value of CN ratio at the input to the demodulator. The CN ratio expresses noise as a power ratio relative to the received carrier. The noise in the receiver is assumed to be additive white Gaussian noise (AWGN), leading to a description of the satellite link as an AWGN channel. The assumption that the noise is additive, white, and has a Gaussian voltage distribution is necessary to simplify the analysis of error probability. In satellite links, the assumption is usually valid provided the noise is thermal in origin (e.g., from the receiver front end and transponder). If the noise is actually interference from another communication link, the assumption may not be valid, but often AWGN conditions are assumed for want of a better way to analyze interference. For the analysis of error probabilities, we need to work with voltages. The noise voltage at the output of the demodulator is given by n0(t). At the sample instant, we will assume a noise voltage n0 volts at the demodulator output. The decision circuit will make an error if noise changes the sign of v(t) at the sample instant. This is illustrated in Figure 5.12, where the received signal has sample voltages of V and V volts. At the sampling instant, the output of the demodulator v0 will be the sum of the signal sample V and the noise sample n0. v0  V  n0

(5.49)

There are two possible ways that an error can occur, depending whether a V or a V signal was transmitted. If a signal V was sent and n0 V (i.e., the magnitude of the noise sample is negative and larger than V) the sum of the signal and noise will be less than zero, giving a symbol error. If the transmitted signal was V and noise at the sampling instant was greater than V, an error will occur because the sum is greater than zero volts. We can calculate the probability that an error will happen, and thus the symbol error rate Pe by the following argument. At the correct decision time, the amplitude of the signal will be V, where V is the peak magnitude of the waveform at the output of the demodulator. The Gaussian distribution is symmetrical about zero volts, so the probability of an error occurring is Pe  12 P1output is 7 0 when V was sent2  12 P1output is 6 0 when V was sent2  P1output is 7 0 when V was sent2  P1n0 7 V2 (5.50) Thus the probability of an error occurring in the transmission of symbols reduces to the simple condition that, at the instant the receiver output waveform is sampled, the noise voltage at the receiver output is larger in the wrong direction than the sample value of the signal. Since the noise is defined to be Gaussian, we can find the probability that the noise exceeds a given value V volts. The probability that the sampled value of the

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P (vo )

Decision circuit output is “0” when vo < 0 volts

Decision circuit output is “1” when vo > 0 volts

ui = +1 “1” transmitted

n (t ) < (−V ) causes error area =

2 ∞e−u 2du πx

∫

vo V

0

Demod output voltage vo = n (t ) + Vui

P (vo ) ui = −1 “0” transmitted

n (t ) > (+V ) causes error 2 ∞e−u 2du πx

∫

vo −V

0

Demod output voltage vo = n (t ) + Vui FIGURE 5.12 Illustration of errors in a binary decision circuit caused by additive Gaussian noise. The threshold is at zero volts.

AWG noise voltage exceeds a value V is given by the integral of the PDF of the noise, from V to infinity1. P1n0 7 V 2 

s 12p





V

exp c

l2 d dl 2s2

(5.51)

where is the rms noise voltage. The integral in Eq. (5.51) cannot be solved analytically. Numerical or approximate solutions must be used, and one such expression is known as the Q function, Q(z). An alternative form is the complementary error function, erfc(x), which is closely related to Q(z). Fortunately, there are relatively simple approximate expressions available for these functions when the probability of an error is small—which is usually the case for a workable digital link, where we expect bit errors to occur infrequently. The condition for probability of error to be small is that V W . The sampled signal must be much larger than the rms noise at the receiver output. A value of V  3 makes the following approximation valid1. Q1z2 

1 12p





z

exp c

l2 d dl  1 3z  12p4  exp1 z222 2

(5.52)

The probability that n0 exceeds V at the sample instant is given by P1n0 7 V2  Q3Vs4

(5.53)

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The complementary error function erfc(x) can also be used to find the probability of an error with a Gaussian noise voltage. The probability that n0 exceeds V at the sample instant is given by P1n0 7  V2  12 erfc3V 1s 122 4

(5.54)

Appendix C gives the relationship of the complementary error function erfc(x) to Q(z), and tables of values for both the complimentary error function and the Q function. Note that the Q function and erfc function are defined for a normalized rms value of the Gaussian variable of one. When applied to symbol error analysis, the Q function has an rms noise voltage of one volt. Errors occur whenever noise at the sample instant exceeds either V or V, depending on the transmitted symbol. The threshold for an error is therefore given by a signal to noise voltage ratio of V1  V. The probability of error for each transmitted data state is given by the Q(z) or erfc(x) functions in Eqs. (5.52) and (5.54). We should ensure that we send equal numbers of data 1 and data 0 states to make the probability of an error occurring in the 1 states the same as the probability of an error occurring in the 0 states. This usually requires a randomizer or scrambler to be inserted in the data stream at the transmitter to prevent the occurrence of long strings of data 1s or 0s which would violate the required condition. The scrambler also helps the symbol clock in the receiver to stay synchronized by providing frequent phase transitions in the received signal. One symbol lasts for Ts seconds. The power in the symbol waveform is V 22R where R is the input resistance of the decision circuit. We will assume a resistance R  1 ohm, as is commonly done in the analysis of communication signals. We will assume a constant amplitude V for the carrier waveform (ignoring the effects of the Nyquist RRC filters on pulse shape), so the energy per symbol, Es, is given by Es  12 V 2  Ts joules

(5.55)

Assuming that we have a matched filter receiver, the sampled signal voltage at the demodulator output is V volts where V  112EsTs 2 volts

(5.56)

The noise power at the demodulator output is N  2R  2 watts, relative to a resistance of 1 ohm. The noise is assumed to be white and therefore has a constant NPSD, N0 watts/Hz, across the noise bandwidth Bn Hz of the receiver. In a receiver with ideal RRC filters, Bn  1 Ts. Hence the single sided noise power spectral density is given by N0  N Bn  s2Bn  s2  Ts joules

(5.57)

s  11N0 Ts 2 volts rms

(5.58)

Hence

Combining Eqs. (5.56) and (5.58) yields V 1s122  2 32EsTs  12 TsN0 4  2 3EsN0 4

(5.58)

The probability that a symbol error occurs is therefore Pe  12 erfc3 11Es N0 2 4  Q3 112Es N0 2 4

(5.59)

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BPSK Bit Error Rate For BPSK a bit and a symbol are the same, so Eq. (5.59) can be written as Pb  12 erfc c

Eb 2Eb d  Qc d A N0 A N0

(5.60)

The analysis in Chapter 4 provides methods by which the carrier-to-noise ratio in an earth station receiver or satellite transponder can be calculated for any satellite link. The results are in terms of the ratio of carrier power to noise power at the input to a demodulator, with the ratio CN usually given in decibels. We need to relate the CN ratio for a receiver to the EsN0 ratio that provides us with a way to calculate the probability of a symbol error. In a receiver with ideal RRC filters, regardless of the value of the roll-off factor , the noise bandwidth of the filter is equal to the symbol rate, which is the reciprocal of the symbol period: Bn  Rs  1Ts, or BnTs  1

(5.61)

A result from matched filter theory states that the energy per symbol is the carrier power times the symbol duration if the transfer function of the receiving filter matches the spectrum of the received signal. A correctly designed digital radio link with RRC filters meets this criterion, so we have Es  C  Ts joules

(5.62)

and the single sided noise power spectral density N0 W/Hz is just the noise power N watts divided by the noise bandwidth Bn in hertz N0  NBn W/Hz

(5.63)

Hence for the ideal conditions specified above where Ts Bn  1 EsN0  CTsBnN  CN

(5.64)

Applying the result of Eq. (5.64) we find that the bit error rate for a BPSK signal in an ideal RRC filtered link is Pe BPSK  12 erfc c

C 2C d  Qc d AN AN

(5.65)

Note that the CN value used in Eq. (5.65) is a linear power ratio, not a decibel value. Using decibel CN ratios in BER equations is a frequent source of error for beginning communications engineers. Since coherent detection is the most efficient way of demodulating direct BPSK, Eq. (5.65) is the relation normally used to determine EbN0, and hence the CN ratio that a satellite link must maintain to meet a specified bit error rate requirement.

QPSK Bit Error Rate QPSK is simply two BPSK links operating on the same radio channel with their carriers in phase quadrature. The BER for each BPSK link is identical, and given by Eq. (5.65). When the bit stream at the transmitter is split into two to drive the I and Q channel of a QPSK transmitter, the symbol rate on the link is halved. But the error rate remains the same as if the signal had been sent as a BPSK transmission at twice the symbol rate, with the same transmitter power. This is because BER is probability of error per bit, and the probability of a bit error is the same for all bits in a QPSK system, regardless of which

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channel (I or Q) they travel through. So QPSK ends up with the same BER as BPSK when considered in terms of EbN0. The total energy per symbol of a QPSK signal is therefore twice that of either of the constituent BPSK signals, or a single BPSK signal sent over the same link with the same EbN0 ratio. Hence Es QPSK  2  Eb BPSK

(5.66)

and therefore to obtain the same error rate for a QPSK signal that we can achieve with a BPSK signal in an RF channel with a noise bandwidth Bn Hz, we require 1CN2 QPSK  2  1C N2 BPSK

(5.67)

Thus for QPSK, transmitted at a rate Rb bits/second in a channel with noise bandwidth Bn Hz Pe QPSK  12 erfc c

Eb 2Eb C d  Qc d  Qc d A N0 A N0 AN

(5.68)

The analysis of the system performance of a radio link using BPSK is always carried out in terms of CN ratio, not EbN0. Almost all communication system textbooks leave the BER results for radio links in terms of EbN0, and state that the BER performance for BPSK and QPSK are the same. However, when BER is considered as a function of CN ratio, BPSK and QPSK do not have the same BER. It takes twice as much transmitter power to deliver two BPSK data streams as to deliver one. Therefore, QPSK requires 3 dB more CN ratio than BPSK to achieve the same error rate when transmitting at twice the bit rate in the same channel bandwidth. If a link is operated with QPSK, the CN required for a given error rate is 3 dB higher than when the same link is operated with BPSK. The advantage of QPSK is that it can send twice as many bits per second relative to BPSK using a channel with a specified bandwidth. This advantage of QPSK can only be exploited if the CN at the receiver is sufficiently high. EXAMPLE 5.4.1 A satellite link achieves a CN ratio in the receiver under clear air conditions of 14.0 dB. (14.0 dB  power ratio of 25.) The receiver has a RRC filter with a noise bandwidth of 1.0 MHz and a roll-off factor of 0.3, with ideal correlation detection BPSK and QPSK demodulators. What are the bit rate, symbol rate, occupied (absolute) bandwidth of the link, and BER when the link is operated: (i) with BPSK modulation and (ii) with QPSK modulation? If rain attenuation on the link causes the received signal to be attenuated by 3 dB, what are the new BER values for BPSK and QPSK modulations? Assume that ideal RRC filters are used. For all radio links using band-pass RRC filters, the symbol rate is equal to the noise bandwidth of the RRC filter. Thus, for both BPSK and QPSK: The symbol rate for this link is Rs  Bn  1 Msps The occupied bandwidth of the RF signal is Bocc  Rs 11  r2  1.3  Rs  1.3 MHz Probability of error can be found from Eqs. (5.65) and (5.68) using the tables in Appendix C.

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(i) BPSK The bit rate Rb  symbol rate Rs  1 Mbps.

BER in clear air  Pe BPSK  Q3 12CN2 4  Q1 112  252 4  Q17.072  7.8  10 11

(ii) QPSK The bit rate Rb  2  symbol rate Rs  2 Mbps.

BER in clear air  Pe QPSK  Q 3 1CN2 4  Q1 1252  Q152  2.8  10 7

When rain attenuation reduces the received signal by 3 dB, the receiver CN  11 dB. (Power ratio  12.59.) The resulting BER values are: (i) BPSK BER in rain  Pe BPSK  Q3 112CN2 4  Q1 12  12.592  Q15.022  2.8  10 7 (ii) QPSK BER in rain  Pe QPSK  Q3 11CN2 4  Q1 112.592  Q13.552  2.2  10 4 All the BERs are acceptable except the last value for QPSK. With a bit rate of 2 Mbps, and a BER of 2.2  10 4 there are hundreds of errors occurring every second. Forward error correction would be needed in the QPSK link to maintain an acceptable BER. Using QPSK rather than BPSK over a link with a noise bandwidth defined by the RRC filter in the receiver doubles the bit rate. However, the CN ratio must be higher to sustain an acceptable error rate. A decision on whether to implement BPSK or QPSK on a given link will rest on the CN values that can be maintained, and the length of time for which the CN ratio might fall to levels at which an unacceptably high BER results. 

Figure 5.13 shows bit error rate for an ideal system with RRC filters having a fixed bandwidth B Hz, carrying BPSK and QPSK signals. The QPSK system carries twice as 10−2

QPSK Probability of bit error (BER)

196

10−4 NCFSK

DBPSK

BPSK 10−6

10−8 0

5

10 Receiver C/N in dB

15

FIGURE 5.13 Bit error rate as a function of CN for a link with ideal RRC filters and no intersymbol interference or timing jitter. The curves shown are for BPSK and QPSK with coherent detection (BPSK, QPSK), differential BPSK (DBPSK), and noncoherent FSK (NCFSK). The implementation margin in each case is 0 dB.

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much information (twice the number of bits carried by the BPSK system) but needs that extra 3 dB of CN ratio to achieve the same BER as BPSK. In a practical BPSK or QPSK radio link, which must have real filters, and inevitably suffers phase jitter in the carrier recovery circuit and timing jitter in the bit clock when the CN ratio is low, the ideal results shown in Figure 5.13 cannot be achieved. An implementation margin must be added to the CN ratio to account for the difference between a real system and the ideal system for which the results of Figure 5.13 apply. In low bit rate systems, such as SCPC channels in VSAT systems and LEO mobile satellite links, implementation margins as low as 0.5 dB have been reported in the literature. For high bit rate systems carrying multi-megabit per second QPSK data streams, implementation margins as high as 2 dB may be required. Hence BER for practical BPSK and QPSK satellite links is calculated from the following relationships 1C N2 eff  1CN2 0 Implementation margin dB 1C N2 eff ratio  101CN2 eff10 as a ratio C 2C BERBPSK  12 erfc c d  Qc d A N eff ratio A N eff ratio C C d  Qc d BERQPSK  12 erfc c A 2N eff ratio A N eff ratio

(5.69) (5.70) (5.71) (5.72)

EXAMPLE 5.4.2 A satellite link uses a bandwidth of 10 MHz in a 52 MHz wide Ku-band transponder. The transmitter and receiver have RRC band-pass filters with roll-off factor   0.25. The overall (CN)0 ratio for the carrier in the receiver is 16.0 dB in clear air, falling below 13.0 dB for 0.1% of an average year. The transmitter and the receiver have both BPSK and QPSK modulators and demodulators. The implementation margin for the BPSK demodulator is 0.8 dB and for the QPSK demodulator is 1.2 dB. Determine the bit rate that can be sent through the link with BPSK, and with QPSK. Find the bit error rate for each modulation in clear air conditions and for the 0.1% of the year conditions. Which modulation would you recommend for this system? The symbol rate for the link is 101.25  8.0 Msps. With BPSK the bit rate equals the symbol rate, so Rb BPSK  Rs  8.0 Mbps. With QPSK the bit rate equals twice the symbol rate, so Rb QPSK  2Rs  16.0 Mbps. For the link using BPSK, the BER is found from Eq. (5.71), BERBPSK  12 erfc3 11CN2 eff ratio 4  Q3 112CN2 eff ratio 4 In clear air (CN)eff  16.0 0.8  15.2 dB, hence (CN)eff ratio  101.52  33.1 and BERBPSK  12 erfc3 133.1 4  12 erfc3 5.754

Using the erfc(x) table in Appendix C and interpolating between the values of erfc(x) for x  5.7 and x  5.8, the BER can be estimated as BERBPSK  12  5  10 15  2.5  10 15  0 With BPSK, the link delivers 8  106 bits per second. With a BER of 2.5  10 15 a bit error will occur, on average, once every 1(8  106  2.5  10 15)  5  107 s. This is a time of about 112 years, so in reality there will be no bit errors on this link. Anytime that x  5.5 in the erfc(x) expression [or z  8 in the Q(z) function] the bit error rate on the link is effectively zero. For 0.1% of the year (CN)eff 13.0 0.8  12.2 dB, (CN)eff ratio 101.22  16.60, and BERBPSK  12 erfc3 116.6 4  12 erfc3 4.074

Using the erfc(x) table in Appendix C and interpolating between the values of erfc(x) for x  4.0 and x  4.1, the BER will exceed 5  10 9 for 0.1% of an average year when BPSK is used as

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the modulation on this link. At a bit rate of 8 Mbps, a bit error occurs, on average, every 25 s with a BER of 5  10 9. When the modulation on the link is changed to QPSK, the bit error rate will increase, as indicated by Eq. (5.72) BERQPSK  12 erfc3 11C 2N2 eff ratio 4  Q3 11CN2 eff ratio 4 In clear air (CN)eff  16.0 1.2  14.8 dB, and (CN)eff ratio  101.48  30.20 BERQPSK  12 erfc3 1130.2 22  12 erfc33.894 Using the erfc(x) table in Appendix C, the BER can be estimated as BERQPSK  12  4  10 8  2  10 8 With QPSK, the link delivers 16  106 bits per second. With CN  16.0 dB there is a bit error, on average, once every 1(16  106  2  10 8)  3.12 s. For 0.1% of the year (CN)eff 13.0 1.2  11.8 dB and (CN)eff ratio 101.18  15.14 BERQPSK  12 erfc3 1115.1422 4  12 erfc32.75 4

Using the erfc(x) table in Appendix C, BER will exceed 5  10 5 for 0.1% of an average year when QPSK is used as the modulation on this link. At a bit rate of 16 Mbps, there are 800 bit errors every second, on average, when CN  13.0 dB. What is an acceptable bit error rate depends on the particular application. For financial transactions, a BER of 10 12 is typically required. Satellite systems don’t often guarantee such a low error rate, so some form of error detection is needed, with retransmission of any data that are found to be in error. For general applications bit error rates of 10 8 to 10 6 are acceptable. Digital voice transmission can withstand occasional error rates as high as 10 4, which typically lead to baseband SN ratios of 34 dB. In this example, BPSK modulation has BER 10 8 for 99.9% of an average year, and would therefore be a satisfactory choice. QPSK can deliver an acceptable error rate in clear air conditions, 2  10 8, but when the CN starts to fall the BER increases quickly, so that with 3-dB attenuation in the link the BER is 5  10 5. This is not adequate for general applications, but would suffice for digital voice links with a requirement that SN  30 dB for 99.9% of the year. The obvious advantage of using QPSK is that twice as many voice channels can be carried by a 16-Mbps bit stream modulated with QPSK, compared to BPSK modulation which can carry only 8 Mbps, within the available channel bandwidth (Bocc) of 10 MHz. 

Generation of Quadrature Phase Shift Keying (QPSK) Signals In QPSK the phase, , of the carrier is set by the modulator to one of four possible values. We may write the result as v1t2  V121cos vct f2

(5.73)

where  takes on the values 4, 34, 54, and 74 rad. The factor 12 is for our later convenience. Using trigonometric identities to expand Eq. (5.73) we obtain v1t2  V12 cos vct cos f  V12 sin vct sin f

(5.74)

The first term is a BPSK signal in phase with the carrier, and is called the I channel. The second term is a BPSK signal in quadrature with the carrier and is called the Q channel. Thus a QPSK waveform may be generated by combining two BPSK waveforms in quadrature. We may write the result as v1t2  uiV cos vct  uqV sin vct

(5.75)

where ui, represents a binary data stream modulating the I channel and uq represents a binary data stream modulating the Q channel. For both of these signals a logical 1 corresponds

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TABLE 5.1 The Relationship between the Modulating Bit Streams ui, uq, and the Phase Angle  of the Modulated QPSK Waveform ui

uq



1

1

1 1

1 1

1

1

4 34 54 74

to ui or uq  1, and a logical 0 corresponds to ui or uq  1. The relationship between ui , uq, and  is given by ui  12 cos f uq  12 sin f

(5.76) (5.77)

and is summarized in Table 5.1. Note that  is conveniently visualized as the phase angle of a phasor whose in-phase component is ui, and whose quadrature component is uq. See Figure 5.14. The bits ui and uq are selected alternately from the input bit stream. For example, ui may represent the odd-number bits and uq the even. In this case one binary data channel enters the QPSK modulator and the outgoing symbol rate is equal to half of the incoming bit rate. QPSK modulators and demodulators are basically dual-channel BPSK modulators and demodulators. One channel processes the ui bits and uses the reference carrier; the other processes the uq bits and uses a 90° phase shifted version of the reference. Figures 5.15 and 5.16 show generalized block diagrams of a QPSK modulator and demodulator. More detailed information is available in reference 13.

QPSK Variants We noted that QPSK may be visualized as the sum of two BPSK signals whose carriers are in phase quadrature. In conventional QPSK the bits ui and uq that modulate these carriers both make step changes at the same time. If the bit changes are staggered so that ui makes step changes at the beginning of each symbol period and uq makes step changes at the midpoint of each symbol period, the result is called stagg-

uq −1, 1 φ = 3π /4

1, 1 φ = π /4

ui

−1, −1 φ = 5π /4

−1, 1 φ = 7π /4

FIGURE 5.14 Phasor diagram showing the phase of a QPSK waveform modulated with all possible pair combinations of the bits (ui, uq). Each phasor is formed by summing a ui and a uq signal.

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I channel Multiplier NRZ data input

LPF Bit select

Σ QPSK output LPF

Multiplier Q channel 90° phase shifter

π /2

IF oscillator FIGURE 5.15 QPSK modulator. The bit select block sends alternate bits to the I andQ channels.

ered QPSK (SQPSK) or offset QPSK (OQPSK). If instead of steps the bits make sinusoidal transitions between their allowed values of 1 and 1, the result is minimum shift keying (MSK) or fast frequency shift keying (FFSK). These modulation systems produce spectra that are slightly different from conventional QPSK and that would appear to have some advantages over QPSK for satellite transmission. While they have received considerable academic attention, OQPSK, SQPSK, MSK, and FFSK have not been widely adopted for commercial GEO satellite applications. The prevailing attitude in the industry seems to be that any theoretical advantages that they might have over conventional QPSK either vanish when these techniques are used over a real transponder or else are so slight as not to justify the added expense that their I channel Multipliers IF QPSK input

LPF splitter

logic to sampler LPF

Carrier recovery

Q channel

π /2

90° phase shifter

IF oscillator FIGURE 5.16 QPSK demodulator. The logic block outputs two bits for each received QPSK symbol.

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implementation would require. For additional information the reader should consult references 2, 13, and 14.

5.5 DIGITAL TRANSMISSION OF ANALOG SIGNALS The previous sections have discussed techniques for transmitting and receiving digital information via satellite. Now we will turn our attention to the problem of putting analog voice and television signals into digital form for transmission, and recovering the analog signals after reception. While the material presented is generally applicable to all analog signals, we will emphasize baseband voice channels because virtually all telephone channels are now digital. Digital television transmission is covered in Chapter 11.

Sampling and Quantizing The basic processes in digital transmission of analog information are sampling, quantizing, and encoding. The principles underlying sampling are routinely presented in beginning courses in communications theory, and we will not reproduce them here. See references 1 and 3 for details. The sampling theorem states that a signal may be reconstructed without error from regularly spaced samples taken at a rate fs samples/second, which is at least twice the maximum frequency fmax present in the signal. Instead of transmitting the continuous analog signal, we may transmit the samples. For example, voice signals on satellite links are normally filtered at baseband to limit their spectra to the range 300 to 3400 Hz. Thus, one voice channel could be transmitted with samples taken at least 6800 times per second or, as it is usually expressed, with a minimum sampling frequency of 6800 Hz. Common telephone system practice is to use a sampling frequency of 8000 Hz. While transmitting the samples requires more bandwidth than transmitting the original waveform, the time between samples of one signal may be used to transmit samples of other signals. This is time division multiplexing (TDM), and we will discuss it later in this chapter. The samples to which the sampling theorem refers are analog pulses whose amplitudes are equal to that of the original waveform at the time of sampling. The original waveform may be reconstructed without error by passing the samples through an ideal low pass filter whose transfer function is appropriate to the sampling pulse shape. A communications system that samples an input waveform and transmits analog pulses is said to use pulse amplitude modulation (PAM). Figure 5.17 sketches this process. Analog pulses are subject to amplitude distortion, and they are also incompatible with conventional baseband digital signals in which pulses take on only one of two possible values. Hence pulse amplitude modulation is not used in communication links. Instead, the analog samples are quantized—resolved into one of a finite number of possible values—and the quantized values are binarily encoded and transmitted digitally. Thus each sample is converted into a digital word that represents the quantization value closest to the original analog sample. Quantization may be uniform or nonuniform depending on whether or not the quantized voltage levels are uniformly or nonuniformly spaced. At the receiver a digital-toanalog (DA) converter converts each incoming digital word back into an analog sample; these analog samples are filtered and the original input waveform is reconstructed. A communications system that transmits digitally encoded quantized values is called a pulse code modulation (PCM) system, a rather antiquated name for a process that is not really a modulation or a code. When the PCM process is applied to voice and TV signals, they are often referred to simply as digital voice and digital TV.

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v

v Input signal

v

Transmitted pulses

Samples

t

t

t

(a) Basic waveforms

Input speech signal

BPF 300-3400 Hz

PAM signal

Analog gate

A/D converter

PCM signal

LPF 0-3400 Hz

Recovered speech signal

D/A converter

Sampling pulse generator

(b) FIGURE 5.17 Pulse amplitude modulation (PAM). (a) Basic waveforms. (b) Block diagram of a PAM communications system. Dashed lines show the additional components that would convert it to a PCM system.

The standard word used in digital telephone systems has 8 bits, so with sampling at 8 kHz, the bit rate of a digital telephone (PCM) channel is 64 kbps. Twenty-four telephone channels are often transmitted on a single twisted pair telephone line using a system known as T1. The T1 system (described later in this chapter) creates a frame of 24 PCM words, one from each of the 24 voice channels, and adds a single framing bit to mark the end of the frame. This gives a 193-bit frame, and frames are transmitted at the rate of 8000 frames per second. The resulting bit rate is 1.544 Mbps, which has become a digital transmission standard in North America, regardless of whether digital voice or digital data are being transmitted. The quantization process, illustrated in Figure 5.18, prevents exact reconstruction of the digitized waveform. (The sampling theorem requires that analog rather than quantized samples be transmitted.) The error introduced is called quantization error; and a person listening to a reconstructed speech signal perceives the quantization error as an added noise called quantization noise. A uniform quantizer (Figure 5.19) operates with L levels spaced A volts apart. The input signal is amplitude limited to lie between (L2) and (L2). The quantizer determines in which level an incoming sample falls and puts out the identification number of that level. This identification number is the digital word that represents the sample. Transmitting L levels requires N bits where N  log2 L

(5.78)

L  2N

(5.79)

or The levels are normally numbered 0 through L 1. Thus an 8-bit PCM (N  8) system would quantize its incoming samples into one of 256 (L  2N  256) levels numbered 0 through 255. Samples of the analog signal are transmitted as binary words ranging

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v

v

t

t

(a)

(b )

v

t (c ) FIGURE 5.18 The quantizing process. (a) The input waveform and the quantization levels. (b) Quantized samples. (c) Quantized pulses. The pulse amplitude is encoded digitally for PCM transmission.

v 4∆ Signals with 3∆ < v ≤ 4∆ are transmitted as 3.5∆, encoded as 111

Level 7 3∆

Signals with 2∆ < v ≤ 3∆ are transmitted as 2.5∆, encoded as 110

Level 6 2∆

Signals with ∆ < v ≤ 2∆ are transmitted as 1.5∆, encoded as 101

Level 5 ∆ Level 4

Signals with 0 < v ≤ ∆ are transmitted as 0.5∆, encoded as 100

0 Signals with −∆ < v ≤ 0 are transmitted as −0.5∆, encoded as 011

Level 3 −∆ Level 2 −2∆ Level 1 −3∆ Level 0

Signals with −2∆ < v ≤ ∆ are transmitted as −1.5∆, encoded as 010 Signals with −3∆ < v ≤ −2∆ are transmitted as −2.5∆, encoded as 001 Signals with −4∆ ≤ v ≤ −3∆ are transmitted as −3.5∆, encoded as 000

−4∆ FIGURE 5.19 Levels and encoding for a uniform 3-bit (8 level) PCM quantizer.

203

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from 00000000 (decimal 0) through 11111111 (decimal 255). If the input signal amplitude is uniformly distributed with an rms value of Vrms, the signal-to-noise ratio of the reconstructed analog signal (assuming that only quantization noise is present) is given by1 1SN2  121Vrms  ¢ 2 2

(5.80)

For uniform quantization and a signal input that has equal probability of any voltage level, the quantization noise added to the recovered analog signal gives a baseband signal to noise ratio of (SN)Q where 1SN2 Q  6N dB

(5.81)

Thus a standard digital telephone channel using an 8-bit word and uniform quantization will have an average quantization SN ratio of 48 dB, using linear quantization.

Nonuniform Quantization: Compression and Expansion Uniform quantization introduces more noise when a signal is small and one quantization interval is large in comparison with the signal than it does when the signal is large and one quantization interval is insignificant. The problem is most apparent with the quiet talker. The quiet talker produces a voice signal with 30 dB lower power than the design level of the telephone system. As a voltage ratio relative to 1 V, 30 dB is 0.0314 V. Thus a telephone system in which the nominal power level is 0 dBm, and the impedance is 600 ohms (the standard values) has a nominal voltage range of 0.775 V. With an 8-bit word and 255 quantization levels, the step size is 2  0.775255  6.1 mV. The quiet talker produces a voltage level of 31.4 Vrms, with an equivalent peak sine wave level of 44.4 mV. Thus the quiet talker uses only the lowest 15 steps of the digital quantizer, equivalent to using a 4-bit quantizer. A 4-bit quantizer gives a quantization baseband SN ratio of 24 dB, so the quiet talker is producing signals that have, at best, a SN ratio of 24 dB rather than 48 dB. Improved noise performance can be obtained using nonuniform quantization in which the size of the quantization intervals increases in proportion to the signal value being quantized. The same effect can be obtained from a uniform quantizer if the input signal is compressed before quantization. The distortion introduced by the compressor must be removed at the receiver by an expander. The transfer functions of the compressor and expander are complementary, that is, their product is a constant and the amplitude distribution of a signal that has passed through both a compressor and an expander is unchanged. Compression at the transmitter followed by expanding at the receiver is called companding. Companding was first employed on terrestrial telephone systems using analog compressors that had logarithmic transfer functions. These were the so-called -law and A-law compressors. Later developments in digital technology allowed digital implementation of the compression and expansion functions and permitted the sampling, compression, quantization, and encoding operations to be combined into one piece of equipment called a coder. Companding with 8-bit digital voice channels leads to a subjective improvement in baseband SN ratio which is typically taken to be 15 dB. Thus a companded digital telephone link gives a baseband quantization SN ratio of 63 dB in the absence of bit errors. Figure 5.20 shows the compression characteristic of a typical analog compression circuit. The entire process of converting an analog voice signal to a 64-kbps digital bit stream, and converting a 64 kbps digital voice channel back to an analog voice signal is now done in a single integrated circuit. The telephone wire from a telephone subscriber (the twisted pair of the last mile) is taken to a digitizing IC as close to the customer’s

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Vout

Vout

Vin

Compressor

205

Vin

Expander

FIGURE 5.20 The transfer characteristic of a compressor and an expander.

premises as possible. The digital side of the IC connects to a four wire circuit which has separate go and return pairs. Telephone exchanges (now called switches) are digital computers that cannot handle analog signals. All telephone voice signals must be converted to digital form before they can be handled by a switch, so the conversion takes place close to the customer. The companding process improves the perceived SN ratio in the baseband channel for the quiet talker by increasing the number of steps in the quantizer at small signal levels. However, with a fixed number of steps (typically 255 in an 8-bit system) the steps must be larger for large signals. This increases the quantization noise that is present with large signals, and therefore lowers the SN ratio for the loud talker. The effect of companding is therefore to even out the impact of quantization noise over the dynamic range of the baseband signal. When baseband SN ratio is calculated from signal and noise powers taking companding into account, the SN ratio is relatively constant with signal level across the whole baseband. However, this is not what the listener perceives. When presented to a human ear, loud sounds appear to mask the increase in quantization noise at high signal levels, and the perceived SN is much better than the calculated values would indicate. With an equiprobable signal reaching the highest permitted signal level, the baseband SN of a companded PCM speech channel appears to be about 15 dB higher than the calculated value of 48 dB. This is called a subjective improvement in SN ratio, because the effect depends on the physiological characteristics of the human ear, not on calculations of signal power and noise power. The compact disc (CD) used for sound recordings is another example of a digital audio system. The CD is intended to reproduce music with high quality and therefore requires a much better quantization SN ratio in the analog sound output than a telephone channel. When a CD is recorded, a 16-bit linear quantizer is used, giving a baseband quantization SN of 96 dB. The dynamic range of the human ear is about 120 dB, but most sound reproduction systems have thermal noise SN ratios of less than 100 dB, so the quantization noise in a CD is inaudible. Linear encoding is more accurate than companding, where a match is needed between the compressor and the expander. Consequently companding is not used when high quality sound reproduction is required. CDs are recorded in stereo using 44 kHz sampling and 16-bit words, giving a bit rate of 1.408 Mbps. The actual bit rate of the recorded material on the CD is much higher because error detection and correction coding is applied to the digital data stream before it is recorded.

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Digital telephone signals are transmitted at 64 kbps or lower rates, because this is sufficient to achieve speaker recognition and intelligible speech. There is no attempt to make a telephone channel a hi-fi sound system because this would require the transmission of a much higher bit rate, and therefore fewer channels per megabit.

Signal-to-Noise Ratio in Digital Voice Systems Thermal noise causes bit errors in digital communication links, as discussed in Section 5.4. In a PCM system, the digital data are converted back to a baseband analog signal at the receiver. We need to know the signal-to-noise ratio that corresponds to a given probability of a bit error occurring in the digital data at the receiver. The analysis is straightforward when only one bit error occurs in each PCM word; provided the BER is below 10 4 and we have 7 or 8 bits per word, the likelihood of two bit errors occurring in one word is very small. We will assume this to be the case in the analysis that follows. When a bit is in error in a PCM word, the recovered sample of the baseband analog signal will be at the wrong level. This adds an impulse of amplitude Vn and duration Ts, the period of one sample, to the true analog signal. The bit that is in error may be located in any position in the PCM word. If the least significant bit is in error, Vn is small and equal to , the analog to digital converter step size; if it is the most significant bit that is in error, Vn will be large and equal to 2N 1  . The resulting average signal-to-noise ratio in the baseband analog channel caused by random bit errors is (SN)t, the subscript t standing for thermal noise, which is assumed to be the cause of the random bit errors, where1 1SN2 t 

3L2 1  41L2 12Pb

(5.82)

We can combine thermal noise from Eq. (5.82) with quantization noise from Eq. (5.81) to find the overall PCM output signal-to-noise ratio 1SN2 PCM 

22 N 1  4 Pb  22 N

(5.83)

When Pb is small, for example, less than 10 4, the quantization noise will dominate and (SN)PCM  22N. For N  8 bits, this gives SN  48 dB. When Pb is larger, thermal noise dominates; for example, with Pb  10 4 and N  8 and (SN)  14Pb  34 dB. Figure 5.21 shows the transition from thermal noise to quantization noise as the predominant noise source as the probability of a bit error decreases, for PCM systems using 7 and 8 bit words, with linear quantization. Clearly, for BERs below 10 6, quantization noise is dominant. However, when the bit error rate is around 10 5 there is a bit error roughly once per second in a 64-kbps PCM voice link. The bit error will be heard as a click, so the calculation of the thermal noise SN is not meaningful. Since most PCM links operate with BERs below 10 6 most of the time, it is worthwhile using nonlinear encoding (companding) to improve the baseband (SN) by reducing quantization noise. The curves for baseband SN in a digital voice channel shown in Figure 5.21 by the solid lines are for a system with linear encoding. When nonlinear encoding is used, higher perceived baseband SN ratios are achieved as shown by the dotted lines provided that thermal noise is not present. As soon as the probability of bit errors exceeds 10 6, thermal noise becomes the dominant source of noise in the channel and the subjective improvement obtained by companding disappears. Figure 5.21 is plotted on a baseline of BER. Comparison with Figure 5.13 in Section 5.4 shows that BER rises very quickly when the CN of a digital radio system falls.

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Baseband S/N ratio (dB)

60

207

With companding

50 Without companding 40

30 10−4

10−5

10−6 Bit error rate

10−7

10−8

FIGURE 5.21 Baseband SN in digital speech system using 8-bit PCM. Signal values between zero and maximum are equally probable.

For example, a QPSK satellite link with an overall receiver (CN)0  17 dB and an implementation margin of 1 dB gives a BER of 10 10. In a speech channel using 64 kbps bit rate, a BER of 10 10 gives one bit error every 2 days; the channel is essentially error free and quantization noise will set the channel SN ratio at a subjective value of 63 dB with -law or A-law companding. If the receiver CN ratio falls by 3 dB, due to rain attenuation in the path, for example, the BER will increase to 4  10 6, giving a thermal noise SN ratio 14Pb  48 dB. This is much lower than the companded quantization SN ratio of 63 dB. The link therefore operates in one of two regimes. For about 99% of the time on a typical satellite communication link, when clear air conditions prevail, there are no bit errors and the baseband SN ratio will be 63 dB. For the remaining 1% of the time when attenuation occurs on the link, the baseband SN will be below 63 dB, but will only fall below 40 dB for a very small fraction of the time when the BER exceeds 4  10 4. The techniques described in Chapter 4 are used to design satellite communication links so that bit error rates can be maintained above 4  10 4 for all but a very small percentage of the time. Mobile satellite systems are based on narrow bandwidth channels, and send voice signals at bit rates as low as 2.4 kbps. Typical bit rates for satellite telephone channels are around 4.8 kbps14. The reduction in bit rate from a standard 64-kbps digital voice channel is achieved using speech compression techniques. Analysis of speech waveforms shows that vowel sounds tend to have frequencies that are usually below 500 Hz in speech, and last for 50 ms or longer. Consonant have a much wider spectral content but last for shorter periods. There are also gaps between words and sentences. This allows considerable room to compress speech by transmitting what amounts to a spectral analysis of the waveform, rather than the waveform itself. At the simplest level, for example, a 50 ms vowel sound at a fundamental frequency of 200 Hz can be described as a sine wave at 200 Hz with a specific amplitude, with a number of harmonics of specified amplitude, lasting for a duration of T milliseconds. This information can be sent as 10 words of 8 bits giving a total of 80 bits, instead of 400 words of 8 bits that result from 8-kHz sampling of the waveform. At the receiving end of the link, a vocoder reconstructs the signal from the information sent over the link. The above example is oversimplified, but it illustrates that a considerable reduction in the transmission rate of digital speech signals is possible. The most widely used speech compression technique is linear predictive encoding (LPE). Speech compression is

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achieved by the use of digital signal processing (DSP) integrated circuits. The DSP IC contains a fast microprocessor and a large memory, and may be executing many millions of operations a second. The analog speech waveform is sampled and digitized, typically at 64 kbps for telephony, and then processed by the DSP compression IC to reduce the bit rate. The low bit rate signal is transmitted to a decoding IC in the receiver which regenerates the 64 kbps bit stream and thus recovers the original speech waveform. The challenge in developing speech compression systems is to provide a natural sounding voice at the receive end of the link which works well in any language. Once the compression technique is reduced to a pair of ICs, albeit very complex and fast devices, the integrated circuits can be manufactured in large quantities at reasonable prices. Systems designers can simply insert a pair of DSP devices in the transmitter and receiver to reduce the bit rate on the radio link.

Digital Television Digital television is rapidly replacing analog TV, with direct broadcast satellite television, digital cable TV, and high definition television (HDTV) all using digital transmission. A television signal is made up of two separate parts: the video signal which creates the picture at the receiver, and the audio part that carries the accompanying sound signals. The video and audio signals are digitized separately, and then multiplexed into a serial bit stream for transmission. The audio signal is typically digitized with more bits than a telephone channel to provide sound of good quality. The video signal bandwidth for a TV signal is large: 4.2 MHz for the NTSC system and 5.25 MHz for the PAL (phase alternate line) system. A NTSC video signal sampled at 10 MHz and encoded with 8-bit words produces a bit stream at 80 Mbps. Transmission of this digital TV signal requires a bandwidth of 50 MHz using QPSK and RRC filters with   0.25. This is a large RF bandwidth, which would require the whole of one 54-MHz Ku-band transponder, making transmission of full bandwidth digital TV very expensive. To overcome the high bit rates required when an NTSC or PAL video signal is sampled and digitized, compression techniques have been developed that reduce the bit rate by a factor of 12 or more. The most important video compression techniques is known as MPEG 2, where MPEG stands for the Motion Pictures Experts Group, an industry standards body. MPEG compression techniques are used in high definition television, digital videodiscs, and direct broadcast satellite TV. The MPEG 2 system divides the picture into 8 by 8 pixel blocks and takes a discrete cosine transform (DCT) of each block. Only the significant coefficients of the DCT are then transmitted, which greatly reduces the bit rate to send each 8  8 pixel block. MPEG 2 also uses frame-to-frame comparison of the video signal to determine which blocks within each frame need to be transmitted because a change has occurred between frames. Compression factors of up to 75 can be achieved with full motion video using MPEG 2 techniques. If degradation of the picture is allowed, even higher compression ratios are possible, but definition is degraded and motion becomes blurred. Similar compression techniques are used for the transmission of pictures over the Internet using JPEG (Joint Picture Experts Group) standards. The human eye is less sensitive to noise and distortion in a video signal than the human ear is to distortions in audio signals. This allows video and picture transmissions to use fewer bits and higher quantization noise levels than audio transmissions. One major disadvantage of MPEG 2 compression is that it introduces a delay of about 1.5 s in the transmission of the TV signal. The delay is of no significance in recorded material, but is excessive for two-way live transmissions.

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The digital video standard used in direct broadcast satellite television uses an average transmission rate of 6.2 Mbps for live video and 1.6 Mbps for prerecorded material. This allows up to 10 video signals to be transmitted by a single 20 MHz bandwidth transponder. For more details of direct broadcast satellite television see Chapter 11.

5.6

TIME DIVISION MULTIPLEXING In time division multiplexing (TDM) a group of signals take turns using a channel. This contrasts to frequency division multiplexing, presented earlier, where the signals occupy the channel at the same time but on different frequencies. Since digital signals are precisely timed and consist of groups of short pulses with relatively long intervals between them, TDM is the natural way for combining digital signals for transmission.

TDM Terminology: The U.S. T1 24-Channel System In this section we will describe the U.S. Telephone T1 24-channel TDM system and use it to introduce the terminology of time division multiplexing. While T1 was developed for terrestrial microwave circuits, it is also used on digital satellite links. We present it here as a convenient vehicle for explaining TDM operation. In pure TDMA systems, the multiplexing blends into the multiple access process. A TDM system transmits a digital word from each channel in turn. Each word is a group of bits that identify the quantization interval of the current sample. The words are organized into frames. One frame contains one word from each channel plus some synchronizing information that serves to identify the start of the frame. A frame is then a series of bits numbered sequentially from zero that carry synchronizing information plus the quantized values of one sample from each channel. The bits within a frame are grouped into slots. A slot contains all the bits from a common source. The slots within a frame are numbered sequentially from zero. In the T1 system illustrated in Figure 5.22, there are 25 slots, numbered 0 through 24. Slot 0 contains a single bit and carries synchronization information. Slots 1 through 24 each contain 8 bits and carry telephone channels 1 through 24. Thus a T1 frame contains 1  (8  24)  193 bits. Standard digital telephone (PCM) systems sample at an 8-kHz rate. Said another way, they transmit one sample of each channel every 125 s. This is the frame interval; the frame rate is the reciprocal of the frame interval and is always equal to the sampling rate, or an integer multiple of the sampling rate. The T1 system must transmit 193 bits in the 125-s frame interval; hence its bit rate is 193 bits divided by 125 s or 1.544 megabits per second (Mbps). Frame synchronization is established and maintained by transmitting a known bit pattern in slot 0. This bit pattern constitutes what is called the frame alignment word (FAW ). When frames are transmitted sequentially, slot 0 of one frame is at the end of the previous frame. The bits of the FAW are therefore both start of frame (SOF) and

Slots

0

1

1 bit 8 bits

2

3

4

24

8 bits

8 bits

8 bits

8 bits

193 bits

Time

FIGURE 5.22 Slot organization of one T1 frame.

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end of frame (EOF) markers. There are several different FAWs that can be used in T1 systems. One T1 FAW is 100011011100; it contains 12 bits and requires 12 frames for transmission. The group of frames that transmit the FAW make up a superframe. Thus in the first frame of a superframe, slot 0 contains a 1. In the second frame slot 0 contains a 0, and so on. At this point let us summarize the operation of a T1 digital multiplexer. It receives digitized voice signals from 24 telephone channels, which, for now, we will assume are perfectly synchronized with each other and have exactly the same bit rate. The 8-bit word samples from each channel flow into buffers and wait for the multiplexer to read them out. The multiplexer reads them out and inserts them into outgoing frames as follows. The first frame is transmitted by sending a 1(the first bit of the FAW), then one 8-bit word from channel 1, then one 8-bit word from channel 2, and so forth through one 8-bit word from channel 24. Then a new group of samples flows into the buffers. The multiplexer forms frame 2 by sending a 0 (the second bit of the FAW) followed by the words from each channel. A third group of samples enters and the process continues. When the buffers have been filled and emptied 12 times, one superframe has been sent and the multiplexer begins a new FAW. At the receiving end of the link a demultiplexer must sort out the bits in each frame and route the appropriate words to each outgoing channel. It must also keep track of the number of the frame (within the superframe) that it is receiving. We may visualize the demultiplexing process by assuming that the incoming bits are clocked serially into a shift register. At the instant the last bit has entered the register, its contents match the bits in the frame of Figure 5.22. The multiplexer then does a parallel transfer of the bits for each channel into their own individual registers for subsequent serial transmission over their separate paths. At this point the digital channels have been demultiplexed. The frame alignment bits go into a shift register, which, at the completion of the superframe, should contain the frame alignment word. If it does not, then the multiplexer and demultiplexer are out of sync. When this occurs the demultiplexer seeks to regain alignment; the process is called reframing. In it the demultiplexer looks at candidate frame alignment bits until it finds one that is going through the requisite 100011011100 100011011100 100011011100 pattern. Obviously there is a trade-off between the number of frame alignment bits and the time required for refraining. If the entire FAW is transmitted within each frame, then refraining time is much shorter than when the FAW is transmitted with one bit per frame. Typical refraining times are about 50 ms,14 which is sufficiently short not to cause significant degradation of speech in the 24 channels when misalignment occurs. Along with the information carried in the 24 channels must go the signaling information necessary to route, initiate, and terminate the data channels. In the T1 system this information is transmitted by “robbing” the least significant bit from slots 6 and 12 and using these to form signaling channels A and B, respectively. Thus, channels 6 and 12 are actually carried by a form of 7-bit PCM and channels A and B convey signaling information at an 86  1.333 kbps rate. The T1 system was developed by AT&T in the 1960s to expand the capacity of twisted pair telephone cables between their telephone exchanges. A single twisted pair could carry 24 voice channels using a T1 system instead of one baseband voice channel. This allowed a 24-fold expansion of channel capacity between exchanges without the need to lay new cables. The development of many thousands of T1 links in the 1960s and 1970s provided AT&T with a capability to transmit any digital signal at 1.544 Mbps. As a result, 1.544 Mbps became a standard transmission rate for any digital signal, not just voice. When used for data, a 1.544 Mbps link is officially called a DS-1,

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TABLE 5.2 North American T-Carrier Digital Transmission Standards System designation

Bit rate (Mbps)

Voice channels

Digital signal

T1 T1C T1D T1G T2 T3 T4 T5

1.544 3.152 3.152 6.443 6.443 44.736 274.176 560.160

24 48 48 96 96 672 4,032 8,064

DS-1 DS-1C DS-1C DS-2 DS-2 DS-3 DS-4 DS-5

rather than a T1 system. However, with the growth of the Internet and the widespread use of high-speed digital links for Internet access, T1 and fractional T1 (12 T1 and 14 T1) links are in common use. The capacity of T1 and higher rate links used for digital speech transmission can be increased by the use of a compression technique known as differential PCM (DPCM). In DPCM the difference between adjacent 8-bit samples of the speech waveform are transmitted, rather than the samples themselves. An improved form of DPCM, adaptive differential PCM (ADPCM) can reduce the transmission rate for digital speech to 32 or 16 kbps. This can double or quadruple the capacity of a T1 link to 48 or 96 speech channels17. A DS-0 link runs at 64 kbps and there is a heirarchy of bit rates from DS-0 through DS-5, at 560.160 Mbps which forms the North American digital transmission standard1. Table 5.2 shows the T-system heirachy for voice and data circuits. Although the T-system was developed for terrestrial circuits, it is often important for a satellite system to be able to carry data between two terrestrial circuits, so it must be able to operate at the standard T-n rates. There are very few satellite transponders that can carry a T-4 or T-5 signal at 274 or 560 Mbps.

Other TDM Systems Outside the United States, most countries use digital transmission standards recommended by the ITU-T (formerly CCITT), the international body that coordinates telephone system standards. The ITU-T has recommended a standard 30 voice channel system with a bit rate of 2.048 Mbps. The system has 32 channels running at 64 kbps each, with two channels reserved for signaling and synchronization. For details on their slot and bit organization the reader should consult reference [1]. The rapid development of optical fiber systems allowed very high-speed digital signals to be transmitted over long distances. Optical fibers have very large bandwidths compared to radio links, and transmission has been demonstrated at rates exceeding 10 Gbps. The most widely used transmission rate seems to be OC-48, which transmits digital data at 2488 Mbps. Since optical fiber cables never contain a single fiber, but more usually several hundred fibers, the capacity of an optical cable system is very large. The largest and heaviest GEO satellites in orbit in 2000 could carry less than the equivalent of four OC-28 circuits. For point-to-point communication, satellites cannot compete with optical fibers; that is why the majority of earnings from commercial operation of satellites comes

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TABLE 5.3 System OC-1 OC-3 OC-9 OC-12 OC-18 OC-24 OC-36 OC-48

Fiber-optic Cable Transmission Standards

Bit rate (Mbps)

64-kbps Voice channel capacity

51.84 155.52 466.56 622.08 933.12 1,244.16 1,866.24 2,488.32

672 2,016 6,048 8,064 12,096 16,128 24,192 32,256

from broadcasting or point to multipoint transmission. Table 5.3 shows the hierarchy of digital signals transmitted using fiber-optic cables. Interconnection of satellite links and terrestrial circuits is feasible at rates up to OC-3, especially with the later generation of Ka-band GEO satellites.

Channel Synchronization in TDM Our explanation of the TI system made the tacit assumption that all 24 incoming PCM channels were synchronized with each other and running at the same bit rate. This condition would hold if the voice channels had reached the originating earth station in analog form and had been digitized by modulators running on a common clock. But if the channels came into the station in digital form, their synchronization would not be guaranteed. They may be resynchronized for TDM transmission by a technique called pulse stuffing1,13. In pulse stuffing the incoming words for each channel flow into an elastic buffer. There is one such buffer per channel, and each buffer can hold several words. The multiplexer reads words out of the buffer slightly faster than they come in. Periodically the multiplexer will go to the buffer and find less than a full word remaining. When that happens it inserts a dummy word called a stuff word into the frame in place of the word it would have taken from the buffer. At the same time it places a message on the signaling channel that states that a stuff word has been inserted. When the demultiplexer at the other end of the link receives the message it ignores the stuff word. When it is time for the next frame to be sent the buffer will have more than a full word waiting for transmission. Pulse stuffing is normal on satellite links that transmit digital signals between different terrestrial TDM systems, because the TDM systems will not be synchronized. The satellite link is run at a bit rate slightly higher than either of the terrestrial TDM systems that it joins. Stuffing bits and words are added to the satellite data stream as needed to fill empty bit and word spaces.

5.7

SUMMARY

Frequency modulation is used for analog signal transmissions in satellite links, primarily for video distribution to older receiving equipment at cable TV head ends. Wideband FM is used to obtain an improvement in SN ratio relative to the received CN ratio. An

FM demodulator is characterized by a threshold. Provided that a satellite link’s overall carrier-to-noise ratio (CN) is above this threshold, the signal-tonoise ratio of the demodulated signal be significantly greater than the incoming (CN). Additional

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REFERENCES

improvement in baseband SN ratio may be obtained through pre-emphasis and de-emphasis. De-emphasis decreases the noise power output of an FM demodulator; pre-emphasis distorts the transmitted baseband signal before transmission to compensate for the deemphasis at the downlink earth station. In analog television (TV) transmission by satellite, the baseband video signal and one or two audio subcarriers constitute a composite video signal that frequency modulates an uplink carrier. This system requires a very wide bandwidth for transmission (usually either a full transponder or a half transponder), but it provides the FM improvement necessary to achieve required (SN) values. Digital modulation is obviously the modulation of choice for transmitting digital data. Digitized analog signals may conveniently share a channel with digital data, allowing a link to carry a varying mix of voice and data traffic. While baseband digital signals are often visualized as rectangular voltage pulses, careful pulse shaping is required to prevent intersymbol interference (ISI) and to permit reasonably distortionless transmission through the limited band width of a transponder. Square root raised cosine (RRC) filters are used in the design of digital radio links to create the required zero ISI waveforms at the receiver. The RRC filter does not exist, so real filters which approximate the RRC filter shape must be used, with consequent nonzero ISI at the receiver output. Equalizers must also be used in digital transmitters and receivers to compensate for changes in signal spectra that can cause ISI. The common digital modulation schemes used on digital satellite links are binary phase shift keying (BPSK) and quadrature phase shift keying (QPSK). In these modulations, an incoming data stream sets the phase of a sinusoidal carrier to one of two (BPSK) or four (QPSK) values. The performance of a BPSK or QPSK link is described by its bit error rate. Digital

213

links are designed to meet bit error rate requirements in the same way as analog links are designed to deliver minimum SN values. Analog signals must be digitized for transmission over a digital link. This involves sampling the signal at a rate that is at least twice the highest frequency present and converting the sample values to digital words. Standard practice with telephone channels is to use nonuniform quantization with a sampling rate of 8 kHz, and to transmit 8-bit words giving a serial transmit bit rate of 64 kbps. This system for transmitting digital speech is often known by the old name of PCM, for pulse code modulation. Adaptive differential PCM (ADPCM) allows the speech to be transmitted at 32 or 16 kbps. Linear predictive encoding can further reduce the bit rate of digital speech signals to 4.8 kbps and is used in some LEO satellite telephone systems. Television signals are transmitted digitally using compression techniques such as MPEG 2 to reduce the bit rate to a manageable level. Digital signals from different channels are interleaved for transmission through time division multiplexing (TDM). Data, or digitized samples of analog signals from each channel, are transmitted in turn. The time interval in which one sample or word from each channel is sent is called a frame. Channels are identified by their position in the frame and individual frames are identified by the presence of synchronization bits that repeat with a known pattern. Important TDM standards are the North American T-system developed by AT&T for the transmission of digital telephony on terrestrial circuits, and now adopted for digital data transmission using fractions and multiples of the 1.544 Mbps T-1 rate, the ITU-T system with 30 digital speech channels and two signaling channels, using a bit rate of 2.048 Mbps, and the optical fiber transmission standards OC-1 through OC-48 with bit rates from 51.84 to 2,488.32 Mbps.

REFERENCES 1. L. W. COUCH, Digital and Analog Communication Systems, Prentice-Hall, Englewood Cliffs, NJ, 6th Ed., 1998. 2. SIMON S. HAYKIN, Digital Communications, John Wiley & Sons, New York, 1988. 3. FERREL G. STREMLER, Introduction to Communication Systems, Addison-Wesley, Reading, MA, 3rd Ed., 1984. 4. H. TAUB and D. L. SCHILLING, Principles of Communications Systems, McGraw-Hill, New York, 1971. 5. W. H. BRAUN and J. E. KEIGLER, “RCA Satellite Networks: High Technology and Low User Cost,”

6. 7. 8. 9. 10.

Proceedings of the IEEE, 72, 1483 – 1505 (November 1984). FTHENAKIS, EMMANUEL, Manual of Satellite Communications, McGraw-Hill, New York, 1984. K. S. SHAMNUGAM, Digital and Analog Communication Systems, John Wiley & Sons, New York, 1979. DAVIDOFF, ed., Amateur Satellite Handbook, American Radio Relay League, 225 Main Street, Newington, CT, 2000. H. NYQUIST, “Certain Topics in Telegraph Transmission Theory,” AIEE Transactions, 47, (April 1928). LEON W. COUCH, Modern Communication Systems, Prentice-Hall, Englewood Cliffs, NJ, 1995.

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11. W. C. LINDSEY and M. K. SIMON, Telecommunication Systems Engineering, Prentice-Hall, Englewood Cliffs, NJ, 1973. 12. V. K. BHARGAVA, D. HACCOUN, R. MATYAS, and N. NUSPL, Digital Communications by Satellite, John Wiley & Sons, New York, 1981. 13. J. J. SPILKER, Jr., Digital Communications by Satellite, Prentice-Hall, Englewood Cliffs, NJ, 1977. 14. www.globalstar.com

15. B. HONARY and G. MARKARIAN, Trellis Decoding of Block Codes, Kluwer, Dordrecht, 1984. 16. E. BIGLIERI, D. DIVSALAR, P. J. MCLANE, and P. SIMON, Introduction to Trellis Coded Modulation and Applications, Macmillan, New York, 1991. 17. I. A. GLOVER and P. M. GRANT, Digital Communications, Prentice-Hall, Englewood Cliffs, NJ, 1998. 18. www.hns.com/spacway 19. www.cyberstar.com

PROBLEMS 1. A C-band satellite link sends a single NTSC-TV signal through a 36-MHz transponder on a C-band GEO satellite. The NTSC video signal is modulated onto the carrier using wideband frequency modulation, and the bandwidth of the transmitted RF signal is 32 MHz. The baseband bandwidth of the TV signal is 4.2 MHz. a. Calculate the peak frequency deviation of the FM carrier using Carson’s rule. b. Calculate the unweighted FM improvement factor for the video signal. c. The overall CN in an earth station receiving the FM-TV transmission is 17 dB. What is the unweighted video SN ratio at baseband? d. De-emphasis and weighting factors improve the quality of the CN ratio by a subjective factor of 17 dB. What is the weighted SN of the baseband video signal? 2. When overall CN is sufficiently high, it is possible to transmits two FM-TV signals in one 36-MHz transponder. The signal-to-noise ratio improvement is reduced when two TV signals are transmitted rather than one because the frequency deviation must be reduced. Two NTSC FM-TV signals are transmitted through a 36-MHz bandwidth transponder. The bandwidth of each signal is 16 MHz. a. Calculate the peak frequency deviation of the FM signal using Carson’s rule. b. Calculate the unweighted SN in the baseband video bandwidth of 4.2 MHz for an overall CN ratio in the earth station receiver of (CN)0. c. What value must (CN)0 have to ensure that the unweighted (SN) of the video signal is 33 dB? d. Use the value of (CN)0 you found in part (c) above to find the baseband video SN ratio in clear air conditions. The de-emphasis and subjective weighting factors for the video signal total 17 dB. If the value of (CN)0 at the earth station receiver falls by 4 dB because of rain in the downlink path, what is the weighted baseband video SN? How would you rate the quality of the video signal?

3. In Problem 2, two NTSC video signals are transmitted as FM carriers in a bandwidth of 36 MHz. Each FM carrier occupies a bandwidth of 16 MHz. A digital T1 carrier with a bandwidth of 2.0 MHz can be sent through the same transponder by using a gap between the two FM carriers. Some of the transponder power must be devoted to the T1 carrier, with the result that the FM-TV carriers have reduced CN at the earth station and lower video SN at baseband. This question asks you to determine the reduction in video SN. You will need to solve Problem 2 before attempting this problem. a. The power at the output of the transponder must be shared between the three RF signals in proportion to bandwidth occupied by each signal. For convenience, assume that the transponder radiates a total power of 20 W. Calculate the power allocated to each signal when only two FM-TV signals are transmitted, and when all three signals are transmitted. b. Using the results from part (a) above, determine the reduction in CN of the FM-TV signals. Hence find the reduction in baseband video SN and the new value of unweighted video SN ratio, based on results from part (d) of Problem 2. c. What is the overall CN ratio at the earth station receiver for the T1 carrier? 4. An NTSC-TV video signal with a baseband bandwidth of 4.2 MHz is modulated by FM onto an RF carrier with a peak frequency deviation of 10 MHz. a. What is the bandwidth of the FM signal according to Carson’s rule? b. The video signal in part (a) above is transmitted through the transponder to another earth station where the CN in the receiver is 19.8 dB in clear air in the bandwidth you calculated in part (a) above. If subjective improvement factors for this video signal are P  8 dB, Q  7 dB, what is the baseband video SN? c. The transponder is reconfigured to carry two video signals, each occupying 18 MHz, which are received with CN  18.0 dB. If only the frequency deviation

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and CN change for the two FM signals, compared to the case in part (b) above, what is the SN of each baseband video signal? 5. A satellite telemetry link operating in S band uses frequency modulation to transmit the value of an analog voltage on the satellite to a receiving earth station. The voltage has a range from 1.0 to 1.0 V, and a maximum frequency of 1000 Hz. The FM modulator on the satellite has a constant of 10,000 Hz per volt. At the receiving earth station the CN ratio of this signal is 10 dB measured in the Carson’s rule bandwidth, and is 3 dB above the FM threshold of the FM demodulator. a. What is the Carson’s rule bandwidth for the FM signal? b. What is the baseband SN ratio at the earth station receiver output for the recovered analog signal? 6. A satellite link has an RF channel with a bandwidth 2.0 MHz. The transmitter and receiver have RRC filters with   0.5. What is correct symbol rate (pulse rate) for this link? 7. A Ku-band satellite uplink has a carrier frequency of 14.125 MHz and carries a symbol stream at Rs  16 Msps. The transmitter and receiver have ideal RRC filters with   0.25. What is bandwidth occupied by RF signal, and what is the frequency range of the transmitted RF signal? 8. A T1 data transmission system transmits data at 1.544 Mbps over a GEO satellite link. At the receiving terminal the clear air value of overall (CN)0 is 16.0 dB. The modulation used on the link is BPSK and the implementation margin of the BPSK demodulator is 0.5 dB. a. Find the BER at the receiver output and the average time between errors. b. Rain affects the downlink from the satellite and the overall CN ratio in the receiver falls by 6.0 dB to 10.0 dB. What is the bit error rate now? 9. A satellite data transmission system transmits data from two T1 carriers as a single 3.088-Mbps bit stream using QPSK. The symbol rate on the link is 1.544 Msps. The satellite link uses ideal RRC filters with   0.25. At the receiving terminal the clear air value of overall (CN)0 is 16.0 dB and the implementation margin of the QPSK demodulator is 1 dB. a. What is the bandwidth occupied by this signal, and the noise bandwidth of the receiver for this signal? b. Find the BER at the receiver output and the average time between errors. c. Rain affects the downlink from the satellite and the overall CN ratio in the receiver falls by 6.0 dB to 10.0 dB. What is the bit error rate now?

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10. a. A 36 MHz bandwidth transponder is used to carry digital signals. A 20 MHz bandwidth in the transponder is occupied by a QPSK signal generated by a transmitter with ideal Nyquist filters with parameter   0.25. What is the symbol rate of the QPSK signal in Msps? What is the bit rate of the QPSK signal? b. Under clear air conditions, the overall (CN)0 ratio in the earth station receiver is 18.0 dB. If the QPSK demodulator has an implementation margin of 1.5 dB, what is the bit error rate of the baseband digital signal in clear air conditions? How often does a bit error occur. (Give your answer in days, hours, minutes, or seconds, as appropriate.) c. Under rain conditions, the overall (CN)0 ratio of the QPSK signal in part (a) above falls to 14.3 dB at a receiving station. What bit error rate would you expect in the recovered bit stream? How often does a bit error occur? 11. A satellite communication system is built as a star network with one large hub station and many remote small earth stations. The system operates at Ka band using the K9 geostationary satellite, and carries digital signals which may be voice, data, or compressed video. The K9 satellite has transponders with a bandwidth of 60 MHz that can be operated in either of two modes: as a bent pipe or with a 40 Msps QPSK baseband processor. The outbound link from the hub to the remote stations has an uplink from the hub station to the satellite that is the input of transponder 1. Signals from the hub are transmitted using a single TDM carrier and QPSK modulation with a symbol rate of 40 Msps. In the initial system design the remote earth stations use receivers capable of receiving 40-Msps QPSK signals. The transponder is operated in bent pipe mode with sufficient back-off to make it linear. The hub transmitter operates at an output power of 100 W, which gives CN  30 dB in the transponder in clear air, measured in the correct noise bandwidth of a 40 Msps QPSK receiver equipped with RRC filters having   0.4. In clear air conditions, the resulting CN of the earth station receiver, ignoring noise transmitted by the satellite, is 20 dB. The receiver has a QPSK demodulator with an implementation margin of 1.0 dB. For the purposes of this question you may assume that all transmitters and receivers in the network and on the K9 satellite have ideal RRC filters. a. Find the overall CN in the earth station receiver in clear air conditions and estimate the bit error rate for the recovered data signal, assuming that FEC is not used. What is the correct noise bandwidth for the earth station receiver that receives the QPSK signal, and what is the bit rate of the link?

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b. An uplink fade occurs which causes an attenuation of 10 dB between the hub station and the satellite. The transponder is operated in bent pipe mode. Find the overall CN in the remote earth station receiver and estimate the BER of the recovered data. c. An uplink fade occurs which causes an attenuation of 10 dB between the hub station and the satellite. Transponder 1 is switched to operate with the 40 Msps baseband processor. The QPSK demodulator in transponder 1 has an implementation margin of 1 dB. Find the overall CN in the remote earth station receiver and estimate the BER. d. Rainfall statistics for the location of the hub station show that attenuation at the uplink frequency will exceed 20 dB for 0.01% of an average year. If the hub station uses uplink power control to mitigate the effects of uplink rain attenuation, determine the maximum uplink transmitter power (in watts) that must be transmitted to ensure that the link BER does not exceed 10 6 at the remote earth station receiver output for 99.99% of an average year when: (i) A linear bent pipe transponder is used (ii) A 40-Msps QPSK baseband processing transponder is used. e. Discuss the value of UPC at the hub station transmitter in this application. Would you recommend a linear transponder or a baseband processing transponder be used on the K9 satellite? Give reasons for your answer. 12. A Ku-band VSAT station receives a TDM data stream at 1.544 Mbps from a GEO satellite. The modulation is QPSK and under clear air conditions the downlink CN in the VSAT receiver is 20 dB (ignoring noise from the satellite). The CN in the satellite transponder is 30 dB. A nearby terrestrial LOS link causes interference with the VSAT such that the carrier-to-interference ratio CI in the VSAT receiver is 19.6 dB. All CN and CI values are quoted for the optimum noise bandwidth of the VSAT receiver. The receiver uses ideal RRC filters with   0.4 and its QPSK demodulator has an implementation margin of 1 dB. a. What is the symbol rate of the QPSK signal and the noise bandwidth of the VSAT receiver? b. What is the clear air overall CN ratio in the VSAT receiver, assuming that the interference can be considered AWGN? What BER would you expect at the data output of the VSAT receiver assuming no FEC is applied to the signal? c. The system is redesigned and half rate FEC is added to the signal so that the bit rate at the transmitter is doubled, but transmitter power is not increased. In the receiver, the FEC decoder has a coding gain of 6 dB.

For the case when FEC is used, determine the overall CN ratio and the expected BER during a rain fade that causes the CN ratio of the received signal to fall by 5 dB but which does not attenuate the interfering signal. d. If the extra bandwidth to implement half rate FEC is available at the satellite, would you recommend that FEC be used in this case? Give reasons for your answer. e. What are the advantages and disadvantages of using forward error correction in satellite links? Illustrate your answer using the above example of a high data rate signal sent to a small earth terminal. 13. A T1 data transmission system transmits data at 1.544 Mbps over a GEO satellite link. At the receiving terminal the clear air value of overall (CN)0 is 16.0 dB. The modulation used on the link is BPSK and the implementation margin of the BPSK demodulator is 0.5 dB. a. Find the BER at the receiver output, and the time that elapses, on average, between bit errors. b. Rain affects the downlink from the satellite and the overall CN ratio in the receiver falls by 6.0 dB to 10.0 dB. What is the bit error rate now? What is the average time between bit errors? c. The modulation method is changed to QPSK and the bit rate is increased to 2  T1  3.088 Mbps, and the symbol rate on the link is 1.544 Msps (Mbaud). What is the bit error rate now? What is the average time between bit errors? d. Rain affects the downlink from the satellite and the overall CN ratio in the receiver falls by 6.0 dB to 10.0 dB. What is the bit error rate and average time between errors now for the QPSK link? e. Changing the modulation method to QPSK and increasing the bit rate to 3.088 Mbps is likely to lead to an unacceptably high bit error rate when the satellite downlink was affected by rain because the receiver (CN)0 will fall by 6 dB. We could retain a bit rate on the link of 1.544 Mbps when using QPSK by changing the transmitter and receiver RRC filters to operate at a symbol rate Rs  1.5442  0.772 Msps. What is the bit error rate and average time between errors now for the QPSK link? f. Rain affects the downlink from the satellite in part (e) above, and the overall CN ratio in the receiver falls by 6.0 dB to 10.0 dB. What is the bit error rate now for the QPSK link? 14. The baseband average SN ratio for a probability of bit error Pe (BER) with N-bit PCM is given by SN  10 log10 c

22N d dB 1  4Pe  22N

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a. The effective CN in a digital receiver with QPSK modulation is 15.6 dB under clear air conditions. What is the baseband SN for 8-bit PCM coded speech? b. In moderate rain conditions the effective CN falls to 13.6 dB. What is the baseband SN for the 8-bit PCM signal? c. In heavy rain, the effective CN falls to 11.6 dB. What is the baseband SN for the 8-bit PCM signal? d. The minimum acceptable baseband SN in a speech channel is usually set at 30 dB. What is the corresponding minimum allowable effective CN for a QPSK link carrying 8-bit PCM coded speech? 15. Direct Broadcast Satellite TV In this question you are asked to analyze the performance of the TV system using frequency modulation. The uplink station delivers a signal to the satellite which conforms to the following specification: Transponder and satellite characteristics Transponder bandwidth

25 MHz

(CN)up in 20-MHz noise bandwidth

24 dB

Saturated output power

200 W

Downlink frequency

12.5 GHz

Downlink antenna gain, on axis

39.0 dB

Atmospheric clear air loss

0.4 dB

All other losses

0.5 dB

Receive station parameters Antenna diameter

18 inches

Aperture efficiency

70%

Antenna noise temperature (clear air)

40 K

Receiver noise temperature 90 K a. The uplink master station transmits an NTSC video signal with a baseband bandwidth of 4.2 MHz to one transponder on the satellite using FM. The transponder is operated with 1 dB of output back-off and the FM signal occupies a bandwidth of 24 MHz. For an earth station with a high gain LNA, at a distance of 38,000 km from the satellite, on the 4 dB contour of the satellite antenna beam. Find: (i) The peak frequency deviation of the FM signal. (ii) The power at the input to the earth station LNA. (iii) The downlink (CN)dn in a noise bandwidth of 24 MHz. (iv) The overall (CN)0 in the earth station receiver. b. For the FM video signal in part (a) above: (i) The unweighted video SN ratio at the baseband output of the receiver,

217

(ii) The weighted SN ratio after pre-emphasis and subjective improvements are added, (iii) The link margin for the downlink given an FM threshold at 8.5 dB. c. Heavy rain affects the uplink to the satellite causing the CN in the transponder to fall to 18 dB. Assuming linear bent pipe operation of the transponder, find: (i) The overall (CN)0 ratio in the earth station receiver and (ii) The video SN ratio. Is this an acceptable SN for viewing a television picture? d. Heavy rain affects the downlink from the satellite causing 4 dB of rain attenuation. The uplink is operating in clear air conditions. Assuming a medium noise temperature of 270 K in rain and 100% coupling of sky noise into antenna noise temperature, find: (i) The new value for (CN)dn in the earth station receiver, (ii) The corresponding overall (CN)0 ratio in the earth station receiver, (iii) The video SN ratio. Is this an acceptable quality television picture? e. Draw a block diagram for the earth station receiver, showing only the parts that relate to reception and output of the NTSC video signal. Your block diagram must show the center frequency, gain, and bandwidth of each block, as appropriate. Do not specify filters with Q exceeding 50. 16. This problem examines the design and performance of a digital satellite communication link using a geostationary satellite with bent pipe transponders, used to distribute digital TV signals from one central (hub) earth station to many receiving stations throughout the United States. The link uses QPSK digital transmission at 20 Msps with half rate forward error correction. The half rate FEC gives a coding gain of 5.5 dB. The design requires that an overall CN ratio of 9.5 dB be met in the earth station receiver to ensure that noise in the video signal on the TV screen is held to an acceptable level. The uplink transmitter power and the receiving antenna gain and diameter must be determined. The available link margins for each of the systems must be found and the performance of the system analyzed when rain attenuation occurs in the satellite–earth paths. The system is specified in Table P.16. a. Uplink design Find the uplink transmitter power to achieve the required (CN)up  30 dB in the transponder in clear air atmospheric conditions. Find the noise power in the transponder for a noise bandwidth of 20 MHz,

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TABLE P.16

System and Satellite Specification

Ku-band satellite parameters Total RF output power Antenna gain, on axis, Ku band (transmit and receive) Receive system noise temperature Transponder saturated output power: Ku band Transponder bandwidth: Ku band Earth station receiver IF noise bandwidth Minimum permitted overall C/N in receiver Transponder HPA output backoff Transmitting Ku-band earth station Antenna diameter Aperture efficiency Uplink frequency Required C/N in Ku-band transponder Miscellaneous uplink losses Location: 2 dB contour of satellite receiving antenna Receiving Ku-band earth station Downlink frequency Receiver IF bandwidth Aperture efficiency Antenna noise temperature LNA noise temperature Required overall (C/N)0 in clear air Miscellaneous downlink losses Location: 3 dB contour of satellite transmitting antenna Rain attenuation and propagation factors at Ku-band Clear air attenuation Uplink 14.15 GHz Downlink 11.45 GHz Rain attenuation Uplink 0.01% of year Downlink 0.01% of year

and then add 30 dB to find the transponder input power level. Calculate the earth station transmit antenna gain, and the path loss at 14.15 GHz. Generate an uplink power budget and find the required power at the transponder input to meet the (CN)up  30 dB objective in the transponder. Don’t forget the various uplink losses. b. Downlink design Assume a high gain LNA and ignore the noise generated in other parts of the receiver. Calculate the downlink (CN)dn to give overall (CN)0  17 dB when (CN)up  30 dB. Hence find the receiver input power to give the required (CN)dn using a value of receiving antenna gain Gr.

3.2 kW 31 dB 500 80 54 20 9.5 1

K W MHz MHz dB dB

5m 68% 14.15 GHz 30 dB 0.3 dB

11.45 GHz 20 MHz 68% 30 K 110 K 17 dB 0.2 dB

0.7 dB 0.5 dB 6.0 dB 5.0 dB

Calculate the path loss at the downlink frequency of 11.15 GHz. Don’t forget the downlink losses. The transponder is operated with 1 dB output backoff. Find the transponder output power and then generate a downlink power budget. Hence find the receiving antenna gain Gr and diameter for a frequency of 11.45 GHz. This diameter antenna will provide the required (CN)0 in the earth station receiver under clear air conditions. c. Rain effects A practical system must continue to operate under adverse weather conditions, so we need a margin for rain attenuation and increase in sky noise temperature during rain. In the following section you will

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determine the margins available on the uplink and downlink to combat rain attenuation and increase in sky noise temperature. d. Uplink rain attenuation Under conditions of heavy rain, the Ku band path to the satellite suffers an attenuation of 6 dB for 0.01% of the year. We must find the uplink attenuation margin and decide whether uplink power control would improve system performance at Ku band. The uplink CN was 30 dB in clear air. With 6-dB uplink path attenuation, the CN in the transponder falls to 24 dB. (Rain on the earth has no effect on the satellite transponder system noise temperature.) Assume linear transponder characteristic and no uplink power control. Find the transponder output power with 6 dB of rain attenuation in the uplink. Hence find the overall (CN)0 in an uplink rain fade of 6 dB, and the link margin available on the uplink. Is this an adequate uplink margin, given the rain attenuation for most of the United States? e. Downlink attenuation and increase in sky noise in rain The 11.45 GHz path between the satellite and the receive station suffers rain attenuation exceeding 5 dB for 0.01% of the year. Assuming 100% coupling of sky noise into antenna noise, and 0.5 dB clear air gaseous attenuation, calculate the overall CN under these conditions. Assume that the uplink station is operating in clear air. Calculate the available downlink fade margin. Find the sky noise temperature that results from a total excess path attenuation of 5.5 dB (clear air attenuation plus rain attenuation); this is the new antenna temperature in rain, because we assumed 100% coupling between sky noise temperature and antenna temperature. Evaluate the change in received power and increase in system noise temperature in order to calculate the change in CN ratio for the downlink. In clear air, the atmospheric attenuation on the downlink is 0.5 dB. The corresponding sky noise temperature is approximately 0.5  7  35 K, which leads to the antenna temperature of 30 K given in the Ku-band system specification. When the rain causes 5 dB attenuation, the total path attenuation from the atmosphere and the rain is 5.5 dB. The sky noise will be much higher in rain. Find the increase in noise power caused by the increase in sky temperature. Hence find the new (CN)dn rain value with 5.5 dB attenuation on the downlink path. Find the overall CN by combining the clear air uplink (CN)up of 30 dB with the rain faded downlink (CN)dn rain to give overall (CN)0 in rain.

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Is the downlink link margin acceptable? If not, calculate the gain and diameter of an earth station antenna that will ensure an overall CN value that does meet the specification. f. Summarize your design for the Ku-band earth station and uplink and downlink. Compare the earth station receiving antenna diameter for the Ku-band system with the antenna for a similar C-band system. If the Ku-band antenna is larger (and therefore has a much higher gain) explain why. 17. A satellite communication system uses a single 54 MHz bandwidth Ku-band transponder to carry 400 two-way telephone conversations (800 RF channels) using analog modulation with single channel per carrier frequency modulation (SCPC-FM). The parameters of any one channel are: Voice channel bandwidth

100–3400 Hz

RF channel bandwidth

45 kHz

RF channel spacing

65 kHz

Downlink path loss (incl. atmos. loss) Satellite downlink antenna gain (on axis) Demodulator FM threshold

206.5 dB 29 dB 5 dB

The transponder has a saturated power output of 40 W, but is run with 3-dB output backoff to achieve near-linear operation. The uplink stations which transmit the SCPC-FM signals to the transponder achieve (CN)up  25 dB in the 45 kHz channel noise bandwidth of the earth station receiver. The system noise temperature of the receiving earth station is 120 K in clear air. a. Calculate the power per RF channel at the transponder output. b. Calculate the gain of the antenna at a receiving earth station that is located on the 3 dB contour of the satellite footprint which will provide an overall CN  10 dB in a receiver for a single RF channel with a noise bandwidth of 45 kHz, in clear air conditions. c. Calculate the diameter of the receiving antenna with a circular aperture having 65% aperture efficiency at a frequency of 11.5 GHz. d. The receiver applies a de-emphasis weighting of 6 dB to the recovered voice signal and a psophometric weighting of 2.5 dB. Calculate the weighted SN at the baseband output of the receiver. e. Comment on the performance of the system. Is the SN adequate in clear air? If the downlink fades by 5 dB because of rain, what is the SN at baseband? Is this acceptable for voice communications?

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18. In Problem 5, an analog voltage was transmitted from a satellite to earth using frequency modulation. The signal could have been sent digitally using a digital to analog converter and PSK modulation. This problem compares the performance of the digital link to the analog link of Problem 5. The digital link is allocated an RF bandwidth of 25 kHz, and uses BPSK modulation. At the receiving terminal, the CN ratio is 10 dB. The link has ideal RRC filters with   0.25 and the BPSK demodulator has an implementation margin of 0.5 dB. a. The analog voltage is sampled at 2.5 kHz and converted to a series of digital words with an analog to digital converter. Determine the maximum number of bits in each word and the average

quantization signal-to-noise ratio of the recovered analog signal. b. Find the BER for the recovered bit stream at the output of the BPSK demodulator, and hence calculate the average SN ratio in the analog voltage due to bit errors. c. Solve Problem 5 for the FM version of this link. Which link has the better performance? What changes should be made to the link with the poorer performance to make the SN ratios approximately equal for the FM and BPSK links? If an RF bandwidth of 50 kHz could be used for the BPSK signal, would the addition of half rate forward error correction with a coding gain of 6 dB improve the performance of the BPSK link?

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CHAPTER

6

MULTIPLE ACCESS

6.1

INTRODUCTION Satellites are always built with the intention that many users will share the bandwidth allocated to the satellite, allowing many separate communication links to be established through the satellite’s transponders. In a large geostationary satellite operated by an international carrier, the satellite can carry tens of thousands of simultaneous telephone conversations between people in many different parts of the world. A domestic satellite carrying television program material can broadcast its signals to thousands of cable television companies serving millions of homes. The ability of the satellite to carry many signals at the same time is known as multiple access. Multiple access allows the communication capacity of the satellite to be shared among a large number of earth stations, and to accommodate the different mixes of communication traffic that are transmitted by the earth stations. The basic form of multiple access employed by all communications satellites is the use of many transponders. A large GEO satellite may have a communication bandwidth of over 2000 MHz within an allocated spectrum of 500 MHz. Through frequency reuse with multiple antenna beams and orthogonal polarization, the spectrum can be reused several times over—as many as seven times in the case of INTELSAT IX satellites. The frequency spectrum used by the satellite is divided into smaller bandwidths which are allocated to transponders, allowing separate communication links to be established via the satellite on the basis of transmit frequency. Transponder bandwidths of 36, 54, and 72 MHz have been commonly employed on GEO satellites. The individual transponders may carry one signal, a single analog television program, for example, or hundreds of signals, as with mobile satellite telephone systems. The use of multiple transponders to divide up a frequency band is not considered as multiple access, although the reason for their use is to make it easier for different earth stations to share the available frequency spectrum efficiently. The signals that earth stations transmit to a satellite may differ widely in their character—voice, video, data, facsimile—but they can be sent through the same satellite using multiple access and multiplexing techniques1,2. Multiplexing is the process of combining a number of signals into a single signal, so that it can be processed by a single amplifier or transmitted over a single radio channel. Multiplexing can be done at baseband or at a radio frequency. The corresponding technique that recovers the individual signals is called demultiplexing. Multiplexing is a key feature of all commercial long-distance communication systems, and is part of the multiple access capability of all satellite communication systems. The designer of a satellite communication system must make decisions about the form of multiple access to be used. The multiple access technique will influence the capacity and flexibility of the satellite communication system, its cost, and its ability to earn revenue. The basic problem in any multiple access system is how to permit a changing group of earth stations to share a satellite in such a way that satellite communication 221

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Guard band Ch 1

Ch 2

Ch 3

Ch 4

Transponder bandwidth FDMA Ch 1

Ch 2

Ch 3

Frequency Guard time Ch 4

Frame period

Time

TDMA Power Station N Station 4 Station 3 Station 2 Station 1

Transponder bandwidth CDMA

Frequency

FIGURE 6.1 Multiple access techniques: FDMA, TDMA, and CDMA. Note that in the direct sequence form of CDMA shown here, all the channels overlap in both time and frequency.

capacity is maximized, bandwidth is used efficiently, flexibility is maintained, and cost to the user is minimized while revenue to the operator is maximized. The multiple access system should also allow for changing patterns of traffic over the 10 or 15 years of the expected lifetime of the satellite. Usually, all of these requirements cannot be satisfied at the same time and some may have to be traded off against others. There are three basic multiple access techniques, illustrated in Figure 6.1. In frequency division multiple access (FDMA) all users share the satellite at the same time, but each user transmits at a unique allocated frequency. This approach to sharing the frequency spectrum is familiar to us all, as it is the way that radio broadcasting has always shared the air waves. Each radio station is allocated a frequency and a bandwidth, and transmits its signals within that piece of the frequency spectrum. FDMA can be used with analog or digital signals. In time division multiple access (TDMA) each user is allocated a unique time slot at the satellite so that signals pass through the transponder sequentially. Because TDMA causes delays in transmission, it is used only with digital signals. In code division multiple access (CDMA) all users transmit to the satellite on the same frequency and at the same time. The earth stations transmit orthogonally coded spread spectrum signals that can be separated at the receiving earth station by correlation with the transmitted code. CDMA is inherently a digital technique. In each of the multiple access techniques, some unique property of the signal (frequency, time, or code) is used to label the transmission such that the wanted signal can be recovered at the receiving terminal in the presence of all other signals. The distinction between multiplexing and multiple access is sometimes blurred. Multiplexing applies to signals that are generated at one location, whereas multiple access refers to signals from a number of different geographical locations. For example, an earth station might use time division multiplexing (TDM) to create a high-speed digital data stream from many digital speech channels delivered to that earth station, and then modulate the data stream onto an RF carrier and transmit the carrier to the satellite. At the satellite, the carrier can share a transponder using time division multiple access (TDMA) or frequency division multiple access (FDMA) with other carriers from earth stations anywhere within the satellite’s coverage zone. The resulting signal is called TDM-TDMA or TDM-FDMA. Note the distinction between TDM and TDMA: signals at one earth station are combined by multiplexing, and then share a satellite transponder with signals from other earth stations by multiple access.

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In all three of the classic multiple access techniques, some resource is shared. If the proportion allocated to each earth station is fixed in advance, the system is called fixed access (FA) or preassigned access (PA). If the resource is allocated as needed depending on changing traffic conditions, the multiple access technique is called demand access (DA). Demand access blurs some of the distinctions between FDMA and TDMA, since stations in an FDMA-DA system transmit only when they have traffic. Demand access with FDMA is widely used in VSAT systems, where earth stations may have traffic to send only intermittently 3. Fixed assignment would be wasteful of transponder capacity, so demand assignment is used. Similarly, a group of earth stations may access part of the bandwidth of a transponder using TDMA, while other TDMA groups of earth stations share different sections of the transponder bandwidth. This approach has been used in both VSAT and mobile satellite systems. (See Chapter 9.) Demand assignment can also be used with CDMA to reduce the number of signals in the transponder at any one time. The Globalstar LEO mobile satellite system uses CDMA with demand assignment4. Systems which combine both FDMA and TDMA techniques are sometimes called hybrid multiple access schemes or multifrequency TDMA (MF-TDMA). In the sections that follow, we will first discuss FDMA, TDMA, and CDMA as fixed assignment schemes, and then cover demand access and hybrid multiple access.

6.2 FREQUENCY DIVISION MULTIPLE ACCESS (FDMA) Frequency division multiple access was the first multiple access technique used in satellite communication systems. When satellite communications began in the 1960s, most of the traffic carried by satellites was telephony. All signals were analog, and analog multiplexing was used at earth stations to combine large numbers of telephone channels into a single baseband signal that could be modulated onto a single RF carrier. Individual telephone channels can be shifted in frequency from baseband to a higher frequency so that they can be stacked into a group of channels using frequency division multiplexing (FDM). The process begins by limiting individual telephone channels to the frequency range 300–3400 Hz, and then frequency shifting 12 channels to the frequency range 60–108 kHz with 4-kHz spacing between channels by generating single sideband suppressed carrier signals with 12 carrier frequencies spaced 4 kHz apart. The 12 channels occupying 60–108 kHz are known as a basic group. Five basic groups can be frequency shifted to the range 60–300 kHz to make a 60-channel supergroup occupying a baseband bandwidth of 240 kHz. Supergroups can be stacked in the baseband to make up single signals that consist of 300, 600, 900, or 1800 multiplexed telephone channels. The analog FDM technique is now obsolete in the United States and many other countries, but it was the primary method of multiplexing telephone channels for transmission over terrestrial cable or microwave links for about 50 years. Early satellite systems used FDM to multiplex up to 1800 telephone channels into a wide baseband occupying up to 8 MHz, which was modulated onto an RF carrier using frequency modulation (FM). Appendix B describes the process of voice multiplexing using FDM techniques, and provides some details of satellite transmission of analog signals using FDM-FM-FDMA. The FDM-FM RF carrier was transmitted to the satellite, where it shared a transponder with other carriers using FDMA. The technique is known as FDM-FMFDMA, and was the preferred method for the transmission of telephone channels over Intelsat satellites for more than 20 years. The main advantage of FDMA is that filters can be used to separate signals. Filter technology was well understood when satellite

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Transponder 1 Transponder bandwidth 36 MHz 10 USA

3714

10 USA

10 Chile

3726

3738

Carrier frequency in MHz Transponder 2 Transponder bandwidth 36 MHz 20 USA

10 Chile

3759

3771

Carrier frequency in MHz FIGURE 6.2 Frequency plan for two C-band transponders using FDMA. The triangles are symbols representing the bandwidth occupied by the signals, not power spectral densities. The places and figures within the triangles are the transmitting station location and carrier RF bandwidth. Frequencies shown are for the downlink from the satellite.

communications began, and microwave filters were used in earth stations to separate the FDMA signals within a given transponder. In a fixed assignment system, each transmitting earth station was allocated a frequency and bandwidth for each group of signals it wished to send. Figure 6.2 shows a typical fixed assignment FDMA plan for two C-band transponders. The triangles represent RF carriers with the transmitting earth station and RF bandwidth shown inside the triangle. Frequencies shown are for the downlink from the satellite. Within each transmission, signals (primarily telephone channels) for different destinations are multiplexed using FDM. Typical Intelsat FDM carriers with a bandwidth of 10 MHz carried 132 to 252 telephone channels. If a small group of channels is intended for a given earth station, the entire carrier must be received and demultiplexed to recover those channels. Channels sent by the same carrier but intended for other earth stations are discarded. The 36 MHz transponder bandwidth can be used to send one or two television signals instead of hundreds of telephone channels. The use of microwave filters to separate channels made the fixed assignment approach to FDMA very inflexible. Changing the frequency assignment or bandwidth of any one transmitting earth station required retuning of the microwave filters at several receiving earth stations. The fixed assignment FDM-FM-FDMA scheme illustrated in Figure 6.2 also makes inefficient use of transponder bandwidth and satellite capacity. As an example, suppose an earth station in the west of the United States uses a Pacific Ocean GEO satellite to send telephone channels to earth stations in Korea, Japan, and Chile. The time difference between North America and the Pacific Rim countries means that the channels will be busy for only a few hours per day, and at a different time of day than the U.S.–Chile links. With fixed assignment, the frequencies and satellite capacity cannot be reallocated between routes, so much of the satellite capacity remains idle. Estimates of average loading of Intelsat satellites using fixed assignment are typically around 15%. It is not possible to achieve 100% loading of satellites used for international

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Transponder bandwidth 54 MHz

1

2

3

4

5

6

7

40 kHz digital speech channels

8

9

10 kHz guard bands

FIGURE 6.3 Illustration of a Ku-band transponder bandwidth filled with a large number of FDMA-SCPC digital speech channels. RF bandwidth of each channel is 40 kHz with 10 kHz guard bands between channels.

traffic, or even for domestic traffic in many cases. Demand assignment and single channel per carrier (SCPC) techniques allow higher loadings and therefore give satellite operators increased revenue. There has been a steady move away from fixed assignment systems as a result. Every earth station that operates in an FDMA network must have a separate IF receiver for each of the carriers that it wishes to receive. SCPC systems can have a very large number of carriers in one transponder; as a result, FDMA earth stations tend to have a very large number of IF receivers and demultiplexers which select individual carriers using narrowband IF filters. Figure 6.3 shows how the intermediate frequency bandwidth of a receiving earth station could be configured to receive 1000 digital speech channels, each with a bandwidth of 40 kHz from a 54 MHz wide Kuband transponder. The 10-kHz frequency spaces between the channels are called guard bands. Guard bands are essential in FDMA systems to allow the filters in the receivers to select individual channels without excessive interference from adjacent channels. All filters have a roll-off characteristic, which describes how rapidly a filter can change from near zero attenuation in its pass band to high attenuation in the stop band. Typically, guard bands of 10 to 25% of the channel bandwidth are needed to minimize adjacent channel interference. FDM-FM-FDMA was a telephone transmission technique well suited to analog telephone signals. Telephony has largely become digital, and frequency division multiplexing has been replaced by time division multiplexing. Digital speech is now used throughout telephone systems, so multiple telephone channels are always transmitted as a high-speed digital signal. The T1 or DS-1 carriers are examples of lower speed digital multiplexed carriers. Optical networks carry OC–48 digital signals at rates up to 2.7 Gbps and beyond. Appendix B discusses the techniques used in FDM-FMFDMA systems, and the calculation of system capacity using this multiple access scheme. Apart from the analog nature of FDM-FM-FDMA which has rendered it obsolete, it is a rather inflexible way to allocate satellite transponder capacity and is not easily adapted to demand access. FDMA is widely used as a method of sharing the bandwidth of satellite transponders. Ideally, a satellite would carry a very large number of transponders, each of which could be allocated to a single RF carrier. In the case of telephony, each transponder would have a bandwidth exactly matched to the RF spectrum of the transmitted telephone channel, with tight filtering to ensure that each signal can be separated from adjacent signals. This approach is impractical: thousands of transponders would be needed and the satellite could be used only for telephony. The builders and operators of satellites have

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historically shown a strong preference for wideband transponders that can carry any type of traffic—the bent pipe transponder that can carry voice, video, or data as the marketplace demands. As a result, transponders have always had wide bandwidths, with bandwidths of 36, 54, and 72 MHz commonly employed. When an earth station has a carrier that occupies less than the transponder bandwidth, FDMA can be used to allow that carrier to share the transponder with other carriers. Allocating a wideband transponder to a single narrow bandwidth signal is clearly wasteful, so FDMA is a widely used technique. When an earth station sends one signal on a carrier, the FDMA access technique is called single channel per carrier (SCPC). Thus a system in which a large number of small earth stations, such as mobile telephones, access a single transponder using FDMA is called a single channel per carrier frequency division multiple access scheme. Not surprisingly, this lengthy descriptor is abbreviated to SCPC-FDMA. Hybrid multiple access schemes can use time division multiplexing of baseband channels which are then modulated onto a single carrier. A number of earth stations can share a transponder using frequency division multiple access, giving a system known as TDM-SCPC-FDMA. Note that the sequence of abbreviations is baseband multiplexing technique first, then multiple access technique next. TDM-SCPC-FDMA is used by VSAT networks in which the earth stations carry more than one baseband signal. FDMA has a disadvantage in satellite communications systems when the satellite transponder has a nonlinear characteristic. Most satellite transponders use high-power amplifiers which are driven close to saturation, causing nonlinear operation. A transponder using a traveling wave tube amplifier (TWTA) is more prone to nonlinearity than one with a solid state high-power amplifier (SSHPA). Equalization at the transmitting station, in the form of predistortion of the transmitted signal, can sometimes be employed to linearize the transponder when fixed assignment is used. Linearization of solid-state and TWT HPAs on the satellite is also possible. Nonlinearity of the transponder HPA causes a reduction in the overall (CN)0 ratio at the receiving earth station when FDMA is used because intermodulation (IM) products are generated in the transponder. Some of the IM products will be within the transponder bandwidth and will cause interference. The IM products are treated as though they were thermal noise, adding to the total noise in the receiver of the receiving earth station.

Intermodulation Intermodulation products are generated whenever more than one signal is carried by a nonlinear device. Sometimes filtering can be used to remove the IM products, but if they are within the bandwidth of the transponder they cannot be filtered out. The saturation characteristic of a transponder can be modeled by a cubic curve to illustrate the generation of third-order intermodulation. Third-order IM is important because third-order IM products often have frequencies close to the signals that generate the intermodulation, and are therefore likely to be within the transponder bandwidth. To illustrate the generation of third-order intermodulation products, we will model the nonlinear characteristic of the transponder HPA with a cubic voltage relationship and apply two unmodulated carriers at frequencies f1 and f2 at the input of the amplifier Vout  AVin  b1Vin 2 3

(6.1)

V1 cos v1t  V2 cos v2t

(6.2)

where A W b. The amplifier input signal is

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The amplifier output signal is

Vout  AVin  b1Vin 2 3  AV1 cos v1t  AV2 cos v2t  b1V1 cos v1t  V2 cos v2t2 3 linear term

(6.3)

cubic term

The linear term simply amplifies the input signal by a voltage gain A. The cubic term, which will be denoted as V3out, can be expanded as V3out  1V1 cos v1t  V2 cos v2t2 3  b3V 13 cos3 v1t  V23 cos3 v2t  2V22 cos2 v2t  V2 cos v2t  2V22 cos2 v2t  V1 cos v1t4

(6.4)

The first two terms contain frequencies f1, f2, 3f1, and 3f2. The triple frequency components can be removed from the amplifier output with band-pass filters. The second two terms generate the third-order IM frequency components. We can expand the cosine squared terms using the trig identity cos2 x  12 [cos2x  1]. Hence the IM terms of interest become VIM  bV 21 bV 22  bV 21 bV 22

 V2 3cos v2t  1cos 2v1t  12 4   V1 3cos v1t  1cos 2v2t  12 4  V2 3cos v2t cos 2v1t  cos v2t4   V1 3cos v1t cos 2v2t  cos v1t4

(6.5)

The terms at frequencies f1 and f2 add to the wanted output of the amplifier, so the thirdorder intermodulation products are generated by the f1  2f2 and f2  2f1 terms. Using another trig identity cos x cos y  cos1x  y2  cos1x  y2 The output of the amplifier contains IM frequency components given by

VIM ¿  bV 21  V2 3cos 12v1t  v2t2  cos 12v1t  v2t2 4  bV 22  V1 3cos 12v2t  v1t2  cos 12v2t  v1t2 4

(6.6)

We can filter out the sum terms in Eq. (6.6), but the difference terms, with frequencies 2f1  f2 and 2f2  f1 may fall within the transponder bandwidth. These two terms are known as the third-order intermodulation products of the high-power amplifier, because they are the only ones likely to be present at the output of a transponder which incorporates a narrow bandpass filter at its output. Thus the third-order intermodulation products that are of concern are given by V3IM where V3IM  bV 21V2 cos 12v1t  v2t2  bV22V1 cos 12v2t  v1t2

(6.7)

The magnitude of the IM products depends on the parameter b, which describes the nonlinearity of the transponder, and the magnitude of the signals. The wanted signals at the transponder output, at frequencies f1 and f2, have magnitudes AV1 and AV2. The wanted output from the amplifier is Vout  AV1 cos v1t  AV2 cos v2t The total power of the wanted output from the HPA, referenced to a 1 ohm load, is therefore Pout  12 A2V 21  12 A2V 22  A2 1P1  P2 2 W

(6.8)

where P1 and P2 are the power levels of the wanted signals. The power of the IM products at the output of the HPA is PIM  2  1 12 b2V 61  12 b2V 62 2  b2 1P 31  P 32 2 W

(6.9)

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It can be seen that IM products increase in proportion to the cubes of the signal powers, with power levels that depend on the ratio (bA)2. The greater the nonlinearity of the amplifier (larger bA ratio), the larger the IM products.

Intermodulation Example Consider the case of a 36-MHz bandwidth C-band transponder which has an output spectrum for downlink signals in the frequency range 3705–3741 MHz. The transponder carries two unmodulated carriers at 3718 and 3728 MHz with equal magnitudes at the input to the HPA. Using Eq. (6.7), the output of the HPA will contain additional frequency components at frequencies f31  12  3718  37282  3708 MHz f32  12  3728  37182  3738 MHz Both of the IM frequencies are within the transponder bandwidth and will therefore be present in an earth station receiver that is set to the frequency of this transponder. The magnitude of the IM products will depend on the ratio bA, a measure of the nonlinearity of the HPA, and on the actual level of the two signals in the transponder. Now consider the case where the two signals carry modulation which spreads the signal energy into a bandwidth of 8 MHz around each carrier. Carrier 1 has frequencies 3714 to 3722 MHz and carrier 2 has frequencies 3726 to 3734 MHz. Denoting the band of frequencies occupied by the signals as fnlo to fnhi, the intermodulation products cover the frequency bands 12 f1lo  f2hi 2

to

12 f1hi  f2lo 2

12 f2lo  f1hi 2

and

to

12 f2hi  f1lo 2.

The IM products are spread over bandwidths (2B1  B2) and (2B2  B1). Hence the third-order IM products for this example cover these frequencies: 3706  3730 MHz and

3716  3740 MHz with bandwidths of 24 MHz.

The location of the 8 MHz wide signals and 24 MHz wide IM products is illustrated in Figure 6.4. The intermodulation products now interfere with both signals, and also cover the empty frequency space in the transponder. Third-order IM products grow rapidly as the output of the transponder increases toward saturation. Equation (6.9) shows that IM power increases as the cube of signal power: in decibel units, every 10 dB increase in signal power causes a 30 dB increase in IM product power. Consequently, the easiest way to reduce IM problems is to reduce the level of the signals in the HPA. The output power of an operating transponder is related to its

Transponder bandwidth 36 MHz Carrier 2

Carrier 1 IM products

3705

3714

IM products

3722 3726

3734

3741

Frequency in MHz FIGURE 6.4 Intermodulation between two C-band carriers in a transponder with third-order nonlinearity.

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saturated output power by output backoff. Backoff is measured in decibel units, so a transponder with a 50 W rated (saturated) output power operating with an output power of 25 W has output backoff of 17 dBW  14 dBW  3 dB. Intermodulation products are reduced by 9 dB when 3 dB backoff is applied, so any nonlinear transponder carrying more than one signal will usually have some backoff applied. Since a transponder is an amplifier, the output power level is controlled by the input power, and there is a saturated input power level corresponding to the saturated output level. When the transponder is operated with output backoff, the power level at its input is reduced by the input backoff. Because the transponder characteristics are not linear, input backoff is always larger than output backoff. Figure 6.5 illustrates the operating point and input and output backoff for a transponder with a nonlinear TWTA. The nonlinearity of the transponder causes the input and output backoff values to be unequal. In the example shown in Figure 6.5, the transponder saturates at an input power of 100 dBW. The transponder is operated at an input power of 102.2 dBW, giving an input backoff of 2.2 dB. The corresponding output backoff is 1.0 dB, giving an output power of 16 dBW (40 W), 10 W below the saturated output power of 50 W (17 dBW). Note that the TWTA has slightly different characteristics when operated with a single carrier and multiple carriers. The generation of intermodulation products when multiple carriers are present robs the wanted output of some of the transponder output power. For the nonlinearity shown in Figure 6.5, the reduction in output power is 0.6 dB at saturation. In the example above, both carriers had equal power. If the powers are unequal, the weaker signal may be swamped by intermod products from the stronger carrier. This can be seen from Eq. (6.9); the IM products that tend to affect Carrier 1 have voltages proportional to the square of the voltage of Carrier 2.

Pout 20

Outback backoff

Single carrier

Saturated output 17 dBW

Multicarrier

15 Transponder output power dBW

Operating point Input backoff

Input saturation 10 −105

−100

Pin

Transponder input power dBW FIGURE 6.5 Typical input–ouput characteristic of a transponder using a traveling wave tube amplifier (TWTA). To maintain quasi-linear operation and minimize intermodulation problems with multiple carriers, the transponder must be operated below its saturated output power level. In this example, the saturated output power from the transponder is 50 W (17 dBW) with a single carrier. The saturated output power is slightly lower with multiple carriers because some output power is converted to intermodulation products.

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Operation of a nonlinear transponder with multiple carriers requires careful balancing of the power levels of each carrier so that intermodulation products are evenly spread across the transponder’s bandwidth. Judicious spacing of the carriers can be used to place the highest intermods in gaps between carriers. The process is known as loading the transponder. Sophisticated computer programs are used by satellite operators to optimize the backoff level of a transponder such that intermodulation is minimized while output power is maximized. When a very large number of carriers access a transponder using FDMA, as might happen with a network of VSAT stations or a transponder used with mobile satellite telephones, the transponder must operate in a quasi-linear region of its characteristics. Quasi-linear means almost linear, either by equalization or by the application of a large output backoff. Earth station HPAs can also cause intermodulation if they carry multiple carriers and operate close to saturation. In large earth stations where multiple carriers are more likely to be transmitted, the HPA is often rated at a much higher level than the expected transmit power. This allows substantial backoff to be used, keeping the amplifier in its linear region. In the above analysis of third-order intermodulation, only two carriers were considered. If there are three (or more) carriers present in a nonlinear transponder, intermodulation products at frequencies such as f1  f2  f3 can be generated that are likely to be within the transponder bandwidth. When many carriers are present, as with a transponder carrying narrowband SCPC signals, there will be a very large number of IM products, making quasi-linear operation essential.

Calculation of C/N with Intermodulation Intermodulation between carriers in a nonlinear transponder adds unwanted products into the transponder bandwidth that are treated as though the interference were Gaussian noise. For wideband carriers, the behavior of the IM products will be noiselike; with narrowband carriers, the assumption may not be accurate, but is applied because of the difficulty of determining the exact nature of the IM products. The output backoff of a transponder reduces the output power level of all carriers, which therefore reduces the (CN) ratio in the transponder. The transponder CN ratio appears as (CN)up in the calculation of the overall (CN)0 ratio in the earth station receiver. IM noise in the transponder is defined by another CN ratio, (CN)IM, which enters the overall (CN)0 ratio through the reciprocal formula (using linear CN power ratios) 1C N2 0  1  31  1CN2 up  1  1CN2 dn  1 1CN2 IM 4

(6.10)

Techniques for the calculation of (CN)IM are beyond the scope of this text. Full knowledge of the transponder nonlinearity and the signals carried by the transponder is required to permit (CN)IM to be calculated5,6. There is an optimum output backoff for any nonlinear transponder operating in FDMA mode. Figure 6.6 illustrates the effect of the HPA operating point on each CN ratio in Eq. (6.10) when the operating point is set by the power transmitted by the uplink earth station. The uplink (CN)up ratio increases linearly as the transponder input power is increased, leading to a corresponding nonlinear increase in transponder output power. As the nonlinear region of the transponder is reached, the downlink (CN)dn ratio increases less rapidly than (CN)up because the nonlinear transponder is going into saturation. Intermodulation products start to appear as the nonlinear region is approached, increasing rapidly as saturation is reached. With a third-order model for nonlinearity, the intermodulation products increase in power at three times the rate at which the input power to

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C/N 30 (C/N)IM 25 C/N ratio in dB

(C/N)up

20

(C/N)dn 15 (C/N)o −110

−105

−100

Transponder input power dBW

Pin

231

FIGURE 6.6 CN ratios for a link using the nonlinear transponder illustrated in Figure 6.5. Overall (CN)0 at the receiving earth station is the combination of the three CN ratios shown in this figure. As the power level at the input of the transponder is increased, (CN)up in the transponder increases linearly, but (CN)dn in the earth station receiver increases less rapidly as the transponder saturates. Third-order intermodulation products are generated in the transponder as it saturates, causing overall (CN)0 to peak at an input level of 104 dBW. This is the optimum operating point for this transponder. The dashed lines show CN ratios for a transponder that does not saturate.

the transponder is increases, causing (CN)IM to decrease rapidly as saturation is approached. When all three CN ratios are combined through Eq. (6.10), the overall (CN)0 ratio in the receiving earth station receiver has a maximum value at an input power level of 104 dBW in the example in Figure 6.6. This is the optimum operating point for the transponder. The optimum operating point may be many decibels below the saturated output level of the transponder under some conditions5. VSAT networks and mobile satellite telephones often use single channel per carrier (SCPC) FDMA to share transponder bandwidth. Because the carriers are narrowband, in the 10 to 128 kHz range typically, a 36 or 54 MHz transponder may carry many hundreds of carriers simultaneously. The balance between the power levels of the carriers may not be maintained, especially in a system with mobile transmitters that can be subject to fading. The transponder must operate in a linear mode for such systems to be feasible, either by the use of a linear transponder or by applying large output backoff to force operation of the transponder into its linear region. EXAMPLE 6.2.1 Power Sharing in FDMA Three identical large earth stations with 500 W saturated output power transmitters access a 36 MHz bandwidth transponder using FDMA. The transponder saturated output power is 40 W and it is operated with 3 dB output backoff when FDMA is used. The gain of the transponder is 105 dB in its linear range. The bandwidths of the earth station signals are Station A: Station B: Station C:

15 MHz 10 MHz 5 MHz

Find the power level at the output of the transponder, and at the input to the transponder, in dBW, for each earth station signal, assuming that the transponder is operating in its linear region with 3 dB output backoff. Each earth station must transmit 250 W to achieve an output power of 20 W from the transponder. Find the transmit power for each earth station when the transponder is operated with FDMA by the three earth stations.

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The output power of the transponder must be shared between the three signals in proportion to their bandwidths. The output backoff of 3 dB means that the output power from the transponder is Pt where Pt  10 log10 40  3  16  3 dBW  13 dBW  20 W The total bandwidth used is 15  10  5  30 MHz. The output power must be shared in proportion to bandwidth used, so the transponder output power allocated to each earth station’s signal is Station A: Station B: Station C:

B  15 MHz Pt  1530  20 W  10.0 W  10 dBW B  10 MHz Pt  1030  20 W  6.67 W  8.2 dBW B  5 MHz Pt  1530  20 W  3.33 W  5.2 dBW

The transponder gain is 105 dB, in its linear range, so for linear operation the transponder input power for each earth station signal is Station A: Station B: Station C:

Pin  10.0  105  95.0 dBW Pin  8.2  105  96.8 dBW Pin  5.2  105  99.8 dBW

The EIRP at each earth station must be set to give the correct input power at the input to the transponder. A single earth station must transmit 250 W  24 dBW to achieve a transponder output of 20 W. For the transponder output power levels of each signal calculated above, the earth station transmitter powers are Station A: Station B: Station C:

Pt  24.0  3.0  21.0 dBW  126 W Pt  24.0  4.8  19.2 dBW  83 W Pt  24.0  7.8  16.2 dBW  42 W



EXAMPLE 6.2.2 Channel Capacity with Demand Access FDMA A large number of satellite telephones can access a single transponder on an LEO satellite using FDMA-DA. Data transmitted from the satellite on initial access by the telephone is used to set the transmit frequency and output power of the satellite telephone. The telephones transmit BPSK signals in L band with an occupied bandwidth of 12 kHz and an output power level between 0.05 and 0.5 W, such that the power level at the input to the transponder is always 144 dBW for any uplink signal. The resulting CN ratio in clear air conditions for any one signal in the transponder is 16 dB. The transponder has a bandwidth of 1.0 MHz, a gain of 134 dB, and a maximum permitted output power of 5 W. The center frequencies of the telephone transmitters are spaced 16 kHz apart to provide a 4 kHz guard band between each signal. Determine the maximum number of satellite telephones which can simultaneously access the transponder. Is the transponder power or bandwidth limited? If the transponder is power limited, what change could be made to increase the number of signals the transponder carries? What effect would the change have on overall (CN)0 for the link? If the transponder is bandwidth limited, the maximum number of signals, Nmax, that it could carry is the available bandwidth divided by the signal bandwidth plus the guard band width Nmax  1000 kHz16 kHz  62 The value of Nmax must be rounded down to the next lowest integer because we cannot send fractional signals. The power level of each signal at the input to the transponder is 144 dBW. The gain of the transponder is 134 dB, so the output power for each signal is Pt  144  134  10 dBW  0.1 W

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If we have 62 signals, each at a power level of 0.1 W, the total power at the output of the transponder is 6.2 W. This exceeds the maximum permitted output power of the transponder, which was set at 5.0 W. Hence the maximum number of satellite telephones that can simultaneously access the transponder is 50, and the transponder is power limited. We can increase the number of signals in the transponder to 62, which is the maximum possible number of telephones that can share the transponder at the same time because of the bandwidth limit, by reducing the input power level by 10 log10 (6250)  0.9 dB. Then the output power from the transponder, per signal, is Pt  144.8  134  10.9 dBW  0.081 W We can now transmit 62 signals from the transponder with a total output power level of 62  0.081  5.0 W, which meets the power limitation for the transponder. The CN ratio in the transponder will be reduced by 0.9 dB because the input signal is 0.9 dB weaker. Hence (CN)up  16  0.9  15.1 dB. The transponder now transmits 0.9 dB less power per signal, which will reduce the (CN)dn ratio at the receiving earth station by 0.9 dB. Hence the overall (CN)0 ratio for the link will be reduced by 0.9 dB when the number of satellite telephones sharing the transponder is increased from 50 to 62. 

6.3

TIME DIVISION MULTIPLE ACCESS (TDMA) In time division multiple access a number of earth stations take turns transmitting bursts of RF signals through a transponder. Since all practical TDMA systems are digital, TDMA has all the advantages over FDMA that digital signals have over analog. TDMA systems, because the signals are digital and can be divided by time, are easily reconfigured for changing traffic demands, are resistant to noise and interference, and can readily handle mixed voice, video, and data traffic. One major advantage of TDMA when using the entire bandwidth of a transponder is that only one signal is present in the transponder at one time, thus overcoming many of the problems caused by nonlinear transponders operating with FDMA. However, using all of the transponder bandwidth requires every earth station to transmit at a high bit rate, which requires high transmitter power, and TDMA is not well suited to narrowband signals from small earth stations. Nonlinearity in the transponder can cause an increase in intersymbol interference with digital carriers; equalizers can be used at the receiving earth stations to mitigate the effect. Many of the concepts developed in Chapter 5 for time division multiplexing (TDM) also apply to TDMA. The difference between TDM and TDMA is that TDM is a baseband technique used at one location (for example, a transmitting earth station) to multiplex several digital bit streams into a single higher speed digital signal. Groups of bits are taken from each of the bit streams and formed into baseband packets or frames that also contain synchronization and identification bits. At a receiving earth station, the high-speed bit stream must first be recovered using the techniques discussed in Chapter 5, which requires demodulation of the RF carrier, generation of a bit clock, sampling of the received waveform, and recovery of the bits. The synchronization bits or words in the packets or frames must then be found so that the high-speed bit stream can be split into its original lower speed signals. The clock frequency for the bit stream is fixed, and the frame length is usually constant. Packet lengths can vary, however, which is the main difference between frames and packets. The entire process requires considerable storage of bits so that the original signals can be rebuilt, leading to delays in transmission. In a GEO satellite system, the largest delay is always the transmission time to the satellite and back to earth, typically 240 ms. The transmission delay is unavoidable, but any additional delays should be minimized.

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Satellite

TDMA stream from satellite One frame Bursts from each earth station

Incoming bit streams

Earth terminals

FIGURE 6.7 Illustration of TDMA with three earth stations. Transmitting earth stations must time their burst transmissions so that they arrive at the satellite in the correct sequence. The signal transmitted by the satellite is a continuous sequence of bursts separated by short guard times.

TDMA is an RF multiple access technique that allows a single transponder to be shared in time between RF carriers from different earth stations. In a TDMA system, the RF carrier from each earth station sharing a transponder is sent as a burst at a specific time. At the satellite, bursts from different earth stations arrive sequentially, so the transponder carries a near continuous signal made up of a sequence of short bursts coming from different earth stations. The principle of TDMA is illustrated in Figure 6.7. The burst transmission is assembled at a transmitting earth station so that it will correctly fit into the TDMA frame at the satellite. The frame has a length from 125 s to many milliseconds, and the burst from the earth station must be transmitted at the correct time to arrive at the satellite in the correct position within the TDMA frame. This requires synchronization of all the earth stations in a TDMA network, adding considerable complexity to the equipment at the transmitting station. Each station must know exactly when to transmit, typically within a microsecond, so that the RF bursts arriving at the satellite from different earth stations do not overlap. (A time overlap of two RF signals is called a collision and results in data in both signals being lost. Collisions must not be allowed to occur in a TDMA system.) A receiving earth station must synchronize its receiver to each of the sequential bursts in the TDMA signal and recover the transmission from each uplink earth station. The uplink transmissions are then broken down to extract the data bits, which are stored and reassembled into their original bit streams for onward transmission. The individual transmissions from different uplink earth stations are usually sent using BPSK or QPSK, and will inevitably have small differences in carrier and clock frequencies, and different carrier phases. The receiving earth station must synchronize its PSK demodulator to each burst of signal within a few microseconds, and then synchronize its bit clock in the next few microseconds so that a bit stream can be recovered. In high-speed TDMA systems, operating at 120 Mbps, for example, these are demanding requirements.

Bits, Symbols, and Channels A potential source of confusion in the discussion of TDMA systems is that QPSK (or possibly QAM) modulation is typically used by transmitting earth stations, and data rates can

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then be described either by bit rate or symbol rate. Both bit rates and symbol rates need to be used in the discussion of digital radio transmission and TDMA systems, so the reader must be clear on the distinction between a bit and a symbol. A bit is the fundamental unit in digital transmission. Data are generated by terminals (e.g., a personal computer) as bits, or by conversion of an analog speech or video signal to digital form as a serial bit stream. The bit stream is described by its bit rate, in bits per second, bps, thousands of bits per second, kbps, or millions of bits per second, Mbps. (Note that the k and M prefixes are in units of 103 and 106, not the binary digital version of 1024 and 1,048,576.) The bit stream must be modulated onto an RF carrier for transmission to the satellite. Phase shift keying is invariably used as the modulation technique. In binary phase shift keying, BPSK, the logical data states of the bits, 1 and 0, are converted into two opposite phase states of the RF carrier, say 0° and 180°. In quadrature phase shift keying, QPSK, two bits at a time are converted into one of four phase states of the RF carrier (see Chapter 5 for details of QPSK). The state of the RF carrier is called a symbol, and the symbol rate is in units of bauds, or symbols per second. For BPSK, bit rate and baud rate are the same. For QPSK, the baud rate (symbol rate) is one half the bit rate. The importance of symbol rate in any digital radio system is that it is the symbol rate, not the bit rate, that determines the bandwidth of the RF signal, and consequently the bandwidth of the filters in the receiver. Occasionally, QAM modulation may be used on a satellite link. In QAM, carrier symbols are generated from the four phase states of QPSK, but can also have different amplitudes. This allows one symbol to convey more than 2 bits, which reduces the RF bandwidth required for a given bit rate. However, an increased (CN)0 ratio is required in the receiver to recover bits from QAM signals, so QAM can only be used in satellite links with higher than usual (CN)0 ratios.

TDMA Frame Structure A TDMA frame contains the signals transmitted by all of the earth station in a TDMA network. It has a fixed length, and is built up from the burst transmissions of each earth station, with guard times between each burst. The frame exists only in the satellite transponder and on the downlinks from the satellite to the receiving earth stations. Figure 6.8 shows a simplified diagram of a TDMA frame for four transmitting earth stations. Each station transmits a preamble that contains synchronization and other data essential to the operation of the network before sending data. The earth station’s transmission is followed by a guard time to avoid possible overlap of the following transmission. In GEO satellite systems, frame lengths of 125 s up to 20 ms have been used, although 2 ms has been widely used by stations using Intelsat satellites. Earth stations must be able to join the network, add their bursts to the TDMA frame in the correct time sequence, and leave the network

Next frame

Frame period T, µs Stn 1

Stn 2

Stn 3

Stn 4

Stn 1

Guard time Traffic: N bits Preamble

FIGURE 6.8 TDMA frame with four transmitting earth stations.

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without disrupting its operation. They must also be able to track changes in the timing of the frame caused by motion of the satellite toward or away from the earth station. Each earth station must also be able to extract the data bits and other information from burst transmissions of other earth stations in the TDMA network. The transmitted bursts must contain synchronization and identification information that help receiving earth stations to extract the required information without error. These goals are achieved by dividing TDMA transmissions into two parts: a preamble containing all the synchronization and identification data, and a group of traffic bits. Synchronization of the TDMA network is achieved with the portion of the preamble transmitted by each earth station that contains carrier and bit clock synchronization waveforms. In some systems, a separate reference burst may be transmitted by one of the stations, designated as the master station. A reference burst is a preamble followed by no traffic bits. Traffic bits are the revenue producing portion of each frame, and the preamble and reference bursts represent overhead. The smaller the overhead, the more efficient the TDMA system, but the greater the difficulty of acquiring and maintaining network synchronization. The preamble of each station’s burst transmission requires a fixed transmission time. A longer frame contains proportionally less preamble time than a short frame, so more revenue producing data bits can be carried in a long frame. Early TDMA systems were designed around 125 s frames, to match the sample rate of digital speech in telephone systems, in exactly the same way that T1 24-channel systems operate. A digital telephone channel generates one 8-bit digital word every 125 s (8 kHz sampling rate), so a 125-s frame transmits one word from each speech channel. However, it is more efficient to lengthen the frame to 2 ms or longer so that the proportion of overhead to message transmission time is reduced. It must be remembered that a longer frame requires multiple 8-bit words when transmitting digital speech. For example, in a time period of 2 ms, a digital terrestrial channel will deliver sixteen 8-bit words, so a 2-ms TDMA frame requires sixteen 8-bit words from each terrestrial channel in each transmitted burst. Figure 6.9 shows a typical TDMA frame with 2.0 ms duration used by some earth stations operating in TDMA through Intelsat satellites. All of the blocks at the start of the

CBTR

UW

TTY

VOW VOW

SC

Digital speech channels

M satellite channels 1

2

1

3

2 3

4

5

4 5 6

6

7

7

8

M

9 10 11 12 13 14 15 16

Sixteen 8-bit samples form each satellite channel FIGURE 6.9 Structure of an Intelsat traffic data burst. A satellite channel is a block of sixteen 8-bit samples from one terrestrial speech channel. Other blocks in the traffic burst are used to synchronize the PSK demodulator, the bit clock, and the frame clock in the receiver (CBTR, UW) and to provide communication links between earth stations (TTY, SC, and VOW). CBTR, carrier and bit timing recovery; UW, unique word; TTY, teletype; SC, satellite channel; VOW, voice order wire.

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frame, labeled CBTR through VOW, are preamble. Speech channel data transmission begins with satellite channel 1 and continues as a serial bit stream through channel M. A satellite channel is made up of the number of bits delivered to the earth station by a single digital speech channel during one frame period. The frame can equally well send digital data of any form as a serial stream of bits occupying the space taken up by M satellite channels. For the specific case of digital speech channels using serial transmission at a rate rsp, the number of speech channels, n, that can be transmitted in a TDMA frame shared equally by N earth stations can be calculated from the duration of the frame, Tframe in seconds, the guard time and preamble length, tg and tpre, in seconds, and the transmitted bit rate of the TDMA system, Rb. The time, Td, available in each station burst for transmission of data bits is Td  3Tframe  N1tg  tpre 2 4 N seconds

(6.11)

In 1 s, the total number of bits, Cb, transmitted by each earth station is Cb  3Tframe  N1tg  tpre 2 4  RbTframe

(6.12)

Since each digital speech channel requires a continuous bit rate of rsp bps, the number of speech channels that can be carried by each earth station is given by n where n  3Tframe  N1tg  tpre 2 4 

Rb Tframe  rsp

(6.13)

EXAMPLE 6.3.1 TDMA in a Fixed Station Network A TDMA network of five earth stations shares a single transponder equally. The frame duration is 2.0 ms, the preamble time per station is 20 s, and guard bands of 5 s are used between bursts. Transmission bursts are QPSK at 30 Mbaud. Calculate the number of 64 kbps voice channels that each TDMA earth station can transmit. If the earth stations send data rather than digital speech, what is the transmission rate of each earth station in Mbps? What is the efficiency of the TDMA system expressed as Efficiency  100%  Message bits sentMaximum possible number of bits that could be sent? Using Eq. (6.11) we can find the data burst duration for each earth station, Td, in microseconds Td  32000  5  15  202 4 5  375 ms A burst transmission rate of 30 Mbaud is 30 million symbols per second, and QPSK symbols carry 2 bits. Hence the transmitted bit rate in each burst is Rb  2  30 Mbps  60 Mbps. The capacity of each earth station in bits per second is Cb where Cb  375  60  1062000  11.25 Mbps Since these bits are sent as 500 bursts each second, the number of message bits per burst is 11.25  106500  225,000 kbits. The number of 64 kbps digital speech channels that can be carried by one earth station is n  11,250,00064,000  1751.7812 We have to discard the fractional channel since we can send only whole channels. The 0.781 fraction of a speech channel represents 50,000 bits per second that cannot be transmitted by each earth station, or 100 bits per burst. At a bit rate of 60 Mbps, 100 bits is equivalent to 1.67 s, so the guard time per burst would increase from 5 to 6.67 s when 175 speech channels are sent. 

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It is always a good idea to check the answer to any problem against a reference value. If the earth stations transmitted without any guard times or preambles, the maximum number of 64-kbps speech channels that could be sent by each of the five earth stations using a transmission burst rate of 60 Mbps would be n  Rb(N  rs)  60  106(5  64  103)  187.5. The calculated capacity per station of 175 channels is a little lower than the maximum number of 187 channels; the difference—12 channels in this case—is the number of speech channels lost by the need to include guard times and preambles in the TDMA frame structure. If we send data rather than digital speech, each earth station can transmit 11.25 Mbps. Overhead accounts for the equivalent of 0.75 Mbps, leading to an efficiency of Efficiency  100%  11.2512.0  93.75% Example 6.3.1 shows that there are diminishing returns to be obtained by increasing the duration of the frame beyond 2.0 ms in this system. There is a 6.25% loss of potential data transmission time when the 2.0-ms frame length of Example 6.3.1 is used with the other parameters of this TDMA system. Doubling the frame length to 4.0 ms would reduce the loss to about 3%, but would add additional delay, and increase the complexity of the earth station equipment. In this case, a 2.0 ms frame period appears to be a good compromise between delay, complexity, and efficiency.

Reference Burst and Preamble Figure 6.9 shows the baseband content of a typical TDMA burst. The segments marked CBTR and UW contain the carrier recovery waveform, bit clock synchronization, unique word, and station identifier. The remaining blocks are part of the preamble and will be described below. CBTR is the portion of the burst from a given earth station that enables a receiving earth station to recover the remainder of the burst. CBTR stands for carrier and bit timing recovery. Carrier recovery is required at the receiver of any radio link in which coherent phase shift keying is the modulation technique. A local carrier must be generated in the IF portion of the receiver from the received BPSK or QPSK signal. This is typically achieved with some form of phase locked loop (PLL), which must lock up quickly in a TDMA system. The local carrier drives the multipliers in the demodulator to achieve coherent demodulation of the PSK signal. (See Chapter 5 for details of the modulation and demodulation process in PSK systems.) Once carrier phase lock is achieved, the demodulator will produce a baseband waveform corresponding to the bits that were modulated onto the carrier at the transmitting earth station. In a TDMA system, the bursts of RF signal received sequentially from different earth stations do not have the same carrier frequency, phase, or bit rate. The differences will be small, but sufficient to require the receiver to relock to each new carrier, and to resynchronize the bit clock. The CBTR contains a sequence of predetermined signals that ensure rapid lock to the carrier and fast synchronization of the bit clock. The carrier recovery portion of the CBTR sequence may consist of unmodulated carrier followed by a number of symbols that follow a specific pattern, or the entire CBTR burst may be modulated. The first part of the CBTR burst is used to obtain lockup of the PLL, and the remaining portion is used for bit clock synchronization. In Figure 6.9, the CBTR burst has a duration of 176 symbols with a transmission rate of 30 Msps, giving a burst duration of 5.86 s. Within this very short time period the carrier recovery circuit must achieve precise lock on the received signal and the bit clock must be brought into synchronization. Carrier phase lock and bit synchronization must be achieved within the CBTR burst time

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even when the carrier-to-noise ratio is very low, perhaps as low as 5 dB in VSAT links employing half rate forward error correction to improve the BER of the received data. Typical TDMA systems using QPSK send the first 48 or 50 symbols of the CBTR burst as all ones or all zeroes on both the I and Q phases of the carrier, which correspond to sending an unmodulated carrier, followed by a sequence of ones and zeros in both channels7,8. The CBTR burst is followed by a unique word (UW) of typically 20 to 48 bits which serves several purposes. It acts as a transmit station identifier, a start of frame (SOF) or burst marker, and as a carrier phase ambiguity detector.

Unique Word The received bit sequence at the demodulator output is continuously run through a correlator which looks for the appearance of the unique word in the bit stream. Figure 6.10 illustrates a simplified unique word correlator. The correlator is looking for a match between the incoming bit pattern and one of four stored sequences, corresponding to the correct UW sequence and three variants in which the I and Q bits of the sequence are inverted due to phase ambiguity in the carrier recovery circuit. QPSK carrier recovery can result in ambiguity if the carrier recovery circuit locks up in the incorrect phase, which is possible in many QPSK demodulators. When this happens, one or both of the I and Q bit streams is inverted. A known bit sequence is required in the received signal for ambiguity resolution. The pattern of ones and zeroes in the CBTR sequence and the unique word allow the receiver to check for phase ambiguity and to invert the appropriate bit stream (I, Q, or both) if ambiguity is found. In the correlator diagram shown in Figure 6.10, when the correct UW sequence is present in the correlator in the correct position (i.e., at the correct time within the incoming signal burst) the correlator output will maximize and trigger the threshold detector. When a new RF burst is received, the carrier recovery circuit locks up the local carrier PLL, and the bit clock then synchronizes to the bit rate of the received signal.

Input from receiver

Shift register

Summer

Σ Bit clock

0

1

0

1

1

1

Output to threshold circuit

Modulo 2 multipliers (exclusive OR gates)

Unique word memory FIGURE 6.10 Unique word correlator. The example shown here has a 6 bit unique word for illustration—practical satellite systems use unique words of 24–48 bits. The bits stream from the receiver output is clocked into the shift register serially. When the contents of the shift register match the stored unique word the output of the summer is a maximum and exceeds the threshold, marking the end of the unique word. This provides a time marker for the remainder of the earth station’s transmission.

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Bits then begin to flow into the correlator, which detects one of the four possible forms of the UW and sets logical inverters that invert the appropriate bits, if necessary. The resulting bit stream after the end of the UW is then output correctly and can be used by the receiver. The time at which the threshold detector is triggered marks a known point in the TDMA burst—the end of the UW sequence. This time is critical, because all subsequent bits from the demodulator will be demultiplexed based on a count that begins when the UW is detected. If the UW is detected at the wrong time, the recovered data in the entire burst will be scrambled and the burst is lost. The UW and the correlator circuits must therefore be designed to ensure that the UW is detected correctly in every burst with a very low probability of a timing error. An incorrectly detected UW is known as a miss, and the probability that a miss occurs can be calculated from the bit error probability (BER) of the recovered bit stream and the length of the UW. The length of the UW and the correlator design are important factors in the design of a TDMA system. The detection of the UW occurs when there is a match between the UW output by the demodulator and the stored sequence. However, if the CN ratio of the RF signal is low, there will be bit errors present in the UW, so precise correlation cannot be guaranteed. A long UW allows a predetermined number of bits to be in error while still ensuring that correct timing of the bit sequence is achieved. The probability that the specific bit sequence of a UW appears within the bit stream of traffic data must also be low, so that the probability of a false alarm is small. A false alarm occurs when a UW is inadvertently detected in the traffic data, causing the timing of the burst to be reset. There are several ways that false alarms can be prevented, but the use of a long UW reduces the likelihood of a false alarm occurring and is therefore desirable. Once the time position of a UW within the TDMA frame is determined, a window can be placed over the UW so that the correlator is operated only during a period slightly longer than the UW duration. This will greatly reduce the chances of a false alarm. As an example of the probability of a missed detection, a 24 bit UW with a bit error rate of 103 can be recovered with a miss probability of 1010 when three of the bits in the UW are allowed to be in error, corresponding to a detection threshold of 21249. Although a miss probability of 1010 may appear sufficiently small at first sight, a large number of bits is lost whenever a UW is detected incorrectly. The TDMA system with five earth stations in Example 6.3.1 used a 2 ms frame with a traffic data time of 375 s per station. At a bit rate of 60 Mbps there are 22,500 traffic bits in each burst. With a probability of missing the UW of 1010, the average BER of the traffic data resulting from one missed UW is 2.25  108. This BER is too high to be acceptable in many applications, so a longer UW would have to be used to lower the probability of a missed detection. For further details of unique word detection and TDMA burst design, the reader is referred to one of the references9–11. There must be as many UW correlators as there are TDMA uplink stations in the network when the UW is also used as a station identifier. If there is a large number of stations it is simpler to use a single UW and a separate station identifier word. The bit stream for each received burst is tagged with a station identifier so that the demultiplexers in the earth station can route the data accordingly. The remaining segments of the preamble provide the receiving earth station with bit streams that are used in the management of the TDMA system. Referring to Figure 6.9, there are groups of bits identified as TTY, SC, VOW. Sixteen bits (eight QPSK symbols) in each burst are allocated to a teletype link (TTY) between the earth stations, and a

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further 16 bits to a service channel (SC). There are two 64-bit voice order wire (VOW) segments that are used in digital voice links between the earth stations. The TDMA burst shown in Figure 6.9 is designed for use in a network of large earth stations in which there are personnel at the earth stations or network control centers. The teletype and voice channels within the preamble of each burst provide a closed communication network between the control stations that is used to manage the TDMA system. In a VSAT network using TDMA, for example, there would be minimal communication requirement between terminals for management purposes, so a VSAT TDMA burst would omit most of these segments. The final part of the TDMA burst carries the traffic data. In Example 6.3.1 which used a 2.0 ms frame, speech data could be carried in satellite channels as sixteen 8-bit words from each terrestrial channel, giving a satellite channel 128 bits in each burst. The bits associated with each satellite channel are demultiplexed by counting the 128 bits of each satellite channel, beginning at the end of the preamble. Once again, the bit timing established by the unique word becomes all important in extracting the satellite channels correctly. There are many different formats for preambles in TDMA systems, depending on the design of the particular system. Figure 6.9 illustrates a TDMA preamble structure designed for a large fixed network with high-speed bit streams. A network of mobile earth terminals using TDMA would have different requirements and would require a different preamble structure. However, the earlier segments that control carrier and bit timing recovery, phase ambiguity removal, and station identification must always be present. A large earth station carrying high-speed data must link into a terrestrial data network to deliver received bits to customers, and to receive incoming data for transmission over the satellite link. The satellite link connects two earth stations which may be thousands of kilometers apart, each interconnecting to its own high-speed terrestrial network. The individual terrestrial networks are not synchronized, and will therefore inevitably run at slightly different rates. This makes the interconnection process difficult, because a data stream delivering bits at 1.00001 Mbps cannot be connected directly to another data stream running at 0.99999 Mbps. Twenty bits would have to be discarded every second if a direct connection were made, which upsets the customer who is paying to transfer the data. A mechanism must be developed that allows for a difference in bit rates at the two ends of the link. The usual solution is to run the bit clock of the satellite link slightly faster than the fastest of the terrestrial link clocks, and to allow additional bit slots for stuffing bits. Stuffing bits are inserted when there are no data bits available from the source because of the difference in bit rates. At the receiving end of the link, the stuffing bits are removed and the received data stream is retimed to match the outgoing terrestrial data channels.

Guard Times Guard times must be provided between bursts from each earth station so that collisions are avoided. Earth stations must transmit their bursts at precisely the correct instant so that the burst arrives at the satellite in the correct position within the TDMA frame. This requires burst transmission timing to microsecond accuracy and tracking of the position of the burst within the TDMA frame by the transmitting earth station. Long guard times make it easier for the earth stations to avoid collisions, but waste time that could be used to send revenue-earning traffic data. Typical guard times in high-speed satellite networks

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appear to be in the range 1 to 5 s. The transmission time between an earth station and a GEO satellite is about 120 ms. If a 2-ms frame time is used, there are typically 60 bursts between the earth station and the satellite at any time. The bursts must arrive at the correct time to mesh between the bursts that arrive from other earth stations. If the satellite range from the earth station increases by 300 m, due to eccentricity or inclination of a GEO satellite’s orbit, or E–W drift in the orbital plane, the transmission delay increases by 1 s. Thus earth stations must monitor the guard times before and after their bursts to ensure that transmission timing is correct. In LEO satellite networks that use TDMA, range to the satellite is changing continuously and much larger guard times are allowed.

Synchronization in TDMA Networks Earth stations operating in a TDMA network must transmit their RF bursts at precisely controlled times such that bursts from each of the earth stations arrive at the satellite in the correct sequence12. This poses two problems: how to start up a new earth station that is joining the TDMA network, and how to maintain the correct burst timing. If the satellite is in low earth orbit, or if it is a GEO satellite with a rapidly changing range, each earth station will perceive a different carrier frequency and frame rate, and even a different frame length. It is usual for the bit rate of transmitted bursts to be an integer multiple of the frame rate, which means that different earth stations must transmit at slightly different bit rates. Maintaining synchronization with the TDMA frame is easier than initial synchronization when an earth station joins a TDMA network. One station is typically designated as the master station, and may generate a reference burst to mark the start of the frame. Each of the stations within the network has a time slot within the frame, and must maintain its transmissions within that time slot. There are guard times at each end of each station’s burst, which define the accuracy that the burst timing must achieve. If the guard times are 2 s, each earth station in the network must keep its bursts timed to within 1 s. This is usually done by monitoring the TDMA frame at the transmitting station and adjusting the burst timing to keep the transmitted burst in the correct time slot in the frame. The start of the reference burst, or the start of the master station’s preamble, marks the start of transmit frame, SOTF, which is the master timing mark for all transmissions. All earth stations in the TDMA network synchronize their clock timing with the SOTF marker. When an earth station monitors its own transmissions to maintain the correct burst timing, this is called satellite loop-back synchronization. The TDMA frame is established at the satellite, so an earth station receiving the frame must subtract the transmission delay from the satellite to the earth station to obtain the SOTF timing at the satellite. It must then transmit its bursts ahead of the SOTF by the same delay time so that the bursts arrive at the correct instant at the satellite. Knowledge of the range of the satellite from the earth station is crucial in calculating delay times. The range can be calculated from the orbital elements of the satellite, which can be determined by a control station that repeatedly measures range to the satellite. There are several ways that earth stations can enter a TDMA network. In fixed networks, the precise time at which an earth station should transmit can be calculated. Provided the calculation is accurate, the earth station can transmit a reference burst (no traffic) timed to fall in the center of its time slot. When the frames containing the reference burst arrive back at the earth station, the actual position of the burst can be checked

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and corrections to the timing made if necessary. The station can then transmit traffic bursts with the correct timing. In TDMA networks which lack sophisticated timing control, an earth station wishing to join the network can transmit a CDMA sequence at a low level, at an arbitrary time. The CDMA sequence will inevitably collide with another earth station’s traffic burst, causing minor interference. The transmitting earth station can use a correlator to compress the CDMA sequence into a single timing pulse using exactly the same process as is used to find a unique word. However, in this case the CDMA sequence is overwritten by interference from the traffic burst with which it collided. Given a suitably long sequence, coding gain can overcome the interference and a pulse will appear at the correlator output marking the end of the CDMA sequence. Alternatively, the transmitting station can use a shorter sequence and step the timing of the CDMA burst until it falls in the empty slot allocated to that station. The position of the pulse within the TDMA frame gives the transmitting earth station the timing information needed to transmit its bursts at the correct time. If the signal transmitted by the satellite cannot be monitored by the transmitting earth station, cooperative synchronization must be used instead. This situation arises when a satellite has multiple beams, or when satellite switched TDMA is used. A TDMA burst received in one beam can be retransmitted by the satellite in another beam that does not cover the transmitting station. A control station is required that can monitor the timing of each of the earth station’s bursts as they arrive at the satellite and send out instructions to the earth stations when changes in timing are needed. In the Intelsat TDMA system, the control station determines a delay time, DN, for each earth station that gives the time between the start of a receive frame and the start of a transmit frame at that earth station. The correct transmit time is then determined by the position of earth station’s burst within the transmit frame. If transmitting earth stations fall out of sync, the control station must send a do not transmit (DNTX) code to the station to tell it to stop transmitting because serious loss of data will occur to other users of the network when a station sends its bursts at the incorrect time. In satellite switched and multiple beam satellite systems, the cooperating control station must provide information to a new earth station that wishes to join the network. The same techniques described above can be used, but an earth station within the receiving beam must determine the timing of test transmissions and send that information to the transmitting station. The availability of a global GPS time standard with better than 1-s accuracy has made some of these tasks easier. (See Chapter 12 for details of the GPS system.)

Transmitter Power in TDMA Networks TDMA works well in fixed networks carrying high-speed data streams. Transponders can be more heavily loaded because less backoff is needed with TDMA; only one RF signal is present in the transponder at any time and there is no third-order intermodulation, so backoff is needed only to keep PM–AM conversion at an acceptable level. Burst lengths can be made variable to accommodate stations that have different bit rates. High uplink transmitter power is required because every station must transmit bursts at a high bit rate and high bit rate signals occupy a wide bandwidth. Maintaining an adequate CN ratio in the transponder forces the uplink earth station to use a high-power transmitter. For small earth stations such as VSATs and satellite phones, this is a major disadvantage compared to SCPC-FDMA.

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EXAMPLE 6.3.2 TDMA in a VSAT Network As an example, consider a typical VSAT earth station in the United States which is part of a TDMA network using a 54 MHz bandwidth transponder on a domestic Ku-band GEO satellite. The VSAT earth station has a 1 m antenna that transmits a single 64 kbps signal at 14 GHz. Let’s assume that the TDMA network uses QPSK modulation and that all transmitters have a symbol rate of 30 Mbaud. We will set (CN)up at 20 dB, and then calculate the required uplink transmit power. The following system parameters will be used: Earth station antenna gain  41.5 dB, satellite antenna gain (on axis)  32.0 dB, edge of beam loss  3 dB, path loss at 14 GHz  207.1dB, receiver noise bandwidth  30 MHz, transponder noise temperature  500 K, atmospheric and other losses  1.0 dB. The uplink power and noise budgets are Earth station transmit power  Pt dBW Earth station antenna gain at 11 GHz  41.5 dB Satellite antenna  32.0 dB Edge of beam loss  3.0 dB Other losses  1.0 dB Path loss at 11 GHz  207.1 dB Power at transponder input  Pt  137.6 dBW Boltzmann’s constant  228.6 dBW/K/Hz Transponder noise bandwidth  74.8 dBHz Transponder noise temperature  27.0 dBK Transponder input noise power  126.8 dBW We require (CN)up  PrkTsBn  20 dB; hence Pt  137.6  126.8  20 dBW and Pt  30.8 dBW or Pt  1200 W. Now consider the same earth station transmitting the same 64-kbps signal in a SCPC-FDMA VSAT network using QPSK with a symbol rate of 32 kbaud and a receiver noise bandwidth of 32 kHz. The uplink power budget is unchanged, but the noise power in the transponder, measured in a bandwidth of 32 kHz is 156.5 dBW. To achieve (CN)up  20 dB in the transponder now requires an uplink transmitter power of Pt  20  137.6 156.5  1.1 dBW  1.3 W. 

The above example illustrates a key problem with TDMA for any small earth station: uplink transmit power. No one is going to equip a 1-m VSAT station with a 1200-W transmitter. Apart from the excessive cost, FCC regulation in the United States do not allow small VSAT stations to transmit more than 2 W to limit interference to adjacent satellites. If we change the multiple access technique for just two earth stations, so that each transmits a burst of QPSK signal at 64 kbaud for half the time, the uplink transmitter power requirement is doubled to 4.1 dBW or 2.6 W. This makes wideband TDMA an unlikely choice in VSAT networks, and limits the number of stations that can share a TDMA frame in a low earth orbit satellite telephone system. The Iridium LEO system was designed to use a hybrid TDMA–FDMA multiple access scheme at L band to combine a small number of digital telephone transmissions into a 50-kbps QPSK signal. Similar techniques are used in some VSAT networks.

EXAMPLE 6.3.3 TDMA in a Fixed Earth Station Network In Example 6.2.1, three identical large earth stations shared a single 36-MHz bandwidth transponder using FDMA. The three earth stations transmitted signals with powers and bandwidths given

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by B  15 MHz Pt  125 W  21.0 dBW B  10 MHz Pt  83 W  19.2 dBW B  5 MHz Pt  42 W  16.2 dBW

Station A: Station B: Station C:

The transponder total power output was 16 dBW with 3-dB output backoff and 105-dB transponder gain. The three earth station accesses to the transponder are changed to TDMA, with a frame length of 1.0 ms, a preamble time of 10 s, and a guard time of 2 s. There is no reference burst in the TDMA frame. The signals are transmitted using QPSK, and within the earth stations the bit rates of the signals are Station A: Station B: Station C:

Rb  15.0 Mbps Rb  10.0 Mbps Rb  5.0 Mbps

Calculate the burst duration and symbol rate for each earth station, and the earth station transmitter output power required if the transponder output backoff is set at 1.0 dB and the gain of the transponder with this output backoff is 104 dB. Compare the uplink (CN) ratios in the transponder for FDMA and TDMA operation given that station A’s transmission has a (CN)up ratio of 34 dB when the transponder is operated in FDMA. The transponder must carry a total bit rate of 15  10  5  30 Mbps within the 1.0-ms frames. Thus each frame carries 30 Mbps  0.001 s  30 kb. The three preamble and guard times take up 3  (10  2)  36 s in each frame, leaving 1000  36  964 s for transmission of data. Hence the burst bit rate is Rb burst  30 kb964 ms  31.12 Mbps. Since we are using QPSK for the transmissions, the burst symbol rate on the link is Rs burst  31.12 Mbps2  15.56 Msps Each of the stations must transmit at the same burst rate of 15.56 Msps. The burst lengths can be calculated from the time available in each frame for data transmission and the number of bits each station must send in a 1 ms TDMA frame. The time available for data transmission is 964 s, which must be shared in proportion to the number of bits each station sends in a frame. The number of bits in a frame is 30,000, so the sharing of bits and times within a frame is given below Station A: Station B: Station C:

Nb  15,000 bits TA  482.0 ms Rb  10,000 bits TB  321.3 ms Rb  5,000 bits TC  160.7 ms

We can easily check to see if these results are correct. Each earth station must have the stated average bit rate, so if we multiply the burst duration for each earth station by the burst bit rate for the transponder, 31.12 Mbps, we must have the correct number of bits/frame for each station. Station A: Station B: Station C:

TA  482.0 ms Nb  482.0 ms  31.12 Mbps  15,000 TB  321.3 ms Nb  321.3 ms  31.12 Mbps  10,000 TC  160.7 ms Nb  160.7 ms  31.12 Mbps  5,000

Each station must transmit at the same symbol rate of 15.56 Msps, regardless of the number of bits sent per frame. In the previous FDMA example, a transponder output power of 20 W  13 dBW was achieved with a total earth station power of 250 W  24 dBW and a transponder gain of 105 dB. With TDMA, we are using a 1 dB transponder output backoff, and a transponder gain of 104 dB, so the transponder output power is now 16  1  15 dBW, an increase of 2 dB, and we have lost 1 dB of gain in the transponder. This requires an earth station output power, from each earth station, of Pt es  24  2  1  27 W  500 W

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TDMA requires a substantial increase in earth station transmitter power, relative to FDMA, for a low capacity earth station which joins high-capacity stations in a TDMA system. In this example, a station that was transmitting 42 W when the transponder was operated in FDMA must now transmit 500 W when the transponder is operated in FDMA. The uplink (CN)up for station A’s 15 MHz signal was 34.0 dB when the transponder was operated in FDMA. With QPSK and a burst rate of 15.56 Msps, the noise bandwidth of the earth station receiver, assuming ideal RRC filters, will be 15.56 MHz. The output of station A has been increased from 21 to 24 dBW, so the input power level at the transponder will also have increased by 3 dB. Hence the uplink (CN)up ratio in the transponder for station A signals is 1CN2 up  34.0  10 log10 115.0 15.562  3  36.8 dB

Since all earth stations transmit at the same power level and with the same burst rate, and all the signals have the same noise bandwidth, the (CN)up ratio for each of the signals in the transponder is identical, at 36.8 dB. This is 2.8 dB higher than for the FDMA operation, but at the expense of a large increase in the total uplink power from the three earth station transmitters. 

Satellite Switched TDMA One advantage that TDMA has when used with a baseband processing transponder is satellite switched TDMA. Instead of using a single antenna beam to maintain continuous communication with its entire coverage zone, the satellite has a number of narrow antenna beams that can be used sequentially to cover the zone. A narrow antenna beam has a higher gain than a broad beam, which increases the satellite EIRP and therefore increases the capacity of the downlink. Uplink signals received by the satellite are demodulated to recover the bit streams, which are structured as a sequence of packets addressed to different receiving earth stations. The satellite creates TDMA frames of data that contain packets addressed to specific earth stations, and switches its transmit beam to the direction of the receiving earth station as the packets are transmitted. Note that control of the TDMA network timing could now be on board the satellite, rather than at a master earth station. In the above example, the VSAT earth station could transmit data to a baseband processing satellite using SCPC-FDMA to permit the use of a small antenna and low power transmitter. The satellite could then use satellite switched TDMA to send that data to multiple earth stations, creating a mesh VSAT network. It is difficult to create a mesh VSAT network using SCPC-FDMA. Baseband processors are considered in more detail in the next section.

6.4

ONBOARD PROCESSING The discussion of multiple access so far has assumed the use of a bent pipe transponder, which simply amplifies a signal received from earth and retransmits it back to earth at a different frequency. The advantage of a bent pipe transponder is flexibility. The transponder can be used for any combination of signals that will fit within its bandwidth. The disadvantage of the bent pipe transponder is that it is not well suited to uplinks from small earth stations, especially uplinks operating in Ka band. Consider a link between a small transmitting earth station and a large hub station via a bent pipe GEO satellite transponder. There will usually be a small rain fade margin on the uplink from the transmitting station because of its low EIRP. When rain affects the uplink, the CN ratio in the transponder will fall. The overall CN ratio in the hub station receiver cannot be greater than the CN ratio in the transponder, so the bit error rate at the hub station will increase quickly as rain affects the uplink. The only available solution is to use forward error correction

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coding on the link, which lowers the data throughput but is actually needed for less than 5% of the time. The problem of uplink attenuation in rain is most severe for 3020 GHz uplinks with small margins. Outages are likely to be frequent unless a large rain fade margin is included in the uplink power budget. Onboard processing or a baseband processing transponder can overcome this problem by separating the uplink and downlink signals and their CN ratios. The baseband processing transponder can also have different modulation schemes on the uplink and downlink to improve spectral efficiency, and can dynamically apply forward error control to only those links affected by rain attenuation. All LEO satellites providing mobile telephone service use onboard processing, and Kaband satellites providing Internet access to individual users also use onboard processing.

Baseband Processing Transponders A baseband processing transponder has a receiver and transmitter similar to those found in an earth station. The received signal from the uplink is converted to an intermediate frequency and demodulated to recover the baseband signal, which is then processed and reassembled. The baseband signal is modulated onto a carrier at a downlink frequency and transmitted back to earth. The signals are invariably digital, although that is not a requirement of a baseband processing transponder. The advantage of this process is that uplink and downlink signal formats need not be the same, and that different forms of error correction can be applied to the uplink and downlink. The CN ratios of the uplink and downlink are not tied together through the reciprocal formula (Eq. 6.10). If the CN on the uplink is low, because of an uplink rain fade, for example, bit errors will be present in the recovered data in the transponder. The BER will depend only on the uplink CN ratio. If the CN on the downlink is high, as is usually the case for all star networks working with a large hub earth station, no additional bit errors will occur on the downlink. Separation of the uplink and downlink signals allows different modulation methods to be used, as well as flexible error correction codes. In star networks, the CN ratio of the uplink and downlink between the satellite and the hub station is usually high because of the large antenna gain and high transmit power of the hub station. The high CN ratio can be traded for a high level modulation such as 16-QAM, which reduces the bandwidth required for the uplink and increases the spectral efficiency of the communication system. 16-QAM sends four bits per symbol, and requires only half the bandwidth of QPSK. As an example, consider a system in which half rate forward error correction and QPSK is used on the uplink from a small earth terminal. For a message data rate of Rd bits per second, the transmitted bit rate will be Rb  2 Rd bps. An RF bandwidth of Rb2  (1   up)  Rd (1   up) Hz will be required on the uplink, where  up is the rolloff parameter of the (assumed) RRC filter in the transmitter. On the downlink to the hub station, where the CN ratio is high, 16-QAM can be used without forward error correction. The RF bandwidth required for the downlink will be Rd4  (1   dn), where  dn is the roll-off parameter of the RRC filter in the transponder transmitter section. If we assume the same roll-off parameter, , in both the uplink and the downlink transmitters and receivers, the downlink requires only one-fourth of the uplink bandwidth to send the same number of bits. The fourfold reduction in downlink bandwidth represents a considerable improvement in the spectral efficiency of the satellite system. Star networks with small uplink earth stations are widely used for VSAT systems and satellite links that carry Internet traffic. One hub station linked to an Internet Service Provider (ISP) can connect to hundreds of user earth stations located at the users’ homes. This is a major growth area for Ka-band GEO satellites, a number of which are planned

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for launch in the 2002–2005 time frame. The user terminal is typically a 0.5- to 0.8-m antenna with a low power transmitter, 0.5–1 W output, that can send data to the satellite at rates up to 400 kbps. The uplink operates in the 30 GHz band with a small rain fade margin, resulting in low CN at the transponder input and requiring the use of error correction on the uplink during rain fades. It may be possible to remove all or most of the error correction when the uplink is operating in clear air, allowing higher data speeds except during rain fades. Rain fading occurs for less than 5% of the time, and affects only a small number of users at any instant. Under most conditions, only a small number of users will suffer severe rain fading at the same time, so most of the links to the satellite will not need heavy error correction. This flexibility can increase the capacity of the satellite links by a factor of two relative to a bent pipe transponder, at the cost of considerably increased complexity in both the satellite and the user terminal.

Satellite Switched TDMA with Onboard Processing Baseband processing is essential in satellites using satellite switched TDMA, because data packets must be routed to different antenna beams based on the address of the destination earth station. The data in such systems is always sent in packets which contain a header and a traffic section. The header contains the address of the originating station and the address of the destination earth station. When satellite switched TDMA is used, the transponder must extract the destination information and use it to select the correct downlink beam for that packet. The satellite is operating much like a router in a terrestrial data transmission system. Switched beam operation of an uplink from a small earth station is more difficult to achieve because it requires synchronization of the earth station transmit time with the satellite beam pointing sequence, in much the same way that a TDMA uplink operates. However, the uplink can operate in a small bandwidth which overcomes the chief disadvantage of classic TDMA—the requirement for high burst rate transmissions and high transmit power. Satellite switched TDMA can greatly increase the throughput of a transponder. Consider, for example, a satellite providing Internet access to individual users in the United States. The uplink and downlink beams at the satellite must provide coverage over an area approximately 6° by 3°, as seen from the satellite. Antenna gain and beamwidth are related by the approximate relationship G  33,000(product of beamwidths in degrees). This limits the maximum achievable satellite antenna gain to approximately 32.5 dB. A satellite with switched beam capability can have much narrower beams with higher gain than a satellite with a single fixed beam. The limitation on gain is the diameter of the antenna, which must fit inside the launch vehicle shroud. For launchers available in 2000, this limit is about 3.5 m. At 20 GHz, the uplink frequency for Ka band, an antenna with a circular aperture of diameter D  3.5 m and aperture efficiency of A  65% has a gain G  A (D)2  55.4 dB, and its beamwidth is approximately 75 D degrees  0.32°. The corresponding downlink antenna for 30 GHz that has a beamwidth of 0.32° and a gain of 55.4 dB has a diameter of 2.33 m. The switched beam satellite has an antenna gain almost 23 dB higher than the single beam satellite, which can be traded directly for reduction in uplink or downlink transmit power, and uplink or downlink data rate. However, the satellite must generate at least 170 beams to cover all of the United States with 0.32° beams, with a consequent increase in satellite antenna complexity. Satellite switched TDMA and multiple beam antennas are a feature of most of the proposed Ka-band Internet access satellites13,14. The Astrolink satellites, for example, have

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105 spot beams for links to small user terminals. The satellite uplink (30 GHz) antenna has a diameter of 2.5 m and the downlink antenna has a diameter of 3.25 m. There are five spot beams for links to hub stations; the large antennas used by the hub stations allow a lower gain antenna with a broader beam to be used on the satellite. Coverage of the United States with multiple beams is not always provided uniformly. Differences in population densities and the frequency of heavy rainfall make it advantageous to provide more system capacity to metropolitan areas, and also to provide higher link margins to areas with more frequent heavy rainfall, such as Florida and the southeastern states. In the most sophisticated of large GEO satellites, a steerable phased array antenna can be used, with control of beam pointing from the ground via the satellite’s telemetry and command link. The antenna beams can then be moved to provide coverage of areas with highest demand for traffic. The growth of the terrestrial optical fiber network will eventually fulfill the need for high-speed access to the Internet. Where direct access to an ISP is available via optical fiber, the transmission rate is likely to be higher and the cost to the user is likely to be lower. As the fiber network spreads through metropolitan areas, an Internet access satellite can concentrate its service on less well populated and rural areas. A steerable beam antenna allows the geographical capacity of the satellite to be reconfigured throughout its lifetime.

6.5 DEMAND ACCESS MULTIPLE ACCESS (DAMA) Demand access can be used in any satellite communication link where traffic from an earth station is intermittent. An example is an LEO satellite system providing links to mobile telephones. Telephone voice users communicate at random times, for periods ranging from less than a minute to several minutes. As a percentage of total time, the use of an individual telephone may be as little as 1%. If each user were allocated a fixed channel, the utilization of the entire system might be as low as 1%, especially at night when demand for telephone channels is small. Demand access allows a satellite channel to be allocated to a user on demand, rather than continuously, which greatly increases the number of simultaneous users who can be served by the system. The two-way telephone channel may be a pair of frequency slots in a DA-SCPC system, a pair of time slots in a TDM or TDMA system, or any combination or FDMA, TDM, and TDMA. Most SCPC-FDMA systems use demand access to ensure that the available bandwidth in a transponder is used as fully as possible. In the early days of satellite communication, the equipment required to allocate channels on demand, either in frequency or time, was large and expensive. The growth of cellular telephone systems has led to the development of low cost, highly integrated controllers and frequency synthesizers that make demand access feasible. Cellular telephone systems use demand access and techniques similar to those used by satellite systems in the allocation of channels to users. The major difference between a cellular system and a satellite system is that in a cellular system the controller is at the base station to which the user is connected by a single hop radio link. In a satellite communication system, there is always a two hop link via the satellite to a controller at the hub earth station. Controllers are not placed on the satellites largely because of the difficulties in determining which links are in use, and who will be charged for the connection. As a result, all connections pass through a controlling earth station that can determine whether to permit the requested connection to be made, and who should be charged. In international satellite communication systems issues such as landing rights require the owner of the system to ensure

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that communication can take place only between users in preauthorized countries and zones. The presence of the signals from all destinations at a central earth station also allows security agencies the option of monitoring any traffic deemed to be contrary to the national interest. Demand access systems require two different types of channel: a common signaling channel (CSC) and a communication channel. A user wishing to enter the communication network first calls the controlling earth station using the CSC, and the controller then allocates a pair of channels to that user. The CSC is usually operated in random access mode (see Section 6.6) because the demand for use of the CSC is relatively low, messages are short, and the CSC is therefore lightly loaded, a requirement for any DA link. Packet transmission techniques are widely used in demand access systems because of the need for addresses to determine the source and destination of signals. Section 6.7 discusses the design of packets for use in satellite communication systems. Bent pipe transponders are often used in demand access mode, allowing any configuration of FDMA channels to be adopted. There seem to be few standards for demand access systems in the satellite communication industry, with each network using a different proprietary configuration. Figure 6.11 shows a typical 54 MHz bandwidth Ku band transponder frequency plan for the inbound channels of a VSAT network using frequency division multiple access with single channel per carrier and demand access (FDMA-SCPCDA) on the inbound link. The individual outbound RF channels are 45 kHz wide, to accommodate the occupied bandwidth of 64-kbps bit streams transmitted using QPSK and RRC filters with   0.4. A guard band of 15 kHz is allowed between each RF channel, so one RF channel requires a total bandwidth of 60 kHz. A 54 MHz bandwidth transponder can accommodate 900 of these 60 kHz channels, but it is unlikely that all are used at the same time. Many VSAT systems are power limited, preventing the full use of the transponder bandwidth, and the statistics of demand access systems ensure that the likelihood of all the channels being used at one time is small. Considerable backoff is required in a bent pipe transponder with large numbers of FDMA channels, as discussed earlier in this chapter. The outbound link of this particular VSAT network is a continuous TDM bit stream transmitted through a separate transponder. A second transponder is used to allow for the differences in transponder gain needed for the inbound and outbound channels of the VSAT system. In VSAT systems, the inbound and outbound channels are usually symmetric, offering the same data rate in opposite directions. Internet access systems are often

Transponder bandwidth 54 MHz 64 kbps QPSK channels

CSC

2

3

4

60 kHz channel spacing

5

6

898

899

CSC

15 kHz guard bands between channels

FIGURE 6.11 Frequency plan for a 54-MHz transponder carrying 900 demand access channels. Each channel has an occupied RF bandwidth of 45 kHz and carries one 64-kbps signal. Channel 1 and channel 900 are common signaling channels (CSC) used by the demand assignment system to set up access to the other 898 channels.

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asymmetric, because requests for information can be short but the resulting replies may be lengthy. The packet length of the TDM signal in the outbound direction may be fixed, which suits a symmetrical network, or variable, which better suits an Internet channel capable of downloading large files or video from the Internet. The common signaling channels (CSC) shown in the inbound transponder in Figure 6.11 are located at the ends of the transponder bandwidth. When a VSAT earth station wants to access the satellite, it transmits a control packet to the satellite on the CSC frequency and waits for a reply. The control packet is received by the hub earth station and decoded. The control packet contains the address of a terrestrial or satellite destination for the call, DA, the address of the station requesting the connection, RA, any other relevant data (such as a character, CP, to indicate that this is a control packet with no traffic data), and a cyclic redundancy check (CRC) that is used in the receiver to check for errors in the packet. The control station records both origination and destination station addresses and measures the duration of the connection in order to generate billing data. In a true demand access system, the control station allocates the VSAT an uplink frequency and a time slot of specified duration in the outbound TDM frame. If the hub station has a large volume of data to send to a particular VSAT station, it can allocate a longer time slot in the TDM frame to that station. This is important in Internet access systems where a large file of video or other multimedia data may have to be sent. The timeslots usually come in multiples of a fixed minimum duration so that clock rates and buffer sizes are compatible. If the system becomes busy and many stations are requesting large files, throughput to any one station will slow down toward the standard minimum rate, exactly as in a terrestrial Internet server. The outbound link transmits a continuous bit stream so that receivers can maintain carrier phase and bit clock synchronization. The data is organized into a sequence of packets, addressed to the receiving stations, and organized into a frame. One frame contains one packet for each earth station, as illustrated in Figure 6.12. In many TDM systems there is always a time slot with addresses and other information for delivery of data to each earth station in the network, but there may be no data to send. In this case, the packet is assigned a special character to indicate no traffic. Once an inbound frequency and an outbound time slot are allocated to the VSAT station, the connection can be completed and data transfer or voice communication can begin. The hub earth station designed to receive FDMA-SCPC signals has multiple receivers operating at different center frequencies and allocated to the many transponders on the satellite. The block diagram of a receiver for one transponder with a bandwidth of 54 MHz is illustrated in Figure 6.13. The receiver amplifies and down-converts the received signal to an intermediate frequency of 700 MHz and then to a second IF at 70 MHz. Individual FDMA-SCPC channels within the band 43 to 97 MHz are down-converted to a standard IF frequency of 2 MHz in this example using local oscillators with frequencies 41 MHz through 95 MHz in steps of 60 kHz. There are a total of 900 such 2-MHz IF receivers to cover all the frequency slots in the transponder bandwidth.

End of TDM frame

A 896

Data

A 897

Data

A 898

Data

A 899

Data

C R C

E O F

Next frame

A 2

Data

FIGURE 6.12 End of a TDM frame outbound to VSAT stations. A, station address; CRC, cyclic redundancy check; EOF, end of frame marker.

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Downlink at f c GHz LNA

Image rejection BPF

Mixer

First LO f c − 700 MHz

700 MHz BPF

700 MHz IF amplifier

70 MHz BPF

Mixer

Second LO 630 MHz

70 MHz IF amplifier

D

B Ch 1 data

D

B Ch 2 data

D

B Ch N data

Channel selection filters Third LOs spaced 60 kHz

Mixers

2 MHz 2 MHz PSK Bit recovery BPFs IF amplifiers demodulators circuits

FIGURE 6.13 Receiver for FDMA-SCPC hub station. Channel selection is by bandpass filters in the 70 MHz IF section of the receiver and the third local oscillator. Each channel selection filter and third local oscillator is at a different frequency, spaced in 60 kHz increments across the IF band. D, demodulator; B, bit recovery circuit.

The hub earth station transmits a continuous TDM signal to all VSAT stations in the network. The symbol rate and the bandwidth of the TDM signal depends on the maximum bandwidth that the VSAT receiver can use, based on noise considerations. If the network is symmetrical and uses all 900 possible FDMA-SCPC channels at a bit rate of 64 kbps per channel, the TDM signal must have a bit rate of 900  64 kbps  57.6 Mbps, ignoring packet overhead. This bit rate is likely to be much too high for a VSAT station, resulting in low CN ratio in the receiver. The transponder can be partitioned to carry multiple groups of TDM signals with lower bit rates better suited to VSAT receivers. EXAMPLE 6.5.1 FDMA-SCPC-DA A VSAT network consists of 250 Ku-band VSAT earth stations sharing one inbound and one outbound transponder on a GEO satellite. The transponder bandwidth is 54 MHz. The transmit data bit rate for the VSAT stations is 64 kbps. Statistics for the VSAT network show that each VSAT generates an average data bit rate of 5 kbps with random time of arrival of data. The average outbound data rate per VSAT station is 20 kbps. The inbound data link operates in FDMA-SCPC using demand access, QPSK modulation with   0.5 RRC filters, and half rate forward error correction. The outbound data link uses a single continuous TDM stream, QPSK modulation, and a 16-bit CRC word in each packet. Determine the bit rate and bandwidths of the VSAT and hub station receivers, suggest a frame and packet size for the TDM link, an SCPC-FDMA frequency plan, and a demand access method. Inbound Link: VSAT to Hub The data rate at the VSAT station averages 5 kbps, but arrives in the form of variable length messages with random arrival times. The data are transmitted from the VSAT station on a half rate FEC

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bit stream at 128 kbps carrying 64 kbps of data, as a QPSK symbol stream at 64 kbaud. The link has   0.5 RRC filters, so the occupied bandwidth of the QPSK signal is Rs (1  )  96 kHz. The IF receivers in the hub earth station will have filters with a noise bandwidth of 64 kHz (equal to the symbol rate of the signal) and total bandwidth of 96 kHz. Allowing a 20% guard band between RF channels requires a carrier-to-carrier spacing of 115 kHz. The maximum number of channels that can be carried by the inbound transponder is 54,000115  469. Several of these channels would have to be designated as common signaling channels, so the maximum usable communication channels would be about 460. If the transponder is power limited, fewer channels can be carried. Each VSAT channel transmits data at 64 kbps, but data arrives at an average rate of 5 kbps. Assuming the data does not require transmission in real time, a buffer can store data until a total of 120 kb has been collected. The data are then packetized (see Section 6.7) and overhead bits are added for addresses, CRC, etc. The VSAT station can then send a request for a channel to the hub station and the data can be downloaded in about 2 s. The average time to collect 120 kb at the VSAT is 24 s. This scenario favors a SCPC-FDMA multiple access scheme using a CSC to obtain access to the satellite once every 24 s. If we assume that it takes 2 s to establish the connection and 2 s to transmit the packets, a total of 4 s of satellite transmission is used every 24 s, giving a VSAT station loading of 16.6%. This means that (in theory) six VSAT stations can share the same RF channel. In practice, it is impossible to load the channels to 100% of their capacity because data arrives at random time intervals causing temporary overload when a large volume of data arrives at the same time. If we assume a 66% load factor for the inbound link, we can share one inbound channel between four VSAT stations, which requires a total bandwidth of 544  13.5 MHz in the transponder. Demand access has clearly achieved considerable savings in bandwidth and power in this case. Outbound Link: Hub to VSAT The hub station transmits a continuous TDM frame consisting of many sequential packets addressed to each VSAT station. There are 250 VSAT stations with an average outbound data rate of 20 kbps each. If we apply half rate FEC to the outbound data stream and use QPSK modulation, we will transmit at 20 kbaud per station. With 250 stations, the average outbound symbol rate is 5 Mbaud, and the outbound data rate is 5 Mbps because we are using QPSK modulation with half rate FEC. The occupied bandwidth of the signal with   0.5 RRC filters is 7.5 MHz, and the VSAT receiver noise bandwidth will be 5.0 MHz. Let us assume a frame length of 5 s. On average each VSAT will receive 100 kbps during a 5-s period, but the statistics of the traffic suggest that there will be megabits of data for some stations and none for others in any given time period. Within the 5-s frame, if the frame time is divided equally, each earth station could receive data for a period of 20 ms, and would receive 10,000 data bits at 5 Mbps. If we assume a 5% overhead allowance for the packet, there will actually be 9500 data bits delivered during the 20-ms period, plus 500 overhead bits. If there are no data bits to be delivered to a given station, only the overhead portion of the packet, 500 bits, needs to be transmitted. That allows other stations to use the spare time in the frame to send additional data at a much higher than average rate. Demand access is most valuable when the traffic mix changes a great deal. The multiple access system described here was designed to meet the needs of the average data rates transmitted on the inbound and outbound links. If many of the stations are inactive, the other stations can have increased data rates. For example, suppose only 50 of the VSAT stations are active. Each VSAT station can transmit at its maximum data rate of 64 kbps, and will deliver data as fast as the terminal can supply it. To increase the inbound data rate above 64 kbps requires a wider channel bandwidth, and a receiver in the hub station with a wider IF filter. Alternatively, the VSAT station could transmit two carriers. The limitation on inbound data rate is likely to be VSAT EIRP and the resulting uplink CN ratio in the transponder. SCPC-FDMA does not offer as much flexibility to change data rates as TDM. On the outbound link with only 50 VSAT stations active, the packet length for the active stations can be increased by a factor of five. Short packets must still be sent to all stations to maintain synchronization of the VSAT receiver, but need consist only of a control packet, which is 500

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bits in length. The outbound link has a bit rate of 5 Mbps. Each VSAT station receives a control packet of 500 bits every 5 s if there is no data to be sent to that station. Hence the 200 inactive stations receive an average bit rate of 100 bps, and the 50 active stations can share the remaining bits. The bit rate for the 50 active stations is 5 Mbps  200  100 bps  4.98 Mbps, an average rate of 99,600 bps for each active station. The normal packet length is 20 ms and delivers 10,000 bits. The packet can now be extended to almost 100 ms for each of the active stations. The actual packet length is 100 ms  80 s  99.92 ms to allow for the time required in each frame to send a single control packet to each inactive VSAT. Thus the active stations receive 499,600 bits in each packet, of which 500 are control bits, giving a data rate of 99.82 kbps. TDM frames which offer variable packet length can easily accommodate a widely changing mixture of data rates delivered to each VSAT station. A field within the packet header can tell the VSAT station how many data bits are in the packet, allowing great variability. 

6.6

RANDOM ACCESS (RA) Random Access is a widely used satellite multiple access technique where the traffic density from individual users is low. For example, VSAT terminals and satellite mobile telephones often require communication capacity infrequently. These users can share transponder space without any central control or allocation of time or frequency, provided the average activity level is sufficiently low. In a true random access network, a user transmits packets whenever they are available. The packet has a destination address, and a source address. All stations receive the packet and the station with the correct address stores the data contained in the packet. All other stations ignore the packet, unless it is designated as a broadcast packet with information for all stations. In satellite communication systems, the network is more usually a star configuration, with a single hub and many small earth stations or portable terminals. Inbound packets are received by the hub earth station and forwarded to their destinations. Early work on random access techniques for radio channels was done at the University of Hawaii, where the system was called Aloha and was known by the generic term packet radio. Random access cannot be used when traffic density exceeds 18%, and therefore makes inefficient use of the bandwidth available in the transponder. Although there is a saving in transmission time because no call set up is required, the low throughput and poor spectral efficiency has restricted random access use in satellite systems to cases where traffic bursts are short and highly intermittent. In general, it is used on single SCPC-FDMA channels, rather than on whole transponders. The common signaling channel, described in the previous section, is an example of an SCPC-FDMA random access channel within a transponder that can successfully use random access because it is lightly loaded.

6.7 PACKET RADIO SYSTEMS AND PROTOCOLS Data transmission between computers or terminals requires agreed methods by which connections are established and data is transferred. When we make a telephone call, there are conventions and etiquette which define how a telephone connection is established and when each person speaks. For example, you decide to call your friend John Doe. You lift the telephone handset and hear dial tone. The telephone system is telling you that it is ready for you to dial a number. You dial the telephone number of your friend and wait to hear a ringing tone. The telephone system is telling you that it is trying to attract the

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attention of your friend. If your friend answers, you expect to hear “Hello, this is John Doe.” You might reply, “Hello, this is Bill Smith. How are you?” If the person who answers just says “Hello,” you cannot be sure that you have reached John Doe, and you might have to say “Hello, this is Bill Smith, is that John Doe?” If you dialed a wrong number and have connected to the wrong person, you might say “Sorry, I must have dialed a wrong number.” Then you would put the phone down and try again. If you weren’t sure whether the answerer said “John Doe” or “John Roe” because of noise on the line, you would ask for a repeat transmission. The process of creating a telephone connection to a friend makes use of signals provided by the telephone system and a set of procedures based in human etiquette and intelligence. Humans can readily determine whether they have reached the correct person, and how to proceed if the call does not go through correctly. A data transmission system lacks the intelligence of a human, and cannot readily adapt to changing responses in the way that humans can. Data transmissions make use of packets and protocols to ensure that automatic connection and transfer of data can be achieved reliably without human intervention. One dictionary definition of protocol is, “The code of ceremonial forms and courtesies of precedence, etc., accepted as proper and correct in official dealings, as between heads of state or diplomatic officials”15. It is this sense of the word protocol that is used to describe the rules by which two data terminals can connect to each other through a communication system and then transfer data. The creation of protocols for data transmission is a very large subject, with many books and papers devoted to the design and performance of different schemes. In this text, only the briefest summary of the subject is included. The widespread ownership of personal computers and growth of the Internet helped spur development of efficient and powerful protocols like TCPIP. The International Standards Organization (ISO) has created a seven layer model for machine to machine communication known as the open systems interconnection (OSI) which separates the functions of different parts of the system. The ISO-OSI model is shown in Figure 6.14. Although the model is widely quoted as describing the structure of data communication systems, it rarely seems possible to identify seven separate layers within any given system. The lowest layer, the physical layer, is the one with which this text is concerned—the transport of bits from one point to another. Regardless of the method of transportation, the ISO model assumes that bits are carried across the physical layer, in both directions, possibly arriving with some errors. The

Application Presentation Session Transport Network Data link control Physical

FIGURE 6.14 ISO-OSI seven layer model for digital communications links.

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remaining six layers of the model are embedded in the hardware and software of the terminals at each ends of the link. The second layer of the model provides error detection and correction, either in hardware or software, and the remaining layers are responsible for organizing the data transfer, from making connections to billing for the service. Terrestrial data communication has evolved through a series of protocols, beginning with a standard by IBM, known as HDL, through X.25 to ATM (asynchronous transfer mode). ATM uses a 53-byte packet and was designed to be transmitted over fiber-optic line networks using the DS-3 44.736 Mbps transmission rate according to an IEE standard (P802.6). Digital cellular radio systems, although using three different standards, are compatible, and any cellular telephone designed for TDMA can work with any provider’s TDMA system. Satellite communication systems have not evolved a set of standards, and many systems use proprietary protocols. For example, the Iridium, Globalstar, and ICO Global LEO/MEO satellite communication systems all use digital transmission with different protocols. Handsets designed for use with the Iridium system cannot communicate with Globalstar or ICO Global satellites. There is now intense interest in designing GEO satellite systems that are compatible with the ATM protocol so that wide band satellite links can be connected directly to terrestrial data networks16, 20 –23. Generically, this is known as wideband by satellite. Typically, the long delay time inherent in transmission via a GEO satellite creates problems when interfacing to a terrestrial protocol designed for much shorter delays. Special interfaces are needed at the earth stations that allow the protocol to be adapted for satellite use. Data cannot be transmitted as a continuous bit stream in most cases, because additional bits for addresses, error control, and other additional information that is not part of the message data must be inserted into the bit stream. Data are sent in packets, usually with an agreed length and content, using a structure that is very similar to the TDMA frame described in Section 6.3. Each packet typically consists of a header, which contains address and control information, a block of message data, and a closing section with error control bits and an end of packet flag. One protocol and packet design that has been widely accepted for use in amateur satellite systems is AX.25. The AX.25 protocol is based on X.25, a protocol developed for terrestrial data communications, and is used by amateur radio operators in a terrestrial data communication network. The protocol was adapted for use in amateur radio LEO satellites with VHF and UHF transponders operating in a store and forward mode. Several of these satellites were built and orbited, providing a method for amateur radio operators to send messages by satellite17. Figure 6.15 shows the structure of the AX.25 packet. All packets begin and end with a unique word, called a flag, 01111110, which is not allowed to appear in any other part of the packet. The flag marks the end of one packet and the start of the next packet, so that the receiving data terminal can extract the packet contents correctly. The general format of the packet contents is a header, followed by message bits, followed by a cyclic redundancy check (CRC). The header contains addresses, in the form of amateur radio call signs, for the sender and intended recipient, and control information that helps the receiving station identify the contents of the packets. The control bits, for example, tell the receiving terminal how long the packet is, and define whether this is a broadcast packet, intended to be viewed by all receiving stations, or a packet for a specific recipient. Control bits also specify the type of packet—some packets contain no message bits and are sent to convey system information. The CRC allows the receiver to check whether the packet was received correctly, and to call for a retransmission if an error is detected. The interested reader should refer to reference 17 for further details of the amateur radio satellite system. All data transmission system must have some form of protocol, and data is almost always sent in packet form. Thus whenever multiple access techniques are discussed and digital data are transmitted, it can be assumed that some form of packet transmission is used.

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Flag

Address

Control

FCS

Flag

8 bits

112–560 bits

8 bits

16 bits

8 bits

257

AX.25 Un-numbered frame

Flag

Address

Control

PID

Message

8 bits

112–560 bits

8 bits

8 bits

N × 8 bits

FCS

Flag

16 bits 8 bits

AX.25 Information frame Flag = 01111110

FCS = Frame check sequence

FIGURE 6.15 AX.25 packets. The unnumbered frame (packet) is used only for control, and contains no message data. All packets (frames) start and end with Flag, a unique word with six ones. Data messages are processed to ensure that six ones do not appear as a string anywhere except Flag. The value of N can be anywhere from 1 to 256. FCS; Frame control sequence.

6.8 CODE DIVISION MULTIPLE ACCESS (CDMA) Code division multiple access is a scheme in which a number of users can occupy all of the transponder bandwidth all of the time. CDMA signals are encoded such that information from an individual transmitter can be recovered by a receiving station that knows the code being used, in the presence of all the other CDMA signals in the same bandwidth. This provides a decentralized satellite network, as only the pairs of earth stations that are communicating need to coordinate their transmissions. Subject to transponder power limitations and the practical constraints of the codes in use, stations with traffic can access a transponder on demand without coordinating their frequency (as in FDMA) or their time of transmission (as in TDMA) with any central authority. Each receiving station is allocated a CDMA code; any transmitting station that wants to send data to that earth station must use the correct code. CDMA codes are typically 16 bits to many thousands of bits in length, and the bits of a CDMA code are called chips to distinguish them from the message bits of a data transmission. The CDMA chip sequence modulates the data bits of the original message, and the chip rate is always much greater than the data rate. This greatly increases the speed of the digital transmission, widening its spectrum in proportion to the length of the chip sequence. As a result, CDMA is also known as spread spectrum. Direct sequence spread spectrum (DSSS) is the only type currently used in satellite communication; frequency hopping spread spectrum (FH-SS) is used in the Bluetooth system for multiple access in short range local area wireless networks18. CDMA was originally developed for military communication systems, where its purpose was to spread the energy of a data transmission across a wide bandwidth to make detection of the signal more difficult (called low probability of intercept). Spreading the energy in a signal across a wide bandwidth can make the noise power spectral density (NPSD) in the receiver larger than the power spectral density (PSD) of the received signal. The signal is then said to be buried in the noise, a common feature of DS-SS signals,

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and the signal is much harder to detect than a signal with a PSD greater than the receiver’s NPSD. The correlation process that recovers the original data bits from a DS-SS spread spectrum signal is also resistant to jamming, the deliberate transmission of a radio signal at the same frequency to blot out someone else’s transmission. Both of these attributes are valuable in tactical military communication systems. CDMA has become popular in cellular telephone systems where it is used to enhance cell capacity. However, it has not been widely adopted by satellite communication systems because it usually proves to be less efficient, in terms of capacity, than FDMA and TDMA. The Globalstar LEO satellite system was designed to use CDMA for multiple access by satellite telephones; the advantage of CDMA in this application is soft handoff. More details of the Globalstar system can be found with reference 4. The GPS navigation system uses DS-SS CDMA for the transmission of signals that permit precise location of a receiver in three dimensions. Up to 12 GPS satellites could be visible to a receiver close to the earth’s surface at any one time. CDMA is used to share a single RF channel in the receiver between all of the GPS satellite transmissions. Chapter 12 gives details of the GPS signal structure and describes the process of data recovery from the DS-SS satellite signals.

Spread Spectrum Transmission and Reception This discussion of CDMA for satellite communications will be restricted to direct sequence systems, since that is the only form of spread spectrum that has been used by commercial satellite systems to date. The spreading codes used in DS-SS CDMA systems are designed to have good autocorrelation properties and low cross-correlation. Various codes have been developed specifically for this purpose, such as Gold and Kasami codes1. The DS-SS codes will all be treated as pseudonoise (PN) sequences in this discussion. Pseudonoise refers to the spectrum of code, which appears to be a random sequence of bits (or chips) with a flat, noiselike spectrum. The generation of a DS-SS signal is illustrated in Figure 6.16. We will begin by assuming that the system uses baseband signals. Most DS-SS systems generate spread spectrum signals using BPSK modulated versions of the data stream, but it is easier to see how a DS-SS system operates if the signals are first considered at baseband. In Figure 6.16, a bit stream containing traffic data at a rate v

v Modulator

1

t

−1

1 −1 −1 −1

Incoming bit stream Rb bps

1 1 1 1 1 −1

1 −1

−1 −1

Outgoing spread bit stream NRb bps

v

1 1 1

1 −1

1 1 1 −1 −1

1 −1

−1 −1

t

Spreading PN sequence NRb bps FIGURE 6.16 The basic principle of a direct sequence spread spectrum (CDMA) system. Each incoming message data bit is multiplied by the same PN sequence. In this example the message sequence is 1 1 and the PN sequence is 1 1 1 1 1 1 1.

t

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Rb, converted to have levels of 1 and 1 V corresponding to the logical states 1 and 0, is multiplied by a PN sequence, also with levels 1 and 1 V, at a rate M  Rb chips per second. Each data bit results in the transmission of a complete PN sequence of length M chips. In the example shown in Figure 6.16, the seven chip spreading code sequence is 1110100, which is converted to 1 1 1 1 1 1 1. The spreading sequence multiplies the data sequence 0 1, represented as 1 1, leading to the transmitted sequence 1 1 1 1 1 1 1 1 1 1 1 1 1 1 shown at the right in Figure 6.16. Recovery of the original data stream of bits from the DS-SS signal is achieved by multiplying the received signal by the same PN code that was used to generate it. The process is illustrated in Figures 6.17 and 6.18.

Shift Demodulated signal input

Shift register

b1 b 2 b 3 b4 b 5 b 6 b 7

−

−

+

−

+

+

+ Phase shifters

Output vo = −b1 − b 2 + b 3 − b4 + b 5 + b 6 + b 7

Adder Bits clocked Register contents in

Output

Bits clocked Register contents in

Output

1 1

1

1

1 1

vo = 1

8 1

1 1 −1 1 −1 −1 vo = 1

1 −1 1 1

1

1

1 1

vo = 3

9 1

1

2 −1 −1 1 1

1

1 1

vo = 5

10 1

3 −1 −1 −1 1

1

1 1

vo = 3

11 −1 1 1

1

1 1 −1 vo = 1

4 1 −1 −1 −1 1

1 1

vo = 3

12 1 −1 1

1

1

1 1

5 −1 1 −1 −1 −1 1 1

vo = 1

13 −1 1 −1 1

1

1

0 1

6 1 −1 1 −1 −1 −1 1 vo = −1

7 1

1 1 −1 1 −1 vo = 3

1 1

1 1 −1 1

14 −1 −1 1 −1 1

vo = 1

vo = 3

1 vo = 1

1 1

vo = 7

1 −1 1 −1 −1 −1 vo = −7

FIGURE 6.17 Data bit recovery using an IF correlator (matched filter). In this example the PN sequence is seven bits long for illustration. The CDMA chips from the receiver are clocked into the shift register serially and the shift register contents passed through phase shifters and added. The phase shifters convert 1 chips to 1 when the correct code is in the shift register such that all the voltages add to a maximum when the received sequence is correct. This figure shows the shift register contents and adder output for the chip sequence in Figure 6.16. Note that a high spurious output of 5 occurs at the third clock step, indicating that the seven bit sequence used here for illustration has poor autocorrelation properties.

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v

v Modulator

t

1 1 1 1 1 1 1 −1 −1 −1 −1 −1 −1

Incoming spread bit stream

t

Recovered bit stream

v

1 1 1

1 −1

1 1 1 −1 −1

1 −1

−1 −1

t

Despreading PN sequence FIGURE 6.18 A baseband correlator for dispreading CDMA signals. The original bit stream is recovered by multiplying the received signal by a synchronized copy of the PN sequence that was used in the transmitter.

At a CDMA receiver which knows the seven bit code, there will be a correlator that has the code stored as multiplier settings. Figure 6.17 illustrates the correlation process. Received chips are clocked into a shift register of length equal to the code sequence— seven stages in this case. The word in the shift register is identified as b1 b2 . . . b7. At each clock cycle the seven chip word with chip values bi in the correlator shift register are multiplied by 1 or 1, corresponding to the chips in the code sequence, by the blocks marked phase shifters in Figure 6.17. Note that received chips are clocked into the correlator from the left, so the code sequence appears reversed (written from right to left) in the phase shifters in Figure 6.17. The outputs of the phase shifters are added to give the output word v0  p1b1  p2b2  p3b3  p4b4  p5b5  p6b6  p7b7. The value of v0 will be 7 or 7 when the correct code sequence exactly fills the seven stages of the shift register. Figure 6.17 shows one process by which the code sequence can be detected. The shift register is originally filled with 1 chips, giving an output from the adder of v0  1. The sequence generated in Figure 6.16 is clocked in from the left. The adder output yields values of v0  1 5 3 3 1 1 as the chip sequence moves into the register. On the next clock cycle, when all seven bits of the sequence are in the seven stage register, the output of the adder is 7. A threshold detector after the adder with a threshold level of 6 would detect the threshold crossing and output a logical 0 for the first data bit. As the next 7 bits are clocked through the shift register, the output of the adder fluctuates between 1 and 5, reaching v0  7 or 7 when the code sequence fills the shift register. Note that the seven bit sequence used for illustration here would be a poor choice for a CDMA system because of the high spurious output from the adder at the third clock step. PN sequences used in CDMA systems are required to have good autocorrelation and good cross-correlation properties to minimize false threshold crossings. Multiple shift registers like the one shown in Figure 6.17 can be operated in parallel with each input delayed by an increment of one bit. If there are N shift registers, N possible code positions are tested at each chip clock cycle giving faster code acquisition. When the code sequence is long, the multistage shift register detector shown in Figure 6.17 becomes unwieldy. Chip-by-chip multiplication is used instead. The multiplier has inputs bi and pi, where bi is the received chip and pi is a stored PN sequence chip.

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The output of the multiplier is then integrated over the duration of the code sequence to yield a value N or N where N is the number of chips in the PN sequence. The process is illustrated in Figure 6.18 for the seven chip code of Figure 6.16. The incoming signal at the left side of the figure is multiplied by the despreading code sequence to give the output at the right of the figure. In practice, a low pass filter is used instead of an integrator to avoid the need to synchronize discharge (dumping) of the integrator contents when a bit is detected. If the correct code is present in the input signal, the output of the multiplier is 1 or 1 as each chip is present in the multiplier. In a practical DS-SS CDMA system, there will be several other CDMA signals present at the correlator input as well as the wanted code. The integrated value of the multiplier output for cross-correlation of any other code with the stored code will yield a value 1 or 1 if the codes have ideal crosscorrelation properties. Given a sufficiently large value for M (i.e., a long PN code sequence), it is possible to detect the bits of the wanted code in the presence of a large number of unwanted CDMA signals. The seven chip code used to illustrate DS-SS code correlation is not a good code for a CDMA system. It exhibits poor autocorrelation because at one position in the shift register the adder output is 5, too close to the peak value of 7. Noise added to the incoming code could easily push the correlator output above the threshold causing a detected bit error. Ideally, we would like to have CDMA codes of length M chips that have autocorrelation values of 1 or 1 everywhere except when the code is aligned correctly, when the value should be M. When a different code is clocked into the shift register in Figure 6.17, the cross-correlation should be 1 or 1 on all clock cycles. Very few known codes have these ideal properties. There are Barker codes with sequence lengths up to 23 chips that meet these requirement, but practical DS-CDMA systems normally use longer codes with nonideal correlation properties. For example, GPS satellites use 1023 chip Gold codes for the CA (course acquisition) code sequence that are built up from maximal length sequences which are easy to generate with a shift register. Practical CDMA systems use BPSK waveforms and correlate the received signals at IF rather than baseband. The shift register shown in Figure 6.17 is typically a single stage multiplier, as shown in Figure 6.18, and the incoming signal and the PN sequence are BPSK waveforms with 0° or 180° phase shifts. Multiplication of two identical, cophased BPSK waveforms yields an output of 1. If the phase of the input waveform is reversed (indicating that the original data bit was a 0 rather than a 1) the output is 1. Coherent phase detection is required so that the IF waveforms can be added in phase, but the correlation principle is the same. The main difficulty in DSSS CDMA receivers is that the received signal is buried in the noise, so the usual techniques for carrier recovery cannot be used. Baseband correlation is rarely used in DS-SS CDMA systems because the signals entering the correlator have CN ratios less than 0 dB (negative CN ), so the signals always look like noise. A complete DS-SS receiver and correlator for the GPS CA DS-SS signal is described in Chapter 12. The GPS CA code sequence is 1023 chips in length, so chip-by-chip multiplication is used in the receiver. The bandwidth occupied by the original data signal with bit rate Rb, if transmitted using BPSK and   0.4 square root raised cosine (RRC) filters, would be 1.4 Rb Hz. The spread spectrum signal occupies a bandwidth M  1.4 Rb Hz, and must be received through an IF RRC filter with a noise bandwidth of M  Rb Hz. Suppose that a BPSK receiver with the appropriate RRC filter with noise bandwidth Rb Hz receives the BPSK signal with CN  11 dB. If we do not change the power level of the original BPSK signal by the process of spreading it into a bandwidth of 1.4 M Rb Hz, the CN in the spread

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spectrum receiver will be 11  10 log10 M dB. If M is large (e.g., 1023 as in the GPS CA code) the CDMA signal will have a CN ratio in the receiver much less than 0 dB; 19.1 dB for the GPS CA signal example. The despreading process using a correlator to recover the original signal adds a processing gain equal to the numerical value M to the (CN)SS ratio of the received spread spectrum signal. Hence the signal-to-noise (SN)out ratio in the spread spectrum receiver after the correlator is given by 1SN2 out  1C N2 SS  10 log10 M

(6.14)

This SN ratio must be sufficiently high for the receiver to recover the bits of the transmitted signal with a reasonable bit error rate. For example, if a BER no larger than 106 is required, (SN)out must be greater than 11.0 dB, allowing a 0.4-dB implementation margin with no forward error correction.

DS-SS CDMA Capacity In a DS-SS CDMA system where there are a number of CDMA signals present at the input to each receiver, it is usual to treat the unwanted (interfering) CDMA signals as noise. If a receiver has an input containing Q signals, each at a power level C watts, and the receiver thermal noise power is Nt watts, the (CN)in ratio for the wanted signal is approximately 1CN2 in  10 log10 3C  1Nt  1Q  12  C2 4 dB

(6.15)

1SN2 out  10 log10 3C 1Nt  1Q  12  C2 4  10 log10 M dB

(6.16)

where [Nt  (Q  1)  C) watts is the total noise at the receiver input. The term (Q  1)  C  I watts is the power of the Q  1 interfering CDMA signals. (Note that Nt and C must be added in watts, not decibel units.) The correlator in the receiver adds a processing gain of 10 log10 M dB to the input CN, as seen in Eq. (6.14), and outputs a correlated signal with a signal-to-noise ratio (SN)out. Hence the output SN ratio for the bit stream in the receiver is given by If Q is a large number, it is probable that [Nt  (Q  1)  C]  (Q  1)  C watts, and then Eq. (6.16) reduces to 1SN2 out  10 log10 31  1Q  12 4  10 log10 M  10 log3M 1Q  12 4 dB (6.17) If Q is also large such that M W 1 then 1SN2 out  10 log10 1MQ2 dB

(6.18)

Examination of Eq. (6.18) shows that M, the number of chips in the spreading code must be 10 times larger than Q if the output SN ratio is to be greater than 10 dB, and that the system capacity is independent of the thermal noise power in the receiver. The bit rate of each signal is given by Rb  RcN  B 3N  11  a2 4

(6.19)

where Rc is the chip rate and B is the transponder bandwidth. The total bit rate for the transponder is given by M  Rb  B  M[Q  (1  )]. If M must be 10 times larger than Q to allow demodulation of the spread signal without many bit errors, the total bit rate through the transponder in bits per Hertz using CDMA will be numerically less than one-tenth of the bandwidth in hertz. This results in poor utilization of the RF bandwidth when CDMA is used, compared to FDMA or TDMA, as the following example demonstrates.

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EXAMPLE 6.8.1 CDMA in a Fixed Earth Station Network A DS-SS CDMA system has a number of earth stations sharing a single 54 MHz bandwidth Ka-band transponder. Each station has a different 1023 bit PN sequence which is used to spread the traffic bits into a bandwidth of 45 MHz. The transmitters and receivers use RRC filters with   0.5 and the chip rate is 30 Mcps. Determine the number of earth stations that can be supported by the CDMA system if the correlated output SN  12 dB. Equation (6.17) gives 1SN2  12 dB  10 log3 M 1Q  12 4  30.9  10 log 1Q  12 Hence 10 log 1Q  12  18.9 dB Q  77  1  76 Each of the carriers has a bit rate of 30 Mbps1023  29.33 kHz, so the transponder carries a total bit rate of 77  29.33 kbps  2.258 Mbps. A 54-MHz bandwidth transponder operated in FDMA or TDMA would have a much higher capacity. The capacity of the system can be improved by adding half rate forward error (FEC) control to the baseband signal to reduce the SN required for detection of the bits in the receiver. If the FEC system has a coding gain of 6 dB, we can use SN  12  6  6 dB. Using Eq. (6.18), because we now know M W Q 6 dB  10 log 1MQ2 gives Q  M4  255 channels. The data bit rate of each channel (before application of half rate FEC) is now 14.66 kbps and the total throughput of the transponder is 255  14.66 kbps  3.74 Mbps. This is still well below the capacity of a FDMA or TDMA system. We can conclude that CDMA is useful in commercial systems only where efficient use of satellite capacity is not important, or where the ease with which stations can leave and join the network outweighs the loss of efficiency, or where power limitations in the transponder ensure that it cannot be heavily loaded. 

EXAMPLE 6.8.2 CDMA in an LEO Satellite Network An LEO satellite communication systems uses direct sequence CDMA as the multiple access method for groups of terminals within each of its multiple antenna beams. The terminals generate and receive compressed digital voice signals with a bit rate of 9.6 kbps. The signals are transmitted and received at a chip rate of 5.0 Mbps as BPSK modulated DS-CDMA. In the absence of any other CDMA signals, the input power level at the receiver input is 146.0 dBW for one CDMA signal, and the noise temperature of the receiving system is 300 K. The satellite transmits 31 simultaneous CDMA signals. Find the SN ratio for the 9.6-kbps BPSK signal after despreading, and estimate the BER of the data signal, given a system implementation margin of 1 dB. If two of the multiple beams from the satellite overlap, so that a second group of 31 DS-CDMA signals is present at the receiver, find the BER of the wanted signal. The thermal noise power in the receiver is Nt  kTsBn. For the chip rate of 5.0 Mbps with BPSK and ideal RRC filters, Bn  5.0 MHz. Hence Nt  228.6  24.8  67.0  136.8 dBW  2.09  1014 W There are 30 interfering CDMA signals overlaid in the 5-MHz bandwidth of the receiver filter. The total interfering power is I  30  Pr  146.0  14.8 dB  131.2 dBW  7.59  1014 W The carrier-to-noise plus interference ratio must be calculated in watts, not dBW, because we cannot add noise and interference in decibel units, only in watts. The carrier-to-noise ratio in the

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receiver for the wanted CDMA signal is C 1Nt  I2  2.51  1015  12.09  1014  7.59  1014 2  2.5196.8  0.0259  15.9 dB The carrier power is well below the noise plus interference power, so the wanted carrier is hidden below the noise and interference. This is called a low probability of intercept signal. CDMA was first used by military radio communication systems because detection of a signal which is below the noise floor is difficult. The coding gain, Gc, for the CDMA receiver is given by the chip rate divided by the bit rate Gc  RcRb  5.0 Mcps9.6 kbps  520.8  27.2 dB Hence, after correlation of the wanted code (despreading), the SN ratio of the 9600-bps BPSK signal is SN  15.9  27.2  11.3 dB With an implementation margin of 1 dB, the effective SN is 10.3 dB  10.7 as a power ratio. For BPSK, the BER is Pe  12 erfc3 11CN2 eff ratio 4  12 erfc33.274  2  106 If a second group of 31 signals is present at the receiver from an overlapping satellite beam, there will be additional interference which lowers the C(N  I) ratio. The interfering power from 31 signals is I  31  Pr  146.0  14.9 dB  131.1 dBW  7.76  1014 W Hence the new C(Nt  I) ratio is

C 1Nt  I 2  2.51  1015  12.09  1014  7.59  1014  7.76  1014 2  2.51174.4  0.0144  18.4 dB

After correlation of the wanted code the SN ratio of the 9600 BPSK signal is SN  18.4  27.2  8.8 dB With an implementation margin of 1 dB, the effective SN is 7.8 dB  6.02 as a power ratio. For BPSK the BER is Pe  12 erfc3 11CN2 eff ratio 4  12 erfc32.454  3  104 We would need to add forward error correction to the baseband signal to improve the bit error rate. To achieve a BER of 106 in this case, a coding gain of about 3 dB would be adequate. With half rate convolutional coding, a coding gain of 5.5 to 6 dB is typical, which would provide a margin of 3 dB over a BER of 106 and a baseband data rate of 4.8 kbps. This bit rate will support a single digital speech channel with LPC linear predictive coding compression. The advantage of overlapping beams in a mobile satellite system is that the wanted signal can be transmitted by both satellites (using different CDMA codes) and blockage of one beam by an obstruction on the ground does not cause loss of the signal if the other beam can still be received. The wanted signal from both satellites can be combined at baseband using a rake receiver, which improves the BER. With optimum combining of the same baseband signal, the BER will be the same as for a single beam with 31 users. 

EXAMPLE 6.8.3 GPS The Global Positioning System (GPS) uses direct sequence CDMA for both the CA and P code transmissions. The design and operation of GPS is discussed in detail in Chapter 12, from which this example of a direct sequence CDMA system is drawn.

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The CA code transmissions from GPS satellites are 1023-bit PN sequences, formed into 64 Gold codes. At any given time, no more than 10 GPS satellites are visible, so interference with the wanted signal is limited to no more than nine overlaid CDMA signals. In this example we will assume for simplicity that the signals are all received with equal power. There are variations in the transmitted power between satellites, and a satellite close to the horizon has a longer path length, so there will be variations in the received power level of individual satellite signals in practice. The received power level for a typical CA signal is given by the downlink budget in the table below, assuming 0 dB receiving antenna gain. The CA code is transmitted at a bit rate of 1.023 Mbps using BPSK modulation. The receiver noise bandwidth is assumed to be 2 MHz. Satellite EIRP (dBW) Path loss (dB) Receive antenna gain (dB) Pr (dBW)

26.8 186.8 0 160.0

The interference from nine satellite spread spectrum signals of equal power for the CA code is given by I  160.0  9.5  150.5 dBW  8.91  1016 W The thermal noise power in a noise bandwidth of 2 MHz for a system noise temperature of 273 K is kTs Bn where Nt  141.2 dBW  7.59  1015 W The noise and interference powers must be added in watts, not in decibels: Nt  I  8.48  1015 W  140.7 dBW The nine interfering satellites have lowered the CN ratio for the wanted signal by 0.7 dB, a relatively small decrease. In a GPS CA code receiver, thermal noise is the dominant factor, not interference from other satellites as in Example 6.8.2. The CN for one CA code signal with nine interfering signals is C 1Nt  I2  160.0  1140.72  19.3 dB The theoretical coding gain for a 1023-chip code sequence is 10 log10 1023  30.1 dB. Hence the SN ratio of the correlated CA signal, assuming ideal correlation, is SN  19.3  30.1  10.8 dB The GPS CA code signal recovered from the correlator is a 1-kbps polar binary waveform with amplitude V volts. The primary use of this signal in a commercial GPS receiver is to determine the time of arrival of the CA code sequences, as it is the time that code sequences arrive from each of the GPS satellites that provides the pseudorange information from which the position of the receiver is calculated. The CA code also provides vital navigation message data that is needed in the position calculation. (See Chapter 12 for details.) Navigation data are modulated onto the 1-kbps signal at 50 Hz, so there are 20 V or 20 V samples in succession for each 1 or 0 data bit of the navigation message. Allowing an implementation margin of 1 dB in recovery of the signal, the effective SN after the correlator is 9.8 dB which will give a BER around 6  106. Averaging 20 samples will improve the error rate by a factor of 120, giving a BER close to 106. At a 50-bps data rate, errors in the navigation message will rarely occur. (Theoretically, there will be a bit error in the navigation message once every 2 days. The entire message repeats every 12.5 min, and vital data repeats every 30 s. The error is quickly overwritten by new data.) In this example we see that GPS satellites make excellent use of CDMA as a way to obtain a high-speed bit stream from which timing information can be obtained, an essential ingredient for

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any time of arrival position location system, and also a low speed data stream that provides the navigation information for the solution of the location problem. The example here uses nine interfering satellites with the same power level. Most GPS receivers select the four strongest satellite signals to use in the position location solution. A more realistic scenario would have four satellites at the maximum receive power level and the remainder at a lower level, since GPS satellites orbit in constellations of four, with one constellation always visible, to improve the accuracy of position location measurements. Thus we should expect less than 0.7 dB degradation in CN due to interference by other satellites’ CDMA signals, and the probability of a bit error in the navigation message then becomes very small. Interestingly, the eastern European satellite navigation system known to the Western world as GLONASS uses FDMA for multiple access. There are a maximum of 64 individual CA signals in the GPS system, from a possible 64 GPS satellites, each of which could be allocated its own RF frequency within a 1 MHz band. If all the parameters of the GPS system were held the same, but multiple accesses were changed from DS-CDMA to FDMA, a multichannel receiver could receive four (or more) 1 kbps BPSK data streams with CN ratio of 10.8 dB. This is essentially how the GLONASS system works, with accuracy comparable to that achieved in the GPS system and a much simpler receiver. 

6.9

SUMMARY

Multiple access is the process by which a large number of earth stations interconnect their links through a satellite. In frequency division multiple access (FDMA), stations are separated by frequency, while in time division multiple access (TDMA), they are separated in time. In code division multiple access (CDMA), stations use spreadspectrum transmissions with orthogonal codes to share a transponder without interference. Multiple access may be preassigned or demand (DAMA), depending on whether or not it responds to changing traffic loads. Frequency division multiple access is the most widely used multiple access scheme. In it each earth station is assigned frequency bands for its uplink transmissions. Because of the TWT backoff required to reduce intermodulation distortion with bent pipe transponders, the spectral efficiency (i.e., the number of channels that can be carried per megahertz of bandwidth) degrades with the number of stations that access a transponder. FDMA is widely used with VSAT earth stations and SCPC systems where the uplink from the earth station is at a low power level. In time division multiple access (TDMA), earth stations transmit in turn. Since only one carrier is present at a time, no TWT backoff is required and thus full transponder EIRP is available. TDMA performance does not degrade with the number of accesses. TDMA transmissions are organized into frames; a frame contains one or two reference bursts that synchronize the network and identify the frame and a series of traffic bursts. Each participating sta-

tion transmits one traffic burst per frame. Frames and individual traffic bursts are identified by standardized bit sequences called unique words. One of the major technical problems in implementing TDMA is synchronization. Once synchronization is acquired, it must be maintained dynamically to compensate for orbital motion of the spacecraft. TDMA is often combined with FDMA, so that a small number of earth stations share a TDMA frame forming one FDMA access to a transponder. This is called MF-TDMA. In code division multiple access (CDMA) stations transmit at the same time and in the same frequency bands using spread-spectrum (SS) techniques. CDMA avoids the centralized network control required for synchronization in TDMA, but tends to achieve rather poor spectral efficiency. The Globalstar LEO satellite system was designed to use CDMA, with the advantage that an earth station can receive the same signal from more than one satellite at the same time, allowing soft handoff between satellites. Random access is used in systems that have low traffic requirements and can tolerate less than 18% utilization of the RF channels. The advantage of random access is that no central network control is needed. Digital links between computers require protocols to ensure efficient transfer of data, and invariably use some form of packet communication. Satellite systems have tended to use proprietary protocols, with the result that different satellite systems are not compatible.

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REFERENCES 1. L. W. COUCH, Digital and Analog Communication Systems, Prentice-Hall, Englewood Cliffs, NJ, 6th Ed., 1998. 2. FERREL G. STREMLER, Introduction to Communication Systems, Addison-Wesley, Reading, MA, 3rd Ed., 1984. 3. J. L. EVERETT, (ed.) VSATs, Peter Peregrinus, IEE UK, 1992. 4. www.globalstar.com 5. K. MIYA, ed., Satellite Communications Technology, KDD Engineering and Consulting, Tokyo, Japan, 1981. 6. TRI T. HA, Digital Satellite Communication, McGrawHill, New York, 1990. 7. Intelsat TDMAIDSI System Specification (TDMAIDSI Traffic Terminals), (BG-42-65E Rev. 2), Intelsat, Washington, DC, June 23, 1983. 8. Technical Requirements for Inmarsat Standard-A Ship Earth Stations (Issue 2), International Maritime Satellite Organization, London, UK, February 1983. 9. ROBERT M. GAGLIARDI, Satellite Communications, Lifetime Learning Publications, Belmont, CA, 1984. 10. K. FEHER, Digital Communications: Satellite Earth Station Engineering, Prentice-Hall, Englewood Cliffs, NJ, 1983. 11. V. K. BHARGAVA, D. HACCOUN, R. MATYAS, and P. NUSPL, Digital Communications by Satellite, John Wiley & Sons, New York, 1981.

12. J. J. SPILKER, Jr., Digital Communications by Satellite, Prentice-Hall, Englewood Cliffs, NJ, 1977. 13. Astrolink URL. 14. Spaceway URL. 15. Webster’s New World Dictionary, 2nd Ed., William Collins  World Publishing Company, 1980. 16. IAN F. AKYILDIZ and SEONG-HO JEONG, “Satellite ATM Networks: A Survey,” IEEE Communications Magazine, Vol. 35, No. 7, pp 30–39, July 1997. 17. DAVIDOFF, Amateur Satellite Handbook, American Radio Relay League. AARL, Newton, CT. 18. www.Bluetooth.com 19. www.globalstar.com 20. “Broadband via Satellite,” IEEE Communications Magazine, Special Issue, July 1997. 21. “Broadband Satellite Network Performance,” IEEE Communications Magazine, Special Issue, March 1999. 22. C. K. TOH and V. O. K. LI, “Satellite ATM Network Architectures,” IEEE Network, September/October, 1999. 23. B. G. EVANS and R. TAFAZOLLI, “Future Multimedia Communications via Satellite,” International Journal of Satellite Communications, 14, 467–474, 1996.

PROBLEMS 1. You are designing an FDM-FM-FDMA analog link that will occupy 36 MHz of an INTELSAT VI transponder. The uplink and downlink center frequencies of the occupied band are 5985.5 MHz and 3760.5 MHz. The distance from the satellite to your earth station is 40,000 km. The saturation uplink flux density for your uplink is 75 dBW/m2 and the satellite’s GT is  11.6 dBK1. At saturation the transponder EIRP for your downlink is 29 dBW and the earth station’s GT is 41 dBK1. The transponder is linear in that the EIRP in dBW is BO dB below the saturation value when the uplink flux density is backed off BO dB below saturation. The intermodulation carrier-to-noise ratio, (CN), in dB, is related to the backoff BO in dB by 1CN2 I  7.86  0.714  BO In other words, at saturation the value of (CN)I is 7.86 dB. Find the maximum overall carrier-to-noise ratio (CN), in dB that this link can achieve. What backoff must be used to achieve it? (When you need a frequency in your calculations, use the uplink or downlink center frequency as appropriate.) Make your calculations for beam center.

Problems 2 through 5 all involve a satellite and earth stations with the same specifications. Five earth stations share one transponder of a 64 GHz satellite. The satellite and earth station characteristics are given below: Satellite

Transponder BW

 36 MHz

Transponder gain

 90 dB (max)

Input noise temp.

 550 K

Saturated output power  20 W (max) 4-GHz antenna gain

 20.0 dB

6-GHz antenna gain

 22.0 dB

Earth station 4-GHz antenna gain

 60.0 dB

6-GHz antenna gain

 63.0 dB

Receive system temp.  100 K Path loss

At 4 GHz, Lp At 6 GHz, Lp

2. The stations all operate in a Speech signals are sampled at 8 bits/sample. The sampled signals multiplexed into 40-Mbps streams using QPSK.

 196 dB  200 dB TDMA mode. 8 kHz, using (PCM) are then at each station,

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a. Find the bit rate for each PCM signal. b. The number of speech signals (as PCM) that could be sent by each earth station, as a single access, with no overhead (i.e., no header or CRC, etc.). This is a TDM data stream. c. The shortest frame time for any TDMA scheme. 3. Assume that the TDMA system uses a 125-s frame time. Find the number of channels that each earth station can send within the TDMA frame when: a. No time is lost in overheads, preambles, and the like. b. A 5-s preamble is added to the beginning of each earth station’s transmission. c. A 5-s preamble is added to each station’s transmission and 2-s guard band is allowed between every transmission. 4. A 750-s frame time is used instead of a 125-s frame. Find the new channel capacities of the earth stations for the cases in Problem 3 above. 5. Find the earth station transmitter power and received (CN) when the system is operated: a. In TDMA with the transponder saturated by each earth station in turn. b. In FDMA with 3-dB input and output backoff. 6. A digital communication system uses a satellite transponder with a bandwidth of 54 MHz. Several earth stations share the transponder using QPSK modulation using either FDMA or TDMA. Standard message data rates used in the system are 80 kbps and 2.0 Mbps. The transmitters and receivers in the system all use ideal RRC filters with   0.25, and FDMA channels in the satellite are separated by 100kHz guard bands. When TDMA is used, the TDMA frame is 125 s in length, and a 2-s guard time is required between each access. A preamble of 148 bits must be sent by each earth station at the start of each transmitted data burst. a. What is the symbol rate for the 80-kbps and 2.0Mbps QPSK signals sent using FDMA? b. What is the symbol rate of each earth station’s transmitted data burst when TDMA is used? c. Calculate the number of earth stations that can be served by the transponder when 80-kbps channels are sent using (i) FDMA and (ii) TDMA. d. Calculate the number of earth stations that can be served by the transponder when 2.0-Mbps channels are sent using (i) FDMA and (ii) TDMA. 7. The capacity of the TDMA system described in Problem 6 can be increased substantially by using satellite switched TDMA. In a group of earth stations, each station sends a 2.0-Mbps signal to every other

earth station in every frame. It takes 1 s to reposition the satellite antenna beam from one earth station to another. Only the downlink antenna beam is switched; the uplink uses a common zone beam. The frame length to be used is 1000 s, with a 148-bit preamble and 2-s guard times between transmissions arriving at the satellite The extra antenna gain at the satellite is traded for an increase in the data rate by using 16-QAM on the downlink. Other parameters of the system are unchanged. a. Find the number of earth stations that can share the transponder. b. Find the total data throughput of the transponder after all preamble bits have been removed. 8. An LEO satellite system transmits compressed digital voice signals to handheld terminals (satphones). The satphones work in groups of 10. The inbound bit stream from the satphone to the satellite is at 10 kbps. The data are sent as a BPSK signal. The outbound bit stream from the satellite is at a bit rate of 100 kbps, and consists of packets addressed to each of 10 satphones. This signal is sent using QPSK, and all 10 satphones receive the 100 kbps bit stream. The system operates in L band where rain fading can be ignored, but blockage from buildings and trees is a significant factor. The satellite uses onboard processing and multibeam antennas. The links use square root raised cosine (RRC) filters with   0.5. In this question we will be concerned only with the links between the satellite and the satphones, and ideal RRC filters will be assumed. a. What is the noise bandwidth of the narrowest bandpass filter in: (i) the satphone receiver and (ii) the satellite receiver for the inbound link? b. What is the occupied RF bandwidth of the radio signals of: (i) the inbound link (phone to satellite) and (ii) the outbound link (satellite to phone)? c. The inbound link has clear air (CN)0  18.0 dB and the BPSK demodulator on the satellite has an implementation margin of 0.5 dB. What is the clear air BER in the baseband of the satellite receiver? d. What is the available fade margin [for (CN)0 on the uplink to the satellite] if the inbound link operating threshold is set at BER  104? e. The outbound link has clear air (CN)0  18.0 dB and the QPSK demodulator in the satellite phone has an implementation margin of 0.8 dB. What is the clear air BER? f. What is the available fade margin [for the overall (CN)0 on the downlink to the satphone] if the outbound link operating threshold is set at BER  105?

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9. A Ka-band satellite broadcasts digital television signals over the United States. The nominal bit rate of the signal is 28 Mbps. The digital signal can convey up to 10 prerecorded NTSC video signals. QPSK modulation is used, and error mitigation techniques are employed that provide an effective coding gain of 6 dB. [Coding gain of 6 dB means that when the (CN)0 value of the received signal is X dB, the BER corresponds to CN  (X  6) dB.] The QPSK demodulator in the receiver has an implementation margin of 1.6 dB. The transmitters and receivers use ideal RRC filters with   0.25. a. What is the occupied bandwidth of the RF TV signal? b. What is the symbol rate of the transmitted QPSK signal, and the noise bandwidth of the earth terminal receiver? c. The minimum permitted BER after error mitigation in the receiver is 106. What is the minimum permitted (CN)0 for the digital TV receiver? d. The Ka-band link suffers rain attenuation that reduces (CN)0 in the receiver by 7 dB for 0.1% of the year. If the BER is 106 under the 0.1% year conditions, what is the clear air (CN)0 value? e. A new coding algorithm is developed that provides a coding gain of 7 dB with a bit rate that increases to 30 Mbps. Assuming that the RRC filters in the system can be changed to match the new symbol rate, does implementation of the new coding algorithm improve the system performance? If so, what is the new (CN)0 margin? 10. This problem is about multiple access techniques in the inbound link of a VSAT network. This set of questions compares the operation of a Ku-band satellite transponder in FDMA, in TDMA, and in FDMARA. There are three parts to the problem. Part 1 100 VSAT stations in a star network share one 54MHz transponder using FDMA. Each VSAT station has a solid-state transmitter with an output power of 1 W and an EIRP of 41 dBW from a 1.1-m diameter antenna. The transmitted data signals have a bit rate of 128 kbps and are transmitted using QPSK modulation and half rate FEC, giving a symbol rate of 128 ksps. At the hub station, the overall CN ratio for each signal received from a VSAT station is 16 dB in clear air. The (CN)up ratio for one channel in satellite transponder is 19.0 dB, and the (CN)dn ratio for one channel in the hub receiver is 19.0 dB. The threshold CN ratio in any hub station receiver for BER  106 is 9.0 dB. This includes the receiver implementation margin of 0.5 dB.

269

The stations share the transponder using FDMA, with 51-kHz guard bands between the edges of the RF signals. The RRC filters used in the VSAT transmitters and the hub station receivers have a roll-off factor   0.4. To minimize intermodulation between signals, the transponder is operated with 3-dB output back off. a. Calculate the RF bandwidth occupied by each VSAT transmission. b. Calculate the maximum number of VSAT stations that can be included in the network if the transponder is bandwidth limited. c. Calculate the clear air CN ratio for a received signal at the hub station, and the link margin, if the number of VSAT stations in the network is increased to the number you calculated in (b) above. Remember that the power available from the transponder is fixed. Adding more stations to the network lowers the power per channel at the transponder output. Part 2 The VSAT network described in Part 1 is modified to be operate with TDMA on the VSAT uplinks instead of FDMA. There are 100 VSAT stations in the network. The TDMA frame has a duration of 2 ms and is made up of 100 bursts from the 100 VSAT stations. There is a preamble of 100 symbols at the start of each VSAT station burst, and each burst is separated from the next burst by a guard time of 1.0 s. a. There are 100 VSAT station RF bursts in each frame of 2.0 ms, and 100 guard times of 1.0 s. What is the duration of each station’s burst? b. Each VSAT station must deliver 128 kbps of data, in the form of 128 k symbols, every second. How many data symbols are there in each RF burst, and what is the total number of symbols per burst after accounting for the 100 symbol preamble at the beginning of each burst? Hence find the burst rate for the VSAT transmissions in symbols per second. c. If all the VSAT stations, and the hub receiver, have RRC filters with roll-off factor   0.4, what is the RF bandwidth occupied in the transponder? If the symbol rate of transmissions were increased until all 54-MHz bandwidth of the transponder were filled, what is the maximum number of VSAT stations in the network? d. The transponder can be operated with 1-dB output backoff when TDMA is used, and the implementation margin of the hub receiver is 1.5 dB. The EIRP of the VSAT stations must be increased because the noise bandwidth of the hub receiver has increased. By comparing the symbol rate with 100 FDMA VSAT

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stations in Problem 1 with the TDMA symbol rate for 100 VSAT stations in part (b) above, estimate the decibel increase in EIRP required from each VSAT transmitter. Comment on the feasibility of transmitting this power level from a VSAT station. Part 3 The FDMA system described in Part 1 is used with random access to serve a very large number of VSAT stations. All the parameters of Part 1 are the same, except that each station has a small amount of data to send at varying intervals of time. The average message data rate for each VSAT station is 5.0 kbps and the maximum permitted channel loading is 12%. a. How many VSAT stations can share each RF frequency? b. What is the maximum number of VSAT stations in the network when the number of RF channels is the value you calculated in Part 1(c)? 11. This problem examines the use of a Ka-band satellite to provide connection to the Internet from a small two-way terminal. The problem is in three parts. The first part establishes the design of the communications links and terminals. The second part examines the capacity of the satellite. The third part looks at changes that must be made to support portable terminals. Part 1 Communication links Description of the satellite communication system A Ka-band GEO satellite is located at longitude 100° W. Star networks can be built with a single hub station, two transponders on the GEO satellite, and a number of earth stations, identified here as VSATs. The major parameters of the system components are given below. You may not need all of these parameters to answer the questions, and additional parameters are given in the individual questions. The Ka-band satellite serves the United States. Coverage of the 48 contiguous states is achieved by a regional beam, but the satellite also carries an advanced antenna system with satellite switched spot beams that allow data packets to be transmitted to small earth stations with a high EIRP. This allows high-speed data transmission in the outbound link. The system is designed primarily to support Internet access via satellite, with highly asymmetrical links. Requests for access to the Internet are made by users at a low data rate through the satellite’s region beam. Replies from the Internet can be received at a high data rate using the satellite’s spot beam. System values Uplink frequency for transponder 1

28.2 GHz

Downlink frequency for transponder 1 21.7 GHz

Uplink frequency for transponder 2

28.1 GHz

Downlink frequency for transponder 2 21.6 GHz Range to satellite (all stations)

38,000 km

Satellite transponders Saturated output power

30 W

Transponder bandwidth

54 MHz

Transponder input noise temperature

500 K

Antenna gain, on axis, regional beam

33 dB

Antenna gain, on axis, switched spot beam

48 dB

VSAT station parameters Transmitter output power

1.0 W

Transmit frequency

28.2 GHz

Receive frequency

21.7 GHz

Antenna diameter Aperture efficiency Receiver system noise temperature (clear air)

0.5 m 65% 250 K

Receiver system noise bandwidth

TBD

Hub station parameters Maximum transmit power

100 W

Transmit frequency

28.1 GHz

Receive frequency

21.6 GHz

Receiver system noise temperature (clear air)

250 K

Antenna diameter Aperture efficiency

4.0 m 65%

Receiver system noise bandwidth

TBD

Atmospheric losses and miscellaneous losses In clear air at 28 GHz

2.0 dB

In clear air at 21 GHz

2.0 dB

Constants: Boltzmann’s constant, k,  1.38  1023 JK  228.6 dBW/K/Hz Part 1 Problems CN ratios in clear air conditions Make all calculations for the worst case of a VSAT station that is located on the 3 dB contour of the satellite antenna beam (regional or spot), and for a hub station on the 2 dB contour of the regional beam. The spot beam is used only for transmissions at 21 GHz from the satellite to the customers’ earth stations. All other links use the satellite’s regional beam. a. Calculate the free space path loss for a 38,000-km path at 28.2 and 21.7 GHz. b. Calculate the gains of the hub and VSAT antennas at frequencies of 28.2 and 21.7 GHz.

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PROBLEMS

c. Calculate the inbound overall CN in the hub station receiver in a noise bandwidth of 128 kHz when the VSAT has a transmitter output power of 1 W and accesses the regional beam on the satellite. Make the overall CN calculation for a single QPSK signal which is transmitted by transponder 1 at an output power of 1 W. d. Calculate the outbound overall CN in transponder 2 with a hub station transmit power of 1 W. Make your calculation in a receiver noise bandwidth of 1 MHz, for a single QPSK signal, with the output power of transponder 2 set at 1 W and the spot beam of the satellite transmitting to the customers’ terminals. Estimate the beamwidth of the spot beam from the satellite. Using a map of the United States, estimate the minimum number of spot beam positions required to serve the entire United States. Part 2 System performance Connection to the Internet is achieved by the following procedure. The customer sends a connection request, in the form of a data packet, to the hub station via the satellite and its regional beam. The hub station decodes the request and notes the location of the station. The connection between the Internet and the hub is established, through an Internet Service Provider (ISP) and the public switched telephone network (PSTN). A response from the ISP is sent back to the customer using the satellite’s spot beam. Since the packet from the customer contains the VSAT station location, the hub station can send instructions to the satellite to point the spot beam in the correct direction when transmitting packets to the customer. Note that with a linear transponder (bent pipe) on the satellite, the beam pointing instructions must be sent to the satellite at the same time as the data packet. The links between the ISP and the customer in this system are highly asymmetric. The customer can send only short requests at a low data rate. The ISP can dump data to the customer at a high data rate, mainly because of the high EIRP of the satellite’s spot beam transmissions. This mode of operation suits applications where the customer is browsing the Internet for information, or is requesting large files or video frames from the Internet. It works less well for sending files from the customer to the Internet—as is done with e-mail, for example. In this problem you are asked to design a VSAT network based on your results from Part 1. Ka-band links are subject to high attenuation in rain. The outbound link is required to achieve a 99.9% availability for a typical VSAT station for which slant path attenuation exceeds 7 dB at 21.7 GHz and 12 dB at 28.2 GHz, for 0.1% of an average year. The inbound

271

link is required to achieve a 99.7% availability for a typical VSAT station for which slant path attenuation exceeds 4 dB at 21.7 GHz and 7 dB at 28.2 GHz, for 0.3% of an average year. The link is declared unavailable if the BER exceeds 106 in the data stream supplied to the customer, or output by the hub station. Begin your analysis by assuming that 20 active VSAT stations share the output power of transponder 1 equally at all times using QPSK-SCPC-FDMA. Half rate FEC coding is used in the inbound and the outbound link and provides a coding gain of 5 dB at a BER of 106 in the recovered data stream. The implementation margin of the QPSK demodulators in the hub receiver is 0.5 dB, and in the VSAT receiver the implementation margin is 0.8 dB. Assume that there are always 20 active VSAT stations receiving data from the outbound link in packet form, using TDM and a single QPSK carrier. Assume linear operation of the transponders, but include the effect of increased sky noise when rain is present on the uplink. Transponder 1 (inbound, SCPC-FDMA) is operated with 2-dB output backoff. Transponder 2 (outbound, TDM) is operated with 1-dB back off. Part 2 Problems a. Determine the clear air overall CN required on the inbound uplink and downlink required for one VSAT transmission to meet the 99.7% availability criterion, and the corresponding clear air CN in the hub station receiver with (i) rain in the inbound uplink (ii) rain in the inbound downlink. Remember to include the effect of increased sky noise. b. Using the results you obtained in Part 1, and Part 2 problem (a), determine the maximum data rate for the VSAT request packets to meet the 99.7% availability criterion with access to the transponder through the satellite’s regional beam, with 20 active VSATs at any time. c. Determine the clear air overall CN in the VSAT station receiver for an outbound data rate of 1 Mbps using QPSK-TDM to meet the 99.9% availability criterion, for (iii) rain in the outbound uplink (iv) rain in the outbound downlink. Remember to include the effect of increased sky noise. d. Using the results you obtained in Part 1, determine the maximum data rate that can be supplied to each VSAT station with 20 active stations in the network at the same time, for the 99.9% availability criterion.

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Note that for the small percentages of time used here, you may assume that rain never occurs simultaneously in both the uplink and downlink. e. If your results from parts (b) and (d) above show that either transponder 1 or 2 is not bandwidth limited, it is possible to optimize the system to transmit at higher bit rates. Redesign the VSAT and hub stations to increase the bit rates in either the inbound link, the outbound link, or both links, within the limits that the VSAT antenna diameter cannot exceed 1 m, and the transmit power cannot exceed 2 W. The hub station antenna diameter cannot exceed 5 m and the transmit power cannot exceed 200 W. You might also consider whether the number of simultaneous users can be increased. The satellite is leased and cannot be changed, except that the gain of the transponders can be adjusted to suit the earth stations used in the network. Part 3

Portable terminals

The one advantage of radio systems over wired communications links is portability. This question asks you to design a portable Ka-band terminal which can be used to connect to the Internet (provided the customer has a clear view of the southern sky). The critical element in a portable communications link is the antenna. A large antenna provides a high data rate, but is cumbersome and must be pointed accurately at Ka-band frequencies. A small antenna is easier to set up, but cannot provide a high data rate. Let’s assume that the dimensions of the antenna are limited to the dimensions of a typical laptop computer—0.25 m  0.2 m—with an aperture efficiency of 25%, and that some method is provided that helps the customer point the antenna beam toward the satellite so that there is no more than 1-dB loss of gain due to antenna mispointing. Because the portable terminals cannot achieve the same CN ratios as the fixed terminals, separate transponders will be needed to service the portables. For convenience, we will call these transponders 3 (inbound) and 4 (outbound) and use the same frequencies as transponders 1 and 2. The ability of the system to operate during rain fades on the outbound link is relaxed with an availability of 99.7% required in each direction. a. Calculate the gain and the beamwidth of the portable antenna at frequencies of 28.2 and 21.7 GHz. b. Using your results from Part 1, find the inbound and outbound overall CN ratios in the hub station

and portable receivers using the conditions in Part 1 in clear air conditions. Don’t forget to allow an extra 1 dB of loss to account for antenna mispointing. c. Assume that 10 active stations share each transponder. Determine the maximum data rates that customers can achieve on the inbound and the outbound links with 99.7% availability of the inbound and outbound links. d. Transponders 3 and 4 can be switched into baseband processing mode. In this mode, the incoming QPSK signal is demodulated to baseband, the data bits are recovered and then remodulated onto a carrier for transmission as a new QPSK signal. This allows the transponder to transmit at its rated output power at all times despite uplink attenuation. The bit error rate for the link is then the sum of the BERs on the uplink and the downlink. Rework your solution to part (c) above using baseband processors for both inbound and outbound links and determine the new data rates for the inbound and outbound links. e. Draw a block diagram of transponder 3 when used in its baseband processing mode. Your block diagram should include all the filters, amplifiers, mixers, oscillators, modulators and demodulators, and all other important blocks. Label each filter and amplifier with a center frequency and bandwidth, and indicate the gain of each amplifier. Label all oscillators with their frequencies. f. Comment on the performance of the fixed and portable Ka-band Internet link system. If the transponders on the GEO satellite cost $1.5 M per year each to lease, and the service provider’s costs to support the customer base that shares these transponders are $ 0.5 M per year, what would you expect to have to charge the customer for access to the Internet when using the fixed terminal and the portable terminal? You can establish a charging structure made up of a monthly fee plus a per minute access charge. Assume that you can achieve a continuous level of activity of 20 fixed or 10 portable terminals for 12 h per day. Each user can be assumed to connect to the Internet for 15 min once each day, but is active (in the sense of data transfer over the satellite) for 1 min per day. How do the data rates and the charges you propose for the portable Internet access service compare to typical charges for cable modem service?

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7

ERROR CONTROL FOR DIGITAL SATELLITE LINKS 7.1

ERROR DETECTION AND CORRECTION The transmission of information over a satellite communication system always results in some degradation in the quality of the information. In analog links the degradation takes the form of a decrease in signal-to-noise ratio. We saw in Chapter 5 that by using wideband FM we can trade bandwidth for power and achieve a good baseband signal-to-noise ratio with a low carrier-to-noise ratio in the RF signal. In digital links we measure degradation of the information content of a signal in terms of the bit error rate. By using phase shift keying, usually coherent QPSK, we can again trade bandwidth for signal power and achieve good bit error rates with low carrier-to-noise ratios. A fundamental difference between analog and digital signals is that we can improve the quality of a digital signal by the use of error correction techniques. No such technique is available for analog signals since once the information is contaminated by noise, it is extremely difficult to remove the noise, as we cannot in general distinguish between the signal and the noise electronically. (There are techniques that attempt to distinguish between signal and noise in television pictures, by using the correlation properties of the picture. They have been used successfully to enhance the quality of images of the moon and other planets obtained by the Voyager and similar space probes. However, the time taken to process the picture and the computer power needed make such techniques impractical for regular TV and voice transmissions.) In a digital system, we can add extra redundant bits to our data stream, which can tell us when an error occurs in the data and can also point to the particular bit or bits that have been corrupted. Systems that can only detect errors use error detection. Systems that can detect and correct errors use forward error correction (FEC). Systems which have only error detection must make a decision about what action to take when an error is detected. The options are to ignore the error, to flag the error, to send a block of information again, or to estimate the error and replace the corrupted data. Which option is selected depends on the nature of the signal that is transmitted. Collectively, these techniques are known as error control. Error control may be implemented at the earth station as a permanent part of the satellite communication link, or it may be applied by the end user. In advanced digital satellite communication links the FEC may be switched in and out on demand, depending on the measured bit error rate or CN ratio at a terminal. Some confusion surrounds the term coding, since it is applied to several different processes, not all of which are concerned with error detection and correction. In the popular sense, coding is used to describe the rearrangement of information to prevent unauthorized use. This process is known technically as encryption. It is widely used on both analog and digital signals that are sent by cable and radio links. Digital signals are much more amenable to encryption, which can be achieved by convolving the data bits with a long bit sequence to destroy the intelligibility of the baseband data. To recover the 273

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information, the bit sequence used in the encryption process must be known to the recipient; this information is contained in the key to the code, which may be changed at frequent intervals to maintain good security. We will not be concerned any further, in this chapter, with encryption. Coding is also a name applied to many processes that change data from one form to another. For example, pulse code modulation (PCM) changes analog data into binary words for transmission over a digital link. It is fundamental to the transmission of voice and video signals by digital techniques, and uses a device commonly called a codec, short for coder–decoder. The term coding is also applied to devices that scramble a digital data stream to prevent the occurrence of long strings of 1s or 0s. Throughout this chapter we shall use the term coding to refer to error detection or error correction. This implies that additional (redundant) bits are added to the data stream to form an error-detecting or error-correcting code. It is possible, in theory, to generate codes that can detect or correct every error in a given data stream. In practice, there is a trade-off between the number of redundant bits added to the information data bits and the rate at which information is sent over the link. The efficiency of a coding scheme is a measure of the number of redundant bits that must be added to detect or correct a given number of errors. In some FEC systems the number of redundant bits is equal to the number of data bits, resulting in a halving of the data rate for a given channel transmission rate. This is called half rate FEC. The loss of communication capacity is traded for a guaranteed low error rate. This technique is widely used in VSAT systems where the links to and from the small antenna terminal have low CN ratios. By adding half rate FEC to the data streams using convolutional coding, the reduction in BER at the baseband output of the receiver is roughly equivalent to a 3-dB improvement in CN ratio. A 3-dB improvement in CN ratio can be obtained by increasing the antenna diameter of the VSAT by 41%, but is an expensive and unwieldy option compared to inserting a half rate FEC integrated circuit into the terminal’s bit stream. Consequently, all satellite terminals which tend to have low CN ratios (VSATs, satellite telephones, DBS-TV terminals) make use of forward error correction to improve the bit error rate on the links to and from the small terminal. The operator of a digital satellite communication link has an option of providing FEC as part of the link, or of providing only a basic transmission channel. At 64 GHz and on many 1411 GHz links, the channel is provided without error correction or detection. A minimum BER is guaranteed by the operator for a specified percentage of time, based on the link design and projected performance. The user is then free to add error detection or FEC to the data sent to and received from the link. If digital speech is sent, error detection or correction is rarely applied. With digital data, some measures must be taken to guard against error, and the user will normally provide the necessary equipment. Financial transactions and records are required to be transmitted with a BER of 1012. Few communications links guarantee such low error rates and a customer sending financial data over a satellite link must use an automatic repeat request (ARQ) protocol to ensure that the probability of an undetected bit error becomes extremely low. Links operating at frequencies above 10 GHz are subject to increases in BER during propagation disturbances. The link will be designed with a margin of a few decibels so that the BER falls below an acceptable level, typically 106, for only a small percentage of any month or year. The total time for which the margin is exceeded by propagation effects will be less than 0.5% of any month in a well-designed system. During the remaining 99.5% of the month, the CN ratio of the received signal will be well above threshold, and very low BER will result. There may, in fact, be no errors for long periods of time and billions of bits can be transmitted with complete accuracy. Under

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these conditions, FEC and error-detection systems do nothing for the communication system. However, unless we can detect a falling CN ratio directly, or an increase in BER, coding may have to be applied all the time to be certain it is available when the CN ratio approaches threshold. To that extent, coding is an insurance against the possibility of bit errors; for most of the time it is unnecessary, but when it is needed, it proves invaluable. Common carriers, who supply communication links to users on a dial-up or leased basis, do not generally apply FEC to their links, nor do they define the protocols to be used. (A protocol defines an operating procedure in a link.) These are user-supplied services and must be defined by the user for the data to be sent. In such cases, the error detection and correction equipment will be located at the customers’ premises, whereas the earth station may be a long distance away and accessed via terrestrial data links. The situation may be very different in a single-user network such as a direct broadcast television system, where the coding and format of the transmitted data are specified by the company that operates the uplink and satellites. A similar situation can arise in carefully controlled systems such as a military communication system where the link operator specifies the user’s earth station and operating parameters in detail.

7.2

CHANNEL CAPACITY In any communication system operating with a noisy channel, there is an upper limit on the information capacity of the channel. Shannon1 examined channel capacity in mathematical terms, and his work led to significant developments in information theory and coding. For an additive white Gaussian noise channel, the capacity H is given by H  B log2 31  P 1N0 B2 4 bps

(7.1)

where B is the channel bandwidth in hertz, P is the received power in watts, and N0 is the single sided noise power spectral density in watts per hertz. Equation (7.1) is commonly known as the Shannon–Hartley law. We can rewrite Eq. (7.1) specifically for a digital communication link by putting H  1Tb where Tb is the bit duration in seconds. The energy per bit is E joules, giving E  PTb  PH

(7.2)

Then substituting EbN0  PHN0 in Eq. (7.1) yields Eb H H  log2 a1  b B N0 B

(7.3)

The ratio HB is the spectral efficiency of the communication link, the ratio of the bit rate to the bandwidth of the channel. Figure 7.1 shows the ratio log2(HB) plotted against EbN0 in decibels for the case when H  B and the link operates at a bit rate H bps. Regardless of the bandwidth used, the EbN0 cannot go below 1.6 dB (ln 2) if we are to operate at capacity. This is known as the Shannon bound. It sets a lower theoretical limit on the EbN0 we can use in any communication link, regardless of the modulation or coding schemes. A link operating with H  B is said to be power limited because it does not use its bandwidth efficiently. Recently developed powerful FEC schemes such as turbo codes2 allow links to operate down to CN ratios of 0 dB at acceptable bit error rates. This is 1.6 dB above the Shannon limit, leaving only a little room for further improvement.

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log2 H B −4

( )

−6

−2

0

(NEb0 ) 0 (dB)

−0.5

−1.0

−1.5 −1.6 dB

FIGURE 7.1 Relationship between HB and Eb N0 for power limited case and low EbN0 ratio.

Figure 7.2 shows the case for a link with H  B, operating at capacity. In this case, we can increase the capacity for a given bandwidth without limit, but only by providing large EbN0 ratios, implying high transmitter power. When H  B, the link is said to be bandwidth limited, implying that we could increase capacity by using the available transmitter power in a wider bandwidth. Practical links using PSK do not achieve capacities close to the Shannon theoretical capacity H. Shannon’s theory assumes essentially zero bit errors; to achieve a bit error rate of 106, a QPSK link requires a theoretical EbN0 of

60

Eb N 0 40 (dB)

20

0

0

2

4

6

log2 H B

( )

FIGURE 7.2 Relationship between HB and EbN0 for bandwidth limited case high EbN0 ratio.

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10.6 dB with a spectral efficiency of 2 bits/Hz. Equation (7.3) predicts a bound of EbN0  1.77 dB for this case. What coding, in particular FEC, can do for us is to improve the link performance under conditions of low CN ratio, such as during periods of rain attenuation, so that the BER of the link does not rise excessively. This takes us closer to the Shannon capacity in the region of low CN ratios while not increasing excessively the bandwidth required for transmission. A theoretical spectral efficiency of 8 bits/Hz can be achieved with EbN0  15.0 dB4. A satellite link could use 256-QAM modulation to achieve this spectral efficiency, but would need CN of 30.3 dB (EbN0  21.3 dB) to give a bit error rate of 106. This is well below the theoretical efficiency for the Shannon bound, even when FEC is applied to the data stream.

7.3

ERROR CONTROL CODING Error detection coding is a technique for adding redundant bits to a data stream in such a way that one or more errors in the data stream can be detected. One redundant bit is added for every N data bits; this allows a single error within that block of N bits to be detected, whatever the number of bits in the block. A simple example of an error detecting code system that has been in use for many years is the single bit parity applied to the 7-bit ASCII code3. The ASCII code is widely used for transmission of data over telephone lines and radio links. The 7-bit ASCII code consists of 128 characters that have internationally agreed interpretation and represent the alphabet in uppercase and lowercase letters, numerals 0 to 9, and a set of useful commands, symbols, and punctuation marks. An eighth bit, the parity bit, is used for detection of error in the 7 data bits of the character. For example, in a system using even parity, the parity bit is 0 when the sum of the data bits is even, and 1 when the sum is odd. Thus the sum of the data bits plus the parity bit is always made even, or 0 in modulo-2 arithmetic. Figure 7.3 shows an example of even and odd parity coding. In odd parity, the sum of the data bits plus the parity bit is always odd, or 1 in modulo-2. Errors in the 7 data bits, or the parity bit, are detected at the receiving end of the link by checking the 8 received bits of each character for conformity with the parity

Data bits

Parity bit

Sum (Modulo-2)

Even parity

0101101

0

0

Odd parity

0101101

1

1

Received codeword

Sum of bits

Error detected?

One error

01010010

1

Yes

Two errors

01010110

0

No

Three errors

11010110

1

Yes

(a)

(b) FIGURE 7.3 (a) Example of even and odd parity for a 7-bit ASCII word. (b) Example of error detection in a 7-bit ASCII word with even parity. Error bits are underlined.

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rule. In modulo-2 arithmetic, 0  0  0, 0   1, 1  0  1, and 1  1  0. This is the exclusive OR function of digital logic. Similarly, 0  0  0, 0  1  1 and 1  0  0, and 1  1  1. All the codes that we will be considering are binary, and modulo-2 arithmetic will be used throughout this chapter. Suppose we have a system using even parity, which transmits the character A in ASCII code, as illustrated in Figure 7.3. At the receiving end of the link we check the 8 bits by modulo-2 addition. If the sum is 0, we conclude that the character is correct. If the sum is 1, we detect an error. Should 2 bits of the character be corrupted, the modulo2 sum is 0, so we cannot detect this condition. We can easily calculate the improvement in error rate (assuming that we discard corrupted characters) that results from adding a parity bit to a 7-bit word. For example, let the probability of a single-bit error occurring in the link be p, and let us suppose that p is not greater than 101. The probability Pe(k) of k bits being in error in a block of n bits is given by the binomial probability function n Pe 1k2  a b pk 11  p2 nk k

(7.4)

where p is the probability of a single-bit error occurring, and n n! a b k k!1n  k2!

(7.5)

For example, with single parity and the 7-bit ASCII characters, we have one parity bit, which allows us to detect one error, and 7 data bits. Two errors cannot be detected, although three can. The most likely error that goes undetected is a 2-bit error. The probability that there are four or more errors in the 8-bit word is much lower than the probability of 2-bit errors, provided the BER is no higher than 102, so when a single parity bit is used, the probability of an undetected error occurring in an ASCII word is approximately Pwc where 8 Pwc  Pe 122  a b p2c 11  pc 2 6 2

(7.6)

where pc, is the single bit error probability for the 8-bit word (i.e., the BER on the link). When pc is small, 11  pc 2 6  1 and 8 Pwc  a b p2c  28 p2c 2

(7.7)

If we had not used parity, the probability Pwu of a single error in the 7-bit word with bit error probability pc is 7 Pwu  a b pu 11  pu 2 7  7 pu 1

(7.8)

Thus the improvement in rate of undetected errors for the ASCII words is approximately 4p, provided pc  pu. EXAMPLE 7.3.1 A data link transmits 7-bit ASCII words at a bit rate of 1 Mbps, with a single parity bit. The probability of a bit error on the link is p  103. Find the probability of an undetected error when uncoded data is transmitted and when a single parity bit is added to each 7-bit word. What is the

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probability of an undetected bit error when the BER on the link is 106? How many undetected character errors would be present if a 500-page textbook were transmitted this link, using single parity? Using the result in Eq. (7.8), the probability of error for uncoded 7-bit words is 7  103. If we add a single parity bit, the probability of an undetected error is given by the result of Eq. (7.7) as Pwc  28p2c  28  106. With a BER of 106, the undetected character error rate is 7  106 for no parity and 28  1012 with single bit parity. Based on 2000 characters per page (including spaces) and 500 pages in the book, there are one million characters in the text. Without error detection, there would be seven typos in the text caused by transmission over the link. With single bit parity error detection, the text could be transmitted 35,700 times before a single undetected character error occurred. At a bit rate of 106 bps, the entire book can be transmitted in 7 s without parity and 8 s with single bit parity. The single undetected error—which will appear in the text as a typo—would occur after 79 h of transmission when single bit parity is used. 

The above example illustrates how powerful even a single parity bit can be in the detection of bit errors in a link with low BER. Some caution is needed in making the assumption that the bit error rate is the same for uncoded and coded transmissions. When we added the single parity bit to a 7-bit ASCII character, the transmission bit rate went up from 7 bits per character to 8 bits per character. The increase in bit rate will result in an increase in BER because the channel bandwidth must be increased and this increases the noise power in the receiver and reduces the CN ratio, so not all the expected decrease in character error rate will be achieved in practice. Example 7.3.1 above for parity error detection is one case of block error detection. We have transmitted our data as blocks, in this case 8-bit blocks consisting of 7 data bits and 1 redundant parity bit. In general, we can transmit n bits in a block, made up of k message bits plus r parity bits. There are two ways in which the blocks of data can be formed. In a packet transmission system (see Chapter 6), the block is usually long, typically 64 to 2048 bits, and checksum or cyclic redundancy check (CRC) bits are added to the block that generate an error condition at the receiver if there are errors in the block. Alternatively, the blocks may be members of a set of n-bit codewords. Coding schemes in which the message bits appear at the beginning of the codeword, followed by the parity bits, are called systematic block codes.

Linear and Cyclic Block Codes Linear block codes are codes in which there are 2n possible codewords containing k message bits and (n  k) redundant check bits. In a systematic linear block code the first k bits of the codeword are the message and the remaining (n  k) bits are the parity bits, or possibly the other way round with the parity bits first. A codeword with n bits of which k bits are message data is written as (n, k). Early work on single error correction linear block codes was done by Hamming at Bell Labs in the 1940s3,5. A subset of linear block codes called binary cyclic codes has been developed for which implementation of errorcorrection logic is relatively easy. The codes can be generated using shift registers, and error detection and correction can also be achieved with shift registers and some additional logic gates. A large number of binary cyclic codes have been found, many of which have been named after the people who first proposed them. The best known are the Bose–Chaudhuri–Hocquenghem codes (BCH codes), which were independently proposed by three workers at about the same time in 1959–19606,7. Other examples of block codes in widespread use are the Reed–Solomon codes, used on compact discs and DBS-TV signals, and the Golay codes. Most of the block codes were developed in the late 1950s for

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error detection and correction with early computer memories, which were prone to cause data errors in the recording and recovery of data. Subsequently, the codes were applied to digital transmission of data. The general form of a linear block codeword C is C  DG

(7.9)

where D is the k-bit message and G is a generator matrix that creates the parity check bits from the data bits. There are 2k valid codewords within a set of 2n codewords generated using Eq. (7.9). For example, if we create a (7, 4) linear block code that has four message bits and three parity bits, there are a total of 27  128 codewords in the set, but only 24  16 codewords are valid. If we receive an invalid codeword we know that an error has occurred on transmission, although we do not necessarily know how many bits have been corrupted, or which of the message bits are wrong. One way to correct errors in received codewords is to compare the received codeword with the valid set of codewords and select the correct codeword that is closest to the received codeword. Some linear block codes are better for error detection or correction than others. There are some basic theorems that define the capabilities of linear block codes in terms of the weight, distance, and minimum distance. The weight, w, of a codeword C is the number of nonzero components of C. The distance, d, between two codewords C1 and C2 is the number of components by which they differ. The minimum distance of a block code is the smallest distance between any pairs of codewords in the entire code. With a single error detection code, the number of errors that can be detected in a code with minimum distance dmin is (dmin  1). The number of errors that can be corrected is (dmin  1)2, rounded to the next lowest integer if the number is fractional. Thus it is always easier to detect errors than to correct them with linear block codes, a feature which is exploited in the Reed–Solomon codes used in CDs and the digital video transmissions of DBS-TV systems. If we are very clever about the generator matrix, G, that we use to generate the code, we may be able to determine which bit is in error when we receive a corrupted codeword. There are no rules for forming generator matrices, so “good” codes are found by inspiration, and a lot of trial and error, using weight, distance, and minimum distance for guidance. A good error detection code is one that detects as many bit errors as possible in a codeword of given length, preferably when the bit errors are sequential (a burst error), and a good error correction code corrects as many errors as possible, even when they are burst errors. Table 7.1 shows some examples of burst error correcting block codes.

Golay Codes The Golay code and the single error-correcting Hamming codes are examples of perfect codes, in which all possible patterns of a given number of errors are corrected8. The Golay code is a (23, 12) cyclic code with a minimum distance of 7 that corrects all patterns of 3 or fewer bit errors. A closely related form of the Golay code has one overall parity check bit added to form a (24, 12) code with a minimum distance of 8. The (24, 12) code will detect all patterns of 7-bit errors and correct all patterns of 3-bit errors. It also has the advantage of a coding rate of one-half; rate one-half coding is easier to implement than other rates because the 2:1 ratio between message bits and code bits simplifies clock synchronization between the input data stream and the output coded stream.

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TABLE 7.1 Examples of Burst Error Correction Block Codes

Code (n, k)

No. of parity check bits

Burst error correcting ability q

Code rate

7, 3 15, 9 511, 499 1023, 1010 131, 119 290, 277 34, 22 169, 155 103, 88 96, 79

4 6 12 13 12 13 12 14 15 17

2 3 4 4 5 5 6 6 7 8

3/7 9/15 499/511 1010/1023 119/131 277/290 22/34 155/169 88/103 38/56

59, 39

20

10

39/59

Source: Reprinted from K. S. Shamnugam, Digital and Analog Communication Systems, John Wiley & Sons, New York, 1979, Table 9.5, p. 475, used with permission.

7.4 PERFORMANCE OF BLOCK ERROR CORRECTION CODES Calculation of the improvement of bit error rate with block encoding requires a comparison of the uncoded error rate to that obtained after correction of blocks of encoded data. For the perfect codes given in Section 7.3, the improvement can be determined exactly; when other codes are used, only an upper and lower bound on the error rate after coding can be established. Figure 7.4 shows a comparison of the performance of a number of cyclic codes when implemented in a coherent PSK link9. The curves are for an ideal link with no modem implementation margin and show error probability as a function of EbN0 at the demodulator input. The (127, 64) code corrects 10 errors, the (1023, 688) code corrects 36 errors. The (127, 113) BCH code is a double-error-correcting code that has been specified for use in 120-Mbps transmission using the Intelsat TDMA system9. Also shown are BER curves for half rate convolutional coding with constraint length 7 and half rate turbo coding with 18 iterations. Note that by the use of FEC, we can reduce the EbN0 needed to achieve a 106 error rate from a theoretical 10.6 dB without coding to 7 dB with BCH (127, 64) coding. The reduction that can be made in EbN0 by adding FEC to the link while maintaining the same BER is called coding gain. Coding gain varies with BER, tending to be greatest at low BER, and sometimes becoming negative when EbN0 becomes small. In Figure 7.4, for example, coding gain for the (127, 64) BCH code is negative when EbN0 is less than 7 dB. When the BER exceeds 2  103, errors are no longer corrected and additional errors are inserted into the bit stream. Convolutional codes perform better at low EbN0 ratios, and are therefore widely used on satellite links. Coding gain must be used with care in satellite links, because to obtain coding gain we must increase the bit rate on the link, which results in a lower EbN0 because the bandwidth

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BCH (1023, 688)

Probability of bit error (BER)

10−2

10−4

Uncoded BPSK

Rate 1/2 convolutional code, soft decision, K = 7

10−6

BCH (127, 64)

1/2

Rate turbo code, 18 iterations 10−8

0

5

10

15

Eb /No in dB FIGURE 7.4 Probability of bit error in a BPSK link with coherent demodulation. Note that these are theoretical curves for EbN0. To apply these results to a practical communication system, an implementation margin must be added to the EbN0 values to obtain CN values. For QPSK links, add 3 dB to the BPSK CN ratios.

of the receiver has to be increased, resulting in lower CN ratios and thus a higher BER. This is particularly true when we apply half rate FEC to a link. If we want to achieve the full coding gain (about 6 dB for many convolutional codes) we can send only half as many bits. If we double the bandwidth of the link in order to send the same data rate, the CN ratio will fall by 3 dB and the effective increase in CN will be only 3 dB instead of 6 dB.

7.5

CONVOLUTIONAL CODES Convolutional codes provide better error correction performance than block codes on satellite links with low CN ratios in the receiver, and have simpler decoding structures. Figure 7.4 shows that the BCH (1023, 688) block code and the constraint length 7 convolutional code have near equal correction capability when the BER is below 106. However, as the BER increases, the convolutional code performs better. Decoding cyclic codes with large block lengths is difficult, and the choice of forward error correction code is often based on the availability of coding and decoding devices as well as performance. Convolutional codes are generated by a tapped shift register and two or more modulo2 adders wired in a feedback network. The name is given because the output is the convolution of the incoming bit stream and the bit sequence that represents the impulse response of the shift register and its feedback network10. Figure 7.5 illustrates an example of a simple convolutional encoder using a three stage shift register10–13. As each

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+

Switch Input

3 stage shift register

0011100111000110

01010101

Bit clock

Output

+

FIGURE 7.5 A simple convolutional encoder. The switch changes state at each input bit clock transition.

incoming information bit propagates through the shift register, it influences several outgoing bits, spreading the information content of each data bit among several adjacent bits. The output data stream at the right of Figure 7.5 is different from the input stream at the left; the data bits are not present in the coder output as in a block code. An error in any one output bit can be overcome at the receiver without any information being lost. The process is somewhat like forming an image from a hologram, where information is distributed more or less uniformly over a two-dimensional field. The image can be reconstructed from only a portion of the field, although resolution is lost if a significant part of the hologram is discarded. The state of a convolution encoder is defined by the shift register contents that will remain after the next input bits are clocked in. If the shift register is K bits long and input bits enter in groups of k, then the encoder has 2Kk states. Putting in a group of k input bits causes the encoder to change states; a change of state is called a state transition. From a given state, a convolutional encoder can go to only 2k other states (although one of these k options may be to remain in the starting state). Each state transition is associated with a unique sequence of input bits and is accompanied by a unique sequence of output bits. The quantity K, which measures the length of the shift register, is called the span or the constraint length of the encoder. If v output bits are transmitted for every k input bits, then the encoding rate is kv. A ratio kv  12 is widely used, giving half rate coding. A decoder for convolutional codes keeps track of the encoder’s state transitions and reconstructs the input bit stream. Transmission errors are detected because they correspond to a sequence of transitions that could not have been transmitted. When an error is detected, the decoder begins to construct and keep track of all the possible tracks (sequences of state transitions) that the encoder might be transmitting. At some point, which depends on its speed and memory, the decoder selects the most probable track and puts out the input bit sequence corresponding to that track. The other tracks that it had been carrying are discarded. One of the best algorithms used for decoding is named for A. J. Viterbi (see reference 14), and for this reason convolutional codes are sometimes called Viterbi codes (Figure 7.6).

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10−2

K=3

K=4 10−3 Bit error rate

K=5 K=6

K=7 10−4

K=8

10−5

4

5

6

7

Eb /No in dB FIGURE 7.6 Performance of rate one-half convolutional codes with constraint lengths from 3 to 8 and 32-bit paths. (Reprinted with permission from Jerrold A. Heller and Irwin Mark Jacobs, “Viterbi Decoding for Satellite and Space Communications,” IEEE Transactions on Communications, COM-19, 835–848 (October 1981). Copyright © 1981 IEEE17).

Convolutional or linear block coding and digital modulation can be combined using trellis coding. Trellis coding employs a high level modulation such as 8-PSK or 16-PSK and allows only certain sequences of modulation states to be transmitted. The receiver can detect errors in the received data strings if an invalid sequence of modulation states is received3,15,16. The advantage of trellis coding is that coding gain can be achieved with a smaller increase in bit rate than with conventional convolution encoding. However, trellis coding does not seem to have been adopted very widely. Recently, turbo codes2 have been developed that allow links to operate down to CN ratios of 2 dB at acceptable bit error rates. The decoding algorithms for turbo codes require powerful DSP integrated circuits, but these are now becoming available.

7.6 IMPLEMENTATION OF ERROR DETECTION ON SATELLITE LINKS Error detection is invariably a user-defined service, forming part of the operating protocol of a communication system in which the earth–satellite–earth segment may only be a part. It allows the user to send and receive data with a greatly reduced probability of error, and

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a very high probability that uncorrected errors are identified and located within a block of data, so that the existence of an error is known even if the exact bit or word in error cannot be determined. The penalty for the user is a reduced transmission rate, just as in FEC. However, the transmission rate is reduced only when errors occur, so there is no bit rate penalty under normal conditions when no errors occur, unlike FEC. Implementation of error correction by use of error detection and retransmission requires the use of protocols. A protocol is an agreed-upon set of actions that define how each end of the data link proceeds so that data are transmitted in an accurate and ordered fashion through the link. Error detection is readily accomplished using the coding techniques described in the previous sections. In many communications systems it is not sufficient simply to detect an error; it must be corrected. However, some systems such as speech and picture transmission may simply count errors to determine whether the link is above an operating threshold. Excessive error counts may cause the link to automatically shut down. When a succession of pictures is transmitted, for example, weather maps, the previous value of a pixel can be used when a new value is in error. When an error-detection code or CRC is used and the error must be corrected, a retransmission of the block of data containing the error must be made so that the correct data are acquired at the receiving terminal. If an error is detected in the block, a not acknowledge (NAK) signal is sent back to the transmit end of the link, which triggers a retransmission of the erroneous block of data. This is called an automatic repeat request (ARQ) system. In terrestrial data communication systems, it is common practice for the receiving terminal to send an acknowledge (ACK) signal to the transmitting end whenever it receives an error-free block of data. Such protocols works well on terrestrial data links with relatively low data rates and short time delays, but the long transmission delays in satellite communication systems make it highly undesirable to send ACK signals for every error free block that is received, so Internet access via satellites must use a different access protocol. Often a satellite terminal receives data using a terrestrial protocol and generates the acknowledgements, then transmits the data over the satellite link using a different protocol. This is called spoofing. There are three basic techniques for retransmission requests, depending on the type of link used. In a one-way, simplex link, the ACK or NAK signal must travel on the same path as the data, so the transmitter must stop after each block and wait for the receiver to send back an NAK or ACK before it retransmits the last data block or sends the next one. With a one-way delay of 240 ms on a GEO satellite link, the data rate of such a system will be very low and is suitable only for links in which data are generated slowly, as when someone is typing on a terminal keyboard. Satellite links usually establish two-way communication (duplex channel) by the use of SCPC-FDM, or TDM, as discussed in Chapter 6. The ACK and NAK signals can be sent on the return channel while data are sent on the go channel, However, if the data rate is high, the acknowledgment will arrive long after the block to which it relates was transmitted. In a stop-and-wait system, the transmitting end sends a block of data and waits for the acknowledgment to arrive on the return channel. The delay is the same as in the simplex case, but implementation is simpler. Figure 7.7 shows an example of a stopand-wait sequence. In a continuous transmission system using the go-back-N technique, data are sent in blocks continuously and held in a buffer at the receive end of the link. Each data block is checked for errors as it arrives, and the appropriate ACK or NAK is sent back to the transmitting end, with the block number appended. When a NAK(N) is received, the transmit end goes back to block N and retransmits all subsequent blocks, as illustrated in Figure 7.8. This requires the transmitter on a satellite link to hold at least 480 ms of data, to allow time for the data to reach the receive end and be checked for

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Retransmitted block

Transmitted blocks 1

2

2

3

NA

AC K

K

Time

AC K

286

1

2

ACK

2

3 Time Received blocks

Transmission delay, t d FIGURE 7.7

Errors detected in this block

Stop-and-wait ARQ system.

errors and for the acknowledgment to be sent back to the transmit end. Since there is a delay in transmission only when a NAK is received, the throughput on this system is much greater than with the stop-and-wait method. Throughput is the ratio of the number of bits sent in a given time to the number that could theoretically be sent over an ideal link. If sufficient buffering is provided at both ends of the link, only the corrupted block need be retransmitted. This system is called selective repeat ARQ. In Figure 7.8, block 2 is corrupted, and blocks 4, 5, 6, and 7 are transmitted before the NAK message is received. At this point, we could transmit block 3 only if blocks 4, 5, 6, and 7 are stored at the receive end. On receiving a correct version of block 3, the receive buffer substitutes it for the corrupted version and releases the data for retransmission. In systems handling data rates of megabits per second, the buffer requirements for continuous transmission systems become quite large because of the delay on the link via a geostationary satellite. The Internet protocol TCP/IP (transmission control protocol/internet protocol) uses selective repeat of blocks that contain errors. The relatively short delays on terrestrial paths used for Internet traffic allow the use of selective repeat ARQ. The TCP/IP protocol cannot be used over a geostationary link without modification because it was designed for terrestrial links with short delays. The protocol times out before repeat transmissions and replies are received when a GEO satellite link is used. Transmitted blocks 1

2

Retransmitted blocks

3

4

5

2

3

4 Time

ACK

NAK

NAK

NAK

NAK ACK

0

1

2

3

4

5

2 Time

td Transmission delay FIGURE 7.8 example.

Received blocks Errors detected in this block

Example of a go-back-N-blocks ARQ system. N is three blocks in this

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Reference 19 contains a good survey of error-detection techniques for use in satellite communication systems and of the various ARQ systems that can be implemented. Some hybrid ARQ systems are described that combine FEC with retransmission of blocks when uncorrected errors are detected. This combines the error-correction properties of the FEC code for a limited number of errors with the error-detection properties of the same code when too many errors are present for all of them to be corrected20. EXAMPLE 7.6.1 Calculate the frequency of retransmission, throughput, and buffer requirements of a satellite link capable of carrying data at rates of (a) 24 kbps and (b) 1 Mbps when a block length of 127 bits is used and the one-way path delay is 240 ms, for a bit error rate of 104 and a double error detecting code (127, 120), using the following ARQ schemes: 1. Stop and wait. 2. Continuous transmission with transmit buffer only (go-back-N ). 3. Continuous transmission with buffers at both ends of the link (selective repeat). Comment on the three ARQ schemes and their suitability for a satellite communication link. For a 127-bit code block, the probability of one or two errors is given by Eq. (7.4) n Pe 1k2  a b pk 11  p2 nk k where k  2, n  127, and p is the probability of a single bit error, which is 104 in this example. The probability of one or two errors being detected in a block of 127 bits is P 1one or two errors2  127  0.9875  104  127  1262  0.9876  108  0.01262 Thus, on average, 1 in every 79 received blocks has a detectable error. 1. Stop-and-wait. We send 127 bits and wait for an acknowledgement, which takes 0.485 s at 24 kbps and 0.4801 s at 1 Mbps. We therefore send only about two blocks each second at either data rate, since the time is dominated by waiting for an acknowledgement. After 79 blocks have been sent, on average, we detect an error; that is after about 39.5 s. The average data rate on the link is approximately 254 bps for both transmission bit rates. 2. Go-back-N. a. The time required to send 79 blocks of 127 bits at 24 kbps is 0.418 s. The 79th block has a detected error: while the NAK signal goes back to the send end to request a retransmission of block 79, a further 91 blocks arrive. These are discarded when the retransmission arrives 0.48 s later, starting at block 79. This slows the throughput by about 54%, to about 11.2 kbps. The buffer at the transmit end must hold 0.48 s worth of bits, about 11,600 bits. b. At a bit rate of 1 Mbps, the time to send 79 blocks is 0.01 s. We then wait 0.48 s for the retransmission. The average data rate is then 20.46 kbps. The transmit buffer must hold 490 kbits of data. 3. Selective repeat system. a. The only time lost in a selective repeat system is in the retransmission of blocks which have errors. On average, one block in every 79 has to be retransmitted, so the rate efficiency of the system is 7879, or 98.73%. At 24 kbps, the average data rate is 23.70 kbps, and the buffer needed is 11,600 bits. b. At 1 Mbps, the average data rate is 987.3 kbps, and the buffer must hold 490 kbit.



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7.7 CONCATENATED CODING AND INTERLEAVING Sophisticated error correction and detection systems are used on some satellite links to overcome burst errors and the effects of low CN ratios. Burst errors can occur when the signal is temporarily blocked, or the CN ratio has become unusually low and the BER is very high. Convolutional codes are limited in the number of sequential bit errors that can be corrected, so long strings of incorrect bits will not be corrected by the FEC decoder in the receiver. Burst errors can be overcome by using interleaving. Interleaving is usually applied ahead of error control coding where a single coding operation is used. When two separate error control coding operations are placed in series, the process is called concatenation. Two coding operations can be applied in sequence to a data stream to decrease the probability of undetected errors occurring because of low CN ratio on a satellite link. Concatenated coding can achieve coding gains up to 9 dB3. With concatenated coding, the interleaver may be placed between the two stages of error control coding. This is the strategy used in audio compact disc recordings and also in direct broadcast satellite TV signals using the Digital Video Broadcast Standard for satellite systems (DVB-S)18. The purpose of interleaving is to spread out the errors that occurred in a burst and thus to make it easier for an error correction system to recover the original data. Figure 7.9 shows a simple interleaver with 5 rows and 5 columns. Bits are read in to the rows and read out by columns, which spreads out the bits in the resulting bit stream. Interleaving can be illustrated more easily using letters of the alphabet rather than binary data. Consider the following message: The cat sat on the table. Let us suppose that we transmit the message with single bit error detection coding, and that two and three bit burst errors occurs so that this message, without interleaving, is received as: The *** sat ** the *able. where the * indicates an error. Because the English language is highly redundant, and only certain letter combination make valid English words, we can guess at the message based on what we received, although we might guess dog or cup instead of cat, or cable instead of table. Now consider what happens when we use the 5  5 interleaver shown in Figure 7.9 at the transmitter. The message is read into the interleaver by row, as shown. The message is read out by column to give: Tattaht hbe oel sn eca t. When this message is sent over the satellite

T

h

a

t

t

e s o

t

h

e

a

b

l (a )

c

T

*

a

a

*

n

e

t

e

c s

*

t

t

h

*

.

*

b

l

a

n t

*

.

(b )

FIGURE 7.9 (a) A 5  5 interleaver at a transmitter. Letters are used for illustration purposes. (b) A 5  5 de-interleaver at the receiver. Letters are used for illustration purposes.

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link, it is received with bit errors in the same positions as previously. We receive the following message: Tatt*** hbe **l sn *ca t. The message is read into the de-interleaver at the receiver by column, and read out by row. The message at the receiver is then: T*e ca* sat *n th* t*bl*. We could correct the message with a single character error correction FEC code, or we could make an improved guess at the true message based on the received message which now has no burst errors. Since there are no burst errors here, a single (character) error correction code would recover the original message correctly. The high level of redundancy in the English language ensures that the message: T*e ca* sat *n th* t*bl*. can be read with little error, although we cannot be certain whether ca* is cat, cam, can, cap, or car. However, the rest of the message is unambiguous after de-interleaving. There is no such redundancy in a stream of bits. Ones and zeroes are equally likely; indeed the bit stream is usually scrambled to make certain that this is the case, so we cannot guess at the correct bit when an error occurs. Interleavers used in digital communication systems work on the same principle as the simple example shown in Figure 7.9 (but using binary data) to break up burst errors and make possible correction of the received data. A key element in the digital transmission of analog data is that analog interpolation can be used to reconstruct an analog waveform containing single word errors. In the example above using letters, interpolation between the correctly received characters is much easier when the errors occur singly. We can make a much better guess at the message received after interleaving than when there is no interleaving. The same principle holds true for binary transmission of analog data. Interpolation of an analog waveform is simple when only one point (a digital word) is missing, as illustrated in Figure 7.10. This approach is used very successfully on audio compact discs and in digital video broadcast transmissions. Compact discs store audio data bits as changes in the reflectivity of the disc. The bits are about a micron long, so a scratch or a speck of dust on the surface of the CD will cause a lengthy burst error. A long interleaver is used to spread out the burst error and this makes reconstruction of the analog waveform possible for burst errors as long as several thousand bits. The interleavers used in satellite transmission of digital video signals are generally shorter than those used on a CD, typically 12  7. Figure 7.11 shows the coding and decoding structure of a typical DBS-TV signal. In this system, the digital signal is first coded using a 188204 rate Reed–Solomon code, then interleaved with a 12  7 interleaver, and finally encoded with a rate three-quarters convolutional code with constraint length 7. At the receiver, the convolutional code is used to correct some errors, which improves the BER, but when the CN ratio in the receiver is low not all the errors will be corrected. The signal is de-interleaved, which spreads out any uncorrected errors so that the probability of a burst error longer than a couple of bits becomes small. The Reed–Solomon

Vr (t ) Straight line interpolation

Errors

X

X

X = error detected

X

X

t

FIGURE 7.10 Illustration of interpolation to fill in missing data in an analog waveform.

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Input

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Reed–Solomon encoder

Interleaver

Convolutional encoder

Link

Convolutional decoder

De-interleaver

Reed–Solomon decoder

Output

FIGURE 7.11 Coding and decoding structure for data in a DBS-TV system.

code is then used to detect the location of the remaining errors and to flag video words which have errors. The video decoder looks at the digital words on each side of a word that has been flagged with an error and replaces the flagged word with a new word that is found by interpolation between the two neighboring words, as illustrated in Figure 7.10. The resulting error in the analog video waveform is small. Thus the Reed–Solomon coding, which has a high code rate and uses only error detection properties allows errors in the video signal to be corrected. Overall, the combined coding of the DVB signal increases the bit rate by 45% when rate three-quarter convolutional encoding and 188204 Reed–Solomon coding is employed. The coding gain is typically 5.5 dB with BER  106 at the output of the convolutional decoder, and very few errors are left in the analog waveform after Reed–Solomon decoding and analog interpolation. One of the features of the DBS-TV systems is that the software in the customers’ receivers can be modified via the satellite. Packets that are tagged as software can be sent to all receivers, and changes made to the signal processing. This allows the system to change the coding and decoding methods to implement improved error control strategies. One DBS-TV system in the United States changed from rate two-thirds to rate three-quarters convolutional encoding, freeing up additional bits for program material and other uses.

7.8

TURBO CODES Turbo codes are the most powerful available codes for forward error correction (FEC). The turbo code was first proposed by C. Berrou and R. Pyndiah from the Ecole Nationale Superieure des Telecommunications de Bretagne (ENST), in France21,23. Turbo codes have been demonstrated that can achieve a BER of 106 with a received signal at EbN0 of 0.7 dB, an improvement in coding gain of 1 dB over the most powerful concatenated and interleaved convolutional codes22,24. The basic form of turbo code generator uses two component codes, separated by an interleaver. Message bits are read into an interleaver by row and then simultaneously read out by rows and by columns into two separate encoders that use either block coding or convolutional coding. One encoder is driven by the row message bits and the other by the column message bits from the interleaver, so that an entirely different bit sequence is applied to each encoder, but both encoders are sending the same message bits. Since the row output of the interleaver is the original data stream, one encoder has an input which is the original message bit sequence and the other encoder input is an interleaved version of the message bits. The outputs of the two encoders are combined by multiplying the 2-bit sequences (modulo two). Alternatively, the two outputs can be added and sent sequentially. Turbo codes based on convolutional codes are usually known as CTC (convolutional turbo codes) and those based on block codes as BTC (block turbo codes).

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At the receiver, the incoming symbol stream is sampled to create a soft input to the two decoders. A soft decoder creates a digital word from each sample using an ADC so that information about the magnitude as well as the state of the received symbol is retained. Recovered bits are given a weighting in the decoding process according to the confidence level from the sampling process. For example, in a BPSK receiver, we expect to recover symbols with sample magnitude V or V volts. We have greater confidence that a symbol with a magnitude close to the expected value of V or V volts is correct than for a received symbol with a magnitude close to zero volts because we can guess that a significant amount of noise has been added to a received symbol with a value close to zero volts. The symbol with a value close to zero volts is much more likely to be in error. The magnitude of the received symbols is retained through the decoding process, which is one of the strengths in turbo codes. The process is known as soft input soft output (SISO) decoding. The soft input symbol stream is input to an interleaver of the same size as the interleaver used in the transmitter and then read out by row and by column into two SISO decoders corresponding to the two encoders at the transmitter. The outputs of the decoders are two versions of the original message data, one of which was interleaved and the other direct. The outputs from the decoders are compared, using the soft output values to apply confidence levels to the decoded bits. The decoding process is then repeated to obtain a better estimate of the original transmitted data. The characteristics of the encoding schemes and the decoding methods is such that repeated processing of the interleaver output through the decoders reduces the number of errors remaining in the recovered data, thus improving the BER. The disadvantage of this approach is that many iterations in the decoders creates latency in the bit stream and the soft decoder must run at a clock rate many times higher than the bit rate of the message data. The overall performance of turbo codes can be improved by adding an outer layer of Reed–Solomon encoding, just as with concatenated convolutional encoding. Much research has been carried out to optimize the two codes used by a turbo encoder and the soft decoding process at the receiver. The aim is to achieve the highest coding gain with the smallest latency and the least complex decoder. With lower speed bit streams such as compressed digital speech, it is possible to use fast DSP integrated circuits to perform the decoding process, making turbo coding attractive for cellular telephone systems. The additional coding gain achieved by turbo codes and the possibility of operating at CN ratios as low as 0 dB makes turbo coding a good choice for a fading radio channel. With higher bit rates, field programmable gate arrays (FPGAs) are needed to perform parallel processing. Commercial turbo code products in the form of coder and decoder boards became available in the late 1990s, and are now available as single chip codecs. The Jet Propulsion Lab (JPL) in the United States has developed turbo codes for use on its deep space research spacecraft24. The signals from spacecraft at interplanetary distances are very weak, so powerful error correction coding at very low received CN ratios is essential. The JPL turbo code system uses a 16-bit block code and sends an uncoded version of the message bits as well as the usual two encoded versions, and iterates in the decoder using the three received versions of the message. JPL reports that the turbo code implemented for the Voyager spacecraft uses a 16,384 bit interleaver and 10 iterations in the decoder giving a half rate code with a BER of 105 at a received SN ratio of 0.7 dB. Even more powerful codes are planned for the Cassini spacecraft with BER of 106 at an input CN of 0 dB using a rate 16 code. Given Shannon’s limit of 1.6 dB for error free recovery of data, turbo codes are approaching this theoretical performance limit. Reference 24 contains a listing of web sites worldwide with information about turbo code development, and a list of commercially available turbo code products.

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7.9 SUMMARY The transmission of data over a satellite communication link is likely to result in some errors occurring in the received data, for at least a small percentage of time, because of noise added by the transmission system. Many links guarantee only 106 bit error rate and may not achieve this accuracy during periods of rain or other propagation disturbances. Bit errors contribute to the baseband (SN) ratio when digital speech is sent, but it is rarely necessary to correct bit errors in speech; the listener can make such corrections because there is a lot of redundancy in speech and for error rates up to 104 speech remains intelligible. When data are sent over a link, the receiving terminal does not know in advance what form the data take and can only detect or correct errors if extra, redundant bits are added to the transmitted data. Coding of data provides a means of detecting errors at the receiving terminal. Error detecting codes allow the presence of one or more errors in a block of data bits to be detected. Error correcting codes allow the receiving terminal equipment to locate and correct a limited number of errors in a block of data. When error detection is employed, a retransmission scheme is often needed so that the data block can be sent again when it is found to be in error. Retransmission schemes use ARQ (automatic repeat request) techniques and are easiest to apply in packet switched data networks, where data are not transmitted in real time. The long round-trip delay (480

ms) in a satellite link makes simple stop-and-wait systems unattractive for real-time data transmission. Throughput can be increased by providing data storage at both ends of the link and using continuous transmission in which corrupted data blocks are retransmitted by interleaving them with subsequent data block transmissions. Forward error correction (FEC) provides a means of both detecting and correcting errors at the receiving terminal without retransmission of data. FEC codes add redundant parity check bits to the data bits in a way that allows errors to be located within a codeword. In general, twice as many errors can be detected by a FEC code as can be corrected. FEC has the advantage over error detection that a single unit at each end of the link (a codec) can insert and remove the FEC code and make corrections as required. FEC is used on all satellite links where the CN ratio at the receiver is likely to be low. This includes satellite telephones, VSAT terminals, and direct broadcast satellite television. All of these systems make use of the coding gain from rate one-half or rate three-quarters FEC to achieve low BER. Interference tends to cause burst errors in which many sequential bits are corrupted. Special burst-error correction codes are available with the capability of correcting errors in a number of adjacent bits. Scrambling and interleaving of data bits are other ways in which the effect of burst errors can be reduced.

REFERENCES 1. C. E. SHANNON, “A Mathematic Theory of Communications,” Bell System Technical Journal, Part 1, 379–423; Part 11, 623–656 (1948). 2. G. DRURY, M. GARIK, and K. PICHAVANO, Turbo Codes: Principles and Applications, Kluwer, Dordrecht, June 2000. 3. L. W. COUCH, Digital and Analog Communication Systems, Prentice-Hall, Englewood Cliffs, NJ, 5th Ed., 1993. 4. K. S. SHAMNUGAM, Digital and Analog Communication Systems, John Wiley & Sons, New York, 1979. 5. S. LIN and D. J. COSTELLO, Jr., Error Control Coding: Fundamentals and Applications, Prentice-Hall, Englewood Cliffs, NJ, 1983. 6. A. HOCQUENGHEM, “Codes Corecteurs d’Erreurs,” Chiffres, 2, 147–156 (1959). 7. R. C. BOSE and D. K. RAY-CHAUDHURI, “On a Class of Error Correcting Binary Group Codes,” Information Control, 3, 68–79, March 1960. 8. W. W. PETERSON and E. J. WELDON, Jr., Error Correcting Codes, 2nd Ed., MIT Press, Cambridge, MA, 1970.

9. K. FEHER, Digital Communications: Satellite Earth Station Engineering, Prentice-Hall, Englewood Cliffs, NJ, 1983. 10. T. MARATANI, H. SAITHOH, K. KOGA, Y. MIZUNO, and Y. J. S. SNYDER, “Application of FEC Coding to the Intelsat TDMA Systems,” Proceedings of the 4th International Conference on Digital Satellite Communications, Montreal, October 1978. 11. MARLIN P. RISTENBATT, “Alternatives in Digital Communication,” Proceedings of the IEEE, 61, 703–721, June 1973. (Reprinted in reference 11, pp. 212–230.) 12. HARRY L. VANTREES, ed., Satellite Communications, IEEE Press, New York, 1979. 13. J. J. SPILKER, Jr., Digital Communications by Satellite, Prentice-Hall, Englewood Cliffs, NJ, 1977. 14. G. DAVID FORNEY, Jr., “The Viterbi Algorithm,” Proceedings of the IEEE, 61, 268–278, March 1973. (Reprinted in reference 11, pp. 286–296.) 15. G. UNGERBOECK, “Trellis-Coded Modulation with Redundant Signal Sets,” Parts I and II, IEEE Communications Magazine, 25, 5–21, February 1987.

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16. S. LIN, T. KASAMI, T. FUJIWARA, and M. FOSSORIER, Trellises and Trellis Based Decoding Algorithms for Linear Block Codes, Kluwer, Dordrecht, April 1998. 17. JERROLD A. HELLER and IRWIN MARK JACOBS, “Viterbi Decoding for Satellite and Space Communications,” IEEE Transactions on Communications, COM-19, 835–848, October 1971. (Reprinted in reference 11, pp. 273–286.) 18. G. DRURY, G. MARHAVIAN, and K. PICHAVANO, Coding and Modulation for Digital TV, Kluwer, Dordrecht, November 2000. 19. S. LIN, D. J. COSTELLO, Jr., and M. J. MILLER, “Automatic-Repeat-Request Error Control Schemes,” IEEE Communications Magazine, 20, 5–7, December 1984.

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20. S. LIN and P. S. YU, “A Hybrid ARQ System with Parity Retransmission for Error Control of Satellite Channels,” IEEE Transactions on Communications, COM30, 689–694, July 1982. 21. C. BERROU, A. GLAVIEUX, and P. THITIMAJSHIMA, “Near Shannon Limit Error-Correcting Coding and Decoding: Turbo Codes,” ICC, pp. 1064–1070, 1993. 22. SERGIO BENEDETTO and GUIDO MONTORSI, “Performance of Continuous and Blockwise Decoded Turbo Codes,” Communications Letters, Vol. 1, No. 3, May 1997, pp. 77–79. 23. R. PYNDIAH, “Near Optimum Decoding of Product Codes: Block Turbo Codes,” IEEE Transactions on Communications, August 1998. 24. http://www331. jpl.nasa.gov/public/JPLtcodes.html

PROBLEMS 1. Alphanumeric characters are transmitted as 7-bit ASCII words, with a single parity bit added, over a link with a transmission rate of 9.6 kbps. a. How many characters are transmitted each second? b. If a typical page of text contains 500 words with an average of five characters per word and a space between words, how long does it take to transmit a page? c. If the bit error rate on the link is 105, how many characters per page are detected as having errors? How many undetected errors are there? d. On average how many pages can be transmitted before (i) a detected error occurs or (ii) an undetected error occurs? e. If the BER increases to 103, how many detected and undetected errors are there in a page of the text? 2. A (6, 3) block code has a minimum distance of two. a. How many errors can be detected in a codeword? b. How many errors can be corrected in a codeword? 3. A QPSK data link carries a bit stream at 1.544 Mbps and has an overall (CN)0 ratio of 16 dB in the receiver at the VSAT earth station in clear air. The QPSK demodulator at the VSAT station has an implementation margin of 1.0 dB. For 0.1% of the year rain attenuation causes 3.0 dB reduction in the receiver (CN)0 ratio. For 0.01% of the year rain attenuation causes 6.0 dB reduction in the receiver (CN)0 ratio. a. Calculate the BER in clear air, and the BER exceeded for 0.1% and 0.01% of the year. b. Repeat the calculation when data are transmitted using half rate FEC with a coding gain of 5.5 dB (at all BERs). The bit rate on the link remains at 1.544

Mbps when coding is added. What is the data rate with half rate FEC applied to the data? c. Repeat the calculation of part (b) above when the bit rate on the link is increased to 3.088 Mbps with no increase in transmitter power. 4. The analysis of a 56-kbps data link shows that it suffers burst errors that corrupt several adjacent bits. The statistics for burst errors on this link are given in Table P.4. a. Using Table P.4, select a burst error correcting code that will reduce the probability of an uncorrected burst error below 1010. b. Calculate the data rate for messages sent over the link using the code you selected. c. Estimate the average bit error rate for the coded transmission. 5. A satellite link carries packet data at a rate of 256 kbps. The data are sent in 255-bit blocks using TABLE P.4 Statistics for Burst Errors on a Link in Problem 4 No. adjacent bits Corrupted

Probability of occurrence

2 3 4 5 6 7 8 9 10

4  102 2  103 3  104 1  106 2  109 5  1011 1  1012 3  1014 2  1017

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a (255, 247) code that can detect three errors. The probability of a single bit error, p, varies from 106 under good conditions to 103 under poor conditions. The one-way link delay is 250 ms. a. If no error detection is used, what is the message data rate for the link? b. For a link BER of 106, find the probability of detecting an error in a block of 255 bits when error detection is applied. Hence find how often an error is detected. c. Estimate the probability that a block of 255 bits contains an undetected error when the link BER is 103 and error detection is applied. d. Find the message data throughput when the link BER is 106 and a stop-and-wait ARQ system is used, assuming one retransmission always corrects the block.

6. Repeat Problem 5 using a block length of 1024 bits and a (1024, 923) code that can detect 22 errors in a block. The (1024, 923) code can correct 10 errors. Find the average number of blocks that can be transmitted before an uncorrected error occurs when the BER is 103. Repeat the analysis for a BER of 102. Note: The probability of an unlikely event (11 or more errors in a block of 1024 data bits with Pb  103 in this case) can be calculated from the Poisson distribution more easily than from the binomial distribution. The Poisson distribution is given by

P1x  k2 

lkel k!

where   NPb. N is the block length, and k is the number of bits in error.

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8

PROPAGATION EFFECTS AND THEIR IMPACT ON SATELLITE–EARTH LINKS Communications system design requires the development of a link budget between the transmitter and the receiver that provides an adequate signal level at the receiver’s demodulator to achieve the required level of performance and availability. The performance of a link is usually defined for time percentages in excess of 99% over periods of at least a month and is, for digital systems, determined by the bit error rate (BER) that provides the minimum level of service. For analog systems, the CN at the demodulator input that provides the minimum signal quality required defines the performance level for that link. The availability of a link is usually defined for low outage time percentages (typically between 0.04 and 0.5% of a year, or between 0.2 and 2.5% of the worst month, for satellite systems) and is, for digital systems, specified by the BER at which an outage is declared for the link. For analog systems, the CN at the demodulator input at which no usable signal can be demodulated defines the limit of availability. Figure 8.1 illustrates the concept of performance and availability for a digital system with BER as the determinant. The link budget was covered in Chapter 4, as was link margin: the difference in power level between clear sky conditions (essentially the performance level) and that which exists at the threshold of the demodulator when the link is under impaired conditions (the availability level). Actually, there are two margins to consider in a link budget: (1) the margin between the “clear sky” level and the performance threshold; and (2) the margin between the performance threshold and the availability threshold. Figure 8.2 illustrates these two concepts of margin for a typical digital Kuband downlink (11 GHz) located in the Mid-Atlantic region of the United States. As can be seen, the attenuation experienced on the link varies with time percentage, gradually falling through the performance threshold and then the availability threshold. It is the link designer’s task to ensure that loss of signal occurs for no longer than the time permitted for that service. The development of an accurate link budget, which includes losses due to the passage of the signal through the atmosphere, is therefore critical. The key equation in the development of the link power budget in Chapter 4 was Eq. (4.11), repeated here in modified form as Eq. (8.1). Pr  EIRP  Gr  Lp  La dBW

(8.1)

The complete volume(s) of the ITU material can be obtained from International Telecommunications Union Sales and Marketing Division Places des Nation—CH-1211 Geneva 20 Switzerland Email: [email protected] http://www.itu.int/publications

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10−10

Performance threshold

10−8

BER

Degraded performance region Availability threshold

10−6

10−4

10−2 0.001

0.01

0.1 1.0 10.0 Time percentage BER level measured

100%

FIGURE 8.1 Schematic of the bit error rate (BER) statistic for a typical communications link. A link is normally designed to provide a given performance specification for a very high percentage of the time. In this example, a BER of 108 is the performance required for 99.9% of the time. The time period over which the statistics are taken is usually a year or a month. Atmospheric constituents (gases, clouds, rain, etc.) will cause the BER in clear sky conditions to degrade. At some point, the BER will reach the level at which an outage is declared. This point defines the availability specification. In this example, a BER of 106 is the availability threshold and it must be met, in this example, for a minimum of 0.01% of the time. 10

Availability threshold

Attenuation, dB

8 Degraded performance region

Performance threshold

6

4

2 0.001

0.01

0.1

1.0

10.0

100%

Time percentage attenuation level measured FIGURE 8.2 Schematic of the loss statistics encountered by a signal on transmission through the atmosphere for a typical Ku-band communications link. In most communications links, an allowance in power margin is built into the link so that the received signal is above the threshold for satisfactory demodulation and decoding. This power margin is commonly referred to as the fade margin since the signal, on occasion, appears to fade below the level established in clear sky conditions. In the schematic above, the link experiences an equivalent fade of about 6 dB before it reaches the performance threshold level established for the link (see Figure 8.1). A further fade of 2 dB, making a total reduction in signal level of 8 dB, takes the link below the availability level established for the link (see Figure 8.1). The relationship between power level, fade margin, and BER, will depend on the modulation used. It will also depend on the amount of channel coding used. In the example above, no inner (FEC) or outer (Reed–Solomon, interleaved) coding has been assumed for the link and the modulation is QPSK. For most heavily coded links, the difference between good performance and an outage (a change on the order of 2 to 3 decades of BER) will occur for a change in signal level of less than 1 dB.

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This equation indicates how the received power, Pr, in dBW depends on the transmitter EIRP (the Effective Isotropic Radiated Power, which is a combination of the output amplifier power, the gain of the transmitting antenna, and the losses associated with that antenna system), the receiving antenna gain, Gr, (which includes, in this case, all losses associated with the receiving antenna), the path loss, Lp, (given by 20 log10 [4R], with  being the wavelength of the signal and R the distance between the transmitting antenna and the receiving antenna) and the attenuation contribution due to the atmosphere, La. Of the terms on the right hand side of equation (8.1), the only one that is not essentially constant with time is the atmospheric loss, La. The component La, usually referred to as propagation loss, determines the margin required by the communications link to meet both the performance and availability specifications.

8.1

INTRODUCTION There are many phenomena that lead to signal loss on transmission through the earth’s atmosphere. These include: Atmospheric Absorption (gaseous effects); Cloud Attenuation (aerosol and ice particle effects); Tropospheric Scintillation (refractive effects); Faraday Rotation (an ionospheric effect); Ionospheric Scintillation (a second ionospheric effect); Rain Attenuation; and Rain and Ice Crystal Depolarization. Rain attenuation is by far the most important of these losses for frequencies above 10 GHz, because it can cause the largest attenuation and is usually, therefore, the limiting factor in Ku and Ka band satellite link design. Raindrops absorb and scatter electromagnetic waves. In Ku and Ka bands, rain attenuation is almost entirely caused by absorption. At Ka band, there is a small contribution from scattering by large raindrops. The various propagation loss mechanisms are illustrated in Fig. 8.3. We will discuss each of these loss mechanisms briefly; for a detailed treatment the reader should refer to references 1 and 2. Figures 8.1 and 8.2 introduced the concept of a time varying BER (or excess link attenuation). Fig. 8.3 indicates where each of the loss mechanisms can be found along the slant path to the satellite. It is also very useful to develop an appreciation for the various time percentages over which each of the propagation loss mechanisms is significant. Figure 8.4 illustrates this schematically, using the same curves from Fig. 8.1. Signal loss—i.e. attenuation—affects all radio systems; those that employ orthogonal polarizations to transmit two different channels on a common, or partially overlapping, frequency band may also experience degradations caused by depolarization. This is the conversion of energy from the wanted (i.e., the co-polarized) channel into the unwanted (i.e., the cross-polarized) channel. Under ideal conditions, depolarization will not occur. When depolarization does occur, it can cause co-channel interference and cross-talk between dual-polarized satellite links. Rain is a primary cause of depolarization. Both attenuation and depolarization come from interactions between the propagating electromagnetic waves and whatever is in the atmosphere at the time. The atmospheric constituents may include free electrons, ions, neutral atoms, molecules, and hydrometeors (an arcane term that conveniently describes any falling particle in the atmosphere that contains water: raindrops, snowflakes, sleet, hail, ice-crystals, graupel, etc.); many of these come in a wide variety of sizes. Their interaction with radio waves depends strongly on frequency, and effects that dominate 30 GHz propagation, for example, may be negligible at 4 GHz. The converse is also true. With one major exception (ionospheric effects) almost all propagation effects become more severe as the frequency increases.

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Sun Satellite Ionosphere

Rain cloud

Depolarization Attenuation

Gases Sky noise emission

Rain

Melting layer

Refractive effects VSAT FIGURE 8.3 Illustration of the various propagation loss mechanisms on a typical earth–space path. The earth terminal (in this example a very small aperture terminal or VSAT) is directed toward a satellite. Refractive effects (causing tropospheric scintillation); gases; a rain cloud, melting layer, and rain, all exist in the path and cause signal loss. The absorptive effects of the atmospheric constituents cause an increase in sky noise to be observed by the VSAT receiver. While atmospheric gases and tropospheric scintillation do not cause signal depolarization, collections of nonsymmetrical ice crystals and rain particles can depolarize the transmissions through them. Above the lower (neutral) atmosphere is the ionosphere, which begins at about 40 km and extends well above 600 km. The ionosphere can cause the electric vector of signals passing through it to rotate away from their original polarization direction, hence causing signal depolarization. At certain times of the day, year, and 11-year sunspot cycle, the ionosphere can cause the amplitude and phase of signals passing through it to change rapidly, i.e., to scintillate, about a general mean level. The ionosphere has its principal impact on signals at frequencies well below 10 GHz while the other effects noted in the figure above become increasingly strong as the frequency of the signal goes above 10 GHz. Finally, if the sun (a very “hot” microwave and millimeter wave source of incoherent energy) is in the VSAT beam, an increased noise contribution results which may cause the C/N to drop below the demodulator threshold. Note: The above picture is not drawn to scale. Most rainstorms occur below 10 km altitude and the ionosphere is not normally present below 40 km, and extends to more than 1000 km above the earth.

8.2 QUANTIFYING ATTENUATION AND DEPOLARIZATION Attenuation, A, is the decibel difference between the power received, Pr, at a given time t and the power received under ideal propagation conditions (often referred to as “clear sky” conditions). With all values in decibel units, we have A1t2  Prclearsky  Pr 1t2

(8.2)

Attenuation, A(t), on satellite communications links operating at C, Ku, and Ka-band is primarily caused by absorption of the signal in rain. On most satellite links above 10 GHz,

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Tropospheric scintillation and gaseous attenuation Cloud attenuation and effects of melting layer Heavy thunderstorm rain 10−10

Performance threshold

10−8

BER

Degraded performance region Availability threshold

10−6

10−4

10−2 0.001

0.01

0.1 1.0 10.0 Time percentage BER level measured

100%

FIGURE 8.4 Approximate range of annual time percentages that various atmospheric impairments affect a link (after Figure 2 of reference 3 © John Wiley & Sons, Inc. Reprinted with permission). Tropospheric scintillation (a refractive effect in the lower atmosphere) and gaseous attenuation are pervasive phenomena that occur all of the time, but at different levels of impact depending on the climate, elevation angle, and time percentage of interest. Clouds exist at various time percentages, depending on the climate, but are generally present for at least 30% of the time in most locations. As the concentration of the frozen particles in the cloud increases, many will start to fall and will melt on reaching the 0°C isotherm. This will lead to enhanced attenuation in the melting layer. Drizzle rain will fall when the water vapor concentration reaches saturation levels. Such rain is usually stratiform and falls for between 1 and 10% of the time, depending on the climate. During hot periods, convective rain will fall, often in the form of thunderstorms. Heavy thunderstorms account for the highest rainfall rates, and hence the highest path attenuations encountered, but they exist for only small time percentages in a year. Not shown in the above figure are ionospheric effects, which have a diurnal, seasonal, and 11-year cyclical impact, again depending on where the earth station is and the precise earth–space path used.

rain attenuation limits the availability of the system and, to develop an adequate link margin, the rain attenuation to be expected for a given time percentage needs to be calculated. This can be a complicated process, but there are basically three steps: (a) determine the rainfall rate for the time percentage of interest; (b) calculate the specific attenuation of the signal at this rainfall rate in dB/km; and (c) find the effective length of the path over which this specific attenuation applies. The difficult part of this process is part (c) because rain falls in two broad categories: stratiform rain and convective rain. These two separate atmospheric mechanisms have different effects on satellite paths. Stratiform rain is generated in cloud layers containing ice, and results in widespread rain or snow at rainfall rates of less than 10 mm per hour. Convective rain is generated by vertical air currents that can be very powerful, leading to thunderstorms and high rainfall rates. Convective rain is very important for satellite communication systems because it is the major cause

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Path to the satellite

Stratiform rain event

Melting level Rain Ground level (a )

Convective rain event

Melting level

Path to the satellite

Rain

Ground level (b)

FIGURE 8.5 (a) Stratiform rain situation. In this case, a widespread system of stratiform rain—that is rain that appears to be stratified horizontally—completely covers the path to the satellite from the ground up to the point where the rain temperature is 0°C. This level is called the melting level because, above it, the precipitation is frozen and consists of snow and ice crystal particles. Frozen precipitation causes negligible attenuation. In general, the signal path in stratiform rain will exit the rain through the top of the rain structure. (b) Convective rain situation. In this case, a tall column of convective rain enters the satellite-toground path. In some cases the storm will be in front of the earth station; in others, behind it. Convective storms normally occur in the summer, thus the melting level is much higher than in winter. In many cases, the melting level is not well defined, as the strong convective activity inside the storm will push the liquid rain well above the melting level height. Except for paths with very high elevation angles (70°), the signal path in convective rain will most often exit from the side of a convective rainstorm.

of link outages. Stratiform rain consists of a generally constant rainfall rate over a very large area while convective rain is generally confined to a narrow, but tall, column of rain. Figure 8.5 illustrates the two rain processes and Figure 8.6 gives the concept of the path attenuation calculation procedure for both rain types. Stratiform rain occurs typically ahead of a warm front in an area of low pressure. Large areas of cloud exist in which ice crystals are sufficiently large to slowly fall and join other ice crystals to form snowflakes, which fall more quickly as their size increases. If there is a high concentration of moisture in the clouds, in the form of ice, large snowflakes may form. The snow falls until it reaches the melting layer. The melting layer is simply the region of the atmosphere where the temperature transitions from below 0°C

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301

Path to the satellite Melting level

hr

θ

Ground level

(a )

Path to the satellite Melting level

Effective path length box

Height adjustment factor Horizontal adjustment factor

Ground level (b)

FIGURE 8.6 (a) Stratiform rain attenuation calculation procedure. In the case of stratiform rain, the rainfall rate along the path can be considered to be uniform and the path completely immersed in the rain. The effective path through the rain—the path over which the rain may be considered to be uniform—is therefore the same as the physical path length in stratiform rain. The path attenuation A is therefore the specific attenuation (i.e., dB attenuation per km) multiplied by the physical path length in the rain (i.e., hrsin ). (b) Convective rain attenuation calculation procedure. In the case of convective rain, the melting level and elevation angle are used to develop two adjustment factors: a height adjustment factor and a horizontal adjustment factor. Once these factors have been used, a smaller box is created inside which it may be assumed that the rainfall rate is uniform. The length of the path that exists inside this box is the effective path length and it is this that is used to multiply the specific attenuation with. In this case, the path exits through the top of the effective path length box. In other cases, it may exit through the side.

to above 0°C. Snow falling into air at a temperature greater than 0°C melts and forms raindrops. If the air at the earth’s surface is below 0°C the snow does not melt, but continues to the ground. The stratiform cloud mechanisms that generate snow result in low rainfall rates, always less than 10 mm per hour, and widespread (stratiform) rain or snow. This leads to generally constant attenuation of the slant-path signals over the entire path length from the ground to the melting layer. EXAMPLE 8.2.1 An earth station at sea level communicates at an elevation angle of 35° with a GEO satellite. The melting level height of the stratiform rain is 3 km. Find (a) the physical pathlength through the rain; (b) find the path attenuation if the specific attenuation is 2 dB/km.

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Answer (a) The vertical height, hr, of the rain is the difference between the melting level height (3 km) and the height of the earth station (0 km, since it is at sea level), which gives hr  3 km. Since the elevation angle is 35°, the physical pathlength, L, through the rain is given by L  hr(sin 35)  3(sin 35)  5.23 km. (b) The rain is stratiform and so it can be considered to be uniform over the path. The specific attenuation is therefore uniform along the path through the rain. If the specific attenuation is 2 dB/km, the path attenuation, A is given by A  (2 dB/km)  (5.23 km)  10.46 dB  10.5 dB 

Convective rainstorms are very complex, and have both horizontal and vertical structure. A convective cell becomes established when a mass of warm moist air is pushed up into colder air at a higher altitude. Adiabatic expansion of the air mass occurs, which cools the air. When the air is cooled below its dew point, it condenses forming clouds, and drops of water start to fall under gravity. The falling drops collide and coalesce with other drops to make larger drops, leading to a drop size distribution. The maximum size of stable raindrops is about 6 mm—larger drops (sometimes exceeding 10 mm in average diameter) are unstable and quickly up break into a collection of smaller drops under wind shear conditions. Large raindrops fall quickly, with terminal velocities up to 8 or 9 m/s. If the falling drops encounter supercooled water as they fall, hailstones can form. Hailstones can exceed the 10 mm diameter limit of raindrops, and may reach golf ball size in severe thunderstorms in the Great Plains. The accretion process can occur in an updraft as well as for falling drops, and in a vigorous thunderstorm, updraft velocities can exceed 100 mph. Since cold air is denser than warm air, once an updraft dies away at the top of a thunderstorm, cold air tends to flow downward, and can create a streamer, a narrow region of intense rain and cold air. Streamers can be a few hundred meters wide or a kilometer wide. At the surface, the streamer is observed as a microburst, which has strong wind shear as the vertical down flow of cold air hits the ground and spills out in all directions. We are all familiar with microbursts. Shortly before heavy rain falls there is often a cold wind, followed by a downpour. The cold wind we feel is the outflow of cold air as it hits the earth’s surface. The effect of convective rain on a satellite slant path depends on the angle at which the path intersects a streamer. Streamers are rarely vertical, so if a slant path

SIDEBAR Microbursts are dangerous for aircraft flying close to the ground, especially when taking off and landing. Several serious accidents to passenger aircraft in the 1980s were attributed to the wind shear associated with microbursts, and extensive research was carried out to develop ways to detect microbursts and wind shear. Networks of anemometers, which measure wind speed, can be deployed around an airfield to detect wind shear, and terminal Doppler radar can be used for the same purpose. An aircraft on final approach to a runway is flying slowly and descending on a 3° glide slope. If the aircraft encounters a microburst, it first experiences a headwind, which increases its speed relative to the air

and tends to slow the rate of descent. The natural reaction of the pilot is to reduce engine power to maintain a constant rate of descent on the 3° glide slope. However, as soon as the aircraft passes the center of the microburst, the wind direction is opposite, and is now a tailwind, which reduces the speed of the aircraft relative to the air and increases the rate of descent of the aircraft. If the engine power has been cut, the airplane may sink into the ground before the engines have developed enough power to keep the airplane aloft. Wind shear detection equipment at airfields and improved pilot awareness of the dangers of microbursts has reduced the incidence of accidents caused by microbursts.

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303

Height above surface (km)

6

4

Heavy rain

Medium rain 2

18° slant path

Light rain

0

0

4

8

12

Range (km) FIGURE 8.7 Example of an RHI scan through a rain storm. Radar reflectivity contours in a rainstorm on June 15, 1986, measured with an S-band radar in Blacksburg, Virginia. The contours represent light, medium, and heavy rain in a narrow vertical column. The radar and a receiving station were collocated at the (0,0) point. Note the narrow extent of heavy rain in a sloping column, and the effect on the slant path to the satellite at an elevation angle of 18°. The statistical rain height Hi for Blacksburg is 4.1 km. In this example, rain is present up to an altitude of 5.6 km above sea level.

is parallel to a streamer it will suffer very heavy attenuation if the streamer envelops the path. If the slant path cuts across the streamer, the path length within the heavy rain may be quite short, leading to relatively little attenuation despite the high rainfall rate. Figure 8.7 shows an examples of a convective rain cell observed with an S-band radar at Virginia Tech’s satellite tracking station. The radar was used to make vertical scans across the slant path to a satellite (known to radar people as an RHI scan, for range-height indicator, a WWII radar display mode). The complex shape of the storm cell requires the use of artificial “adjustment factors” to convert the physical path through the rainstorm to an effective path length over which the rain may be considered to be uniform. As well as causing significant attenuation, rain and ice crystals can cause depolarization. Depolarization is more difficult to quantify than attenuation. All signals have a polarization orientation that is defined by the electric field vector of the signal. (See Figure 8.8.) In general, signals are never purely polarized; the direction of the electric field will never be perfectly oriented or constant. Successful orthogonal polarization frequency sharing— usually called dual-polarization frequency reuse—requires that there be sufficient isolation between two orthogonal polarization states to permit the separation of the wanted polarization (the copolarized signal) from the unwanted polarization (the cross-polarized signal) at the receiving antenna5. The difference between the copolarized and the cross-polarized signal energy will determine the cross-polarization discrimination at the receiver, the XPD, and hence the level of interference between two orthogonally polarized signals. To illustrate the process by which depolarization is measured; imagine a dualpolarized antenna transmitting orthogonally polarized signals. We will call the two polarizations V (for vertical) and H (for horizontal) for convenience, although there

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FIGURE 8.8 Orthogonally polarized waveguide horn antennas. The polarization of an electromagnetic wave is defined by the orientation of the electric vector. In the example above, two waveguide horns, excited in the TE10 mode, are radiating in the same direction. The top horn is oriented such that the electric vector is vertically polarized; the bottom horn is turned on its side compared with the top horn and so the electric vector is horizontally polarized. The arrows indicate the electric field vector. Since the electric polarization vectors are oriented 90° with respect to each other in the two horns, the transmitted signals are considered to be orthogonally polarized. Orthogonally polarized signals do not interfere with each other, even if they are at exactly the same frequency, provided they are “purely” polarized (i.e., there is no component of the signal present in the other, orthogonal, polarization). In all cases, however, the transmitted signals are not purely polarized, due to antenna imperfections, so a component exists in the unwanted polarization. In addition, some of the energy in one polarization can “cross” over to the other polarization due to asymmetric particles (e.g., large, oblate raindrops) existing in the propagation path. This cross-polarized energy can give rise to interference between the two, mutually orthogonal polarizations. The degree of cross-polarization to be expected along a given path is predicted using cross-polarization models that are usually based on the rain attenuation along the path.

are infinitely many orthogonal polarization pairs. Let the complex phasor amplitudes of the transmitted electric field vectors with polarization V and H be a and b, respectively, as shown in Figure 8.9. The transmitting antenna is excited so that a and b are equal. If the transmission medium between the transmitting and receiving antennas were clear air, phasor a would give rise to a V polarization wave of amplitude ac at the receiving antenna and phasor b would cause an H polarization wave of amplitude bc. The subscript c stands for copolarized; these fields have the same polarization sense as their transmitted counterparts. (See Figure 8.10.) If asymmetrical rain or ice crystal particles exist in the transmission medium, some of the energy in a will couple into a small (cross-polarized) H polarized field component

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305

a

V Direction of propagation

H b

FIGURE 8.9 Fields excited by a dual-polarized antenna. The field radiated by the V horn has the vertically polarized electric field vector indicated by a and the field radiated by the H horn has the horizontally polarized field vector indicated by b. In most antenna systems, one horn is used to radiate both polarizations simultaneously rather than two. This permits the single feed horn to be located at the prime focus of the antenna to generate the best far field pattern5. The two polarization senses to be transmitted are excited in separate parts of the transmitter and are then coupled together via an ortho-mode transducer into a single waveguide section. This waveguide section, which can support both polarizations simultaneously, is then used to couple the signals into a waveguide horn that is capable of radiating both polarizations senses equally.

whose amplitude at the receiving antenna is ax, and b will give rise to a small (cross-polarized) V polarized component bx. An ideal receiving system that introduces no cross-polarization will have a V channel output (ac  bx) and an H channel output (bc  ax). The unwanted bx term represents interference with the wanted signal ac and the unwanted ax term is interference with the wanted signal bc. This interference will cause cross talk on an analog link and increase the BER on a digital link. This generation of unwanted cross-polarized components is called depolarization. Vertical

Vertical

a

ac

At transmitting antenna

At receiving antenna bx

ax bc

b Horizontal

Horizontal

FIGURE 8.10 Illustration of signal depolarization in the transmission path. The transmitted fields a and b produce copolarized components ac and bc at the receiving antenna. The transmission medium in this instance is not clear sky, nor is the transmitting antenna perfectly polarized, and the anisotropy of the transmission medium and imperfections in the transmitting antenna induce cross-polarized components of the transmitted signal to be received. These cross-polarized components at the receiving antenna are ax and bx. With perfect antennas and in the absence of depolarization ax and bx would be zero.

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The measure of depolarization that is most useful for analyzing communications systems is the cross-polarization isolation, XPI. In terms of the complex phasor field amplitudes, it is given by Eq. (8.3) for the V polarized channel and by Eq. (8.4) for the H polarized channel. XPIV  ac bx XPIH  bc ax

(8.3) (8.4)

The XPI values are commonly expressed in decibels; for example, XPIV  20 log10 0ac bx 0 dB

(8.5)

Physically, the XPI is the decibel ratio of wanted power to unwanted power in the same channel. The larger the XPI value, the less interference there is and the better the communications channel will perform. XPI is difficult to measure. It requires the simultaneous transmission of signals at the same frequency in both polarization senses. The COMSTAR series of satellites had a beacon that rapidly switched between two orthogonal polarization senses, thus permitting the measurement of XPI [e.g., 6, 7]. More recently, the ACTS8 and OLYMPUS9 satellites also incorporated a switched beacon to permit XPI measurements. Most propagation experiments are much simpler than this and measure simultaneously the wanted (the copolarized) and the orthogonal, unwanted (the crosspolarized) signals that are received from a satellite beacon that transmits in only one polarization. In this case (referring to Figure 8.10) the experiment would measure (say) signals ac and ax that are derived from a singly polarized signal a that is transmitted from the satellite. Measuring received signals bc and bx simultaneously from a singly polarized signal b would provide the same result. This process allows the cross-polarization discrimination, XPD to be derived XPDV  ac ax

(8.6)

XPDV  20 log10 0ac ax 0 dB

(8.7)

or in decibels

In most transmission situations encountered in practice, the values calculated for XPI and XPD are the same10 and they are sometimes simply called the “isolation.” In practice, real antennas do not transmit polarization pairs that are exactly orthogonal, nor does the isolation remain the same over the 3-dB beamwidth of the antenna. Receiving antennas can also introduce cross-polarization. There is therefore a residual XPD component present even in clear sky conditions. This must be accounted for in the link budget of a dual-polarized, frequency reuse system. The residual XPD on axis is normally better for linearly polarized antennas (30 to 35 dB) than for circularly polarized antennas (27 to 30 dB). These values represent antennas carefully designed for dual-polarized operation; inexpensive antennas will typically exhibit about 20 dB XPD for linear or circular polarizations.

8.3 PROPAGATION EFFECTS THAT ARE NOT ASSOCIATED WITH HYDROMETEORS In this section we will discuss propagation effects that are not associated with raindrops or ice crystals: atmospheric absorption, cloud attenuation, refractive effects that include tropospheric scintillation and low angle fading/multipath effects, Faraday rotation, and ionospheric scintillation.

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307

Atmospheric Absorption At microwave frequencies and above, electromagnetic waves interact with molecules in the atmosphere to cause signal attenuation. At certain frequencies, resonant absorption occurs and severe attenuation can result. Figure 8.11 (from Figure 6 of reference 11) shows these resonant absorption peaks on a zenith path (that is, a path at an elevation angle of

103 5

a

2

102 5

Zenith attenuation (dB)

2

10 5

B

2

1 5

A

2

10−1 5

2

10−2

3

5

10

2

5

102

2

3.5

Frequency (GHz) FIGURE 8.11 Total zenith attenuation due to atmospheric gases calculated from 3 to 350 GHz (from Figure 6 of reference 11 © ITU, reproduced with permission). The two curves represent the gaseous attenuation that would be observed looking straight up from sea level (i.e., on a zenith path) right through the neutral atmosphere on a satellite–earth path. Curve A is for a dry atmosphere (i.e., no water vapor present) while curve B is for a standard atmosphere. A standard atmosphere consists of a surface pressure of 1013 hPa [a hPa has the same numerical value as the old pressure unit of millibars], a surface temperature of 15°C, and a surface relative humidity of 7.5 mg/m3. Curve A shows only the resonant absorption peaks of the oxygen molecules (a broad peak at 60 GHz and a narrow peak at 118.75 GHz). Curve B includes the resonant absorption peaks due to the water vapor molecule at 22.235, 183.31, and 325.153 GHz. The shaded portion “a” indicates a range of values since there are many individual resonant absorption lines in this frequency region.

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90°) from a sea level location right through the neutral atmosphere. Neutral means that no ionization is present. The first absorption band in Figure 8.11 is that due to water vapor at 22.235 GHz. The K-band sets of frequencies are on both sides of this absorption band, which has led to the terminology of Ku band (signifying frequencies under the absorption band) and Ka band (signifying frequencies above the absorption band). It is common to specify a satellite frequency band by the uplink frequency. From Figure 8.11, it can be seen that gaseous absorption accounts for less than 1 dB on most paths below 100 GHz that lie outside the absorption bands. However, in many new systems that employ very small system margins, it is important to account for the gaseous losses along the anticipated path. New prediction procedures that attempt to account for all attenuating phenomena along the path (e.g., [12]) include gaseous absorption.

Cloud Attenuation Once considered to be largely irrelevant for satellite communications paths, clouds have become an important factor for some Ka-Band paths and all V-Band (5040 GHz) systems. The difficulty with modeling cloud attenuation is that clouds are of many types and can exist at many levels, each type having a different probability of occurrence. The water droplet concentrations in each cloud will also vary, and clouds made up of ice crystals cause little attenuation. Two models have been proposed13,14, both of which have similar accuracy. Typical values of cloud attenuation for water-filled clouds are between 1 and 2 dB at frequencies around 30 GHz on paths at elevation angles of close to 30° in temperate latitudes. In warmer climates, where clouds are generally thicker in extent and have a greater probability of occurrence than temperate latitudes, cloud attenuation is expected to be higher. As with most propagation effects, the lower the elevation angle, the higher the cloud attenuation.

Tropospheric Scintillation and Low Angle Fading The atmosphere close to the ground, sometimes called the boundary layer, is rarely still. Energy from the sun warms the surface of the earth and the resultant convective activity agitates the boundary layer. This agitation results in turbulent mixing of different parts of the boundary layer, causing small-scale variations in refractive index. Figure 8.12 illustrates the process.

Stratified Layers (calm conditions)

Turbulent Mixing (convective conditions)

(a )

(b)

FIGURE 8.12 Schematic of stratified and turbulent conditions in the boundary layer of the atmosphere. In (a), the air is calm and the lower atmosphere next to the earth’s surface (the boundary layer) forms into layers. Each layer has a slightly different refractive index, decreasing in general with height. In (b), the earth’s surface has become heated by energy from the sun and the resultant convective activity has mixed the formerly stratified layers into “bubbles” that have different refractive indices. The turbulent mixing of the lower atmosphere will cause relatively rapid fluctuations in a signal passing through it, which are called scintillations.

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0 −2 −4 −6 −8 −10 11.00

28/07/76 11.10 11.20 Time GMT

Copolar (dB)

Copolar (dB)

When a signal encounters a turbulent atmosphere, the rapid variation in refractive index along the path will lead to fluctuations in the received signal level. These fluctuations are generally about a fairly constant mean signal level and are called scintillations. Because the bulk of the fluctuations are caused within 4 km of the earth’s surface, they are referred to as tropospheric scintillations. Tropospheric scintillations occur in many weather conditions, as can be seen in Figure 8.13. Tropospheric scintillation does not cause depolarization. The magnitude of the scintillations becomes generally larger as the frequency increases, the path elevation angle reduces, and the climate becomes warmer and more humid. Prediction models exist to calculate this phenomenon with good accuracy15. On paths below 10° elevation angle, tropospheric scintillation can be performance limiting; below 5° elevation angle it can become availability limiting. When the elevation angle falls below 10°, a second propagation effect becomes noticeable: low angle fading. Low angle fading is the same phenomenon as multipath fading on terrestrial paths. A signal transmitted from a satellite arrives at the earth station receiving antenna via different paths with different phase shifts. On combination,

11.30

0 −2 −4 −6 −8 −10 16.35

0 −2 −4 −6 −8 −10 11.30

06/07/76 11.40 11.50 Time GMT

12.00

0 −2 −4 −6 −8 −10 −12 −14

18.30

20/01/76 07.35 07.45 Time GMT (c )

19/07/76 18.40 18.50 Time GMT

19.00

(e )

07.55

Copolar (dB)

Copolar (dB)

(b) 0 −2 −4 −6 −8 −10 07.25

17.05

(d )

Copolar (dB)

Copolar (dB)

(a )

19/04/76 16.45 16.55 Time GMT

0 −2 −4 −6 −8 −10 09.00

03/05/76 09.10 09.20 Time GMT

09.30

(f )

FIGURE 8.13 Scintillations observed under a variety of weather conditions on a 30-GHz downlink from ATS-6 (from reference 16). Scintillations with various amplitudes can be observed under different weather conditions. Two of the data sets were taken in clear weather, two in cloud conditions, and two during rain, as follows: (a) clear-weather copolar signal with low scintillation; (b) clear-weather copolar signal with high scintillation; (c) copolar scintillation in cloud; (d) copolar scintillation in cloud; (e) copolar scintillation and attenuation in rain; (f ) copolar scintillation and attenuation in rain. Note the difference in scintillation amplitude under what are apparently similar weather conditions along the path.

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the resultant waveform may be enhanced or attenuated from the normal clear sky level. Signal enhancement has been observed to exceed 8 dB on a 3.3° path at 11.198 GHz17, while cancellation can cause complete link dropout. The mechanism for low angle fading has been interpreted as atmospheric multipath and also as the “defocusing and focusing” of the incoming signal. Both explanations have merit: the received signal is made up of components that have arrived via different paths (i.e., multipath), but the mechanism for developing the different paths is one of refraction rather than reflection at the atmospheric layer boundaries. Low angle fading is only significant in very still air on very low elevation angle paths. It is normally not considered for satellite paths when the elevation angle is above 10°. Note that the multipath effect referred to here is occurring in the atmosphere, and is therefore different from multipath effects in terrestrial radio links which are caused by reflections from the ground, buildings, trees, etc.

Faraday Rotation in the Atmosphere The ionosphere is that portion of the earth’s atmosphere that contains large numbers of electrons and ions. At its lowest, it reaches down to close to 40 km above the earth; there is no distinct upper boundary, but it exists well above 600 km above the earth. The ionosphere completely dominates radio propagation below about 40 MHz, but its effects on the frequencies used by most communications satellites are minor. Electrically, the ionosphere is an inhomogeneous and anisotropic plasma and an exact analysis of wave propagation through it is extremely difficult. For a given frequency and direction of propagation with respect to the earth’s magnetic field, there exist two characteristic polarizations. Waves with these polarizations, called characteristic waves, propagate with their polarization unchanged. Any wave entering the ionosphere can be resolved into two components with the characteristic polarizations. The phase shift and attenuation experienced by the characteristic waves can be calculated at any point along the propagation path, and the total field can be computed as the vector sum of the fields of the characteristic waves. This total field can be interpreted as an attenuated and depolarized version of the wave that entered the ionosphere. Thus, when a linearly polarized (LP) satellite path signal reaches the ionosphere, it excites waves with the two characteristic polarizations. These travel at different velocities, and when they leave the ionosphere their relative phase is different from when they entered. The wave that leaves the ionosphere has a different polarization from the LP wave that was transmitted. This is called Faraday rotation, and its effect is essentially the same as if the field vector of the transmitted LP wave had been rotated by an angle . For a path length through the ionosphere of Z meters, the rotation angle  is given by f



a

2.36  104 b ZNB0 cos u dz rad f2

(8.8)

Here,  is the angle between the geomagnetic field and the direction of propagation, N is the electron density in electrons/cubic meter, B0 is the geomagnetic flux density in Teslas, and f is the operating frequency in Hz. The rotation angle  varies inversely with f 2. Table 8.1 gives the value of  and some other parameters with frequency15. The polarizations of an earth station antenna can be adjusted to compensate for the Faraday rotation observed under average conditions. However, the rotation of the uplink will be in an opposite sense to that on the downlink and so, to compensate in both directions at the same time, a feed will be required that is able to rotate the relevant sections in opposite

20 5 dB 1 dB 0.4 ps/Hz See Rec. ITUR P.531

1f2 1/f2 1f3 See Rec. ITUR P.531

30 rotations 25 s 1°

0.1 GHz

1f2

1f 1f2 1f2

2

Frequency Dependence

0.8 dB 0.16 dB 0.026 ps/Hz See Rec. ITUR P.531

3.2

4.8 rotations 4 s 0.16°

0.25 GHz

0.2 dB 0.04 dB 0.0032 ps/Hz See Rec. ITUR P.531

48

1.2 rotations 1 s 2.4

0.5 GHz

0.05 dB 0.01 dB 0.0004 ps/Hz 20 dB peakto-peak

12

108° 0.25 s 0.6

1 GHz

0.12

5  104 dB 104 dB 4  107 ps/Hz 4 dB peakto-peak

6  103 dB 0.001 dB 1.5  105 ps/Hz 10 dB peakto-peak

1.1° 0.0025 s 0.36

10 GHz

1.32

12° 0.028 s 4.2

3 GHz

(a) The estimated values are based upon a total electron content (TEC) of 1018 electrons/m2, which is a high value of TEC encountered at low latitudes in day-time with high solar activity. (b) The scintillation values in the last entry are maximum values observed near the geomagnetic equator during the early nighttime hours (local time) at equinox under conditions of high sunspot number.

NOTES:

Variation in the Direction of arrival (r.m.s.) Absorption (auroral and/or polar cap) Absorption (mid-latitude) Dispersion Scintillation

Faraday Rotation Propagation Delay Refraction

Effect

TABLE 8.1 Estimated Ionospheric Effects for Elevation Angles of About 30° One-Way Traversal (from TABLE 1 of [15] © ITU, reproduced with permission)

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directions. The XPD that results when the polarization angle of an LP wave changes by an amount  is given by XPD  20 log10 1cot ¢f2

(8.9)

Hence, a 6° change from average conditions would reduce the XPD on the link to about 19.6 dB.

Ionospheric Scintillations Energy from the sun causes the ionosphere to “grow” during the day, increasing the total electron content (TEC) by two orders of magnitude, or more. The TEC is the total number of electrons that would exist in a vertical column of area 1 m2 from the surface of the earth all the way through the earth’s atmosphere. Typical values of TEC range from 1018 during the day to 1016 during the night. It is the rapid change in TEC from the daytime value to the nighttime value, which occurs at local sunset in the ionosphere, that gives rise to irregularities in the ionosphere. The irregularities cause the signal to vary rapidly in amplitude and phase, which leads to rapid signal fluctuations that are called ionospheric scintillations. The magnitude of the ionospheric scintillations varies with time of day, month in the year, and year in the 11-year sunspot cycle. The greatest scintillation effects are observed just after local sunset in the equinox periods during the sunspot maximum years. The effects are also worst within about 20° of the geomagnetic equator and over the poles. The length of the cycles averages at around 11 years, but has been as short as 9.5 years and as long as 12.518. Solar sunspot cycle 22 was from 1986.8 to 1996.4.

8.4

RAIN AND ICE EFFECTS At frequencies above 10 GHz, rain is the dominant propagation phenomenon on satellite links. Many experiments have been conducted on geostationary satellite links, using experimental satellites such as SIRIO, OTS, and CTS (Hermes) at Ku Band and ATS-6, OLYMPUS, and ACTS at Ka Band. One experimental satellite, Italsat, also allowed 5040 GHz (V Band) experiments to be conducted in Europe. References 1, 2, 8, and 9 provide detailed results and explanations of all the propagation phenomena.

Characterizing Rain Most farmers, hydrologists, and city planners need to know how much total rain will fall in a given period: that is, the rain accumulation. Indeed, most weather forecasts are given in terms of how much precipitation will fall (or accumulate) over a given region. Rain accumulation, unfortunately, is of little use to satellite link designers, since it is the rate at which the rain is falling that is important: that is, the rainfall rate. Rainfall rate is measured by a rain gauge, the most common of which is a tipping bucket rain gauge. This is fairly accurate between rainfall rates of 10 to 100 mm/h. Peak values of 100 to 150 mm/h may be expected for short periods during summer thunderstorms in the mid-Atlantic region of the United States. Higher rainfall rates are observed in tropical regions. The long-term behavior of rainfall rate is described by a cumulative probability distribution or by a cumulative distribution function (cdf ). The cdf for rainfall rate is commonly referred to as an exceedance curve. This gives the percentage of time (usually the percentage of 1 year) that the rainfall rate exceeds a given value. Climate related parameters tend to be very variable. Rain accumulation can vary significantly from year-to-year,

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10.0000 3-year average 1979 1980 1981

Percentage of time rain rate is exceeded

1.0000

0.1000

0.0100

0.0010

0.0001

0

20

40

60

80

100

120

140

160

180

200

Rain rate (mm/h) FIGURE 8.14 Typical rainfall rate cumulative probability distributions or “exceedance” curves. These sets were measured at Virginia Tech, Blacksburg, United States as part of a 3-year experiment with the Italian satellite, SIRIO. The 1979 data indicate a relatively dry year, while those of 1981 indicate a relatively wet year. Despite this, a single, rare thunderstorm in 1979 produced much higher rainfall rates than those observed in 1981 at low time percentages. The availability level the link has to operate at will determine what rainfall rate is of most importance and it will also give a range over which the design must cope. For example, if 0.01% was the availability requirement, in 1979 the rainfall rate for this time percentage was 38 mm/h while in 1981 it was 58 mm/h. This shows the value of long-term statistics so that one year’s data do not bias the link design.

as can the exceedance curves, particularly at the low time percentages of interest to satellite link designers. Three annual exceedance curves taken from an experiment performed at Blacksburg, Virginia, are shown in Figure 8.14 and it can be seen that the rainfall rate at the 0.01% point varies between 38 and 58 mm/h over the 3 years. The cumulative attenuation curves for the 3 years showed similar trends. Depending on the elevation angle of the link, this can make a significant difference in the attenuation measured at the same time percentage in each of the years. For this reason, link designers prefer to use values averaged over many years of measurements for their propagation models. There have been two approaches to developing these long-term statistics: rain climate maps and exceedance contour maps.

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Rain Climate Maps Rain climate maps were the first approach to developing long-term rainfall rate statistics that could be used for both propagation predictions and interference calculations. These maps divide the world into regions where the average rainfall rate statistics are the same, within a margin of about 10%. An example of a rain climate map is shown in Figure 8.15 for the Americas19. The rainfall rate statistics for the 15 rain climate

165° 75°

135°

105°

75°

45°

A

15° 75° N

A

E G C

C B

D

B

E

60°

C

B E

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E

G

60°

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314

Latitude 45° in degrees N

F F

K

K

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M

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0° P

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K E D

45° D A

60° 165°

15°

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Latitude 45° in degrees S

A

135°

105°

75°

45°

60° S 15°

Longitude in degrees W FIGURE 8.15 Rain climatic zones for the Americas (from Figure 1 of reference 19 © ITU, reproduced with permission).

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TABLE 8.2 Rainfall Rate Intensities for the Rain Climatic Zones (From TABLE 1 in [19] © ITU, reproduced with permission) Percentage of Time (%)

A

B

C

D

E

F

G

H

J

K

L

M

N

P

Q

10 0.3 0.1 0.03 0.01 0.003 0.001

0.1 0.8 2 5 8 14 22

0.5 2 3 6 12 21 32

0.7 2.8 5 9 15 26 42

2.1 4.5 8 13 19 29 42

0.6 2.4 6 12 22 41 70

1.7 4.5 8 15 28 54 78

3 7 12 20 30 45 65

2 4 10 18 32 55 83

8 13 20 28 35 45 55

1.5 4.2 12 23 42 70 100

2 7 15 33 60 105 150

4 11 22 40 63 95 120

5 15 35 65 95 140 180

12 34 65 105 145 200 250

24 49 72 96 115 142 170

NOTE: See Figure 8.12 for the Rain Climatic Zones in N. and S. America

regions worldwide are given in Table 8.2. (Note that not all of these regions exist in the Americas). The ease with which the tables and rain climate maps can be used is offset by the clear inaccuracies that occur when large parts of the earth are given the same climate classification. The step-changes across the climate boundaries are also arbitrary and are not supported by measured data. Wherever possible, it is always best to use measured rainfall rate data as the attenuation prediction model input whenever these data exist.

Rainfall Rate Exceedance Contour Maps In an effort to overcome the inaccuracies of the rain climate maps, the ITU developed a set of comprehensive rainfall rate exceedance curves for the whole world. An initial set for the Americas is shown in Figure 8.16. These maps are updated periodically and the latest set can be found on the ITU web site20. An example is shown in Figure 8.1722.

Raindrop Distributions Rain attenuation and depolarization occur because individual raindrops absorb energy from radio waves. The drops absorb some of the incident energy and some is scattered. The size and shape of raindrops have been measured2. The most common mathematical description of the distribution of raindrop sizes is exponential and of the form N1D2  N0e1DDm2 mm1 m3

(8.10)

where Dm is the median drop diameter and N(D) dD is the number of drops per cubic meter with diameters between D and D  dD mm. The rainfall rate R is related to N(D) and also to the terminal velocity V(D) of the falling drops in meters per second with diameter D by23 R  0.6  103p

 D V1D2N1D2dD mm/h 3

(8.11)

The details of scattering and absorption by a single raindrop, and the summation over the drop population that the calculation of path attenuation from the drop size distribution requires are beyond the scope of this text.

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165° 75°

150°

135°

120°

105°

90°

75°

60°

45°

30° 75° N

10

15 15 10

60°

60° 20 30 40 50

45°

60 70 80 0 9 0 0 1 0 12

30° 100

15°

0

15°

14 110

110

130

8 70 0

Latitude

30°

45°

110 15°

0°

90

0°

15°

100 90

30°

20 30 40

45°

80 70 60 50

30°

45°

30

60° 165°

150°

135°

120°

105°

90°

75°

60°

45°

60° S 30°

Longitude in degrees W FIGURE 8.16 Rainfall rate exceedance contours for the Americas (from reference 21 © ITU, reproduced with permission). This was the first of a set of three rainfall rate exceedance contours that were developed for the world. In this version, the contours only existed over land. The latest versions20 include data over all of the surface of the earth (see Figure 8.17).

SIDEBAR The first measurements of raindrop size distributions were made by Laws and Parsons33 in 1944 with a very ingenious experiment. The experiment was designed to find the sizes of raindrops in typical rainstorms. A baking pan (typically 1 m  0.5 m) was filled with flour and placed out in the rain for a minute. The pan of flour was then baked in an oven, and the loose flour sifted out. What remained were pellets of baked flour where raindrops had fallen and

316

been absorbed by the flour, with dimensions that corresponded to the raindrops that hit the tray in that particular storm. From their flour measurements, Laws and Parsons derived an empirical exponential relationship between the rain rate and the drop size distribution that is still used today. The experiment is repeated occasionally—by schoolchildren studying earth sciences—as a demonstration of rainfall characteristics.

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317

90°

80°

70°

Latitude in degrees N

60°

50°

40°

30°

20°

10° 120

0°

300°

320°

340°

0°

20°

40°

60°

80°

Longitude in degrees E FIGURE 8.17 Rain intensity (mm/h) exceeded for 0.01% of the average year (Figure 2 of reference 22). This map provides rainfall rate contours for the Northern Hemisphere between longitudes 300° E and 80° E (Europe, North Africa, the Middle East, and parts of Russia, India, and China).

8.5

PREDICTION OF RAIN ATTENUATION Attenuation by rain can be predicted accurately if the rain can be precisely described all the way along the path. Path attenuation is essentially an integral of all the individual increments of rain attenuation caused by the drops encountered along the path. This is the physical approach to predicting rain attenuation. Unfortunately, rain cannot be described accurately along the path without extensive meteorological databases, which do not exist in most regions of the world. Most prediction models therefore resort to semiempirical approaches, which calculate an effective path length through the rain, Leff, over which the rainfall rate is assumed to be constant. This constant rainfall rate leads to a constant specific attenuation, R, and the path attenuation, A, is simply given by A  specific attenuation  effective path length in rain  gRLeff dB

(8.12)

The semiempirical approach is based on two premises: (1) Rainfall rate measured at a point on the surface of the earth is statistically related (over a period of at least a year) to the attenuation encountered along the path to a satellite from that same point; (2) The actual

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length of the path through the rain medium can be adjusted such that an “effective” path length is developed over which the rain can be considered to be homogeneous (see Figure 8.6). The estimation of attenuation on the slant path to a satellite is essential to the process of establishing a margin in the link budget that ensures the required availability of the link is met. Over a period of many years, several attenuation models have been developed that have been widely used. These include the Crane model35, the simple attenuation model (SAM)36, the Dissanayake, Haidara, Allnutt (DAH) model4 and several models published by the CCIR and ITU-R15,21,25. The ITU-R model, based on the DAH model15, is discussed in detail here, because it provides the most accurate statistical estimate of attenuation on slant paths, worldwide, at the time of writing. Appendix D discusses the simple attenuation model (SAM), developed at Virginia Tech by Warren Stutzman and Keith Dishman. The SAM model is less accurate than the ITU-R model, but allows the user to quickly obtain an estimate of the slant-path attenuation at any frequency and rain rate. A power law equation describes the relationship between point rainfall rate R and specific attenuation, R, the attenuation measured over 1 km24 gR  k1R0.01 2 a dB/km

(8.13)

In Eq. (8.13), the suffix 0.01 to R denotes the rainfall rate measured for 0.01% of the average year, a typical input time percentage for most models. Equation (8.13) holds for all values of rainfall rate, however. The parameters k and  are frequency dependent. Table 8.3 gives values for k and  for frequencies between 4 and 50 GHz25. TABLE 8.3 Regression Coefficients for Estimating Specific Attenuation (from TABLE 1 in [25] © ITU, reproduced with permission) Frequency (GHz)

kH

kV

H

V

4 6 8 10 12 20 30 40 50

0.000650 0.00175 0.00454 0.0101 0.0188 0.0751 0.187 0.350 0.536

0.000591 0.00155 0.00395 0.00887 0.0168 0.0691 0.167 0.310 0.479

1.121 1.308 1.327 1.276 1.217 1.099 1.021 0.939 0.873

1.075 1.265 1.310 1.264 1.200 1.065 1.000 0.929 0.868

NOTES: (1) The suffices V and H refer to vertical and horizontal polarization, respectively (2) Values of k and  at frequencies other than those in the table can be obtained by interpolation using a logarithmic scale for frequency, a logarithmic scale for k, and a linear scale for . (3) Values have been tested and found to be accurate up to a frequency of 40 GHz; values between 40 and 50 GHz are expected to be accurate but have not yet been tested. (4) For linear and circular polarization, and for all path geometries, the coefficients in equation (8.13) can be calculated using the values in the above table and the following equations25 k  3 kH  kV  1kH  kV 2 cos2 u cos 2t4 2

a  3 kHaH  kVaV  1 kHaH  kVaV 2 cos2 u cos 2t4 2k

where  is the path elevation angle and  is the polarization tilt angle relative to the horizontal (  45° for circular polarization).

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EXAMPLE 8.5.1 What is the specific attenuation at 10 GHz if the rainfall rate is 40 mm/h and linear vertical polarization is used? Answer From Table 8.3, kV  0.00887 and V  1.264 at a frequency of 10 GHz. Using Eq. (8.13), we therefore have Specific attenuation  gR  0.00887 1402 1.264  0.9396  0.94 dB/km



If the rainfall rate were constant along the path, as it generally is in light, stratiform rain (see Figure 8.5), then calculating the total attenuation for a given rainfall rate would be simple. The physical path length through the rain, L, would be the same as the effective path length and the total attenuation, A, is given by A  gR  physical path length in rain  gRL dB

(8.14)

On short terrestrial paths (5 km, although this varies with rainfall rate: the lower the rainfall rate, the longer the path), the path length through relatively constant rain can be taken as the distance between the transmitting and receiving antennas. The path through the rain is also at almost the same height along the whole path. This is not the case with satellite paths where the signal follows a slanting path path through the atmosphere, and encounters rain of different types and intensities on the way. Rain can take more than 10 min to fall from a height of 5 km (the approximate upper limit of liquid water in a severe thunderstorm) to the ground. If there are updrafts present, as is always the case in convective rain, it can take even longer. There is therefore no instantaneous relationship between attenuation measured along a path to a satellite and the rainfall rate measured at the earth station site. However, there is a strong statistical relationship between the long-term cumulative statistics of rainfall rate and the long-term statistics of slant-path attenuation. Many models of rain attenuation use equiprobable values of rainfall rate and path attenuation to determine the cumulative statistics of attenuation from those of rainfall rate. Figure 8.18 illustrates the procedure for finding equiprobable values of rainfall rate and path attenuation. The assumption that point rainfall rate on the ground is statistically related (over a period of at least a year) to the attenuation observed on a satellite path to that same point has been validated in many experiments worldwide. Since the path encounters highly variable drop sizes and rainfall rates, the physical length L used in Eq. (8.14) has usually to be replaced by an effective path length Leff. We therefore find that the total path attenuation, A, for a given satellite link is given by Eq. (8.12), which is repeated below for completeness A  gR  effective path length in rain  gR Leff dB

(8.15)

The procedure by which the effective path length is calculated uses the statistical height of rain (i.e., the melting level height), the height of the earth station above mean sea level, and the elevation angle. See Figure 8.19 and earlier Figures 8.5 and 8.6. In Figure 8.19, the rain is shown as filling the complete slant path up to the melting layer. This is a correct assumption in stratiform rain, which exists over large areas and has a relatively constant rain rate along the slant path. It is rarely correct when convective rain is present. The rain rate and drop distributions are not constant, and the path may not pass through the top of the rain cell. Figure 8.20 illustrates the problem. The ITU-R procedure for predicting slant-path rain attenuation for GEO satellite paths is contained in Section 2.2.1.1 of Rec. 61815. It uses a semiempirical approach to the prediction of rain attenuation. Rather than attempt to predict attenuation by inputting rainfall rate at every time percentage, it inputs only the rainfall rate measured (or predicted) for 0.01% of a year. It then extrapolates from this time percentage to other time percentages. While this “one size fits all” approach is nonphysical, it removes the inherent inaccuracies

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Percentage of time rain exceeds R (mm/h) 100 10 1 P 0.1 0.01 0.001

R (mm/h)

R (P )

0

Percentage of time attenuation exceeds A (dB) 100 10 1 P 0.1 0.01 0.001

0

A (dB)

A (P )

FIGURE 8.18 Cumulative statistics of rainfall rate and path attenuation illustrating equiprobable procedures. For a given time percentage, P, the rainfall rate is read off the rainfall rate statistics and the path attenuation is read off the path attenuation statistics. If the data for the two parameters have been taken over a long enough period (at least a year; longer periods in multiples of years), R(P) and A(P) are strongly related. Some models use the full rainfall rate statistics to develop path attenuation statistics. Others use one time percentage to relate the two statistics (e.g., the 0.01% point) and develop the second set of statistics from that single point. The disadvantage of this approach (i.e., it is nonphysical) is outweighed by the improved accuracy obtained by extrapolating to both low and high time percentages, where the rainfall rate measurements are somewhat suspect.

To satellite

L El

He

H0 Earth station

Sea level

FIGURE 8.19 Geometry of a satellite path through rain. The height of the melting layer, shown as He here, is normally considered to be the highest point at which rain attenuation occurs. The rain fills the volume between the melting layer height and the ground. The height of the earth station above mean sea level is given by H0.

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(a )

321

( b)

FIGURE 8.20 Example of different path length geometries. In both cases, a similar rainstorm exists in the slant path. In case A, the path to the satellite exits through the side of the storm cell while in case B it exits through the top (in a similar geometry as in Figure 8.5). The only difference between the two paths is the elevation angle to the satellite. To develop an effective path length to use in Eq. (8.12), use is made of both a “vertical adjustment factor” and a “horizontal adjustment factor” to account for the possibility of either case A or case B occurring.

of using very low rainfall rates for time percentages of 0.1% (and higher) or very high rainfall rates for time percentages of 0.001%. The ITU-R is seeking to change this one size fits all approach. The current procedure (early 2002) is reproduced below (the equation and figure numbers have been changed to correspond with those in this chapter). The following procedure provides estimates of the long-term statistics of the slantpath rain attenuation at a given location for frequencies up to 55 GHz. The following parameters are required: R0.01: point rainfall rate for the location for 0.01% of an average year (mm/h) hS: height above mean sea level of the earth station (km) : elevation angle (degrees) : latitude of the earth station (degrees) f: frequency (GHz). Re: effective radius of the earth (8500 km) The geometry is illustrated in Figure 8.21.

A B C

Ls

θ

(hR′ − hs )

hR′

D

hs LG A: frozen precipitation B: rain height C: liquid precipitation D: Earth–space path FIGURE 8.21 Schematic presentation of an earth–space path giving the parameters to be input into the ITU-R rain attenuation prediction procedure (Figure 1 of reference 15 © ITU, reproduced with permission).

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Step 1: Calculate the rain height, hR , which is equivalent to h0 as given in Recommendation ITU-R P.839. Step 2: For  5° compute the slant-path length, LS, below the rain height from: LS 

1hR¿  hS 2 sin u

(8.16)

km

For   5°, the following formula is used:

21hR¿  hS 2 (8.17) km 21hR¿  hS 2 12 asin u  b  sin u Re Step 3: Calculate the horizontal projection, LG , of the slant-path length from: LS 

2

LG  LS cos u km

(8.18)

Step 4: Obtain the rainfall rate, R0.01, exceeded for 0.01% of an average year (with an integration time of 1 min). If this long-term statistic cannot be obtained from local data sources, an estimate can be obtained from the maps of rainfall rate given in Recommendation ITU-R P.837. Step 5: Obtain the specific attenuation, R, using the frequency-dependent coefficients given in Recommendation ITU-R P.838 and the rainfall rate, R0.01, determined from Step 4, by using: gR  k1R0.01 2 a dB/km

(8.19)

Step 6: Calculate the horizontal reduction factor, r0.01, for 0.01% of the time: 1 (8.20) LGgR 2L G 1  0.78  0.3811  e 2 A f Step 7: Calculate the vertical adjustment factor, v0.01, for 0.01% of the time: r0.01 

hR¿  hS b deg LGr0.01 LGr0.01 km LR  cos u 1hR¿  hS 2 km LR  sin u x  36  0f 0 deg x  0 deg 1

z  tan1a For z 7 u Else If 0f 0 6 36 Else v0.01 

1  1sin 1u2 a31 11  e1u11x22 2

1LRgR f2

 0.45b

Step 8: The effective path length is LE  LRv0.01 km

(8.21)

Step 9: The predicted attenuation exceeded for 0.01% of an average year is obtained from: A0.01  gRLE dB

(8.22)

Step 10: The estimated attenuation to be exceeded for other percentages of an average year, in the range 0.001% to 5%, is determined from the attenuation to be exceeded for 0.01% for an average year: If p 1% or 0f 0 36°: If p 6 1% and 0f 0 6 36° and u 25°:

b0 b  0.005 1 0f 0  362

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Otherwise: Ap  A0.01 a

323

b  0.005 1 0f 0  362  1.8  4.25 sin 1u2

 10.6550.033 ln 1 p2 0.045 ln 1A0.012 b 11p2 sin 1u22

p dB (8.23) b 0.01 This method provides an estimate of the long-term statistics of attenuation due to rain. When comparing measured statistics with the prediction, allowance should be given for the rather large year-to-year variability in rainfall rate statistics (see Recommendation ITU-R P.678).

EXAMPLE 8.5.2 A Ku-band satellite is to be used in a video broadcasting system. The uplink will be from Miami, Florida, where the studios of the company are located. Since the uplink will be used to feed more than a million home receivers, the uplink availability must be 99.99% in the average year. The question is therefore: what is the rain attenuation on the Miami uplink path for 0.01% of the average year? The information on the link is as follows: Uplink frequency Polarization Coefficients for calculating specific attenuation at 17.80 GHz Rain climate regions for Miami Elevation angle Height of rain hR Height of Miami earth station site a.m.s.l. (above mean sea level)

17.80 GHz Vertical kV 0.0510 V 1.0927 Climate region M 45° Assume 4 km in Miami 0.05 km

Answer Step 1: We already know the rain height (given as 4 km). Step 2: Find LS, the slant-path length below the rain height. LS 

1hR¿  hS 2 sin u

, thus LS  5.5861 km

(Note: Keep all the significant figures at present.) Step 3: Find LG, the horizontal projection of the slant-path length. LG  LS cos u, thus LG  3.95 km Step 4: Find R0.01, the rainfall rate for 0.01% of an average year (mm/h). From the Rain Climatic Zone information (Table 8.2) we have R0.01  63 mm/h. Step 5: Find R, the specific attenuation, along the path for Miami for the rainfall rate encountered at 0.01% of an average year. gR  k1R0.01 2 a,

giving gR  4.7175 dB/km

Step 6: Find r0.01, the horizontal reduction factor for Miami. r0.01 

1 LGgR 1  0.78  0.3811  e2LG 2 A f

Thus r0.01  0.7051 for Miami. Step 7: Calculate the vertical adjustment factor, v0.01, for Miami. To do this we need some intermediate parameters.

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Calculate , where z  tan1 a

hR¿  hs b LGr0.01

Thus,  54.81230 for Miami. This is greater than the elevation angle, , which is 45°. Part (b):

Find LR, an intermediate parameter in calculating the effective path length. Since  , LR 

LGr0.01 cos u

giving LR  3.9388 km. Part (c):

Find , the second intermediate parameter for calculating effective path length. x  36  0f 0

where  is the latitude of the site. Thus   36  25  11.0 for Miami. Finally, calculate v0.01, from v0.01 

1 1  1sin1u2 a31 a1  e1u11x22 b

1LRgR f2

 0.45b

This gives v0.01  1.0332 for Miami. Step 8: Calculate LE, the effective path length for Miami. LE  LRv0.01 which gives LE  4.0696 for Miami. Step 9: Calculate A0.01, the predicted attenuation exceeded for 0.01% of an average year along the path in Miami. A0.01  gRLE and this gives A0.01  19.1983 dB for Miami 1 19.2 dB The rain attenuation on the uplink from Miami for 0.01% of the average year will be 19.2 dB, which is the answer to the question posed. This value, however, pertains to a fixed link that does not change significantly with time. Such a situation would not apply to non geostationary orbit (NGSO) satellite systems. A double-probabilistic approach is required for estimating the statistical impact of rain attenuation on NGSO paths: the probability that attenuation will occur for a given elevation angle and the probability that the satellite will be at that elevation angle. The first approach is documented in ITU-R Rec. 61815 and is abstracted below. 

Calculation of Long-Term Statistics for NGSO Systems For non-GSO systems, where the elevation angle is varying, the link availability for a single satellite can be calculated in the following way: a) calculate the minimum and maximum elevation angles at which the system will be expected to operate; b) divide the operational range of angles into small increments (e.g., 5° wide); c) calculate the percentage of time that the satellite is visible as a function of elevation angle in each increment);

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d) for a given propagation impairment level, find the time percentage that the level is exceeded for each elevation angle increment; e) for each elevation angle increment, multiply the results of c) and d) and divide by 100, giving the time percentage that the impairment level is exceeded at this elevation angle; f) sum the time percentage values obtained in e) to arrive at the total system time percentage that the impairment level is exceeded. In the case of multi-visibility satellite constellations employing satellite path diversity (i.e., switching to the least impaired path), an approximate calculation can be made assuming that the spacecraft with the highest elevation angle is being used.

Scaling Attenuation with Elevation Angle and Frequency Experience has shown that, if long-term attenuation data already exist at a site, it is more accurate to scale measured results to another frequency or another elevation angle, instead of predicting the path attenuation at the new frequency and/or elevation angle from rainfall rate data. Two fairly simple (and surprisingly accurate) rules of thumb exist for scaling over small changes in frequency and elevation angle: (i) For a uniform rainfall rate environment (i.e., stratiform rain) and assuming a “flat earth,” path attenuation in decibels scales with the path length through the rain (i.e., it follows a cosecant law); (ii) Between about 10 and 50 GHz, attenuation in decibels scales as the square of the frequency. These two laws are expanded below.

Cosecant Law The attenuation in decibels at the same frequency at elevation angles El1 and El2 from the same site are approximately related by A1El1 2 cosecant1El1 2  A1El2 2 cosecant1El2 2

(8.24)

This formula breaks down when the elevation angle is low (10°) where its implicit flat earth and uniform rainfall rate assumptions fail to hold. EXAMPLE 8.5.3 A 12-GHz direct broadcast satellite link was found to experience 4 dB of rain attenuation at an elevation angle of 45° for 0.01% of the time in an average year. What would be the rain attenuation measured at the same time percentage for the same site if the elevation angle were 10°? Answer Let suffix 1 in Eq. (8.24) refer to the new elevation angle (i.e., 10°) and suffix 2 to the old elevation angle. Thus, A110°2  3cosecant110°2 cosecant145°2 4  A145°2  3 5.75881.4142 4  4 dB  16.2883 1 16.3 dB The impact of elevation angle on a given link is clear from this example.



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Squared Frequency Scaling Law If A( f1) and A( f2) are the attenuations that would be measured on the same path at frequencies f1 and f2 GHz, they are approximately related by 1 f1 2 2 A1 f1 2  A1 f2 2 1 f2 2 2

(8.25)

This formula relates the long-term statistics (i.e., annual statistics). It should not be used for short-term frequency scaling (i.e., from second to second) on a link or for frequencies that are close to any resonant absorption line. EXAMPLE 8.5.4 A user measures rain attenuation statistics along a satellite link as 6 dB for 0.01% of a year when using a carrier frequency of 10.7 GHz. The satellite operator wants to move the user from the current transponder to a new one, which would change the carrier frequency to 11.4 GHz. What would be the new rain attenuation value, all other link parameters remaining the same? Answer Let suffix 1 in Eq. (8.25) refer to the new frequency (i.e., 11.4 GHz) and suffix 2 refer to the old frequency (i.e., 10.7 GHz). Thus, A111.42  3 111.42 2 110.72 2 4  A110.72  3 129.9600114.4900 4  6 dB  6.8107 1 6.8 dB A more accurate form of frequency scaling can be found in ITU-R Rec. 61815, and is summarized below. 

ITU-R Long-Term Frequency Scaling of Rain Attenuation If A1 and A2 are the equiprobable values of rain attenuation, in dB, at frequencies f1 and f2 (GHz), respectively, the attenuation at frequency f2 can be found from that at frequency f1 from A2  A1 1f2 f1 2 1H 1f1,f2, A12

(8.26)

where f1 f 2 

f2 4

1  10 f 2 H1f1,f2, A1 2  1.12  10 3 1f2 f1 2 0.5 1f1A1 2 0.55

8.6

(8.27) (8.28)

PREDICTION OF XPD Any particle that has spherical symmetry will cause no depolarization of an incident signal. Rain in the atmosphere starts as very small droplets. The surface tension within these droplets is so strong that they retain their spherical shapes. As the drops collide, they coalesce into larger drops. The larger the drop, the more likely it is to distort out of a spherical shape due to wind effects. In convective events, particularly severe thunderstorms, the drops can become relatively large (many millimeters in average diameter) and so they will distort into ellipsoidal forms, generally flattening out in the horizontal axis. Figure 8.22 illustrates the process.

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Very small droplet: wind effects cannot overcome surface tension and droplet remains perfectly spherical in shape. Many droplets have collided and coalesced into larger drop. As drop falls, pressure on underside overcomes surface tension forces and drop begins to distort from spherical shape, taking on elliptical profile in vertical axis. Viewed from below, drop is still essentially circular in cross section. Raindrop has grown in size through further collisions with other raindrops. Larger mass causes raindrop to fall at higher velocity and increased wind forces on underside cause drop to become strongly ellipsoidal in vertical profile. This symmetrical, ellipsoid shape will only exist in very still air and, even then, vertical motion will cause drop to start to oscillate, forming oblate and prolate spheroids alternately. Large, oscillating drops will distort into shapes with no true axes of symmetry. Drops will hollow out underneath due to drop’s fall and, since such large raindrops generally form as result of severe convective activity, turbulent air motion will cause large, oscillating drops to break up. Smaller drops resulting from this breakup then cause additional raindrop formation to occur. FIGURE 8.22 Schematic of the shape of an individual raindrop from formation to maturity.

If all of the ellipsoidal drops in a rainstorm were aligned, then waves propagating with their electrical field vectors parallel to the raindrops’ minor axes (for all practical purposes, vertically polarized waves) would experience the minimum attenuation for that rainfall rate, and waves propagating with their electric field vectors parallel to the major axes (i.e., horizontally polarized waves) would experience the maximum attenuation. In these two special cases, no depolarization would occur. The difference between the attenuations experienced by waves with horizontal and vertical polarization is small—rarely greater than a decibel. It is called the differential attenuation. In a like manner, waves with horizontal and vertical polarization can experience differential phase shift as they pass through an anisotropic medium. At frequencies below about 10 GHz, differential phase shift is the more important phenomenon. At frequencies above about 30 GHz, differential attenuation is more important. Between 10 and 30 GHz, either differential phase or differential attenuation will be the major effect, depending on the elevation angle of the link and the climate31. Imagine now the case of a wave whose linear polarization is intermediate between horizontal and vertical. We can resolve this wave into its vertically polarized and horizontally polarized components as in Figure 8.23. These components propagate through the rain with their polarizations unchanged, but the horizontal component is attenuated more than the vertical component. If at any point we recombine the vertical and horizontal components to reconstruct the wave, we find that its polarization has rotated toward the vertical and a cross-polarized component is now present. This process is a simplification of a complicated problem in electromagnetic wave scattering. For details of the process, the reader should consult the extensive publications of T. Oguchi, the pioneer researcher in the field (e.g., [26]). Depolarization, while it is dependent to a great extent on the volume of rain that is present in the path, the shape of the raindrops in the path and the orientation of their major and minor axes also significantly affect it. The orientation will have two independent features: one that is due to the rain medium, and is referred to as the canting angle; and one that is due to the path geometry, and is referred to as the tilt angle.

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Vertical

Ei

E iV Er

E rV

θ

E rH

E iH

Horizontal

FIGURE 8.23 A simplified explanation of rain depolarization based on a drop with an elliptical cross section. An incident electromagnetic wave with electric field vector Ei strikes a raindrop. We resolve it into a horizontal component E iH and a vertical component E iV. The horizontal component is attenuated more than the vertical component because it encounters more water. Thus, when we recombine the horizontal and vertical field components E rH and E rV that arrive at the receiver, we find that the received wave Er has had its polarization rotated toward the vertical by the angle .

Canting Angle Falling raindrops orient themselves so as to minimize the aerodynamic forces. In steady fall, the minor axis of the drop is parallel to the net wind force and so their major axis is horizontal when the raindrop is falling in still air. Under windy conditions, the aerodynamic force will have two components: one due to the raindrop fall velocity (i.e., vertical) and one due to the prevailing wind direction (i.e., horizontal). The resultant of these two forces will lead to the raindrops’ major axis being canted out of the usual horizontal orientation. The prevailing wind speed lessens with altitude, becoming zero at the ground. The raindrop orientation will therefore vary with altitude. Since the horizontal wind direction with respect to the path varies, the net horizontal component measured over a long interval will be close to zero. The canting angle will therefore have a mean of zero. In any given rainstorm, however, the canting angle will have a finite probability of being nonzero, thus leading to enhanced depolarization for horizontal or vertical polarized waves over short time intervals. Figure 8.24 illustrates the canting angle process schematically.

Tilt Angle The tilt angle refers to the angle between the local horizontal (or vertical) and the actual orientation of the electric field vector of the transmitted signal. The orientation of the electric field vector transmitted by a geostationary satellite is referenced to the equator at the subsatellite point. Horizontal polarization is parallel to the equator and vertical polarization is perpendicular to the equator. An earth station that lies on the same longitude as the GEO satellite (say, to the north) would receive signals polarized in the local vertical direction if the satellite is transmitting a vertically polarized signal. If the location of the earth station is moved either east or west from the longitude of the GEO satellite, the vertically polarized signal transmitted by the satellite is now received out of the local vertical at the earth station. That is, the polarization vector would appear to be tilted away from the original orientation. The process is illustrated in Figure 8.25.

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Canting angle

Raindrop

Local horizontal

Vertical wind force due to drop velocity

Net wind force

Horizontal wind force due to prevailing wind FIGURE 8.24 Illustration of canting angle. The resultant of the prevailing wind force and the wind force due to the raindrop fall velocity leads to a net wind force that is out of the vertical direction. The raindrop, which has already distorted into an ellipsoid due to the wind force induced by the drop velocity, will now orient itself to minimize drag forces. This means that the raindrop will cant out of the horizontal and orient its minor axis to be parallel to the net wind force. N Earth as viewed from GEO

S

Equator

(a )

GEO arc as viewed from earth station

Orientation of local electric field vector of vertically polarized transmissions from GEO satellite

Local horizontal (b )

FIGURE 8.25 Schematic of tilt angle. In (a) above, S, is the subsatellite point of a GEO satellite. Transmissions from the satellite will be horizontally polarized if they are parallel to the equatorial plane. Vertically polarized transmissions will be orthogonal to the equatorial plane. If an earth station were on the satellite longitude (here shown by the broken line SN) it would receive the polarization vector in the orientation transmitted—although the polarization would be undefined at the subsatellite point. In (b) above, the earth station is not on the equator. The arc shows how the GEO orbit would look from the earth station. In this instance, the satellite is transmitting a vertically polarized signal. The orientation of the vertically polarized transmissions may not be received at the local vertical, however. The local orientation will depend on where the satellite is located on the GEO arc as seen by the earth station. The polarization vector may therefore be tilted out of the transmitted orientation by virtue of the link geometry. The polarization will only be vertical (or horizontal) at the earth station site to a GEO satellite if the azimuth to the satellite is 0° or 180° from true north.

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A simple equation that gives the tilt angle with respect to the horizontal, assuming the transmissions from a GEO satellite are polarized in the north–south direction, is27 t  arctan1tan Lesin b2 degrees

(8.29)

where Le is the earth station latitude (positive for the Northern Hemisphere and negative for the Southern Hemisphere) and is the satellite longitude minus the earth station longitude (i.e., L s  Le), with longitude expressed in degrees east. EXAMPLE 8.6.1 What is the perceived polarization tilt angle at an earth station located at 52° N, 1° E, for vertically polarized signals transmitted from a GEO satellite located at 60° E? Answer:

Using Eq. (8.29)

t  arctan1tan 52 sin360  14 2  arctan11.27990.85722  arctan11.49322  56.19° The ITU-R XPD prediction method15 is based on the attenuation measured (or predicted) at the frequency of interest, plus additional terms to take account of the canting angle distribution, the tilt angle, and ice crystal depolarization28. The stepby-step procedure is summarized below. Step 1: Calculate the frequency-dependent term: Cf  30 log f

for 8  f  35 GHz

(8.30)

where f is the frequency in GHz. Step 2: Calculate the rain attenuation dependent term: CA  V1 f 2 log Ap

(8.31a)

where Ap is the rain attenuation in decibels exceeded for the required percentage of time, p, for the path in question, commonly called the copolar attenuation or CPA; V1 f 2  12.8f 0.19 V1 f 2  22.6

for 8  f  20 GHz for 20 6 f  35 GHz

(8.31b) (8.31c)

Step 3: Calculate the polarization improvement factor: Ct  10 log31  0.48411  cos 4t2 4

(8.32)

where is the tilt angle. The improvement factor C  0 for  45° and reaches a maximum value of 15 dB for  0° or 90°. The value  45° corresponds to circular polarization. Step 4: Calculate the canting angle dependent term: Cu  40 log1cos u2

for u  60°

(8.33)

where  is the elevation angle of the link. Step 5: Calculate the canting angle dependent term: Cs  0.0052 s2

(8.34)

where is the effective standard deviation of the raindrop canting angle distribution, expressed in degrees. The value of is 0°, 5°, 10°, and 15° for 1, 0.1, 0.01, and 0.001% of the time, respectively. Step 6: Calculate rain XPD not exceeded for p% of the time: XPDrain  Cf  CA  Ct  Cu  Cs dB

(8.35)



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Step 7: Calculate the ice crystal dependent term: Cice  XPDrain  10.3  0.1 log p2 2 dB

(8.36)

Step 8: Calculate the XPD not exceed for p% of the time, including the effects of ice crystals: XPDp  XPDrain  Cice dB (8.37) The rain attenuation below 8 GHz is fairly low and so the attenuation-dependent XPD prediction method does not provide accurate results. To calculate XPD for frequencies below 8 GHz, it is best to calculate the XPD at 8 GHz and then scale in frequency to the required frequency using15 XPD2  XPD1  20 log c

f2 11  0.48411  cos 4t2 2

f1 11  0.48411  cos 4t1 2

d

for 4  f1, f2  30 GHz (8.38)

Unpublished results from the Italsat experiment29 appear to show that it is possible to predict XPD from 35 GHz up to 50 GHz by amending Eq. (8.30) and changing the values of V( f ) in Eq. (8.31a) to Cf  26 log f V1 f 2  20

(8.39a) (8.39b)

EXAMPLE 8.6.2 What is the value of XPD at 0.01% of the time for a 12-GHz link that experiences 7-dB attenuation for this period of time? The elevation angle is 30°. Calculate the XPD for tilt angles of 20° and 0°. Answer

Using the step-by-step procedure we have:

Step 1: Cf  30 log f  32.3754 Step 2: V( f )  12.8 f 0.19  12.8  1.6034  20.5236 CA  V( f ) log Ap  20.5236  log (7)  17.3445 Step 3: Tilt angle of 20° C  10 log [1  0.484 (1  cos 4 )]  10 log[1  0.484(1  cos 80)]  3.6456 Tilt angle of 0° C  10 log[1  0.484 (1  cos 4 )]  10 log[1  0.484(1  cos 0)]  14.9485 Step 4: C  40 log(cos )  40 log(cos 30)  40 log(0.8660)  2.4988 Step 5: C  0.0052 2  0.0052 102  0.52 Step 6: XPDrain  Cf  CA  C  C  C For  20°  32.3754  17.3445  3.6456  2.4988  0.52  21.6953  21.7 dB For  0°  32.3754  17.3445  14.9485  2.4988  0.52  32.9982  33.0 dB Step 7: Cice  XPDrain  (0.3  0.1 log p)2 For  20°  21.6953  (0.3  0.1 log p)2  21.6953  (0.3  0.1 log 0.01)2  21.6953  (0.3  0.2)2  1.0848 For  0°  32.9982  (0.3  0.1 log p)2  32.9982  (0.3  0.1 log 0.01)2  32.9982  (0.3  0.2)2  1.6499 Step 8: XPDp  XPDrain  Cice For  20°  21.6953  1.0848  20.6105  20.6 dB For  0°  32.9982  1.6499  31.3483  31.3 dB Note: The single, best way to reduce depolarization is to operate with polarization senses that are linear vertical or horizontal as perceived by the receiving antenna. This can be seen from the very different results calculated in the above example when the tilt angle was 0° (i.e., the signal is being received in linear, horizontal polarization) compared with those when the tilt angle is 20°. 

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Ice Crystal Depolarization The calculation procedure for ice crystal depolarization incorporated in the calculation of XPD has been found to have wide variations in accuracy. At high elevation angles and at frequencies below 10 GHz, the procedure tends to agree with measured data. That is, ice crystal depolarization occurs only in severe thunderstorms and so it is a rare occurrence. However, on low elevation angle paths, the contribution due to ice crystals has been observed to occur for quite high time percentages. At frequencies above 30 GHz, it is expected that ice crystal depolarization will be a significant effect, particularly at elevation angles below 30°.

Rain Effects on Antenna Noise At frequencies below about 50 GHz, rain attenuation is mostly caused by absorption rather than by scattering of the signal energy out of the path. Any absorber with a physical temperature greater than absolute zero (0 kelvins) will act as a black body radiator. At frequencies below 300 GHz, the radiation is in the form of white Gaussian noise with a noise power given by kTB, where T is the equivalent noise temperature of the absorber. Raindrops are absorbers at microwave frequencies and, when the raindrops fall through the antenna beam, some of their isotropically radiated thermal energy will be detected by the receiver (see Chapter 4). Rain will therefore cause not only signal attenuation and depolarization; it will also cause an increase in sky temperature, which, in turn, will increase the overall system noise temperature. The impact of the increase in sky noise temperature can be high for low noise receiving systems at Ku band, as illustrated in the examples in

Absorbing medium (e.g., a rain cloud) with effective temperature Tm kelvins

Temperature radiated by medium, Tr , is Tr = (1 − σ )Tm kelvins

FIGURE 8.26 Schematic of the additional radiated sky temperature due to absorption in rain. The added temperature received by the antenna due to radiation from the “hot” rainstorm will cause an additional component to be added to the system noise temperature. This additional component is similar to the noise temperature contribution from a lossy feed. In Chapter 4, in the analysis of system noise temperature, a noise temperature contribution due to signal loss, Tl, was calculated using a “gain” component Gl, where Gl was a linear value. For example, when the component at a physical temperature of 280 K caused a loss of 2 dB (which  1/1.58  0.63 of the original value), Tl  Tp(1  Gl)  280(1  0.63)  103.6 K. The parameter Gl is identical to , i.e. a loss of 2 dB is the same as a fractional transmission of 0.63 of the original signal.

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Chapter 4. Rain attenuation in the 1 to 3 dB range can cause the system noise level to increase by 1 to 3 dB, leading to a reduction in CN ratio (in dB) in rain, which is twice the rain attenuation value. The increase in antenna noise temperature due to rain, Tb, may be estimated by Tb  280 11  eA4.34 2 K

(8.40)

where A is the rain attenuation in decibels and the value 280 K is an effective temperature of the rain medium in kelvins. Values between 273 and 290 K may be used, depending on whether the climate is cold or tropical. An alternative approach is to treat the rain as a passive attenuator with a fractional transmission coefficient of . If the rain totally attenuates the signal,  0; if the rain medium is completely transparent and no attenuation takes place,  1. Figure 8.26 illustrates the process. EXAMPLE 8.6.3 What is the additional noise temperature contribution of an antenna compared with that in clear sky when there is 4 dB of rain attenuation in the path? You may assume that the rain medium is at a temperature equivalent to 285 K. Answer An attenuation of 4 dB causes the signal to be reduced by a factor of 2.5119. The fractional transmission coefficient, , would therefore be 12.5119  0.3981. (Another way of looking at this is to say that only 39.81% of the original signal power is being received during the 4-dB rain event.) The additional sky temperature radiated would therefore be 285 (1  0.3981)  171.5395 K  171.5 K. Note that, if the system noise temperature had been 200 K, the effective system noise temperature is now 200  171.5  371.5 K. In other words, the signal power has decreased by 4 dB and the noise power has increase by 2.7 dB. A 4-dB rain attenuation has thus led to a 6.7 dB reduction in CN. This is somewhat simplistic, since the receiving antenna efficiency is not 100%, and it therefore does not accept all of the radiation that is incident upon it. However, the enhanced sky noise contribution received by the antenna during rain conditions will be close to that radiated by the rainstorm. Careful attention must be paid in the system design to allow for enhanced sky noise contributions as well as signal degradations when developing link budgets. Put another way, the key in link budget calculations is to find the change in carrier-to-noise, CN, rather than just the change in carrier power, C. 

8.7 PROPAGATION IMPAIRMENT COUNTERMEASURES Attenuation Many research groups have investigated the use of fade countermeasures. Fade mitigation has been shown30 to fall into three main classes: • Power control (i.e., varying the EIRP of the signal to enhance CN) • Signal processing (i.e., changing the parameters of a signal to improve BER) • Diversity (i.e., choosing a different path or time to take advantage of decorrelated fading) Interestingly, the three main classes of fade mitigation affect a link differently and are complementary in nature30. For satellite systems that use frequencies at Ka band and above,

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all three classes of fade mitigation techniques might be required for high availability links. We will look briefly at each technique.

Power Control In adaptive power control, the transmitter power is adjusted to compensate for changes in signal attenuation along the path. At its simplest, it is like automatic gain control in a receiver, which adjusts to fluctuations in the received energy so as to hold the receiver output constant. Many satellite links are operated such that the uplink is the critical portion of the connection; that is, the first part of the overall connection that will drop out in a rain fade is the uplink. The overall availability (and performance) of the connection is therefore enhanced if the uplink operates with an increased EIRP in rain. This is referred to as up-link power control (ULPC). ULPC can operate closed loop, where the signal power is detected at the satellite and a control signal sent back to the earth station to adjust the power, or open loop, where the fade on the downlink signal is used to predict the likely fade level occurring on the uplink. Closed-loop operation is always more accurate but is more expensive to implement; hence most ULPC systems are, at present, open loop. Open loop ULPC becomes more difficult the further apart the downlink and uplink frequencies are. It becomes even more difficult at Ka band when the downlink (20 GHz) and uplink (30 GHz) frequencies are on either side of the 22-GHz water vapor absorption line. The ratio of 30-GHz attenuation to 20-GHz attenuation is less than 1 for 20-GHz attenuation values of less than 1.0 dB, since cloud attenuation (i.e., essentially water vapor absorption) is higher at 20 GHz than at 30 GHz due to the proximity of the 22-GHz water absorption line to the 20-GHz downlink. Figure 8.27 gives the average 30 :20 GHz attenuation ratios, with uplink attenuation as parameter. Note that the long-term 30 :20 GHz

4 (30/20) GHz attenuation ratio

334

3

2

1

2

6

10

14

18

22

30 GHz attenuation, dB FIGURE 8.27 Instantaneous 30:20 GHz attenuation scaling ratio with 20-GHz attenuation as parameter. The solid curve above is a prediction of the scaling ratio that takes into account both rain attenuation and tropospheric scintillation. The pair of broken curves are the bounds of individual instantaneous measurements of the uplink and downlink attenuation values. The large range of scaling ratios shows that great care must be taken in developing open loop ULPC algorithms that use only a measure of the amplitude of the downlink frequency.

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attenuation scaling ratio does not become established until the uplink attenuation is above 7 dB. Another major consideration is power flux density variations at the satellite. If many earth stations are operating under rain fade conditions with the same satellite, as could happen in a VSAT network with many hundreds of thousands of earth stations, implementing ULPC can lead to significant received power fluctuations at the satellite, and this has capacity implications. Some of the advanced Ka-band satellites with multiple switched beams can also implement downlink power control, if sufficient bandwidth and power are available.

Signal Processing The move from very large earth stations (e.g., the Intelsat Standard A) to a multiplicity of small earth stations has been accompanied by a shift in the median traffic stream. It is rare to find a non-video or non-Internet network distribution link via a satellite at a rate of more than 2 Mbit/s. The need to make small traffic streams economic by using VSATs has led to the introduction of onboard processing (OBP) techniques. This process typically translates the digital carriers arriving at the satellite to baseband for processing and onward transmission back to earth. The process is generically called MCDDD (multicarrier demodulation, demultiplexing, and decoding). The OBP process is carried out at baseband and allows each individual traffic packet to be switched to the correct output port of the satellite antenna for transmission down to earth following recoding and remultiplexing. By detecting the signal level of each packet on arrival at the OBP, not only can most bit errors be removed but the transmitting earth station can also be alerted if the energy level of the received packet has fallen, so that ULPC can be used at the earth station to correct the signal level (within the power level range of the ULPC system). The use of OBP separates the uplink from the downlink and each part of the link can be treated separately in developing a link budget.

Diversity Many diversity schemes have been proposed, but few have been implemented as yet due to the cost. If OBP techniques are being used on the satellite, a form of time diversity can be used. In this approach, additional slots in the TDMA frame can be assigned to the rainaffected link so that the same signal can be sent at a slower rate, essentially lowering the bandwidth and raising the CN. The FEC rate could also be changed in the OBP payload. If the satellite operates in a number of frequency bands (e.g., C band and Ku band), a rain affected Ku-band link could be switched to C band, which is not attenuated significantly by rain. To be able to do this, spare C-band capacity must be held in reserve on the satellite so that it can be used when required. Similarly, each Ku-band earth station would need to have a dual-band antenna and receiving system so that they could switch between the two bands. The added cost has not justified this approach to date. However, the V-band systems in design at present may find it economic to include a low capacity Ka-band or Ku-band payload to use in those traffic streams that are the highest priority. Of all the diversity schemes, that of site diversity appears to offer the most significant gain in availability. Site diversity is a technique whereby two, or more, earth stations are located sufficiently far apart to ensure that the rain impairments observed at each of the stations are generally uncorrelated. More exactly, it is the paths through the rain that are uncorrelated and so the technique is more accurately described as path diversity. The earth stations are

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connected together so that any one earth station can be used to support the traffic stream while the other(s) is (are) suffering a rain fade. If we assume that there are two earth stations, identified by suffixes 1 and 2, which are operated in a site diversity mode, then the joint attenuation AJ(t) is defined by AJ 1t2  minimum3A1 1t2, A2 1t2 4 dB

(8.41)

The average single-site attenuation AS(t) is the mean of A1(t) and A2(t), namely AS 1t2  3A1 1t2  A2 1t2 4 2 dB

(8.42)

An ideal system that monitors the received downlink signals at both sites and always selects the stronger of the two experiences an attenuation of AJ(t) and the diversity system would perform better than either site alone. How much better is measured by two statistical quantities, diversity gain and diversity improvement. Diversity gain, GD(P), is the decibel difference between the average single-site attenuation AS(P) equaled or exceeded P% of the time and the joint attenuation AJ(P) equaled or exceeded P% of the time. GD 1P2  AS 1P2  AJ 1P2 dB

(8.43)

Diversity improvement ID(A) is the ratio between the percentage of time PS that the average single-site attenuation AS exceeds A dB to the percentage of time PJ that the joint attenuation AJ exceeds A dB. ID 1A2 

PS 1A2 PJ 1A2

(8.44)

Diversity gain determines system margin, and it is the measure of diversity system performance that we will use here. In addition, diversity gain has been shown to be stable from year to year and, as such, is a reliable statistic to use in system design. Diversity improvement, on the other hand, is extremely variable from year to year. Figure 8.28 illustrates these two concepts. The first, and still the best, diversity gain model is that due to Hodge32, which has been adapted by the ITU-R in Rec. 61815. Hodge developed the diversity gain model through an iterative analysis of diversity data available. Intuitively, he assumed site separation was the key element. In this, he has been proved correct. The procedure is abstracted below. Step 1:

Calculate the gain contributed by the spatial separation of the two earth stations from Gd  a11  ebd 2

(8.45)

where d is the separation (km) between the two sites, a  0.78A  1.94 11  e0.11A 2 b  0.5911  e0.1A 2

Step 2:

and A  path attenuation (dB) for a single site. Calculate the frequency-dependent gain from: Gf  e0.025 f

Step 3:

(8.46)

where f  frequency (GHz). Calculate the gain term dependent on elevation angle from: G u  1  0.006 u where   path elevation angle (degrees).

(8.47)

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Percentage of time attenuation exceeds A

100

ID(A) =

P S (A ) P J (A )

P S (A ) P J (A ) 10

G D (P ) = A S (P ) − A J (P )

1.0

G D (P )

P Average single-site

Joint 0.1

A (dB)

A A J (P )

AS(P )

FIGURE 8.28 Illustration of diversity gain and diversity improvement (diversity advantage). At a given percentage of time, P, the diversity gain GD(P) is the decibel difference between the average single-site attenuation exceeded AS(P) and the joint attenuation exceeded AJ(P). At a given attenuation, A, the diversity improvement ID(A) is the ratio of the percentage of time PS(A) that the single-site attenuation exceeds A to the percentage of time PJ(A) that the joint attenuation exceeds A.

Step 4:

Calculate the baseline-dependent term from the expression: Gc  1  0.002 c

Step 5:

(8.48)

where   angle (degrees) made by the azimuth of the propagation path with respect to the baseline between sites, chosen such that   90°. Compute the net diversity gain, G, as the product: G  Gd  Gf  Gu  Gc dB

(8.49)

The use of a site diversity system is very expensive if traditional approaches are used. That is, two large earth stations connected together via a very high-speed terrestrial link. It has only been used operationally to date by the gateway stations of the Iridium network. These gateway earth stations operate in Ka band and are single-point failures for the network. As such, the expense of a diversity setup was well justified. Another proposed approach to site diversity has been to use wide area diversity33, in which a multitude of VSATs are linked via routers to a metropolitan area network.

Depolarization Depolarization compensation is a technique whereby the feed system of the antenna is adjusted in such a way as to correct for depolarization in the path. Alternatively, the

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orthogonal channels may be cross-coupled in the receiver and, provided good samples of the signal in each channel can be obtained, the interfering (i.e., depolarized) signal may be removed by subtracting the correct amount of signal. Few earth stations have implemented depolarization compensation, as it is an expensive undertaking. Those earth stations that have implemented depolarization compensation have done so at C band. At C band, differential phase is the primary cause of depolarization. As the frequency increases, differential attenuation becomes an increasingly significant cause of depolarization until, at V band, differential attenuation dominates completely. For this reason, the amount of rain depolarization observed for each dB of rain attenuation on commercial communications satellite links is largest at C band, reducing monotonically to V band. Most Ka-band systems for direct-to-home (DTH) Internet services have rain margins of less than 10 dB. At this attenuation level, depolarization effects are not significant.

8.8

SUMMARY

The design of radio systems includes a link margin that is intended to provide for changes in the received signal level due to both equipment effects and random changes in the environment between the transmitter and the receiver. The link margin permits the communications system to operate with both the required performance, a measure of the service quality required for a significant fraction of the time, and availability, a measure of the time period when usable service is provided. Developing an adequate link margin is critical to the acceptance of the service. However, each additional dB of link margin that is provided comes with a cost associated with it. A lot of care, therefore, goes into developing an accurate estimate of the likely impairments on any given link that would cause the performance and availability of the service to fall below acceptable levels. A key to this estimate is an understanding of the propagation effects along the path between the satellite and the earth station. Propagation effects cause two principal phenomena to be observed at the receiving terminal: a change in the wanted signal level, which is referred to as signal attenuation or fading; and a change in the unwanted signal level, which is referred to as depolarization or cross-polarization. Attenuation and depolarization effects are a function of the signal frequency, the atmospheric conditions, and the path geometry. In general, the higher the frequency, the warmer and wetter the weather, and the lower the operating elevation angle of the earth station, the worse the propagation effects are. The only time this is not true is for ionospheric effects, where the effects on commercial satellite systems are only of significance at C band or below.

With the exception of ionospheric effects, propagation phenomena are weather dependent. To overcome this problem, statistical models are used. Longterm measurements of rainfall rate are statistically related to long-term path attenuation measurements when taken over the same period and at the same site. In this case, long-term is at least 1 year so that all of the seasons normally experienced in a given year may be included. The prediction of rain attenuation has taken two distinct paths: one uses measured data and develops an empirical model to predict the phenomenon on a worldwide basis; the other attempts to model the physics of the process. Statistical models of rain depolarization, tropospheric scintillation, gaseous losses, cloud attenuation, low angle fading, and related propagation effects have been developed. Most of these models provide usable predictions for frequencies between 4 and 50 GHz, but care must be taken when predictions for unusual path geometries (e.g., 5°) or severe climates (e.g., tropical regions) are required. More recently, the impact of individual rain fades—their occurrence statistics, duration of individual events, time between fades of the same level— has become important for developing user perception for direct-to-home (DTH) services. Counter-measures to rain fades may take many forms—for example, increasing the TDMA frame allocation, changing the modulation index, changing the power level, changing the frequency—and it is likely that some of them will be included in the Ka-band DTH services planned for the first decade of the twenty-first century.

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REFERENCES 1. L. J. IPPOLITO, Propagation Effects Handbook for Satellite Systems Design, ITT Industries, Advanced Engineering and Sciences, Ashburn, Virginia. Developed for NASA as an update of Reference Publication 1082(04), National Aeronautics and Space Administration, Washington, DC, September 2000. 2. J. E. ALLNUTT, Satellite-to-Ground Radiowave Propagation, ISBN 0 86341 157 6, Peter Peregrinus, 1989 [2nd Ed. in preparation]. 3. J. E. ALLNUTT, “Refraction and Attenuation in the Troposphere,” Wiley Encyclopedia of Electrical and Electronics Engineering, John G. Webster, ed., ISBN 0 471 13946 7, Vol. 18, pp. 379–388, 1999. 4. A. W. DISSANAYAKE, J. E. ALLNUTT, and F. HAIDARA, “A Prediction Model that Combines Rain Attenuation and Other Propagation Impairments along Earth–Satellite Paths,” IEEE Transactions on Antennas and Propagation, Vol. 45, No. 10, October 1997, pp. 1546–1558. 5. M. A. B. TERADA, “Reflector Antennas,” Wiley Encyclopedia of Electrical and Electronics Engineering, JOHN G. WEBSTER ed., ISBN 0 471 13946 7, Vol. 18, pp. 360–379, 1999. 6. D. C. COX and H. W. ARNOLD, “Comparison of Measured Cross-Polarization Isolation and Discrimination for Rain and Ice on a 19 GHz Space–Earth Path,” Radio Science, 19, 617–628, March–April 1984. 7. P. N. KUMAR, “Depolarization of 19 GHz Signals,” Comsat Technical Review, 12, 271–293, Fall 1982. 8. ACTS experimental results, presented in a series of proceedings; e.g., Proceedings of the Twenty-third NASA Propagation Experimenters Meeting (NAPEX XXIII) and the Advanced Communications Technology Satellite (ACTS) Propagation Studies Workshop, Nasser Golshan and Christian Ho, eds., Falls Church, Virginia, June 2–4, 1999. For updated information, see the NASA web site. 9. OLYMPUS Propagation Experiment (OPEX), Second Workshop of the Olympus Propagation Experimenters, WPP-083, Volumes 1 (Attenuation Measurement and Prediction), 2 (Depolarization), 3 (Radiometry and Meteorological Measurements), 4 (Radar), and 5 (Data Processing), Noordwijk, The Netherlands, 8–10 November 1994. 10. P. A. WATSON, and M. ARABI, “Cross-Polarization Isolation and Discrimination,” Electronics Letters, 9, November 1973, 516–519. 11. Rec. ITU-R P.676-3, Attenuation by Atmospheric Gases, 1997. 12. A. W. DISSANAYAKE, J. E. ALLNUTT, and F. HAIDARA, “A Prediction Model that Combines Rain Attenuation and Other Propagation Impairments along Earth–Satellite Paths,” IEEE Transactions on Antennas and Propagation, Vol. 45, No. 10, pp. 1546–1558, October 1997. 13. E. SALONEN, and S. UPPALA, “New Prediction Method of Cloud Attenuation,” Electronics Letters, Vol. 27, No. 12, pp. 1106–1108, 1991.

14. A. W. DISSANAYAKE, J. E. ALLNUTT, and F. HAIDARA, “Cloud Attenuation Modeling for SHF and EHF Applications,” Invited paper, Special issue of International Journal of Satellite Communications, revised and accepted, January 2000. 15. Rec. ITU-R P.618-6, Propagation Data and Prediction Methods Required for the Design of Earth–Space Telecommunications Systems, 1999. 16. T. PRATT, and D. J. BROWNING, “Co-polar Attenuation and Radiometer Measurements at 30 GHz for a Slant Path to Central England,” Proceedings ATS-6 Meeting (ESA SP-131), ESTEC, Noordwijk, The Netherlands, 15–20, October 1977. 17. E. C. JOHNSTON, D. L. BRYANT, D. MAITI, and J. E. ALLNUTT, “Results of Low Elevation Angle 11 GHz Satellite Beacon Measurements at Goonhilly,” IEE Conference Publication No. 333, ICAP 91, pp. 366–369, April 1991. 18. K. MURSULA, and T. ULICH, “A New Method to Determine the Solar Cycle Length,” Geophysic Research Letters, 25, 1837–1840, 1998. 19. Rec. ITU-R PN.837-1, Characteristics of Precipitation for Propagation Modelling, 1994. 20. The general web site for the ITU is http://www.itu.int For ITU model software see http://www.itu.int/brsg/sg3/ databanks/tropospheric.html 21. Report 564 of the Recommendations and Reports of the CCIR, Volume V, Propagation in Non-Ionized Media, International Telecommunications Union (ITU), Geneva, Switzerland, 1982. 22. Rec. ITU-R P.837-2, Characteristics of Precipitation for Propagation Modelling, 1999. 23. M. P. M. HALL, Effects of the Troposphere on Radio Communication, Peter Peregrinus, Ltd., Stevenage, UK, 1979. 24. R. L. OLSEN, D. V. ROGERS, and D. B. HODGE, “The aRb Relation in the Calculation of Rain Attenuation,” IEEE Transactions on Antennas and Propagation AP-26, 318–329, March 1978. 25. Rec. ITU-R P. 838, Specific Attenuation Model for Rain for Use in Prediction Methods, 1992. 26. T. OGUCHI, “Electromagnetic Wave Propagation and Scattering in Rain and Other Hydrometeors,” Proceedings of the IEEE, 71, 1029–1078, September 1983. 27. J. E. ALLNUTT, and D. V. ROGERS, “System Implications of 1411 GHz Path Depolarization. Part II: Reducing the Impairments,” International Journal of Satellite Communications, 4, 13–17, 1986. 28. C. W. BOSTIAN, and J. E. ALLNUTT, “Ice Crystal Depolarization on Satellite-to-Earth Microwave Radio Paths,” Proceedings of the IEE, 126, 951–960, 1979. 29. F. BARBALISCIA, A. PARABONI, and C. RIVA, Private communication, December 1999. 30. L. CASTANET, J. LEMORTON, and M. BOUSQUET, “Fade Mitigation Techniques for New SatCom Services at KuBand and Above: A Review,” COST 255 First International Workshop on Radiowave Propagation Modelling

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for SatCom Services at Ku-Band and Above, WPP-146, 243–251, October 1998. 31. (a) D. V. ROGERS and J. E. ALLNUTT, “System Implications of 1411 GHz Path Depolarization. Part I: Predicting the Impairments,” International Journal of Satellite Communications, Vol. 4, No. 1, pp. 1–12, 1986. (b) J. E. ALLNUTT and D. V. ROGERS, “System Implications of 1411 GHz Path Depolarization. Part II: Reducing the Impairments,” International Journal of Satellite Communications, Vol. 4, No. 1, pp. 13–18, 1986. 32. D. B. HODGE, “An Empirical Relationship between Path Diversity and Gain,” IEEE Transactions on Antennas and Propagation, AP-24, 250–251, March 1976; D. B. HODGE, “An Improved Model for Diversity Gain on Earth–Space Paths,” Radio Science, 17, 1393–1399, November–December 1982.

33. J. O. LAWS and D. A. PARSONS, “The Relation of RainSize to 10Intensity,” Trans. Amer. Geophys. Union, Vol. 24, pp. 432–460, 1943. 34. J. E. ALLNUTT, and B. ARBESSER-RASTBURG, “Low Elevation Angle Propagation Modeling Considerations for the INTELSAT Business Service,” ICAP 85, IEE Conf. Publ. 248, 62–66, 1985. 35. R. K. CRANE, “Prediction of Attenuation by Rain,” IEEE Transactions on Communications, Vol. COM-28, No. 9, pp. 1717–1735, September 1980. 36. W. L. STUTZMAN and W. K. DISHMAN, “A Simple Model for the Estimation of Rain Induced Attenuation along Earth–Space Paths at Millimeter Wavelengths,” Radio Science, 17, 1465–1476, November–December 1982.

PROBLEMS 1. a. For a typical satellite-to-ground link, give one propagation impairment that has less impact on the average signal level as the frequency increases from 1 to 30 GHz and give one propagation impairment that has a greater impact on the average signal level as the frequency increases from 1 to 30 GHz. b. Ionospheric scintillation effects are variable in both time and location. (i) What causes the time variability? (ii) Over what periodic interval of time do we expect to observe the worst effects repeating on a given satellite link? (iii) In any given year, when would the worst ionospheric effects be observed? (iv) On any given day, when would the worst ionospheric effects be observed? (v) Within what latitude range on the earth’s surface are these effects the worst for geostationary satellite links? 2. A DTH (direct-to-home) satellite ISP (Internet service provider) consortium is designing a system to provide digital multimedia service to CONUS (CONtinental United States) coverage. They are investigating where in the United States to locate the uplink transmitter for their service. The uplink to the Internet/Multimedia satellite will be at 30 GHz and the downlink to the user terminals will be at 20 GHz. a. Using Table 8.2 and Figure 8.15, determine the approximate region within CONUS where you would expect to obtain the lowest outage time due to rain for the uplink transmitter in the Internet/Multimedia satellite system. b. For time percentages of 0.01 and 0.001% of a year, what are the specific attenuations observed at the site you have selected in part (a), assuming horizontal polarization is employed? (Hint: you will need to refer to Table 8.3 for this part of the problem.)

c. If the elevation angle of the transmitting earth station is 50°, it is 200 m above mean sea level, and the height of the rain hR is 4.5 km, what are the predicted values of rain attenuation measured for time percentages 0.01 and 0.001% of a year for the uplink transmitter? 3. A simple procedure to calculate the effect of changing the elevation angle is to use the cosecant law [see Eq. (8.24)]. To calculate the effect of changing the frequency, either a simple formula may be used [see Eq. (8.25)] or a more complicated procedure may be used [see Eq. (8.26) et seq.]. A DTH satellite ISP consortium is investigating whether it should change location and frequency for its uplink transmitter. Using their currently allocated frequency (30 GHz) on the uplink, they have observed an attenuation of 30 dB at an elevation angle of 50° at the time percentage of interest to their service. They have been notified that they would be permitted to use an uplink frequency at 18 GHz, but they would have to relocate their earth station. The rain climate is the same as their current earth station, but the elevation angle would now be 15° instead of 50°. a. Using the simple frequency scaling formula [Eq. (8.25)], what is the attenuation that would be observed at a frequency of 18 GHz if an attenuation of 30 dB was observed at 30 GHz (both at an elevation angle of 50°)? b. If the elevation angle is reduced from 50° to 15°, what attenuation would be observed at a frequency of 18 GHz, using the value you calculated for 18 GHz for an elevation angle of 50° in part (a) above as input? c. Based on the answer you obtained in part (b), would you recommend moving the uplink transmitter and changing the uplink frequency to 18 GHz?

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d. Would your answer change if you used the more complicated frequency scaling formula in Eq. (8.26) instead of the frequency-squared law in Eq. (8.25)? 4. A cable TV network (CATV) downlink signal to a cable head-end terminal is at approximately 12.5 GHz. The attenuation measured on the CATV downlink for 0.01% of the time in the location of interest is 12 dB. The elevation angle is 20°. The downlink is using dualpolarization frequency reuse to permit all of the channels to be sent through one satellite. In order to meet the QOS (quality of service) guarantees established for the terminal, the XPD must be no lower than 15 dB. a. What is the predicted XPD for 0.01% of the time if the downlink signal is linearly polarized with a tilt angle of 0°? b. What is the predicted XPD for 0.01% of the time if the downlink signal is circularly polarized? c. Is the XPD minimum of 15 dB for the QOS interference criterion met in either of these cases? 5. Tropospheric scintillation is not an absorptive effect and so will not lead to an increase in receiver noise temperature, as rain attenuation will. There will be a slight increase in noise temperature due to enhanced humidity levels that lead to tropospheric scintillation, but we will ignore that aspect for this question. A direct-to-home (DTH) Ku-band receiver has the following design specifications: System noise temperature in clear sky: 100 K Clear sky CN: 11 dB Performance margin: 99% of a year with CN  10 dB Availability margin: 99.9% of a year with CN  6 dB The climate in which the DTH receiver is to operate has been predicted to have the following tropospheric scintillation and rain attenuation statistics: Annual %-age time

10% 1% 0.1% 0.01%

Scintillation fade level

0.5 1.5 2.5 3.5

dB dB dB dB

Rain attenuation fade level

0 1 3 10

dB dB dB dB

a. What is the reduction in CN due only to tropospheric scintillations at the four percentage times? b. What is the reduction in CN due only to rain attenuation at the four percentage times (the effective temperature of the rain medium is 280 K)? c. Is the tropospheric scintillation a performancelimiting phenomenon, when acting in isolation from other effects?

341

d. Is the tropospheric scintillation an availabilitylimiting phenomenon when acting in isolation from other effects? e. Is rain attenuation a performance-limiting phenomenon, when acting in isolation from other effects? f. Is rain attenuation an availability-limiting phenomenon when acting in isolation from other effects? g. When combining tropospheric scintillation and rain attenuation effects for this DTH terminal, does the terminal meet both the performance and the availability specifications? h. If your answer is “no” to either the performance or availability questions in part (g) above, what additional CN margin is required to meet the performance and availability specifications? 6. An earth station complex is being designed to provide high availability access at Ka band to a satellite system used in a DTH Internet/Multimedia service offering. The earth station must meet an annual 99.99% availability level, that is, the maximum outage in any year must not exceed 0.01% of a year. The maximum EIRP from the earth station on the 30-GHz uplink is limited by interference considerations, resulting in a maximum rain fade margin of 30 dB on the uplink. For other reasons, the downlink rain fade margin (which includes the CN reduction due to noise temperature increase) at a frequency of 20 GHz is limited to a maximum of 9 dB. The earth station complex is situated in a region where the rain attenuation at 30 GHz is 50 dB for 0.01% of a year. The approximate frequency scaling formula may be used to find the equivalent 20 GHz fade level for 0.01% of a year. As can be seen, a single earth station will not be able to meet the availability requirements in this location. To overcome the rain attenuation, consideration is being given to operating two earth stations as a site diversity pair. The earth stations will operate at an elevation angle of 40°. The baseline between the earth stations is 90° to the satellite azimuth. a. Calculate the diversity gain achievable at 30 GHz and 20 GHz for site separations of 2, 3, 5, 10, 15, and 20 km in the above diversity system. b. If an additional margin of 3 dB must be allowed to provide a hysteresis band to assist switching between the earth stations on the 30 GHz uplink (i.e., the 50dB rain margin is increased to 53 dB), which is the first site separation calculated in part (a) that will permit the 30-GHz uplink availability target to be met (going from the lowest separation to the highest)? c. If an additional margin of 0.5 dB must be allowed for inefficiencies in the diversity combining on the 20-GHz downlinks between the diversity pair of earth stations (i.e., the 9-dB rain margin is increased to

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9.5 dB), which is the first site separation calculated in part (a) that will permit the 20-GHz downlink availability target to be met (going from the lowest separation to the highest)? 7. The attenuation that would be observed on a satellite path may be estimated by measuring the increase in antenna noise temperature and calculating the associated attenuation A from Eq. (8.40). Instruments that do this are called radiometers. They are useful for attenuation values between 0 and about 10 dB, but not very good for larger values of inferred attenuation. (The effect is called radiometer saturation.) Using Eq. (8.40), show why this occurs. 8. Geostationary communications satellites that use linear transponders have both advantages and disadvantages when compared with satellites that use onboard processing (OBP). One of the advantages of a linear transponder is its flexibility in usage—it can easily accommodate analog and digital traffic with varying capacity streams, provided they fit within the transponder bandwidth and power limitations. Another advantage is the ability to apportion outage to different parts of a given link. A typical two-way link is designed one way at a time. That is, for example, in a service between the United States and Thailand, the link from Thailand to the United States is designed separately from the return link from the United States to Thailand. For high availability services, a one-way

outage of 0.04% of a year is permitted. The 0.04% can be split equally, 0.02% on the uplink and 0.02% on the downlink. Thailand has a much more severe rain climate than the west coast of the United States and it may make the overall link design easier if a nonsymmetrical split is made of the total outage time. If the earth station in Thailand is in climate M and the earth station on the western seaboard of the United States is in Climate D (see Table 8.2); the uplink frequency is 12 GHz in both cases and the downlink frequency is 10 GHz in both cases; linear, vertical polarization is used both ways; the elevation angle is 5° at both the Thailand and the U.S. earth stations, find the following: a. How should the 0.04% outage time be divided up between the 12-GHz Thailand uplink and the 10-GHz U.S. downlink on the Thailand–U.S. one-way link so that the same rain attenuation is experienced on the Thailand uplink as it is on the U.S. downlink? b. How should the 0.04% outage time be divided up between the 12-GHz U.S. uplink and the 10-GHz Thailand downlink on the U.S.–Thailand one-way link so that the same rain attenuation is experienced on the U.S. uplink as it is on the Thailand downlink? [Note: for both parts (a) and (b) in the above question, the answer should be rounded to increments of 0.005%, e.g., 0.03%:0.01% or 0.035%:0.005%.]

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9

VSAT SYSTEMS The acronym VSAT stands for very small aperture terminal and, like many technical terms, its precise meaning has changed over the years. The first earth station antennas used in commercial satellite communications systems were very large and expensive1, with typical aperture diameters of 30 m. These large antennas operated in C band (64 GHz). With the rapid expansion of satellite telecommunications worldwide, there was a need to make access to the satellite more affordable. This came about in two ways: a significant increase in the transmit power capabilities of satellites and the move to frequency bands above C band. Both led to a rapid decrease in the size and cost of the earth station antenna.

9.1

INTRODUCTION Most VSAT systems operate in Ku band, with earth station antenna diameters of 1 to 2 m and transmitter powers of 1 or 2 W. The earth stations are usually organized in a star network, in which the earth stations connect to a central hub station via a GEO satellite. Data rates on the links are from a few thousand bits per second up to 256 kbps, depending on the traffic requirements. VSAT systems are used to link businesses and stores to a central computer system so that sales transactions can be completed more rapidly than by using a telephone line and modem, and so that a central office can rapidly distribute and collect information from a large number of locations in a region or country. In the 1990s there was rapid growth of VSAT networks in the United States. Businesses adopted VSAT networks for the transmission of data, as an alternative to the terrestrial telephone and data systems then available. The next decade is expected to see growth of VSAT networks operating in Ka band as new 3020 GHz GEO satellites become available. These networks may operate directly to the home for Internet connections and delivery of multimedia material. Few VSAT systems are used just for voice traffic, although the data rates are well matched to digital voice bit rates. For this reason, voice over IP (VOIP) has become a growth segment in VSAT operations. Large antennas are usually implemented using a symmetrical configuration, for ease of construction, with the feed on the boresight axis. The feed can either be in front of the antenna (a front-fed design) or behind the antenna, as in Cassegrain or Gregorian designs. Further, these different approaches may be axially symmetric or offset (see Section 9.6). A common break point in the design of antennas is at a main reflector diameter of about 100 wavelengths. If the diameter is larger than this, the additional cost of a Cassegrain or Gregorian design is more than outweighed by the superior off-axis and polarization performance achievable, and also the increased gain (up to 1 dB) that can be achieved by shaping the reflectors. Cassegrain and Gregorian antennas require a subreflector with a minimum diameter of 10 wavelengths. If the main reflector is less than 100 wavelengths in diameter, the subreflector becomes an appreciable fraction of the main reflector diameter and causes 343

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TABLE 9.1 Summary of Characteristics for the Intelsat VSAT IBS Antennas C-band antenna standard G/T (4 GHz), dB/K Typical antenna diameter, m Voltage axial ratio (circular polarization): XPD isolation value, dB: Ku-band antenna standard G/T (11 GHz), dB/K Typical antenna diameter, m Voltage axial ratio (linear polarization): XPD isolation value, dB:

F1

H4

H3

H2

22.7 3.5–5.0

22.1 3.5–3.8

18.3 2.4

15.1 1.8

1.09 27.3 dB

1.09 27.3 dB

1.3 17.7 dB

1.3 17.7 dB

E1

K3

K2

25.0 2.4–3.5

23.3 1.8

19.8 1.2

31.6 30.0 dB

20.0 26.0 dB

20.0 26.0 dB

Source: Intelsat Earth Station Standards (IESS) 207 (C-Band) and 208 (Ku-Band)2.

significant blockage and scattering problems. Earth stations with antenna aperture diameters less than 100 wavelengths were called small aperture terminals and, as the size reduced, the term VSAT was coined and then USAT (ultra small aperture terminal). Table 9.1 gives a summary of the standard antennas used in the Intelsat IBS (Intelsat business service). These are typical sizes of VSAT antennas used in international and domestic services. Figure 9.1 shows a typical VSAT antenna on the roof of a commercial building.

FIGURE 9.1

A typical VSAT antenna.

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The standard VSAT antennas in Table 9.1 are not as small as the Ku-band directto-home (DTH) antennas used for direct broadcast satellite television reception, which are typically 0.5–0.8 m in diameter. DBS-TV satellites use very powerful transponders, typically 160–240 W compared to 20 to 50 W of Ku-band satellites used for VSAT service. VSAT antennas are also much larger than the ultimate USAT—a handheld satellite telephone as used in Iridium, Globalstar, New ICO, and other mobile satellite service (MSS) systems, which have an omnidirectional antenna. While the size of the VSAT antenna is a key factor in making the service both economically attractive to the user and environmentally acceptable to the community, it places severe restrictions on the end-to-end system design. A careful balance has to be drawn between satellite transponder loading, transmitted power (both up and down), VSAT antenna offaxis emission (for interference considerations), clear sky performance, and—especially for Ku-band frequencies and above—availability during impaired propagation conditions (i.e., during rain).

9.2

OVERVIEW OF VSAT SYSTEMS The underlying concept behind most VSAT systems is to bring telecommunications service directly to the end user without any intermediate distribution hierarchy. Historically, traffic from individual users was bundled together into ever-larger groups and carried over trunk transmission lines via terrestrial microwave systems, satellite systems, or optical fiber cables, before being divided up (demultiplexed) into smaller traffic streams and redistributed to the users at the far end. This is still the most economical transmission architecture for point-to-point communications when the services are being brought into areas with relatively high concentrations of users. Such conditions do not always apply, however, and VSAT networks take advantage of the wide area broadcast capabilities of GEO satellites. In many regions of the world, the potential users are either widely distributed or the existing telecommunications infrastructure lacks the capacity to expand quickly to meet the demand for new users. This situation applies to most developing countries and, in many cases, network implementations have been adopted that “leapfrog” the conventional evolution of telecommunications systems. Geostationary satellites allied to microwave cellular technologies have been used to bypass completely the traditional expansion of analog telephony. One such solution is wireless local loop (WLL) coupled with VSAT distribution architectures. Figure 9.2 illustrates the concept schematically3. The VSAT/WLL concept usually has an optimum range of user densities where the economics are most favorable. This is illustrated in Figure 9.3. The information in Figure 9.3 is only approximate and can be affected by local topography (e.g., a large mountain range, which would favor satellite delivery), the availability of optical fibers in the country’s telecommunications network, or significant transportation routes such as a major rail system, which allows a lower cost optical fiber to be laid alongside the railroad tracks or right-of-way. VSAT networks allow multimedia traffic to be brought directly to the end user, but generally handle only small traffic streams (sometimes as little as the equivalent of one voice circuit). The traffic stream is also usually intermittent in nature: the user accesses the satellite in a demand assigned multiple access (DAMA) mode whenever a message is to be sent and receives a short reply in due course. This is typical in a point of sale (POS) VSAT system that is used to transmit credit card information at a gas pump or a store register. Information about the sale and the customer’s credit is sent to a central computer

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Farm

Wireless Public Payphone

Satellite Gateway

VSAT & WLL Terminal Backhaul (VSAT technology)

Village Local Loop (WLL technology)

FIGURE 9.2 Schematic of a VSAT/WLL communications network (from Figure 2-5 of reference 3). The geostationary satellite is used to link a large number of VSATs with the main switching center in a large city. Each VSAT acts as the link to the local switching center in the village or rural community, with the final mile of the telephony link being carried over a wireless local loop.

facility, and an authorization or denial is received in response. The interaction between the VSAT and the main hub earth station in the POS transaction is completely automatic and transparent to the user, the customer in this case. Most VSAT networks do not generate enough traffic to justify a dedicated satellite. Many do not even have enough traffic at any given instant to fill one satellite transponder. For this reason, most VSAT networks are designed around the use of leased transponders,

~0 Users/km2

~10 Users/km2

~100 Users/km2

~1000 Users/km2

User Density in number of users per square kilometer

Uneconomic: Requires large subsidy for any implementation

VSAT/WLL: Appears the best technological implementation

Fiber/microwave FS: Traditional terrestrial fixed service appears the best technological implementation

FIGURE 9.3 Approximate economic break points in the implementation choices for serving new regions with different population densities. Physical distances, major transportation routes, and geographic barriers, as well as the individual country’s demographics and political influences, can alter the break points.

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in the case of a large network, or a fractional transponder lease for a medium to small network. The network architecture selected will depend on a number of factors that are discussed in the following section.

9.3

NETWORK ARCHITECTURES There are three basic implementations of any telecommunications service: one-way; twoway; and split-two-way (sometimes referred to as split-IP, when referring to Internet traffic, since the outbound and inbound channels are routed over different systems). The two-way implementation is further divided into two basic network architectures: Star and Mesh. We will look first at the three basic implementations.

One-Way Implementation This is the mode of a satellite used in the broadcast satellite service (BSS). The introduction of digital technology allows the provider and user much greater flexibility in the operation of a broadcast network. By means of proprietary software in the user terminals, different parts of the downlink can be accessed by different subscribers according to the programs ordered from the supplier (and paid for by the user). This form of channel selection is called narrowcasting. There can be many narrowcasting groups within a larger broadcasting area. Figure 9.4 gives a schematic of this one-way (broadcast) application.

Split-Two-Way (Split IP) Implementation This implementation is used when there is no normal return channel as, for example, with Ku-band broadcast satellite service (BSS) systems that carry Internet traffic7. The relatively

Narrowcasting Group

Master Station

Broadcasting Coverage Area

FIGURE 9.4 Schematic of a broadcast satellite service coverage region in which smaller, narrowcasting groups exist within the broader coverage area (from Figure 2-1 of reference 3). The master control station sends encoded signals within the broadcast stream that enables certain users to have access to particular channel groupings according to the subscriber’s choice.

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high capacity downlink stream is not complemented by an uplink capability from the user terminal. If the BSS downlink is used as the download channel from an Internet service provider, the only option the user has for a return link is via another telecommunications channel, such as a standard telephone line. The Internet protocol (IP) is therefore split between a satellite downlink (outbound) channel and a terrestrial telephone (inbound, or return) channel; hence the term split IP for this implementation. The advantage of this approach is that the VSAT terminal does not require a transmit capability, which significantly reduces its cost and complexity. The disadvantage is that the telephone line connection must usually be through a modem, with a bit rate generally restricted to 56 kbps or less.

Two-Way Implementation In this case, a return link is designed into the service so that two-way communications can be set up over the same satellite, from the hub to the user and from the user back to the hub. The VSAT/WLL implementation illustrated in Figure 9.2 is a two-way service between the hub (in this case the satellite gateway) and any VSAT terminal. The architecture selected is the key to the economics of two-way connections: it can be either Mesh or Star. These two architectures are illustrated in Figures 9.5a,b, with the topology as viewed by the satellite shown in Figures 9.6a,b. Initially, the most common VSAT architectures were Star networks since the very low receive GT (gain-to-noise temperature ratio) of the VSATs, coupled with their limited transmit EIRP, was compensated for by using a large hub with high GT and EIRP. The Satellite

Master Control Station (the hub)

Satellite

VSATS (a )

VSATS (b)

FIGURE 9.5 (a) Illustration of a Star VSAT network. In this network architecture, all of the traffic is routed via the master control station, or hub. If a VSAT wishes to communicate with another VSAT, they have to go via the hub, thus necessitating a “double hop” link via the satellite. Since all of the traffic radiates at one time or another from the hub, this architecture is referred to as a Star network. (b) Illustration of a Mesh VSAT network. In this network architecture, each of the VSATs has the ability to communicate directly with any of the other VSATs. Since the traffic can go to or from any VSAT, this architecture is referred to as a Mesh network. It will still be necessary to have network control and the duties of the hub can either be handled by one of the VSATs or the master control station functions can be shared among the VSATs.

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VSAT VSAT VSAT

Satellite

Hub

VSAT VSAT (a )

VSAT

VSAT

VSAT

VSAT

VSAT

Satellite

VSAT

VSAT

VSAT

VSAT

VSAT (b)

FIGURE 9.6 (a) Topology of a Star VSAT network viewed from the satellite’s perspective. Note how the VSAT communications links are routed via the satellite to the hub in all cases. (b) Topology of a Mesh VSAT network from the satellite’s perspective. Note how all of the VSATs communicate directly to each other via the satellite without passing through a larger master control station (hub).

cost of the hub was therefore quite high and, at least for the smaller VSAT networks, somewhat prohibitive. This led to the concept of a shared hub, where several networks operate through one main hub. The difficulty with this approach for large countries with widely dispersed communities is that the host computers for the small VSAT networks are rarely close to the hub. A high-speed terrestrial data link is required between the host computers of the networks and the hub, which increases the cost of the network. Rather than have one large hub for all of the VSAT networks sharing the same satellite, the overall network evolved to allow each subnetwork to have its own hub as soon as the economics made it attractive. In this way, the host computer of each VSAT network can be co-located with its own hub, thus eliminating the cost of the interconnection between the hub earth station and the computer controlling the service offered through the VSAT network. Whether the hub is shared or dedicated on the one hand or the VSAT is connected to a single user or a local area network (LAN) with multiple users sharing access through an Ethernet connection on the other, in every case there will need to be an access control protocol.

9.4

ACCESS CONTROL PROTOCOLS The International Standards Organization (essentially a standards committee of the United Nations) has specified the open systems interconnection (ISO/OSI) that mandates a sevenlayer model for a data communication system, as shown in Figure 9.7.

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User 1

User 2

Application

Application

Presentation

Presentation

Session

Session

Transport

Transport

Network

Network

Link

Link

Physical

Physical

FIGURE 9.7 The ISO-OSI seven-layer “stack” for interconnecting data terminals. In this example, user one and user two are conducting a two-way communications session with each other. Each user interacts with their local device (e.g., a computer keyboard/visual display unit) at the application layer of the ISO-OSI stack. Their transaction is then routed via the various layers, with suitable conversions, etc., until the content is ready to be transmitted via the physical layer.

A satellite communications link occupies primarily the physical layer, which is where bits are carried between the terminals. A VSAT network must have terminal controllers at each end of the link and these occupy the network and link layers, the two layers above the physical layer. The network control center typically controls the system and is responsible for the remaining layers. Unfortunately, few communications systems conform in an easily identifiable way to the seven layers of the ISO-OSI model. (For example, the IP protocol stack of five layers simply puts the first three layers of the ISO/OSI stack into one layer). It is, however, very useful as a conceptual model which identifies functions that must be performed somewhere in every data communication network. Most data communication networks use some form of packet transmission, in which blocks of data are tagged with an address, error control parity bits, and other useful information before transmission. The receiving end of a link checks arriving packets for errors, and then sends an acknowledgement signal (ACK) that the packet was received correctly, or a not acknowledge signal (NAK) that tells the transmit end to resend a particular packet because the packet had an error. Some systems do not send acknowledgements, only NAK signals to request a retransmission of a packet with an error, since this speeds up data transmission. This is the error control method used in the Internet protocol TCP/IP. Generically, such systems are known as automatic repeat request (ARQ) systems. Chapter 7 discusses packet transmission systems and the problem of error detection and correction in packet networks using satellite links. The ISO-OSI stack was initially developed for terrestrial communications systems. For this reason, the protocols that implement the functions of each layer were designed for use in terrestrial circuits with low delay and low bit error rate (BER), that is, very high performance levels. These are key points when trying to use such protocols over satellites, particularly those in geostationary earth orbit (GEO). Many of the early protocols had a connection time-out of a few milliseconds. If no reply was received from the recipient

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in this interval, transmissions ceased. Similarly, an errored signal received from the source or an intervening node would trigger an automatic error recovery sequence. For example, the X.25 and X.75 packet systems use an ARQ approach, which, on detecting an error in a packet, immediately requests a retransmission and halts further transmissions until the corrected packet is received. Frame relay and ATM (asynchronous transfer mode) systems flag the error but continue the flow of information (continuous transmission ARQ). In both cases, the errored transmission must be corrected and suitable buffers at the receiver end (or intermediate node) used to restore the packets in their original order. The more errors that occur in the link, necessitating many retransmissions of packets, the slower the effective data throughput rate of the link becomes. The potential for delay and (propagation induced) errors are therefore critical design elements in digital VSAT connections.

Delay Considerations A typical slant range to a GEO satellite is 39,000 km. The one-way delay over such a GEO link (earth station to satellite to earth station) is 2  (rangevelocity)  260 ms. The one-way delay in a typical 4000-km transcontinental link via fiber-optic cable is a little over 13 ms. Neither example includes processing delay (e.g., source coding and/or compression, channel coding, baseband processing in the switching elements, frame length) which can add several tens of milliseconds or even over a hundred milliseconds. The time out element of a protocol is often referred to as the window of the connection. As long as the window is “open,” communications can continue without interruption. Figure 9.8 illustrates a continuous transmission ARQ system that has a 60-ms

Packets received by User 2 B

A

C

D Time

10 ms transmission delay

ACK A ACK window A

Time A1 A

Time A2

B

ACK B

ACK C

ACK window B

C

D Time

Packets transmitted by User 1 FIGURE 9.8 Illustration of a communications link with a 10-ms one-way delay and a 60-ms window. In this example, a packet or frame is sent at instant A1 from user 1 to user 2. User 2 receives the transmission without error and sends an acknowledgement back, which is received at instant A2, 20 ms after the initial transmission from user 1. This is well within the time window of 60 ms. The time window rolls forward after each successful acknowledgement. Thus the transmission from user 1 at instant B1 is received by user 2, and the acknowledgement received by user 2 at instant B2, within the new rolling time window of 60 ms. Each packet or frame is successfully received in this example.

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Packets received by user 2 A

B

C

D Time

ACK C ACK A window

ACK B window

ACK A

Time A1

A

B

C

D

ACK B

Time A2 (outside ACK window A)

Time Packets transmitted by User 1 FIGURE 9.9 Illustration of a communications link with a 260-ms one-way delay and a 60-ms window. In this example, a packet or frame is sent at instant A1 from user 1 to user 2. User 2 receives the transmission without error and sends an acknowledgement back, which is received at instant A2, 260 ms after the initial transmission from user 1. Unfortunately, instant A2 is well after the rolling window time-out of 60 ms. Transmissions from user 1 are automatically shut down by the protocol when the time-out of 60 ms is exceeded. Ignoring processing delays in this example, user 1 is only transmitting for 60 ms in every 260 ms, thus drastically lowering the throughput. Again, no propagation errors are assumed to occur.

window with a 10-ms one-way delay and Figure 9.9 illustrates a link with a 60-ms window and a 260-ms one-way delay. Clearly, satellite systems have to operate satisfactorily, and seamlessly (i.e., the user has no idea whether the link is terrestrial or via a satellite), with existing terrestrial networks or their utility is severely compromised. This is particularly true for GEO systems and there are two ways to make terrestrial protocols work with a satellite link. First, the protocols can be changed so that the time-out window is well in excess of 260 ms; second, the satellite element of the packet network can be configured to exist as a separate subnetwork within the global packet network. In practice, both solutions are adopted. Figure 9.10 illustrates the concept4. The VSAT and hub “protocol” equipment act as processing buffers to separate the satellite (VSAT) network from the terrestrial network. This is sometimes known as spoofing because the terrestrial part of the system uses a conventional protocol and is unaware of the VSAT network’s existence. The electronic processing and emulation permit traffic to flow seamlessly between two very different networks without operator intervention. In essence, this is the interface through which the VSAT user is connected to the VSAT network via the physical layer (see Figure 9.11). Once the user’s traffic has moved from the terrestrial network through the interface and is inside the VSAT network, the packet header is reorganized, with the appropriate routing and address of the traffic attached, so that the information can pass successfully over the satellite network to the correct recipient. Network management of the VSAT system, which includes congestion control, is also carried out in this element of the VSAT network, termed the network kernel. In addition, all of the necessary protocol conversions are carried out so that the packet or frame can successfully pass over a satellite connection with a long delay. A typical data link layer protocol (layer 2 in the ISO-OSI stack) that is used in a low delay, terrestrial link employs modulo-8 operation. That is, the protocol will transmit

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User’s higher layers

Gateway

Layer 3

Protocol

Layer 2

Interface

Layer 1

Physical interface

Gateway

353

User’s higher layers

Protocol Layer 3

User terminal

Network kernel

Network kernel

Interface Physical interface

VSAT

Hubstation

Layer 2 Layer 1 User terminal

VSAT network Terrestrial link Satellite link FIGURE 9.10 Protocol architecture of a Star VSAT network (Figure 2.2.1 of reference 4 © ITU, reproduced with permission). VSAT networks are normally maintained as independent, private networks, with the packetization handled at the user interface units of the VSAT terminals. The satellite access protocol (with a larger time-out window) is handled in the VSAT/hub network kernel, which also handles packet addressing, congestion control, packet routing and switching, and network management functions. Protocol conversion and, if necessary, emulation is handled by the gateway equipment.

only 7 unacknowledged frames before it stops transmissions; this leads to the low throughput demonstrated in Figure 9.9, particularly for GEO satellite links. The high level data link control (HDLC) protocol used in layer 2 for satellite systems therefore usually employs a modulo-128 operation. That is, 127 frames may be sent without receiving any acknowledgements before the protocol shuts down transmissions. Moving from modulo-8 to modulo-128 operation significantly increases the “window” size permitted for the link layer control. The concept, called protocol emulation, is demonstrated in Figure 9.114. Another critical function performed in the VSAT interface/kernel sections is to respond to polling activity from the terrestrial packet networks. It is normal for packet networks to poll users to see if there are packets to be sent. The interface/kernel elements in the VSAT network respond to the polling signals of the terrestrial network immediately, thus avoiding the long delay that would occur if the polling signal had to be passed over the satellite link. Negative acknowledgements are made to the polling signals until a request to send data is received over the satellite link. Given that the correct protocols have been inserted at ISO-OSI layer 2 within the VSAT system, and the management functions have been carried out (i.e., polling, switching, routing, addressing, and flow control) so that the link can operate successfully at a protocol level, there still remains the major part of the system design question to answer: how is the physical connection to be established over the satellite? To answer this question we must move from protocol design/emulation to transmission engineering. First, we will cover some of the basic techniques involved in developing a transmission design.

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Low throughput at layer-2 modulo-8 operation

A

B

Protocol emulators inserted at both ends of VSAT network

A

B B′

A′

VSAT

Hub

Mimic B with Optimized for satellite Mimic A with modulo-8 modulo-128 operation modulo-8 FIGURE 9.11 Schematic of protocol emulation to permit a VSAT network to operate seamlessly with a terrestrial network (Figure 2.2.2 of reference 4 ©ITU, reproduced with permission). In the modulo-8 operation shown in the top part of the figure, the VSAT network simply passes on the traffic over the satellite without any change to the protocols in the link layer (layer 2). This results in extraordinarily low throughput for GEO systems. In the lower part of the figure, the bottom two layers of the ISO-OSI stack are formed inside the VSAT network and the modulo-8 operation is changed to modulo-128. The two-layer stack also emulates the other side of the VSAT network so that terrestrial network A believes it is linked directly with terrestrial network B. That is, both terrestrial networks are mimicked at the VSAT interface/kernel.

9.5

BASIC TECHNIQUES Satellite link design has been covered extensively in Chapters 4, 5, and 6. However, some of the major elements that concern VSAT systems design will be reviewed below. These involve: selecting an appropriate multiple access scheme, evaluating signal formats, and establishing effective coding and interference practices.

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Multiple Access Selection As set out in Chapter 6, there are three fundamental multiple access schemes: frequency division multiple access (FDMA), time division multiple access (TDMA), and code division multiple access (CDMA). Within TDMA, there are two broad subdivisions of access: those that are closely controlled in time and access ability and those (like ALOHA and other Ethernet-like connections) that are loosely controlled in time and access ability. Multiple access schemes that do not closely control time, frequency, and/or code are significantly less efficient than those that do5,6. Pure ALOHA, which is a random access scheme, has a maximum throughput of 18.4%5. By combining some aspects of TDMA with the random access of ALOHA, slot reservation ALOHA can have an efficiency exceeding 60%5,6. Slot reservation is akin to a controlled access TDMA scheme with a very large frame. The intended application and the potential interference environment often determine the choice between FDMA, TDMA, and CDMA for VSAT networks, with economics also playing a major part. FDMA generally offers the lowest costs for entry-level VSAT systems from the user’s perspective since the receiver bandwidth and terminal transmit power required are the lowest. These systems carry thin route traffic, typically the equivalent of one digital voice channel at 64 kbit/s. The occupied bandwidth of an RF channel carrying a digital signal with a symbol rate Rs and using error control coding with a code rate Rc is given by B  Rs 11  a2 Rc Hz

(9.1)

where  is the roll-off factor of the root raised cosine (RRC) filters in the link. For example, in a link using QPSK modulation where two bits of information are carried by each transmitted symbol, a message information rate of 64 kbit/s results in a transmission symbol rate of Rs  32 ksps. If the message data bits are encoded with one-half rate FEC, code rate Rc  12, the occupied bandwidth required for a 64 kbit/s signal is Bocc  32,000  11  a2 12 Hz  64  11  a2 kHz

(9.2)

Typical values of  for satellite links lie between 0.25 and 0.5, with the higher value being easier, and thus cheaper, to realize when conventional analog filters are used in the transmitter and receiver. If an   0.5 RRC filter is used, the occupied RF bandwidth of this signal is Bocc  64  11  0.52  96 kHz

(9.3)

A VSAT that is required to transmit a 64 kbit/s stream of data using QPSK modulation, using half rate FEC and a raised cosine filter with a roll-off factor of 0.5, therefore needs an RF channel bandwidth of 96 kHz and has a receiver noise bandwidth of 64 kHz. Note that the roll-off of the RRC filter, while adding additional spectrum requirements, does not alter the noise bandwidth; all RF and IF RRC filters used in digital radio links have a noise bandwidth in hertz equal to the symbol rate in symbols per second. In practice, a guard band will have to be added between FDMA channels so that adjacent signals do not overlap in frequency at the satellite, and to allow the filters in the receiver that extract individual channels to roll off between channels. Most VSATs will operate unattended for most of the time and will be exposed to all weathers. The frequency of the fundamental oscillator may therefore drift. For this reason, fairly large guard bands need to be designed in, which generally leads to a spectrum allocation at the satellite around 120 kHz for each

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Satellite

36 MHz B/W Ku-band downlink Other VSAT channels

Receiver

RF channel from VSAT #N

IF receiver

FEC decoder

QPSK demodulator

Hub station

64-kbps terrestrial channel

Other VSAT channels FIGURE 9.12 Schematic of a 64-kbit/s equivalent voice channel accessing a satellite using FDMA. The 64-kbit/s information rate is contained in a bandwidth of 96 kHz when transmitted to the satellite. The bandwidth of the satellite transponder (from frequency f1 to frequency f2) is divided up, or channelized, into increments of 96 kHz so that a large number of VSATs can access the transponder at the same time. Each of the 96-kHz channels requires a certain amount of spectrum on either side to guard against drift in frequency, poor VSAT filtering, etc. The 96-kHz channels plus the guard bands on either side add up to a channel allocation of about 120 kHz per VSAT. From a spectrum allocation viewpoint, therefore, a typical 36-MHz satellite transponder would permit the simultaneous access of 300 VSATs, each of which is transmitting the equivalent of a 64-kbit/s voice channel. Because each VSAT uses a single channel continuously on the uplink, it is often referred to as SCPC (single channel per carrier)-FDMA.

64-kbit/s (voice) channel of the type described above. This situation is illustrated in Figure 9.12. In Figure 9.12, the 64-kbit/s data stream, equivalent to a PCM digital voice channel, could be derived from a point of sale device, for example, a credit card reader, an Internet access request, a digitized analog voice channel, etc., all of which require onward transmission over a VSAT network. The 64-kbit/s equivalent voice channel shown in Figure 9.12 is the output of the terrestrial–satellite interface equipment, after the required emulations and protocol conversions have taken place prior to transmission over a satellite network. This channel from the VSAT to the hub via the satellite is called the inbound or inroute channel. The RF transmission to the satellite from the VSAT will have a frequency that falls within the bandwidth of a specific transponder on the satellite. If the transponder operates in a bent pipe mode, with no onboard processing, the satellite will retransmit the multiple VSAT channels on the downlink with exactly the same channelization as on the uplink. Thus, in the example used in Figure 9.12, a transponder with 36-MHz bandwidth would transmit 300 channels to the hub station or other VSATs in the network. If the VSAT network is being operated in a Mesh mode, each VSAT must have a frequency synthesizer that allows it to select any of the 300 possible downlink channels, and the network must have a control channel that tells each terminal at which frequencies it should receive and transmit.

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It is more usual, however, for an FDMA VSAT system to operate in a Star mode, illustrated in Figure 9.5a. The hub earth station is therefore designed to receive all 300 downlink channels. The digital signal in each channel is recovered and the address information read off so that the hub can forward the information to the intended user. If the required end user is external to the VSAT network (i.e., in the terrestrial network) the information is passed through the hub interface equipment and on to the public switched telephone network (PSTN). If the required end user is within the VSAT network, or a response has been received through the interface equipment from the terrestrial PSTN, all of the information to be transmitted back to the various VSAT terminals is reassembled at the hub into a return channel. The return link from the hub to the satellite, and from thence to the individual VSAT terminals, is not normally sent as a multitude of narrowband FDMA channels. In most cases, the return channel from the hub to the VSAT terminals, called the outbound or outroute channel, is a single, wideband stream in a time division multiplexing (TDM) format. In the TDM stream, the separate, low data rate, narrowband signals for the individual VSATs are assembled in a predetermined format so that each of the VSATs can extract the required information destined for that VSAT. Figure 9.13 illustrates the TDM downlink concept used in FDMA VSAT Star networks. Note here the important difference between TDM and TDMA, which are often confused. Time division multiplexing, TDM, is not a multiple access technique. Digital signals from various sources are assembled into a single, high-speed data stream at one point, such as the hub of a VSAT network, and then transmitted as a single continuous stream. In TDMA, several sources, such as earth stations, transmit in sequence so that a sequence of RF signals is assembled at the satellite. In the FDMA Star VSAT network example shown in Figures 9.12 and 9.13, the VSAT network is quite large. Separate transponders are used for the inbound and the outbound channels. In many VSAT networks, the total instantaneous capacity required does not justify two separate transponders for the inbound and outbound signals. Such a case, using a shared transponder, is illustrated in Figure 9.14. In designing FDMA links, care should be taken to allocate the correct transmit power per channel in calculating the link budget so that the power spectral density is the same for every channel. For example, if a 54-MHz transponder is operated at an output power of 54 W, the power spectral density at the transponder output is 1 W per MHz. A single inbound 120-kHz downlink channel transmitted by the satellite will therefore have a transmit power level of (120 kHz54 MHz)  54 W  120 mW. This transmit power level is multiplied by the gain of the antenna in the direction of the hub (less feed and other losses) to give the effective isotropic radiated power, or EIRP, per 120-kHz channel. Operating a transponder in an FDMA mode also requires careful power balancing to ensure linear operation, which usually requires the output amplifier to be backed off. FDMA operation of a transponder with a large number of simultaneous RF channels requires significant output backoff in the transponder to obtain near-linear operation. The nonlinearity of the transponder at output levels close to saturation causes the generation of third-order intermodulation products that degrade the CN ratio in the SCPC channels. Amplifier output backoff values of between 3 and 7 dB can be found in the VSAT literature. The value required in any particular case depends on the transponder nonlinearity characteristic, the number of RF channels carried by the transponder, and the extent to which the power spectral densities of each RF channel in the transponder is matched. Backoff at the transponder output lowers the channel EIRP, and therefore degrades the downlink CN ratio. Automated transponder loading plans are used to optimize the backoff at the transponder output such that the overall CN of the link is maximized. The problem of optimizing transponder backoff becomes particularly difficult when the bandwidth is split between the

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“Outbound” TDM stream from hub via satellite

f1′

f2′

36 MHz Satellite transponder

Downlink “outbound” TDM stream from hub, via satellite, to each VSAT terminal

Combined channel rate ~20 Mbit/s

Demodulation and decoding

Transmission bandwidth ~36 MHz

Demultiplexing combined channel into individual equivalent 64 kbit/s channels

Pick off required 64 kbit/s signal that is intended for this VSAT from demultiplexed channel stream

64 kbit/s equivalent voice channel

Terrestrial/VSAT network interface

Terrestrial channel to User equipment

FIGURE 9.13 Schematic of the TDM downlink “outbound” channel from the hub, via the satellite, to the individual VSAT terminals. The 300 individual, narrowband, “inbound” channels received at the hub from the VSATs are sent back to the VSATs in a single, wideband, “outbound” TDM stream at a combined transmission rate of 20 Mbit/s. Each VSAT receives the downlink TDM stream and then demodulates and decodes it (i.e., changes the modulated band-pass signal into a baseband line code and removes the FEC). The line code is then passed through a demultiplexer which is used to extract the required part of the stream that contains the equivalent 64-kbit/s voice channel destined for that VSAT terminal. Carrier recovery and bit recovery circuits are used in the receiver in order to be able to identify the exact position of the required VSAT channel in time. The bandwidth of the satellite transponder (from frequency f1’ to frequency f2’) is fully occupied in this example.

inbound and outbound directions. The end-to-end gain setting of the transponder controls the backoff; it may be impossible to optimize the gain for both directions at the same time. A TDM downlink format is sometimes paired with a TDMA uplink plan, particularly for some advanced multimedia services that operate in Ka band (3020 GHz). Uplink FDMA formats are not as bandwidth efficient as TDMA. On the other hand, a VSAT

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Inbound narrowband VSAT channels

359

Outbound wideband TDM stream

36-MHz satellite transponder FIGURE 9.14 Illustration of a VSAT network frequency assignment in which the inbound and outbound channels share the same transponder in the satellite. In the example here, 18 MHz of spectrum is allocated for each side of the system connection. On the uplink to the satellite, the collection of FDMA narrowband channels transmitted by the VSATs coexists in the same transponder with the wideband TDM stream transmitted up by the hub. On the downlink from the satellite, the hub receives the collection of individual narrowband channels while the wideband TDM downlink stream is received by each VSAT. The precise frequency assignment can vary to suit the capacity of the VSAT network.

that uses TDMA on the uplink is required to transmit at the full burst rate of the TDMA scheme, and must therefore have a much more powerful transmitter than an SCPC-FDMA VSAT. If the average traffic for an individual VSAT is only one equivalent voice circuit (64 kbit/s), having to transmit at 5 Mbit/s, say, instead of 64 kbit/s can pose major difficulties. The VSAT transmit power must be increased by a factor of 500064  78 to maintain the same uplink CN, since the earth station receiver must have a bandwidth that is wider by the same factor. VSAT economics and bandwidth efficiency trade-offs have led to a hybrid TDMA-FDMA approach called MF-TDMA (multifrequency TDMA). This is illustrated in Figure 9.15. In the MF-TDMA example shown in Figure 9.15, each of the VSATs has to transmit at a burst rate that is approximately five times the normal single VSAT single-channel rate. If each VSAT transmits at a message data rate of 64 kbit/s and there are five VSATs sharing

Inbound, downlink TDM stream to hub Inbound, uplink MFTDMA VSAT bursts

Hub

A

B

C

D

E

FIGURE 9.15 Example of a multifrequency TDMA (MF-TDMA) scheme. In this particular case, five VSAT terminals (A, B, C, D, and E) share the same frequency assignment; that is, they all transmit at the same frequency. However, they each have a unique time slot in the TDMA frame when they transmit, so that they do not interfere with each other. The bursts from each VSAT are timed to arrive at the satellite in the correct sequence for onward transmission to the hub.

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the same frequency, the minimum burst rate would be 5  64 kbit/s  320 kbit/s. However, time gaps (the equivalent of frequency guard bands in an FDMA scheme) have to be left in between each of the individual payloads within the TDMA frame to avoid overlaps due to incorrect clock timing. The satellite transponder plan would look very much like Figure 9.14 with the exception that each of the single channel, narrowband, inbound frequency slots would actually be allocated to a number of VSATs sharing a small TDMA frame transmitted at the same frequency. In the same way that the hub used in the FDMA scheme detects all of the individual inbound VSAT frequencies and then bundles the outbound return traffic into one wideband TDM stream, the hub in the MF-TDMA scheme detects each of the inbound MF-TDMA VSAT signals and bundles the outbound traffic into a wideband TDM stream. CDMA schemes were originally employed in VSAT systems for encryption purposes in military applications because they have a very low probability of intercept. However, unless there is a severe interference environment, as in some terrestrial microwave cellular radio systems, CDMA schemes are not normally selected because they are, in general, less bandwidth efficient than FDMA or TDMA. TDMA, in particular MF-TDMA for narrowband applications, is normally more bandwidth efficient than FDMA. Each VSAT operating in a CDMA mode transmits with the same frequency and at the same time, and relies on the orthogonal coding employed in a direct sequence, or frequency-hopping, spread spectrum application, to provide complete mutual separation of the individual communications signals. CDMA comes into its own for VSAT satellite applications when off-axis emissions from the earth terminals are likely to cause interference into another satellite. This is illustrated in Figure 9.16.

2° USAT(2)

2° WSAT

Geostationary orbit arc: satellites at 2° spacing

USAT(1)

Beamwidth of VSAT

VSAT

FIGURE 9.16 Illustration of how a VSAT can cause interference to other satellite systems. In this example, the VSAT is transmitting to a wanted satellite (WSAT) but, because the antenna of the VSAT is small, its beam will illuminate two other adjacent, unwanted satellites (USATs) that are 2° away in the geostationary arc. In a like manner, signals from USAT (1) and USAT (2) can be received by the VSAT, thus causing the potential for interference if the frequencies and polarizations used are the same. Off-axis emission is closely specified by the ITU-R and is a key element in uplink power control design. When LEO constellations are sharing the same frequency bands as GEO systems, the use of CDMA may confer some advantages for coordination purposes at the expense of system capacity.

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In Figure 9.16, if the wanted satellite (WSAT), unwanted satellite one (USAT 1), and unwanted satellite two (USAT 2), all use the same frequencies and polarizations, the use of CDMA would prevent interference between the systems if orthogonal CDMA codes are used. There would, however, be an increase in the noise received since each CDMA channel would appear as a noiselike signal to every other CDMA channel. Thus, each additional CDMA signal will incrementally reduce the CN of the channel. There is no hard and fast CN threshold for CDMA links, as there is for TDMA and FDMA links, when the received signal characteristics become unusable. This soft threshold allows for some design flexibility, but care has to be taken to avoid undue errors induced by excess selfinterference from the VSAT signals. Different satellite multiple access schemes can be used with a range of VSAT network access schemes, offering considerable flexibility to the system designer. Figure 9.17 illustrates this schematically. In Figure 9.17, the satellite transponder is accessed using any of the three satellite multiple access modes: FDMA, TDMA, or CDMA. In addition to the multiple access scheme used, the VSAT network requires some form of satellite network access control.

VSAT Host computer

F E P

H B E Hub station Satellite capacity access protocol (FDMA, TDMA, CDMA)

VSAT

V B P

V B P

Satellite network access protocol (satellite-efficient access protocol such as MF-TDMA/TDM plus addressing, switching, routing, flow control, congestion control) Customer’s data protocol FIGURE 9.17 The different layers of protocols used in VSAT networks (after Figure 3-1 of reference 3). The host computer sends traffic for the VSAT network to the front-end processor (FEP) at the hub station (the master control station of the VSAT network). The FEP passes the traffic to the hub baseband equipment (HBE) to be formatted for transfer over the satellite link via the selected satellite access protocol. The satellite then passes the outbound (or outroute) traffic on the downlink to the VSATs. The VSAT baseband processor (VBP) then extracts the relevant traffic for the user and forwards it after any necessary protocol conversion, etc.

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The satellite network access control ensures that the most efficient access protocol is used, for example, MF-FDMA/TDM or SCPC FDMA, for that particular satellite VSAT network. Not shown in this figure are the protocol emulators that act as the interface between the terrestrial and satellite networks, or the satellite access control that monitors switching, flow and congestion control, addressing, etc. (see Figure 9.10). Most of the signal formats for satellite multiple accesses have been discussed in detail in Chapter 6. The subsection below reviews the generic case for digital satellite multiple access using TDMA schemes.

Signal Formats The VSAT uplink signal, the inbound or inroute channel, in an MF-TDMA multiple access format must contain sufficient information for the intended receiver to acquire the carrier frequency, lock onto the incoming data so that the timing of the bit stream can be obtained, and then identify the start of the payload transmission. This generic procedure is shown in Figure 9.18. More details of various TDMA/packet signal formats can be found in Chapter 6.

Modulation, Coding, and Interference Issues Modulation and channel coding are key considerations in determining the efficient and error-free transfer of information over a communications channel. They also have an impact on the potential for interference to another system and from another system. A modulation that has a large number of bits per symbol (e.g., 64 QAM) will occupy a relatively small bandwidth but it will require relatively high amplifier linearity and a high CN ratio in the receiver. It is also more susceptible to interference than modulations with fewer bits per symbol. High-index modulations require significantly more margin than low-index modulations. In choosing the most appropriate modulation and channel coding for a VSAT system, ease of implementation is also a major factor since VSATs are very cost-sensitive. Faced with these trade-off decisions, the most common forms of modulation used in VSAT systems are QPSK and, when spectrum efficiency is less important, BPSK. In an ideal QPSK system with ideal RRC filters and no channel coding, a value of EbN0 of 10.6 dB will provide a BER of 106, corresponding to a receiver overall CN ratio of 13.6 dB, ignoring any implementation margin. The CN requirement can be significantly reduced if channel coding is applied. Coding aspects are covered extensively in Chapter 7. Only those aspects that touch on VSAT systems will be reviewed below.

End of previous frame

CR

BTR

UW

Payload

FIGURE 9.18 Generic sequence for the start of a burst from a VSAT inbound signal. When the burst is received at the hub, the first part of the packet enables the carrier recovery (CR) to occur, followed by the bit timing recovery (BTR). The unique word (UW) identifies the start of the payload in the new frame.

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Channel Coding Channel coding can take the form of a block code or a convolutional code. Convolutional coding is a process where the encoding and decoding process is applied to a group of bits in sequence rather than a bit at a time, as in a block code. The number of bits in the encoding sequence, k, is called the constraint length of the convolutional code. In the decoding process, k bits are used to evaluate the value of each bit transmitted. Since the encoding process is applied to the signal prior to transmission and is used to detect and correct for bit errors, it is called a forward error correcting (FEC) code. In a like manner, a block FEC code is applied to the channel prior to transmission. Convolutional and block codes can be used together on a channel. One example is a channel that first has an inner convolution code applied to the bit sequence and then has an outer interleaved code such as a Reed–Solomon code applied. Reed–Solomon codes combine good error detection capability with high code rates. This form of concatenated coding is used extensively in many communications systems (see for example, the coding used in direct broadcast satellite television discussed in Chapter 11), since the interleaved coding will counter burst errors while the convolutional FEC coding will counter individual bit errors. Direct broadcast satellite television is one example of such a coding approach, and the recording of music on CDs is another. The encoding and decoding procedure is illustrated schematically in Figure 9.19. For VSAT systems that have small traffic streams, excess processing delay can add significantly to the end-to-end link delay. This is very important for GEO systems and for LEO/MEO systems with satellites that have large onboard processing capabilities. The processing delay due to first interleaving a signal and then de-interleaving it adds a fixed amount of overhead, as well as requiring buffering at both ends of the transmission link. For this reason, Reed–Solomon outer codes are not normally added to signals that have information rates below about 256 kbit/s, even though the lower EbN0 value for a given BER performance is so significant (see Figure 9.20). For links that have no real requirement for instant response times and multimedia interactivity, but require the best BER performance for a given EbN0 (typical of most Internet links), Reed–Solomon codes are a very practical way of reducing the power requirements for a given link and BER specification.

Input bit stream

FEC encoder

Communications channel Reed–Solomon Reed–Solomon interleaver de-interleaver

FEC decoder

Output bit stream

Outer code

Inner code FIGURE 9.19 Schematic of the encoding and decoding process when an “inner” and “outer” code are applied to a communications signal. The Reed–Solomon interleaved block code is applied after the FEC (either block or convolutional) on the encoding side. The reverse occurs on the decoding side. While it might look like the FEC code is outside the Reed–Solomon code, it is the time they are applied that counts. Since the FEC is applied first in the encoding process, it is then wrapped by the Reed–Solomon code, which becomes the outer wrapping of the doubly coded signal.

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10−1

10−2 Coherent PSK w/o FEC 10−3 Viterbi R = 1/2, K = 7 BER

364

10−4 Sequential R = 1/2 10−5 Viterbi R = 1/2, K = 7 + Reed–Solomon 10−6

10−7 0

2

4

6

8

Eb/No (dB)

10

12

14

FIGURE 9.20 BER vs Eb /N0 performance of coherent QPSK for various types of codes (from Figure 3.3.5 of reference 4. ©ITU, reproduced with permission).

Interference Interference between systems operating with similar characteristics (i.e., frequency bands, polarizations, and services) is usually the subject of intense debate, particularly when a new system seeks to operate close to an existing system, in terms of orbital separation or antenna beam directions. The interaction between operators seeking to ensure that no harmful interference is caused by, or to, their respective systems is called coordination. The coordination process is the subject of extensive regulation by the ITU and national frequency management authorities [e.g., the Federal Communications Commission (FCC) in the United States]. The key aspect in such coordination exercises lies in determining the power radiated by the interfering station in the direction of the interfered-with station. The calculation of the received interference power will have four elements: • The output power of the interfering station’s transmit amplifier • The transmit gain of the interfering station’s antenna in the direction of the interferedwith station • The receive gain of the interfered-with station’s antenna in the direction of the interfering transmissions • The path loss between the two stations. The interference geometry is illustrated in Figure 9.21. Recommendation ITU-R S.7288 mandates the maximum permissible levels of offaxis EIRP density from a VSAT transmitting in the 14-GHz band (Ku band). The relevant part of the Recommendation is abstracted below. “VSAT earth stations operating in the 14 GHz frequency band used by the fixed satellite service must be designed in such a manner that at any angle  specified below, off the main lobe axis of an earth station antenna, the maximum EIRP in any

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WSAT

Gain, Gw (dB), in direction of wanted satellite

USAT

Gain of antenna of interferedwith satellite, Gs, (dB), towards the VSAT Path to satellite which will have a fixed path loss and variable loss due to propagation impairments

Main lobe and first sidelobes of VSAT antenna

Gain, Gu (dB), in direction of interfered-with satellite

VSAT with HPA power of P (dBW) FIGURE 9.21 Illustration of the interference geometry between a VSAT and a satellite of another system. The EIRP of the VSAT toward the interfered-with satellite [P(dBW)  Gu(dB)] is the interference power from the VSAT into the interfered-with satellite. To develop the interference link budget, the gain of the interfered-with satellite in the direction of the VSAT, Gs(dB), would be used, plus any additional effects along the path (such as site shielding, if used, expected rain effects for given time percentages, etc.)

direction within 3° of the geostationary-satellite orbit should not exceed the following values: Angle off-axis

Maximum EIRP in any 40-kHz band

2.5°    7° 7°    9.2° 9.2°    48°  48°

33  25 log  dBW 12 dBW 36  25 log  dBW 6 dBW

In addition, the cross-polarized component in any direction  degrees from the antenna main-lobe axis should not exceed the following limits: Angle off-axis

Maximum EIRP in any 40-kHz band

2.5°    7° 7°    9.2°

23  25 log  dBW 2 dBW”

There are two important notes contained in the footnote to the above Recommendation. Since the Recommendation was developed for 3° satellite spacing in GEO, the first note indicates that the off-axis limits may need to be reduced by up to 8 dB where the satellite spacing is 2°. The second note pertains to CDMA VSAT systems. When there are N VSATs expected to transmit simultaneously on the same frequency, the maximum permitted EIRP values should be decreased by 10 log N. The rapid increase expected in satellite delivery of Internet-like traffic direct to homes and offices has led to a multitude of proposed new constellations of satellite systems,

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with the result that interference aspects have received a lot of study within the ITU and ETSI (European Telecommunications Standardization Institute). The off-axis limits have been tightened under a new proposal before ETSI, which is expected to be reflected in a new version of ITU-R S.728 above. The new proposals are directed toward the use of Kaband satellites with VSAT apertures well below 1 m. The new limits proposed to ETSI are given below. “The maximum EIRP in any 40 kHz band within the nominal bandwidth of the copolarized component in any direction  degrees from the antenna main beam axis shall not exceed the following limits under clear-sky conditions, within 3 of the geostationary orbit plane. Angle off-axis

Maximum EIRP in any 40-kHz band

1.8°    7.0° 7.0°    9.2° 9.2°    48°  48°

19  25 log  10 log N dBW 2  10 log N dBW 22  25 log   10 log N dBW 10  10 log N dBW”

For CDMA systems, N is the number of VSAT earth stations transmitting simultaneously on the same frequency. For TDMA and FDMA systems, N  1. The proposed recommendation by ETSI is for satellites that are spaced 2° apart in GEO. The values in both cases (CDMA and TDMA/FDMA) may be relaxed for directions more than 3° away from the geostationary plane since VSAT antenna patterns are normally optimized for the GEO arc. During rain fade conditions, the values in the above equations may be exceeded through the application of uplink power control (ULPC) at affected stations, in order to overcome the rain attenuation. The effective off-axis emission levels received by adjacent satellite systems are not expected to vary substantially from that during the clear sky condition if the uplink power control system is designed and operated properly. Uplink power control systems have the potential to be very inaccurate9, particularly those that rely on open-loop control10,11. Carefully controlled experiments at Ku band12 and Ka band13 have shown that there is an irreducible error on the order of 1.2 dB for open-loop ULPC and, even with closed-loop ULPC, where the power level is measured at the satellite, there can be time-delay constraints that will limit the accuracy14.

9.6

VSAT EARTH STATION ENGINEERING Antennas The key element in a VSAT system is the earth station antenna used at the VSAT earth stations. The small size and low transmit power of a VSAT station are the factors that keep the price of the earth station at a level that makes a VSAT network an economic alternative to a terrestrial data network using telephone lines and modems. Large antennas are usually implemented using a symmetrical configuration, for ease of construction, with the feed on the boresight axis. The feed can either be in front of the antenna (a front-fed design) or behind the antenna, as in a Cassegrain or Gregorian design. Figure 9.22 shows these three antenna configurations. The front-fed antenna has a paraboloidal reflector with a feed at its focus. The feed is often a scalar horn, a circular waveguide horn with a wide flare angle and corrugations on the internal surface. A scalar horn has a circularly symmetric pattern leading to equal beamwidths in all planes, and good aperture efficiency. When only a portion of the

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Symmetric

F

F

F

Offset

F Front fed

F F Cassegrain

Gregorian

FIGURE 9.22 Configuration of front-fed, Cassegrain, and Gregorian antennas. The top three configurations are axially symmetric while the bottom three are offset-fed designs that reduce the aperture blockage considerably.

paraboloid is used, an offset reflector results. An offset reflector fed by a scalar horn is used as the preferred configuration for most DBS-TV receiving antennas. The basic design of the Cassegrain antenna has a paraboloidal main reflector and a hyperboloidal subreflector. One focus of the subreflector is coincident with the focus of the main reflector; the feed is at the other focus of the subreflector. The surface profile of both reflectors can be modified by redistributing energy across the aperture so as to increase the aperture efficiency and reduce blockage caused by the subreflector. This is called a shaped reflector Cassegrain antenna. The Cassegrain configuration is used widely for large earth station antennas. It is preferred over the Gregorian configuration because the subreflector is closer to the main reflector and therefore easier to support. However, where there is a severe off-axis interference environment, an offset-fed Gregorian design is the best to use. Gregorian antennas are also occasionally used on DBS-TV satellites with a phased array feed to create complex coverage regions on earth.

Transmitters and Receivers Historically, large earth stations are assembled as discrete elements. On the receive side, the antenna and feed components are connected by waveguide to the front-end, low noise amplifier (LNA). Behind the LNA, a mixer/down-converter changes the signal from radio frequency (RF) to an intermediate frequency (IF). After filtering and amplification, the IF signal is demodulated, demultiplexed, and decoded, and the baseband signal forwarded to the user. The transmit side is the mirror image of the receive side with the signal input at baseband and the output at RF, with the LNA receiver replaced by a high-power amplifier (HPA) transmitter. This design of earth station is typical of a hub station used in a VSAT network. Much of this discrete component design has changed with the

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ODU

Building roof

IFL IDU

FIGURE 9.23 Schematic of a VSAT user setup. The VSAT outdoor unit (ODU) is located where it will have a clear line of sight to the satellite and is free from casual blockage by people and/or equipment moving in front of it. The interfacility link (IFL) carries the electronic signal between the ODU and the indoor unit (IDU) as well as power cables for the ODU and control signals from the IDU. The IDU is normally housed in a desktop computer at the user’s workstation and consists of the baseband processor units and interface equipment (e.g., computer screen and keyboard). The IDU will also house the modem and multiplexer/demultilexer (mux/demux) units if these are not already housed in the ODU.

Antenna Feed LNC

DEM IFL

HPC

Outdoor unit (ODU)

MOD

Baseband processor (BBP)

To data terminal equipment

Indoor unit (IDU)

FIGURE 9.24 Schematic of the typical configuration of a VSAT earth station (after Figure 4.1.1 of reference 4 ©ITU, reproduced with permission). The low noise converter (LNC) takes the received RF signal and, after amplification, mixes it down to IF for passing over the interfacility link (IFL) to the IDU. In the IDU, the demodulator extracts the information signal from the carrier and passes it at baseband to the baseband processor. The data terminal equipment then provides the application layer for the user to interact with the information input. On the transmit operation, the user inputs data via the terminal equipment to the baseband processor and from there to the modulator. The modulator places the information on a carrier at IF and this is sent via the interfacility link to the high-power converter (HPC) for upconversion to RF, amplification, and transmission via the antenna to the satellite.

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introduction of digital receivers and the need to develop cheap, mass-produced VSAT terminals. The VSAT earth station can now be considered as two basic components: an outdoor unit (ODU) and an indoor unit (IDU). This is illustrated in Figure 9.23 The ODU and IDU units are broken down further in Figure 9.24. Figure 9.24 gives a typical configuration of a VSAT that has the modulator/demodulator (modem) equipment located in the IDU. In some cases, the low noise block (LNB) or low noise converter (LNC) and the HPB (high-power block) or HPC (high-power converter) house the complete RF output and RF input stages of the transmitter and receiver, the up-converters and downconverters and, in many cases, the modem. With the mass production of VSATs, the LNBs and LNCs are being developed on application specific integrated circuits (ASICs), very often grown as a single microwave monolithic integrated circuit (MMIC). The hub station has a more complex design than the VSAT station since it not only has to handle all of the inbound and outbound traffic, rather than just “thin route” traffic of a single or a few users, but it also has to manage the network control functions. Figure 9.25 gives the general layout of the hub earth station.

Outbound TDM channels

Outbound modulators

Hub antenna

IF Interface

HPA

LNA

Inbound deodulators

Transmit PCE Control Bus

UC

Receive PCE

Line interface equipment to host computers

DC Inbound MF-TDMA channels

HUB control interface

FIGURE 9.25 Schematic of a typical hub master control station (after Figure 4.2.1 of reference 4. ©ITU, reproduced with permission). The line interface equipment handles the terrestrial ports to the host computer. The control bus via the hub control interface allows all of the transmit, receive, and switching functions to be carried out. The transmit processing and control equipment (PCE) prepares the TDM stream for the outbound link to the VSATs. This stream passes through the IF interface (the equivalent of the interfacility link of the VSAT in Figure 9.23) to the up-converter (UC) that mixes the IF to RF. The high-power amplifier (HPA) amplifies the TDM stream and the antenna transmits the signal. On the receive side, the antenna passes the individual inbound MF-TDMA signals to the low noise amplifier (LNA) for amplification prior to down-conversion (DC), demodulation, and so on to the user.

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As far as possible, all of the equipment used is purchased as commercial off-theshelf (COTS) to reduce costs. A critical part in the economics of satellite access is the antenna. All satellite systems require controlled access to the satellite and it is usual to specify the frequency tolerances allowed, the range of power flux densities acceptable at the satellite antenna, and the polarization purity of the transmission as a minimum on the transmit side. On the receive side, a minimum antenna G/T will be set for a given elevation angle. Antenna tracking tolerances will also be specified (if needed). To keep costs to a minimum, most satellite systems develop antenna standards (see Table 9.1) and, for a given antenna purchased, performance and availability will be guaranteed by the space segment provider for that antenna standard and for the power levels specified. If a nonstandard antenna is used, however, the satellite system owner will require on-site testing of the complete earth station system to ensure compliance with specifications. This would be extremely expensive for a VSAT user.

9.7 CALCULATION OF LINK MARGINS FOR A VSAT STAR NETWORK The minimum allowed carrier to noise, (CN)0, for a typical inbound VSAT link is 6.0 dB, with BPSK modulation and half rate FEC encoding, giving a BER of 106. This is called the threshold (CN)0 value, and will vary depending on the modulation and FEC methods used on the link. Ignoring system errors, such as mispointing the antennas or satellite problems, the clear sky CN ratio on a given inbound link can be reduced to the threshold (CN)0 by either a rain fade on the uplink or a rain fade on the downlink. In a like manner, there are two rain fade margins for the outbound link: from the hub to the satellite (uplink) and from the satellite to the VSAT (downlink). The entire twoway system drops below the performance minimum when any one of the low margin links drops below threshold (i.e., the design margins are exceeded). We want a link failure to be much less likely on the satellite–hub links (either uplink or downlink), because such a failure affects every VSAT in the network. Lack of performance or availability on a VSAT–satellite link (either uplink or downlink) affects only that individual VSAT connection. The link fade margin is found by using the reciprocal formula (CN) ratio with the (CN)0 value set to its threshold value. The procedure for the calculation of link margin is different for networks using linear, bent pipe, transponders and networks that are connected via a satellite that uses onboard processing (OBP). A bent pipe transponder simply translates the uplink frequency to the downlink frequency, without any regeneration of the signal. A rain fade on the uplink will therefore be reflected in reduced output power of the satellite on the downlink. When OBP is present, the transmitted power from the satellite is always held constant, irrespective of any fade on the incoming, uplink signal. An uplink fade, therefore, causes a worse BER to be received in the transponder on that uplink, which will add errors to the signal to be transmitted on the downlink from the satellite. Hence with OBP, bit error rates on the uplink and downlink add as follows 1BER2 overall  1BER2 uplink  1BER2 downlink

(9.4)

With a bent pipe transponder, link margins are calculated differently for the uplink  downlink, forward (outbound) and return (inbound) links. On the downlinks, the

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reciprocal (CN) formula is used to find the lowest value of (CN)down [ (CN)down min] that, combined with (CN)up, gives the threshold value for (CN)0, which we have called (CN)threshold in the equation below.

or

1 1 1   1C N2 threshold 1C N2 up 1CN2 down min

(9.5)

1 1 1   1C N2 down min 1C N2 threshold 1CN2 up

(9.6)

Rain attenuation on the uplink will cause the power received at the transponder to fall, causing the output power from the transponder to be reduced. On an inbound link with many VSAT stations operating in SCPC-FDMA, rain fading on one VSAT uplink will not significantly affect the total power received by the satellite, so the transponder operating point will not change and the transponder output power for that link will fall linearly with uplink attenuation. Thus (CN)down at the hub station receiver will fall as (CN)up fades, dB for dB, and 1CN2 0 fade  1CN2 0 clear sky  AdB uplink fade

(9.7)

On an outbound link (hub to VSAT), using TDM, rain attenuation on the uplink from the hub will affect the operating point of the satellite transponder, because only one signal (that of the hub station) is present. If the input to output characteristics of the satellite transponder are known, the change in the received uplink power, which causes a change in the operating point of the transponder, can be taken into account in calculating the resulting downlink (CN) ratio. The downlink (CN) ratio will be higher for a nonlinear transponder because, when the transponder is operated at the beginning of its nonlinear region (as most are) to maximize transponder output power, a 1-dB reduction in input power will cause less than a 1-dB change in output power. In general, the difference is less than 0.5 dB so, if the nonlinear characteristics of the transponder are not known, assuming the transponder to be truly linear (1 dB for 1 dB) between input and output will result in no more than 0.5-dB error in the calculation of uplink attenuation margin. The downlink attenuation margin in most VSAT networks tends to be small, and is usually the limiting factor in the network design. Uplink power control (ULPC or UPC) at the hub station can maintain a relatively constant power level at the input to the transponder over the operating range of the ULPC system, which in turn increases the fade margin by the available range of the ULPC system. Most ULPC systems operate in an open-loop mode and so care must be taken not to overdrive the satellite transponder by increasing the power too much above the true rain fade level by incorrect operation of the ULPC system. For this reason, most ULPC systems do not add power to the uplink until at least a 1- to 2-dB rain fade has been confirmed on the downlink, and even then, the addition of power on the uplink is always lagging behind the true rain fade level so as to avoid both transponder saturation problems and interference into other systems. Once the ULPC system has reached its maximum level of added power, any further increase in rain fade level will cause the overall (CN) ratio in the VSAT receiver to fall. Figure 9.26 in Example 9.1 shows typical (CN)0 behavior in a VSAT receiver with and without ULPC at the hub station of a Star network.

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(C/N )0 Outbound link (dB) 14

2-dB uplink attenuation threshold for starting ULPC adjustments

12

10

7-dB ULPC range Up to 7-dB of uplink power control applied

8

(C/N )0 threshold for BER = 10−6

6

4

No uplink power control

2

0

0

2

4 6 8 10 Inferred uplink attenuation (dB)

12

FIGURE 9.26 Outbound link performance with rain attenuation on the uplink, with and without ULPC. Due to the error in predicting the uplink attenuation from a measure of the attenuation on the downlink, the up-link power control (ULPC) does not begin to operate until the inferred uplink attenuation reaches 2 dB. From that point on, for every additional inferred dB of uplink attenuation, a dB of ULPC is added until the ULPC maximum is reached. The ULPC maximum is 7 dB in this case.

9.8 SYSTEM DESIGN PROCEDURE: EXAMPLE 9.1 Few VSAT system designs start from scratch. In most cases, either a satellite already exists and a VSAT network has to be designed around its capabilities or a VSAT network is already in place and the opportunity occurs to upgrade it with a new satellite launch. In both cases, the VSAT system needs to be optimized carefully, balancing the transmission requirements of the inbound and outbound links so that the space segment capacity is used efficiently. In the design example below, Ku-band satellite transponders are to be used in a VSAT system. The general characteristics of the hub and VSATs are known. The design example shows the steps needed to develop a link budget, determine capacity information, and carry out trade-off calculations to optimize the links for a Star VSAT network. The design of VSAT networks is dominated by the low gain and small transmitter power of the VSAT station. The links must operate with low CN ratios at the VSAT receivers and with low fade margins.

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Description of System The satellite communications system consists of a single hub earth station, a geostationary satellite operating with a linear transponder, and a number of VSAT stations. The system uses two Ku-band transponders on the GEO satellite. Each VSAT station sends and receives a 64-kbit/s data stream to and from the hub. Digital data are sent to the hub from the VSATs by the inbound link via one transponder at a message bit rate of 64 kbit/s using binary phase shift keying (BPSK) and half rate forward error correction (FEC) coding, giving a transmitted bit rate of 128 kbps. The occupied RF bandwidth of each VSAT channel is 160 kHz, corresponding to ideal RRC filters with   0.25. Multiple access for the inbound link is by SCPC-FDMA with RF channels spaced 200 kHz apart to allow a 40-kHz guard band between channels. Data from the hub station to the VSATs (the outbound link) are sent as a continuous TDM stream of packets using a second transponder and BPSK with half rate FEC. The VSAT antenna has a diameter of 1 m and a saturated output power of 2 W.

System Parameters System values Uplink frequency for transponder 1 Downlink frequency for transponder 1 Uplink frequency for transponder 2 Downlink frequency for transponder 2 Range to satellite (all stations)

14.02 GHz 11.72 GHz 14.08 GHz 11.78 GHz 38,500 km

Satellite transponders Maximum output power Transponder bandwidth Transponder input noise temperature Antenna gain (maximum, transmit and receive)

20 W 54 MHz 500 K 34 dB

VSAT station parameters Transmitter output power Antenna gain: transmit receive Receiver system noise temperature (clear air) Receiver system noise bandwidth

2.0 W 41.5 dB 40.0 dB 150 K TBD

Hub station parameters Maximum transmit power Receiver system noise temperature (clear air) Antenna gain at 14 GHz (transmit) Antenna gain at 11.7 GHz (receive) Receiver system noise bandwidth

200 W 150 K 50 dB 48.5 dB 128 kHz

Atmospheric losses In clear air at 14 GHz In clear air at 11.7 GHz

0.7 dB 0.5 dB

Constants Boltzmann’s constant, k,  1.38  1023 J/K S 228.6 dBW/K/Hz

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For this VSAT network, the requirement is to determine an appropriate transponder loading (i.e., the number of channels passing through the transponders on the inbound and outbound links) and determine whether the link is power limited or bandwidth limited. It is also necessary to find whether the link is balanced (i.e., the same number of inbound and outbound channels) so that the most efficient use is made of the transponders. The best approach in most design exercises is to develop the basic link calculations so that the operation of the system is understood and then to perform a trade-off analysis. There are four RF links in a two-way VSAT system. The inbound link from the VSAT to the hub station has an uplink to transponder 1 on the satellite and a downlink to the hub station. The outbound link from the hub station to the VSAT has an uplink to transponder 2 on the satellite and a downlink to the VSAT. Each of the four links has its own CN ratio, which is calculated from a separate link budget. The uplink and downlink CN values on the inbound link are combined to give an overall inbound (CN)0 in the hub station IF receiver, and the uplink and downlink CN values on the outbound link are combined to give an overall outbound (CN)0 in the VSAT receiver. Once the CN values are known, bit error ratios for the inbound and outbound links are calculated from the inbound and outbound CN values.

Preliminary Calculations All link budgets require knowledge of the free space path loss between the earth station and the satellite and the noise powers in the operating bandwidths. The free space path loss, Lp, for a range of 38,500 km is calculated as follows. Free Space Path Loss From Chapter 4, the path loss, Lp, is given by Lp  20 log14pRl2 dB

(9.8)

where R is the range (38,500 km) and  is the wavelength, in meters. The path losses at 11.7 GHz (  0.02564 m) and 14.0 GHz (  0.02143 m) are therefore Lp(14.0)  207.1 dB and Lp(11.7)  205.5 dB. (Note that it is usual to state dB values only to one decimal place. If the value is to be used in additional calculations, higher accuracy may be retained until the full answer is known. In this case Lp(14.0)  207.073 S 207.1 dB and Lp(11.7)  205.515 S 205.5 dB.) Noise Powers Noise Power in Transponder 1, Inbound SCPC FDMA Channels The inbound VSAT links that pass through transponder 1 have a message data rate of 64 kbit/s with half rate FEC encoding, giving a transmitted bit rate of 128 kbps, with BPSK modulation. From Section 9.5, the noise bandwidth is therefore 128 kHz, since BPSK has one bit per symbol. From Chapter 4, noise power, Np, is given by Np  kTsBn W

(9.9)

with Ts  500 K S 27 dBK, Bn  128 kHz S 51.1 dBHz, and k  228.6 dBW/K/Hz. This gives the noise power at transponder 1 input as Ntr1  228.6  27  51.1  150.5 dBW. Noise Power in the Hub Station Receiver, Inbound SCPC FDMA Channels The inbound VSAT signals arrive at the hub station after being retransmitted (i.e., transponded) by the satellite. The noise bandwidth is still 128 kHz, since each VSAT channel is received by a separate IF receiver at the hub station. The system noise temperature of the hub station receiver is Ts  150 K S 21.8 dBK.

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Hence Nhub  228.6  21.8  51.1  155.7 dBW

(9.10)

Noise Power in Transponder 2, Outbound TDM Channels The outbound TDM bit stream from the hub station to the VSATs passes through transponder 2. Not all of the VSATs will be transmitting and receiving simultaneously and so the TDM stream should be sized for the likely average activity rate of the network. No information was provided on the network size so, for this example, we will assume a starting value noise bandwidth of 1 MHz in the VSAT receiver as an initial value for the first set of calculations. This corresponds to a BPSK signal with a baseband data rate of 500 kbit/s and half-rate FEC encoding. The system noise temperature of satellite transponder 2 is Ts  500 K S 27 dBK. Hence, for the 1-MHz noise bandwidth of the VSAT receiver, the noise power at the input of transponder 2 is Ntr2  228.6  27  60  141.6 dBW

(9.11)

Noise Power in the VSAT Receivers, Outbound TDM Channel Each of the VSATs receives the outbound TDM stream from the hub station in a noise bandwidth of 1 MHz. The system noise temperature of the VSAT receiver is Ts 150 K S 21.8 dBK. Hence NVSAT  228.6  21.8  60  146.8 dBW

(9.12)

Link C/N Ratios We will now look at the inbound links from the VSATs to the hub. Each VSAT transmits a 128-kbit/s stream made up of 64 kbit/s message bits encoded with half rate FEC and modulated with BPSK. The bit streams from the many VSATs are received via the satellite at the hub station using separate IF receivers for each VSAT channel. Each of these separate channels has a noise bandwidth of 128 kHz, numerically equal to the symbol rate of the BPSK signal. We need to set a minimum allowable CN and an implementation margin (a small factor allowed to compensate for practice vs theory), so that the link margin can be calculated. We could begin by setting a threshold for the overall CN ratio of 10 dB at the demodulator input. This assures that the link will provide a BER in excess of 106. A minimum CN ratio of 10 dB is pessimistic, however, for most digital systems. A more realistic approach for BPSK with a transmission bit rate of 128 kbit/s is to expect a coding gain of 5.5 dB with half rate forward error correction and an implementation margin of 0.5 dB. Given that the theoretical minimum CN ratio for BER  106 using BPSK modulation is 10.6 dB, the threshold (CN)0 ratio will be 10.6 dB  implementation margin  coding gain  10.6  0.5  5.5  5.6 dB. For the TDM wideband downlink data stream, a higher implementation margin is expected, 1.0 dB, which leads to a (CN)0 threshold  6.1 dB. The frequency used for this VSAT network is Ku band. At Ku band and above, rain can cause appreciable signal loss for small time percentages in any given year. Since rain is an absorbing medium, there will also be an increase in sky noise when rain is present in the path. Depending on the rain climate at the VSAT stations and the availability required, a link margin will be needed, over and above the implementation margin required by the demodulator, so that the VSAT system can operate normally under rain conditions. The difference between the threshold (CN)0 ratio and the clear air (CN)0 ratio is the link margin. For this VSAT network, we will use a clear sky (CN)0 overall of 14.0 dB,

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giving a link margin of 8.4 dB on the inbound link and a link margin of 7.9 dB on the outbound link.

Inbound Links The next step in the design is to develop the link power budgets and to evaluate the CN ratios of the inbound and outbound links. A worst-case design is adopted, with the VSAT station on the 3 dB contour of the satellite beam, and losses included in the link budget to account for earth station antenna mispointing, polarization mismatch, etc. If the link works successfully under worst-case conditions, it will always work for any VSAT station under those conditions. VSAT Uplink Power Budget We will assume that the VSAT is located at the satellite beam’s edge-of-coverage, the 3 dB contour of the satellite’s uplink (receiving) antenna pattern. The power received at the satellite from a single VSAT, in dB, is given by Pr  Pt  Gt  Gr  Lp  Losses

(9.13)

where Pt is the transmit power (2 W), Gt is the gain of the VSAT antenna (41.5 dB), Gr is the gain of the satellite receive antenna (34 dB), Lp is the free space path loss at 14 GHz (207.1 dB), and the losses include the satellite antenna edge of beam loss (3 dB), clear air uplink atmospheric loss due to gases (0.7 dB), and miscellaneous losses of 0.5 dB to account for antenna mispointing, feed loss, etc. The link budget is shown below. Pt (VSAT transmit power)  2 W Gt (VSAT transmit antenna gain at 14 GHz) Gr (satellite receive antenna gain at 14 GHz) Lp (free space path loss at 14 GHz) Leoc (edge of coverage loss of satellite antenna) Latmos (atmospheric gaseous loss) Lmisc (miscellaneous losses) Pr (received power at satellite transponder input)

3.0 41.5 34.0 207.1 3.0 0.7 0.5 132.8

dBW dB dB dB dB dB dB dBW

Uplink Inbound C/N in Transponder 1 Each VSAT is transmitting to a separate receiver in the hub station with a noise bandwidth of 128 kHz. At the input to the transponder receiver, the received power of each VSAT carrier signal is 132.8 dBW and the noise power in the receiver for each VSAT channel is 150.5 dBW. The uplink CN ratio in a noise bandwidth of 128 kHz is therefore 1C N2 trans1  132.8  1150.52  17.7 dB

5 S ratio of 58.886

Downlink Inbound VSAT Channel Power Budget at the Hub At present, we do not know the number of uplink (inbound) VSAT channels that will be sharing the same 54-MHz transponder on the satellite. The total power available in the transponder is 20 W. For this design example we will begin by assuming that each of the inbound VSAT signals is retransmitted by transponder 1 at an output power level of 1.0 W, and then develop a VSAT channel downlink power budget using the relevant values of Pt, Gt, Gr, Lp, Leoc, Latmos, and Lmisc. There will be one significant difference, however. It is normal

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to locate the hub earth station well within the satellite antenna coverage. Thus, only 1 dB is assumed for the edge of coverage loss, Leoc. Pt (satellite transmit power)  1 W per channel Gt (satellite transmit antenna gain at 11.7 GHz) Gr (hub station receive antenna gain at 11.7 GHz) Lp (free space path loss at 11.7 GHz) Leoc (1 dB contour of satellite antenna) Latmos (atmospheric gaseous loss) Lmisc (miscellaneous losses) Pr (received power at the hub earth station receiver input)

0.0 dBW 34.0 dB 48.5 dB 205.5 dB 1.0 dB 0.5 dB 0.5 dB 125.0 dBW

Downlink C/N at the Hub from Transponder 1 on the Inbound Links The inbound streams from the VSATs, transmitted down by transponder 1 in the satellite, are each received at the hub in a noise bandwidth of 128 kHz. The VSAT carrier power in the hub is 125.0 dBW and the noise power in the hub station receiver in a noise bandwidth of 128 kHz is 155.7 dBW. The CN is therefore 1CN2 hub inbound  125.0  1155.72  30.7 dB

5 S ratio of 1,174.96

Overall Inbound C/N at the Hub The overall CN of the inbound VSAT channels is calculated by the reciprocal formula where the CN values are power ratios and not their dB values. Namely: 1  3 1C N2 overall 4  1  3 1CN2 hub inbound 4  1  3 1C N2 trans1 4  1 3 111752 4  1 3 158.882 4  0.0008511  0.0169837  0.0178348 giving 1CN2 overall  110.01783482  56.0702824 S 17.5 dB in clear sky Note how the VSAT uplink CN ratio of 17.7 dB dominates the overall inbound CN ratio in the hub station receiver because of the high CN ratio on the downlink to the hub station. Analysis of the Inbound Side of the VSAT System The saturated output power of the satellite transponder was given as 20 W. The VSAT streams are accessing the satellite in an SCPC-FDMA format, prior to amplification in transponder 1. When FDMA is used with a large number of simultaneous channels, the output amplifier must be backed off to avoid the generation of unwanted intermodulation products. Typical output backoff is 2 to 3 dB. With an output backoff of 2 dB, the 20-W (13 dBW) transponder saturated output power is reduced to 11 dBW S 12.6 W. Each VSAT channel was assumed to use 1 W of power on the downlink. Hence, the transponder can share 11 dBW (12.6 W) among a maximum of 12.6 VSAT channels. The calculation of the overall CN ratio on the inbound link from the VSATs showed that the overall (CN)0 at the hub in clear sky was 17.5 dB. A 14-dB clear sky value was assumed earlier to provide sufficient margin above the demodulator threshold to meet the availability requirements. There is therefore excess CN in the link between the VSAT and the hub station and it is possible to share the downlink power between a much larger number of VSATs and still keep the overall CN  14 dB in clear air. Using the reciprocal ratios of CN values (NOT the dB values) and forcing the overall CN to be 14 dB, we can write 1  3 1C N2 overall 4  1  3 1CN2 hub inbound 4  1  3CN2 trans1 4

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which yields

1 3 1252 4  1  3 1CN2 hub inbound 4  1  3C N2 trans1 4

The ratio (CN)trans1, the uplink CN on the inbound link from the VSAT, is fixed by the transmitting parameters of the VSAT at 17.7 dB S a ratio of 58.88. The minimum permissible downlink CN, [(CN)hub inbound], to maintain an overall (CN)0 of 14 dB is 3 1CN2 hub inbound 4  1  3 11 252  11 58.882 4  43.45 S 16.4 dB

However, earlier calculations had shown that the (CN)hub inbound was 30.7 dB. There is therefore an excess CN of 30.7  16.4  14.3 dB S a ratio of 26.9 channels. We can therefore increase the number of channels by a ratio of 26.9 from the 12.6 channels we had before, when we assumed 1 W per channel, to a total of 26.9  12.6  339 channels. The power allocated to each VSAT downlink at the transponder output is now (1 W)26.9 or (12.60 W)339  0.037 W/channel. (Remember that the 20-W output amplifier of the transponder was backed off by 2 dB to 12.6 W.) The next stage of the system calculation is to check whether the inbound link is power or bandwidth limited. In most cases, power may be traded for bandwidth, giving additional flexibility in the system design. Each VSAT transmission requires 200-kHz RF bandwidth (160-kHz plus guard bands either side). The transponder bandwidth is 54 MHz. With 200 kHz per channel, the transponder can handle 54 MHz/200 kHz  270 channels. There is sufficient power available to carry 339 VSAT channels, but there is only space for 270 VSAT channels. The inbound link is therefore considered to be bandwidth limited with a maximum capacity of 270 channels and a clear air link margin of 8.4 dB.

Inbound Links with 270 Channels Downlink VSAT Channel Power Budget at the Hub With 270 channels, the power allocation per channel in the 20-W transponder is 20270  74.1 mW per channel 1 11.3 dBW. The received power will therefore drop from 125 dBW to 136.3 dBW. The noise power will remain the same, giving the CN of the inbound channels at the hub as 1CN2 hub inbound  136.3  1155.7 dBW2  19.4 dB

5 S ratio of 876

Overall Inbound (C/N)0 at the Hub The overall (CN)0 of the inbound VSAT channels with 270 channels arriving at transponder 1 is 1 3 1C N2 overall 4  1  3 1CN2 hub inbound 4  1  3 1CN2 trans1 4  1  3 1872 4  1  3 158.882 4  0.0115  0.0169837  0.0285 giving 1CN2 overall  1  10.02852  35.1197 1 15.5 dB

The overall CN ratio has dropped 2 dB from the case where 20 channels were assumed. The relatively small change in CN (from 17.5 to 15.5 dB) for an 11.3-dB increase in capacity (20 channels to 270) shows that the dominant CN in the inbound link is the uplink channels from the VSAT. The much larger number of channels (270) might, however, require an additional backoff in the output power amplifier on the satellite to avoid intermodulation products for so many closely, and regularly, spaced carriers. This reduction in output power for the inbound carriers might drop their CN to below that of the uplink

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VSAT CN (i.e., below a value of 58.88). If that were the case, the link would be downlink limited (satellite to hub) rather than uplink limited (VSAT to satellite) which would not be a good design.

Outbound Links The outbound, TDM link goes from the hub, via the satellite, to the VSAT stations. The TDM link is a continuous bit stream at a bit rate Rb bits per second using BPSK modulation and half rate FEC. The bit stream conveys data packets that are addressed to the relevant VSATs sequentially. The CN ratio in the VSAT receiver must not be lower than the 6.1-dB threshold that gives BER  106 with this modulation. We will initially design the outbound link for an overall (CN)0  10 dB in clear air in the VSAT receiver, giving a link margin of 3.9 dB to overcome rain attenuation and noise temperature increase (during rain) for the time percentages the availability requirements impose. In the same way as for the inbound links, a link power budget is tabulated and the CN margins evaluated to provide the necessary insights into the performance of the overall system. Hub Uplink Power Budget Pt (hub station transmit power)  100 W Gt (hub transmit antenna gain at 14 GHz) Gr (satellite receive antenna gain at 14 GHz) Lp (free space path loss at 14 GHz) Leoc (edge of coverage loss of satellite antenna) Latmos (atmospheric gaseous loss) Lmisc (miscellaneous losses) Pr (received power at satellite transponder input)

20.0 50.0 34.0 207.1 1.0 0.7 0.5 105.3

dBW dB dB dB dB dB dB dBW

Uplink C/N in Transponder 2 The hub station is transmitting to each VSAT receiver in a noise bandwidth of 1 MHz. We calculated the received power of the TDM stream earlier to be 105.3 dBW and the noise power in the satellite transponder for the TDM stream to be 141.6 dBW. The uplink CN is therefore 1C N2 trans2  105.3  1141.62  36.3 dB

5 S ratio of 4265.86

Outbound Downlink Power Budget at the VSAT Transponder 2 carries a single TDM–BPSK carrier, so we will assume 1-dB output backoff, giving an output power of 12 dBW. This yields the power budget below. Pt (satellite transmit power) Gt (satellite transmit antenna gain at 11.7 GHz) Gr (VSAT receive antenna gain at 11.7 GHz) Lp (free space path loss at 11.7 GHz) Leoc (1 dB contour of satellite antenna) Latmos (atmospheric gaseous loss) Lmisc (miscellaneous losses) Pr (received power at the hub earth station receiver input)

12.0 34.0 40.0 205.5 3.0 0.5 0.5 123.5

dBW dB dB dB dB dB dB dBW

Downlink C/N at the VSATs from Transponder 2 on the Outbound Links The outbound TDM stream from the hub station, transmitted down by transponder 2 in the satellite, is received at each VSAT in a noise bandwidth of 1 MHz. We calculated

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the received power of each VSAT carrier signal in the hub to be 123.5 dBW and the noise power in the VSAT receiver as 146.8 dBW. The outbound downlink CN is therefore 1C N2 VSAT outbound  123.5  1146.82  23.3 dB

5 S ratio of 213.86

Overall Outbound C/N at the VSATs The overall CN of the outbound TDM stream received by the VSAT receiver is calculated, as before, using the reciprocal formula with the CN values as power ratios. Namely: 1  3 1C N2 overall 4  1  3 1CN2 VSAT outbound 4  1 3CN2 trans2 4  1  3 1213.82 4  1 3 14265.82 4 giving 1CN2 overall  203.6 S 23.1 dB in clear sky conditions Analysis of the Outbound Side of the VSAT System The value of 10 dB was assumed for the VSAT (CN)0 ratio in clear sky conditions to meet the availability requirements at a BER of 106. As can be seen from the outbound calculations, a CN value higher than 10 dB exists at the VSAT. This extra margin of CN can be used to provide a wider bandwidth (higher capacity) TDM stream or to increase the rain margin for the same capacity. We will investigate the maximum bit rate that can be used on the outbound link with an overall (CN)0 of 10 dB in the VSAT receivers in clear air. The CN for the outbound link is high from the hub to the satellite (36.3 dB) but lower from the satellite to the VSAT (23.3 dB). To find the maximum bandwidth we can use on both parts of the link, we need to find the reduction in CN ratio, (CN)reduction, that will apply equally to both parts of the link and still provide an overall CN of 10 dB in clear sky conditions. This works out to a CN reduction of 13.1 dB on each part of the link. That is (36.3  13.1)  23.2 dB on the uplink and (23.3  13.1)  10.2 dB on the downlink. A CN reduction of 13.1 dB amounts to a reduction ratio of 20.4. This means that the outbound stream from the hub to the VSAT now has a noise bandwidth 20.4  (the original bandwidth)  20.4  1 MHz  20.4 MHz, and an occupied bandwidth of 30.6 MHz. The equivalent bandwidth of the VSAT channels in the TDM stream is 128 kHz (ignoring any overhead requirements for frame and bit recovery, addressing, etc.), so the maximum number of VSAT channels that can be served by the hub is (20.4 MHz)(128 kHz)  159. Since there is not enough power to provide the required CN ratio if the 54-MHz transponder were full to capacity in terms of bandwidth, the outbound link is power limited. A summary of the system design and analysis so far is given in Table 9.2. Table 9.2 shows that the inbound link has excellent availability, better than 99.99% of an average year for a network located in the eastern United States. The outbound link is much more likely to fail, because of rain on the uplink or downlink. Availability is 99.55%, corresponding to outages totaling about 40 h a year. The outages occur for approximately equal times of 20 h on the uplink and downlink when uplink power control is not used. When the uplink rain attenuation exceeds 3.6 dB, all VSAT terminals in the network have outbound BER 106. Using ULPC at the hub station significantly reduces outages caused by uplink rain attenuation, limiting the outages to about 25 min in an average year.

System Analysis It is now important to look at the total capacity (bit/s) that can be delivered from the VSATs to the hub (inbound link) and from the hub to the VSATs (outbound link) to see if there is a reasonable match. The lower of the two rates will determine the maximum number of VSATs that can be simultaneously served.

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TABLE 9.2 Performance of a VSAT Star Network with 1-m Diameter VSAT Antennas and 2-W Transmit Amplifier Link

Inbound

Outbound

Channel data rate Modulation Multiple access No. of 64 kbit/s channels RF bandwidth occupied in transponder Clear sky (C/N)up Clear sky (C/N)down (C/N)0 clear sky (C/N)0 threshold Uplink fade margin Downlink fade margin Uplink fade margin with 7-dB ULPC Downlink rain attenuation margin Link availability, eastern U.S. (no ULPC)

64 kbit/s BPSK SCPC-FDMA 270 54 MHz 17.7 dB 30.7 dB 17.5 dB 6.1 dB 10.3 dB 24.6 dB Not applicable 20.5 dB 99.995%

10.2 Mbit/s BPSK TDM 159 30.6 MHz 23.2 dB 10.2 dB 10.0 dB 6.4 dB 3.6 dB 3.7 dB 10.6 dB 1.8 dB 99.55%

Link availability, eastern U.S. (7-dB ULPC)

99.995%

99.70%

The inbound link through transponder 1 was found to be bandwidth limited, with a fully loaded capacity of 270 VSAT stations. At an information rate of 64 kbit/s per channel, the transmission rate is 17.28 Mbit/s. The outbound link (transponder 2) was found to be power limited and could carry a maximum of 159 channels. With an equivalent 64-kbit/s data rate per channel, the outbound link can handle 10.176 Mbit/s. There is clearly a large mismatch between the two sides of the links, which may lead to inefficient use of the space segment. The inbound and outbound links can be balanced if we increase the gain of the VSAT antenna. For example, the CN of the receiving terminal can be increased by 3 dB by increasing the diameter of the VSAT antenna by a factor of 12, from 1 m to 1.41 m in diameter. We can maintain the same VSAT EIRP by reducing the transmitter power by 3 dB, from 2 to 1 W, which will keep the outbound link CN unaltered. The lower transmitter power will reduce the cost of the transmitter sufficiently to compensate for the increased antenna diameter. The new downlink (CN)0 ratio is 13.2 dB in clear air with 159 channels, which can be reduced to 10.2 dB with 318 channels. At 128 kbit/s per channel, the TDM bit rate in the outbound channel is 40.7 Mbps. Allowing for overhead in the packet transmissions and using  = 0.4 RRC filters will restrict the data bit rate to around 34.56 Mbps, which matches the inbound data rate for 270 VSAT channels at 128 kbps. The inbound link overall CN is controlled by fading of the uplink signal when rain occurs between the VSAT and the satellite. The link margin for the inbound uplink is 8.4 dB and we can use almost this entire margin for rain attenuation on the uplink, because the satellite input noise temperature is unaffected by rain at the earth’s surface. We can assume a linear response from the transponder for a single VSAT signal, because the rain does not affect the remaining stations, assuming wide geographic separation between terminals. The inbound link’s overall (CN) ratio is dominated by the uplink CN from the low power VSATs. Hence, each dB of rain attenuation on the inbound uplink causes approximately 1 dB reduction in overall CN in the hub station receiver, so we have an uplink rain attenuation margin of 8.4 dB for each VSAT on the inbound link.

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The outbound link CN is controlled by the downlink from the satellite to the VSAT station. When rain occurs in this link, the uplink CN at the transponder remains the same, and the downlink CN fades. We have a fade margin of 4.1 dB on the outbound downlink, which must be shared between rain attenuation and increase in VSAT receiver system noise temperature. In this case, rain attenuation of 2.1 dB will cause the sky temperature to increase from a clear air value of 40 K (based on 0.7-dB clear air attenuation) to 128 K. The VSAT system noise temperature will increase by a similar amount, from 150 K in clear air to 238 K in rain, resulting in a noise power increase of 2.0 dB. Thus the downlink CN ratio will be reduced by 4.1 dB when downlink rain attenuation is 2.1 dB. The annual availability of the VSAT terminal will depend on its location, but typically in the eastern U.S. rain, rain attenuation of 2 dB is experienced 4 h in an average year at Ku band. For locations with more frequent occurrence of heavy rain, a larger antenna can be specified. A 2-m diameter antenna has 3-dB greater gain than a 1.41-m antenna, which translates directly to a 3-dB increase in downlink fading margin. The major parameters of the redesigned VSAT system are detailed in Table 9.3. Table 9.3 shows that the redesigned VSAT network has a balanced capacity on each link and can support 270 channels at 64 kbit/s each. Link availability is improved on the outbound links, with less than 160 min of outage caused by uplink rain attenuation in an average year, without ULPC. Rain on the downlink will cause individual VSAT stations to suffer outages totaling 22 h in an average year. When 7 dB of ULPC is available at the hub station, outages caused by uplink attenuation on the outbound link are reduced to less than 20 min in an average year. Figure 9.26 illustrates the effect of using ULPC at the hub station, based on typical 14-GHz rain attenuation in the eastern United States. Uplink power is increased once the uplink attenuation exceeds 2 dB and then compensates for rain attenuation until the attenuation exceeds 9 dB. Without ULPC, the (CN)0 for the outbound link reaches the 6.4-dB threshold with an uplink attenuation of 4.5 dB. With a ULPC margin of 7 dB, the threshold (CN)0 is reached when the uplink rain attenuation exceeds 11.5 dB.

TABLE 9.3 Performance of an Improved VSAT Star Network with 1.4-m Diameter VSAT Antennas and 1-W Transmit Amplifier Link

Inbound

Outbound

Channel data rate Modulation Multiple access No. of 64 kbit/s channels RF bandwidth occupied in transponder Clear sky (C/N)up Clear sky (C/N)down (C/N)0 clear sky (C/N)0 threshold Uplink fade margin Downlink fade margin Uplink fade margin with 7-dB ULPC Downlink rain attenuation margin Link availability, eastern U.S. (no ULPC) Link availability, eastern U.S. (7-dB ULPC)

64 kbit/s BPSK SCPC-FDMA 270 54 MHz 17.7 dB 30.7 dB 17.5 dB 6.1 dB 10.3 dB 24.6 dB Not applicable 20.5 dB 99.995% 99.995%

17.3 Mbit/s BPSK TDM 270 51.3 MHz 20.9 dB 10.9 dB 10.5 dB 6.4 dB 4.5 dB 4.3 dB 11.5 dB 2.1 dB 99.65% 99.80%

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In some VSAT service applications, however, a much larger outbound capacity may be required than inbound, particularly if the link has been designed for Internet connections where the “download” channel will need a much higher throughput than the “upload” request from the user. If the VSAT system considered above in the design example were matched so that the same transmission rate is on each side (inbound and outbound), the maximum capacity would be limited to 159 equivalent channels. As is the case for most Star VSAT networks, the number of outbound channels is limited by the downlink CN ratio. The downlink CN ratio can be increased by increasing the satellite transmit power, increasing the VSAT antenna diameter, or reducing the rain margin. Additional trade-off analyses can also be performed that consider other aspects of the overall network operation.

9.9

SOME NEW DEVELOPMENTS A number of significant technology enhancements are being planned for new VSAT systems. These are being driven in two main technical areas that seem to develop rapidly in parallel: digital signal processing and microminiaturization. The first enables large amounts of complex “transactions” to take place in microseconds, while the second enables the product to be built into ever-smaller packages. Perhaps the change that had the biggest impact on satellite developments over the 1990s was the shift from military spending being the major financial input for space ventures to entrepreneurial commercial funding. With that major change has come market driven economics that force the end product to be reliable and low-cost. Behind all of the developments has been the surge in demand for communications capabilities that can bring any stream of bits into any terminal, anywhere: the ultimate multimedia portable unit. The term multimedia implies the capability of handling any traffic stream, whether it be voice, video, fax, or data. The mixing of short voice packets with long data packets is now achieved through asynchronous transfer mode (ATM) techniques. The transfer of the ATM packet or frame is typically handled through virtual containers or circuits which exist at the time the connection is set up, but are torn down once the transaction is completed, thus freeing the capacity to handle other traffic. The sizing of the virtual container is set by the link requirement but must be within the overall capacity of the channel: there would be no point in attempting to transfer the equivalent of 1 Mbit/s down a 64-kbit/s channel. The access technology must also be capable of mixing the various virtual containers within the available channel capacity. Equally important will be the ability of the intervening nodes to handle these virtual containers. One such node is the satellite. Most advanced satellites are designed to handle the vast increase in Internet-like traffic and they take advantage of onboard processing (OBP) technology. OBP was once considered to be overly expensive in both mass and power for commercial satellites but is now the enabling technology of most new systems. The Iridium constellation was the first commercial satellite system to employ OBP extensively although, in the civilian area, INTELSAT VI was the first to use onboard, satellite switching between beams. In the Iridium system, the uplink signals from the handheld units are mixed down to baseband within the satellite payload, the header information stripped off the frames, and the traffic reassembled and routed to the appropriate output port, whether it be for downlinking to another handheld unit, downlinking to the gateway (hub) earth station, or cross-linking to another satellite. All of the new satellites being proposed for Internet service at Ka band (3020 GHz) have OBP capabilities. This is not a trivial undertaking. Not only must the

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OBP design allow for a large variety of traffic types and permit flexible connection between any points in the coverage, it must also be incredibly reliable. Most satellites are designed for at least a 10-year lifetime and, given the need to design, develop, and launch satellite networks, system designers must attempt to predict the likely customer requirements 15 to 20 years hence. This is a daunting task given the typical Internet product lifetime of about 2 years. Based on this, a linear transponder, for all its analog antiquity, will likely remain the best satellite resource for the foreseeable future. Further, the simplicity of the network architecture for GEO systems over those required for non-geostationary satellite systems, argues for linear transponders on GEO satellites as the most cost effective solution for almost all commercial undertakings. In the same way that the hub and VSAT interface units for transmission emulate terrestrial packet connections over the satellite, the virtual containers in terrestrial ATM connections must be converted to satellite virtual packets (SVPs) for transmission through a satellite OBP payload. The SVPs will be a common baseband element at the link layer of the ISO/OSI stack. The SVP approach permits fast packet switching architectures to be used and builds on the terrestrial protocols and applications that have already been developed for ATM systems. It also permits application specific integrated circuit (ASIC) chips that are being employed for ATM networks to be used in the user terminals. MFTDMA uplink access protocols will be the typical multiple access technique used and multi-carrier demodulation, demultiplexing, and decoding (MCDDD) of the incoming data stream will precede the OBP payload on the satellite. Since the direct-to-home (DTH) Internet services from these satellites will be at Ka band, they will suffer appreciable degradations in rain. The OBP satellite payload will therefore have capabilities to monitor the incoming bit stream and request changes in power or modulation to counteract the rain impairments detected. These requests are sent to the individual VSATs or to the network control station for onward routing to the user. There is a large potential market for two-way Internet connections to the home by satellite. To be successful, the service must offer high data rates in both directions, at a cost that is acceptable to the customer.

9.10

SUMMARY

Very small aperture terminals (VSATs) have become a part of everyday life around the world. In highly developed regions, they act as links in the retail chain, taking point of sale data from automated terminals (e.g., a gas pump) from the customer to the credit card authorization center. In all regions of the world they have become the fastest growing segment of video distribution: the direct-to-home (DTH) receivers in the direct broadcasting service (DBS) using geostationary satellites. Soon, DTH terminals will be two-way links via satellites in the Global Information Infrastructure, better known as the GII or simply the Internet. The development, and more importantly the universal acceptance, of VSATs required a number of technological breakthroughs, the most important of which were digital compression techniques and very high-density integrated circuits. It can be argued that the remarkable success of

DBS-TV would have been impossible without MPEG 2. In a like manner, digital signal processing and the integration of many functions into one chip set have made DTH terminals a practical reality. Whether the Internet DTH terminals will be a commercial success remains to be seen, as many factors have to be balanced in this endeavor. Some of these factors are relatively straightforward to ascertain: link margins, capacity allocations, and trade-off between outbound and inbound links. Others are more subjective, such as customer acceptance of outages at Ka band and the ability of onboard processing (OBP) payloads to adapt to changes in traffic mixes over the lifetime of a particular satellite system. Nevertheless, VSATs will always be a major part of every satellite system, growing in importance as new enterprises seek to provide multimedia streams directly to customer premises.

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REFERENCES 1. Satellite Communications: the First Quarter Century of Service, DAVID W. E. REES, John Wiley & Sons, New York, 1989 ISBN 0-471-62243-5. 2. Intelsat Earth Station Standards are available from Intelsat, 3400 International Drive, NW, Washington DC, 20008-3098. They are also available on the web at http://www.intelsat.com 3. Intelsat VSAT Handbook, September 1998. Available from Application Support and Training, Intelsat, 3400 International Drive, NW, Washington DC, 20008-3098. 4. “VSAT Systems and Earth Stations,” Supplement No. 3 to the Handbook on Satellite Communications, International Telecommunications Union, Geneva, 1994 (for updates on this handbook, please refer to http://www.itu.int). 5. D. RAYCHAUDHURI and K. JOSEPH, “Channel Access Protocols for Ku-Band VSAT Networks: A Comparative Evaluation,” IEEE Communications Magazine, Vol. 26, No. 5, pp. 34–44, May 1988. 6. Multiple Access Communications: Foundations for Emerging Technologies, N. ABRAMSON, ed., IEEE Press, New York, 1993. 7. H. D. CLAUSEN, H. LINDER, and B. COLLINI-NOCKER, “Internet over Direct Broadcast Satellites,” IEEE Communications Magazine, Vol. 37, No. 6, pp. 146–151, June 1999. 8. Recommendation ITU-R S.728, Maximum Permissible Levels of Off-Axis E.I.R.P. Density from Very Small Aperture Terminals (VSATs), 1992.

9. J. E. ALLNUTT, Satellite-to-Ground Radiowave Propagation, Peter Perigrinus Ltd. (for the IEE), 1989. 10. L. CASTANET, J. LEMORTON, and M. BOUSQUET, “Fade Mitigation Techniques for New SatCom Services at KuBand and Above: A Review,” COST 255 First International Workshop on Radiowave Propagation Modelling for SatCom Services at Ku-Band and Above, WPP-146, pp. 243–251, October 1998. 11. H. VASSEUR, M. CZARNECKI, L. CASTANET, and M. BOUSQUET, “Performance Simulation of a Ka-Band VSAT Videoconferencing System”, COST 255 First International Workshop on Radiowave Propagation Modelling for SatCom Services at Ku-Band and Above WPP-146, pp. 227–234, October 1998. 12. INTEL-1474 Final Report, Demonstration of Advanced Networking Concepts, COMSAT Laboratories, February 1997. 13. A. W. DISSANAYAKE, “Application of Open-Loop Uplink Power Control in Ka-Band Satellite Links,” Proceedings of the IEEE, Vol. 85, No. 6, pp. 959–969, June 1997. 14. D. G. SWEENEY and C. W. BOSTIAN, “Implementation Adaptive Power Control as a 3020 GHz Fade Countermeasure,” IEEE Transactions on Antennas and Propagation, Vol. 47, No. 1, pp. 40–46, January 1999.

PROBLEMS 1. a. What does the acronym VSAT stand for? b. What is the typical range, in meters, of the aperture diameter for a VSAT operating with a Ku-band satellite? c. As a direct broadcast satellite service (DBSS) operator, you want to identify the appropriate receive antenna to use in the home market. Calculate, and set down in tabular form, the gain (in dB) and 1-dB beamwidth of the following antenna diameters: 0.1, 0.2, 0.3, 0.4, 0.5, 0.6, 0.7, 0.8, 0.9, and 1.0 m. Assume a frequency of 12 GHz, an antenna efficiency of 55%, and that the 1-dB beamwidth is half that of the 3-dB beamwidth. d. If users are able to point their antennas to within

0.5° and require a minimum gain of 30 dB, what antenna diameter range is available to the users? e. Given this acceptable range of antenna diameters, which one of these antenna diameters would you choose, stating your reasons? 2. a. Explain in your own words what “leapfrog technology” is.

b. Give three examples of leapfrog technology. A country with an emerging economy is seeking to increase the communications capability in its interior, which, for the present, lacks a significant terrestrial communications infrastructure. They plan to do this, in part, with a VSAT/WLL architecture. A typical VSAT will handle a two-way T1 stream (1.544 Mbit/s), which is capable of incorporating 24, 64 kbit/s digital voice/data channels. c. If a linear satellite transponder, SCPC approach is used, what RF bandwidth will this VSAT T1 stream require on the satellite? Assume no FEC is used, a root raised cosine filter roll-off factor   0.3, and QPSK modulation is employed. d. If half rate FEC is used, what is the occupied satellite bandwidth now? e. What is the noise bandwidth in cases (a) and (b) above? f. If realistic guard bands on the satellite are assumed, and ignoring satellite power issues, how many T1

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streams can be handled by a 72-MHz transponder in cases (a) and (b) above? (Note: state your guard band requirements clearly.) g. If more than 24 channels are required by the WLL through the VSAT, how would you go about accomplishing this if (i) these are all voice channels; (ii) these are a mix of voice and data channels; (iii) these are all data channels? 3. a. Explain what Mesh and Star architectures are in a VSAT network. b. Give two advantages and disadvantages of the Mesh and Star architectures. c. Give the three major types of multiple access schemes that are used in satellite systems. d. Which one of the three multiple access schemes identified in (c) is the closest to the ALOHA multiple access scheme? e. What are the advantages and disadvantages of an MF-TDMA access scheme from a system perspective (i.e., how does this access scheme affect the earth terminal and the satellite payload design)? f. What does MCDDD mean and how will it affect the design of the satellite payload? g. Why has a TDM approach been adopted for most downlink applications for digital VSAT and Internet applications to small terminals? 4. a. What do the symbols ACK and NAK mean when applied to a packet switched communications system? b. What is meant by the term window when used with packet communications systems? c. What does the term spoofing mean when applied to the interface between dissimilar networks. An Internet service provider (ISP) is evaluating a couple of non-geostationary satellite orbit (NGSO) constellations that are being designed to provide global Internet access directly to small office/home office (SOHO) terminals located on the SOHO premises: Skybridge and Teledesic (see Table 10.7). Both of the NGSO systems have opted for orbital altitudes of approximately 1400 km, but there the similarities end. Teledesic employs an onboard processing payload and an end-to-end system architecture using intersatellite links (ISLs) between the satellites to complete the network. Skybridge employs a bent pipe approach on the satellite with the long-distance component being carried over the terrestrial network, and just the end elements of the network employing the satellites (similar to Figure 10.28). In the question below, the information is for illustrative purposes only for use in this example and does not reflect in any way the true system parameters of either of the two systems.

Assume the following: • Altitude of satellites is 1400 km • Minimum operational elevation angle is 40° • Delay introduced by bent pipe satellite is 0 ms • Delay introduced by any switching element (ground or space) is 15 ms • All antennas track accurately (earth terminal and ISL antennas) • Gateway earth station located relatively close to SOHO terminals d. What is the propagation path delay from a terminal on the ground to one of the NGSO satellites when the satellite is viewed at the lowest permissible elevation angle? e. If a typical trans-Atlantic link via Teledesic requires three satellites in the link (up to the first satellite, across to the second, over to the third satellite and down to the end user terminal from the third satellite) and the user terminals at both ends operate at 40° elevation angle, what is the total one-way signal delay if the satellite to satellite to satellite path is 6000 km? f. What is the one-way signal delay for the Skybridge system between the same two terminals in (e) above if the terrestrial link is 8000 km and the satellites are at 40° el. angle to the gateways? g. If no “spoofing” is performed at the satellite segment–earth segment interface, what is the minimum window size required for the Teledesic system in (e) and the Skybridge system in (f), ignoring the length of the messages being sent? 5. When power and bandwidth issues have been optimized, the fundamental limitation for most wireless systems is nearly always interference. Interference can be caused deliberately, as in the jamming of an opposing entity’s signals, or unintentionally, as in a mispointed antenna (due to high wind) or an inadvertent increase in amplifier power (due to operator error). These are usually classified as short-term interferers and there is usually no way to protect a system against such interferers other than to clear them down. Of more interest to commercial systems is the potential for long-term interference caused by nearby systems. A Ku-band VSAT system is being designed for a new service to be offered to two-way SOHO terminals close to a major urban center. One equation used to determine the maximum EIRP permitted in any 40-kHz band at an angle  off the main-beam axis between 2.5° and 7° is given by 33  25 log  dBW. This equation is generally used for satellite spacings of 3°. a. Using the above off-axis equation, what is the maximum off-axis EIRP permitted 3° from the antenna main beam axis?

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There is a range of antennas being considered for the SOHO 14-GHz uplink. Assuming an antenna efficiency of 55%, b. What is the on-axis gain value, in dB, for antennas having antenna aperture diameters of 1, 0.8, 0.6, 0.4, and 0.2 m at a frequency of 14 GHz? c. What are the 3-dB beamwidths of the antennas in part (b) above? If we assume that the 6-dB beamwidth is 1.5  the 3-dB beamwidth, the 10-dB beamwidth is 2  the 3-dB beamwidth, and reasonable interpolations can be used for beamwidth values close to these, d. What is the gain of each of the antennas in part (b) 3° away from the main beam axis? e. If there are three choices of output amplifier, 1, 0.5, and 0.1 W, and assuming no losses between the output amplifier and the antenna, what is the EIRP value on-axis for each of the antennas in part (b) using these three possible output amplifiers? f. What is the EIRP value of each of the antenna plus amplifier combinations in part (e) 3° away from the main beam axis? g. Which of the antenna and amplifier combinations above meet the off-axis limitations of part (a)? h. What additional EIRP allowance needs to be made if the system must now meet the interference requirements for satellites spaced 2° apart rather than 3° apart as in parts (a) through (g)? 6. Many VSAT systems will operate close to acceptable long-term interference limits, both in terms of the interference they can tolerate from similar nearby systems and the interference they cause to similar systems nearby. It is also very possible that many of these VSAT systems will operate close to the performance and availability minima acceptable to the services being offered. There are a number of techniques that may be used to increase the margin available in such cases, one of them being uplink power control (ULPC or UPC). The amplifier power is increased during those periods when rain attenuation occurs in the path so that the received CN remains the same. Increasing

387

the EIRP incorrectly may violate agreed interference limits. A Ka-band VSAT SOHO terminal employs a fixed increment of uplink power control under rain fading conditions. In clear sky, the EIRP is at its nominal level; under rain fade conditions, 7 dB of additional EIRP is switched in as a single-step command. a. If the VSAT uplink EIRP in clear sky conditions operates 3 dB below the agreed interference threshold, what is the smallest rain fade level at which the ULPC may be switched on to provide the fixed increment of 7 dB of additional EIRP without violating the interference limit? In this part of the problem, assume no error in measuring the rain fade level or setting the EIRP level. b. If the rain fade measurement accuracy is 0.5 dB, what is the revised answer to part (a)? c. If the rain fade measurement accuracy is 0.5 dB and the EIRP level may only be set to an accuracy of

0.5 dB, what is the revised answer to part (a)? d. Some ULPC systems require the uplink signal to be detected and measured on the satellite and a downlink data channel from the satellite contains the required ULPC information for the VSAT SOHO terminal. If the round-trip delay, which includes all of the propagation, processing, and routing delays, is 2 s and the maximum rain fade is 1 dB/s, how would this change your answers to parts (a), (b), and (c)? e. Interference rules permit the EIRP commanded by ULPC systems to exceed the long-term interference limits for small time intervals. If (i) this interval is 60 s; (ii) we assume a worst-case scenario of the path attenuation going from a very large value to zero attenuation in zero seconds (which happens occasionally due to intermittent accidental blockage of the antenna aperture); (iii) the limit of the path attenuation measuring equipment has been set so as to accommodate the rain attenuation measurement accuracy, the ULPC level measurement accuracy, and the ULPC amount (7 dB), what is the minimum time constant of the path attenuation measuring equipment that is required to enable the interference criteria to be met?

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CHAPTER

LOW EARTH ORBIT AND NON-GEOSTATIONARY SATELLITE SYSTEMS For 6 years after the launch of Sputnik 1 on the night of October 4/5 1957, all of the space vehicles launched were either inserted into low earth orbit (LEO) or they were small scientific research vehicles dispatched on missions to the earth’s moon or to the inner planets: Mars and Venus. The primary reason for the use of LEO was the generally small throw mass of the launchers then in use. The throw mass of a launcher is the mass of the spacecraft that is ejected from the spent rocket. The throw mass includes both the payload for the mission and the spacecraft bus system within which it will function. It also includes any additional rocket motors and fuel that will be needed to boost the spacecraft into the final trajectory and/or to maintain that trajectory. While the throw mass of the launchers used by the former USSR in that period was significantly higher than that of the U.S. launchers, political reasons dictated their capabilities be used in noncommercial areas. It was therefore left to the United States to open up the geostationary (Clarke) orbit1 in 1963 with Syncom 1 and with it, the commercialization of space. The Clarke orbit, more usually called the geostationary earth orbit (GEO) or geostationary satellite orbit (GSO), is a unique resource that has enabled the generation of many billions of dollars in revenues per year from communications satellites and the associated launchers. Communications satellites and their launchers were the only commercial space ventures in the twentieth century that had any significant return on investment. This may change with the new generation of non-geostationary satellite orbit satellite constellations currently under development, in deployment, and in operation for a variety of commercial ventures, although the first ventures have unfortunately been conspicuous failures. The terms low earth orbit (LEO) and medium earth orbit (MEO) are generally used for specific orbit altitude ranges, for reasons that we will see later. LEO satellites are confined between an upper orbit altitude of about 1500 km and a lower orbit altitude dictated by atmospheric drag (generally around 500 km). MEO satellites have a lower orbit altitude of around 1500 km and an upper bound set by the GEO altitude of around 36,000 km. Most MEO systems, however, orbit in the 10,000 to 15,000 km range. LEO and MEO satellites—now generally referred to as non-geo-stationary orbit (NGSO) satellites—have been used in a variety of roles. From an era in the late 1950s when every launch made front-page news we have now become somewhat blasé about satellites: they have become part of everyday life, much like computers and the Internet. NGSO satellites brought us the first communications satellite (SCORE), the first pictures of our cloud cover for weather forecasting (TIROS), the first navigation aids in space (TRANSIT), the first live television pictures across oceans (TELSTAR), the first Geographic Information Systems pictures of the earth (SPOT), the first infrared, ultraviolet, and X-ray view of the universe from outside the earth’s atmosphere and, of course, the first manned missions (Vostok and Mercury). Each of these missions has been succeeded 388

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by more complex satellites with more advanced capabilities: soon the International Space Station (ISS) will assume scientific missions accompanied by one, or more, free-flying modules and, for the first time, watchers on the earth with their naked eye in daytime may be able to see a satellite as it passes overhead. Already, in early 2002, the ISS had an orbital mass in excess of 250,000 lb and dimensions close to a Boeing 747. As the satellite missions became more complex, the requirements for the specific orbits became more precise. Some satellites have to be very close to the earth, some in highly elliptical orbits, and yet others in orbits with a plane that matches the view angle to the sun. This chapter reviews the different earth orbits available and what missions may use them to advantage.

10.1

INTRODUCTION The geostationary orbit has been the preferred orbit for satellite communication systems for 35 years, and is likely to continue to be the orbit that provides most of the revenue for satellite system operators. The reason is simple: more bits can be sent per dollar of capital investment when a satellite is in a geostationary orbit than in any other orbit. This was realized quite early in the development of satellite communications, and Intelsat, which was the first provider of commercial satellite systems, developed a series of geostationary satellites, beginning in 1965 with Early Bird (INTELSAT I). Commercial and national satellite systems followed in the 1970s and 1980s, all using GEO satellites. Directto-home (DTH) satellite television broadcasting, one of the most successful applications for satellite communication systems, also requires GEO satellites so that customers can use small fixed dish antennas. In such a DBS-TV system, the major investment is in earth stations, not in the satellite. Ten million earth stations bought for $250 each, for example, cost $2.5 billion, well in excess of the cost of a cluster of GEO DBS-TV satellites. There are some specialized applications that require non-geostationary satellites. Surveillance of the earth’s surface, for both military data gathering and earth resources applications, requires satellites in low earth orbit that cover the entire surface of the earth. Satellites providing global navigation, such as the Global Positioning System (GPS) constellation, must utilize orbits that place the satellites in widely spaced positions in the sky, as seen by the receiver. Some of the satellites can be in GEO, but most must be in inclined orbits with an even distribution over the earth’s surface. GPS uses 24 satellites in orbits with an altitude of 20,000 km and an inclination of 55°. Mobile satellite communication systems demand an earth station with a low gain antenna that has a near omnidirectional pattern. A GEO satellite used for communication with a satellite telephone that is handheld, like a cellular telephone, requires a very large antenna with hundreds of beams to achieve a very high gain. The high gain satellite antenna is needed to compensate for the low gain of the antenna employed by the user’s telephone handset. An alternative to a GEO satellite with a high gain antenna is a LEO or MEO satellite constellation with a smaller multibeam antenna. Because the satellite is not geostationary, a large number of satellites is required to maintain continuous coverage. The Iridium system used 66 satellites in LEO, for example, to provide continuous global coverage. Building, launching, and maintaining a constellation of communication satellites in low earth orbit is expensive. When low earth orbit satellite constellations were first proposed for mobile satellite services, the satellites were envisaged to be small, simple, and low cost compared with GEO satellites. Early estimates for the cost of the Iridium system, for example, were between $1 billion and $2 billion. As the development of the LEO systems progressed, the satellites became more and more complex and their cost steadily

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increased, becoming comparable to the cost of GEO satellites. The satellites used by the ICO global system (now called New ICO), for example, are actually modified versions of a large GEO satellite, the Hughes (Boeing) 601 design. Since any LEO or MEO system requires many more satellites than a GEO system serving the same region, the cost of the LEO or MEO system will exceed the cost of the equivalent GEO system. The Iridium system, with 66 satellites in low earth orbit, eventually cost over $5 billion, compared with a typical cost of $250 million to launch and maintain a large GEO satellite. Iridium failed as a commercial venture because the final cost greatly exceeded initial projections, and the system was unable to attract a sufficient number of customers quickly enough. Debt repayments on the high capital cost of the system came due before the customer base had built to a large enough size to provide substantial revenue. However, analysis of the cost per bit transmitted through an Iridium satellite shows that it is much higher than the cost per bit for a GEO satellite, and any LEO system must therefore be able to offer considerable advantages to its customers over that of an equivalent GEO system if it is to succeed commercially. It remains to be seen whether the other nongeostationary mobile satellite systems can succeed where Iridium failed. As the twentyfirst century starts, prospects do not look bright for NGSO systems with Iridium, Orbcomm, Globalstar, and ICO filing for bankruptcy protection. ICO has emerged from the Chapter 11 filing as New ICO, but is still struggling with identifying its mission. Initially conceived as a mobile satellite system provider, emphasis has moved to the provisioning of Internet-like service to mobile customers as a preliminary to Teledesic. This chapter discusses a number of applications and satellite systems that are not in GEO orbit, beginning with those in simple, circular, equatorial orbits; moving through simple inclined orbits to those with high eccentricity; and then reviewing those that take advantage of specific attributes of their orbit for observations (sun synchronous orbits) or the provisioning of navigation services through half-sidereal periodic orbits (GPS). The socalled inclined orbit GEO satellites are not discussed in this chapter. These satellites, once fully stationary GEO satellites, have had their in-orbit operational life extended by removing station keeping in the N–S direction while maintaining E–W station keeping so that the average subsatellite point remains nominally the same. Such inclined-orbit operation was first started after an unusual run of launch vehicle failures in the 1986 time frame when every single type of commercial satellite launcher failed (including the tragic loss of the space shuttle Challenger). The up to 2-year hiatus in some satellite replenishment programs forced inclined orbit operation of GEO satellites on all service providers. Currently, such

SIDEBAR The first spacecraft launches relied on terrestrial radar tracking and guidance commands transmitted from the ground. This, and the relatively crude control capabilities of the rockets themselves, dictated relatively wide error bounds for the intended orbit. Indeed, achieving orbit in those early days—any orbit—was declared a success! Rapid advances in rocketry, which included the ability for multiple restarts of high-energy upper stage engines, and the inclusion of sophisticated onboard guidance computers, quickly enabled spacecraft mission planners to design with some confidence orbits that were mission-specific. That is, the mission

could not be successful unless the designed orbit was achieved within the specified tolerance. In some cases, the mission was for a single spacecraft (such as a meteorological satellite) while, in others, a constellation of spacecraft would be required to achieve the mission goals. In all cases, careful analysis of the mission goals led to the selection of a particular orbit altitude, ellipticity, and inclination and system architecture (number of satellites, number of planes, spacing of satellites within the plane, connectivity, etc.). Quite often, tight launch windows were also dictated—specific time periods when the launches had to be executed.

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problems are avoided by designing the orbital maneuvering life (OML) to be many years longer then the orbital design life (ODL). This aspect is discussed in Chapter 2. In the sections that follow, we will examine the parameters that need to be determined in the selection of an orbit that will achieve given mission goals. Only earth orbit missions are considered; spacecraft missions requiring escape velocity from the earth are beyond the scope of this book.

10.2

ORBIT CONSIDERATIONS Once in orbit, the motion of a satellite is determined by orbital mechanics, as discussed in Chapter 2, Orbital Mechanics and Launchers. However, while the satellite moves in such a way as to balance centrifugal and centripetal forces, the earth is also in motion beneath it. As well as rotating once a sidereal day, the earth also moves around the sun; and the solar system, with the sun at its center, is orbiting around the center of the home galaxy, the Milky Way. There is therefore a complex relationship between the various motions of the natural and artificial bodies. How many of these need to be considered simultaneously will depend on the design goals of the satellite system. A satellite designed to observe the earth’s surface will not need to know where the stars are at any particular time, but the location of the local star, the sun, may be important if the satellite needs to use sunlight to illuminate its coverage region on the surface of the earth. On the other hand, a satellite designed to observe background thermal radiation levels of deep space in the infrared band will need to know the position of each of the neighboring planets. Should the telescope of the satellite inadvertently point toward one of these planets, the temperature viewed would not reflect that of the true background radiation level. In the sections that follow, we will review all of the different NGSO orbits that have been used for scientific, military, and commercial satellite missions. The simplest NGSO orbit is an equatorial orbit.

Equatorial Orbits Equatorial orbits lie exactly in the plane of the geographical equator of the earth. That is, the orbital path lies directly above the equator at all times. In order to take advantage of the 0.45 km/s eastward rotational velocity of the earth, most satellites are launched toward the east into a prograde orbit. A westerly directed orbit is called a retrograde orbit. A satellite in an eastwardly directed equatorial orbit will have two periods: a real orbital period that is referenced to inertial space (the galactic background) and an apparent orbital period that is referenced to a stationary observer on the surface of the earth. The real orbital period, denoted here as T hours, is given by Eq. 2.6. The apparent orbital period to the observer on the equator will be P hours where P  124T2  124  T2 hours

(10.1)

To be exact, 23.9344 h, one sidereal day, should be used in place of 24 h in Eq. (10.1). Table 10.1 (from reference 2) illustrates the difference between P and T for a number of orbital altitudes and elevation angles. It also shows the time the satellite is visible to the observer, neglecting atmospheric refraction and assuming the satellite can be tracked down to 0°, that is, right down to the horizon. Other implications of the observing time are considered in more detail in Section 10.3, Coverage and Frequency Considerations. The plane of a satellite’s orbit must be in the plane of the equator for the satellite to be in equatorial orbit. This can be achieved by launching the satellite in one of two ways. The first launch method is to locate the launch site on the equator and to launch

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TABLE 10.1

Orbital Periods and Observing Time Orbital period

Orbital height (km)

True (hours)

Apparent (hours)

Observing Time (hours)

500 1,000 5,000 10,000 35,786

1.408 1.577 1.752 5.794 23.934

1.496 1.688 1.890 7.645 

0.183 0.283 0.587 2.894 

Source: From Table 1.1 in reference 2.

the spacecraft toward the east along the equatorial plane. The second method is to launch the satellite into an inclined orbit and to execute a maneuver either during the launch trajectory or when the satellite is in an inclined orbit that changes the plane of the initial orbit so that the final orbit is in the plane of the equator. Removing the inclination from the orbit, so that the satellite orbits exactly over the equator, requires significant energy, particularly if the launch site is well removed from the equator. The first two sites from which orbital flights were made, Cape Canaveral in the United States and Baikonur in Kazakhstan, were not close to the equator (approximately 28° N and 46° N, respectively). In addition, the early launch vehicles lacked the ability to alter the trajectory significantly during launch. The first artificial earth satellites were therefore placed into inclined orbits, that is, the planes of the orbits were inclined to the equatorial plane.

Inclined Orbits There are advantages and disadvantages to inclined orbits, depending on the mission goals and the data recovery requirements. The greater the inclination of the orbit is, the larger the surface area of the earth that the satellite will pass over at some time in its flight. Figure 10.1 illustrates this for a LEO satellite. In Figure 10.1b, the inclined orbit will take the spacecraft, at one time or another, over the earth’s entire surface that lies approximately between the latitudes given by  the orbital inclination. For example, an orbit with an inclination of 30° will cover all regions that lie approximately between latitudes 30° north and 30° south. The superior coverage of the earth with an inclined orbit satellite is counterbalanced by the disadvantage that the master control station (MCS) will not be able to communicate directly with the satellite on every orbit as with an equatorial orbit satellite. A LEO satellite orbits the earth with a period of 90 to 100 min and, for an inclined orbit satellite, the earth will have rotated the master control station out of the path of the satellite on the next pass over the same side of the earth. Depending on the quantity of data that need to be passed to the MCS, or if real-time communications are required continuously, a system architecture that employs multiple satellites will need to be considered. The simplest, and lowest cost, solution to pass data between an inclined orbit satellite and an MCS is to design the satellite to store the data acquired over many orbits (when it is out of sight of the MCS) and then, when it passes within radio range of the MCS, to dump the data rapidly to the MCS. This is called store-and-forward and it is one of the capabilities of some LEO systems, including Orbcomm satellites3. It was also the technique used for the very first communications satellite, Project SCORE, in December 1958. In the

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N Coverage of LEO satellite

Equatorial LEO satellite

S (a )

N Coverage of inclined orbit LEO satellite (shaded area)

Inclined orbit LEO satellite

S (b )

FIGURE 10.1 (a) Coverage of an equatorial orbit LEO satellite. The LEO satellite is in an equatorial LEO orbit and so it will only pass over the equator. The coverage of the equatorial LEO satellite will therefore be limited to a swathe of the earth close to the equator, determined by the height of the orbit and the beamwidth of the satellite’s antenna. In this example, the orbit is assumed to be circular and the antenna beamwidth has been ignored. (b) Coverage of an inclined orbit LEO satellite. The LEO satellite is in an orbit that is inclined at approximately 40° to the equator. The satellite will therefore pass over, at one time or another, all regions of the earth between latitudes 40° N and 40° S of the equator. The coverage of the inclined orbit LEO satellite will therefore be a swathe of the earth between about 40° of the equator, determined by the height of the orbit and the beamwidth of the satellite’s antenna. In this example, the orbit is assumed to be circular and the antenna beamwidth has been ignored. Note: The higher the orbit and the greater the inclination, the further the satellite’s total coverage will reach.

Orbcomm system, if a user on the ground is unable to establish contact via an Orbcomm satellite to a gateway earth station (GES) in the Orbcomm system, a “GlobalGram®” may be left stored within the satellite for later transmission to the GES when it comes into view of the satellite. The downlink transmission rate must be high enough to enable all of the stored messages in a LEO satellite to be sent to the MCS in the period when it is within range of the satellite. If a continuous, real-time connection is required between a LEO satellite and the MCS, there are only two approaches that can be used. • The first approach is to locate control stations around the world so that the LEO satellite is never out of sight of at least one of the control stations. Terrestrial or GEO satellite connections are then established between the many control stations and the MCS to bring the LEO data back to the MCS in real time.

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• The second approach is to establish intersatellite links (ISLs) to relay the LEO data traffic back to the MCS. The ISLs can either be set up among the LEO satellites in the constellation, so that the LEO data traffic is relayed between the LEO satellites in orbit via their ISLs, or the ISL link can be set up between the LEO satellite(s) and one or more GEO satellites. The GEO satellite relays the LEO data traffic back to the MCS directly, if it is within sight of the MCS, or via another GEO satellite. Iridium4 and the early design of Teledesic5 adopted the former solution of LEO satellites interlinked via ISLs within their own constellation. Skybridge6 and Globalstar7 chose a different approach. NASA uses the TDRSS satellites for shuttle missions (see sidebar). Geostationary relay satellites have also been used for military reconnaissance missions by at least the United States and Russia—transmitting onward data received from LEO observation satellites—and a similar system was in its implementation stage at the end of 1999 for some civilian earth observation missions. Figure 10.2 illustrates the two concepts. In both of the examples shown in Figure 10.2, the LEO satellite is in a circular orbit. A circular orbit gives a constant dwell time over a given coverage region since the angular velocity of the satellite is the same at any point in the orbit. In many cases, mission goals will dictate different dwell times for different parts of the orbit. A circular orbit will not achieve this result. To accomplish variations in dwell time around the orbit, the satellite must be in an elliptical orbit.

Elliptical Orbits As noted in Chapter 2, an elliptical orbit will have a nonzero eccentricity. The orbit eccentricity, e, is determined by the lengths of the semimajor axis, a, and the semiminor axis, b, of the orbit ellipse e2  1  1b2 a2 2

(10.2)

Alternatively, if Ra is the distance between the center of the earth and the apogee point of the orbit and Rp is the distance between the center of the earth and the perigee point, the eccentricity is e  1Ra  Rp 2  1Ra  Rp 2

(10.3)

Figure 10.3 illustrates the geometry of Eq. (10.3). In Eqs. (10.2) and (10.3), if the orbit is exactly circular, a  b and Ra  Rp, and the eccentricity reduces to zero. In general, no orbit is truly circular for a variety of reasons,

SIDEBAR NASA built and operated a number of relay stations for the Mercury, Gemini, and Apollo programs. In none of these missions was the manned spacecraft ever out of real-time contact with the Manned Spaceflight Center in Houston, United States (which acted as the MCS in this case) except when the Apollo craft was behind the moon or in the re-entry phase where ionized plasma caused a radio blackout for all spacecraft.

Communication with the Space Shuttle is maintained using the Tracking and Data Relay Satellite System (TDRSS). Several TDRSS satellites in geostationary orbit relay data from the Shuttle to several earth stations around the world that then send the data to NASA’s MCS for manned space flight in Houston, Texas.

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Master control station Inclined orbit LEO satellite

Ground tracks of satellite on successive orbits

(a )

Inclined orbit LEO satellite

Master control station

GEO satellite (b)

FIGURE 10.2 (a) Store-and-forward concept. In this LEO application, the satellite stores information it has gathered while orbiting the earth and, once within range of the master control station, it downloads the stored data. The (uplinked) data storage rate is usually low, a few kbit/s at most, while the download is at a much higher rate due to the small time the satellite has available when it is within range of the master control station. (b) Real-time data transfer via a GEO satellite. In this approach, the LEO satellite can transfer data in real time via the GEO satellite to the master control station whenever it can “see” the GEO satellite. If there were a number of GEO satellites equipped with intersatellite links (ISLs) distributed around the geostationary orbit, then the LEO satellite need never be out of real-time contact with the master control station.

but eccentricity values of 103 or less can be considered to correspond to circular orbits for all practical purposes. The eccentricity is another way of describing the variation in the radius of the orbit. If Rav is the average radius of an orbit from the center of the earth, then the variation, R, in the orbital radius, is given by8 ¢ R  eRav

(10.4) 4

For a geostationary satellite (Rav  42,164.17 km) with an eccentricity of 10 , R will be 4.2 km. For a LEO constellation with a circular orbit of approximately 800 km above the earth, with each LEO satellite having an eccentricity of 104, R will be 0.7178 km

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Satellite

Perigee

Apogee

Earth

Rp

Ra

FIGURE 10.3 Schematic of an elliptical orbit illustrating ellipticity. The satellite orbits the earth with a perigee distance from the center of the earth given by Rp and an apogee distance from the center of the earth given by Ra. Note that the perigee and apogee are always exactly opposite each other in the orbit. This is true of any object in any orbit around any other body.

(assuming the earth mean radius is 6378.137 km). If the orbit becomes less circular and the eccentricity increases to 103, R increases to 7.178 km. If the LEO satellites in the constellation pass over (under) each other, then the vertical separation must be sufficient to prevent any likelihood of a collision between satellites. The average orbital altitude and eccentricity of the orbit will determine the likelihood of a collision. One of the more famous orbits has an eccentricity 0.74. This is a special case of a highly elliptical orbit (HEO) known as the Molniya orbit.

Molniya Orbit The former Soviet Union had a difficult communications design problem. Much of the landmass is in far northern latitudes. Archangel, the port on the White Sea, is close to latitude 60° N; immense tracts of Siberia lie inside the Arctic Circle. To compound the problem further, the country was spread across 11 time zones: it was the largest country in the world (and Russia still is). The signals from a geostationary satellite can reach well inside the Arctic Circle if operations at elevation angles below 5° are permitted, but a single GEO satellite cannot reach that far north over 11 time zones simultaneously. A new type of orbit was required to provide good communications coverage over the former USSR. What transpired was the Molniya system. The first Molniya satellite was launched in April 1965 and it gave its name to both the system of satellites and to the unique orbit. The word Molniya means flash of lightning in Russian. The apogee of the Molniya orbit is at an altitude of 39,152 km and the perigee is at an altitude of 500 km. The orbital period is 11 h and 38 min and the orbital inclination is 62.9°. This combination of apogee, perigee, and inclination ensures that the ground track of the Molniya orbit repeats every other orbit. That is, if the orbit passes exactly over Moscow on orbit one, it will do so again on orbit three, five, seven, nine, and so on. Figure 10.4 illustrates the orbit geometry. Two Molniya orbits, with the planes of the orbits separated by 180°, will thus provide coverage over the extreme latitudes of Russia for 24 hours per day using two satellites, correctly phased—one in each of the two Molniya orbits. When one of the satellites

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GEO orbit distance

N

FIGURE 10.4 Schematic of a Molniya orbit. In this example, the trajectory is configured to have a large dwell time over the northern part of the orbit so that it can serve a country that has most of its landmass in this region. This was the design adopted for the original Molniya system of the former Soviet Union. Approximately 60% of this Molniya orbit, which stretches more than 3000 km beyond the height of a GEO orbit, has good look angles for latitudes between 30° N and 90° N. This translates to more than 6 h of the 11 h 38 min orbital period.

is at its apogee over Russia in Molniya orbit 1, the other satellite will also be at its apogee somewhere over North America in Molniya orbit 2. By the time the second satellite has moved once more to its apogee in Molniya orbit 2, the earth will have revolved half a turn under it and Russia will again be spread beneath it. Figure 10.5 illustrates the dual Molniya orbit concept. Satellite 2

Satellite 1

Molniya orbit 1

Molniya orbit 2

N

FIGURE 10.5 Schematic of an operational Molniya system. Satellite 1 in Molniya orbit 1 is providing service over Russia at close to its apogee while the second satellite is also close to its apogee in Molniya orbit 2. Molniya orbits l and 2 are separated by 180° in their orbital planes. By the time satellite 2 has moved around its orbit once and back to its apogee (a period of about 12 hours), the earth will have rotated about 180° and the second satellite will be over Russia.

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The two-satellite, dual Molniya orbit requires that earth station operations be carried out at elevation angles well below 30° for full 24 hours-per-day coverage of one region. Note that, if four satellites are used, one in each of four Molniya orbits that have planes 90° apart, high elevation angle service could be provided to any region at high latitudes in the Northern or Southern Hemisphere. The planes of the four Molniya orbits would be orthogonally distributed around the earth with one satellite in each Molniya orbit, correctly phased in its own orbit to provide coverage from the apogee sections of that orbit as the region rotates beneath the four-satellite constellation. Up to eight Molniya satellites, in eight different Molniya orbital planes separated by 45° and suitably phased around their orbit, have been used to provide continuous coverage over Russia. The Molniya orbit has another advantage for specific services intended for latitudes well away from the equator. The orbit takes the satellite so far away from the equatorial plane and the apogee is so distant from the earth that long dwell times with elevation angles close to 90° can be achieved at high latitudes on the earth. This fact was used in a proposal for a Molniya orbit to deliver mobile satellite service (MSS) to automobiles. A view of the earth, with the plot of the orbit track, abstracted from this proposal9 is shown in Figure 10.6. The Molniya orbit, in addition to the long delay time associated with the communications range when at apogee and the lack of continuous 24-h contact with a single spacecraft from a fixed coverage, also suffers from three drawbacks that increase the overall end-to-end costs. The first is the requirement to track the spacecraft. The second is the need to switch communications to the other Molniya satellite—rather like a mobile radio handoff situation—when the first goes out of coverage as the other comes into coverage. Due to the wideband nature of the traffic and the large angular separation between successive Molniya satellites as seen from one earth station, this requires two reflector antennas at each site. At the end of the twentieth century, phased array antennas still could not provide accurate coincident tracking of both transmit and receive beams simultaneously well away from the (unsteered) electrical boresight over bandwidths that exceed a few percent of the carrier frequency and at a cost that commercial systems can accept. The third drawback to a Molniya orbit is the radiation environment that the satellite has to pass through four times a day—twice on ascent and twice on descent. While the first two drawbacks may be less of an inhibition to commercial success in the long term with direct-to-home services when relatively inexpensive, and efficient, phased array antennas are available that track over the required range of look angles, the third drawback will always be a major factor.

Radiation Effects The effect of radiation on electronics in space is generally separated out into two main aspects10: total dose and single-event upsets. The total dose is simply the cumulative effect of radiation over the lifetime of the electronics in space and is mainly due to trapped electrons and protons in the Van Allen belts. (The Van Allen radiation belts are discussed later in this section.) Eventually, the cumulative effect of radiation will degrade the performance of the transistor junction/chip such that it cannot be relied on to generate the correct responses, etc. This is particularly harmful in the computer elements that control the operation of the satellite and the payload. Single-event upsets are caused by heavy ions ejected from the sun, usually protons, impacting the circuitry at a critical point such that they deposit enough charge to induce an energy (bit) flip, that is, change an open circuit to a closed circuit, create a logical one instead of a logical zero, etc. These single-event upsets are

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(a )

160°

120°

80°

40°

0°

40°

80°

120°

160° 80°

60° 40° 20° 0° 20° 40° 60° 160°

120°

80°

40°

0°

40°

80°

120°

160°

(b)

FIGURE 10.6 View from above the Molniya orbit apogee showing the ground track8. (a) View from the apogee point of a Molniya orbit positioned at almost 0° longitude when at apogee. (b) Ground track of the Molniya orbit shown in Figure 10.6a. Note the two apogees in the orbit, one over close to 0° longitude and the other at close to 180°. The apogee occurs at a high latitude, from which the elevation angles are well above 70° over quite a large region. With these high elevation angles, blockage of buildings would be minimized and thus allow relatively high availability for an MSS system operating to automobiles in most cities. This proposal8 was for a European MSS system, but the apogee could be phased to occur at any longitude so that cities in high latitudes, but arbitrary longitude, could operate to an MSS satellite in Molniya orbit.

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SIDEBAR The Van Allen radiation belts were named after the discoverer of these belts, James A. Van Allen, who was the principal investigator for a radiation measurement package on the United States’ first successful artificial earth satellite, Explorer 1, launched in January 1958.

The radiation belts consist of high-intensity protons and electrons that are temporarily trapped in the earth’s magnetic field. While the trapped electrons can have energies up to 7 MeV (seven million electron volts), the trapped protons can have energies up to 500 MeV10.

more critical if the bit flip is permanent, that is, “latch-up” occurs in a set position from which it cannot be changed. The relative motion between the liquid core and the solid mantle and outer crust above it generates the earth’s magnetic field. The magnetic field lines stretch out around the earth as shown schematically in Figure 10.7. While generally symmetrical close to the earth, the magnetic field lines of the earth become distorted further out from the earth due to interaction with the energy flowing toward the earth from the sun. The boundary where the solar atmosphere and the earth’s magnetic field meet far out in space is called the bow shock, much like the pressure waves concentrating in front of the wing of an aircraft. Since the earth’s magnetic and geographic poles are not coincident, the magnetic equator (and magnetic latitudes) will be different from the geographic equator (and geographic latitudes). The geomagnetic latitude  can be computed from11: f  arcsin 3sin a sin 78.5°  cos a cos 78.5° cos 169°  b2 4

(10.5)

where  is geographic latitude and  is geographic longitude. North and east coordinates are considered positive, and south and west coordinates negative.

Magnetic field lines

NM

FIGURE 10.7 Representation of the magnetic field lines that flow between the north and south magnetic poles of the earth. The earth has a strong magnetic field due to having a liquid core that is spinning at a different rate than the solidified outer shell. The magnetic poles, however, are not coincident with the geomagnetic poles and so the magnetic equator is not located in the same position as the geographical equator. Sometimes the geomagnetic latitudes are referred to as dip latitudes since they will correspond to the dip in the magnetic field at that point. NM is the north magnetic pole.

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The electrons and protons become ensnared in the earth’s magnetic field when their kinetic energy cannot overcome the trapping effect of magnetic lines of force at the given point of encounter in space. Since the magnetic field strength decreases with increase in altitude on a given radial from the center of the earth, only the electrons are trapped in the higher reaches of the earth’s environs (10,000 km altitude above the earth) since the field forces are relatively low at these altitudes. Both electrons and the higher energy protons are trapped lower down in the earth’s atmosphere 200 to 10,000 km10, where the field is relatively more intense. The radiation levels induced by the electrons and protons fluctuate wildly with latitude, longitude, altitude, and with the sunspot cycle.

SIDEBAR Sunspots are disturbances on the surface of the sun. Sunspots appear to generate huge outflows of energy from the sun and the amount of energy closely follows the number of sunspots—or rather groups of sunspots— which can be counted on the surface of the sun. The sunspot count, and hence the level of energy, varies with a mean period of about 11 years, although the actual cycle spans a 22-year Hale cycle as the magnetic field lines associated with the sunspot activity on the sun’s surface reverse every 11 years. The 11-year sunspot

cycle period is not constant. The period has been as short as 9.5 years and as long as 12.5 years12. The first cycle that has been given an official number is the 1755–1766 period. The last full solar cycle of the twentieth century (1986.8–1996.4) was labeled Cycle 22. A schematic of this cycle, showing the large variation in sunspot count that exists, is given in Figure 10.8. The turn of the twentieth century saw us in cycle 23 with the two-to-four year period of peak activity starting in the fall 1998 equinoctial period.

200

Sunspot Number

150

100

50

0

1986

1988

1990

1992

1994

1996

Year FIGURE 10.8 The general variation of the sunspot number over solar cycle 22. The smoothed sunspot number is averaged over several months. The fluctuations in the actual sunspot number are shown about this smoothed average. Not only does the sunspot count vary widely from month to month, it does so also from day to day. The higher the average sunspot number is, the larger the variation in actual sunspot number count is in general. Note the more rapid rise than decline in the average sunspot number count and the fairly long period when the sunspot activity was very high. Because of the “flat” nature of the sunspot maximum period (up to 4 years) it is usual to determine the sunspot periods from their minima.

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Outer Van Allen belt Inner Van Allen belt

N

~15,000 km ~1,500 km FIGURE 10.9 Pictorial representation of the two Van Allen radiation belts. The above schematic is a vertical “slice” through the radiation belts that exist around the equator. The shaded areas are the regions in the two belts where the radiation is at a maximum. The two principal NGSO regions lie “under” the first Van Allen radiation belt—the low earth orbit or LEO region—and “between” the two radiation belts—the medium earth orbit or MEO region—so as to avoid the highest radiation doses. However, radiation never falls to zero and exists in all areas (see reference 10).

The variability of the sunspot cycle leads to large fluctuations in the radiation environment in space. While there are large variations in the radiation environment with latitude, height above the earth, and with orbital inclination, it is normally considered that there are two main Van Allen radiation belts where the effect is more concentrated. The center of the first belt is at a height of about 1500 km above the earth and the second at around 15,000 km, measured around the equator, although these distances are somewhat arbitrary and there is some evidence that the outer belt may actually be two merged belts. The belts can be considered as doughnut-shaped, with the energy at its highest toward the center of the given belt. Figure 10.9 illustrates the concept. The trapped electrons and protons travel northwards and southwards along the magnetic field lines shown in Figure 10.7. They are reflected when they are close to the magnetic poles10 and so, statistically, spend more of their time closer to the equator than the poles; hence the Van Allen belts are positioned around the geomagnetic equator. The closer to the center of the radiation belts a satellite is positioned and the longer it is in space, the higher the total radiation dose becomes. Total dosage for semiconductors that are fabricated using silicon is measured with a unit called the krad(Si). A rad(Si) is a unit of energy absorbed by silicon from radiation and it is equivalent to 0.01 J/kg10. Radiation in near-earth space is highly variable. It changes both with height above the earth and with the inclination of the orbit with respect to the equatorial plane. Since the radiation is concentrated at the equator, satellites that are in equatorial orbits will receive a higher dosage than those that are in polar orbits will. In a like manner, as the orbital height moves from very close to the earth (300 km) outward for the first few thousand kilometers, the radiation dose will increase. Table 10.2 gives some typical examples of total radiation dosage for a LEO satellite designed for a 10-year operational lifetime.

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TABLE 10.2 Typical Total Doses for Various Orbits Orbital height (km) Orbital type (degrees) Polar orbit (90°) Equatorial orbit (0°)

800

1100

2000

30 krad(Si)

100 krad(Si)

500 krad(Si) 2000 krad(Si)

The data are based on a 10-year mission life using silicon-based electronics in a satellite with a 2.5 mm thick aluminum skin. Source: Data extracted from the text of reference 10.

Choosing an orbit that has a reduced level of radiation can therefore reduce the potential for radiation damage. Where this is not possible, then either radiation hardened (rad-hard) devices must be selected for the satellite or suitable shielding employed. Both are expensive options, the former because of the fabrication costs and the latter because radiation shields can be heavy and are nonproductive elements of the payload. Developing electronic devices that can withstand total radiation doses of 1 Mrad(Si) is possible with rad-hard technologies but newer techniques for approaching these levels of radiation hardening of devices will be needed for constellations of dozens of satellites. New, relatively cheap production processes have been shown to provide consistent shielding to 100 krad(Si) total dosage10 and it is likely that such techniques, plus local site shielding with aluminum strips, will be largely employed for the foreseeable future. The same approach is being used for protection against single-event upsets.

Sun Synchronous Orbit A sun synchronous orbit is a special form of low earth orbit where the plane of the orbit maintains a constant aspect angle with the direction to the sun. Some satellite missions require a specific orbit with such a constant relation to the direction of the sunlight. One example is an earth resources satellite that requires a large amount of direct sunlight to illuminate the region below the satellite so that photographs can be taken. This satellite

SIDEBAR With the ever-smaller integrated circuits being developed for flight operations, there is an increased likelihood that the linear energy transfer (LET) that is generated by the heavy ion collision will cause a single-event upset. The potential for an upset depends on the LET generated and the threshold level at which the device will incur a single-event upset. Many spacebound integrated circuits (ICs) have LET thresholds greater than 37 MeV-cm2/mg10, which means that heavy ions with LETs of less than this amount will cause few single-event upsets. Fortunately, ICs can be manufactured relatively inexpensively with LET

thresholds of 37 MeV-cm2/mg and the incidence of heavy ions with a LET exceeding 37 MeV-cm2/mg is very rare10. The potential for latch-up will also reduce as technology advances permit devices to be used that employ a lower drive voltage. In addition, many of the silicon-on-insulator (SOI) and silicon-on-sapphire (SOS) technologies have been found to be immune to latch-up, as there are no parasitic paths10. This is encouraging, as there are some satellites that will require a unique type of orbit that cannot be selected from a radiation dosage perspective. One such orbit is the sun synchronous orbit.

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A

Sunlight

N

B

Sun

Earth

FIGURE 10.10 Examples of two sun synchronous orbits. In the illustration above, the earth is viewed from above the North Pole, N, with the sunlight illuminating the left side of the earth. Two sun synchronous orbits are shown. Orbit B is designed so that it will always have one-half of the orbit with the sun almost directly behind it; orbit A is designed to be always within sight of the sun—the so-called sunset–sunrise orbit since it will be orbiting the terminator. The terminator is the line that divides night from day.

would be in orbit B in Figure 10.10. Another example of a satellite needing this same orbit is a meteorological satellite, where images of the clouds and their directions of motion are critical in developing forecasts. While communications satellites have returned the greatest tangible investment returns for their owners, it is arguable that meteorological satellites have led directly to huge savings in human life, as well as to less property damage and farm animal destruction, in extreme weather situations. The early meteorological satellites (e.g., TIROS) were in LEO sun synchronous orbits, but all recent meteorological satellites are in GEO orbits to provide more instantaneous and continuous coverage. Other satellites that employ sun synchronous orbits are surveillance satellites. Some surveillance satellites use orbit B of Figure 10.10, so that the maximum illumination is provided once per orbit. Others use orbit A of Figure 10.10. This particular “sunset–sunrise” orbit always has the satellite illuminated by the sun while the region below it has the sun at almost grazing incidence. There are two advantages in this orbit. First, the satellite need not have a large battery capacity for eclipse operations since it is always illuminated. Second, since the shadows are so long in the region being surveyed, changes in terrain or structures will be immediately obvious.

SIDEBAR Before synthetic aperture radars were orbited, the sunset–sunrise orbit was used advantageously to detect changes in terrain following natural disasters such as earthquakes. The long shadow of a German V2 rocket allowed it to be detected by a reconnais-

sance aircraft for the first time at Peenemunde toward the end of the Second World War. Similarly, the illfated USSR moon rocket was detected on its launch pad by a surveillance satellite using the shadow it cast.

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B

Alignment error

Movement of earth around sun A

Plane of satellite’s orbit around earth FIGURE 10.11 Illustration of alignment changes of the orbital plane of a satellite due to the movement of the earth around the sun. In the illustration above, a satellite has been launched into an LEO orbit (position A) in which the sun is directly overhead on the sunward side of the orbit. The view is from above the North Pole of the sun, orthogonal to the plane of the earth’s orbit around the sun. When the earth has moved to position B, the plane of the satellite’s orbit—which is fixed in inertial space—now has an alignment error with respect to the sun. If it is essential that the orbital plane of the satellite always be in line with the direction to the sun on a day-to-day basis over a long period, then the plane of the satellite’s orbit will need to change at the same rate that the alignment error is increasing. That is, the plane of the orbit will need to ”precess” to match the movement of the earth around the sun. Sun

Figure 10.11 illustrates how the sun synchronous orbit is achieved. If a satellite is in a perfectly circular LEO orbit over the poles of the earth, a carefully timed launch would put the orbit in such a position that the sun is directly behind the satellite on the sunward side of the first orbit. This is position A of Figure 10.11. However, a short while later, the earth will have moved in its orbit around the sun and the plane of the satellite’s orbit (now in position B) will no longer be aligned with the direction of the sunlight. In order to make the satellite’s orbital plane always keep pace with the apparent change in position of the sun, it must be launched into a retrograde orbit. A retrograde orbit has a velocity component in a westerly direction. In practice, a LEO satellite launched into an orbit with an inclination of close to 98° to the equator (measured counter clockwise from the equator looking east) will move the orbital plane in time to the earth’s movement around the sun. Elliptical orbits with different retrograde inclinations (see the constellation Ellipso in Section 10.5) will also yield sun synchronous orbits. The change (rotation) in the orbital plane is called precession. A key advantage of a sun synchronous orbit is that it will repeat its track every half day. It can therefore be used to make measurements at given times of the day and night so that correlation exercises can be attempted. A sun synchronous orbit will pass over almost all of the earth at one time or another. Determining the instantaneous surface area of the planet seen by the satellite and over which information is required—or to which communications is to be established— is another issue. This portion of the earth’s surface is called the coverage area or coverage region.

SIDEBAR One example of a spacecraft in a sun synchronous orbit was the Mars Explorer spacecraft, which was put into a sun synchronous orbit around Mars in 1998. The orbit was used to measure temperature at 2 A.M.

and 2 P.M. local time equivalents over the same region so that local heating effects and cooling effects could be accurately tracked.

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10.3 COVERAGE AND FREQUENCY CONSIDERATIONS General Aspects In some cases, the designer of a satellite system has few degrees of freedom in designing a payload to provide optimum coverage. This occurs in some missions where a shared spacecraft has to accommodate a number of payloads. Examples exist in the scientific community when (generally) low-cost, multiple missions are being developed for a single spacecraft. In other cases, more freedom exists in the design stage and the mission planners can vary a range of parameters iteratively to arrive at an optimum coverage. The mission goals will directly determine the coverage that has to be achieved by a given satellite system. This in turn leads to the selection of orbit, payload technologies, etc. For example, if a communications satellite system has to provide coverage of the European Union (EU), there is a minimum altitude at which a single satellite can operate and still cover all of the EU at once. If the coverage of the EU must be continuous, a GEO orbit can be selected or a constellation of NGSO satellites can be designed to provide the necessary coverage overlap between successive satellites. The determination of coverage area, while initially an exercise in simple geometry, is eventually heavily influenced by the available technology both on the ground and in space, and other aspects such as the radiation environment. We will consider first the geometrical aspects of determining an optimum coverage. In Figure 10.12, a spacecraft orbits at distance rs from the center of the earth, C. We will assume that the spacecraft is a communications satellite and that it needs to be in contact with an earth station located at E. The elevation angle to the satellite is . Using the sine rule we have 3rs sin 190  u2 4  3d sin 1g2 4

(10.6)

cos 1u2  3rs sin 1g2 4 d

(10.7)

which yields

All three parameters in equation (10.7) have key inputs to the architecture of the satellite system. The angle  will yield the coverage area on the surface of the earth assuming the satellite has a symmetrical coverage about nadir. The distance d will determine the free space path loss along the propagation path, and will be a factor in the link budget design. The elevation angle  will influence the GT ratio of the antenna, the blockage probability from terrain and buildings near the antenna and the likely propagation impairments that will be encountered along the path to the satellite. For systems that operate in frequency bands that suffer significant degradations in rain, the elevation angle can be the critical design element (see Chapter 8 for more details on propagation effects along earth-space paths).

Frequency band Low earth orbit satellite systems providing data and voice service to mobile users tend to use the lowest available RF frequency. The EIRP required by the satellite transponder to establish a given CN ratio in the mobile receiver is proportional to the square of the RF frequency of the downlink, as the analysis in the next paragraph shows. The power that must be transmitted by a mobile transmitter is also proportional to RF frequency squared when the mobile uses an omni-directional antenna. Since the cost of satellites increases as the EIRP of the transponders increases, a lower RF frequency yields a lower cost system. This is one reason why L-band is allocated for mobile satellite services.

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407

S Local horizon at earth station, E, looking toward satellite, S

rs d δ

Z θ γ

E

re

C

FIGURE 10.12 Geometry for calculating coverage area. The satellite, earth station, and the center of the earth are all in the same plane in this figure. SCE is the central angle, , and the elevation angle, , is the angle between the local horizon and the satellite at the earth station in the plane of the figure. The line SC joins the satellite and the center of the earth and cuts the surface of the earth at point Z. To an observer at point Z, the satellite is at zenith. The satellite is at a distance, d, from the earth station and at a distance rs from the center of the earth. The radius of the earth is given by re, a good average value for which is 6370 km.

Consider a LEO satellite with a coverage zone on the earth’s surface that has an area A m2. A transponder on the satellite with output power Pt watts drives an antenna with a linear gain Gt to produce an EIRP from the satellite of PtGt watts. The average flux density across the coverage zone is therefore F  PtGt A watts/m2

(10.8)

The value of the flux density is independent of frequency. The mobile receiver has an antenna that is nearly omnidirectional, with a gain Gr, where Gr is typically less than 3 dB. The effective receiving area of this antenna is given by Ae  Grl24p

(10.9)

The received power at the mobile earth station is given by Pr  F A, hence Pr 

PtGtGrl2 watts 4pA

(10.10)

Thus the received power at the mobile terminal with an omnidirectional antenna increases as the square of the wavelength, or decreases as the square of the frequency. The lower the RF frequency, the greater the received power for any given coverage zone. By reciprocity, the same result will apply when the mobile terminal transmits with an omnidirectional antenna. It therefore makes sense for mobile systems, which are forced to use omnidirectional antennas so as to avoid having to steer a directional antenna, to use the lowest possible RF frequency. That is why Orbcomm’s data relay LEO satellite system uses VHF

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and UHF frequencies. Orbcomm satellites have a single transmit beam at the satellite that serves the entire coverage zone. For the same reasons, L band is allocated for mobile satellite service, but to achieve similar CN ratios with L-band links as Orbcomm satellites achieve with VHF links, the L-band satellite must provide multiple beams from a high gain antenna. One disadvantage of VHF and UHF frequency bands is a high noise power due to the natural environment. For this reason, the antenna noise temperature for a system operating at VHF or UHF will be much higher than the receiver noise temperature. Environmental noise temperature falls with increasing frequency; by L band it is not a significant factor. The worst possible choice of frequency for a mobile system is Ka band (or above). A Ka-band mobile downlink operating at 20 GHz requires 22.5 dB more transmitted EIRP, or receiving antenna gain, than the same system operating at 1.5 GHz. Occasionally proposals are aired for Ka-band mobile systems, but such a system can only succeed with a steered directional antenna on the mobile terminal. Conventional mechanically steered Ka-band dishes are thousands of dollars more expensive than a simple whip antenna, and self-steering phased arrays are even more costly. It is worth noting in passing that it is the omnidirectional antenna of a mobile terminal that drives up the cost of every transmitted bit in a mobile system. Suppose that a fixed terminal with an antenna of gain Grx supports a bit rate of Rb bits per second. A mobile terminal with a low gain antenna, gain Grm, and the same satellite EIRP and path loss can support a much lower data rate of Rb (GrmGrx). For example, an 18-inch DBS-TV antenna operating at 12.5 GHz has a gain of 33 dB. A satellite link using the DBS-TV antenna can receive data at 2000 times the rate of a mobile terminal that has an omnidirectional antenna with a gain of 0 dB, with the same overall CN value in the receiver. Given equal costs for the space segment of the communications link, and a mobile system operating in Ku band, a system operator must charge the mobile user 2000 times as much per delivered bit compared with the DBS-TV terminal owner. Looked at another way, for the same monthly fee, the DBS-TV customer can receive signals at 20 Mbps, equivalent to several compressed digital television signals, while the mobile terminal customer can receive only 10 kbps—a single voice channel. Antenna gain is the system designer’s friend. Mobile systems will become much more attractive economically when a self-steering, self-phasing, phased array is available for mobile terminals which has even a moderate gain. A 10-dB increase in antenna gain translates directly to a 10-fold increase in bit rates.

Elevation Angle Considerations As we have seen in Chapter 8, rain attenuation can cause significant attenuation on a slant path. At Ka Band (3020 GHz), even light rain can cause appreciable signal loss. Light rain is usually stratified, so the higher the elevation angle the lower the rain attenuation for a given rainfall rate. Figure 10.13 illustrates the geometry. Most commercial satellite systems require that earth stations operate above certain minimum elevation angles. For example, Intelsat requires that all earth stations using Intelsat C-band (64 GHz) satellites operate above 5°, otherwise the earth station does not meet Intelsat’s standard specification and must be qualified for operation on an individual basis. To qualify an earth station on an individual basis is an expensive undertaking. At Ku band (1411, 1412 GHz) the standard antennas in the Intelsat system are required to operate above a minimum elevation angle of 10°. In creating the original Teledesic system architecture5, the overriding design input for the coverages was that no earth station should operate at an elevation angle below 40°. This requirement, when coupled with an orbital height of around 800 km, led to an unrealistic initial constellation of 840 operational satellites to

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Paths to satellite

B

409

Height of melting layer

C

Stratiform rain

A Earth station FIGURE 10.13 Illustration of the decrease in the path through rain as the elevation angle to the satellite increases. Light rain is generally formed at the melting layer height in stratiform clouds. The rain then falls fairly uniformly over a wide area. Freezing precipitation (hail, ice crystals, dry snow) does not cause any appreciable attenuation to radio waves. Since the rain is uniform, the attenuation per meter will be constant everywhere in the stratiform rain shower and the total path attenuation will be given by the length of the signal path in the rain. The higher elevation angle path (AC) will therefore suffer less attenuation than the lower elevation angle path AB.

provide full global coverage. Most satellite systems now, whether for the mobile satellite service (MSS) or the fixed satellite service (FSS) at frequencies above 10 GHz, tend to limit the elevation angle of the user to no less than 10° so that reliable service can be provided. Given a minimum elevation angle and an orbital height, the geometry setup in Figure 10.12 can be used to develop a coverage area, assuming that the satellite has a symmetrical beam aimed at nadir. A plot similar to that shown in Figure 10.14 results. Track of sub satellite point along surface of earth

Movement of coverage area under satellite Orbital path of satellite

Earth FIGURE 10.14 Illustration of coverage area under a satellite. In this example, an NGSO satellite moves along a path over the earth with a nadir pointing antenna, that is, the antenna has its electrical axis directed straight down toward the subsatellite point. The antenna will have a finite usable beamwidth which will allow a given portion of the surface to be illuminated at the same time. This is shown as the shaded portion in the figure. Increasing the altitude of the satellite’s orbit will increase the coverage area. Alternatively, the altitude can remain fixed and the beamwidth increased in order to cover a larger surface area.

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The shadowed area on the earth in Figure 10.14 is the maximum instantaneous coverage on the surface of the earth that can be achieved from that satellite. The calculation process for the instantaneous coverage has as input the minimum elevation angle a user can tolerate and the orbital altitude selected. (Instantaneous coverage means that, if a snapshot were taken and the motion of the satellite frozen at an instant in time, the region of the earth covered by the satellite’s antenna at that particular time would be the instantaneous coverage from that satellite. The word instantaneous is used to separate out coverages that are developed by scanning or hopping satellite antenna beams. For the scanning or hopping beam concepts, full coverage is obtained by moving elemental beams around the coverage area to pick up traffic. A full, and instantaneous, coverage of the observed region is therefore not obtained with scanning or hopping beams.) The instantaneous coverage from a satellite, however, is not always served by one beam from the satellite antenna due to the lack of available spectrum and a concomitant need for extensive frequency reuse. This is particularly true for MSS systems which, like terrestrial microwave cellular systems15, have to divide up their coverages into cells covered by separate beams in order to provide enough capacity into a given cellular structure. Each cell, here a separate beam from the satellite antenna, will have a portion of the spectrum allocated to it. The simplest spectrum reuse pattern is a three-cell configuration. The spectrum is divided into three roughly equal portions and a three-cell pattern is built up over the coverage area. Figure 10.15 illustrates the concept. There are many other cell reuse patterns that are possible15.

Spectrum A

Spectrum B

Spectrum C

Instantaneous coverage FIGURE 10.15 Illustration of a three-cell reuse pattern. The instantaneous coverage of the satellite antenna is shown as the circle with a broken line. Within this coverage, individual beams formed by the satellite antenna make a regular pattern that fills up the instantaneous coverage. The spectrum that has been allocated to this satellite has been divided up into three portions, called Spectrum A, Spectrum B, and Spectrum C. These different spectra are indicated with different shadings above. None of the three spectra are adjacent to the same spectral allocation. Note: In general, each of the individual beams will overlap their neighbors for two reasons. First, by overlapping the individual beams, there are no “holes” in the instantaneous coverage. Second, physics will prevent the beams going from full power to zero power over a negligible distance. It is usual to develop coverages using the half-power (3 dB down) point of the beams as the edge of coverage gain/power. There will therefore be energy spilling over into adjacent beam coverages. This is why it is necessary to employ a different spectral allocation in adjacent beams, unless a CDMA access technique is used.

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TABLE 10.3 Frequency, Antenna, and Capacity Characteristics of the Big LEOs Parameter Mobile user link Frequency (up/down) (GHz) Maximum bandwidth (MHz) Spot beams per satellite Nominal capacity per satellite (voice circuits) Orbital altitude (km)

Iridium

Globalstar

New ICO

1.62135–1.6265 5.15 48 1,110

1.619–1.6215/ 2.4835–2.4985 11.35 16 2,400

1.980–2.010/ 2.170–2.200 30 163 4,500

780

1,414 km

10,355 km

Source: [13, 14].

Number of Beams per Coverage The very small spectrum allocation available for MSS systems ( 50 MHz), and the many competing systems that aim to provide mobile satellite services, place a number of constraints on the system design. Table 10.3 shows the spectrum, antenna, and resulting capacity of the three major MSS systems: Iridium, Globalstar, and ICO-Global, known generically as Big LEOs. ICO emerged from Chapter 11 bankruptcy protection in 2000 with a new name: New ICO. At the time of going to press, it was not clear how the New ICO satellite payload design was being affected by the proposal to merge New ICO MSS applications and Teledesic Internet-like services. New ICO is an MSS system spun off from the International Maritime Satellite Organization (Inmarsat). While it uses a medium earth orbit architecture, with 10 satellites in two planes at an altitude of 10,355 km above the earth, New ICO is generally considered together with Iridium and Globalstar as one of three Big LEO MSS systems. Because the New ICO satellites are so much further away from the earth than both Iridium and Globalstar satellites, the New ICO satellites need to generate many more beams per satellite within each instantaneous coverage in order to achieve sufficient capacity/km2 within the coverage. Figure 10.16 presents a comparison of the Iridium and New ICO array of multiple spot beams developed within their respective instantaneous satellite coverages. The requirement placed on the MSS satellite antenna to generate multiple beams within a given instantaneous coverage is a key driver in the payload technology. Traditional satellite antennas have evolved from simple, front-fed reflector antennas with one feed horn, to offset-fed designs with more than a hundred feeds16. Such multiple feed horn reflector antenna designs are necessarily large and heavy. The greater the number of individual beams to be generated, the heavier the reflector antenna and associated feed horns and beam forming network. Depending on the precise spacecraft mission, there is a threshold where the cost and complexity of a phased array antenna implementation will be less than that of the equivalent reflector antenna. A phased array antenna usually has a nonmechanically steered array of radiators. The radiating elements can be passive devices (e.g., dipoles or feed horns) or active devices (e.g., patch elements that include amplifiers). The steering of the beam is carried out by varying the phase (and amplitude for full sidelobe control) of the signal in each radiating element. For a passive device, the phase control is achieved in the feed matrix placed between the high-power amplifier and the radiating antenna elements while, for the active device, there is a phase shifter per element per beam. In many cases, it is possible

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+20° +10° 0° −10° −20° +30° +20° +10°

0°

−10° −20°

(a )

90 60 30 Latitude

412

0 −30 −60 −90 −180

0

180

Longitude (b)

FIGURE 10.16 (a) User spot beams developed by an Iridium satellite (Figure 11 of reference 13 © 1997 IEEE, reproduced with permission). The satellite covers about 40° of the earth’s surface from its orbital height of 800 km, which translates into about 4000-km diameter main coverage. This coverage is divided up into 48 spot beams. Each of the spot beams has the same beamwidth: the curvature of the earth has caused the outer spot beams to appear elliptical. The boundary of each spot beam denotes the 3-dB-down point of that spot beam. (b) User spot beams developed by an ICO-Global satellite (Figure 20 of reference 13 © 1997 IEEE, reproduced with permission). The satellite covers about 110° of the earth’s surface from its orbital height of 10,355 km, which translates into about 12,000-km diameter main coverage. This coverage is divided up into 163 spot beams. Each of the spot beams has the same beamwidth: the curvature of the earth has caused the outer spot beams to appear elliptical. The boundary of each spot beam denotes the 3-dB-down point of that spot beam. Note that the spot beam size of the ICO-Global satellite is similar to that of the Iridium satellite.

to include the amplifier as part of the active phased array radiating element. This particular phased array concept is referred to as a direct radiating phased array. Figure 10.17 illustrates the two phased array approaches. In either approach, the scan angle is often a critical design limitation.

Off-Axis Scanning The design of a point-to-point wireless communications system requires that the antennas at either end be directed toward each other for maximum gain advantage. This was the approach adopted for the fixed service (FS), the terrestrial microwave communications

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Direction of beam

413

Phase front of beam

A

A

A

A

A

A

A

φ

φ

φ

φ

φ

φ

φ

HPA

Input signal (a )

Direction of beam

Phase front of beam

Direct radiating element

G, φ

G, φ

G, φ

G, φ

G, φ

G, φ

G, φ

Input signal (b )

FIGURE 10.17 Illustration of scan angle control mechanisms for phased array antennas. (a) Passive phased array. (b) Direct radiating array. In (a), the high-power amplifier (HPA) has had the power divided up among a number of different feed lines. Each feed line is acted on by a variable phase-change () and a variable attenuator (A) device. The resultant output signal is then fed to a passive feed horn. The sum of the many phases and amplitudes generated by the feed horn cluster will develop the antenna coverage. In (b), the phase and amplitude are controlled by the direct radiating device at the end of the feed line. The amplitude is controlled by the gain of the radiating amplifier, G, and the phase, , can either be controlled within the amplifier unit itself or by a phase element associated with the radiating device. To develop a large number of beams, many signal lines will feed each element and a complex phase front will be developed. Each beam direction will be given by the composite phase of the associated phase front for that signal; each beam shape will be given by the number of individual elements contributing to the development of that phase front.

service. If the transmitting antenna has to communicate with more than one receiving antenna, and these antennas are located in different positions, a compromise must be reached between the gain of the transmitting antenna toward the various receiving antennas. In this case, most, if not all, of the receiving antennas will not be on the boresight (main beam axis) of the transmitting antenna. Figure 10.18 illustrates the problem.

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Main lobe of transmit antenna

Boresight of transmit antenna

Receive antenna

Transmit antenna

(a )

Main lobe of transmit antenna Set of receive antennas

Required total scan angle

Transmit antenna

Boresight of transmit antenna

(b )

FIGURE 10.18 (a) Point-to-point line-of-sight terrestrial communications link. The transmit antenna illuminates the receive antenna along its electrical boresight, providing the maximum gain for the link. The receive antenna has its electrical boresight directed toward the transmit antenna (not shown explicitly here). The transmit and receive gains are therefore maximized. (b) Point-to-multipoint line-of-sight terrestrial communications link. In this plan view of a point-to-multipoint system, one example of which is called LMDS—Local Multipoint Distribution System, the transmit antenna has to cover a number of receive antennas spread over a large total scan angle. There are two main options available to the link designer: use a single, wide-angle beam to cover the receive antennas (as has been illustrated here); or set up several different transmit antennas, each directed toward a given receive antenna. In this latter concept, the function of the transmit antenna can be divided up among a small group of antennas that provide 360° coverage, as in the sectored antenna approach of cellular systems.

Exactly the same design compromise illustrated in Figure 10.18b faces satellite system designers who have to provide coverage over a large instantaneous area from a single satellite. A satellite is a prime example of a point-to-multipoint system. There are two basic input geometrical parameters that are used in the initial design phase of a satellite antenna: the orbital height and the instantaneous coverage requirements for a single satellite. Figure 10.19 presents the three main design options for orbital altitude: LEO, MEO, or GEO and Table 10.4 lists some scan angle requirements for various satellite altitudes, with atmospheric refraction ignored.

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415

Total scan angle

LEO

MEO

GEO

FIGURE 10.19 Schematic of the total scan angles for LEO, MEO, and GEO satellites. The further away the satellite is from the earth, the smaller the satellite antenna scan angle needed to provide an instantaneous coverage out to a given user elevation angle minimum. In the above figure, the satellites are all in an equatorial orbit. The view is from above the earth with one side of the earth in sunlight and the other in darkness. The terminator is directly under all three satellites.

A fixed antenna with a parabolic reflector is able to scan its main beam away from the electrical boresight axis by repositioning the feed transversely from the prime focus. However, the plane wave that is present in the aperture of a focused parabolic reflector antenna becomes distorted when the feed horn is moved away from the focus, resulting in an effect known as coma. Coma causes a reduction in antenna gain, an increase in sidelobe levels, and an increase in cross-polarization. The reduction in gain and polarization purity can be held to relatively small values if the focal length, f, of the antenna is long with respect to the antenna diameter, D, and the off-axis scan angle is small. A value of fD  1 is generally taken as the required design goal, and is usually implemented by employing a double reflector configuration such as the Cassegrain antenna. Cassegrain antennas have a large equivalent fD ratio while being mechanically compact. GEO satellite antenna designs that scan over the full earth coverage (8.7°, which corresponds to TABLE 10.4 Scan Angle and Latitude/Longitude Ranges for Different Satellite Altitudes LEO Orbit and orbital height Scan angle Latitude/longitude range

MEO

GEO

750 km

1,800 km

10,000 km

14,000 km

35,786 km

57.2° 12.8°

47.1° 22.9°

21.5° 48.5°

17.1° 52.9°

8.25° 61.8°

The minimum elevation angle to the user terminal for the data in this table is 20°. The coverage is assumed to be a cone of revolution around a nadir pointing direction. Note that, even though the scan angle is smaller for satellites at higher altitudes, the latitude (and longitude) coverage increases with altitude. Thus, for a given scan angle, instantaneous coverage increases with satellite altitude. Source: Some of the data have been extracted from references 7 and 13.

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0° elevation angle at the extreme earth station locations) have been successfully implemented using reflector antenna technology. However, as the scan angle requirement increases (due to the satellite moving from GEO to a lower orbit) and the number of individual beams that are needed within the instantaneous coverage grows large, phased array antennas prove to be the best match to the system design17. Phased array antennas are used on the Iridium, Globalstar, and New ICO satellites, while Cassegrain or Gregorian (another double reflector configuration) antennas are used on many large GEO satellites. A number of factors influence the coverage of a phased array antenna from a given satellite. While the phase of the signal in the radiating elements determines the steering of the beam, it is usual to have the main beam axis (generally normal to the surface of the antenna array panel) directed at nadir. This will lead to the edge-of-coverage users suffering two loss components that are larger than a signal transmitted in the nadir direction. First, they will be further away from the satellite, and so will suffer a free space path loss that is increased with respect to a nadir user. For the LEO example in Table 10.4 that is at an altitude of 750 km, the 57.2° scan angle leads to a slant range of 1681 km at the 20° edge of coverage. The difference in path loss between the range at nadir (750 km) and edge of coverage (1681 km) is 7.0 dB. The 1800-km orbit LEO satellite has a 5.5dB path loss difference between nadir and edge of coverage. For mobile satellite systems that must operate with this rapid variation in path loss as the satellite passes by the user, power control is employed to offset changes in perceived power level at both the satellite and the earth terminal (the handset). The second higher loss component experienced by edge of coverage users is that the satellite antenna will incur a scan loss as it attempts to direct energy away from the main beam axis (the boresight directed at nadir) out to the edge-of-coverage user. Figure 10.20 illustrates the change in path loss with scan angle. Satellite

d EOC

dN

Total instantaneous coverage angle (scan angle)

Boresight directed at nadir Edge of coverage (EOC)

Edge of coverage (EOC) Instantaneous coverage region

FIGURE 10.20 Illustration of path loss and scan angle loss evaluation for a phased array. The phased array has its prime axis pointed at nadir. The energy received at nadir from the satellite will be greater than that received at edge of coverage (EOC) for two reasons. First, the path loss will be less since the nadir distance, dN, is less than the EOC distance, dEOC. Second, there will be a scan loss associated with the antenna reaching out to cover the EOC region.

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417

Satellite Total instantaneous coverage angle Total instantaneous coverage angle (the scan angle) Scan angle, φ, away from nadir Nadir direction Individual beam formed over edge of instantaneous coverage

3-dB beamwidth of the individual beam

Section through surface of earth

Required instantaneous coverage FIGURE 10.21 Illustration of the scan angle of an individual beam within an instantaneous coverage. The instantaneous coverage is developed through many smaller beams spread over the region to provide sufficient frequency reuse for the users in that area. Only one of the small, individual beams is shown above as a shaded area on the right. This beam is scanned to the edge of the instantaneous coverage. Note that, a user within the small, individual beam will have to factor two components into the link budget: (1) the gain loss due to not being at the center of the individual beam; and (2) the scan loss due to not being at the nadir point of the instantaneous coverage region (here assumed to be the boresight of the phased array antenna on the satellite).

Scan loss for a phased array antenna normally follows the relationship17 Scan loss  1cosine f2 k

(10.11)

Scan loss  1cosine 57.22 1.3  0.4507 1 3.5 dB

(10.12)

where  is the scan angle off boresight and k is an empirical number between 1.2 and 1.5. Note that the negative sign in Eq. (10.11), which was not incorporated in the referenced article, allows the sign on both sides of the equation to agree [see the example in Eq. (10.12) below]. Figure 10.21 illustrates the geometry for Eq. (10.11). A typical value to use for k is 1.317. For example, a LEO system that needs to scan 57.2° away from boresight will have a scan loss

Thus the scan loss is 3.5 dB for a beam transmitted to the edge of an instantaneous coverage of 57.2°. The edge-of-coverage path will also suffer an additional path loss compared with the nadir path of 7 dB for a LEO satellite orbiting at a height of 750 km. The edge of coverage signal is therefore 10.5 dB below the nadir signal in this example. To counteract these two loss components—scan loss and enhanced path loss—the phased array boresight can be redirected toward the edge of coverage. The problem with this approach is that at least three phased array panels are required on the spacecraft to

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330 inches

Bus section

Solar array panel Battery and radiator 180 inches

Main mission antenna panel

Communications section Gateway antenna

Cross-link antenna

FIGURE 10.22 A sketch of an Iridium satellite (Figure 10 of reference 13 © IEEE 1997, reproduced with permission, courtesy of Motorola). One of the three phased array antennas is shown as the “main mission antenna panel” in the above figure. The instantaneous coverage is developed using these three phased array antennas, much like a three-sector microwave cellular coverage within a cell. Iridium uses an FDMA/TDMA multiple access technique. Since communications can be established from two of the antennas into an area that is on the “joint” between the two sector coverages, the signals must be accurately controlled in time so that they arrive at all three antenna array panels at the same instant and the TDMA bursts do not overlap in time. Not shown clearly in this figure are the four ISL antennas that communicate with the other satellites in the constellation. Two of the four antennas link “forwards” and “backwards” within the same orbital plane while the other two link eastwards and westwards. Iridium ISL links are only possible with satellites that are moving in the same general direction. That is, a north-going satellite cannot establish an ISL link to a south-going satellite in another plane.

illuminate the full instantaneous coverage. With three antenna array panels, the scan loss is reduced by at least 1.5 dB17. This solution was adopted by Iridium. The three phased array antenna panels can be seen clearly in Figure 10.224,17. In addition to the required antenna scan angle, the height of the orbit is the other key geometrical parameter that influences the design of a LEO constellation.

Determination of Optimum Orbital Altitude The locations at the edge of coverage within the instantaneous coverage region normally present the greatest problems in the design of a satellite service. It is at the edge of coverage that the power flux density into the user terminal is at its lowest. Even if the user is at the center of the individual beam that serves the edge of coverage (see Figure 10.21) there are still the two additional factors that determine whether the link can provide adequate service: the scan loss and the added free space path loss at edge of coverage when compared with nadir. Minimizing the total additional loss in the transmission path to edge of coverage is a design goal. If the orbital altitude is increased, the free space path loss will increase, but the scan loss will decrease. For a LEO constellation of satellites with a given large instantaneous coverage requirement, as the orbital altitude increases from a minimum of 500 km, the scan loss will decrease faster than the path loss increases. There will therefore be an optimum altitude for a LEO constellation, based on the number of satellites per plane and whether more than one satellite must be in view to any given user

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30

0.

25

−0.1 −0.2

20

−0.3

15

−0.4

10

−0.5 −0.6

5 0

419

−0.7 1000

1400

Relative loss (dB)

Minimum grazing angle (°)

10.3 COVERAGE AND FREQUENCY CONSIDERATIONS

−0.8 1800

Satellite altitude (km) FIGURE 10.23 Relative transmission loss and minimum grazing angle vs satellite altitude for a constellation of LEO satellites with 10 satellites per plane and double coverage (Figure 1 from reference 18 reprinted with permission from Microwave Journal). In the calculations for this figure, it was assumed that there would always be two satellites in view for any user. With 10 satellites per plane, a minimum scan angle at the satellite is calculated and, from this, the elevation angle for the edge-of-coverage user is found. This angle is referred to as the minimum grazing angle in the figure. The scan loss  free space path loss for edge of coverage are normalized to an orbital height of 800 km. As the orbital height increases, the scan loss  free space path loss reaches a shallow minimum between about 1350 km and 1800 km altitude above the earth.

40

0.4

30

0.2

20

0

10

−0.2

0

1000

1400

Relative loss (dB)

Minimum grazing angle (°)

at all times. Figure 10.23 shows this trade-off for a constellation with 10 satellites per plane and double coverage (i.e., two satellites always in view from all possible user sites). Figure 10.24 shows a similar trade-off for a constellation with 15 satellites per plane with the same double coverage requirement18.

−0.4 1800

Satellite altitude (km) FIGURE 10.24 Relative transmission loss and minimum grazing angle vs satellite altitude for a constellation of LEO satellites with 15 satellites per plane and double coverage (Figure 2 from reference 18 reprinted with permission from Microwave Journal). The calculations for this figure are similar to those carried out for Figure 10.23, except for this constellation there are 15 satellites per plane. With 15 satellites per plane, a minimum scan angle at the satellite is calculated and, from this, the elevation angle for the edge-of-coverge user is found. This angle is referred to as the minimum grazing angle in the figure. The scan loss  free space path loss for edge of coverage are normalized to an orbital height of 800 km. As the orbital height increases, the scan loss  free space path loss reaches a more pronounced minimum than that in Figure 10.23. This time the minimum is between about 950 and 1300 km above the earth rather than 1350 km and 1800 km altitude found for a constellation with one-third the number of satellites per plane.

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Figures 10.23 and 10.24 indicate that there are a number of iterative analyses that can be performed to balance scan loss, free space path loss, number of satellites in view at any user site, and orbital altitude. Once these “geometrical” analyses have been performed, it is necessary to look at other factors. From the satellite hardware design aspect, a critical factor is the radiation environment: the higher the LEO orbit altitude, the worse the radiation environment becomes as it approaches the first main Van Allen radiation belt at around 1500 km. Perhaps the most critical factor is the RF transmit power available from the user’s handset. Battery and adaptive handset antenna technology may be able to increase the available EIRP from the user’s phone, but the biological radiation limits imposed for safe usage will place an upper bound on handset EIRP.

Radiation Safety and Satellite Telephones In the United States, the Federal Communications Commission (FCC) mandates strict limits on radiated power levels throughout the spectrum. Their rules are usually issued in dockets (see, e.g., reference 22, which provides the main FCC Web site and the dockets for evaluating the environmental effects of radio frequency radiation). The Office of Engineering of the FCC has also posted an RF Safety Program on its Web site23, which provides guidance on the specific absorption rate (SAR) for wireless phones and devices. Many of these guidelines have been developed through IEEE Committees (see e.g., reference 24), which have made many proposals to the American National Standards Institute (ANSI) on this issue. Safe exposure levels are given as 0.08 W/kg as averaged over the whole body for the general population and 0.4 W/kg for occupational or controlled exposure for professionals working in this area. These values do not provide enough insight into handheld units held close to the head and many studies are still underway, some of which are reported in reference 25. It is clear that handset power levels are well below those that cause ionization damage to tissue. However, while the short-term effects of the power levels used in handsets have been proven to be negligible, there have been insufficient studies at present to provide the long-term effects of such exposure, that is, over more than 10 years of handset use. A number of international groups, in particular ETSI in Europe, are collaborating on such studies.

Projected NGSO System Customer Service Base A single satellite in an NGSO system will not provide continuous 24-h coverage over a given area. If a national or regional coverage is desired, a constellation of NGSO satellites is required with orbits tailored to match the coverage. This was the approach adopted for the Molniya system where a minimum of two satellites in two Molniya orbits could provide continuous 24-h service. Most of the new NGSO systems have been aimed at mobile users. For mobile users, the problem is to generate sufficient transmit power in a handheld terminal without exceeding the limits for electromagnetic radiation from the antenna into the head and body of the user. Low power transmissions from the handheld unit requires either a satellite in low earth orbit or a very large antenna on a MEO or GEO satellite. All three alternatives have been developed13. The driving forces behind the decisions made in choosing a system architecture will be discussed in Section 10.5 and some typical systems will be reviewed in Section 10.6. We will now look at two closely associated elements of an NGSO system—or any telecommunications system for that matter—that can have significant implications on customer acceptance: delay and throughput.

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10.4 DELAY AND THROUGHPUT CONSIDERATIONS Delay in a communications link is not normally a problem unless the interactions between the users are very rapid—a few milliseconds apart in response time. Long delays, such as those associated with manned missions to the moon, required the development of agreed procedures, much like tactical military or police communications requires specific handoff code words such as “over” to signal the end of one user’s input. For most commercial satellite links that are over long distances, particularly those with satellites in geostationary orbit, the main problem was not delay, but echo. A mismatched transmission line will always have a reflected signal. If the mismatch is large, a strong echo will return. Over a GEO satellite link, the echo arrives back in the telephone headset about half a second after the speaker has spoken, and usually while the speaker is still speaking. This will interrupt the speaker and the conversation becomes fragmented. The development of echo suppressors and, even better, echo cancellers, solved the problem. Figure 10.25 illustrates the one-way propagation time for a typical LEO, MEO, and GEO system. Based on the calculations shown in Figure 10.25, the time delay for a signal passing between LEO user 1 and LEO user 2 in the same instantaneous coverage is 5.4 ms (2.7 ms up and 2.7 ms down) and the go and return (round-trip) delay between the two users is twice this at 10.8 ms. It is rare, however, for a user to be immediately underneath a LEO satellite and, for LEO satellites in higher orbits, the round-trip delays due to propagation time can be more than double this. Globalstar, which has a maximum path length from the satellite to the user of 2500 km, will have a maximum round-trip delay time of 33 ms. For GEO users, the up and down (forward) link delay is typically 230 ms with the round-trip delay 460 ms. However, Figure 10.25 does not tell the whole story. Most MSS systems use voice compression to reduce the bandwidth required for a single voice channel. The coded bit rates for a single voice channel range from 2.4 kbit/s for Globalstar to 6.25 kbit/s for Iridium13. The vocoders sample the incoming analog voice signal and produce excellent, low data rate digital reproductions—but at a price in delay. The access scheme can also add additional delay. If the channel is operated in a simplex fashion, i.e., you cannot send at the same time as you are receiving, there can be a delay in response. The Iridium TDMA access mode uses a time division duplexing (TDD) scheme. A TDD scheme allows transmissions to occur for a certain period (while receive functions are off) and then transmissions cease while receive operations are in use. In the present Iridium TDMA access scheme, eight users share a frequency assignment and, within this frequency channel, share a 45-ms transmit frame and a 45-ms receive frame. There can therefore be up to 90 ms between transmission and reception of specific parts of a message. On the Iridium satellite, the onboard processing system translates the received signal to baseband, the header address information is read, and the appropriate route selected for onward transmission. The baseband signal is then reformatted, up-converted to the RF band, and transmitted. All of this takes time. The forward delay (ground-to-satellite plus satellite-to-ground) within the same instantaneous coverage averaged 153 ms in the initial operational tests of Iridium. A transoceanic link delay using intersatellite links averaged 253 ms—almost the same as for a GEO satellite link. Delay can also have an adverse effect on the throughput of the signal, as noted in Chapter 9. If the protocol used in the link is not adapted for the particular delay environment, appreciable reduction in throughput will occur. Customer acceptance of a service has been found to be driven by three prime factors: access ability (i.e., can the required connection be obtained immediately on request?), availability (i.e., once connected, will the call be dropped?), and performance (i.e., is the error rate low and the throughput high?). Pricing will attract customers but it will not keep them for long if all three prime factors are not met.

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GEO satellite: 35,786 km

One-way delay: 119.3 ms

MEO satellite: 10,355 km

One-way delay: 34.5 ms

LEO satellite: 800 km One-way delay: 2.7 ms

FIGURE 10.25 One-way propagation delay for the three orbits: LEO, MEO, and GEO. The one-way delay figures shown above have been calculated assuming the radio signal propagates at the speed of light in a vacuum, i.e., 3 108 ms. That is, no account has been taken of any delay due to the refractive index of the atmosphere not being unity. Also, no account has been taken of any processing delay imposed on the signal from any source coding, channel coding, modulation, or access scheme used.

As a final element in the discussion on delay, it is worth noting the challenges that face system designers when intersatellite links (ISLs) are employed to relay signals around a LEO constellation. It is a fairly straightforward matter to design an ISL to connect two GEO satellites or a LEO satellite to a GEO satellite: the relative motions are not that large. Consider now a LEO system attempting to establish connections across the constellation. The connections will have to be both in plane (i.e., around the same orbit plane of that particular ring of satellites) and across planes. When the satellites are close to the equator, the orbital planes are at their furthest separation and the rate of change between two LEO satellites traveling in the same direction is at a minimum. As satellites move closer to the poles, the more rapidly they have to steer their ISL antennas to maintain contact. In some operational modes, Iridium switches off the acrossplane ISL links when the spacecraft are above latitudes of about 60° 13. In no case, however, can Iridium maintain an ISL link between planes where the satellites are moving

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“Seam” across which microwave ISLs are unable to track LEO satellites in adjacent plane FIGURE 10.26 Schematic of the ISL seam in the Iridium constellation. The Iridium satellites are in an orbit that is close to polar (86.5° inclination). There are four ISL antennas on each satellite that are used to communicate with adjacent satellites. The ISLs operate at 23 GHz and use solid reflector tracking antennas. The inertial mass of the antennas combined with the need to have a stable satellite platform for the normal communications mode to the earth limits the rate of change of the tracking mechanism. Satellites across the seam are traveling at a closing speed of about 36,000 mph (58,000 kmh) and it is likely that only lightweight optical ISLs will be able to track at the angular rates of change required across an LEO seam.

in opposite directions. There will therefore be a seam in the constellation across which no ISL links can operate. This is illustrated in Figure 10.26. The revised Teledesic system (once 840 satellites, then 288 satellites, and now fewer than 200 satellites) has reportedly been designed to operate across the LEO “seam” and so it is likely that the ISLs will be optical and not microwave. Optical ISL antennas are much smaller and lighter than microwave ISL antennas and so impose less tracking restrictions due to inertial forces when under acceleration. Whether or not to use ISLs; whether to design to operate across the seam if ISLs are used; selecting an orbital height, number of satellites visible at any instant, coverage region; etc.; all interact in the overall system design. We will now look at other system considerations that can affect the design of the satellite network in other respects.

10.5

SYSTEM CONSIDERATIONS There are four important factors that influence the design of any satellite communication system: incremental growth, interim operations, (satellite) replenishment options, and endto-end system implementation.

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Incremental Growth The 1964 decision by the Interim Communications Satellite Committee (soon to become The International Telecommunications Satellite Organization Intelsat a few years later) to select a GEO satellite system rather than a 12-satellite MEO system that was supported by major entities on both sides of the north Atlantic at that time was driven by incremental growth plans as well as by launcher technology. The primary international traffic route was across the Atlantic Ocean, followed (a long way second) by the Indian Ocean region, and (an even longer way third, at that time) by the Pacific Ocean region. The system could be grown incrementally with a GEO architecture. The first GEO satellite, Early Bird, was placed over the Atlantic Ocean region in 1965. For the first decade of operations, new satellites were launched into the Atlantic Ocean region to replace satellites that had been operating there. The satellites being replaced were moved to the Indian Ocean region and the satellites replaced in the Indian Ocean region were moved to the Pacific Ocean region. It was not until INTELSAT VII that Intelsat specifically designed a satellite for the Pacific Ocean region from scratch. This approach to incremental growth served Intelsat well. By comparison, the new LEO and MEO mobile service systems now in operation require all of the satellites to be in operation before full operations can begin. However, most of the LEO and MEO system operators developed interim operations plans where a reduced number of satellites could provide useful service.

Interim Operations Interim operations for LEO and MEO systems serve two functions: they can bring a service on line gradually, introducing the technology to the market while teething problems are sorted out; and they can act as fall back plans should multiple satellite failures occur over a short period. Nearly all of the LEO and MEO systems undertook such interim operations. Orbcomm began commercial operations with less than half of its 36-satellite constellation in place, thus becoming the first commercial LEO system to establish a revenue stream. Globalstar began with 32 out of the planned 48-satellite constellation and New ICO plans to start operations with six out of the planned ten-satellite constellation. Iridium, since it uses ISLs to complete the network, required all 66 satellites to be available before beginning beta testing in November 1998. The technical planning for interim operations includes relaxing the number of satellites visible to any user at any particular time, which lowers the number of satellites required to complete the constellation. The elevation angle minimum for users is also usually lowered, the gaps between operational satellites in the same plane are made symmetrical, and the orbits adjusted if possible to maximize coverage over those parts of the day when user service requests are highest. Most LEO constellations have at least four satellites per plane and multiple spacecraft launches are used in the constellation buildup. A Pegasus launch vehicle carries eight Orbcomm satellites into orbit, A Delta II carries five Globalstar satellites, and a Proton carries seven Iridium satellites into LEO. When a satellite fails in service, there is an inorbit spare to take its place. If more than one satellite fails in a plane, additional satellites must be launched to replenish the system.

Replenishment Options Launching five or more satellites to replace one failed satellite makes little economic sense. As a result, the LEO service providers use smaller rockets to replenish their system. Table 10.5 lists the primary and replenishment launchers used by the Big LEO systems and Orbcomm.

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TABLE 10.5 Primary and Replenishment Launchers for the Big-LEO Systems and Orbcomm

LEO/MEO system Iridium Globalstar [note 1] New ICO [note 3] Orbcomm

Primary launchers (number per launch)

Secondary/replenishment launchers (number per launch)

Delta II (5) and Proton (7) Delta II (5) and Soyuz (4) Atlas IIAS (1), Proton (1), Zenit (1), Delta III (1) Pegasus (8)

Long March (2) [Note 2] [Note 4] Taurus (2)

Notes: 1. Globalstar initially selected Zenit but canceled the launch services contract when the first launch failed with 12 satellites aboard. 2. Globalstar has not selected a replenishment launcher. 3. The Zenit rockets for New ICO are to be launched from a floating platform. Sealaunch is a joint venture of Boeing and Hughes. 4. New ICO satellites are so large, and the orbital altitude so high, that only one satellite is launched per rocket. Any of the selected rockets could act as replenishment launch vehicles.

End-to-End System Implementation A communications system can be part of a larger network (e.g., just providing the longdistance portion of the connection) or it can provide the full end-to-end system implementation, from user to user. AT&T and Intelsat, when they were first set up, did not provide end-to-end service: AT&T provided long-distance capacity for local telephone companies and Intelsat provided satellite capacity for entities such as AT&T to carry their international traffic. Neither company interacted directly with the end user. Indeed, specific laws or protocols prevented this from happening. The design of an NGSO system will be heavily influenced by the decision on whether or not to provide service directly to the end user. It will also be impacted by the decision on whether or not to include established telephone companies in the delivery of the service. By their very nature, mobile satellite systems have committed to serve the end user directly. However, different approaches have been taken with regard to including established telephone companies. Two examples of organizations that took opposite decisions are Globalstar and Iridium. Globalstar elected not to bypass existing telephone companies while Iridium did. These decisions led to a very different architecture for the two systems, which will be discussed in the next section.

10.6 OPERATIONAL NGSO CONSTELLATION DESIGNS Seven satellite constellation designs are reviewed briefly in the following discussion, four MSS offerings with multiple beams, one with single beam coverage providing both twoway services and one-way store-and-forward services, and two Internet-multimedia satellite systems.

Ellipso The Ellipso constellation drew from studies of the world’s population distribution and the potential market for MSS users. Figure 10.27 (abstracted from data in reference 19)

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30

25 Percentage of population

426

20

15 10

5

0

S S S S S S S S S N N N N N N N N N 90– 80– 70– 60– 50– 40– 30– 20– 10– 0– 10– 20– 30– 40– 50– 60– 70– 80– 80 70 60 50 40 30 20 10 0 10 20 30 40 50 60 70 80 90 Latitude range south (S) and north (N) in degrees

FIGURE 10.27 Percentage of the world’s population living in the given latitude ranges (data extracted from reference 19 and reproduced with permission). The data in the figure show that more than 85% of the world’s population lives in the northern hemisphere. Designing a satellite system that spends most of its time in the northern hemisphere would therefore cover the world’s population more efficiently.

shows that more than 85% of the world’s population lives north of the equator. Additional studies19 concluded that an equatorial constellation of MEO satellites could serve the bulk of the world’s population. Ellipso therefore adopted an incremental approach to their service offering. The first set of satellites would be in a circular equatorial orbit. The second set would be in elliptical equatorial orbit, with the ellipticity of the orbit designed to provide dwell times over the regions of greater demand. The third set of satellites would be in sun synchronous 3-hour orbits inclined at 116.6° to provide coverage over the highly industrialized northern hemisphere regions. The equatorial orbit groups of the Ellipso system are called Concordia™ and the sun synchronous group is called Borealis™. Details can be found in Table 10.6. The Ellipso spacecraft is based on the Boeing GPS satellite bus and up to five satellites can be launched by a single rocket. No onboard processing is performed; the signals received at the satellite are transponded down to gateway earth stations for onward routing via the terrestrial PSTN or satellite network. No ISLs are used.

Globalstar In a similar manner to Ellipso, Globalstar elected to develop a constellation that was aimed at the populous regions of the earth. The Globalstar orbital planes are therefore inclined at 52° to the equator, thus ignoring the sparsely populated high-latitude regions. To minimize the power requirements of the user handset, the constellation altitude was lowered to just below the first Van Allen radiation belt. This increased the total number of satellites needed to 48. No onboard processing or ISLs are used; the signals received at the satellite are simply transponded down and the gateway earth stations process the signals for the onward routing (see Figure 10.28). Like Ellipso, service over water is

Spot beams per satellite Satellite lifetime

Orbital height (km)

Total complement Orbital inclination Orbit type

16 7.5 years

1414

48 52° Circular

6 8

1S3S5

Number of planes Satellites per plane 1 7 then 1 7 and 2 3 then 1 7, 2 3, 2 5 23 3 at 0°, 2 at 116.6° 1 circular (0°), 2 elliptical (0°), 2 sun synchronous 1 circular 8050, 2 elliptical 6149–8050, 2 sun synchronous 633–7605 61 5 to 7 years

Globalstar

Ellipso

163 12 years

10,255

10 45° Circular

2 5

New ICO

48 5 to 7 years

780

66 86.5° Circular

6 11

Iridium

1 5 to 7 years

775

36 4 at 45°, 1 at 72° Circular (45° and 72°)

4 8 then 4 8 and 1 4

4S5

Orbcomm

System Parameters of Five NGSO Constellations Aimed at Data and Voice Communications

System parameter

TABLE 10.6

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Instantaneous coverage regions of two satellites

User 1 Gateway earth station

PSTN connection (fiber-optic cable or GEO satellite)

User 2 Gateway earth station

FIGURE 10.28 Schematic of end-to-end connection for satellites that have no onboard processing or ISLs. User 1 is in a different instantaneous coverage region than that occupied by user 2. The signal from user 1 is picked up by the gateway earth station and relayed to the gateway earth station of user 2. The signal is then sent up to the satellite from the second gateway earth station and then down to user 2. If user 1 or user 2 is using a fixed telephone (or computer) the signal would simply pass over the regular PSTN circuits and not via the space segment. Because the users must be in line-of-sight contact with a gateway earth station, no maritime traffic can be picked unless the ship is close to land (and a gateway earth station).

restricted to coastal regions where the satellite is within radio range of a gateway earth station.

New ICO ICO Global is the company that was spun off from the International Maritime Satellite Organization (Inmarsat); New ICO is the company that emerged from bankruptcy protection in 2000. Inmarsat was initially set up solely for the purpose of providing reliable communications to maritime traffic. Later, Inmarsat also provided aeronautical services, in addition to priority links for safety communications, whether on land or sea. New ICO, although primarily aimed at the LMS market (Land Mobile Services), also needed to provide capacity for maritime links. New ICO elected not to include ISLs in their system architecture nor any significant onboard processing. Since a LEO constellation would not provide maritime coverage without ISLs, a higher orbit was necessary. If little onboard processing is used, traffic routing from mobile to mobile would have to be carried out at the gateway earth stations (as it is for Ellipso and Globalstar) necessitating a double-hop link. A double-hop link involves two uplinks and two downlinks. (A double hop is used in Figure 10.28; two different earth–space links are used to complete the connection.) A double-hop configuration is not feasible for a GEO constellation since the overall delay would be completely unacceptable at about 1 s. New ICO therefore adopted a MEO constellation. An inclination of 45° is used, but since the orbit altitude is so high, full global coverage is possible. Toward the end of 2000, it was learned that New ICO was modifying the payload and service offerings to provide two-way Internet-like connections in a joint venture with Teledesic. At the time of going to press, it is unclear how this synergy will evolve.

Iridium The genesis of Iridium was formed around the need to communicate from anywhere to anywhere on the surface of the world, even where no telecommunications infrastructure

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existed. The system therefore must be stand-alone. From this—and the need for a low power handset—came the concept of first 77, and then 66, almost-polar orbiting LEO satellites linked via ISLs. Each of the satellites in the constellation acts as a switching node. Uplink signals are received and demodulated at the satellite using onboard processing to recover individual data packets at baseband so that the header information can be read. Using this information and links to the network control stations, the next node for each packet is determined and the packet is reformatted with the next address. The baseband data packet is then processed and up-converted for transmission either to the ground at L band directly to another Iridium user or at 20 GHz to a gateway earth station, or over one of the four ISL links (at 23 GHz) to the next satellite in the chain. Onboard processing is needed to carry out the entire message routing and formatting functions.

Orbcomm Many research organizations and businesses need to obtain data from locations that are either inaccessible on a regular basis or are moving within areas without good cellular telephone coverage. Examples are buoys measuring water characteristics in rivers and at sea, and delivery trucks. Tracking of high value cargo on trucks is another application that needs to send a short message to a central station at regular intervals. A GPS receiver on the cargo determines its location and this information is sent with an ID number via an Orbcomm satellite. If the truck carrying the cargo is hijacked, its route can be followed and the truck intercepted. Much of this information is neither required in real time nor does it need a high capacity link. Orbcomm developed their system around this requirement and have orbited a constellation of satellites with both two-way data communications and store-and-forward capabilities (see Table 10.6). The satellites are lightweight (40 kg) and simple in design and execution3. A single beam is used to develop the instantaneous coverage and no onboard processing is used. A terminal that is within the coverage area of a satellite and a gateway station (which includes almost all of the United States) can send short messages to the gateway station in real time. The message length is limited to a few hundred bytes. A terminal that has data waiting to be uploaded for store-and-forward listens for the passage of a satellite and then uploads its data when the satellite is in view. The data, in the form of a packet with the address of the intended recipient, are stored and transmitted to a gateway station for onward transmission to the recipient when the satellite is within range of the gateway station. Orbcomm satellites carry short messages, with a relatively high cost per transmitted bit. The system is therefore most attractive to users who want to send a small number of high value bits, such as requests for help in emergency situations or tracking information for high value cargo. None of the five NGSO constellations above were initially designed to carry traffic at rates higher than 10 kbps. This is not adequate for Internet access, which has emerged as a potentially important requirement in mobile systems. Two NGSO constellations that addressed this market from the outset are Skybridge and Teledesic.

Skybridge Skybridge evolved a similar approach to coverage as Globalstar, by selecting an inclined orbit that covers the major population densities. Like Globalstar, Skybridge satellites

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carry a nonprocessing payload and do not have intersatellite links; so all traffic is transponded down to the gateway earth stations for processing and onward routing. However, Skybridge satellites are intended to carry wideband traffic and therefore use frequencies above 10 GHz. They chose to employ the same Ku-band frequencies as the FSS service in GEO uses: 12.75–14.5 GHz on the uplink and 10.7–12.75 GHz on the downlink. To allow successful coordination with existing FSS GEO systems, they elected to prevent any operations (up or down) whenever a satellite look angle is within 10° of the GEO orbital plane. This requirement led to a relatively large number of satellites (80 vs 48) for the constellation. The decision not to use ISLs also required a very large number of gateway earth stations (on the order of 200). Skybridge also uses the concept of a fixed earth cell (see Figure 10.29). More details of Skybridge can be found in Table 10.7.

Teledesic Teledesic started from the same precept as Iridium, but is designed for Internet-like data traffic rather than voice communication. Any user can access any other user or ISP (Internet service provider) independent of location and the existing telecommunications Position 1

Track of sub satellite point along surface of earth

Position 2

Orbital path of satellite

Coverage does not move under satellite: cell is a stationary coverage on surface of earth

Position 3

Earth FIGURE 10.29 Concept of a stationary cell. Unlike the coverage of the NGSO satellite show in Figure 10.14, a stationary coverage (or “fixed earth cell”) of an NGSO satellite does not move with the satellite. The phased array antenna on the satellite steers the beam, while the satellite transits, to keep the coverage on the surface of the earth constant. As the satellite moves between positions 1, 2, and 3, stationary coverage is maintained on the surface of the earth. Separate antennas are used for communications coverage and gateway links. In this way, a gateway need not necessarily be within a given stationary coverage.

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TABLE 10.7 System Parameters of Two NGSO Constellations Aimed at Internet Multimedia Communications System parameter

Skybridge

Teledesic

Number of planes Satellites per plane Total complement Orbital inclination Orbit type Orbital height (km) Spot-beams per satellite Satellite lifetime

20 4 80 53° circular 1469 18 7 years

12 24 288 90° circular 1400 — 7 years

infrastructure. The concept of Teledesic is to provide a complete worldwide data communications system above the surface of the earth using satellites, instead of on the earth’s surface using fiber-optic cables. This requirement dictated the use of wideband data links, onboard processing, and ISL links. To avoid the necessity of coordinating with existing systems, Teledesic chose to move their operations completely into Ka band. As noted earlier, to reduce the impact of rain, Teledesic also limited the elevation angle at which users could access the satellites (the mask angle) to 40°. The initial Teledesic constellation had a complement of 840 satellites (22 planes with 40 operational satellites per plane) plus 40 spare satellites in orbit. The orbital altitude was later moved up from 700 km to about 1400 km, which reduced the number of planes to 12, with 24 operational satellites in each plane (see Table 10.7). The early estimates of Teledesic’s system cost were between $9 B and $12 B, using 840 satellites. Reduction in the number of satellites to 288 lowered the cost significantly, and further reductions in the number of satellites seem likely to make the cost of creating the system more acceptable. Other companies are seeking to provide Internet access from satellites with lower cost solutions than those of Teledesic and Skybridge. One company aiming to do so for a “mere” $2.6 B is Virtual Geosatellite20. However, for a given bit rate, no proposed system is lower in cost than a GEO alternative21. The geostationary earth orbit has the unique characteristic of providing data transfer by satellite at the lowest cost per bit. None of the currently scheduled or operating LEO and MEO satellite constellations has been able to demonstrate a significant added value from the use of their particular service when a commercial return on investment is required. There is a clear military requirement for many of the new constellations—from anywhere to anywhere—without any intervening infrastructure, but the growth in terrestrial cellular systems and optical fiber links has removed much of the potential commercial demand for these new services. At the turn of the twenty-first century, more than 90% of all Internet traffic flowed through about 30 metropolitan areas. If these conurbations are connected via optical fibers or through high-powered spot beam antennas from GEO, the remaining traffic is what a LEO or MEO system would pick up. The same is true for cellular telephony: what the major cities do not provide leaves very little traffic for a high priced LEO or MEO alternative. Table 10.8 gives some early 2000 data on Internet traffic centers.

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TABLE 10.8 1. 2. 3. 4. 5. 6. 7. 8.

Internet Traffic Centers

London New York Amsterdam Frankfurt Paris Brussels Geneva Stockholm

(18 terabits/s) (13.2 terabits/s) (10.9 terabits/s) (10.5 terabits/s) (9.7 terabits/s) (6.2 terabits/s) (5.9 terabits/s) (4.4 terabits/s)

9. 10. 11. 12. 13. 14. 15.

Washington, DC San Francisco Toronto Chicago Seattle Vancouver Tokyo

(4.0 (3.9 (3.5 (2.7 (2.6 (2.5 (2.4

terabits/s) terabits/s) terabits/s) terabits/s) terabits/s) terabits/s) terabits/s)

Source: The data above were extracted from Telegeography at http://www.telegeography.com. The data provide the Internet capacity of the top 15 major cities in terms of their Internet traffic, averaged over a day26.

EXAMPLE 10.6.1 System Design A company wishes to develop a LEO constellation that provides continuous global coverage. They are restricted to an orbital height of 750 km due to user terminal power, operating time between battery charges, and satellite launcher capabilities. The following design data are required: • The length of the coverage arc on the surface of the earth within the instantaneous coverage; • The gain of the satellite antenna if one beam is to illuminate this coverage; • The number of satellites needed to complete one plane with a suitable overlap; and • The number of satellites needed to complete a global system. Length of Coverage Arc Figure 10.12 illustrates the geometry of a satellite and user terminal. If the minimum elevation angle is set at 10°, we know rs  re  750 km (the orbital height),   10°, and re  6378 km (average radius of the earth). We need to find the central angle, , (angle ECZ in Figure 10.12) which will allow us to find the length of half of the arc under the coverage-arc EZ. Using the sine rule, we have sin 1d2 re  sin 1angle SEC2 rs

(10.13)

The angle SEC    90°  100° and this yields   61.7859  61.79° If d  61.79°, then g  180  100  61.79  18.21° Arc EZ is therefore given by re (with  in radians)  2027.1 km. The diameter of the instantaneous coverage region is therefore 2 2027  4054 km and the coverage angle measured at the center of the earth is 36.42°. (Note that this assumes the coverage is symmetrical about the nadir pointing direction SC in Figure 10.12.) Alternatively, we could have derived the same result by noting that the circumference of the earth is  diameter of the earth   6378 2  40074 km. The fraction of this illuminated by the satellite is (2)(2) (with  in radians)  0.1012. The total coverage diameter arc  40074 0.1012  4055 km. Gain of Satellite Antenna The angle  in Figure 10.12 is half of the antenna beamwidth. The full angle of the antenna beamwidth at the satellite is therefore 2  61.79° 2  123.6°. The gain of an antenna can be related to the 3-dB beamwidth using the approximate relationship Gain ratio  33,000 13-dB beamwidth in degrees2 2  G

which gives G  33,000 1123.6°2 2  2.16 1 3.3 dB

(10.14)

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Instantaneous coverage arc = 4054 km

Angle γ of instantaneous coverage arc at center of earth = 36.42°

FIGURE 10.30 Coverage results from Example 10.6.1. The coverage of one satellite orbiting at an altitude of 750 km is shown. The coverage arc has been calculated to be 4054 km when a minimium elevation angle of 10° is assumed. The angle at the center of the earth that subtends the instantaneous coverage arc, , has been found to be 36.42°. To complete the coverage around the earth using this satellite configuration would require (360)(36.42) satellites. Rounding to whole number yields a minimum number of 10 satellites per plane.

Number of Satellites per Plane We now have the situation set up in Figure 10.30. Since each of the satellites will cover 36.42360  0.1 of the earth’s circumference, we will need a minimum of 10 satellites in one plane. Total Number of Satellites for a Global System By the same logic used above, if 10 satellites are required to complete coverage around (say) the equator, 5 complete planes of satellites will be needed to complete the full global coverage. (Remember that one plane of satellites, if in a polar orbit, will have satellites on both hemispheres of the earth, some going northwards and some southwards. There will therefore be 10 “slices” around the earth made up of 5 planes of satellites.) The total minimum number of satellites needed is therefore 50. It should be noted that this is an absolute minimum number. In addition to coverage gaps potentially existing, there will be a need to have spare satellites in orbit to take care of satellite failures. Other architecture requirements can now be imposed. One could be the need for simultaneous coverage of any user by two satellites. The coverage of each satellite is unchanged, but the satellites must be half the distance apart so that two satellites always are in view throughout the constellation. A second architecture rule might be that no user is required to operate below 20° (rather than 10°). There are many other possible variations, for example, covering only the latitudes between 65° of the equator, inserting elliptical orbits to increase dwell time over a particular region. It is thus easy to see why some constellations need many dozens of satellites to complete the full architectural requirements. 

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SUMMARY

Launching satellites into a geostationary orbit is a complex task that was not achieved successfully until 6 years after the first satellite was orbited in 1957. The quest for the geostationary orbit grew from a paper by Arthur C. Clarke that established this orbit as the prime location for communications satellites. If the satellite is geostationary, the earth station antenna need not be steerable, greatly reducing the cost of the system. Non-geostationary satellites, however, continued to be launched for a huge range of missions. TIROS satellites photographed the weather over the earth in sun synchronous orbits, orbits that precessed to match the rotation of the earth around the sun. TRANSIT satellites began the experiments into navigational aids that were space borne. MIDAS, the first early warning satellite; IRAS, the first infrared astronomy satellite; and Explorer satellites that probed the inner reaches of space above the earth continued the ever expanding list of experimental spacecraft launched into non-geostationary orbits for research purposes. Molniya, which means flash of lightning in Russian, was a satellite series that established the first regional satellite system in 1965. The Molniya orbit has found many uses for a range of other satellite systems in non-geostationary orbit. In the 1990s, a whole series of proposals arose for constellations of non-geostationary satellite communications systems, some in low earth orbit and others in medium earth orbit. The thrust for moving the satellites “down” from geostationary orbit can be summed up in one word: power (or, more strictly speaking, four words: EIRP). The non-geostationary satellite systems were aimed at the mobile user and so required that the user terminal antennas be essentially nondirectional.

Limited by radiation dosage limits, the telephone handsets for the mobile satellite systems could radiate only feeble amounts of power. To compensate for the low power and minimal GT in the handsets, either technologically adventurous antenna systems were required in geostationary orbit or equally adventurous (as it turned out, but this time in an economic sense) constellations of non-geostationary satellites were required for a global mobile telephony system. Satellite constellation design is a complex mix of coverage requirements, capacity, and connectivity. The iterations in a design will also have to bear in mind the radiation environment in space caused by the Van Allen radiation belts and the need to have both an incremental design philosophy and replenishment options for the satellites that fail. From the design examples shown, it is clear that geostationary satellites will always provide cheaper costs on a per bit basis than a satellite system in any other orbit until technology breakthroughs enable smart antenna design for the user handsets. Even then, care must be taken that terrestrial systems have not taken away the customer base. This is a very interesting period in the evolution of telecommunications. The Internet, and by implication digital data communications, has become the greatest growth area in the transfer of information globally. The extent to which satellites in non-geostationary orbit can fit into the global information infrastructure successfully will determine whether they have any commercial future in this field. That they have a future in navigation (GPS and the European system, Galileo) and geographic information systems is assured, but the revenue streams in these two cases have yet to match the investment requirements.

REFERENCES 1. A. C. CLARKE, “Satellite Communications Systems,” Wireless World, pp. 305–308, 1945. 2. J. E. ALLNUTT, Satellite-to-Ground Radiowave Propagation, Peter Perigrinus, Ltd., UK, 1989. 3. http://www.orbcomm.com 4. http://www.iridium.com 5. http://www.teledesic.com 6. http://www.skybridgesatellite.com 7. http://www.globalstar.com 8. GARY D. MORGAN and WALTER L. MORGAN, Principles of Communications Satellites, ISBN 0-471-55796-X, John Wiley & Sons, 1993. 9. PROFESSOR WATSON, private communication, 1996 (then at York University, now at Bath University, both in the UK).

10. J. M. BENEDETTO, “Economy Class Ion-Defying ICs in Orbit,” IEEE Spectrum, Vol. 35, No. 3, pp. 36–41, March 1998. 11. ITU-R Rec. 435-5, Prediction of Sky-Wave Field Strength between 150 and 1600 kHz, 1986. 12. K. MURSALA and T. ULICH, “A New Method to Determine the Solar Cycle Length,” Geophys. Res. Lett. Vol. 25, pp. 1837–1840, 1998. 13. J. V. EVANS, “Satellite Systems for Personal Communications,” IEEE Antennas and Propagation Magazine, Vol. 39, No. 3, pp. 7–20, June 1997. 14. F. J. DIETRICH, P. METZEN, and P. MONTE, “The Globalstar Cellular Satellite System,” IEEE Transactions on Antennas and Propagation, Vol. 46, No. 6, pp. 935–942, June 1998.

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15. THEODORE S. RAPPAPORT, Wireless Communications: Principles and Practice, ISBN 0-13-375536-3, PrenticeHall, Englewood Cliffs, NJ, 1996. 16. M. A. B. TERADA, “Reflector Antennas,” Wiley Encyclopedia of Electrical and Electronics Engineering, John G. Webster ed., ISBN 0-471-13946-7, Vol. 18, pp. 360–379, 1999. 17. T. J. SCHUSS, J. UPTON, B. MYERS, T. SIKINA, A. ROHWER, P. MAKRIDAKAS, R. FRANCOIS, L. WARLDLE, and R. SMITH, “The IRIDIUM Main Mission Antenna Concept,” IEEE Transactions on Anennas and Propagation, Vol. 47, No. 3, pp. 416–424, March 1999. 18. P. CHIAVACCI, “The Influence of Phased-Array Antenna Systems on LEO Satellite Constellations,”

19. 20. 21. 22. 23. 24.

25. 26.

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Microwave Journal, vol. 42, No. 5, pp. 282–290, May 1999. http://www.ellipso.com http://www.virtualgeo.com http://www.hcisat.com http://www.fcc.gov/oet/dockets/ http://www.fcc.gov/oet/rfsafety/ ANSI/IEEE C95.1-1992, “IEEE Standard for Safety Levels with Respect to Human Exposure to Radio Frequency Electromagnetic Fields, 3 kHz to 300 GHz”, copyright of the IEEE, 1992. K. R. FOSTER and J. E. MOULDER, “Are Mobile Phones Safe?,” IEEE Spectrum, pp. 23–28, August 2000. DR. FENG, Virginia Tech, private communications, 2000.

PROBLEMS 1. What is the preferred orbit [approximate orbital height (maximum and minimum, or average if close to circular), approximate orbital inclination, range of subsatellite points on the equator if geostationary, and approximate orbit eccentricity] for a satellite that needs to do the following: a. Observe the polar ice caps at least every 2 h; b. Observe the Falkland Islands (Islas Malvinas) with an elevation angle to the satellite from the surface of the islands of at least 60° for 6 h every 24 h; c. Observe the development of tropical cyclones in the southern Pacific Ocean and hurricanes in the northern Pacific Ocean for 24 h each day; d. Observe swathes of the earth below the satellite illuminated by sunlight from directly behind the satellite on every pass the satellite makes on the sunlit side of the earth; and e. Make observations of interstellar X-rays for 1 h per day when more than 40,000 miles from the earth, but be able to relay information back to earth once per orbit when less than 1000 miles above the earth; Note: The satellite in question need only do one of the above five missions at a time, not all of them at the same time. 2. Why is it optimum (in terms of launch energy requirements) to do the following? a. Launch a satellite toward the east b. Launch a satellite from the equator A fully steerable earth station antenna is on the equator and it observes a satellite that is in a circular equatorial orbit moving in an easterly direction. It can observe the satellite down to the horizon, which is at an effective elevation angle of 0° in every direction. What is the apparent orbital period of the satellite (i.e.,

the time between successive passes when the satellite is directly over the earth station) if the true orbital period is the following? c. 2 h d. 6 h e. 12 h f. Did anything look strange at first sight with your answer to (e) above? If yes, what was it and can you explain the answer you arrived at for (e), which looks counterintuitive at first sight? Note: (1) For this question, do not use a sidereal day in developing your answers; assume 24 h for the earth’s rotational period. (2) If you saw nothing strange at first sight in your answer to (e) you are either an exceptional orbital mechanic or not very curious! 3. The International Space Station (ISS) has an experimental package that will be located in the freeflying module (FFM). The experiment in the FFM was located there to avoid unnecessary vibrational tremors and thruster accelerations from the ISS impacting the gravity sensitive biological experiments on board the FFM. The experiments require continuous communications with a main research laboratory located in New Mexico, so that real-time monitoring and adjustment to the experiments can be made. This requires huge quantities of data and highresolution video to be transmitted 24 h per day to New Mexico. An average, one-way data rate of OC192 (approximately 10 Gbit/s) is needed for the data/video link to New Mexico. A return link of OC1 (51.84 Mbit/s) suffices for the uplink control path to the FFM. a. What is your preliminary outline system design solution for the data link between the FFM and the earth station in New Mexico? Give justifications for your

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choice of system architecture and frequency bands (see the note at the end of this question regarding frequency bands). b. If 8-phase PSK modulation with 34-rate FEC is employed for the OC-192 downlink from the FFM, approximately what instantaneous bandwidth will be required for the link? c. If the instantaneous bandwidth of a given transponder is limited to about 10% of the carrier frequency, what is the lowest downlink carrier frequency that can be contemplated for this link in the FSS bands? d. If you selected a preliminary answer to part (a) before you looked at parts (b) and (c), did you change your mind once you knew the tentative answers to parts (b) and (c)? e. If occasional shuttle blockages of the transmission path, or other small interruptions occur in the link, that lead to outage intervals of up to 5 min per day, what is the maximum data storage required for the transmission buffers? f. The OC-192 data link has been designed to have some redundancy in the information capacity. If there is a redundancy margin of 0.5% in the data transmission requirement (that is, for every 1000 bits actually sent, normally only 995 bits are required so there is spare capacity to send 5 bits more of information within those 1000 bits, if the need arises), will the 5 min per day of anticipated outage be handled by the OC-192 link transmission rate? Note: Four FSS bands are 64 GHz (C band), 1411 GHz (Ku band), 3020 GHz (Ka band), and 5040 GHz (VQ Band). Remember that the uplink carrier frequency is given first in each paired band. 4. The broadcast of digital radio signals from satellites has a number of advantages and disadvantages. Some of the advantages are: crisp, clear sound; uninterrupted reception regardless of distance from your home location; and the ability to listen, commercial free, to your favorite radio station wherever you travel within the coverage region. The main disadvantage is the possible blockage by tall structures. This last issue may prove to be critical in the success, or otherwise, of two different approaches to the provisioning of such services. In the United States, two companies (XM Radio and Sirius Satellite Radio) have approached the system design of a broadcast satellite radio system in two very different ways. One has adopted a GEO approach (www.xmradio.com) and the other a Tundra approach (www.siriusradio.com). The Tundra orbit is similar in concept to the Molniya orbit—a very

high apogee and a relatively low perigee—but the orbital period of a Tundra orbit is one sidereal day rather than about half that amount. Three satellites are each in a different Tundra orbit, with the planes of the three Tundra orbits 120° apart. By phasing the position of each of the three satellites in its Tundra orbit, there are always two satellites visible at high elevation over the coverage region. The Tundra approach therefore needs three satellites as opposed to the single satellite needed in the GEO approach, but the Tundra approach requires far fewer terrestrial repeaters to achieve full coverage in built-up areas that cause blockage to GEO satellite signals. In the system level example below, the costs given are for example only and do not represent in any way the true costs of either of the proponent broadcast radio systems noted earlier. If programming and content support, and the unit cost of the car radio system, are assumed to be the same for both system approaches; the cost of a GEO satellite is $200 M each, including launch; the cost of the Tundra satellites is $125 M each, including launch; and the average cost of a terrestrial repeater is $25 k each; find: a. If no terrestrial repeaters are needed for either the GEO or Tundra approach, which system is less expensive to bring to operational status, a single GEO satellite or three Tundra satellites? b. If it is necessary to have an in-orbit spare for the GEO approach (i.e., two GEO satellites need to be launched prior to broadcast radio services being offered) but no in-orbit spare is required for the three Tundra satellites (since they provide a strong measure of in-orbit redundancy by virtue of their orbit placement), which system is now the cheapest to bring to operational status? (Still no terrestrial repeaters are required.) c. If the operational control center costs for a GEO satellite are $6 M per year for one satellite and $1 M per year for each additional satellite; the operational control center costs for a Tundra satellite are $12 M per year for one satellite and $4 M per year for each additional satellite; and an operational lifetime of 10 years is assumed in this part of the problem, which system is now the cheaper to bring into service and operate over a 10year lifetime (assuming two GEO and three Tundra satellites)? (Still no terrestrial repeaters are required.) d. Using the assumptions in part (c), if the Tundra approach requires 10 times fewer terrestrial repeaters than the GEO approach, at what number of terrestrial repeaters for the Tundra approach are the total costs

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PROBLEMS

of the two satellite broadcast radio systems approximately equal to each other? e. Given the answer to (d), which approach might be the better to pursue? State your reasons, giving as many parameters that might influence your choice between the two system approaches. f. Using all of the assumptions in (d), which of the two approaches would you select—giving your reasons—for a digital satellite radio broadcast service to be offered in (i) Indonesia, (ii) Europe, (iii) the Pacific Islands, and (iv) South America. 5. This question concerns the effect of radiation on electronic equipment on a spacecraft. a. What are the two principal effects that radiation has on electronic equipment? b. What particles principally cause these effects? c. What is the prime generator of these particles? d. What causes the generation of these particles to fluctuate with time? e. Is there a periodicity in these fluctuations? If yes, what is the approximate period of these fluctuations? f. Is there a particular region, or are there particular regions, in near-earth space where concentrations of these particles are to be found? If yes, what is it (or what are they)? g. What orbital inclination of a satellite circling the earth will cause that satellite to receive the highest radiation dosage compared with other inclinations? h. Which of the radiation particles is the strongest and therefore the hardest to protect against? i. What are two ways to reduce unexpected performance or decreased lifetime in electronic equipment exposed to radiation? 6. The design of a communications satellite’s antenna is fundamental to that satellite’s ability to perform its assigned task. The portion of the earth’s surface illuminated via that antenna is called the coverage. This question considers the definitions of coverage, frequency reuse, and capacity issues. a. What is the difference between total coverage and instantaneous coverage for a satellite antenna illuminating the surface of the earth? b. What is the fundamental difference between a “hopping” beam and a “scanning” beam that might be employed on advanced communications satellites? c. A satellite needs to provide communications capability over a given coverage region. It has been predicted that the average number of users in the cov-

437

erage region that access the satellite is 1% at any given time of those who have signed up with that satellite service. That is, for every 10,000 customers signed up, 100 are using the satellite at any given time. (i) How many instantaneous communications channels must the satellite be able to provide if the coverage region contains a potential user population of 120 million? (ii) If the bandwidth allocation for the satellite communications service permits only 6000 channels to be provided without frequency reuse, how many separate beams will be required for this coverage region, ignoring interference between individual beams? (iii) If interference requirements between individual beams dictate that no frequency may be reused by adjacent beams, what is the approximate new minimum number of separate beams that are required, ignoring for the moment the exact geometry of the beams? (Note: For this part of the question, assume that the bandwidth allocation has been divided into three separate bands to develop the frequency reuse pattern.) (iv) If the physical size of the total coverage region remains the same between cases (ii) and (iii) above, what can you say about the size of the new individual instantaneous beams in part (iii) within the overall coverage? (v) In addition, what can you say about the effect this change in size has on the communications capability and satellite connectivity complexity issues? 7. a. Define the terms “Zenith” from the point of view of an earth station on the surface of the earth and “Nadir” from the point of view of an earth orbiting satellite. b. A geostationary satellite is required to provide communications coverage over the whole of the earth (i.e., a global horn antenna). What is the maximum off-axis angle that needs to be contained within the antenna coverage, measured from nadir? c. A LEO satellite constellation is being designed to provide telecommunications service on a global basis at a downlink frequency of 12 GHz. An evaluation is being carried out to determine what would be the best altitude for the system (which will use circular orbits), what is the best mix of number of beams vs complexity, etc. An initial determination of the optimum orbital height is 1400 km. Assume the minimum operational elevation angle to any part of the coverage on the surface of the earth is 20°.

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(i) What is the scan angle from nadir to edge of coverage? (ii) What is the difference in path loss from nadir to edge of coverage? (iii) What is the scan loss at edge of coverage if the parameter k [see Eq. (10.11)] is 1.4? (iv) If the requirement is to provide the same power density at the edge of coverage as exists at nadir, what additional amount of power (in dB) is needed at the edge of coverage? (v) In what ways do you think you could cope with this large difference in power requirements between nadir and edge-of-coverage power? 8. Historically, nearly every commercial service of whatever nature has been developed incrementally. Most major enterprises that exist today started as small operations in a single location. Global satellite systems do not have that luxury: they have to start global. The designers of the LEO satellite system in Problem 7 above want to examine how they could provide global coverage with different levels of service so that they can have a measure of incremental growth options. They have

selected polar orbit for their system. Find the following: a. What is the minimum number of satellites that can be used to provide continuous coverage around one polar orbit plane assuming operations can continue down to an elevation angle of 0°? b. What is the minimum number of orbit planes that need to be used to provide continuous coverage over the entire globe? c. (i) Repeat (a) above for a minimum elevation angle of 20°. (ii) Repeat (a) above for a minimum elevation angle of 40°. (iii) Repeat (b) above for a minimum elevation angle of 20°. (iv) Repeat (b) above for a minimum elevation angle of 40°. d. What are the advantages and disadvantages to the service provider if they adopt the constellation in part (b) above? e. What are the advantages and disadvantages to the service provider if they adopt the constellation in part (c) (iii) and part (c) (iv) above?

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Geostationary satellites have carried television program material almost since their inception for commercial service in the late 1960s. The limited bandwidth of undersea cables designed for voice communications prevented their use for video signals, so live television signals could not be transmitted beyond the limits of any continent at that time. AT&T engineered microwave links in the 1950s that allowed video signals to be distributed throughout the United States, and other countries quickly followed suit to establish national television networks. The first time that a GEO satellite was used extensively for video transmission was for the Tokyo Olympic Games in 1968, which were broadcast live in the United States using a link through an early Intelsat satellite over the Pacific Ocean. The growth of cable TV (CATV) systems in the United States in the 1970s encouraged the use of domestic North American satellites for distribution of cable TV signals. One transponder on a GEO satellite can send a video signal to thousands of independent cable systems distributed throughout the country, an example of point to multipoint transmission. Satellites are a very effective way to distribute wideband signals, and there was rapid growth in the use of C-band transponders for video signals, using FM and one transponder for each video signal1. Ku-band satellites and compressed digital video signals followed, making it possible to transmit several video signals through one transponder, with corresponding savings in transmission costs. A large fraction of all the transponders in most of the world’s domestic and regional GEO satellite systems are devoted to the distribution of video signals. Figure 11.1 shows the earth station complex at Virginia Tech, in Blacksburg, Virginia, which is equipped to uplink analog and digital video signals for distribution to educational institutions. The large antenna in the left of the photograph is a 9-m C-band Cassegrain antenna used to transmit analog video FM-TV C-band signals to a domestic satellite. The antenna is equipped with two 3-kW C-band transmitters connected to orthogonal polarization ports on the antenna feed, allowing simultaneous transmission of two video signals to two orthogonally polarized transponders on the satellite. The antenna was used for many years to distribute graduate classes to 16 locations in and around the Commonwealth of Virginia. The smaller antenna in the middle of Figure 11.1 is a 5.5-m Ku-band Cassegrain uplink antenna used to transmit multiple digital compressed video signals. The University moved to compressed digital transmission at Ku band when the leasing price of C-band transponders suddenly increased following the failure of a large domestic C-band satellite. The antenna at the right of Figure 11.1 is a Simulasat antenna. The reflector is a parabolic torus antenna aligned with the GEO arc and has seven feeds. Each feed illuminates a section of the reflector which approximates a paraboloid. The antenna is used by the University’s campus cable TV network to receive video signals from seven GEO satellites. Several smaller receive-only antennas can be seen in the background. 439

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FIGURE 11.1 The Virginia Tech earth station complex in Blacksburg, Virginia.

Video distribution and direct broadcast television (DBS-TV) have become a major source of revenue for the satellite communications industry. At the end of 1998, the total revenue earned by all satellite communications entities worldwide was estimated at $30 B2. Of this total, $17 B was estimated to have been earned by video distribution and direct broadcast television. The rapid growth of digital DBS-TV may increase the percentage of the industry’s revenues that come from video services even further during the first decade of the 2000s. Many satellite communication systems that were designed for voice and data transmission have ended up distributing TV signals. The revenues available from entertainment television have become a major driver in the satellite communications industry. In 2001, two direct broadcast satellite radio services began operation in the United States using S-band frequencies. The satellites provide a wide range of radio programming, aimed primarily at drivers of road vehicles. Repeaters are used in city areas to overcome the problem of satellite visibility around tall buildings.

11.1 C-BAND AND KU-BAND HOME SATELLITE TV In the early 1980s, the development of low noise GaAsFET amplifiers for C-band, and improved threshold extension demodulators for video signal receivers, allowed much smaller diameter antennas to be used to receive C-band FM video signals distributed through GEO satellites. A market rapidly developed in the United States for home satellite TV systems using 3-m and 3.6-m dish antennas (10-ft and 12-ft diameter) and set-top

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receivers that could receive the video signals from domestic GEO satellites. At that time, the signals were not scrambled, so owners of satellite dishes could receive a wide range of television programming free of charge. The cable TV industry in the United States became concerned about the growth of home satellite TV receiving systems, and tried to have Congress pass laws which would ban their use. Congress did not pass such laws, but instead told the industry to scramble (encrypt) their signals and to charge customers for the descrambling information, and then passed laws that made the unauthorized use of descrambling equipment illegal. An estimated four or five million C-band and Ku-band FM satellite TV systems were sold in the United States by the time that Ku-band direct broadcast satellite television arrived in the 1990s, using digital transmission and 0.5-m dishes, and offering more channels than the earlier system at a comparable price3,4. DBS-TV originally started in Europe and the United States in the 1980s using analog FM transmission in Ku band. Satellite TV was much more successful in Europe than in the United States in the 1980s, possibly because there were fewer alternative sources of TV programming in Europe. Most European countries offered only a handful of broadcast TV channels, and cable service has never been as widespread in Europe as in the United States. Nevertheless, at least one European satellite based direct broadcast TV system failed during the 1980s, and two satellites built for a U.S. company intending to enter the DBS-TV field were sold to a European company. The market for DBS-TV systems grew slowly in the 1980s, and then very rapidly after the introduction of high capacity digital DBS-TV satellites in the 1990s. Primestar developed an analog (FM) DBS-TV service in the United States using transponders on medium power (50–90 W) Ku-band GEO satellites located at 85° west longitude, and a receiving terminal with a 1-m dish. Primestar offered up to 40 TV channels by subscription for a fee of about $40 per month, and based its business plan on leasing the DBS-TV receiving system to customers rather than requiring outright purchase.

11.2

DIGITAL DBS TV In the 1990s, digital video transmission became feasible, and several systems were developed in the United States in the 12.2- to 12.7-GHz band allocated to DBS-TV services. The development of low cost Ku-band antennas and receivers, and high-speed digital integrated circuits specifically for DBS television that incorporate QPSK demodulation, error control, decryption, and MPEG decoding made DBS-TV practical. The digital signal processing is incorporated in a single integrated circuit that implements the digital video standard used by all the DBS-TV systems, DVB-S. The large volumes in which DBS-TV receivers have been manufactured have allowed the cost of a receiving system to be reduced steadily since the start of DBS-TV service. Figure 11.2 shows the rapid growth in subscribers to DBS-TV systems and the cost of a typical home DBS-TV installation in the United States during the 1990s. Directv, a fully digital DBS-TV system owned by Hughes Electronics Corporation, was developed by a consortium of companies led by Hughes, and began limited service in 1994 with a single GEO satellite at 101° W longitude. The first satellite, called DBS-1, was launched in December 1993, and was followed by two more satellites, DBS-2 and DBS-3, in 1994 and 1995. A fourth satellite was added in 1999, and a fifth satellite was launched in 2000 with a transmit antenna capable of providing spot beams, using locations of 101° W and 109° W 3.

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1200

15 Purchase price

Subscribers

10

800

5

400

0 1995

Cost of DBS-TV system in U.S. $

CHAPTER 11

Number of DBS-TV subscribers in United States (millions)

442

0 1997

1999

2001

Year FIGURE 11.2 Growth in subscribers to DBS-TV systems and fall in typical home DBS-TV installation cost in the United States in the 1990s. By 2001, suppliers were offering complete DBS-TV systems free to anyone who would sign a one year contract.

The Echostar Communications Corporation started service with its Dishnetwork in March 1996 with a single satellite at 119° W. In 2001 there were six Echostar satellites in orbit, at longitudes 61.5° W, 119° W, and 110° W. By 2001, Dishnetwork had 5 million customers4. Table 11.1 summarizes the major parameters of two of the DBS-TV satellites serving U.S. customers in 2001. Directv has grown its customer base very rapidly with over 10 million customers by year-end 2000. Directv transmits 200 TV and audio channels that are available in a mixture of subscription packages, much like cable TV companies offer, with pay per view for individual movies and special events. In rural areas, DBS-TV offers hundreds of television channels in place of the three or four terrestrial broadcasting stations that are typically available. In city areas, DBS-TV offers an alternative to cable television at a similar cost. In 1999, Directv acquired Primestar, which provided an additional orbital slot at 85° W. All of the U.S. DBS-TV satellites use digital video transmission, as do several of the European satellites. The main European DBS-TV provider is SES (Société Européenne de Satellites), based in Luxembourg, which had eight Astra DBS-TV satellites in orbit in 1999. Two further satellites were due to be launched in 2001. The Astra DBS-TV satellites are some of the largest GEO satellites in orbit. Astra 2A, built by the Hughes Space and Communications Co., had an on-orbit weight of 7335 kg when launched in 1998, with a life expectancy of 15 years. Astra 2D is a Hughes (Boeing) 601 high-power satellite, due for launch in 20016. The 12.2- to 12.7-GHz band was set aside for exclusive use by DBS-TV satellites in geostationary orbit so that high-power transponders could be used on specially designed DBS-TV satellites. Typical transponder output levels are 100 to 240 W with flux density at the earth’s surface up to 105 dBW/m2. The satellites can carry up to 32 transponders, giving a total transmitted RF power up to 3.2 kW, higher than for any other commercial satellite in 1999. DBS-TV satellites are typically large and heavy, generally use a three-axis

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TABLE 11.1

443

DBS-1R and Echostar 6 Satellite Specifications

DBS-TV satellite

DBS-1R

Echostar 6

Location in geostationary orbit Launch date Satellite manufacturer

101° W longitude July 14, 2000 Hughes Space and Communications Inc. HS 601-HP Ku band (12.2–12.7 GHz) 16 active (4 spare) 200 W (dual 100 W TWTAs) Two wings, GAAlAs cells 8.7 kW 7.7 kW 27-cell NiH 350-Ahr Liquid apogee motor 110 lbf

119° W longitude July 13, 2000 Space Systems Loral

Type designation Frequency band Transponders Output power Solar power system Beginning-of-life output power End-of-life output power Batteries Propulsion Station keeping thrusters N–S (xenon ion) E–W (bipropellant) N–S (bipropellant) Dimensions in orbit Length over solar arrays Width over antennas Dimensions, stowed Height Width Mass at launch Mass in orbit (beginning of life) Antennas Transmit Receive

SSL 1300 Ku band (12.2–12.7 GHz) 32 125 W (can be paired) Two wings 11.27 kW NiH Bipropellant

4  106 lbf (0.17 N) 4  5 lbf (10 N) 4  2 lbf (22 N) 86 ft (26 m) 23 ft (7.0 m)

102.1 ft (31.1 m) 28.4 ft (8.66m)

13 ft 3 in (4 m) 11 ft 9 in (2.7  3.6 m) 3446 kg (7581 lb) 2304 kg (5069 lb)

8157 lb (3700 kg)

(2) 2.72 m 1.32 m

(2) 2.39 m 1.19 m

stabilized design, and have large solar sails to generate the DC power required by the transponders. Figure 11.3 shows Echostar 6, a large GEO three-axis stabilized DBS-TV satellite built for Echostar by Space Systems Loral. The flux density at the earth surface produced by medium- and high-power transponders used on DBS satellites is in the range 105 to 115 dBW/m2, which allows small receiving antennas (dishes) to be used for DBS-TV reception, with diameters in the range

FIGURE 11.3 Echostar 5 DBS-TV satellite. (Photo courtesy of EchoStar Communications Corporation.)

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0.45–0.9 m. The small dish required for DBS-TV reception played a critical part in the acceptance and success of DBS-TV in the United States. Previously, DBS-TV reception of cable television signals was possible at C band and Ku band with 2.5- to 3.5-m dishes. The local governments of many cities and towns refused to permit these large dishes in residential areas, although they became popular in rural areas. The U.S. Congress passed laws in 1997 that prevented local governments from restricting the use of antennas less than 1 m in diameter, opening up a large market for Kuband DBS-TV services. European DBS systems use similar size dishes, but lower transponder power on the satellite. The EIRP values are similar, but western Europe can be covered by a much narrower antenna beam from a GEO satellite than the United States, allowing higher gain antennas to be used on the European DBS satellites. The combination of a higher gain satellite antenna and lower transponder output power produces similar flux densities at the earth’s surface, and the receiving terminals can therefore use antennas with similar diameters. The small receiving antenna has a wide beam, typically 4° for a 0.45-m dish, which forces wide spacing of DBS-TV satellites to avoid interference at the receiving antenna by the signals from adjacent DBS-TV satellites. A 9° spacing in the GEO arc has been adopted by the United States, which restricts the number of DBS-TV satellites that can be placed in geostationary orbit to serve the United States. In the 1990s the U.S. FCC successfully auctioned spectrum and orbital locations for DBS-TV satellites, raising hundreds of millions of dollars from companies that saw a profitable commercial venture. The first entrant into the high-power DBS-TV field, Directv, spent about $1 B to develop their system and needed about two million customers to break even. That number was quickly passed and the Wall Street Journal described Directv as “one of the most successful business ventures of the century.” Directv uses four satellites in two pairs spaced half a degree apart at a nominal orbital location of 100° W. The early DBS-TV satellites served the entire United States from this GEO location, using relatively broad beams. In 2000, new legislation allowed DBS satellite operators to compete with cable television companies by supplying news from local TV stations to specific regions. This made spot beams serving only a part of the United States very desirable, and later generations of Directv DBS-TV satellites incorporate large transmit antennas that can generate spot beams on centers of population in the United States. Each DBS-TV satellite carries up to 32 high-power transponders covering part of the 12.2- to 12.7-GHz broadcast satellite band (BSS), and the satellites at each orbit location transmit in opposite hands of circular polarization (CP). Signals with opposite hands of circular polarization are orthogonal, and a suitably designed earth station antenna can separate two signals with opposite hands of circular polarization. In the simplest DBSTV receiving terminal, an electronically controlled polarizer is used immediately behind the antenna feed. The polarizer can be set to receive LHCP (left-hand circular polarization) or RHCP (right-hand circular polarization) by changing the voltage supplied to the low noise block converter (LNB) unit at the antenna. Typically, a supply voltage of 7 V will cause the antenna to receive one polarization and reject the other. Increasing the voltage above 14 V causes the antenna to switch polarizations. The polarizer converts the circularly received polarization signal to a linearly polarized signal in a section of waveguide, and a linear probe in the waveguide converts the signals to currents that drive the LNA input. More complex receivers use an orthogonal mode transducer (OMT) with two LNBs so that both hands of circular polarization can be received at the same time by using two LNBs and two receivers. The dual-polarization receiving system is needed when more than one channel must be received at a time. A dual-channel DBS-TV receiver with a

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dual-polarized antenna allows two TV channels to be viewed by different people in the same household at the same time. In 2000, Directv charged an additional $5.00 above the basic monthly subscription fee for this grade of service, rather than requiring the payment of two separate subscriptions. Customers wanting to receive signals from more than one orbital location need an antenna with two feeds. Reception from two satellites spaced 9° apart in GEO can be achieved with a larger antenna, 0.45  0.6 m (18”  24”) that produces two beams separated by the appropriate angle. The two feeds can be seen in the photograph of a Dishnetwork antenna in Figure 11.4b. DBS-TV receiving antennas are typically an offset parabolic reflector design with the feed below the antenna aperture. The offset feed design eliminates blockage of the aperture by the feed which occurs in symmetrical reflector antenna designs, and improves the aperture efficiency of the antenna, and therefore increases its gain. Offset fed parabolic reflectors have a beam squint effect in the plane of symmetry when operated in opposite hands of circular polarizations. For the 0.45 m diameter antenna widely used for DBS-TV reception in the United States, the LHCP and RHCP beams are squinted about 0.25° from the antenna’s boresight. The 3-dB beamwidth of the antenna is around 4°, so the squint effect does not cause significant loss of gain.

(a)

(b)

FIGURE 11.4 DBS-TV receiving antennas. (a) Directv antenna mounted on the wall of a house (b) Dishnetwork antenna mounted on a post. Note the two feeds to allow reception from two satellites at separate locations in the geostationary orbit.

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12.2–12.7 GHz DBS-TV signal

LNA

Image rejection BPF

Low noise block converter mounted on antenna feed

900–1400 MHz IF amplifier

Mixer

Coaxial cable to set top receiver

Local oscillator 11.3 GHz

Ku-band antenna 900–1400 MHz IF amplifier

70 MHz IF amplifier

Mixer

QPSK demod

Baseband amplifier

D

Select polarization

Tuned BPF select transponder

Tuned LO

Input Frequency synthesizer

Microprocessor

D/A Inner decoder

De-interleaver

Outer decoder

Digital demux

MPEG 2 decoder

Video Audio Analog output to TV set

FIGURE 11.5 Block diagram of a DBS-TV receiver.

A Directv receiving antenna mounted on the wall of a house is shown in Figure 11.4a, and a Dishnetwork antenna mounted on a post is shown in Figure 11.4b. Figure 11. 5 shows a block diagram of a DBS-TV receiver. The entire front end of the receiver is located at the antenna feed in the form of an LNB to minimize loss of signal and hence to maintain the lowest possible system noise temperature. The electronic polarizer is switched by changing the voltage supplied to the LNB via the cable that interconnects the antenna and set-top receiver. The entire 12.2–12.7 GHz band is downconverted by the LNB to the 900–1400 MHz band, where cable losses are much lower than at Ku band. The down-converter consists of a dielectric resonator local oscillator and mixer, followed by an IF amplifier and band-pass filter. The high gain LNB can drive 100 m of coaxial cable without any reduction in signal quality. Where longer cable runs are needed, amplifiers for the 900–1400 MHz band can be used to boost the signal strength. The set-top box accepts the entire 500-MHz band and separates out the individual transponder frequencies. Any one of these frequencies (and the corresponding polarization) can be selected on demand by the user. The user enters a desired channel number into the set-top box using an IR remote control, for example, channel 362, which is converted via a stored look-up table in the receiver to an RF channel frequency and polarization. The signal from the required transponder is then selected by the receiver by setting the correct polarization at the antenna and tuning the set-top local oscillator to the appropriate IF channel frequency. The QPSK signal is then demodulated. The result is a multiplexed bit stream, typically at a bit rate up to 40 Mbps, which contains the bits for channel 362 and several other video signals. The bit stream is encrypted and contains error control coding bits and data bits. The bit stream is processed to correct and detect errors, de-interleaved, and decrypted. A digital demultiplexer then extracts the bits for the wanted channel, 362 in this example, sends them to

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a MPEG 2 decoder, and finally generates analog video and audio signals with D/A converters to drive the TV set. The look-up table in the receiver that relates channel numbers to frequencies, polarizations, and instructions for the TDM demultiplexer is downloaded from the satellite on a regular schedule. This allows the service provider to change the transponder which carries a particular signal, and to alter the mix of signals on a given transponder as required, without the customer being aware of the changes. The satellite is also used to address individual receivers and to load another lookup table that specifies which channels the user is authorized to receive. If the user fails to pay his or her bills to the service provider, the receiver will eventually be instructed to show only a message that it has been disconnected for failure to make timely payment for the service. This process involves a smart card, which identifies each receiving system, and enables decryption of the satellite signals. The high level of protection applied to the DBS-TV signals is intended to prevent unauthorized reception by users who have not paid monthly fees. Hackers have reportedly broken the encryption system of Directv from time to time by reprogramming smart cards, but Directv retaliated in early 2001, disabling the pirated cards and shutting down nonpaying viewers5. Pay per view channels are handled differently from broadcast channels. A customer wishing to buy a movie or a sporting event selects the desired channel and authorizes the system to make a charge. The Directv receive terminals have no uplink capability, and must therefore use terrestrial telephone circuits to send charging information to the central office of the TV service provider. The cost of the pay per view event is then added to the customer’s monthly bill. This requires a connection between the DBS-TV receiver and the PSTN at the customer’s premises. The receiver dials a toll free number, and downloads a record of the charges for that customer, and any other information that the receiver is programmed to deliver to the service provider. Such information might include the pattern of channels that the customer selects and watches, which is valuable data for advertisers. Using the satellite to convey instructions to the receiver brings some notable advantages to the DBS-TV customer. A customer who wishes to change the level of service he or she receives need make only a single phone call to obtain a service upgrade. The customer owns the DBS-TV receiving equipment and is responsible for its maintenance, so there are no service calls by satellite TV providers.

11.3

DBS-TV SYSTEM DESIGN The DBS-TV system must provide a received signal power at the small receiving antenna that provides an adequate CN margin in clear air. Heavy rain will cause attenuation that exceeds the link margin, so occasional outages will be experienced, especially during the summer months when thunderstorms and heavy rain are more frequent. The CN margins used in DBS-TV systems are small, to avoid the need for a large receiving antenna. The selection of a CN margin is a design trade-off between the outage level that customers can be expected to tolerate, the maximum allowable diameter of the receiving dish antenna, and the power output from the satellite transponders. Typical designs with receiving antennas in the 0.45 to 0.9 m range and 100 to 250 W satellite transponders yield rain attenuation margins of 3 to 8 dB and outage times totaling 5 to 40 h per year depending on the receiver’s location. However, since most customers don’t watch TV for 24 h per day, they will not be aware of all the outages. Unfortunately, thunderstorms tend

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G 60°

C

C

B E

−4 dB E B

D −6 dB

E

G

B Latitude

448

60°

C

F K −4 dB

F H

−6 dB

−4 dB M

30°

E

M

30°

−2 dB

N

N

E

P 150°

120°

90° Longitude

60°

30°

FIGURE 11.6 Coverage pattern of DBS-TV satellite with rain region contours. The solid lines show ITU-R rain zones for the United States. The dashed lines show the satellite antenna beam contours in decibels relative to maximum gain. Note that the satellite antenna has maximum gain in the southeast of the United States where heavy rain is most frequent.

to occur more often in the late afternoon and evening, resulting in more outages during prime viewing time. The Ku-band transmit beam from the satellite carrying the DBS-TV signals is shaped to deliver more power to those areas that suffer the highest occurrence of heavy rain, such as the states in the southeastern part of the United States. This creates a larger link margin in those areas and helps to keep outages to an acceptable level. Figure 11.6 shows satellite antenna EIRP contours over the United States for the Conus beam of a typical DBS-TV satellite located at 101° W, with U.S. rain zones superimposed. The transmit antennas on the DBS-TV satellites have diameters up to 107 inches (2.71 m) which gives a spot beam gain of 49.3 dB. Multiple spot beams are used to provide local TV programming to selected cities and conurbations, while the Conus beam provides service throughout the contiguous 48 states. The high gain of the spot beam allows the local program services to be transmitted at a lower transponder output power level, and also permits frequency reuse by spatial beam separation. The dual Gregorian reflector system of the transmitting antenna on the satellite is fed by a complex feed structure that produces the Conus beam contours shown in Figure 11.6. The 6 dB contour of the beam is approximately 5.5° degrees wide in the E–W direction and 2.5° wide in the N–S direction, corresponding to a 3 dB contour that is 4.0°  1.8°. The estimated gain of the Conus beam is 36.5 dB. The coverage zone within the 6 dB contour of the Conus beam, taking account of the earth’s curvature, is approximately 4000 km E–W and 2000 km N–S. Florida, Alabama, and Louisiana, for example, are in rain zones M and N of the United States that has a rainfall rate of 50 mm per hour for five times the number of hours per year that this rain rate occurs in Washington, D.C., and much of the eastern portion of the United States. At Ku band, a rain rate of 50 mm/h will cause about 6-dB attenuation on a typical DBS-TV slant path, sufficient to ensure an outage of the DBS-TV signal. This rainfall rate is exceeded for about 5 h per year in Florida and 1 h per year in Washington D.C. The central and western parts of the United States have rainfall rates of 50 mm per hour for much less than 1 h per year, and therefore do not need such large link margins. See Chapter 8 for detailed maps of rainfall rates and the occurrence of heavy

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rain across the United States, and for the techniques used to convert rainfall statistics into attenuation data that can be used to calculate outage times. Note that much of the DBSTV coverage of the United States shown in Figure 11.6 lies inside the 6 dB contour of the satellite antenna beam, giving beam loss values between 4 and 6 dB, so the usual edge of beam loss of 3 db cannot be applied here. The regions within the 4dB to 6 dB contours are those that do not have frequent heavy rainfall. Some manufacturers of DBS-TV receiving systems offer larger dishes for customers living in high rainfall zones. Increasing the antenna diameter from 18 inches to 24 inches, for example, increases its gain by 2.5 dB. This increase in antenna gain adds directly to the rain fade margin of the receiver, and lowers the outage time in heavy rain.

11.4

DBS-TV LINK BUDGET In this discussion, rain attenuation statistics at Ku band will be used that are representative of many locations in the central and eastern parts of the United States, where typical path attenuation in rain exceeds 3 dB for 0.2% (15 h) and 6 dB for 0.01% (52 min) of an average year. The distribution of the fades is random, with some long fades in the heaviest of thunderstorms and numerous shorter fades in brief periods of heavy rain. Directv claims that receiving systems using 0.45 m (18 inch) diameter antennas designed for their DBS-TV transmissions have an availability exceeding 99.7%, which is an outage time of 0.3% of the year, a total of about 25 h in an average year. For much of the United States, this corresponds to a rain attenuation in the slant path of 3 dB and requires a link margin of 5.7 dB when allowance is made for the increase in antenna noise temperature that accompanies 3 dB of rain attenuation. A representative link budget for a GEO DBS-TV system serving the United States is shown in Table 11.2. The path length of 38,500 km is the maximum expected path length for a receiver in the United States and a satellite at longitude 101° W. The threshold CN value is set at 8.6 dB, corresponding to a system using QPSK with an implementation margin of 0.8 dB, forward error correction coding that produces 6 dB of coding gain,

TABLE 11.2

Link Budget for Ku-Band DBS-TV Receiver

Downlink power budget Transponder output power, 160 W Antenna beam on-axis gain (Conus coverage) Path loss at 12.2 GHz, 38,500-km path Receiving antenna gain, on axis Beam contour loss Miscellaneous and gaseous attenuation losses Received power, C Noise power budget Boltzmann’s constant, k System noise temperature, clear air, 143 K Receiver noise bandwidth, 20 MHz Noise power, N C/N in clear air Link margin over 8.6-dB threshold Link availability throughout U.S.

22.0 dBW 36.5 dB 205.9 dB 33.5 dB 3.0 dB 0.8 dB 117.7 dBW 228.6 dBW/K/Hz 21.6 dBK 73.0 dBHz 134.0 dBW 16.3 dB 7.7 dB Better than 99.7%

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and a maximum BER of 106. This requires a clear air CN ratio in the DBS-TV receiver of 8.6  5.7  14.3 dB in clear air. The link budget in Table 11.2 shows that a link margin of 7.7 dB is achieved for a receiver located on the 3 dB contour of the satellite antenna beam. Earth stations close to the 6 dB contour of the satellite beam have a link margin of 4.7 dB. A receiver located in the SE states of the United States, within the 2dB contour of the satellite beam, has a link margin of 8.7 dB. In the link budget shown in Table 11.2, the transponder output power is 160 W, corresponding to a transponder with a saturated output power of 200 W and 1 dB output backoff. A small amount of output backoff is normally used to avoid excessive AM-PM and PM-AM conversion in the transponder. The receiving antenna is a high efficiency design with an offset parabolic reflector 0.45 m in diameter and a circularly polarized feed. The offset design ensures that the feed system does not block the aperture of the antenna, which increases its efficiency. The gain of this antenna is 33.5 dB at 12.2 GHz assuming an aperture efficiency of 66%. The receiver in Table 11.2 is located at the 3dB contour of the transmitting antenna. Miscellaneous losses of 0.4 dB for gaseous attenuation at 12 GHz and 0.4 dB for receive antenna mispointing and other losses are included in the link power budget. The result is a received carrier power of 117.7 dBW in clear air conditions. The noise power budget of the link in Table 11.2 is based on a receiver noise bandwidth of 20 MHz, an antenna noise temperature of 35 K in clear air, and a 12-GHz LNA with a noise temperature of 110 K. The result is a noise power of 134.0 dBW in a noise bandwidth of 20 MHz referred to the input of the LNA, and a clear air CN ratio of 14.3 dB. The noise bandwidth of a digital receiver is set by the bandpass filters in the final IF stage of the receiver, immediately before the demodulator. The filter must be designed to match the symbol rate of the transmitted signal, and has a root raised cosine (RRC) transfer function. (See Chapter 5 for details of digital transmission techniques.) The noise bandwidth of all RRC filters is always equal to the symbol rate of the digital transmission. In the DBS-TV system described in Table 11.2, a QPSK signal with a symbol rate of 20 Msps is assumed, which results in a receiver noise bandwidth of 20 MHz. However, with MPEG-2 encoding of the video signals, the data rate is not constant. The VBS-S digital video standard is designed to allow for variable bit rates, and the figure of 20 Msps is probably a maximum value. Complex coding schemes are used on DBS-TV digital transmissions, making use of Reed–Solomon block encoding, interleaving, and an inner layer of convolutional encoding, as discussed in Chapter 7 and illustrated in Figure 7.11. The error mitigation scheme allows 6 dB of coding gain to be achieved with a code rate greater than half rate FEC, which allows more of the bits in the 40-Mbps bit stream to be allocated to data and fewer to parity bits. Data rates of 23 to 27 Mbps are reported for the data stream of some DBS-TV systems, with rate three-quarter inner convolutional coding. The overall code rate with 188204 outer Reed–Solomon coding and rate three-quarter inner convolutional coding is 0.69, which provides a message data rate of 27 Mbps with a coded bit rate of 39.1 Mbps.

11.5

ERROR CONTROL IN DIGITAL DBS-TV Digital DBS-TV transmissions typically use a transmitted symbol rate of 20 Msps using QPSK, which gives a bit rate of 40 Mbps. Error correction coding and control bits occupy 13 to 17 Mb of the bit stream, leaving 23 to 27 Mb for digital TV data. A 23-Mbps data stream can carry three live compressed digital video signals using MPEG 2 encoding, or up to 10 prerecorded and processed video signals. Prerecorded material, which

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comprises the majority of programming on satellite TV channels, is heavily processed to reduce its bit rate as far as 1.6 Mbps. When prerecorded material such as movies is digitized and processed through MPEG 2 compression, the data rate can be reduced to 1.6 Mbps, but results in digital artifacts appearing in the picture, especially when there is a lot of motion in the scene. A digital artifact appears as a freezing of the entire picture for a fraction of a second, caused by overloading of the MPEG 2 processing, or as a block or pixel of the wrong color. The artifacts can be removed one by one by a digital artist who works on the recorded material to paint out the effects. The final result is a movie or show that can be recorded in digital compressed form at an average rate of 1.6 Mbps for replay over the satellite. Live program material with a lot of motion in the picture can cause the bit rate of an MPEG 2 coded signal to increase above the average value of 6.2 Mbps. Mixing prerecorded and live material in a single transponder helps even out the bursty nature of live material. Error control in the digital video standard is achieved in a similar way to compact disc recordings. The compressed digital video signal bit stream is first split into blocks of bits and encoded with a Reed–Solomon linear block code. The coded bit stream is then interleaved (see Chapter 7 for details of interleaving) and encoded again with a convolutional code. The double layer of error control coding is called a concatenated code. At the receiver, the recovered bit stream is first decoded with a Viterbi decoding algorithm to remove the convolutional coding, and a limited number of errors in the bit stream are corrected. The corrected bit stream is then de-interleaved, and a Reed–Solomon decoding algorithm is applied. Figure 11.7 shows a block diagram of the coding and decoding operations performed on the bit stream. The coding process used for digital video and audio bit streams relies for much of its error correction capability on the fact that the end signal delivered to the user is analog. Bit errors in a digital video or audio signal result in the wrong voltage occurring when the errored word is converted to a voltage by the receiver’s digital-to-analog converter (DAC). If the system knows that a particular word is in error (i.e., the coding scheme detected an error but was unable to correct it), that word can be flagged and the error can be removed by interpolation of the analog waveform. When a word is known to contain a bit error, it is replaced by a new word which is calculated to have a value midway between the two adjacent words in the bit stream, as illustrated in Figure 7.10 in Chapter 7. The result is an interpolated value in the analog voltage waveform output by the DAC. The interleaving process ensures that most bit errors are single errors, increasing the probability that words contain only a single error and that the error will be detected.

Data input

(204, 188) Reed–Solomon encoder

Interleaver

Rate 3 /4 Convolutional encoder

Link

Convolutional decoder

De-interleaver

Reed–Solomon decoder

Data output

FIGURE 11.7 Block diagram of the coding and decoding operations in a DBS-TV signal.

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All error correcting codes are able to detect at least twice as many errors as they can correct. The use of analog waveform interpolation allows the video (and CD) error correction process to take advantage of the error detection capabilities of the outer layer of Reed–Solomon (R-S) coding to correct errors in the analog waveform without using error correction coding. If error correction coding were used instead of analog interpolation, at least twice as many parity bits would be needed in the bit stream, which would reduce the number of data bits drastically. The outer layer of Reed–Solomon linear block coding has good error detection properties, and the decoding process can be implemented in real time in a high-speed microprocessor. This is an important consideration in a realtime system, like TV broadcasting. Decoding of the error control codes must be possible in real time in a low cost processor, and this is one of the driving factors in the selection of the concatenated code used for CDs and for the video broadcast standard. Bit errors that are not corrected or detected by the error control coding process result in individual words that are in error. In an uncompressed digital television picture, a single word error would result in a pixel on the TV screen that is the wrong color. In a digital compressed TV signal, each word influences many pixels on the screen, and a word error may result in a block of the wrong color. MPEG 2 achieves a compression ratio of 8 to 10 with live video, so a single word error in an MPEG 2 encoded bit stream might be expected to cause errors in 8 or 10 pixels on the TV screen, or even more pixels if error propagation occurs. However, all digital compression schemes are designed to minimize the effect of errors in incoming words, so the impact of single word errors is usually confined to a small square on the screen which has the wrong information. As the bit error rate of the recovered bit stream in the receiver increases, the impact of the word errors becomes more severe, and larger blocks of the TV picture are corrupted. The receiving system is able to recognize the high error rate and will blank the screen until an acceptable error rate is restored. Thus a rain fade on a DBS-TV link which goes below the receiver threshold is characterized by the initial appearance of small squares of incorrect color on the TV screen, followed by larger block errors, and then a blank screen. When the rain intensity eases as the storm moves through the slant path, the signal will return above threshold and the picture will reappear on the TV screen. The user of a DBS-TV system is usually aware of a thunderstorm or very heavy rain in the locality when the signal goes below threshold and the TV screen goes blank. This seems to make loss of the TV picture more acceptable to users, and most DBS-TV customers appear to be satisfied with a nominal availability of 99.7%. The actual availability is undoubtedly higher than 99.7% for most of the customers in the United States, and few complaints seem to arise from the loss of signal in heavy rain.

11.6 MASTER CONTROL STATION AND UPLINK Direct broadcast television satellites are relay devices that provide a very large coverage area serving millions of customers. The many signals that are broadcast by the satellites are collected at a master control station and uplinked to the satellites by a group of large antennas with fade margins sufficient to overcome any expected rain fade. The video and audio signals that are uplinked to the DBS-TV satellites are available in prerecorded form on video tape or disc, or are collected from other satellites or fiber-optic lines. This is a large operation which requires substantial resources and a sizable labor force.

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FIGURE 11.8 Dishnetwork uplink earth station. (Photo courtesy of EchoStar Communications Corporation.)

Figure 11.8 shows the master control station of the Dishnetwork system operated by Echostar Communications Corporation; the location is in U.S. rain zone B2, providing a low probability of heavy rain. The statistics of region B2 show that a rain rate of 50 mm/h is exceeded for only 5 min in a typical year. Directv’s uplink earth station is located in Colorado, within the U.S. B2 rain zone. The major European uplink station for DBS-TV, operated by SES, is located in Luxemborg, which is also in the European rain zone B2. The uplink station must transmit hundreds of signals to the DBS-TV satellites 24 h a day, 365 days a year. Most of the signals are prerecorded, either from satellite feeds which are used to distribute new video and audio program material, or from archived material. The uplink stations have hundreds of tape and video disc players, all under computer control, which supply the video and audio signals for each channel. The signals are mainly stored in digital form, allowing direct multiplexing into bit streams for the individual transponders. Analog signals must be digitized and compressed before being multiplexed with other signals into the bit streams that are sent to each transponder. More details of the uplink centers operated by DBS-TV companies can be found from their web sites3–5. A simplified block diagram of the transmitting equipment at an uplink station is shown in Figure 11.9. One uplink antenna will typically transmit up to 16 RF channels to one DBS-TV satellite. Each RF signal is a QPSK modulated Ku-band carrier with a symbol rate of up to 20 Msps, occupying a bandwidth up to 27 MHz. The encoded, compressed, and multiplexed bit stream drives a video exciter which generates QPSK modulation of an intermediate frequency carrier, typically at 70 MHz. The 70 MHz signal is upconverted to the transponder input frequency in the transmitter, which contains a traveling wave tube high-power amplifier (HPA). The HPA is usually rated at a much higher power than its normal operating output power level, which provides sufficient output

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QPSK IF amplifier modulator

Upconverter

LPA

HPA

Multiplexer

M

MUX

70 MHz LO

MPEG 2 encoder ADC Analog video and audio signals

MPEG 2

17 GHz uplink antenna

Other RF signals

RF LO

Digital Reed–Solomon Convolutional multiplexer encoder encoder Interleaver MUX

Coder

I

Coder

Other digital signals

FIGURE 11.9 Simplified block diagram of a DBS-TV uplink earth station.

backoff of the HPA to ensure linear operation. The signals from any number of HPAs are multiplexed together in microwave combiners and sent to the antenna feed for transmission to the satellite.

11.7

INSTALLATION OF DBS-TV ANTENNAS Installation of a home satellite TV system offers an interesting challenge to home owners who do not have much knowledge of microwave antennas and satellite communication systems. A DBS-TV system antenna with a diameter of 0.45 m (18 inches) has a beamwidth of 4°, and needs to be pointed to an accuracy of 0.5° for optimum reception of the satellite signal. The problem is to provide a simple method for pointing the antenna in azimuth and elevation within about 2° so that a signal can be received and peaked. Directv and Dishnetwork offer an installation kit and book of instructions that make the process quite easy. The antenna is mounted onto a 2-inch diameter tube with a swiveling clamp. The lower end of the tube has a mounting bracket that is bolted to any convenient surface that provides a clear view of the southern sky, and the tube is set vertical using a plumb line or level. The antenna can then be rotated in azimuth. Elevation angle is set by rotating the dish about a horizontal axis provided by a bolt that is part of the swiveling clamp, and using a simple angle scale marked on the mounting. When the dish is set to the correct azimuth and elevation angles the bolts in the clamp are tightened down and the antenna is permanently set to the correct look angles. Directv provide an azimuth and elevation look angle calculator in the set-up menu of their receiver, and also a signal strength meter with both numerical and audio outputs. The azimuth and elevation look angles can also be found from tables and maps in the DBS-TV receiving system installation guides, and by using software that can be downloaded from web sites. The Directv on-screen calculator provides elevation and azimuth angles for the Directv satellites at longitude 101° W based on the zip code of the user, or the latitude and longitude of the earth station. The azimuth angle is given relative to magnetic north, and the installation kit includes a small compass that allows the user to set an approximate azimuth angle. The elevation angle can be set within 1° by careful adjustment

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of the clamp, which ensures that the satellite will be within the elevation plane 3 dB beamwidth of the antenna when the azimuth angle is correct. The procedure used to find the satellite is quite simple. The antenna is rotated in azimuth until a tone is heard from the TV receiver, indicating that a signal is being received. The antenna azimuth angle is adjusted to maximize the loudness of the tone and the signal strength value (a number between 0 and 100) shown on the screen, which ensures that the satellite is within the azimuth 3 dB beamwidth of the antenna. (The procedure typically requires two people because the TV set is rarely close to the antenna installation point.) Once the satellite signal has been acquired, the azimuth and elevation angles are adjusted alternately to maximize the signal strength, and the clamp is tightened down to hold the antenna at the correct angles. When the above procedure is followed with care, the antenna can be set to the correct azimuth and elevation angles in a few minutes.

11.8

SATELLITE RADIO BROADCASTING In the United States in 2001, two companies commenced transmission of digital radio signals from satellites, each offering 50 radio channels for a monthly subscription of about $10–13. Generically, the system is called Satellite Digital Audio Radio Service (SDARS)8. The target audience is in automobiles and other road vehicles, which is where most radio listening occurs in the United States. A vehicle equipped with an SDARS receiver can receive the same program anywhere in North America, a selling point that the companies hope will make their systems financially successful. Although subscription television, both satellite and cable, has been very successful in the United States, SDARS is the first attempt to create a subscription radio service—in contrast to terrestrial radio broadcasting which has always been free to the listener, supported by advertising revenue. SDARS vehicle radios have initial pricing in the $300–$500 range. The SDARS satellites have high-power transponders to compensate for the low gain omnidirectional antenna on the vehicle, and both systems use terrestrial repeaters in large cities to augment the satellite signal when blockage occurs by tall buildings. XM Satellite Radio Inc.9 based in Washington D.C. uses two satellites in GEO at 85° W and 115° W longitudes, appropriately named “Rock” and “Roll.” Each satellite transmits in a separate 3.7 MHz wide band in the frequency ranges 2332.5–2336.5 MHz and 2341–2345 MHz. Sirius Satellite Radio Inc.10 based in New York city, has three satellites equally spaced in a 24 h polar elliptical orbit centered at a longitude of 100° W with its apogee over North America. The satellites are above the horizon for listeners in the United States for approximately 16 h in each 24 h, with two of the three satellites transmitting in separate 4.2 MHz wide bands in the frequency ranges 2320–2324 MHz and 2328.5–2332.5 MHz. The highly elliptical orbit of the Sirius satellites can provide a higher elevation angle than a GEO satellite, which is desirable in cities to minimize blockage by tall buildings, but requires a handoff between satellites. Terrestrial repeaters operate in the same frequency bands, in the gaps between the two satellite downlink frequencies. Table 11.3 provides some details of the SDARS systems. XM repeaters receive their signals directly from the SDARS satellites; Sirius repeaters are fed by relay via a Ku-band GEO satellite. Because of the high probability of the satellite signals being blocked by buildings in a city and trees in rural areas, both systems utilize time diversity to overcome short interruptions in signal. The transmissions from the two satellites, and from the terrestrial repeaters, are delayed by varying amounts up to 5 s. The satellite radio receiver delays the

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TABLE 11.3 U.S. Satellite Digital Audio Radio Service Parameter

XM Satellite Radio Inc.

Sirius Satellite Radio Inc.

Number of satellites

Two in GEO at 85° W and 115° W 2332.5–2336.5 MHz 2341.0–2345.0 MHz 7050–7075 MHz 1500 in 70 cities 100

Three in highly elliptical 24-h orbit at 100° W 2320.0–2324.0 MHz 2328.5–2332.5 MHz 7060–7072.5 MHz 105 in 46 cities 100

4.0 Mbps

4.4 Mbps

TDM-QPSK

TDM-QPSK

Downlink frequencies Uplink frequencies Terrestrial repeaters Total number of audio channels Transmission rate before FEC Satellite downlink modulation

signals to achieve a common timing and then selects or combines the signals to achieve the best SN ratio. Signal transmission formats from the satellites are very similar to those used in DBS-TV: TDM-QPSK modulation is used to send multiple signals as a high-speed digital data stream, and concatenated Reed–Solomon outer layer and half rate convolutional inner layer coding is used for error control.

11.9

SUMMARY

Satellite broadcasting of television has become a major part of the satellite communications industry. In 1999, DBS-TV and video distribution earned more than half the revenues of the satellite communications industry, worldwide. Most DBS-TV and distribution of video signals is now digital, and Directv and Echostar in the United States have been major success stories with a total of 15 million customers by the end of 2000. DBS-TV systems operate with small antennas and low cost receiving systems, and offer a very large number of video and audio channels, making them attractive to customers. The link budget for a typical DBS-TV signal shows that the link margin is in the 4 to 8 dB range, which yields a better than 99.7% availability in the United States. Shaping of the transmitted beam from the satellite provides higher clear air CN ratios in regions where heavy rainfall occurs most often, such as the southeast of the United States. Digital DBS-TV signals are transmitted as a 20-Msps QPSK signal occupying about 27 MHz of transponder bandwidth. The 40-Mbps signal has a data rate between 23 and 27 Mbps with the remaining bits used for error control and system operation. DBS-TV digital signals make extensive use of error correction and error detection techniques in the form of a double layer of error control coding with

interleaving. The recovery of a good quality analog signal from a bit stream with bit errors relies on deinterleaving to spread out bursty errors, correction of some errors by the inner layer of convolutional coding, and detection of remaining, uncorrected, errors by the outer layer of Reed–Solomon linear block coding. Detected errors that remain after the decoding of the digital signal typically result in single word errors. Delivery of a bit stream through a direct broadcast satellite can be adapted to serve Internet users who require the download of large blocks of data. The development of DBS-TV satellites with spot beams for local news broadcasts has provided the higher satellite antenna gain needed for uplinks from 0.5-m dishes. Both Directv and Dishnetwork offer Internet access terminals with uplink capability, although the service is not available outside the spot beam coverage areas, where the connection to the ISP must be made through a terrestrial telephone link and the PSTN. Satellite radio broadcasting commenced in 2001 from three Sirius satellites in elliptical orbits and two XM satellites in GEO. The signals are transmitted in S band at 2.3 GHz and are aimed primarily at automobiles, which is where most people listen to the radio. Repeaters are used in city areas to overcome signal blockage by tall buildings.

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REFERENCES 1. LEON W. COUCH, Modern Communications Systems: Principles and Applications, Prentice Hall, Englewood Cliffs, NJ, 1995. 2. WTEC Panel Report on Global Satellite Communications Technology and Systems, International Technology Research Institute, Baltimore, MD, December 1998, ISBN 1-883712-51-3. Available from NTIS as NTIS report PB99-117954. Available online at http://itri.loyola.edu

3. 4. 5. 6. 7. 8.

www.directv.com www.dishnetwork.com www.slashdot.com www.ses.com.lux www.ssloral.com DAVID H. LAYER: “Digital Radio Takes to the Road,” IEEE Spectrum, Vol. 38, No. 7, pp. 40–46. July 2001. 9. www.xmradio.com 10. www.siriusradio.com

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12.1

INTRODUCTION The Global Positioning Satellite System (GPS) has revolutionized navigation and position location. It is now the primary means of navigation for most ships and aircraft and is widely used in surveying and many other applications. The GPS system, originally called NAVSTAR, was developed as a military navigation system for guiding missiles, ships, and aircraft to their targets. GPS satellites transmit L-band signals that are modulated by several codes. The CA (coarse acquisition) code was made available to the public in the mid-1980s. The secure high accuracy P code allows authorized users (mainly military) to achieve positioning accuracy of 3 m. This was the accuracy that the military users wanted for targeting smart bombs and cruise missiles, but such accuracies are also useful for autolanding aircraft in fog and for docking ships in bad weather. The first commercial use of GPS was in surveying, but by 1990 several companies had produced low-cost, handheld GPS receivers for general position location and navigation. Increased sales and larger volume production quickly brought down the price of a GPS receiver, and the market expanded rapidly. GPS receivers are now a consumer product, and will soon be found in every car and cellular telephone. The GPS system has been successful because it provides a direct readout of the present position of a GPS receiver with a typical accuracy of 30 m. There are other position location systems, such as LORAN, (a contraction of long range navigation) that can also provide direct readout of position, but not with the accuracy and reliability of GPS. The success of GPS is an excellent example of what satellites do best: broadcasting. An unlimited number of GPS receivers can operate simultaneously because all that a GPS receiver has to do to locate itself is to receive signals from four GPS satellites. The GPS space segment consists of 24 satellites in medium earth orbit (MEO) at a nominal altitude of 20,200 km with an orbital inclination of 55º. The satellites are clustered in groups of four, called constellations, with each constellation separated by 60º in longitude. The orbital period is approximately one-half a sidereal day (11 h 58 min) so the same satellites appear in the same position in the sky twice each day. The satellites carry station-keeping fuel and are maintained in the required orbits by occasional stationkeeping maneuvers, just like GEO satellites. The orbits of the 24 GPS satellites ensure that at any time, anywhere in the world, a GPS receiver can pick up signals from at least four satellites. Up to 10 satellites may be visible at some times, and more than four satellites are visible nearly all of the time. Replacement satellites are launched as needed, so there may be more than 24 operational GPS satellites at any given time. Figure 12.1 shows a GPS satellite. The satellites weigh 1877 kg at launch and have a design lifetime of 10 years. In 2000, there were 30 GPS satellites in orbit, some of which were spares. Because GPS is an integral part of the defense of the United States, spare GPS satellites are kept in orbit and more spares are ready for immediate launch. The GPS

458

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FIGURE 12.1 GPS block II-F satellite.

system is operated by the U.S. Air Force from the GPS master control station (MCS) at Falcon Air Force Base in Colorado Springs, Colorado. The MCS and a series of subsidiary control stations around the globe continuously monitor all GPS satellites as they come into view and determine the orbit of each satellite. The MCS and other stations calculate ephemeris data for each satellite, atomic clock error, and numerous other parameters needed for the navigation message. The data are then transmitted to the satellite using a secure S-band link and used to update onboard data. There are five GPS monitor stations located in Hawaii, Colorado Springs, Ascension Island in the Atlantic Ocean, Diego Garcia in the Indian Ocean and Kwajalein in the Pacific Ocean1. The monitor stations have precise cesium time standards and make continuous measurements of range to all visible satellites. These measurements are performed every 1.5 s, and used to provide updates for the navigation messages. The position of a GPS receiver is found by trilateration, which is one of the simplest and most accurate methods of locating an unknown position. In trilateration, the distance of the unknown point from three known points is measured. The intersection of the arcs corresponding to three distances defines the unknown point relative to the known points, since three measurements can be used to solve three equations to give the latitude, longitude, and elevation of the receiver. The distance between a transmitter and a receiver can be found by measuring the time it takes for a pulse of RF energy to travel between

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the two. The distance is calculated using the velocity of electromagnetic waves in free space, which is assumed to be equal the velocity of light, c, with c  299,792,458 m/s. Time can be measured electronically more accurately than any other parameter by the use of atomic clocks, and this is how the GPS position location system can achieve a measurement accuracy of 1 m in a distance of 20,000 km. To achieve a position location accuracy of 1 m, timing measurements must have an accuracy better than 3 ns. This is possible with modern digital circuitry and a great deal of averaging. Each satellite carries several high accuracy atomic clocks and radiates a sequence of bits that starts at a precisely known time. A GPS receiver contains a clock that is synchronized in turn to the clock on each satellite that it is receiving. The receiver measures the time delay of the arrival of the bit sequence, which is proportional to the distance between the satellite and the GPS receiver. When the distance of a GPS receiver from three satellites has been measured, the remaining piece of information that is required is the position of each satellite. This is calculated in the GPS receiver using the ephemeris for the satellite orbits that are broadcast by each satellite in its navigation message. Since the time at which the transmitted bit sequence started is known at the receiver, the position of the satellite at that time can be calculated from its orbital data. Making the calculation for four satellites provides the receiver with sufficient information to determine its position with very good accuracy. Four satellites, rather than three, are needed because the clock in the receiver is not inherently accurate enough. The fourth distance measurement provides information from which clock errors in the receiver can be corrected and the receiver clock synchronized to GPS time with an accuracy better than 100 ns. GPS satellites transmit two signals at different frequencies, known as L1 and L2. The L2 signal is modulated with a 10.23 Mbps pseudorandom (PN) bit sequence called the P code that is used by military positioning systems. The P code is transmitted in an encrypted form known as the Y code, which restricts the use of the P code to authorized users. The L1 frequency carrier is modulated by a 1.023 Mbps PN sequence called the CA code that is available for public use, and also carries the P code as a quadrature modulation. The higher bit rate of the P code provides better measurement accuracy than the 1.023 Mbps CA code. CA stands for coarse acquisition and P stands for precise. GPS systems using the secure Y code require the CA code as an intermediate step in making distance measurements with high accuracy. The accuracy of CA code receivers was deliberately degraded some of the time by a process called selective availability (SA). SA causes variations in the CA code satellite transmissions that result in less accurate calculation of position. SA was discontinued in May 2000, but can be reinstituted if the President of the United States declares a National Emergency. The GPS system provides two categories of service. The precise positioning service (PPS) receivers track both P code and CA code on L1 and L2 frequencies. The PPS is used mainly by military users, since the P code is encrypted into the Y code before transmission and requires decryption equipment in the receiver. Standard positioning service (SPS) receivers track the CA code on L1. This is the service that is used by the general public. The P(Y) and CA codes transmitted by each satellite create direct sequence spread spectrum signals which occupy the same frequency bands. Both the CA codes and the P codes are publicly available, but the P code cannot be recovered in a GPS receiver without a knowledge of the Y code decryption algorithm. In this discussion we will concentrate on the CA code and its use in position location. The former USSR built and operated a global navigation system that is very similar to GPS, known in the West as Glonass for global navigation satellite system. Almost everything about Glonass is similar to GPS except the multiple access technique. Glonass

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uses FDMA, with a different transmit frequency at each satellite. The equivalents of the P code and CA code can be transmitted by Glonass satellites in RF bandwidths of 20 kHz and 2 kHz, so 100 satellites can be accommodated in a bandwidth of 2 MHz. An FDMA receiver with 100 channels is simpler than a CDMA receiver. A frequency synthesizer that can be tuned to the unique frequency of each satellite is required, rather than the digital correlators that recover the GPS signals in a CDMA receiver. The European Union is considering building a similar satellite navigation system called Galileo, scheduled for operation by 2008, to provide precise navigation signals without dependence on the United States.

12.2

RADIO AND SATELLITE NAVIGATION Prior to the development of radio, navigation was by compass and landmarks on land, and by the sun and stars at sea. Neither technique provides high accuracy, and shipwrecks caused by inaccurate navigation and foggy weather were a common occurrence. On land, people often got lost in wilderness areas (and still do). Pilots of light aircraft, relying solely on a map and landmarks, would get lost and run out of fuel before they found somewhere to land. With a GPS receiver and a map, it is impossible to get lost. GPS receivers are very popular with airplane pilots, owners of sea-going boats, and wilderness hikers. The development of aircraft that could fly above the clouds, and particularly the building of large numbers of bomber aircraft in the 1930s, made radio navigation essential. Military thinking after WWI, and during WWII, placed high reliance on the ability of bomber aircraft to win a war by destroying the weapon manufacturing capability of the enemy. During WWII, the allies sent 1000 bomber aircraft at a time to targets in Germany, causing immense destruction to many cities. The philosophy of mass destruction continued after WWII with the development of nuclear bombs, intercontinental ballistic missiles (ICBMs), and cruise missiles. However, bomber aircraft, ICBMs, and cruise missiles must find their targets, so accurate navigation is an essential part of each of these weapon systems. This demand for accurate targeting of airborne weapons led to the development of GPS. However, the majority of GPS users are now civilian, and the worldwide market for GPS equipment is projected to be worth $25 B by 2005. Commercial aircraft fly on federal airways using VOR (VHF omni range) beacons. The airways are 8 miles wide to allow for the angular accuracy of VOR measurements, which is better than 4°. GPS will eventually replace VOR navigation, allowing aircraft to fly directly from point of origin to destination, but the system of VOR beacons in the United States is likely to remain for many years as a backup to GPS. GPS can provide a single navigation system with better accuracy and reliability than all earlier radio navigation aids. It can provide navigation of aircraft directly between airports, instead of indirectly via airways, while providing absolute position readout of latitude and longitude. Differential GPS can be used instead of ILS to provide the required straight line in the sky for an instrument approach to a runway, and can be linked to an autopilot to provide automatic landing of aircraft in zero visibility conditions. Ships can safely navigate and dock in treacherous waters in bad weather by using differential GPS. Eventually, GPS will replace all other means of navigation, although some may be retained as backup systems in case of failure of the GPS receiver(s) or jamming of the signals. GPS was preceded by an earlier satellite navigation system called Transit, built for the U.S. Navy for ship navigation, which achieved much lower accuracy and became obsolete when GPS was introduced. Transit satellites were in low earth orbits and the

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SIDEBAR The earliest radio navigation systems, developed in the 1930s and 1940s, were simple transmitting stations operating in the mf (AM) radio band. A special antenna and receiver can make an indicator needle point at the transmitter, so that an aircraft or a ship can home in on the beacon. The beacons used by aircraft are called non-directional beacons (NDB) and the receiver is called an automatic direction finder (ADF). Most commercial aircraft carry ADF receivers, which are still required for instrument approach procedures at some airports. NDBs and an ADF receiver were the main means of radio navigation for aircraft in the 1920s and 1930s, but have several serious disadvantages. If a strong wind is blowing across the path that the aircraft is taking to the NDB, it may fly a curved path instead of a straight line. The NDB was largely superseded by the VOR beacon in the 1940s. VOR stands for VHF omni range. A VOR transmitter generates a rotating VHF radio beam and also radiates a continuous sine wave signal that is phase referenced to the time that the rotating beam sweeps through north. A VOR receiver on the aircraft synchronizes to the reference signal and measures the angle of the beam relative to north at the time it is received. With two VOR transmitters and a map showing their locations, the aircraft can determine its position. Many VOR stations have DME (distance measuring equipment). The aircraft DME equipment transmits a pair of pulses and measures the time for the round-trip to the VOR and back, which provides a measurement of range to the VOR. With knowledge of range and angle to a VOR, navigation is possible using a single VOR-DME station. WWII aircraft needed to navigate to targets over enemy territory where there were no VORs available. Hyperbolic navigators were developed in Germany, Britain, and the United States during WWII to provide radio navigation at longer ranges than can be achieved in the VHF band by using frequencies between 100 kHz and 2 MHz. The frequencies can propagate round the earth’s curvature making longrange navigation possible. A hyperbolic navigator has

three radio transmitters that transmit at the same frequency, or very close frequencies. In the earlier forms, the phase of the radio wave was used, but later systems like LORAN time the arrivals of pulse transmissions. The receiver compares the phase or time of arrival of the radio signals from two transmitters and calculates the difference in distance between the two transmitters. A line with a constant difference in the distance between two points is a hyperbola with the two transmitters at the foci. A third transmitter provides two more hyperbolas, and their intersection locates the receiver, hence the name hyperbolic navigator. LORAN uses pulse transmissions in the 100–500 kHz RF band, and can provide reliable navigation with accuracy of a fraction of a mile at ranges of hundreds of miles from the transmitters. The U.S. Coast Guard built LORAN stations along the coastline of the United States to provide navigation assistance for ships in coastal waters where the danger of running aground is greatest. LORAN is steadily being replaced by GPS, and the Coast Guard now has differential GPS systems in place to help ships navigate in estuaries and rivers. Instrument landing systems (ILS) are essential when aircraft must land in conditions of poor visibility. An ILS installation at an airport provides two radio beams that allow the aircraft to fly an approach along a straight line to the runway threshold. The localizer, a VHF transmitter and antenna at the end of the runway, provides two modulated beams in the horizontal plane. A vertical needle on a course deviation indicator (CDI) in the aircraft cockpit shows the aircraft’s lateral position relative to a line leading to the runway threshold. A glide slope transmitter at the side of the runway transmits another radio beam which points upward at about 3°. A horizontal needle on the CDI shows the position of the aircraft relative to the glide slope, with a sensitivity of 10 ft as the aircraft approaches the ground. The pilot of the aircraft tries to keep both the CDI needles centered, so that the aircraft flies a straight line down the glide slope and arrives at a height of 50 feet above the runway threshold.

system used the Doppler shift observed at the receiver when a beacon signal was transmitted by the satellite. Because of the high velocity of LEO satellites—about 7.5 km/s— their signals are significantly shifted up in frequency when the satellite appears over the horizon with a component of velocity toward the receiver. The Doppler shift falls to zero as the satellite passes the observer, and then becomes negative as the satellite flies away.

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Observation of the Doppler shift with time, which may need to be as long as 10 min, and a knowledge of the satellite orbit, allows calculation of the receiver’s position. There was never a sufficient number of Transit satellites to provide continuous position data, and the long time required to obtain an accurate position fix was a disadvantage. A similar system called SARSAT, for search and rescue satellite, is used to find emergency locator transmitters (ELTs) on aircraft that have crashed. Most general aviation aircraft carry an ELT, which turns on at a frequency of 121.5 MHz when subjected to high G forces, as might be experienced if the aircraft crashes. Certain LEO satellites carry 121.5-MHz receivers that relay the signals to earth stations at rescue coordination centers. If an aircraft ELT turns on, a SARSAT satellite will eventually fly by and relay a Doppler shifted signal to the rescue station. Analysis of the Doppler shift over the observation period provides information about the location of the ELT, but with an accuracy of only 1 or 2 km. Almost 97% of ELT locations turn out to be false alarms—the ELT was dropped or accidentally turned on. It seems probable that GPS and cellular phones or satellite phones will eventually replace the SARSAT system.

12.3

GPS POSITION LOCATION PRINCIPLES The basic requirement of a satellite navigation system like GPS is that there must be four satellites transmitting suitably coded signals from known positions. Three satellites are required to provide the three distance measurements, and the fourth to remove receiver clock error. Figure 12.2 shows the general arrangement of position location with GPS. The three satellites provide distance information when the GPS receiver makes three measurements of range, Ri, from the receiver to three known points. Each distance Ri can be thought of as the radius of a sphere with a GPS satellite at its center. The receiver lies at the intersection of three such spheres, with a satellite at the center of each sphere. Locally, at the receiver, the spheres will appear to be planes since the radii of the spheres are very large. A basic principle of geometry is that the intersection of three planes completely defines a point. Thus three satellites, through measurement of their distances to the receiver, define the receiver location close to the earth’s surface. There is another point in outer space where the three spheres intersect, but it is easily eliminated in the calculation process.

S2 S3 S1

S4

FIGURE 12.2 General arrangement of position locations with GPS. The aircraft must receive signals from four GPS satellites to be able to determine its position.

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Although the principles by which GPS locates a receiver are very simple, requiring only the accurate measurement of three ranges to three satellites, implementing the measurement with the required accuracy is quite complex. We will look first at the way in which range is measured in a GPS receiver and then consider how to make the measurements. Range is calculated from the time delay incurred by the satellite signal in traveling from the satellite to the GPS receiver, using the known velocity of EM waves in free space. To measure the time delay, we must know the precise instant at which the signal was transmitted, and we must have a clock in the receiver that is synchronized to the clock on the satellite. GPS satellites each carry four atomic clocks which are calibrated against time standards in the GPS control stations around the world. The result is GPS time, a time standard that is available in every GPS satellite. The accuracy of an atomic clock is typically 1 part in 1011. However, it is too expensive to include an atomic clock in most GPS receivers, so a standard crystal oscillator with an accuracy of 1 in 105 or 1 in 106 is used instead. The receiver clock is allowed to have an offset relative to the GPS satellite clocks, so when a time delay measurement is made, the measurement will have an error caused by the clock offset. For example, suppose the receiver clock has an offset of 10 ms relative to GPS time. All distance measurements will then have an error of 3000 km. Clearly, we must have a way to remove the time error from the receiver clock before we can make accurate position measurements. CA code receivers can synchronize their internal clocks to GPS time within 170 ns, corresponding to a distance measurement uncertainty of 50 m. Repeated measurements and integration improve the position location error to well below 50 m. It is surprisingly easy to remove the clock error, and this removal is one of the strengths of GPS. All that is needed is a time measurement from a fourth satellite. We need three time measurements to define the location of the receiver in the three unknown coordinates x, y, and z. When we add a fourth time measurement we can solve the basic position location equations for a fourth unknown, the receiver clock offset error . Thus the four unknowns in the calculation of the location of the receiver are x, y, z, and .

Position Location in GPS First, we will define the coordinates of the GPS receiver and the GPS satellites in a rectangular coordinate system with its origin at the center of the earth. This is called the earth centered earth fixed (ECEF) coordinate system, and is part of the WGS-84 description of the earth. WGS-84 is an internationally agreed description of the earth’s shape and parameters, derived from observations in many countries4. GPS receivers use the WGS-84 parameters to calculate the orbits of the GPS satellites with the accuracy required for precise measurement of the range to the satellites. The Z-axis of the coordinate system is directed through the earth’s North Pole and the X- and Y-axes are in the equatorial plane. The X-axis passes through the Greenwich meridian—the line of zero longitude on the earth’s surface, and the Y-axis passes through the 90° east meridian. The ECEF coordinate system rotates with the earth. The receiver coordinates are (Ux, Uy, Uz), and the four satellites have coordinates (Xi, Yi, Zi), where i  1, 2, 3, 4. There may be more than four satellite signals available, but we use only four signals in a position calculation. The measured distance to satellite i is called a psuedorange, PRi, because it uses the internal clock of the receiver to make a timing measurement that includes errors caused by receiver clock offset. The geometry of a GPS measurement is illustrated in Figure 12.3.

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465

S2

S1

S3

R2

R1

FIGURE 12.3 Position location by measurement of the distance to three satellites. The GPS receiver is located at point X, where three spheres with radii R1, R2, and R3 intersect. The centers of the spheres are the three GPS satellites S1, S2, and S3. If the distances R1, R2, and R3 are measured, the location of the point X can be uniquely defined.

R3

X

Psuedorange, denoted as PRi, is measured from the propagation time delay Ti between the satellite (number i) and the GPS receiver, assuming that EM waves travel with velocity c. PRi  Ti  c

(12.1)

The distance R between two points A and B in a rectangular coordinate system is given by R 2  1xA  xB 2 2  1yA  yB 2 2  1zA  zB 2 2

(12.2)

The equations which relate pseudorange to time delay are called ranging equations: 1X1  Ux 2 2 1X2  Ux 2 2 1X3  Ux 2 2 1X4  Ux 2 2

   

1Y1 1Y2 1Y3 1Y4

 Uy 2 2  Uy 2 2  Uy 2 2  Uy 2 2

   

1Z1 1Z2 1Z3 1Z4

 Uz 2 2  Uz 2 2  Uz 2 2  Uz 2 2

   

1PR1 1PR2 1PR3 1PR4

 tc2 2  tc2 2  tc2 2  tc2 2

(12.3)

where  is receiver clock error (offset, or bias). The position of the satellite at the instant it sent the timing signal (which is actually the start of a long sequence of bits) is obtained from ephemeris data transmitted along with the timing signals. Each satellite sends out a data stream that includes ephemeris data for itself and the adjacent satellites. The receiver calculates the coordinates of the satellite relative to the center of the earth, (Xi, Yi, Zi), and then solves the four ranging equations for the four unknowns using standard numerical techniques for the solution of nonlinear simultaneous equations. (The equations are nonlinear because of the squared terms.) The four unknowns are the location of the GPS receiver, (Ux, Uy, Uz), relative to the center of the earth and the clock offset —called clock bias in GPS terminology. The receiver position is then referenced to the surface of the earth, and can be displayed in latitude, longitude, and elevation. Typical accuracy for a low-cost GPS receiver using the GPS CA code is 30 m defined as a 2DRMS error. The term DRMS means the distance root mean square error of the measured position relative to the true position of

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the receiver. If the measurement errors are Gaussian distributed, as is often the case, 68% of the measured position results will be within a distance of 1DRMS from the true location and 95% of the results will be within 2DRMS of the true location. Accuracy in GPS measurements is usually defined in terms of 2DRMS, in the horizontal or vertical plane. In practice, the error surface that encloses 68 or 95% of all measurements is not a circle but an ellipse, and the error in any dimension is affected by several dilution of precision (DOP) factors. DOP is discussed later in this chapter. The U.S. Department of Defense has the ability to degrade the position measurement accuracy of CA code receivers by applying selective availability (SA). SA exists to allow the accuracy of CA code receivers to be degraded in the event of a national emergency (i.e., enemy action) affecting the United States and was applied to the CA signals most of the time until May 2000. SA was switched off on May 1, 2000, and will not be used again unless the security of the United States is threatened. With SA off, the accuracy of GPS position measurements with the CA code increased dramatically, particularly in the vertical dimension. Variation in elevation readout of a typical CA code receiver with SA on could be as large as 200 m. With SA off, the variation may be as small as 10 m. Selective availability and atmospheric propagation effects (tropospheric and ionospheric) all cause errors in the timing measurements made by a GPS receiver, leading to position location errors. The atmosphere and the ionosphere introduce timing errors because the propagation velocity of the GPS signals deviates from the assumed free space value. The errors can be largely removed if a number of GPS reference stations are built at precisely known locations. The stations observe the GPS signals and compute the current error in position as calculated from GPS data. This information can then be broadcast to all GPS users as a set of corrections to be applied to GPS measurements. The system is called a wide area augmentation system (WAAS). A network of 24 WAAS stations built in North America for the U.S. Federal Aviation Administration (FAA) provides aircraft with improved position measurement accuracy. Using WAAS, accuracies of a few meters can be obtained with CA code receivers. In the event of a national emergency, WAAS would be switched off to prevent enemies using GPS for accurate targeting of weapons. WAAS also includes an integrity monitoring system to ensure that the GPS signals used by aircraft do not contain errors which could cause false readings. WAAS is required to send a warning of possible errors within 5.6 s if a problem is detected with any GPS satellite signal. Similarly, a single reference station at a known location—for example, an airport— can determine the local measurement error in GPS and broadcast this information to GPS users so that greater accuracy can be obtained with a CA code receiver. This is one (simple) form of differential GPS (DGPS). More complex forms of differential GPS use a reference station which transmits the signals received from each GPS satellite so that phase comparisons can be made by the receiver. With lengthy integration times and a sophisticated phase comparison receiver, differential GPS accuracies of 1 cm can be obtained. With DGPS, the receiver computes its position relative to the reference station rather than in latitude and longitude. Differential GPS is used when a vehicle needs to be positioned accurately with respect to a fixed point, such as an aircraft with respect to a runway or a ship with respect to a berth.

GPS Time The clock bias value  which is found as part of the position location calculation process can be added to the GPS receiver clock time to yield a time measurement that is synchronized

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to the GPS time standard. The crystal oscillator used in the GPS receiver is highly stable over a period of a few seconds, but will have a frequency which changes with temperature and with time. Temperature changes cause the quartz crystal that is the frequency determining element of a crystal oscillator to expand or contract, and this changes the oscillator frequency. Crystals also age, which causes the frequency to change with time. The changes are very small, but sufficient to cause errors in the clock time at the receiver when the clock is not synchronized to a satellite. Calculating the clock bias by solving ranging equations allows the receiver clock time to be updated every second or two so that the GPS receiver time readout is identical to GPS time. Every GPS receiver is automatically synchronized to every other GPS receiver anywhere in the world through GPS time. This makes every GPS receiver a super clock, which knows time more accurately than any other time standard. Prior to the widespread use of GPS receivers, standard time transmissions were broadcast by the U.S. National Institute of Science and Technology (NIST, formerly the Bureau of Standards). The broadcasts were made in the HF (shortwave) band, and could be received throughout the United States. However, the HF signals propagate over long distance by reflection from the ionosphere, which introduces an uncertain delay into the time of arrival of the signal. The time standard provided by GPS is typically accurate to better than 170 ns, and has been used to synchronize electric power generators across the United States, for scientific applications that require synchronized clocks in different locations, and as a long-term frequency standard. The time standard on board each GPS satellite consists of two cesium clocks plus two rubidium clocks (atomic clocks). An atomic clock uses the fundamental resonance of the cesium or rubidium molecule as a frequency reference to lock a crystal oscillator. In the GPS satellites, the master oscillator is at 10.23 MHz; all code rates, the L1, and the L2 RF frequencies are multiples or submultiples of 10.23 MHz. The atomic clocks are updated by the controlling ground stations to keep them within 1 s of Universal Time Coordinated (UTC), and the navigation message broadcast by each satellite contains information about its current clock errors relative to GPS time. (UTC is a worldwide time standard. Greenwich Mean Time (GMT) is equal to UTC.)

12.4

GPS RECEIVERS AND CODES GPS satellites transmit using pseudorandom sequence (PN) codes. All satellites transmit a CA code at the same carrier frequency, 1575.42 MHz, called L1, using BPSK modulation. The L1 frequency is 154 times the master clock frequency of 10.23 MHz. The CA code has a clock rate of 1.023 MHz and the CA code sequence has 1023 bits, so the PN sequence lasts exactly 1.0 ms. The exact values of the frequencies are about 0.005 Hz lower than stated here to allow for relativistic effects caused by the high velocity of the satellites in their orbits (3.865 km/s). (GPS measurements are one of the few examples where relativistic effects must be taken into account, because the clocks are mounted on platforms moving at very high speeds.) The P code is transmitted using BPSK modulation at the L2 carrier frequency of 1227.6 MHz (120  10.23 MHz), and is also transmitted with BPSK modulation on the L1 carrier frequency, in phase quadrature with the CA code BPSK modulation. Figure 12.4 shows the way in which the L1 and L2 signals are generated on board a GPS satellite. The CA and P code transmissions from all GPS satellites are overlaid in the L1 and L2 frequency bands, making GPS a direct sequence spread spectrum (DS-SS) system

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Q

π/2 L1 1575 MHz

I

Σ

C/A code C/A + NAV

1.023 Mbps Navigation message

50 bps

L1 + P + C/A + NAV L1 output

multipliers P + NAV

P code

10.23 Mbps

L2 1227 MHz

L2 output L2 + P + NAV

I

FIGURE 12.4 Signal generation in a GPS satellite.

(see Chapter 6 for details of spread spectrum techniques). The receiver separates signals from individual GPS satellites using knowledge of the unique CA code that is allocated to each satellite. At most, 12 GPS satellites can be seen by a receiver at any one time, so the coding gain in the spread spectrum receiver must be sufficient to overcome the interference created by 11 unwanted signals while recovering the twelfth wanted signal.

The CA Code The CA codes transmitted by GPS satellites are all 1023 bit Gold codes. GPS CA Gold codes are formed from two 1023 bit m-sequences, called G1 and G2, by multiplying together the G1 and G2 sequences with different time offsets. An m-sequence is a maximum length pseudorandom (PN) sequence, which is easy to generate with a shift register and feedback taps. A shift register with n stages can generate a PN sequence 2n  1 bits in length. The bit pattern is set by the feedback taps and combining logic of the shift register. The PN sequences G1 and G2 are both generated by 10-bit shift registers and are therefore both 1023 bits long. The clock rate for the CA code is 1.023 MHz, so each sequence lasts 1.0 ms. Figure 12.5 shows a generator diagram for the CA code. The CA code for a particular satellite is created with an algorithm that includes the identification number of the GPS satellite, thus creating a unique code for each satellite. The satellite with ID number i has a CA code sequence Ci(t) Ci 1t2  G11t2  G21t  10iTc 2

(12.4 )

where Tc  clock period for the CA code. There are 64 Gold sequences available for satellites numbered 1 through 64. A total of 100 Gold sequences can be created using the algorithm in Eq. (12.4), but not all the

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+

G1 3

10

10 bit G1 shift register

X1

Reset to 1

+ Phase selector

÷10

+

S1

2

C/A code

S2

3

10.23 MHz Master clock

6

+

8 9 10

G2

X1

1023 decode

1 kHz clock

÷20 50 Hz clock

10 bit G2 shift register FIGURE 12.5 C/A code generator.

sequences have sufficiently low cross-correlation properties, and reference 4 states that only 37 are actually used in the GPS system. Low cross-correlation of the sequences is a requirement because the GPS receiver can pick up signals from as many as 12 satellites at the same time. A correlator in the receiver looks for one of the sequences and must reject all other sequences that are present. Two CA code sequences with zero cross-correlation would achieve a rejection ratio of 1023, but the 64 available CA code sequences will not all have zero cross-correlation. The selected group of 37 are the sequences with the lowest levels of cross-correlation among the available set of 100 Gold code sequences. They also have low autocorrelation time sidelobes, another requirement of direct sequence spread spectrum systems. The CA code sequence length of 1.000 ms gives range ambiguity of 300 km, since the code travels at a velocity of approximately 3  108 m/s and therefore has a length in space of 3  105 m. The entire CA code sequence repeats in space every 300 km, leading to ambiguity of position only if the GPS receiver is in outer space. The ambiguity is easily resolved if the receiver knows roughly where it is; just knowing that the receiver is located close to the earth’s surface is usually sufficient. The user can enter the approximate location of the GPS receiver when it is first switched on to help resolve any ambiguities quickly. Figure 12.6 shows a simplified block diagram of a CA code GPS receiver. The antenna is typically a circularly polarized patch antenna with an LNA mounted on the printed circuit board. A conventional superhet receiver is used to generate an IF signal in a bandwidth of about 2 MHz, which is sampled using I and Q sampling techniques and processed digitally. The digital portion of the receiver includes a CA code generator, a correlator,

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Receiving antenna LNA

Digital signal Down-converter IF amplifier A/D converter processing A/D

DSP Clocks

Navigation message

Timing data

Local oscillator Microprocessor

Display FIGURE 12.6 Simplified GPS receiver.

and a microprocessor that makes the timing measurements and calculates the receiver’s position. Most GPS receivers make use of a 12-channel IC chip set that can be purchased for about $25.00 (Year 2000 prices).

12.5

SATELLITE SIGNAL ACQUISITION The GPS receiver must find the starting time of the unique CA code for each of four satellites. This is done by correlating the received signal with stored CA codes, as in any direct sequence spread spectrum system. (See Chapter 6 for details of this process.) Usually, the receiver will automatically select the four strongest signals and correlate to those. If the geometry of the strongest satellites is poor, that is, the satellites are close together and have pseudoranges that are nearly equal, the receiver may also use several weaker signals. If the receiver is making a cold start, with no information about the current position of GPS satellites, or its own location, it must search all 37 possible CA codes until it can correlate with one. Once correlation is obtained, the data stream (called the navigation message) from that satellite can be read by the receiver. The data stream contains information about the adjacent satellites, so once one signal is correlated, the receiver no longer needs to search through all the other 36 possible codes to find the next satellite; it can go directly to the correct code. Searching all 36 CA codes of 1023 bits for correlation can be a slow process. In the worst case, 36 codes might have to be searched before a correlation could be obtained. However, available satellites in 2000 all had numbers between 13 and 45 5, so, on average, 16 codes might have to be searched before correlation is successful. A direct sequence spread spectrum receiver locks to a given code by matching the locally generated code to the code received from the wanted satellite. Since the start time of the code transmitted by the satellite is not known when the receiver commences the locking process, an arbitrary start point must be selected. The locally generated code is compared to the received code, bit by bit, through all 1023 bits of the sequence, until either lock is found, or the receiver concludes that this is not the correct code for the satellite signal it is receiving. If the starting time for the locally generated code was not selected correctly, correlation will not be obtained immediately. (This will occur with a probability of 99.9% when

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the timing of the locally generated sequence is selected at random.) The locally generated code is then moved forward one bit in time, and correlation is attempted again. The process is continued 1023 times until all possible starting times for the locally generated code have been tried. If the satellite with that particular CA code is not visible, no correlation will occur and lock will not be achieved. It takes a minimum of 1 s to search all 1023 bit positions of a 1023 bit CA code, so in a typical case, it will take at least 15 s to acquire the first satellite. Many receivers search for a given CA code several times before moving to the next code, so several minutes may elapse before the correct CA code is found, given no other information. Once one CA code is found, the remaining satellites can then be acquired in a few seconds because their IDs are known from the data transmitted in the navigation message of each satellite. Although it takes only 20 s on average to lock to the CA code of one satellite, the receiver must find the Doppler frequency offset for at least one satellite before correlation can occur. The receiver bandwidth is matched to the bandwidth of the CA code. The theoretical noise bandwidth of the CA code receiver is 1.023 MHz and the velocity of the satellites is 3.865 km/s. The angle between the spacecraft velocity vector and a receiver on earth is 76.1° when a GPS satellite is at the horizon, so the maximum velocity component toward a receiver is vr  928 m/s, giving a maximum Doppler shift in the L1 signal of vr  4.872 kHz, ignoring the effect of earth rotation. Allowing the satellite to reach an elevation angle of 5° before it is used for a position measurement limits the Doppler shift that must be accommodated by the receiver to 4 kHz. From a cold start, the receiver must try eight Doppler frequency shifts of up to 4 kHz in 1-kHz steps when searching for the signal from a satellite. This can increase the acquisition time of the first satellite to several minutes. Figure 12.7 illustrates the search process. There are eight possible Doppler shifts for each signal, and 1023 possible code positions, giving 8184 possible signal states that must be searched. Once any of the GPS satellites has been acquired, the navigation message provides sufficient information about the adjacent satellites for the remaining visible satellites to be acquired quickly. The receiver may need to search in Doppler shift because the position of

Doppler shift

+4 kHz

Signal 0

BIF = 1 kHz

−4 kHz 0

1 ms

1 µs C/A code timing

FIGURE 12.7 Code synchronization and Doppler tracking matrix.

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the receiver relative to the satellites is not known, but their CA codes are. The GPS receiver retains the information from the navigation message when switched off, and it also runs its internal clock. When next switched on, the receiver will assume that its position is close to its last known position when it was switched off, calculate which satellites should be visible, and search for those first. This greatly speeds up the acquisition process. If the receiver has been moved a large distance while turned off, a cold start may be needed. The correlation process described above assumes that each satellite is acquired sequentially. Some lower cost GPS receivers use sequential acquisition of the satellites, and also make timing measurements sequentially, one satellite at a time. More sophisticated receivers have parallel correlators which can search for and acquire satellites in parallel. Twelve parallel correlators guarantee that all visible GPS satellites will be acquired, and start-up time is much shorter than with sequential acquisition. Accuracy is also better with parallel processing of the signals. Integrity monitoring of the GPS position measurement is possible by using a fifth satellite to recalculate the receiver position. With five satellite signals there are five possible ways to select four pseudoranges to use in the ranging equations, leading to five calculations of position. If there is disagreement between the results, one bad measurement can be eliminated. If more than one result disagrees with the others, the integrity of the measurements is compromised. GPS receivers used for navigation of aircraft in instrument meteorological conditions (IMC, in the clouds) and for instrument landings are required to have integrity monitoring to guard against receiver or satellite failures and interference with or jamming of GPS signals. The P code for the ith satellite is generated in a similar way to the CA code. The algorithm is Pi 1t2  X11t2  X21t  iTc 2

(12.5)

where Tc is the period of the X1 sequence, which contains 15,345,000 bits and repeats every 1.5 s. The X2 sequence is 37 bits longer. The P code repeats after 266.4 days, but is changed every 7 days for security reasons. The long length of the P code sequence makes the distance measurements unambiguous. P code sequences cannot be acquired easily because they do not repeat, a deliberate feature to prevent unauthorized users from operating high accuracy GPS receivers. The CA code provides information to authorized users on the starting time of the P code; this is contained in the navigation message as an encrypted handover word. If the current feedback tap settings for the P code generators are known, and the handover word is decrypted, the receiver can start the local X Code generators close to the correct point in the P code sequence. This allows rapid acquisition of the P code, and is the origin of the name coarse acquisition for the CA code.

12.6

GPS NAVIGATION MESSAGE A key feature of the GPS CA code is the navigation message. The navigation message contains a large amount of information that is used by GPS receivers to optimize the acquisition of satellite signals and calculate position. The navigation message is sent at 50 bps by BPSK modulation of the CA and P codes. Effectively, 20 CA code sequences form one navigation message bit. The phase of the 20 sequences is inverted between the 1 and 0 bits of the message by modulo-2 addition of the navigation message data to the CA and P code sequences. The navigation signal is extracted by a 50-bps BPSK demodulator that follows the CA or P code correlator. The narrow bandwidth of the navigation message ensures a high SN ratio at the demodulator input and correspondingly low probability of

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12.7 GPS SIGNAL LEVELS

TABLE 12.1 Header Subframe Subframe Subframe Subframe

473

GPS Navigation Message: Subframe Details

1 2 and 3 4 5

Telemetry message: health of satellite, handover word Satellite clock correction data. Age of transmitted data Ephemeris for this satellite Almanac data for satellites 25 and higher. Ionospheric model data Almanac data for satellites 1–24. Health data for satellite 1–24

Note: Subframes 1, 2, and 3 repeat all data every 30 s. Subframes 4 and 5 repeat every 30 s, but transmission of the full data set requires 25 subframes over a period of 12.5 min.

bit errors in the navigation message. Satellites with elevation angles above 10° will typically give a SN ratio of greater than 17 dB at the output of the correlator. The complete navigation message is 1500 bits, sent as a 30-s frame with 5 subframes. However, some information is contained in a sequence of frames, and the complete data set requires 12.5 min for transmission. The most important elements of the message are repeated in every frame. The subframes contain the satellite’s clock time data, orbital ephemeris for the satellite and its neighbors, and various correction factors. Details of the subframes are given in Table 12.1. The calculation of position in a GPS receiver requires very accurate knowledge of the location of the satellite at the time that the measurements of pseudoranges are made. If the pseudorange is measured to an accuracy of 2.4 m, we must know the satellite position to an even greater accuracy, and that requires very accurate calculation of the GPS satellite orbits. By comparison, the orbit of a communication satellite does not need to be known to the same level of accuracy. As described in Chapter 2, the GPS system uses modified WGS-84 data to define the earth’s radius, Kepler’s constant, and the earth’s rotational rate. Data on the speed of EM waves is taken from the International Astronomical Union. The WGS-84 data set also includes a very detailed description of the earth’s gravitational field, which is essential for precise location of the satellites in their orbits. All of these parameters and corrections are stored in every GPS receiver, and used in calculating position.

12.7

GPS SIGNAL LEVELS GPS receiver antennas have low gain because they must be omnidirectional. We will assume a worst-case gain of G  0 dB, corresponding to an isotropic antenna. In practice, G  0 dB in many directions, but may fall to 0 dB in some directions. The omnidirectional antenna picks up radiated noise from the environment, making the antenna temperature close to 273 K. LNA temperatures can be as low as 25 K, so a system noise temperature of 273 K will be used as a typical value. Typical GPS antennas are circularly polarized patches or quadrafilar helices that have carefully shaped patterns that cut off quickly below 10° elevation to minimize noise pick up from the ground. The LNA is mounted directly below or behind the antenna to avoid the increase in noise temperature caused by lossy antenna cables. GPS satellites have an array of helical antennas that provide gain toward the earth, and 10 W transmitters, leading to EIRP values in the range 19 to 27 dBW. The CA code transmitted by the satellite is a direct sequence spread spectrum signal, so the CN ratio in the CA code’s RF bandwidth will be less than 0 dB. This is typical of systems that use direct sequence spread spectrum signals. The low CN ratio of the spread spectrum signal is converted to a usable SN by correlation of the code sequences, which adds a despreading (processing) gain to the CN ratio. The theoretical processing gain of a direct sequence spread spectrum signal is equal to the ratio of the chip rate to the bit rate in the

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spreading sequence, but losses in the correlation process always make practical gains a little lower. For the CA code transmitted at 1.023 Mbps and a 1-ms correlation time, the theoretical processing gain is 1023, or 30.1 dB. The corresponding processing gain for the P code is 40.1 dB. The GPS receiver can pick up signals from up to 10 satellites at the same time. The RF energy from the satellite spread spectrum transmissions adds to the noise in the receiver as an interference term, I. For simplicity, in the following analysis we will assume that there are 10 GPS satellites visible, that there are 9 interfering satellites generating random signals (noise) out of which the receiver must extract the 10th signal, and that all the received signals are of equal strength. The signals from interfering satellites are treated as random noise because the Gold codes that they transmit have very low cross-correlation with the code from the wanted satellite. Noise has zero cross-correlation with the wanted signal, and the Gold codes used by GPS satellites are chosen because they closely approximate noise. Nine interfering GPS satellites represents a worst case; in practice the number of visible satellites varies between four and ten, and the signal strengths also vary depending on the elevation angle of the satellite and the antenna pattern at the receiver. The worst case is actually when a weak signal from a satellite at a low elevation angle must be extracted from stronger signals from satellites at higher elevation angles. GPS receivers automatically select the strongest signals for processing so that the worst case can be avoided, but if the sky is partially blocked by obstructions, a weak signal may have to be used. Table 12.2 shows the downlink signal power budget for the L1 and L2 carriers. A receiving antenna gain of 0 dB is assumed. The interference from nine CA code spread spectrum signals of equal power is given by the sum of the received power (in watts) from each satellite I  9  1016 W 1 150.5 dBW The thermal noise power, N, in a noise bandwidth of 2 MHz for a noise temperature of 273 K is kTBn watts, where N  141.2 dBW 1 7.59  1015 W The noise and interference powers must be added in watts, not in decibels: N  I  8.49  1015 W 1 140.7 dBW Hence the worst case CN for one CA code signal in this scenario is C 1N  I2  160.0  1140.72  19.3 dB

(12.6)

Similar analysis yields the C(N  I) values for the two P code signals, as shown in Table 12.3. Note that thermal noise is the major factor in setting C(N  I), since in TABLE 12.2 Link Budget for L1 and L2 Carriers Carrier L1 (1575 MHz) Code

Carrier L2 (1227 MHz)

CA code

P code

P code

EIRP (dBW) Path loss (dB)

26.8 186.8

23.8 186.8

19.7 185.7

Receive antenna gain (dB) Pr (dBW)

0 160.0

0 163.0

0 166.0

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TABLE 12.3 CN, Noise, and Interference Budget for L1 and L2 Carriers Carrier L1 (1575 MHz) Code Ts (dBK) Bn (dBHz) N (dBW) I (dBW) N  I (dBW) Pr (dB) C(N  I) (dB) Gproc (dB) SN (dB)

Carrier L2 (1227 MHz)

CA code

P code

P code

24.4 63.0 141.2 150.5 140.7 160.0 19.3 30.1 10.7

24.4 73.0 131.2 153.0 131.1 160.3 31.9 40.1 8.2

24.4 73.0 131.2 (thermal) 156.0 (nine satellites) 131.2 166.0 34.8 40.1 5.3

the worst case of interference caused by nine visible satellites, all received at maximum power, the interference power level is 9.3 dB below the thermal noise power. The CN ratio at the receiver is 0.7 dB lower when the interference from the nine visible satellites in included. A more realistic scenario would have four satellites at the maximum receive power level and the remainder at a lower level, since GPS satellites orbit in constellations of four, with one constellation always visible, to improve the accuracy of position location measurements. Thus we would expect less than 0.7 dB degradation in the CN ratio due to interference by other satellites’ CDMA signals for almost all of the time. The SN at the correlator output is 10.7 dB for the CA code and 8.2 dB for the L1 P code, using the CN values in Table 12.3 and the theoretical processing gains for each code with no losses in the correlation and filtering of the signals. Historically, the earlier generations of GPS satellites have had transmitter EIRPs up to 3 dB higher than indicated by Table 12.3. Receiving antennas with gain greater than 0 dB also help to increase the CN ratio, so CA code SN ratio can be up to 6 dB higher than the specification value of 10.7 dB for the CA code. The navigation message has a 50-bps bit rate, and each bit extends over 20 CA code correlation periods. The CA code correlator output is passed through a 50 Hz bandwidth filter which integrates the 20 pulses from the correlator to give a single message bit, in the form of a 50 bps BPSK signal. The SN ratio of the BPSK message signal will theoretically be 13 dB higher than the SN at the correlator output, at 23.3 dB. However, the correlation and filtering processes are not perfect and an implementation margin of several dB must be allowed. Nevertheless, the SN ratio of the BPSK signal will be above 20 dB in most cases, guaranteeing error free detection of the navigation message.

12.8

TIMING ACCURACY The position location process requires an accurate measurement of the time of arrival of the code sequence at the receiver. The output of the CA code correlator is a 1 s wide pulse that repeats every millisecond. The accuracy with which a timing measurement can be made on a single pulse is given by the approximate relationship6 dt 

1 3Bn 1S N4

seconds

(12.7)

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where t is the rms timing error, Bn is the noise bandwidth of the RF channel, and SN ratio is the signal-to-noise power ratio (not in dB) for the pulse in the noise bandwidth Bn. The SN ratio after the correlator is SN  CN  Gp  losses

(12.8)

where Gp is the correlator processing gain. For the CA code Gp  1023  30.1 dB and SN  19.3  30.1 dB  losses  11.7 dB  losses If we assume the specification value for SN of 11.7 dB and losses of 1.7 dB, SN  10 dB, a power ratio of 10. The theoretical noise bandwidth of the correlator is Bn  1 MHz (IF noise bandwidth) so dt  1 3106 1104 s  0.316 ms

(12.9)

A typical GPS receiver will update the display no more than twice a second, so the pulses from the correlator can be averaged over a period of half a second, which will decrease the rms error by 1500  22.4 to an rms value of 14 ns, assuming randomly distributed errors. The 14 ns rms timing error translates to an rms distance error of 4.2 m. However, four distance measurements are needed to obtain a position measurement, so with no other errors accounted for, the basic position measurement accuracy of the CA code receiver is about 8.4 m 14.2  142 measured as an rms value. A higher CN ratio in the receiver will improve the accuracy, but other errors, discussed later in Section 12.10, will lower the accuracy. The accuracy achieved by commercial CA code GPS receivers was better than expected by the designers of the GPS system. Military strategists became concerned that CA code GPS receivers could be used to target weapons against the United States with considerable accuracy. The U.S. Department of Defense (DOD) introduced selective availability (SA), a scheme to deliberately degrade the accuracy of CA code receivers by varying some of the parameters of the GPS satellites. Selective availability was switched off on May 1, 2000, and will be turned on only if the security of the United States is threatened.

12.9

GPS RECEIVER OPERATION A CA code GPS receiver must be able to correlate signals from at least four satellites, calculate time delays, read the navigation message, calculate the orbits of the GPS satellites, and calculate position from pseudoranges. The key to accurate position determination is accuracy in the timing of the arrival of the Gold code sequences from each satellite in view. All GPS receivers use a microprocessor to make the required calculations and to control the display of data. There are many different ways that this can be done, depending on the application for which the receiver is intended. The tasks of the microprocessor will not be considered here—it is assumed that once accurate timing data is available and the navigation message read that the microprocessor can complete its required tasks. Most CA code GPS receivers use an IC chip set that contains 12 parallel correlators. This allows the receiver to process signals from up to 12 satellites at the same time, which helps keep all the signals synchronized. Some simpler receivers use a single

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correlator and process four satellite signals sequentially, with consequent lower accuracy. The received GPS signals are converted to a suitable IF frequency in the front end of the receiver, and then processed to recover the CA codes. In more recent GPS receivers, much or all of the signal processing is done digitally using DSP techniques. The explanation of the signal processing techniques used in GPS receivers that follows is based on block diagrams that can be implemented in analog or digital form. The blocks presented in this discussion are those that would be found in an analog receiver. Most GPS receivers implement them using digital signal processing techniques (DSP). We will start the analysis by considering the signal received from the satellite at the output of the IF stage of the receiver. The IF signal in the GPS receiver will consist of the sum of a number (up to 12) of signals from visible GPS satellites. The IF carrier signal has several BPSK modulations applied to it by the satellite, and when received on earth has been Doppler shifted by satellite and earth motion. The IF signal from N GPS satellites in view is s1t2  a 5AiCi 1t2Di 1t2 sin 3 1vi  vd 2t  fi 1li 2  fi 4 6 N

(12.10)

i1

where Ai is the amplitude of the received signal. Ci(t) is the Gold code modulation Di(t) is the navigation message modulation i is the IF frequency of the received carrier d is the Doppler shift of the received signal i(li) is the phase shift along the path i is the phase angle of the transmitted signal The key to successful measurements in a GPS CA code receiver is to generate a signal in the receiver that is identical to the signal received from satellite i, but without the navigation data that is modulated onto the transmitted signal. When the correct signal is generated in the receiver it has the correct CA code for satellite i, the code has the correct starting delay, and the correct Doppler shift has been applied. The locally generated signal is multiplied by the received signal, which contains several other signals from visible GPS satellites, and the output is integrated over the CA code length of 1 ms. The result is a constant output for a period of 20 ms, corresponding to the duration of a navigation data bit. The precise matching of the locally generated signals to the received signals from four visible GPS satellites ensures that the local receiver’s chip clocks and CA code generators are exactly in sync with the received signals. When this condition is achieved, the start time of each CA code sequence and the corresponding chip clock transition provide the high accuracy time marker that makes GPS time delay measurement possible. The receiver must measure i(li) in Eq. (12.10) as a time delay in order to obtain the pseudorange for each of the N satellites in view, and it must recover the Ci(t) modulation by correlation. The Di(t) modulation contains the navigation message as a 50 bps BPSK modulation of the Ci(t) signal. Both the C(t) and D(t) signals are modulated onto the carrier of the satellite signal by binary phase shift keying (see Chapter 5) and therefore have values 1. Demodulation of BPSK signals requires a locally generated carrier which is locked to the phase of the received carrier, and recovery of the data signal requires a bit clock that is locked to the bit rate of the received signal. The wanted signal is buried below the receiver noise and CDMA interference. We must multiply the signal and noise by the wanted CA code sequence to despread the

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signal and to bring it above the noise. The nominal bandwidth of the signal is 1 kHz after the correlator, since the 1023 bit sequence of the CA code repeats every millisecond. However, the IF carrier can be shifted in frequency by up to 4 kHz because of Doppler effects. The receiver must therefore first search in Doppler frequency space— eight 1 kHz frequency offset steps—until a signal is found. This is done as part of the signal acquisition process by incrementing the frequency of the locally generated carrier in 1 kHz steps. Part of a typical receiver structure for the GPS CA code is shown in Figure 12.8. The function of the non-coherent delay lock loop is to set the frequency of the voltage controled oscillator (VCO) in the receiver to match CA code rate of the received signal, and to align the received chip transitions correctly. GPS satellites generate all their signals from a master clock, which means that there is phase coherence between the chips, the codes and the RF frequencies of all GPS signals from a particular satellite. The delay lock loop shown in Figure 12.8 takes advantage of the coherent nature of the GPS CA signals, so that the VCO becomes both a time reference for the CA code signals and also the chip clock. The PN code generator in Figure 12.8 must be set to the correct code, and its start time must also be set correctly, for the loop to lock. When the IF CA code in the receiver is correctly generated and has the correct frequency and timing, it will exactly match the received CA code at the input to the delay lock loop. The delay lock loop has three paths: punctual, early (half chip ahead), late (half chip behind). The delay lock loop steers the chip clock so that the punctual output can be used to drive the CA code generator. The CA code chip rate is generated by the VCO. The incremental process of trial and error which eventually finds the correct sequence and timing was described above. The early-late channels in the delay lock loop generate output

Punctual BPF

C(t − τ )

Output D i sin(ω IFt )

B = 50 Hz BPF

Envelope detector

Early + Σ

C(t − τ − Tc /2) BPF

Input D i C(t − τ ) sin(ω IFt + φ )

Envelope detector

− Late

C(t − τ + Tc /2) PN code generator

VCO (chip clock)

Code select

Loop filter

B = 1 kHz

FIGURE 12.8 Noncoherent code lock loop and navigation message recovery. VCO, voltage controlled oscillator.

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signals which steer the phase of the VCO so that the navigation message is recovered correctly. The locally generated carrier that is used to demodulate the C(t) signal must be Doppler shifted to match the Doppler offset of the received signal, and modulated with the correct CA code sequence, starting at the correct time. The correct Doppler shift, code sequence, and start time are all unknown when the receiver is first switched on. The signal is buried below the noise, so it is not possible to determine the correct parameters by direct analysis of the received signal. The receiver must therefore be designed to search all possible Doppler shifts, code sequences, and code start times until an output is obtained from the correlator indicating that a satellite signal has been found. Once one GPS satellite signal has been found, information contained in the navigation message can be used to steer the receiver to the parameters needed to acquire the other visible satellites. If the receiver is turned off and then turned on again, the microprocessor memory has the last known satellite configuration stored, and can derive expected signal parameters by allowing for the time for which the receiver was off. The output of the CA code correlator with Doppler corrected IF frequency for the satellite signal with code number M is x1t2  Am R 1tm  t2Dm 1t2 sin3vm 1t2  fm 1lm 2  fm 4  n1t2

(12.11)

y1t2  AmR 1tm  t2 Dm 1t2 sin 3vm 1t2  fm 1lm 2  fm 4  n1t2

(12.12)

y¿1t2  R 1tm  t2 Dm 1t2 sin3vm 1t2  f¿m 4  n1t2

(12.13)

where R(m  ) is the autocorrelation function of the wanted code number M, and n(t) is the output from cross-correlation with all other codes. The time shift (tm  ) to the correlation peak is the wanted measurement that provides the pseudorange to the satellite. The output of the correlator is a despread signal at baseband, which is modulated with the 50 bps navigation message. With the CA code removed by the correlation process, it is a straightforward process to demodulate the navigation message D. Passing this signal through a narrow bandwidth bandpass filter improves the SN ratio and ensures that the message is recovered without errors. The IF carrier is recovered with a special type of phased locked loop (PLL) called a Costas loop. A Costas loop compensates for the arbitrary phase of the received signal. The despread IF carrier is BPSK modulated by the navigation message Dm(t) The IF carrier signal is limited to remove any amplitude variations, which sets Am  1. Then

The navigation message D(t) is recovered by multiplying the IF signal y (t) by sin[m(t)   m] and low pass filtering to obtain the 50 bps signal. The reference carrier for the BPSK demodulator can be derived from the output of the Costas loop. The demodulated message signal is z(t) where z 1t2  R 1tm  t2Dm 1t2  n¿1t2

(12.14)

Provided that the correlation peak of z(t) crosses the threshold and n (t) doesn’t, we can recover the data message Dm(t) correctly. If everything works correctly in the receiver, the SN of the signal y (t) is at least 17 dB, so there will be no bit errors. Even if a bit error occurs in the navigation message, it is removed when the next message is received about 30 s later. Figure 12.9 shows a Costas loop which is often used as the demodulator for low speed BPSK signals such as the 50 bps GPS navigation message. The loop has an I channel and a Q channel driven by a VCO. The VCO frequency is set by the sum of the outputs

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LPF

1 2

D(t ) cos φ

Navigation message

B = 50 Hz Input D i sin(ω IFt + φ )

sin(ω IFt + φ ) VCO ω IF

π /2

Loop filter 1 8

sin(2 φ )

cos(ω IFt + φ ) LPF

1 2 D(t )

sin φ

FIGURE 12.9 Costas loop. LPF, low pass filter.

from the I and Q channel detectors, which steers the VCO phase such that the I channel is in phase with the signal. The I channel output is then (ideally) a zero ISI waveform which can be integrated and sampled to recover the navigation message bits.

12.10

GPS C/A CODE ACCURACY

The major sources of error in a GPS receiver that calculates its position are: Satellite clock and ephemeris errors Selective availability (when switched on) Ionospheric delay (advance) Tropospheric delay Receiver noise Multipath The accuracy that can be achieved with a GPS CA code receiver can be found by using a range error budget. The figures in square brackets [ ] are for the case when selective availability (SA) is turned off. Typical values of range error are given in Table 12.4. All values are in meters (m). Note that a value of 2.4 m error is assigned to receiver noise. The value calculated in Section 12.8 was 4.2 m, for a worst-case received signal strength. The range error introduced by the ionosphere and the troposphere can be partially removed by receiving identical signals at two different carrier frequencies. This technique is used by high precision P code receivers. The P code signal is transmitted on the L1 carrier at 1575.42 MHz, in phase quadrature with the CA code signal. The P code is also transmitted on the L2 carrier at 1227.60 MHz. Algorithms are used in the P code receiver to calculate the net delay of the signal caused by the ionosphere and the atmosphere, and to then remove the errors from the calculated ranges. CA code receivers use a standard atmosphere and ionosphere and assume a constant delay at a given elevation angle. Variations in the density of the atmosphere with atmospheric pressure changes, and in the free

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TABLE 12.4 Range Error for C/A Code Measurements (m) Satellite clock error Ephemeris errors Selective availability Ionospheric delay Tropospheric delay Receiver noise Multipath RMS range error [RMS range error

3.5 4.3 32.0 6.4 2.0 2.4 3.0 33.4 m with SA 9.5 m without SA]

[0]

Brackets indicate SA off.

electron content of the ionosphere, lead to departure from the standard values and hence to errors in the pseudorange calculation. There are plans to transmit the CA code at a third and a fourth L-band frequency from later GPS satellites to provide improved accuracy with CA code receivers8. The range error shown in Table 12.4 is for one satellite–earth path, for the pseudorange that is calculated from the timing measurements using the receiver clock. However, four pseudorange measurements are needed to make a position determination. Thus the position location output of the GPS receiver combines four path errors, which are not necessarily equal because of the geometry of the satellites in the sky and the different signal strengths at the receiver input. Receiver position is calculated in (x, y, z) coordinates, and the errors in x, y, and z depend on the elevation angle of satellites, the satellite geometry, and the other parameters in the error budget. The calculated position will have different levels of error in the x, y, and z directions. To account for these differences several dilution of precision factors (DOP) are defined. A DOP factor multiplies the basic position measurement error to give a larger error caused by the particular DOP effect.

Dilution of Precision: HDOP, VDOP, and GDOP Horizontal dilution of precision is one of the most important DOP factors for most GPS users. It provides an error metric for the x and y directions, in the horizontal plane. A typical HDOP value is 1.5, and it is often the smallest of the DOPs. Horizontal measurement error for a CA code receiver is typically 14.3 m with SA off (1DRMS) and 50 m with SA on (1DRMS). GPS practice uses 2DRMS as the quantifier for accuracy in position determination giving a 2DRMS accuracy of 28.6 m with SA off. The 2DRMS accuracy figure means that 95% of all measurements yield a position within 28.6 m of the true location of the GPS receiver, in this example. There are many DOP factors in GPS. The more important ones are horizontal dilution of precision, HDOP, vertical dilution of precision, VDOP, and geometric dilution of precision, GDOP. Other DOPs include position dilution of precision, PDOP, and time dilution of precision, TDOP. In general, VDOP and GDOP are most likely to degrade the accuracy of GPS position measurements. VDOP accounts for loss of accuracy in the vertical direction caused by the angles at which the satellites being used for the position measurement are seen in the sky. If the satellites are all close to the horizon, the angles

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between the satellites and the receiver are all similar and VDOP can be large. In the worst possible case, if all the satellites were at the horizon, it would be impossible to make an accurate measurement in the vertical direction. A change in range to at least one satellite must occur when the receiver is moved, otherwise the receiver cannot detect that change. If all the satellites are at the horizon, no range change occurs for vertical movement of the receiver and consequently vertical accuracy is very poor. Similarly, if all the satellites were clustered directly overhead, HDOP would be large. VDOP is important in aircraft position measurements, where height above the ground is a critical factor, especially when landing. CA code receivers suffer from significant VDOP and cannot provide sufficient vertical accuracy for automated landing of aircraft. CA code GPS receivers cannot guarantee sufficient vertical accuracy unless operated in a differential GPS mode. The GPS satellites are configured in orbit to minimize the probability that a DOP can become large, by arranging the orbits to provide clusters of four satellites with suitable spacings in the sky. However, if the receiver’s view of the sky is restricted, for example, by buildings, the geometry for the position calculation may not be ideal and GDOP can become large. This causes all the other DOP values to increase. Aircraft, and ships at sea always have a clear view of the sky, but automobiles often do not. CA code receivers may revert to two-dimensional measurements (x and y) using three satellites when the sky is obstructed.

12.11

DIFFERENTIAL GPS

The accuracy of GPS measurements can be increased considerably by using differential GPS (DGPS) techniques. There are several forms of DGPS, all of which are intended to increase the accuracy of a basic GPS position measurement, and to remove the effects of selective availability. A second, fixed, GPS receiver at a reference station is always required in a differential GPS system. In the simplest forms of DGPS, a second GPS receiver at a known position continuously calculates its position using the GPS CA code. The calculated (x, y, z) location is compared to the known location of the station and the differences in x, y, and z are sent by a radio telemetry link to the first GPS receiver. The accuracy of the CA code position measurement can be increased from 100 m to about 10 m, with SA in effect, but this technique works well only if the two stations are close together and use the same four satellites for the position calculation. In a more sophisticated form of differential GPS, the monitoring station at a known location measures the error in pseudorange to each satellite that is visible at its location, and telemeters the error values to users in that area. This allows other GPS users to select which satellites they want to observe, and extends the area over which the DGPS system can operate. The accuracy of a CA code measurement can be increased to 5 m for receivers within 10 km of the reference station and to 10 m for receivers within 500 km of the reference station. The most accurate forms of differential GPS use the relative phase of the many signals in the GPS transmissions to increase the accuracy of the timing measurements. Suppose that you could count the number of cycles of the 1575 MHz L1 carrier wave between a satellite and a GPS receiver, and that the GPS satellites are stationary for the length of time it takes to make the count at two separate locations. The wavelength of the L1 carrier is 0.19043 m, so movement of the receiver by 0.01 m directly away from the satellite would change the phase angle of the received wave by 18.9°. If the total number of cycles between the satellite and the receiver is known, and fractional cycles are measured

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with a phase resolution of 20°, the true distance to the satellite can be found to 0.01 m accuracy. In principle, measurements which compare the phase angle of the received L1 carriers from several GPS satellites could therefore be used to detect receiver movements at the centimeter level. This is called differential phase or kinematic DGPS. The obvious difficulty is that we cannot count the number of cycles of the L1 carrier between the satellite and the receiver. However, we can make phase measurements and time of arrival comparisons for various GPS signals at two different locations and resolve motion between the two locations. If one of the receivers is a fixed reference station, it is then possible to locate the second GPS receiver very accurately with respect to that fixed location. This technique is valuable in land surveying, for example, where a reference station can be set up at a known location, such as the corner of a plot of land, and the position of the plot boundary relative to that point can be measured. The same technique can be used to find the position of an aircraft relative to an airport runway so that a precision approach path can be established. The difficulty with DGPS phase comparison measurements is that the L1 carrier has cycles which repeat every 0.19043 m, and one cycle is identical to the next. This creates range ambiguity which must be resolved by reference to the wavelengths of other signals. The 10.23 MHz P code transmission of the L1 carrier has a P code chip length in space of 29.326 m, which is 154 cycles of the L1 carrier. The ambiguity of the carrier waveform can be resolved within the 29.326 m length of a P code chip by comparison of the time of arrival of a particular cycle of the L1 carrier with the time since the start of the P code chip. Similar ambiguity resolution for the 29-m P code chips is possible using the length of the CA code chip and the CA code sequence. The length of a CA code chip at 1.023 MHz is 293.255 m, and the length of a CA code sequence is 293.255 km. When ambiguity resolution is applied using all of these waveforms, very small movements of the receiver can be detected and ambiguity out to 293 km can be removed. Aircraft flight paths have been tracked to an accuracy of 2 cm over distances of tens of kilometers using phase comparison DGPS techniques. This explanation of kinematic differential GPS is oversimplified, because the satellites are moving and measurements over a considerable time are required to resolve ambiguity to the centimeter level. The P code can be used for real-time differential measurements without knowledge of P code itself, because only a comparison of the time of arrival of the code bits is required. Selective availability is not applied to the P code, so differential measurements made with the P code cannot be affected by SA. In the Wide Area Augmentation System (WAAS) developed by the FAA for aircraft flying in North America, 24 WAAS receive stations continuously monitor their position as calculated from the CA codes of all visible satellites in the GPS system. The stations also use the P code transmissions to make accurate differential measurements of the pseudorange to each visible satellite. The actual position of the WAAS stations is known very accurately from prior survey data, so each WAAS station can calculate the error in the pseudorange to each visible satellite. The 24 WAAS stations send their data to a central station with an uplink to a GEO satellite. The central station validates the data, combines all the information, and sends a sequence of pseudorange correction data to all GPS users via the satellite. The central station also determines whether any of the data is in error, and sends a warning signal called an integrity message to instruct aircraft not to use the GPS system, or a particular satellite, because the data are not reliable. This is an essential part of the FAA strategy for using GPS as the primary means of aircraft navigation. If the aircraft is relying on GPS information alone to determine its position, that information must have a very high reliability.

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The WAAS GEO satellite transmits signals which are in a similar format to the L1 signals transmitted by a GPS satellite. A conventional GPS receiver with suitable software can extract the pseudorange error values from the WAAS satellite transmission and obtain markedly improved accuracy in its position determination. Thus no hardware changes are needed to convert a GPS receiver to use WAAS data. The GEO satellite can also be used to augment GPS satellites for position measurements, since it radiates the same signal format. The calculation of pseudorange error from the P code sequence, rather than (x, y, z) position data error from the CA code, significantly increases the accuracy of the WAAS DGPS system. Eventually, it seems probable that local area augmentation systems (LAAS) using differential GPS will be established at many airports to replace or augment existing ILS precision approach systems. Advanced LAAS DGPS systems have been demonstrated to achieve better than 1-m accuracy in three dimensions, with update rates sufficiently fast to control a passenger aircraft. This is sufficient to allow DGPS position data to be coupled to the aircraft autopilot so that blind landings can be made automatically in zero visibility conditions. Several demonstrations of autoland using DGPS were made in the late 1990s using Boeing 737 and 757 aircraft. Aircraft used by overnight delivery companies will likely be fitted with GPS blind landing systems first, since cargo aircraft are subject to fewer restrictions than passenger aircraft and overnight delivery is subject to delays when airports are closed by low visibility weather. Typically, a good autoland system fitted to a large aircraft can achieve more consistent landings than a skilled pilot, so autoland may eventually become as common for landings as autopilot use is for en route operation. Weather may eventually be less of a factor in causing delays to passenger aircraft arrivals and departures.

12.12

SUMMARY

The Global Positioning System has revolutionized navigation and become a consumer product with many applications. A system that was originally conceived for targeting nuclear weapons has become a worldwide commodity. Eventually GPS receivers will be present in all aircraft, ships, and automobiles so that navigation will become a simple task, and no one with a GPS receiver (and the ability to read a map) will ever be lost. GPS receivers will be embedded in all cellular telephones so that the user will know his or her location, and an emergency call will always contain location information. GPS may eventually replace all existing aircraft and ship radio navigation systems and become the sole means of navigation, permitting aircraft to select their own routes between airports and to be landed automatically in zero visibility conditions under the control of a local area differential GPS system. The principles of GPS position location are simple, but its execution with the required accuracy is complex. Position location with GPS is based on trilateration; measurement of the distances from an unknown point to three known points (GPS satellites)

yields a unique solution for the position of the unknown point. The apparent distance between each satellite and the receiver is determined by measuring the time-offlight of a signal transmitted by the satellite, using real-time clocks on the satellite and at the receiver. Electromagnetic waves travel at 3  108 m/s, or 300 m per microsecond. To achieve position location accuracy of a few meters, the time-of-flight of GPS signals must be determined to one-tenth of a microsecond or better accuracy. GPS satellites have four atomic clocks with extreme accuracy, but most commercial GPS receivers use low-cost quartz crystal controlled clocks that are not accurate to a microsecond. The signals from a fourth GPS satellite are used to calculate the offset error in the clock of the GPS receiver, which allows the receiver to track time with an accuracy better than one-tenth of a microsecond. Thus four GPS satellites are needed to make an accurate threedimensional position measurement. The distances between the GPS satellites and the receiver are known as pseudoranges, because they may be in error by

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PROBLEMS

thousands of kilometers. The receiver clock error data provided by the fourth psuedorange measurement allows correction of pseudoranges to true ranges. The satellites all transmit navigation messages that contain the ephemeris for each GPS satellite; these data are used by the GPS receiver to calculate satellite orbital positions. GPS satellites use direct sequence spread spectrum techniques, transmitting two pseudorandom sequences known as CA and P codes at two frequencies. Each satellite is assigned a unique set of codes which are used by the receiver to make timeof-flight measurements and also to identify the satellites. Commercial GPS position measurement receivers use the course acquisition (CA) code transmitted at a chip rate of 1.023 Mbps by GPS satellites on the L1 carrier at a frequency of 1575.42 MHz. The CA code is never changed. Authorized users, primarily military, use the P code which has a chip rate of 10.23 Mbps to provide greater positioning accuracy. The P code is encrypted and changes once a week. P code signals are transmitted on the L1 carrier, in phase quadrature with the CA code signal,

485

and also on the L2 carrier at a frequency of 1227.60 MHz. Receivers that can use both L1 and L2 transmissions are able to remove some timing errors introduced by atmospheric and ionospheric delays. The accuracy achievable with a CA code GPS receiver is typically 30 m, but can be degraded to 200 m by selective availability (SA), a DOD applied effect that deliberately reduces the accuracy of the CA code position measurements. SA was introduced as a security measure to prevent CA code receivers from achieving very high accuracy, but was removed on May 1, 2000. Differential GPS makes use of a fixed GPS receiver in a known location to calculate errors in the GPS position measurement, or errors in the pseudoranges to all visible satellites, which are then sent over a radio link to other GPS receivers. The errors do not vary greatly over a wide area, allowing CA code GPS receivers to achieve far greater accuracy than is possible with a single receiver. Differential GPS is the basis of the U.S. FAA’s Wide Area Augmentation System (WAAS) that will improve the accuracy of CA code receivers in aircraft to 10 m.

REFERENCES 1. B. W. PARKINSON, and J. J. SPILKER, The Global Positioning System-Theory and Applications, American Institute of Aeronautics and Astronautics, Washington, DC, 1996. 2. E. D. KAPLAN, Understanding GPS Principles and Applications, Artech House, Norwood, MA, 1997. 3. B. H. HOFFMAN-WELLWENHOF, H. LICHTENEGGER and J. COLLINS, GPS Theory and Practice, Springer Verlag, New York, NY, 1992.

4. G. STRANG, and K. BORRE, Linear Algebra, Geodesy, and GPS, Wellesley-Cambridge Press, Wellesley, MA, 1997. 5. Aviation Week, Aerospace Source Book, p. 162, January 17, 2000. 6. M. I. SKOLNIK, Introduction to Radar Systems, 2nd Ed., McGraw-Hill, New York, NY, 1981. 7. B. CLARKE, Aviation Application of GPS, McGraw-Hill, New York, NY, 1996. 8. Aviation Week, January 28, 2002.

PROBLEMS 1. Find the exact altitude of a GPS satellite that has an orbital period equal to precisely one half of a sidereal day. Use a value of mean earth radius re  6378.14 km and a sidereal day length of 23 h 56 min 4.1 s. 2. Find the maximum Doppler shift of the L1 signal frequency for a GPS satellite at an altitude of 20,200 km when the satellite has an elevation angle of 10°. Hint: Maximum Doppler shift occurs when the observer is in the plane of the satellite orbit. Find the velocity of the satellite and the component of velocity toward the observer.

3. An observer at the geographical North Pole has a GPS receiver. At an instant in time, four GPS satellites all have the same range from the observer, and the GPS receiver records a measured delay time for the CA signal of 0.17097528 s for each satellite. The four satellites’ coordinates are calculated to be (0, 13280.5, 23002.5), (0, 13280.5, 23002.5), (13280.5, 0, 23002.5), (13280.5, 0, 23002.5), where all distances are in km. Assuming an earth radius of 6378.0 km at the North Pole, so that the observer’s coordinates are (0, 0, 6378), determine the clock offset error in the GPS receiver. [Use Eqs. (12.1) and (12.3), and take the velocity of light in free space to be 2.99792458  108 m/s.]

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4. Accurate position location using GPS requires precise knowledge of the speed of light. In most applications, we use a velocity of light of 3.0  108 m/s. Solve Problem 3 above and then recalculate the clock offset using c  3  10 m/s instead of the more precise value given in Problem 3. What is the error in the clock offset? What is the difference in the ranges to the satellites when the approximate value for c  3  10 m/s is used? Discuss the corresponding position error due to the approximation. Why is it essential to use the exact value of the velocity of EM waves? 5. A CA code GPS receiver is located at the geographic South Pole, coordinates (0, 0, zp). Four GPS satellites are used to determine the radius of the earth at the South Pole. At the instant of time that the measurement is made, the satellites have the following coordinates: 1, (0, 13280.500, 23002.500); 2, (0, 13280.500, 23002.500); 3, (13280.500, 0, 23002.500); 4, (0, 0, 26561.000).

The corresponding measured delay times for the CA code sequences from the satellites are: 1, 0.12102731 s; 2, 0.12102731 s; 3, 0.12102731 s; 4, 0.11738995 s. Find the clock offset in the GPS receiver, and determine the radius of the earth at the South Pole. Use a value for the velocity of light in free space c  2.99792458  108 m/s, and work your solution to a precision of 1 m. You will need to solve two simultaneous nonlinear equations from the set in Eq. (12.3) in which the unknowns are the clock offset and the value of zp. Start with an estimated value zp  6378 km, and then solve the two simultaneous equations. This will give two unequal values for the clock offset. Use iteration of the value of zp to find the correct values for clock offset and earth radius at the South Pole.

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APPENDIX A

DECIBELS IN COMMUNICATIONS ENGINEERING Most readers of this book will be familiar with the practice of expressing power ratios in decibels, abbreviated dB. The dB ratio A of two power levels p1 and p2 is given by A  10 log10 a

p1 b dB p2

(A.1)

provided that p1 and p2 are expressed in the same units. Although the decibel is formally defined only for a power ratio, p1 and p2 and A can also be expressed in terms of many combinations of voltage, current, resistance, electric field strength, magnetic field strength, and so on. It is common practice in communications engineering to use decibels and the mathematical properties of the logarithm to transform multiplicative equations to additive equations, to manipulate the additive equations into particularly convenient forms, and to define new logarithmic units with dB in their names for some of the quantities that appear. When first presented, this practice is confusing to many people, and we hope to clarify it here. Consider the simple voltage divider circuit with resistors RS and RL shown in Figure A.1. The rms voltages across the source and the load resistance are VS and VL, respectively; the rms power supplied by the source is pS W and the rms power delivered to the load is pL W. From elementary circuit theory, these quantities are related by V 2S RS  RL V 2L pL  RL VS RL VL  RL  RS pLRL pL  RL  RS pS 

(A.2) (A.3) (A.4) (A.5)

letting RL  RS  RT

(A.6)

then pL 

pS RL RT

(A.7)

Equation (A.7) is a multiplicative equation, and pS and pL must have the same units. Expressed another way, whatever units we substitute in for pS, pL will have the same units. The reader should keep this in mind for what comes next. 487

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APPENDIX A

RS

+

+

VS

RL

VL −

−

FIGURE A.1 Voltage divider circuit used as illustration in Appendix A.

Solving for the ratio pLpS and expressing the result in decibels, we have 10 log10 a

pL RL b  10 log10 a b pS RT

(A.8)

Invoking the properties of logarithms we may rewrite this as 10 log10 1 pL 2  10 log10 1 pS 2  10 log10 a

RL b RT

(A.9)

Without affecting the correctness of this equation or of its predecessor we may divide pL and pS by 1 W. Expressed in the form of Eq. (A.9), the result is 10 log10 a

pS pL RL b  10 log10 a b  10 log10 a b 1W 1W RT

(A.10)

The first term above is the decibel ratio of pL to 1 W. This is defined as pL expressed in units of decibels above 1 W, or pL in dBW. If we represent this quantity as PL then PL 1dBW2  10 log10 a

pL b 1W

(A.11)

pS b 1W

(A.12)

Likewise the source power in dBW is PS where PS 1dBW2  10 log10 a Substituting PL and PS into Eq. (A.10) yields PL 1dBW2  PS 1dBW2  10 log10 a

RL b RT

(A.13)

If we had expressed the powers in Eq. (A.9) in milliwatts and then divided both of them by 1 mW, the only effect would have been to express PL and PS in decibels above 1 mW or dBm PL 1dBm2  10 log10 a

pL b 1 mW pS b PS 1dBm2  10 log10 a 1 mW

(A.14) (A.15)

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and Eq. (A.9) would have become PS 1dBm2  PL 1dBm2  10 log10 a

RL b RT

(A.16)

Equations (A.13) and (A.16) differ only in the logarithmic power units that appear on their left-hand sides. These equations would be true so long as PL and PS were both expressed in the same logarithmic units. This happens because units that cancel by division in multiplicative equations like Eq. (A.7) cancel by addition or subtraction in additive decibel equations like Eqs. (A.13) and (A.16). We can use this to write a general form for both these equations: PS  PL  10 log10 a

RL b dB RT

(A.17)

The “dB” after the equation means that the quantities involved must be expressed in a consistent set of logarithmic units. Equation (A.17) is typical of many of the equations used in this text that contain a mixture of decibel and nondecibel units. We can change it to one involving only decibel quantities if we divide both the resistances by 1 ohm () and transform the ratio on the right-hand side to a difference PS  PL  10 log10 a

RL RT b  10 log10 a b 1 1

(A.18)

Now let us define a new unit for our own use called the dB for decibels above 1 ohm. This is a dubious but expedient use of the term decibel! Thus RL 1in dB2  10 log10 a

RL b 1

(A.19)

We can also express RT in dB by taking its log. Thus in these new units Eq. (A.16) becomes PL 1dBm2  PS 1dBm2  RL 1dB2  RT 1dB2

(A.20)

Likewise we could have expressed the resistance in kilohms (k) and then divided all of the resistance terms by 1 k, inventing a new unit that we will call the dBk. The result could have been either or

PL 1dBm2  PS 1dBm2  RL 1dBk2  RT 1dBk2

(A.21)

PL 1dBW2  PS 1dBW2  RL 1dBk2  RT 1dBk2

(A.22)

The units in these two equations cancel by addition and subtraction rather than by multiplication and division. Hence so long as both powers are in the one common decibel unit and so long as both resistances are in another common decibel unit, we may write a general form of these equations PL  PS  RL  RT dB

(A.23)

A common alternative is to rearrange Eqs. (A.21) and (A.22) so that input quantities and output quantities are on opposite sides of the equation. For example, if the usual problem is to find the load power, then Eq. (A.21) would most usefully be expressed PL 1dBW2  PS 1dBW2  RL 1dB2  RT 1dB2

(A.24)

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APPENDIX A

Readers seeing expressions like Eq. (A.24) for the first time object to its apparent addition of dBW and dB, since they have learned by sad experience that quantities with different units should not be added. But logarithmic units are different: quantities with different logarithmic or decibel units are added; the test of correctness is whether or not the units cancel by addition or subtraction. Thus in Eq. (A.24) the dB units cancel in the subtraction of RT in dB from RL in dB and the dBWs cancel because they appear on both sides of the equal sign. The approach we have followed in this text is to use decibel units where the practice is common (principally for power) and to call the reader’s attention to other common db units at the point of first introduction. Besides the dBW and the dBm, a common power unit is the dBp, which is power expressed in dB above 1 picowatt. We must emphasize again that decibel units of resistance are irregular and that we have introduced them here only as a teaching tool.

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APPENDIX B FDM/FM/FDMA ANALOG TELEPHONE TRANSMISSION While digital modulation has many inherent advantages over analog frequency modulation (FM) for telephone signals, much of the early investment in the Intelsat network was for FDM/FM telephone systems, and these are still widely used outside the United States where telephone systems are less well developed and the transition to an all digital network has not yet occurred. This Appendix will discuss analog multiplex FDM/FM system design and characteristics in detail. Satellite FDM/FM analog telephone links resemble the terrestrial microwave pointto-point links that carried most long-distance telephone traffic between 1950 and 1985. Figure B.1 sketches a typical system. In it a multiplexer takes the baseband signals from many individual telephone conversations, translates them to adjacent channels in the RF spectrum, and combines them. Essentially, the multiplexer stacks the individual channels in non overlapping spectral bands. The resulting composite FDM signal frequency modulates an IF carrier (usually at 70 MHz) to create an FM (frequency modulation) multiplex signal. The IF carrier is converted to the appropriate uplink frequency, amplified, and transmitted to the satellite. At the satellite the signal is amplified, down-converted to the downlink band, and retransmitted. At the receiving earth station the downlink signal is amplified and downconverted to IF. The frequency modulated IF signal drives an FM demodulator, which recovers multiplex signals with the voice channels stacked in frequency. Then a demultiplexer uses product detectors and filters to translate each channel back to baseband.

Baseband Voice Signals A baseband voice signal is the voltage generated by an individual telephone set. While its detailed characteristics depend on the speaker, the telephone system in the United States assumes a flat spectrum extending from 300 to 3100 Hz. ITU-T recommends a baseband bandwidth of 300 to 3400 Hz, but some designs assume a 0–3000 Hz spectrum. We will assume the ITU-T baseband spectrum of 300 to 3400 Hz. Schematically the spectrum of a baseband voice signal is often represented by the triangle shown in Figure B.2; in a normal spectrum the peak of the triangle is to the reader’s right and in an inverted spectrum (one in which the order of frequencies has been reversed) the peak is to the reader’s left. The spectrum is not really triangular; this is just a convenient symbol. The amplitude of a voice signal in a communications link depends on where and how it is measured. In telephone engineering practice, signal powers are expressed in terms of transmission levels, their decibel levels with respect to a reference point. At the reference point, the signal power in dBm is indicated by the unit dBm0; the 0 stands for the zero transmission level point or the test point. Thus, a 2 dBm0 signal is one that produces an average power of 2 dBm at the reference point. A suitable power meter placed at the 5 dB transmission level point would measure the absolute power in the 2 dBm0 signal as 7 dBm. 491

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A

f Incoming voice channels

Frequency modulator

Multiplexer Composite FDM signal 0 to f max

Each channel has bandwidth b Hz

70-MHz FM signal Bandwidth = BIF

To upconverter and transmitter

70-MHz IF carrier (a )

70-MHz From FM signal IF amplifier (C/N)i

Demodulator

Demultiplexer Composite FDM signal (S/N)o

Outgoing voice channels (S/N)wc

(b ) FIGURE B.1 Transmitting and receiving ends of a typical FDM system. (a) At uplink earth station. (b) At downlink earth station.

Frequency (a )

Frequency (b)

492

FIGURE B.2 Representations of the spectrum of a telephone system baseband voice signal. (a) Normal spectrum. (b) Inverted spectrum.

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When telephone engineering began, the test point was accessible and meters could be connected to it. The Bell System standardized the transmission level at the outgoing side of the toll transmission switch as 2 dB below the test level point or 2 dBm0 (2 dB with respect to the zero test level point). With later switchboards, the test point lost its accessibility and disappeared. But the 2 dB standard transmission level at the toll transmission switch remained, and transmission levels are defined from this reference exactly as if the zero test level point still existed. Under Bell System standards that prevailed for many years, the long-term average power carried by a single voice channel in a telephone system was taken to be 18 dBm0 (ITU-R assumes 15 dBm0). The peak instantaneous power in the channel is about 18 dB higher or 0 dBm0. Thus, telephone equipment is often adjusted by applying a 1-kHz tone at 0 dBm0 to the system to simulate peak power on one channel. This is called the test tone. We will return to it later in our discussion of multiplexing. The reader should be aware that the original Bell and ITU-R values of 18 dBm0 and 15 dBm0 for the average power level in a single telephone channel are very conservative and that many carriers use other values in some applications. In general, the number of voice channels that a transponder can carry varies inversely with the average power level per channel1.

Voice Signal Multiplexing The process of shifting analog voice channels in frequency and combining them for transmission is called frequency division multiplexing (FDM). The procedure is hierarchical; individual channels are combined into groups; the groups are combined into larger groups; the larger groups are combined into still larger groups, and so on. The names of the groups and their internal channel arrangements vary between administrations and countries. In this section we will use terminology largely drawn from reference 4. The first step in voice channel multiplexing is to combine the baseband signals into a basic group (often called simply a group) extending from 60 to 108 kHz. The channels are stacked one above the other at 4 kHz intervals. The stacking is done by double-sideband suppressed-carrier (DSBSC) amplitude modulating each voice channel onto an appropriate carrier, filtering out the upper sideband, and saving and summing the lower sidebands. The result is a single-sideband suppressed-carrier (SSBSQ) signal. See reference 2 for a detailed explanation of SSBSC and DSBSC techniques. Figure B.3 illustrates the process. The carrier frequency in kHz of the nth channel is given by 112  4n; thus channel 1 is at the top of the spectrum and channel 12 is at the bottom. Since each channel occupies only 3.1 kHz, there are 0.9 kHz guard bands between channels. These prevent interference and simplify the filtering process when the baseband signals are recovered at the receiver. Selecting the lower sideband in the modulation process inverts the spectra of the channels, but they will be inverted again and put back in the right order at the receiver. While a single basic group could be transmitted by itself, most satellite and terrestrial microwave links carry significantly more channels. Terrestrial systems use a rigid hierarchy of channel combinations extending from the 12 channel basic group through the 60 channel basic supergroup to 600 channel basic mastergroups and beyond. The largest named combination in the Intelsat hierarchy is the basic supergroup, five groups stacked in a 240 kHz band. The stacking is done by SSBSC modulating the individual groups onto appropriate carriers and summing the resulting lower sidebands. Figure B.4 illustrates generation of a 312 to 552 kHz supergroup. The carriers are spaced 48 kHz apart; that for group 5 is highest at 612 kHz. The spectra of the individual groups are inverted when the

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APPENDIX B

1

Incoming channels

104–108 kHz BPF 108 kHz

100–104 kHz BPF

2 •

+

Multiplexed signal

•

104 kHz

•

•

•

• 60–64 kHz BPF

12

64 kHz (a )

12 60

11 64

10 68

2 72

100

1 104

108

Frequency (kHz) (b) FIGURE B.3 Multiplexing 12 telephone channels to form a basic group. (a) The basic hardware. (b) The spectrum of the multiplex signal showing individual channel spectra.

basic supergroup is formed. But the individual channel spectra in the groups are themselves inverted; the second inversion puts them back in the original order and the individual channel signals in the supergroup are frequency-shifted versions of the original baseband signals. Supergroups are normally separated by 12 to 14 kHz guard bands. Since transponder bandwidth in a satellite system is usually limited, the arrangement of channels in the Intelsat system follows a more flexible format than that used by terrestrial microwave links, In INTELSAT V, for example, earth stations are allowed 12, 24, 36, 48, 60, 72, 96, 132, 192, 252, 312, 432, 492, 552, 612, 792, 972, 1092, 1332, or 1872 voice channels. All of these numbers are divisible by 12 (the number of channels in a basic group), but the channel arrangement varies with the number of channels in order to make efficient use of the transponder. For example, 132 channels are multiplexed by combining a basic group in the 12 to 60 kHz band with one basic supergroup from 60 to 300 kHz and a second basic supergroup from 312 to 552 kHz. Table 5.13 summarizes the

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Incoming 12-channel basic groups 312–360 kHz BPF

1

420 kHz

360–408 kHz BPF

2

468 kHz

408–456 kHz BPF

3 •

516 kHz

Outgoing supergroup +

•

•

•

•

•

5

504–552 kHz BPF 612 kHz

FIGURE B.4 Schematic hardware for multiplexing five 12-channel basic groups to form a basic supergroup.

channel combinations available on the INTELSAT IV through VI spacecraft and lists some of the associated system specifications. The baseband spectrum between 0 and 12 kHz is reserved for an intrasystem channel called the order wire, which carries housekeeping information, and for the energy-dispersal signal, if used. The amplitude of a multiplexed telephone signal is a random function of time whose characteristics depend on N, the number of channels. For N greater than or equal to 24, the signal amplitude is usually represented by a Gaussian probability distribution with zero mean and rms value a. The probability p(v) that the instantaneous voltage has the value v is given by p1v2 

1 2 2 ev 2s s12p

(B.1)

For design purposes, the rms value  is usually taken as that which will result in 1 mW (0 dBm) total power at the impedance level of the system. Usually the amplitude of a multiplexed voice signal with N  24 is hard limited to lie between 3.16 times the rms value. This causes clipping for something less than 0.2% of the time and introduces negligible distortion. Thus it is common practice to equate the peak value of the signal to 3.16 times the rms values. For N  24 the multiplexed spectrum is more “peaky” and a much higher peak-to-rms ratio may be used.

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496

APPENDIX B

Frequency Modulation with Multiplexed Telephone Signals The signal-to-noise ratio described by Eq. (5.9) in Chapter 5 exists at the output of the FM demodulator and describes the ratio of the total power in the multiplexed telephone channels to the total thermal noise power. Let us now consider the FM detector output signal-to-noise ratio for a single telephone channel located at the high-frequency end of a multiplex signal. This is the (SN) at the output of the demultiplexer. If the channel bandwidth is b Hz, then the noise power output of interest is that between fmax  b and fmax. Hence the term f2 – f1 in Eq. (5.10) becomes fmax  ( fmax – b). 3 3 Since fmax W b, ( fmax  b)3  f max (1  3bfmax)  f max  3bf 2max and Eq. (5.10) becomes BIF 1¢frms 2 2 dc 2 d b f max

(B.3)

1S N2 wc  1C N2 0  10 log10 1BIFb2  20 log10 1¢frms fmax 2

(B.4)

1SN2 wc  1C N2 0 c

3BIF 1¢frms 2 2 13bf 2max 2

d  1C N2 0 c

In decibel form, Eq. (B.3) is

where the subscript wc means worst channel. See Figure B.5. Note that b is the bandwidth of one baseband voice channel (nominally 3100 Hz), and (CN)0 is the overall carrier-to-noise ratio of the link. For the narrowband voice channel indicated by the shaded region, the noise power N may be calculated by multiplying the noise spectral density at fm by b. This leads to Eq. (B.3). For wideband channels the noise power must be calculated by integrating from f1 to fm. Equation (B.3) describes the ratio of signal power to thermal noise power in a telephone channel at the upper end of a multiplexed baseband signal. But the frequency response of neither the human ear nor a telephone receiver is flat, and a telephone listener will respond differently to noise in different parts of the audio spectrum. Some of the noise that is present in bandwidth b Hz will be unnoticed, and the effective signal-to-noise ratio will be higher than that given by Eq. (B.3) by a weighting factor. Its value depends on the frequency response of the telephone receiver and of the user’s ear. ITU-T and common satellite practice use Power spectral density of noise at demodulator output

Proportional to f 2

b

f1 FIGURE B.5

fm − b fm

f2

Frequency, f

Noise power spectral density at the output of an FM demodulator.

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psophometric weighting. We will adopt the latter and use the symbol p for the psophometric weighting factor. The numerical value of p is 1.78; this corresponds to 2.5 dB4. Noise at the high-frequency end of the input spectrum to an FM detector is demodulated with greater output than noise at the low end, as discussed in Chapter 5. The rising noise at the detector output can be suppressed using a de-emphasis filter with the proper characteristics in the receiver, and the complimentary pre-emphasis filter in the transmitter. The pre-emphasis and de-emphasis filters have inverse characteristics for frequencies above a specified frequency f1. In practice the characteristics shown in Figure 5.1 in Chapter 5 need to be maintained only up to the highest baseband frequency present. The use of pre-emphasis in the transmitter and de-emphasis in the receiver improves the overall (SN) of the demodulated signal. The degree of improvement depends on the filters and the modulating waveforms used. In ITU-R standard satellite and microwave links, pre-emphasis improves the output signal-to-noise ratio of a telephone system by a factor of 2.5 (4 dB) over that given by Eq. (B.3)5. Other values apply to SCPC and TV transmission. Generally the more nonuniform the modulation spectral density, the larger the pre-emphasis improvement. Reference 4 provides several circuits for pre-emphasis and de-emphasis filters. Pre-emphasis improvement and the improvement due to psophometric weighting are independent of each other; hence the right side of Eq. (B.3) may be multiplied by p and by w to yield the psophometrically weighted signal-to-noise ratio on the worst multiplexed telephone channel at the output of an FM link using pre-emphasis and having an overall carrier-to-noise ratio of (CN)0. 1SN2 wc  1CN2 0 c

BIF 1 ¢frms 2 2 dc 2 d pw b f max

(B.5)

In decibel form, Eq. (B.5) is 1SN2 wc  1C N2 0  10 log10 1BIFb2  20 log10 1¢frms fmax 2  P  W

(B.6)

where P is 2.5 dB and W is 4 dB.

Bandwidth Calculation for FDM/FM Telephone Signals We derived Eq. (B.6) and its predecessors assuming a sinusoidally modulated waveform with an rms frequency deviation frms and requiring a transmission or IF bandwidth BRF Hz. In this section we will relate these quantities to the number of channels N carried by a multiplexed telephone signal and to the available transponder bandwidth. For link performance calculations, the rms frequency deviation frms that should be used is the rms test-tone deviation. This is the rms carrier deviation that a single 1kHz 0dBm sine wave called the test tone would produce when supplied to the modulator input, and it represents a standardized test signal in one telephone channel. Putting this another way, the transmitter is designed and adjusted to produce this rms carrier frequency deviation when the modulator input signal is a standard 1-kHz 0-dBm test tone. The rms test-tone deviation is related to the rms deviation that a multiplexed telephone signal will cause by the loading factor, l. For N voice channels, l is given by reference 5 as 20 log10 l  L  15  10 log10 1N2 20 log10 l  L  1  4 log10 1N2

for N 7 240 for 12 N 240

The product frms is called the rms multicarrier deviation.

(B.7) (B.8)

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APPENDIX B

The ratio of the peak frequency deviation fpk to the rms multicarrier deviation l  frms is given by the peak factor, g. For a large number of channels (typically N  24), g is taken as 3.16 (corresponding to 10 dB) and for small numbers of channels (typically N  24), a value of 6.5 (18.8 dB) may be used5. If necessary, the incoming voice signals may be amplitude limited to force a true peak-to-rms ratio of 3.16 on the multiplex signal. Thus for an analog FDM/FM telephone link ¢fpk  lg¢frms

(B.9)

The use of a high peak-to-mean ratio results in a low rms frequency deviation and consequently a low (SN) ratio improvement factor for an average baseband level. Amplitude limiting will cause distortion on large signal peaks, but it allows a higher average frequency deviation and a greater (SN) ratio improvement. In SCPC systems where the peak-to-mean ratio of the single telephone signal is large, companding is often used to reduce the peak-to-mean ratio of the signal before it is applied to an FM modulator. The 3.16 factor used in calculating the peak deviation of the carrier frequency represents the 0.1% extreme of a Gaussian distribution of signal voltages. For 0.2% of the time, the signal voltage will exceed 3.16 times the rms value, assuming a Gaussian probability distribution. Because we have to restrict the bandwidth of our RF signal to avoid interference with adjacent channels, the FM modulator at the transmitter will have to be preceded by a limiter that prevents large peaks of signal from overdeviating the carrier frequency. This distorts the multiplexed signal, but for N  24 the effects of limiting are small. The maximum modulating frequency, fmax, depends on the multiplexing scheme used, that is, on the number of channels multiplexed and how they are organized into basic groups, basic supergroups, and so on. When the standards of a satellite system are established, fmax is tabulated for the allowed values of N. Table B.1 contains the fmax values for INTELSAT IV through VI. If fmax is not known or if a new satellite link is being designed, a good estimate to use for fmax in kHz is 4.2 N 5. For a minimum required worst channel (SN)wc ratio, (typically about 50 dB) and an overall (CN)0 fixed by the link power budget, a satellite systems engineer may trade off values of N, BRF, and fpk. The number of channels, N, determines BRF through Carson’s rule by BRF  2 1lg¢ frms  fmax 2

(B.10)

where l and fmax depend on N. The minimum (SN)wc and BRF in turn determine the required value of ( frmsfmax) in Eq. (B.5). A satisfactory solution requires that BRF not exceed the allocated transponder bandwidth and that the rms test-tone deviation frms be achievable by the modulator. After discussing minimum (SN) ratio requirements, we will present an example calculation that illustrates the interdependence of all the variables involved.

Telephone Performance Specifications While U.S. engineers tend to think of system performance requirements in terms of decibel signal-to-noise ratios, international practice often expresses system specifications in terms of absolute channel noise levels measured in picowatts (psophometrically weighted), abbreviated pWp, or in dB above a 1-pWp reference level, abbreviated dBp. (Unfortunately the dBp abbreviation is used both for weighted and for unweighted picowatts.) Picowatts are particularly useful when noise power contributions from several sources must be combined. Decibel powers cannot be added directly. To convert between picowatts and dBp and milliwatts and dBm, it is necessary first to remember that a psophometric weighting filter reduces the power level of white noise

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by 2.5 dB and second that 0 dBp (unweighted) corresponds to 90 dBm (unweighted). If P is an absolute power level to be expressed in different units, then

and

P in dBp 1unweighted2  10 log10 1P in pWp2  2.5

(B.11)

P in dBm 1unweighted2  P in dBp  90

(B.12)

Assuming a standard 0-dBm signal level, then

1SN2 unweighted  P1dBm2  87.5  10 log10 1P in pWp2 1SN2 weighted  90  10 log10 1P in pWp2

(B.13) (B.14)

Thus a 7500-pWp channel noise level corresponds to a weighted (SN) of 51.25 dB and an unweighted (SN) of 48.75 dB. Typical satellite link designs allow 7500 to 10,000 pWp total thermal noise for the space segment (up and down links including the intermodulation noise added at the spacecraft). The Intelsat specification for the INTELSAT IV, IV-A, and V space segments is 8000 pWp 3.

Practical Examples In this section we will apply the equations that describe FDM/FM analog telephone transmission to several examples involving the INTELSAT V spacecraft. The numbers describing the satellite are taken from Table B.1. EXAMPLE B.1 An INTELSAT V transponder using a global beam achieves a (CN)0 ratio of 17.8 dB in clear air at an earth station. The transponder carries 972 channels on a single carrier; the FDM/FM signal fully occupies a 36-MHz bandwidth in the transponder. If the weighted (SN) on the top baseband channel is 51.0 dB, find the rms test-tone deviation and the rms multicarrier deviation that must be used. Compare these with the tabulated values. If the weighted (SN) on the top baseband channel is 51.0 dB, find the rms test-tone deviation and the rms multicarrier deviation that must be used. Compare these with the tabulated values. First we will illustrate the procedure to follow if the multiplexing scheme is not known. Estimating fmax as 4200  N  4.082 MHz and using standard pre-emphasis and psophometric weighting values of 4.0 and 2.5 dB, substituting into Eq. (B.6) we have 51.0  17.8  10 log10 a

¢frms 36  106 b  20 log10 a b  6.5 3 3.1  10 4.082  106

Solving for frms 51.0  17.8  40.6  6.5  13.9  20 log10 a

¢frms 4.082  106

b

Hence frms  778 kHz is the rms test-tone deviation. Under the loading rule of Eq. (B.7), L  15  10 log10 (972)  14.88 and l  10(14.8820)  5.55. Thus the rms multicarrier deviation is frms  5.55  778 kHz  4.32 MHz. We may check this answer by computing the Carson’s rule bandwidth BRF. BRF  2  (3.16  4.32 MHz  4.082 MHz)  35.5 MHz, which is close to the 36 MHz allowed by the transponder bandwidth and the original design. These are slightly different from the published values because the true value of fmax (determined by the multiplexing hierarchy) is 4.028 MHz. Using this value of fmax we find frms to be 813 kHz and the occupied bandwidth is 36.6 MHz. The published value of frms is 802 kHz; this leads to an occupied bandwidth of 36.2 MHz and a weighted (SN) of 50.9 dB. 

500

300 804 1052 1300 2044 2540 3284 3284 4028 5884

72 192 252 312 492 612 792 792 972 1332

2.5 5.0 7.5 10.0 15.0 20.0 20.0 25.0 25.0 36.0

Allocated satellite BW unit (MHz) bs 2.25 4.5 6.75 9.0 13.5 17.8 18.0 22.4 22.5 36.0

Occupied bandwidth (MHz) b0 125 180 260 320 377 454 356 499 410 591

Deviation (rms) for 0-dBm0 test tone (kHz) fr 261 459 733 1005 1488 1996 1784 2494 2274 3834

Multichannel rms deviation (kHz) fmc

Carrier-to-noise ratio in occupied BW (dB) (CN) 23.4 25.8 23.2 22.0 22.9 21.9 26.2 22.3 25.7 23.8

141.7 136.3 137.1 137.1 134.4 134.2 129.9 132.8 129.4 129.3

Source: (Reprinted with permission of the International Telecommunications Satellite Organization from Standard A Performance Characteristics of Earth Stations in the INTELSAT IV, IV-A, and V Systems Having a G T of 40.7 dB/K (BG-28-72E Rev. 1), Intelsat, Washington, DC, December 15, 1982.)

Top baseband frequency (kHz) fm

Carrier capacity (number of channels) n

Carrier-to-total noise temperature ratio at operating point (8000  200 pW 0p) (dBWK) (CT)

TABLE B.1(a) INTELSAT IV-A, V, V-A, and VI Transmission Parameters (High-Density FDM/FM Carriers)

appB.qxd 10/08/02 10:26 Page 500

Top baseband frequency (kHz) fm

60 108 156 204 252 252 300 408 552 408 552 804 552 804 1052 1052 1300 1548 1796 1796

Carrier capacity (number of channels) n

12 24 36 48 60 60 72 96 132 96 132 192 132 192 252 252 312 372 432 432

1.25 2.5 2.5 2.5 2.5 5.0 5.0 5.0 5.0 7.5 7.5 7.5 10.0 10.0 10.0 15.0 15.0 15.0 15.0 17.5

Allocated satellite BW unit (MHz) bs 1.125 2.00 2.25 2.25 2.25 4.0 4.5 4.5 4.4 5.9 6.75 6.4 7.5 9.0 8.5 12.4 13.5 13.5 13.0 15.75

Occupied bandwidth (MHz) b0 109 164 168 151 136 270 294 263 223 360 376 297 430 457 358 577 546 480 401 517

Deviation (rms) for 0-dBm0 test tone (kHz) fr 159 275 307 292 276 546 616 584 529 799 891 758 1020 1167 1009 1627 1716 1645 1479 1919

Multichannel rms deviation (kHz) fmc

13.4 12.7 15.1 18.4 21.1 12.7 13.0 16.6 20.7 12.7 14.4 19.9 12.7 14.7 19.4 13.6 15.6 18.4 21.2 18.2

154.7 153.0 150.0 146.7 144.0 149.9 149.1 145.5 141.4 148.2 145.9 140.6 147.1 144.4 139.9 144.1 141.7 138.9 136.2 138.5

(continued )

Carrier-to-noise ratio in occupied BW (dB) (CN)

Carrier-to-total noise temperature ratio at operating point (8000  200 pW 0p) (dBWK) (CT)

TABLE B.1(b) INTELSAT IV-A, V, V-A, and VI Transmission Parameters (Regular FDM/FM Carriers)

appB.qxd 10/08/02 10:26 Page 501

501

502

1796 2044 2292 1796 2044 2292 2540 3284 4028 4028 4892

432 492 552 432 492 552 612 792 972 972 1092

20.0 20.0 20.0 25.0 25.0 25.0 25.0 36.0 36.0 36.0 36.0

Allocated satellite BW unit (MHz) bs 18.0 18.0 18.0 20.7 22.5 22.5 22.5 32.4 32.4 36.0 36.0

Occupied bandwidth (MHz) b0 616 558 508 729 738 678 626 816 694 802 701

Deviation (rms) for 0-dBm0 test tone (kHz) fr 2279 2200 2121 2688 2911 2833 2755 4085 3849 4417 4118

Multichannel rms deviation (kHz) fmc

Carrier-to-noise ratio in occupied BW (dB) (CN) 16.1 18.2 20.0 14.1 14.8 16.6 18.1 16.5 19.7 17.8 20.7

Carrier-to-total noise temperature ratio at operating point (8000  200 pW 0p) (dBWK) (CT) 139.9 137.8 136.0 141.4 140.3 138.5 136.9 137.0 133.8 135.2 132.4

Source: (Reprinted with permission of the International Telecommunications Satellite Organization from Standard A Performance Characteristics of Earth Stations in the INTELSAT IV, IV-A, and V Systems Having a G/T of 40.7 dB/K (BG-28-72E Rev. 1), Intelsat, Washington, DC, December 15, 1982.)

Top baseband frequency (kHz) fm

Carrier capacity (number of channels) n

TABLE B.1(b) (continued )

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REFERENCES

503

EXAMPLE B.2 A single carrier that will occupy (when modulated) 9 MHz of an INTELSAT V transponder can produce a (CN)0 of 14.7 dB in clear air at a standard earth station using the satellite’s global beam. Assuming an 8000-pWp space segment noise allocation, how many telephone channels can the transponder carry? This example illustrates the kind of analysis a systems engineer would perform to determine telephone channel allocations for a proposed spacecraft. It requires an iterative solution. First, by Eq. (B.14), 8000 pWp corresponds to a weighted (SN) of 51.0 dB. Substituting the value for (CN)0  14.7 dB, the 3.1-kHz channel bandwidth, and an IF bandwidth equal to the 9-MHz occupied bandwidth into Eq. (B.6), we obtain 51.0  14.7  10 log10 a

¢frms 9  106 b  20 log10 a b  6.5 3 fmax 3.1  10

Solving for ( frmsfmax) 51.0  14.7  34.6  6.5  4.8  20 log10 a

¢frms b fmax

or a

¢frms b  0.57 fmax

For N channels fmax in Hz is approximately 4200N. Putting all these numbers, a g value of 3.16, and l for N  240 from Eq. (B.9) into Eq. (B.10), we obtain 9  106  2  510 114 log10 N220  3.16  0.57  4200N  4200N6

This reduces to 1071.43  N  31.61  1010.2 log10 N2  14 Substituting a few values and using trial and error to match N to the result from the above equation, we find that N  191.2 solves the equation. The tabulated value for INTELSAT V is 192. Solving this problem required a preliminary assumption that N  240. Suppose instead we had assumed N  240 when getting l from Eq. (B.7). The equation to be solved for N then would have become 9  106  2  510 11510 log10 N220  3.16  0.57  4200N  4200N6 1071.43  N  30.320  10 10.5 log10 N 2  1 4

The solution to this equation is approximately N  195.6 and it violates the hypothesis that N  240. At this point in the process the incorrect initial assumption becomes apparent. 

REFERENCES 1. W. H. BRAUN and J. E. KEIGLER, “RCA Satellite Networks: High Technology and Low User Cost,” Proceedings of the IEEE, 72, 1483–1505 (November 1984). 2. H. L. KRAUSS, C. W. BOSTIAN, and F. H. RAAB, Solid State Radio Engineering, John Wiley & Sons, New York, 1980. 3. Standard A Performance Characteristics of Earth Stations in the INTELSAT IV, IV-A, and V Systems

Having a GT of 40.7 dBK, Publication (BG-28-72E Rev. 1), Intelsat, Washington, DC, December 15, 1982. 4. Recommendations and Reports of the CCIR, 1978, Vol. IV, International Telecommunication Union, Geneva, Switzerland, 1978. 5. H. L. VAN TREES, Satellite Communications, IEEE Press, New York, 1979.

appC.qxd 28/08/02 19:41 Page 504

APPENDIX C COMPLEMENTARY ERROR FUNCTION erfc(x) AND Q FUNCTION Q(z) Equivalence Formulas and Tables of Values The complementary error function erfc(x) and the Q function Q(z) both give the area under the tail of a Gaussian distribution. The parameters x and z define the lower limit of integration of the Gaussian function, with an upper limit of infinity. The functions are important in digital communications because they define the probability that additive white Gaussian noise with a normalized rms value of 1 volt exceeds a threshold set at x or z volts, giving the probability of a symbol error due to noise (see Chapter 5 for details). A useful approximation to erfc(x) for x  1.5 is1 erfc1x2 

exp 1u2 2 1pu

where u  x 12 and  is the rms value of the Gaussian variable. An approximation for Q(z) with   1 for z  3 is2 Q1z2 

1 2 ez 2 12pz

The equivalence between erfc(x) and Q(z) is erfc1x2  2Q1 12z2 x Q1z2  12 erfc a b 12

REFERENCES 1. SIMON HAYKIN, Digital Communications, John Wiley & Sons, New York, 1988.

504

2. LEON W. COUCH, Digital and Analog Communication Systems, Macmillan, New York, 1990.

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TABLE OF Q FUNCTION Q(z)

Table of Q Function Q (z) z 0 2.0 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3.0 3.1 3.2 3.3 3.4 3.5 3.6 3.7 3.8 3.9 4.0 4.1 4.2 4.3 4.4 4.5 4.6 4.7 4.8 4.9

Q(z) 0.5 2.280 1.791 1.394 1.075 8.220 6.227 4.674 3.476 2.562 1.871 1.354 9.702 6.889 4.847 3.378 2.332 1.595 1.081 7.252 4.821 3.174 2.070 1.337 8.558 5.423 3.404 2.117 1.303 7.948 4.800

E2 E2 E2 E2 E3 E3 E3 E3 E3 E3 E3 E4 E4 E4 E4 E4 E4 E4 E5 E5 E5 E5 E5 E6 E6 E6 E6 E6 E7 E7

z 5.0 5.1 5.2 5.3 5.4 5.5 5.6 5.7 5.8 5.9 6.0 6.1 6.2 6.3 6.4 6.5 6.6 6.7 6.8 6.9 7.0 7.1 7.2 7.3 7.4 7.5 7.6 7.7 7.8 7.9 8.0

Q(z) 2.872 1.701 9.981 5.799 3.372 1.902 1.073 6.000 3.320 1.820 9.979 5.310 2.827 1.490 7.778 4.021 2.058 1.043 5.236 2.603 1.281 6.244 3.014 1.440 6.816 3.194 1.482 6.810 3.098 2.396 6.226

E7 E7 E8 E8 E8 E8 E8 E9 E9 E9 E10 E10 E10 E10 E11 E11 E11 E12 E12 E12 E12 E13 E13 E13 E14 E14 E14 E15 E15 E15 E16

505

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506

APPENDIX C

Table of Function erfc(x) x

erfc(x)

x

0.05 0.0 0.10 0.15 0.20 0.25 0.30 0.35 0.40 0.45 0.50 0.55 0.60 0.65 0.70 0.75 0.80 0.85 0.90 0.95 1.00 1.05 1.10 1.15 1.20 1.25 1.30 1.35 1.40 1.45 1.50 1.55 1.60 1.65 1.70 1.75 1.80 1.85 1.90

0.94363 1.00000 0.88754 0.83200 0.77730 0.72367 0.67137 0.62062 0.57161 0.52452 0.47950 0.43668 0.39614 0.35797 0.32220 0.28884 0.25790 0.22933 0.20309 0.17911 0.15730 0.13776 0.11979 0.10388 0.08969 0.07710 0.06599 0.05624 0.04771 0.04030 0.03389 0.02838 0.02363 0.01962 0.01621 0.01333 0.01091 0.00889 0.00721

1.95

0.00582

2.0 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3.0 3.1 3.2 3.3 3.4 3.5 3.6 3.7 3.8 3.9 4.0 4.1 4.2 4.3 4.4 4.5 4.6 4.7 4.8 4.9 5.0 5.1 5.2 5.3 5.4 5.5 5.6 5.7 5.8 5.9

erfc(x) 5.167 3.267 2.029 1.237 7.408 4.357 2.515 1.426 7.932 4.331 2.321 1.220 6.297 3.187 1.583 7.713 3.687 1.729 7.951 3.587 1.587 6.889 2.932 1.224 5.012 2.013 7.925 3.060 1.159 4.303 1.567 5.596 1.959 6.727 2.265 7.476 2.420 7.680 2.390 7.291

E3 E3 E3 E3 E4 E4 E4 E4 E5 E5 E5 E5 E6 E6 E6 E7 E7 E7 E8 E8 E8 E9 E9 E9 E10 E10 E11 E11 E11 E12 E12 E13 E13 E14 E14 E15 E15 E16 E16 E17

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APPENDIX D

THE SIMPLE ATTENUATION MODEL The first edition of Satellite Communications included an attenuation model developed by Warren Stutzman and Keith Dishman of Virginia Tech that has proved remarkably useful and accurate over the years1. The model provides an estimate of the slant-path attenuation that can be expected to occur for a given percentage of a year based on published statistics for the occurrence of heavy rain. The simple attenuation model (SAM) has the advantage over the ITU-R model in Chapter 8 that it is easy to find the attenuation that is likely to be exceeded for any rain rate, slant-path elevation angle, and frequency between 2.9 and 180 GHz. It may be less accurate than the ITU-R model in a statistical sense, but is nevertheless a very useful tool for the analysis of slant-path attenuation. The model uses an empirically derived relationship for the specific attenuation A1 on a slant path with a given rain rate R mm/h A1  aRb

dB/km

(D.1)

Specific attenuation is defined as the attenuation that is caused by rain over a distance of 1 km. The values of a and b are empirically determined constants established from the analysis of slant-path propagation measurements of attenuation and rain rate taken over a period of several years. The Satellite Communications Group at Virginia made slant-path measurements with a series of experimental satellites in the 1970s and 1980s. It was from these measurements and the work of others in the same field that the SAM was developed. The values of a and b are found from the following relationships a  4.21  105 f 2.42 4.09  102 f 0.669 b  1.41 f 0.0779 2.63 f 0.2.72

2.9  f  54 GHz 54  f  180 GHz

8.5  f  25 GHz 25  f  164 GHz

(D.2) (D.3)

Note that the values of f used in Eqs. (D.2) and (D.3) must be in GHz. The attenuation on a slant path through rain depends on two factors: the specific attenuation A1 given by Eq. (D.1) and the path length L through the rain. Hence the total attenuation on the path, A dB, is given by A  A1L  aRbL dB

(D.4)

where L is in km. On terrestrial paths, L is usually known and evaluation of Eq. D.4 is straightforward. On a slant path, the value of L is generally highly variable and unknown. Raindrops take a surprisingly long time to fall from the upper part of a storm to the ground, and usually do not fall vertically. A rain gauge located close to a receiving terminal, or under the slant path, may measure a rain rate that differs considerably from the rain rate in the slant path at high altitudes. A radar pointed along the slant path can reveal the intensity and extent of the rain, and has been used successfully to estimate slant path attenuation2, but for statistical estimation purposes we must use an effective path length, Leff. 507

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508

APPENDIX D

The concept of a statistical equivalent path length is valid because rainfall rate and slant-path attenuation show the same statistical behavior over a period of 1 year. Comparison of exceedance curves for rain rate at a receiving site and slant-path attenuation, as illustrated in Figure D.3 and in Figure 8.14 in Chapter 8, for example, shows that the shapes of the curves are very similar. We can define a path average rain rate Rp(t) such that A1t2 



L1t2

0

aR1y,t2 dy  aRp 1t2 L1t2 dB b

b

(D.5)

where y is along the slant path and A(t) and L(t) are the attenuation along the path and the path length, which are varying with time. All the rain that passes through the propagation path should eventually reach the ground. Hence the exceedance curve for the path average rain rate should be the same as the exceedance curve for the ground (point) rain rate. Therefore the attenuation value A(P) equaled or exceeded for P percent of the time should be proportional to the value of ground rain rate R(P) equaled or exceeded for the same P percent of time. The proportionality factor between A(P) and a[R(P)]b is the effective path length, Leff. Thus we can find the statistical path attenuation for any rain rate as A1P2  a3R1P2 4 bLeff dB

(D.6)

The effective path length can be found from measured slant-path attenuation and rain rate by identifying and tabulating corresponding values of A(P) and R(P) with P as a parameter. However, it is much more convenient to calculate the effective path length from an effective rain height which can be expressed by an equation. Effective rain height is a fictitious altitude at which all rain suddenly ceases. In stratiform rain, with R  10 mm/h, the SAM model calculates effective rain height He km and then sets the path length in rain as L

He  H0 km sin 1EL2

(D.7)

where H0 is the height of the earth station above mean sea level. This concept is illustrated in Figure D.1. Temperature decreases with altitude in the lower atmosphere, eventually reaching 0oC, the zero degree isotherm, at a height Hi km. Above this level, liquid water is assumed To satellite

L El

He

H0 Earth station

Sea level

FIGURE D.1 Geometry for the calculation of rain attenuation on a slant path in the SAM model. H0 is the earth station’s height above mean sea level and He is the rain height. The total propagation path length in rain is L.

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THE SIMPLE ATTENUATION MODEL

509

to be absent. Frozen water, in the form of snow or ice pellets, may exist above the zero degree isotherm, but causes much less attenuation than rain and is ignored in the SAM model. (In reality, water is often present above an altitude Hi in convective storms, carried upward by strong updrafts.) Heavy rain is always associated with convective storms, which occur primarily during the spring, summer, and fall months in temperate climates. The zero degree isotherm height Hi depends on the latitude of the earth station and is estimated by Hi  4.8 km  7.8  0.1 0 ¶e 0

km

0 ¶e 0  30° 0 ¶e 0 7 30°

(D.8)

where e is the latitude of the earth station. Note that in winter, if it snows, the zero degree isotherm is at ground level, so the value of Hi is a typical value for use in calculating slant-path attenuation, not a real value. In the SAM model, there are two rain models which have different effective rain heights. In stratiform rain, for which R  10 mm/h, the rain height is constant and equal to Hi, and the effective path length is equal to L in Eq. (D.7). In convective rainstorms, when R  10 mm/h, the effective rain height depends on the rain rate because strong storms push rain higher into the atmosphere, lengthening the slant path. However, the rain rate is not uniform with altitude, so the model creates an effective path length that may be longer or shorter than the value found in Eq. (D.7). Based on empirical data, the following expressions for effective rain height, He, were derived He  Hi km He  Hi 10 log10 1R 102 km

R  10 mm/h R 7 10 mm/h

(D.9)

In convective rain, when R  10 mm/h, a modified value of effective path length must be used Leff  µ

1  exp c gbloge a

R b L cos 1El2 d 10

R gbloge a b cos 1El2 10

∂ km R 7 10 mm/h

(D.10)

where the empirical constant   122. The procedure for determining the expected attenuation exceeded on a given slant path using the SAM model is simple. Here are the steps. 1. Determine the location of the earth station, its latitude ¶e, and the elevation angle to the satellite, El. (See Chapter 2 for the calculation of El.) 2. Locate the rain zone of the earth station using Figure D.2 (reproduced from Figure 8.15 in Chapter 8). 3. Go to Table D.1 (reproduced from Table 8.2 in Chapter 8) and find the average rain rates exceeded for the required percentages of a year. 4. Calculate the specific attenuation, A1, for the frequencies used by the satellite from Eq. D.4 using the appropriate values of a and b from Eqs. (D.2) and (D.3). 5. Find the height of the zero degree isotherm, Hi, from Eq. (D.8) and the effective rain height, He, from Eq. (D.9). 6. Calculate the effective path length, Leff, from Eq. (D.7) if R  10 mm/h, or from Eq. (D.10) if R  10 mm/h. 7. Finally, calculate the average attenuation exceeded for the specified percentage of time from Eq. (D.4): A  A1 Leff dB.

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510

APPENDIX D

165°

135°

75°

105°

45°

A

15°

A

E G C

C

G

60° B

B

E

60°

C

B E

D

E

F

K

K

Latitude in degrees N

F

H

M

30°

E

M

30°

N E

N P N

0° Equator

0° P

N

E N

C

D

E

30°

30°

K E D D A

60° 165°

Latitude in degrees S

A

135°

105°

75°

45°

60° 15°

Longitude in Degrees W FIGURE D.2 Rain climate zones for the Americas. (From Figure 1 of reference 19 of Chapter 8, © ITU. Reproduced with permission.)

Remember that the value of slant-path attenuation calculated with any statistical model is an estimated value of attenuation that will be exceeded under the specified conditions in an average year. There are no average years, so considerable fluctuation about the estimated value should be expected, especially around the 0.01% of a year level, as illustrated in Figure D.3. This is one of the factors that makes the design of satellite links difficult: the models provide only estimates of slant-path attenuation, but designs have to be based on these estimates with just a small margin added. In the first year of operation of a given satellite link, rain attenuation might exceed the estimates by several decibels, causing the link to be out of service for longer than the expected percentage of the year.

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THE SIMPLE ATTENUATION MODEL

TABLE D.1

Rainfall Rate Intensities for the Rain Climatic Zones

Percentage of Time (%)

A

B

C

D

E

F

G

H

J

K

L

M

N

P

Q

10 0.3 0.1 0.03 0.01 0.003 0.001

0.1 0.8 2 5 8 14 22

0.5 2 3 6 12 21 32

0.7 2.8 5 9 15 26 42

2.1 4.5 8 13 19 29 42

0.6 2.4 6 12 22 41 70

1.7 4.5 8 15 28 54 78

3 7 12 20 30 45 65

2 4 10 18 32 55 83

8 13 20 28 35 45 55

1.5 4.2 12 23 42 70 100

2 7 15 33 60 105 150

4 11 22 40 63 95 120

5 15 35 65 95 140 180

12 34 65 105 145 200 250

24 49 72 96 115 142 170

Note: See Figure D.2 for the Rain climate zones. Source: From Table 1 in Reference 19 of Chapter 8.

The owner of the link may believe that the design is incorrect, and complain that the performance of the link is below specification. It takes several years of operation to determine the average attenuation exceeded for a given percentage of a year, and thus to find out whether the design is correct after all. Some owners are unwilling to wait that long, so the designer of the link may be tempted to include a larger than necessary rain attenuation margin to guarantee that the performance specification is met every year. EXAMPLE D.1 The SAM model is used here to estimate the rain attenuation that will be exceeded for 0.01% of an average year on a 45o elevation angle slant path from an earth station in Blacksburg, Virginia, to a satellite operating at a frequency of 28 GHz. The latitude of the earth station is 37.229o N with an elevation H0  0.640 km. From Figure D.2, the earth station is in ITU region K, and Table D.1 gives a rainfall rate of 42 mm/h exceeded for 0.01% of an average year. From Eqs. (D.2) and (D.3), the calculated values of the coefficients a and b at a frequency of 28 GHz are (independent of the rainfall rate) a  4.21  105 f 2.42  0.134  1.062 b  2.63 f 0.272 The latitude of the earth station is 37.229, so Eq. (D.8) gives the average zero degree isotherm height as Hi  7.8  0.1  37.229  4.08 km For 0.01% of the year the average rain rate exceeded is 42 mm/h, which is greater than 10 mm/h, so Eq. (D.9) gives He  Hi log10 1R 102  4.08 0.62  4.70 km

The effective path length exceeded for 0.01% of an average year can then be found from Eqs. (D.7) and (D.10) He  H0 4.70  0.64   5.74 km sin 1EL2 sin 45° R 1  exp c gbloge a b L cos 1EL2 d 10 Leff  µ ∂ R gbloge a b cos 1EL2 10 L

gbloge 1R 102 cos1EL2  1 22  1.062  loge 14.22  cos 45°  0.0490 Leff  11  e0.049  e 5.74 2 0.049  5.0 km

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APPENDIX D

10.0000 11.7 GHz 19.04 GHz 28.56 GHz

Percentage of time attenuation is exceeded

1.0000

0.1000

0.0100

0.0010

0.0001 0.0

5.0

10.0

15.0

20.0

25.0

30.0

35.0

Attenuation (dB) FIGURE D.3 Cumulative attenuation distribution for the calender year 1978 as measured at the authors’ earth station on satellite downlinks at three frequencies. The 11.7-GHz path was at 33° elevation; the other two were at 45°.

The attenuation exceeded on the slant path is then given by Eq. (D.4) A  aRbLeff  0.134  421.062  5.0  35.5 dB Comparison with measured data for the author’s earth station in the year 1978 in Figure D.3 shows the measured value to be 32 dB. A variation of a few decibels can be expected when comparing attenuation measured for a specific year to the statistical value. The SAM model works well for higher rain rates and attenuation levels. At low rain rates it tends to underestimate attenuation, partly because it does not take account of attenuation occurring in clouds and above the zero degree isotherm. 

REFERENCES 1. W. L. STUTZMAN and W. K. DISHMAN, “A Simple Model for the Estimation of Rain-Induced Attenuation along Earth–Space Paths at Millimeter Wavelengths,” Radio Science, 17, 1465–1476, November–December 1982. 2. R. L. OLSEN, D. V. ROGERS, and D. B. HODGE, “The aRb Relation in the Calculation of Rain Attenuation,” IEEE

Transactions of Antennas and Propagation, AP-26, 318–329, March 1978. 3. Rec. ITU-R PN.837-1, Characteristics of Precipitation for Propagation Modelling, 1994.

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GLOSSARY OF TERMS AND ACRONYMS Definitions of Common Packet Network Abbreviations The acronyms set out below have come from many documents. However, a large number have been abstracted from two sources: (a) a collection put together by Dr. Gory Fairhurst and Dr. Tim Spracklen, and presented at a spring 1996 VSAT course given at the University of York, England; and (b) from references 3 and 4 in Chapter 9. 10B2 10-MHz baseband LAN using thin coaxial cable (of IEEE 802) 10B5 10-MHz baseband LAN using thick coaxial cable (of IEEE 802) 10BF 10-MHz baseband LAN using fiber (with repeaters) (of IEEE 802) 10BT 10-MHz baseband LAN using twisted pair cable (of IEEE 802) 802 Family of network standards specified by the IEEE 802.3 A LAN standard based on Ethernet (see 802, 10BX) 802.11 A wireless LAN standard for the 2.4-GHz ISM band A Address field (usually follows the flag at the frame header) ABM Asynchronous Balanced Mode (a CO-DL protocol in HDLC) AC Alternating Current ACK (i) Sequence number indicating correct reception in a CO protocol ACK (ii) A flag in TCP indicating that an acknowledgment is present ACTS Advanced Communications Technology Satellite ADC, A/D Analog to Digital Converter (converts analog signal to digital stream) ADF Automatic Direction Finder (radio navigation device) ADPCM Adaptive Differential Pulse Code Modulation

AFC Automatic Frequency Control AKM Apogee Kick Motor ALOHA Shared random packet access channel AM Amplitude Modulation AM-PM (Unwanted conversion of) Amplitude Modulation to Phase Modulation AMI Alternate Mark Inversion coding (bipolar) (in high-speed PL) AOCS Attitude and Orbit Control System AOR Atlantic Ocean Region ANSI American National Standards Institute ARP Address Resolution Protocol (translated IP 1 HA) (cf. RARP) ARQ Automatic Repeat reQuest (i.e., frame retransmission procedure) ASI Adjacent Satellite Interference ASCII American Standard for the Computer Interchange of Information ASIC Application Specific Integrated Circuit ASK Amplitude Shift Keying ATM Asynchronous Transfer Mode (of B-ISDN) AUI Automatic Unit Interface (standard interface to an LAN MAU) AWGN Additive White Gaussian Noise Abort 8-bit sequence, which ends an invalid HDLC frame (cf. Flag, idle) Az Azimuth BBP Baseband Processing bps Bits Per Second (unit of clock, utilization, throughput); also bits/s B Byte (group of 8 bits) (sometimes known as an octet) BCC Block Checksum Character in asynchronous PL (cf. CRC) BCH Bose–Chaudhuri–Hocquenghem (family of error correcting codes) BER Bit Error Ratio (or Rate) in digital circuit (cf. SNR in an analog circuit) BFN Beam Forming Network 513

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514

GLOSSARY OF TERMS AND ACRONYMS

BISYNC An IBM protocol using one-by-one frame acknowledgement (BSC) B-ISDN Broadband Integrated Services Digital Network BPF Bandpass filter BPSK Binary Phase Shift Keying BSC An IBM protocol using one-by-one frame acknowledgement (BISYNC) BSS Broadcast Satellite Service BTC Block Turbo Codes BTR Bit Timing Recovery CA Course Acquisition (one of the codes transmitted by GPS satellites) CATV CAble Television (Originally Community Antenna Television) C & M Control and Monitoring C band Radio frequency band 4–8 GHz CBTR Carrier and Burst Timing Recovery (used in preamble of TDMA frames) CCIR An international committee of the ITU on radio (now called the ITU-R) CCITT An international standards committee of the ITU (now called the ITU-T) CCMF Centralized Control and monitoring Functions CD Compact Disc CDI Course Deviation Indicator (radio navigation display) CDC Control error Delay Channel CDM Code Division Multiplexing (form of spread spectrum multiplexing) CDMA Code Division Multiple Access (common form of spread spectrum access) CL Connection-Less (datagram) (e.g., UI, LLC1, Ethernet, IP, UDP) CMC Convolve Multiply Convolve (a process used in Acousto-Optical MCDs) CO Connection-Oriented (e.g., HDLC ABM, TCP) COTS Commercial-Off-The Shelf CPU Central Processor Unit (or microprocessor) CR Carrier Recovery CRC Cyclic Redundancy Check (check for bit errors) CSC Common Signaling Channel CUG Closed User Group (common VSAT private user groups) CI Carrier power to Interference noise power ratio

CI0 Carrier power to Interference noise power (per Hz) ratio CIM Carrier power to Intermodulation noise power ratio CN Carrier power to thermal Noise power ratio CN0 Carrier power to thermal Noise power (per Hz) ratio CPFSK Continuous Phase Frequency Shift Keying CT Carrier power-to-noise Temperature ratio CTC Convolutional Turbo Codes CSMA/CD Carrier Sense Multiple Access with Collision detection (LAN) DA Demand Assignment (allocation of resource for communications) DAC, DA Digital-to-Analog converter (converts digital stream to analog signal) DAMA Demand Assigned Multiple Access (for efficient use of resource) DBS-TV Direct Broadcast Satellite Television dBHz Decibel-Hertz [10 log10(bandwidth in Hz)] dBK Decibel-Kelvins [10 log10(noise temperature in K)] dBW Decibel-Watts [10 log10(power in W)] dBm Decibel-milliwatts [10 log10 (power in mW)] DC Direct Current DC (Frequency) Down Converter DCE Data Circuit-terminating Equipment (modem side of a PL) DGPS Differential GPS DISC Disconnect request frame in HDLC (CO-DL) (cf. FIN in TCP) DL Data Link layer (Layer 2 of OSI) (e.g., Ethernet, HDLC) DM Disconnected Mode frame in HDLC (rejects a request) DME Distance Measuring Equipment (radio navigation aid) DNS Domain Name Service (converts IP name to/from IP address) DNTX Do Not Transmit DOD U.S. Department Of Defense DOP Dilution of Precision (GPS accuracy parameter) DPLL Digital Phase Locked Loop clock recovery for synchronous circuits DSAP Destination Service Access Point (part of IEEE LLC frame) DSBSC Double-SideBand Suppressed Carrier

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GLOSSARY OF TERMS AND ACRONYMS

DS-CDMA Direct Sequence Code Division Multiple Access DS-SS Direct Sequence Spread Spectrum (same as DS-CDMA) DS-n Hierachy for digital transmission rates DSP Digital Signal Processing DTE Data Terminal Equipment (computer side of a PL) DTH Direct-To-Home (VSAT terminal in a private home) DVB-S Digital Video Broadcast standard for Satellite systems Demux Demultiplexer (separates previously multiplexed circuits) (cf. Mux) E1 European data rate and framing standard for circuits at 2.048 Mbit/s EEPROM Electrically Erasable Programmable Read Only Memory EIRP (e.i.r.p.) Equivalent Isotropically Radiated Power (combines power and antenna gain) El Elevation ELT Emergency Locator Transmitters ELV Expendable Launch Vehicle EM Electromagnetic EbN0 Energy per bit over thermal Noise power (per Hz) ratio ENST Ecole Nationale Supérieure des Telecommunications de Bretagne EODL End Of Design Life EOF End Of Frame EOML End Of Maneuvering Life ES Earth Station ES End System (i.e., User’s computer) supporting all OSI layers ESA European Space Agency ETSI European Telecommunications Standardization Institute EU European Union E–W East–West (station-keeping maneuver) F Flag (cf. Flag) FAW Frame Alignment Word FCC Federal Communications Commission FCS Frame Check Sequence (identical to CRC) (cf. BCC) FDDI Fiber Distributed Data Interface (Ring LAN operating at 100 Mbit/s) FDM Frequency Division Multiplexing (a method of combining signals at different frequencies into a single signal)

515

FDM-FM-FDMA Frequency Division Multiplexed Frequency Modulation Frequency Division Multiple Access FDMA Frequency Division Multiple Access FEC Forward Error Correction FEP Front End Processor FET Field Effect Transistor FFSK Fast Frequency Shift Keying (similar modulation to MSK) FIFO First-In First-Out (i.e., a queue or buffer) FIN TCP flag bit indicating the end of a TCP connection FM Frequency Modulation FPGA Field Programmable Gate Array FPLMTS Future Public Land Mobile Telecommunications System FR Frame Relay (connection-oriented datagram circuit) FRMR Frame Reject frame in CO-DL HDLC (indicates link error) FSK Frequency Shift Keying FSS Fixed Satellite Service FTP File Transfer Protocol (of IP stack) (also generic term) Flag (i) The 8-bit delimiter between HDLC frames (cf. SYN, abort, idle) Flag (ii) A bit which indicates a binary state (e.g., status flag, SYN, FIN) G universal gravitational constant (6.672  1011 Nm2/kg2) GaAsFET Gallium Arsenide Field Effect Transistor (Low noise RF transistor) GEO Geostationary Earth Orbit (GSO) GES Gateway Earth Station GHz GigaHertz (units of 109 Hz) GMSK Gaussian Minimum Shift Keying (MSK using Gaussian shaped pulses) GLONASS GLObal NAvigation Satellite System (Russian Federation equivalent of GPS) GPS Global Positioning System GSM Global System for Mobile communications (ETSI standard) GSO Geostationary Satellite Orbit (GEO) GTO Geostationary Transfer Orbit GT Gain-to-noise Temperature ratio (of a receiving system) HA Hardware Address in IP and Ethernet (PL address of an interface)

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GLOSSARY OF TERMS AND ACRONYMS

HBE Hub Baseband Equipment HCI Hub Control interface HDTV High Definition Television HDB3 High Density Bipolar 3 coding (in highspeed PL) (cf. AMI) HDLC High Level Data Link Control (family of DL protocols) HEO Highly Elliptical Orbit HF High Frequency HP Horizontally Polarized HPA High-Power Amplifier HPC High-Power amplifier and (frequency) Converter Hub The central earth station of a VSAT STAR network I Information frame in HDLC (carries data in a CO-DL) (cf. UI) I In-phase component of a signal IC Integrated Circuit ICBM InterContinental Ballistic Missile ICO Intermediate Circular Orbit Idle Sequence of digital 1’s indicating no frames in HDLC (cf. Flag, abort) IM InterModulation IO Input or Output of a computer interface ICMP Internetworking Control and Monitoring Protocol (of IP suite) IDU Indoor Unit (of a VSAT earth station) IEE Institution of Electrical Engineers (UK) IEEE Institute of Electrical and Electronic Engineers (USA) IF Intermediate Frequency (between baseband and RF) IFL Interfacility Link (cable/fiber link between ODU and IDU) IFRB International Frequency Registration Board IHL Internet Header length (length of the PCI in an IP datagram) ILS Instrument landing system IM Intermodulation (two or more frequencies creating unwanted products) IMC Instrument Meteorological Conditions (in the clouds) Inbound Channel establishing a connection from a STAR VSAT toward the hub Inroute Channel establishing a connection from a STAR VSAT toward the hub

IP

(i) Internetworking Protocol (CL network protocol of IP suite) IP (ii) Suite of Internet protocols using IP network protocol IPX Popular proprietary network protocol (Novell) (cf. IP) IS Intermediate System (e.g., router, switch) supporting up to OSI NL ISI InterSymbol Interference ISL InterSatellite Link ISDN Integrated Services Digital Network (all digital public network) ISO International Standards Organization (standards committee of the UN) ISS International Space Station ITU International Telecommunications Union ITU-R Radiocommunication sector of the ITU (formerly CCIR) ITU-T Telecommunications Standardization Sector of the ITU (formerly CCITT) JPEG Joint Picture Experts Group JPL Jet Propulsion Lab kbps Kilobits/second (1000 bps) of clock, utilization, and throughput (also kbit/s) kHz kiloHertz (units of 103 Hz) kB KiloByte (1000 bytes) (210 bytes) (cf. B for Byte) K band Radio frequency band 16–24 GHz Ka band Radio frequency band 24–36 GHz Ku band Radio frequency band 8–16 GHz L band Radio frequency band 1–2 GHz LAAS Local Area Augmentation System (enhanced GPS) LAN Local Area Network (e.g., Ethernet) (cf. WAN, MAN) LAP Link Access Protocol (defined by X.25) (see also LAPB) LAPB Link Access Protocol Balanced (defined by X.25) (see also LAP) LAPD Link Access protocol on the D (i.e., data) Channel LC Line Code (the digital stream after the AD used to drive the modulator) LEO Low Earth Orbit LET Linear Energy Transfer LHCP Left-Hand Circular Polarization LIE Line Interface Equipment LIM Terrestrial Line Interface Module

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GLOSSARY OF TERMS AND ACRONYMS

LLC Logical Link Control (DL of IEEE stack) (similar to HDLC) LLC1 CL-DL protocol using UI frames based on HDLC (IEEE 802 suite) LMDS Local Multipoint Distribution System LO Local Oscillator (see VCXO) LORAN LOng RAnge Navigation LNA Low Noise Amplifier (used at the front end of an earth station receiver) LNB Low Noise Block (front end receiving equipment with LNA, VCXO, etc.) LNC Low Noise amplifier and (frequency) Converter LP Linearly Polarized LPC Linear Predictive Coder LPE Linear Predictive Encoding (used in speech and video compression) LPF Low Pass Filter LRE Low bit Rate Encoding lsb Least significant bit (rightmost bit of a byte or word) LT Line Termination LU Logical Unit (of SNA) MAC Medium Access Control (lower part of DL in IEEE 802 stack) MAN Metropolitan Area Network (or regional network) (cf. LAN, WAN) MAU Medium Attachment Unit (LAN transceiver electronics (PL) MBONE Multicast IP backbone network (used for multimedia networking) MCD MultiCarrier Demodulator MCDD MultiCarrier Demodulator and Demultiplexer MCDDD MultiCarrier Demodulation, Demultiplexing, and Decoding MCPC Multiple Circuits Per Carrier (medium-tohigh capacity link) MCS Master Control Station MEO Medium Earth Orbit MF-TDMA MultiFrequency Time Division Multiple Access MHz MegaHertz (units of 106 Hz) MMIC Microwave Monolithic Integrated Circuit MODEM Modulator/Demodulator converts digital signal from/to line code (cf. LC) MPEG Moving Picture coding Expert Group (created video bit rate compression)

517

msb Most significant bit (leftmost bit of a byte or word) MSK Minimum Shift Keying (form of FSK) MSM Microwave Switching Matrix (used for interconnecting links at RF or IF) MSS (i) Maximum Segment Size (largest allowed payload at TCP layer) MSS (ii) Mobile Satellite Service MTBF Mean Time Before Failure MTU Maximum Transfer Unit (max. size of DL frame in IP) (cf. MSS) Mux Multiplexer (combines circuits into a single bit stream) (cf. Demux) NACK Negative ACK in a CO protocol (retransmission request) (e.g., REJ) NASA National Aeronautical and Space Agency (U.S. space agency) NBFM NarrowBand Frequency Modulation NCC Network Control Center NDB NonDirectional Beacon (radio navigation aid) NF Noise Figure N-ISDN Narrowband ISDN based on circuits at 64 kbps (64 kbits/s) NIST National Institute of Science and Technology (formerly the U.S. Bureau of Standards) NL Network Layer (layer 3 of the OSI stack) (see IP, X.25) Non-GSO Non-Geostationary Satellite Orbit NPSD Noise Power Spectral Density NRZ Non Return to Zero N–S North–South (station keeping maneuver) NRZ Non Return to Zero coding within the bit period (1  high, 0  low) (PL) NT Network termination NTSC National Television Standards Committee (Established color TV standards in the United States) OBP OnBoard Processing (used on advanced satellites to connect links, etc.) OC-n Hierachy for fiber-optic bit rates ODU Outdoor Unit (of a VSAT terminal) OEM Original Equipment Manufacturer OMT Orthomode Transducer (separates modes/polarizations in antenna feeds) OQPSK Offset keyed QPSK (QPSK with a half symbol time shift between I and Q signals)

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518

GLOSSARY OF TERMS AND ACRONYMS

OSI (i) Open Systems Interconnection (reference model) OSI (ii) Suite of protocols defined using the OSI reference model (cf. IP) Outbound Channel establishing a connection from the hub to a STAR VSAT Outroute Channel establishing a connection from the hub to a STAR VSAT PABX

Private Automatic Branch Exchange

Packet Information block identified by a label at layer 3 of ISO-OSI stack PAD Packet Assembler/Disassembler (interface between packet and nonpacket networks) PAL Phase Alternate Line (color TV standard used in Europe and elsewhere) PBX Private Branch Exchange PCE Processing and Control Equipment (part of the hub BBP equipment) PCI Protocol Control Information (header) (of OSI reference model) PCM Pulse Code Modulation PCN Personal Communications Network PCS Personal Communications System PDN Public Data Network PDU Protocol Data Unit (of OSI) (e.g., a packet at NL, or frame at DL) PFD Power Flux Density (power per m2 at a given plane) PISO Parallel-in Serial-out shift register used in serial transmitter (PL) ping Network testing application protocol (ICMP/IP) PLMN Public Land Mobile Network PL Physical Layer (layer 1 of OSI stack) (e.g., RS-232, RS-449) PLL Phase Locked Loop clock recovery at PL (see also DPLL) PN PseudoNoise POS Point Of Sale POTS Plain Old Telephone Service PSK Phase Shift Keying PSTN Public Switched Telephony Network PVC Permanent Virtual Circuit Q Quadrature phase component of a signal QAM Quadrature Amplitude Modulation (QPSK and ASK combined) Q band Radio frequency band 36–46 GHz QPSK Quadrature phase Shift Keying

RA Random Assignment (also called ALOHA) RARP Reverse Resolution Protocol (translates HA 1 IP address) (cf. ARP) REJ REJect request frame (Go-Back-N ARQ in HDLC) (cf. SREJ) RF Radio Frequency RG58U Type of flexible coaxial cable used for IEEE 802.3 10B2 LANs RHCP Right-Hand Circular Polarization RLV Reusable Launch Vehicle RNR Receiver Not Ready HDLC supervisory frame (CO-DL) (positive ACK) RRC Root Raised Cosine (Frequency response of zero-ISI filter) RS Reed–Solomon (block error detecting and correcting code) RS-232 V.24 serial interface using 25-pin Dconnector (cf. RS-449) RS-449 Differential synchronous serial interface using 37-pin D-connector) RTT Round-Trip Time (time to receive a response from remote system) RZ Return to Zero coding within the bit period (i.e., 0 between bits) (in PL) S band Radio frequency band 2–4 GHz SA Selective Availability (applied to GPS signals until May 2000) SABM Set Asynchronous Balanced Mode in HDLC (modulo-8, CO-DL) SABME SABM Extended frame in HDLC (modulo-128, CO-DL) SAM Simple Attenuation Model SAP Service Access Point (of OSI) (e.g., port in UDP or TCP) SAR Specific Absorption Rate SARSAT Search And Rescue SATellite SAW Surface Acoustic Wave SC Service Channel SC SpaceCraft (satellite) SCADA Supervisory Control And Data Acquisition SCC (i) Subnetwork Control Center (ii) Satellite Control Center (controls satellite payload when in orbit) SCPC Single Channel Per Carrier (nonmultiplexed, low capacity FDMA) SCPC-FDMA-DA Single Channel Per Carrier Frequency Division Multiple Access Demand Access

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GLOSSARY OF TERMS AND ACRONYMS

SCU

Subnetwork Control Unit

SDH Synchronous Digital multiplexing Hierarchy (European, see SONET) SDLC Synchronous Data Link Control (An IBM version of HDLC (usually a polled link) allowing multiple unacknowledged frames) SDU Service Data Unit (of OSI reference model) (cf. PDU) SER Symbol Error Rate SES

Société Européenne de Satellites

SIPO Serial-In Parallel-Out shift register used in serial receiver equipment SISO Soft Input–Soft Output (used in error correction decoders) SLIP Serial Line IP (DL protocol using asynchronous communications) SNA Proprietary suite of protocols used by IBM (cf. IP) SNAP SubNetwork Access Protocol supports IP over LLC (IEEE 802.3) SNR (SN) Signal-to-Noise Ratio (analog quantity measured in dB) (cf. BER) SMPT Simple Mail Transfer Protocol (TCP/IP e-mail application) SOHO Small Office/Home Office (relatively low capacity user) SONET Synchronous Optical Network; U.S. standard similar to SDH SOTF

Start Of Transmit Frame

S-PCN Satellite Personal Communications Network sps Symbols per second SREJ Selective Reject request frame (in HDLC CO-DL) (cf. REJ) SSAP Source Service Access Point (part of IEEE LLC DL frame) SSB Single SideBand modulation (a narrowband form of AM) SSBSC Single-SideBand Suppressed-Carrier modulation SS-FDMA Satellite Switched FDMA SSMA Spread Spectrum Multiple Access (see also CDMA) SSPA

Solid-State Power Amplifier

SS-TDMA Satellite Switched TDMA SSTO Single Stage To Orbit STDM Statistical Time Division Multiplexing (cf. TDM or packet mode)

519

SYN (i) Synchronization character (8 bits) delimiting a block (cf. Flag) SYN (ii)TCP flag bit indicating the start of a TCP connection) TCP Transmission Control Protocol (CO transport protocol of IP suite) TCP/IP Transmission Control Protocol–Internet Protocol TDD Time Division Duplexing TDM Time Division Multiplexing (sharing a link in time) TDM-SCPC-FDMA Time Division Multiplexing Single Channel Per Carrier Frequency Division Multiple Access TDMA Time Division Multiple Access (sharing a resource in time) TDRSS Tracking and Data Relay Satellite System TE Terminal Equipment TEC Total Electron Content telnet Remote log-in and terminal emulation application protocol (TCP/IP) THz TerraHertz (units of 1012 Hz) TNC Terminal Node Controller T1 U.S. data rate and framing standard for circuits of 1.544 Mbps (Mbits/s) T-n U.S. heirachy for digital transmission rates ToS Type of Service in IP (indicates type of data being transported) TR Token Ring (IEEE 802.5) LAN protocol using a ring architecture TTC Telemetry, Tracking, and Control TTC&M Telemetry, Tracking, Control, & Monitoring TTL (i) Time To Live in IP (number of hops before packet is discarded) TTL (ii) Transistor–Transistor Logic family of integrated circuits TST Time domain, Space domain, Time domain switching TTY TeleTYpe TV TeleVision TVRO TeleVision Receive Only TWTA Travelling Wave Tube Amplifier TX-PCE Transmit Processing and Control Equipment UA Unnumbered Acknowledgement frame in CO-DL HDLC UART Universal Asynchronous Receiver Transmitter (device)

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520

GLOSSARY OF TERMS AND ACRONYMS

UC (Frequency) Up-Converter UDP Universal Datagram Protocol (CL transport protocol of IP suite) UHF Radio frequency band 300 MHz–1 GHz UI Unnumbered Information CL frame in HDLC (cf. LLC1) ULPC Uplink Power Control (one countermeasure against signal fade) (UPC) UMTS Universal Mobile Telecommunications System UNIX A common “open” operating system used by many computers UPC Uplink Power Control (one countermeasure against signal fade) (ULPC) UPT Universal Personal Telecommunications USART Universal Synchronous/Asynchronous Receiver Transmitter UT Universal Time (Equal to GMT) UTP Unshielded Twisted Pair cabling (used in 10BT LANs) UW Unique Word V band Radio frequency band 46–56 GHz VCO Voltage Controlled Oscillator (used with mixer to change freq.) VCXO Voltage Controlled Crystal Oscillator VHF Radio frequency band 30–300 MHz VOIP Voice Over Internet Protocol VOR VHF Omnidirection Range beacon (radio navigation beacon) VOW Voice Order Wire (Station–Station voice link) VP Vertically Polarized VSAT Very Small Aperture Terminal VSB Vestigial Sideband (form of AM used in TV broadcasting) V.24 Recommendation (of ITU-T) for serial communications (RS-232) VT Virginia Tech WAAS Wide Area Augmentation System (enhanced GPS) WAN Wide Area Network (e.g., the Internet) (cf. LAN, MAN) WARC World Administrative Radio Conference WBFM WideBand Frequency Modulation WGS-84 World Geodetic System 1984 WLL Wireless Local Loop WWI World War I

WWII World War II XPD Cross-Polarization Discrimination (a measure of signal polarization) XPI Decibel ratio of wanted power to unwanted power X.25 Recommendation (of ITU-T) for packetswitched communications (specifies interface between DTE and DCE for terminals operating in the packet mode and connecting PDNs by dedicated circuit)

ITU-R Standards Applicable to VSAT Systems Existing Standards and Recommendations • Recommendation ITU-R S.725, Technical Characteristics for Very Small Aperture Terminals (VSATs), 1992. [Earth station RF aspects] • Recommendation ITU-R S.726-1, Maximum Permissible Level of Spurious Emissions from Very Small Aperture Terminals (VSATs), 1992–1993. [Permitted emitted power in given bands] • Recommendation ITU-R S.727, Cross-Polarization Isolation from Very Small Aperture Terminals (VSATs), 1992. [Cross-polarized power to be used in interference calculations] • Recommendation ITU-R S.728, Maximum Permissible Levels of Off-Axis E.I.R.P. Density from Very Small Aperture Terminals (VSATs), 1992. [Power levels away from the main beam axis to be used in interference calculations] • Recommendation ITU-R S.729, Control and Monitoring Function of Very Small Aperture Terminals (VSATs), 1992. [Functions that the VSAT should be able to control in its operation] Draft Standards and Recommendations Being Considered • Draft New Recommendation, Connection of VSAT Systems with Packet-Switched Public Data Networks (PSDNs) Based on ITU-T Recommendation X.25, 1995. [ITU-R standard on the interconnection of VSATs with Public Switched Networks] Standards and Recommendations under Preparation • Connection of Private VSAT Networks to the Public ISDN.

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GLOSSARY OF TERMS AND ACRONYMS

• ETS 300 160 Control and Monitoring at a Very Small Aperture Terminal (VSAT). • ETS 300 161 Centralized Control and Monitoring for VSAT Networks. • ETS 300 194 The Interconnection of VSAT Systems to Packet Switched Public Data Networks (PSPDNs).

European Technical Standards (ETS) for VSAT Systems Existing Standards • ETS 300 157 Receive-Only Very Small Aperture Terminals (VSATs) Used for Data Distribution Operating in the 1112 GHz Frequency Bands. • ETS 300 159 Transmit/Receive Very Small Aperture Terminals (VSATs) Used for Data Communications Operating in the Fixed Satellite Service (FSS) 111214 GHz Frequency Bands. • ETS 300 333 Receive-Only Very Small Aperture Terminals (VSATs) Used for Data Distribution Operating in the 4 GHz Band. • ETS 300 332 Transmit/Receive Very Small Aperture Terminals (VSATs) Used for Data Communications Operating in the Fixed Satellite Service (FSS) 6 GHz and 4 GHz Frequency Bands.

521

Physical Constants a  re Me G k

Radius of GEO (42,164.17 km) Kepler’s constant (3.986004418  105 km3/s2) Mean earth radius (6378.137 km) Mass of the earth (5.98  1024 kg) Universal Gravitational constant (6.672  1011 Nm2/kg2) Boltzmann’s constant (1.39  1023 J/K  228.6 dBW/K/Hz)

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INDEX A A-law, 169 ACK see Acknowledgement signal AKM see Apogee Kick Motor ALOHA, 355 AMCS-1, 7 AMPS see analog mobile phone service AOCS see attitude and orbit control system ARQ see automatic repeat request ASCII code, 277 AT&T, 425 ATM see asynchronous transfer mode ATS-6, 86, 309 AWGN see added white Gaussian noise Absolute bandwidth, 180 Absorption atmospheric, 297 resonant, 307 water vapor, 308 Adaptor, booster, 58 Acceleration, 17 Access Control Protocols, 349 Access, random, 254 Accuracy CA code, 480, 481 GPS, 464 timing (GPS), 475, 476 2DRMS, 464 AceS Asia Cellular, 9 Acknowledgment signal, 286, 350 Across plane ISL seam, 422 Active device, 411 Acquisition sequential, 472 Added white Gaussian noise, 191 Adjustment factor height, 301 horizontal, 301 Afristar, 8

522

Agila 2, 11 Allocations, frequency, 97 American Mobile Satellite Corp., 7 Ameristar, 8 Amos 1, 12 Amplifier back-off, 77, 115, 124, 357 front end, 107 IF, 107 noiseless, 105 quasilinear, 124 RF, 107 Analog FM transmission, satellite, 164 mobile phone service, 157 signals, digital transmission of, 201 TV transmission (from satellite), 115 video, 439 Angle azimuth, 31 canting, 327 central, 33 elevation, 31 minimum grazing, 419 phased array scan, 412 tilt, 327 Angular distance, 30 momentum, 20 velocity, 25 Anomaly eccentric, 25 mean, 26 true, 25 Antenna aperture, 81 array, 80 axially symmetrical, 367 beam, hemi, 81 beam, spot, 75 beam, zone, 75 beamwidth, 82

beamwidth and gain, 115 boresight, 101, 366 Cassegrain, 367 cell structure, 410 coma, 415 contour, 3 dB, 121 DBS-TV, 442, 446 deployment, 86 dual polarized, 445 edge of coverage loss, 117 efficiency, 82 fD ratio, 415 feed, 107 front-fed, 367 gain, 80, 82, 101 gain, non-circular coverage, 117 global, receive horn, 58 global, transmit horn, 58 Gregorian, 367 hopping beam, 410 horn, 80 horn, scalar, 366 inflatable, 85 isotropic, 101 main lobe, 414 mispointing loss, 115 multiple beam, 221, 248 noise, 332 nulls, 80 offset fed, 121, 367 omnidirectional, 80 parabolic torus, 439 paraboloid, 80 pattern, 80, 125 phased array, 411 radiating elements, 411 reflector, 80 satellite, 59, 80 satellite, gain, 432 scan loss, 416 scanning beam, 410 sidelobe level (mask, ITU-R), 125 Simulasat, 439 smart, PCS, handset, 145

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INDEX

solid angle, 101 spot beam, 248, 412 steerable phased array, 249 telemetry and command bicones, 58 wire, 80 Aperture antenna, 80 (antenna) efficiency, 82 effective, 102 Apogee, 22, 23 Apogee kick motor, 44, 48, 63 Apollo, 394 Arab satellite, 9 Arabsat, 9 Ariane -5 -40 -42P -44P -42L -44LP, 45 Ariane 44, 5 Ariane 5, 46 Aries, first point of, 27 Arthur C. Clarke, 1, 4 ANIK IA, 4 ANIK E1, E2, 12 Apparent orbital period, 391 Apstar-V, 6 Arc coverage, 432 jets, 63 Argument of perigee west, 28 Ascending node, 28 Ascension, right, 27, 28 Asia Broadcasting and Communications, 9 Asia satellite, 9 Asiasat, 9 Asiastar, 8 Astra DBS-TV satellites, 442 IH, 5 1A, 1B, etc., 11 K, 14 Astrolink, 248 Asynchronous transfer mode (ATM), 256 Atlas II, IIAS, 45 Atlas IIAS, 425 Atlas IIAS, IIIA, IIIB, V, 46 Atmospheric absorption, 297 Attenuation downlink, rain, 130 rain, 15, 116, 297, 317, 507–510 rain margin, 120 uplink, rain, 130

Axially symmetric antenna, 367 Axis semimajor, 23 semiminor, 23 Atmospheric attenuation, 103 drag, 17 loss, 103 multipath, 310 Atmosphere neutral, 298 standard, 307 Attenuation, atmospheric, 103 cloud, 297, 308 differential, 327 gaseous, 307 rain, 15, 116, 297, 507–510 rain, at Ku-band, 147 rain, prediction, 317 scaling (frequency and elevation), 126, 325 specific, 297, 318 total path, 122 zenith, 307 Attitude and orbit control system, 57, 60 sensor, 65 Automatic repeat request, 287, 350 go-back-N, 287 selective repeat, 287 stop-and-wait, 287 Availability link, 295 threshold, 296 Azimuth angle, 31 B BAPTA see bearing and power transfer assembly BER see bit error rate BER vs. Eb No, 364 BPSK see binary phase shift keying BS-3N, 9 BTR see bit timing recovery Back-off, output power, 77, 115, 124, 129, 357 Background, galactic, 24 Baikonur (Cosmodrome), 392 Bandlimited channel, 182

523

Bandpass filter, 75 transmission of digital data, 179 Bandwidth absolute, 180 channel, 180 FDM/FM, 497 limited, 276 noise, 115, 355 occupied, 180, 355 occupied, equation, 180 Base, customer service, 420 Baseband digital signals, 172 onboard processing transponder, 59 noise power, FM, 160 processor (VSAT), 368 SN ratio, FM signals, 159 transmission, digital signals, 172 Basic transmission theory, 100 Bathtub curve (failure rates), 89 Batteries, 71 Nickel-Hydrogen, 72 capacity, 72 Baud (rate), 172, 235 Baudot, 172 Beamwidth 1 dB, 37 3 dB, 82, 306 Bearing and power transfer assembly, 58 Beginning of life, 71 Bell System, 3 Bent pipe (transponder), 59, 78, 113, 370 Big LEOs, 411 Binary phase shift keying, 180, 187, 362 Bit error rate, 100, 188, 296 vs. CN, 196 Bit flip (radiation effect), 398 redundant (FEC), 187 timing recovery, (TDMA), 362 Bipropellant (maneuvering fuel), 63 Black body radiation, 105 Block code, 280

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524

INDEX

Blockage path, L-band, 140 PCS, body, 145 PCS, trees, 145 Bluetooth, 257 Boltzmann’s constant, 115 Booster adapter, 58 BorealisTM, Ellipso orbit, 426 Boresight, antenna, 101, 366 Box, station-keeping, 41 Brahe, Tycho, 22 Brasilsat, 9 Broadcast satellite radio, direct, 440 satellite television, direct, 72, 118, 439 Broadcasting satellite system, 9 Bsat-1A, -1B, 9 Budget link, 113, 295 noise power, 134 power, 134 Bulges, equatorial, 39 Burst error, 281 C CA code see coarse acquisition code and code CBTR, 237 CCIR now called ITU-R CDMA, 14, 258, 262, 355 capacity, 262 DS-SS, 262 frequency hopping, 257 Gold code, 258, 266 Kasimi code, 258 spreading codes, 258 CN see carrier-to-noise CN margin DBS-TV, 447 CN ratio DBS-TV, 450 GPS receiver, 475 overall, 113, 123 rules of thumb, 128 CONUS see continental United States CP see circular polarization CRC see cyclic redundancy check C-band, 3 GEO link budget example, clear air, 114 GEO link budget example, worst case, 114

Cable TV, 439 Canting angle, 327 Capacity, channel, 275 Carrier recovery BPSK, 190 TDMA, 362 Carrier (signal), 75 Carson’s rule, FM signals, 159 Cassegrain antenna, 102, 367 Cape Canaveral, 3, 44, 392 Carrier-to-Noise, 15, 100, 362 overall, 100, 113 negative, 261 reciprocal formula, 128 vs. BER, 196 Cartesian coordinate system, 20 Catalyst (hydrazine propellant), 63 Celestial mechanics, 17 Cell fixed earth, 430 structure (antenna), 410 Cells, solar, 59, 71 Cellular telephony, 157 Central angle (satellite coverage), 33 Centrifugal force, 18 Centripetal force, 18 Challenger, 44 Chandra (X-Ray telescope), 44 Channel capacity, 275 capacity, FDM/FM, 232, 488–493 coding, 363 I and Q, 183 synchronization, TDM, 212 Channelization, of transponder, 76 Characteristic waves (ionosphere), 310 Chinese Long March rocket, 47 Circular polarization DBS-TV, 444 depolarization, 297 Circuit carrier recovery, 190 hypothetical reference, 128 Circumscribed circle (orbit), 25 Clarke, Arthur C., 1, 4 Clarke orbit, 388 Clear air (or sky), 116 sky (level), 295

Climate parameters (rainfall rate), 314 Clock atomic, 467 bias, 465 offset, 464 Cloud attenuation, 297, 308 Coarse acquisition code, see code CA Code ASCII, 277 BCH, 280, 283 binary cyclic, 280 block, 280 block, systematic, 280 CA, 264, 460, 467–470 convolutional, 284, 450 distance, 281 efficiency, 274 Golay, 281 Gold, 468 Kasami, 258 locally generated, 471 outer, 296 P, 460, 467, 472 Reed-Solomon, 291 spreading, see CDMA trellis, 285 turbo, 275, 285, 292 weight, 281 Code division multiple access see CDMA Coding, 273 channel, 363 concatenated, 290 gain, 146, 282 Coefficients, regression, 318 Coherent detector, 189 Cold start (CDMA correlation), 470 Columbia, 7 Coma, antenna, 415 Combining CN and CI, 127 Command (satellite control), 70 Communications satellite act, 3, 4 Communications subsystem, 59, 72 Companding, 204 Complementary error function (erfc), 192 Compressed digital video, 439 Compression and expansion, 204

ind.qxd 28/08/02 19:33 Page 525

INDEX

Compression, speech, 207 Comsat, 3, 7 ConcordiaTM, Ellipso orbit, 426 Constant Boltzmann’s, 105, 115 Kepler’s, 19 gravitational, 19 Constraint length (convolutional code), 284 Continental United States (coverage), 118 Contour, antenna, 3 dB, 121 Control attitude, satellite, 64 packet, 251 power, uplink, 126, 371 system, orbit, 66 system, thermal, 59 Control structure (satellite maneuvers) command word, 70 control code, 70 execute, 70 quality (components), 87 Controller, terminal node (data packet), 171 Conus beam, 448 Convective rain, 298 Convolutional encoding, 450 Coordinate system cartesian, 20 geocentric, 20, 27 orbital plane, 21 polar, 21 rectangular, 27 Co-polarized, 297 Correlation detector, 189 Cosines, law of, 34 Costas loop, 479 Coverage arc, 432 (area and region), 406 central angle, 33 edge of satellite antenna beam, 115 instantaneous, 410, 415 stationary, 430 Criterion, Nyquist, 174, 179 Cross-correlation, 469 Cross polarization, 297 Cross polarization discrimination (XPD), 303 isolation (XPI), 305

Cumulative distribution function, 312 Cumulative probability distribution, 312 Customer service base, 420 Cyclic redundancy check (CRC), 256 D DAMA see demand assigned multiple access DA-SCPC see demand assigned DBS see direct broadcast satellite DBS-TV see Direct Broadcast Satellite Television Direct Broadcast Satellite Television, 6, 15, 389 Congress, law, 120 DBS-1 R, 443 DBS-TV link budget, 449 DME see distance measuring equipment DOP see dilution of precision DSBSC see double sideband suppressed carrier DS-SS see direct sequence–spread spectrum DS-SS CDMA, 262 DTH see direct to home DVB-S see digital video standard Damper, nutation, 58 Data transmission, analog FM channels, 170 Date, Julian, 28 Day Julian, 28 sidereal, 25 mean solar, 25 Decibels, 487–490 Declination, 27 Decoding soft input, soft output (SISO), 293 Viterbi, 284 Defocusing, 310 Degraded performance, 296 Delay lock loop, 478 Delay, propagation, 311, 421 Demand assigned multiple access, 139, 249, 345

525

Demodulator BPSK, coherent, 190 FM, threshold, 119, 169 QPSK, 200 Demultiplexing, 221 Delta II, 45, 425 Delta III, 425 Delta III, IV, 46 Demodulator, FM, 160 Density, flux, 101, 120, 125 Depolarization ice crystal, 297, 332 rain, 297 Descending node (orbit), 28 Determination, orbit, 42 Design combining CN and CI, 127 example, PCS, LEO system, 137 examples, system, 132 life, end of, 38 uplink, 124 to specified CI ratio, 127 to specified CN ratio, 127 system, to specific performance, 131 Despun antenna, 61 shelf, 58 Deutsche Telekom Geschaftsbereich, 9 Detector coherent, 189 correlation, 189 Deviation see frequency deviation Deviation frequency (FM), 158 over (FM TV), 166 ratio, FM, 161 Device active, 411 passive, 411 Differential attenuation, 327 modulation, PSK, 187, 190 PCM, 211 phase, 327 Digital carriers, US standards, 211 compressed TV, 452 demodulation, 187 heirarchy, 211 modulation, 187

ind.qxd 28/08/02 19:33 Page 526

526

INDEX

Digital (Continued) transmission, 172 transmission of analog signals, 201 video standard, 441 voice, SN, 206 Dilution of precision (GPS errors) DOP, 466 GDOP, 481 HDOP, 481 VDOP, 482 Dip latitude (magnetic), 400 Direct broadcast satellite radio, 440 broadcast satellite television, 72, 118, 439 insertion launch, (satellite), 48, 49 PSK modulation, 187 radiating phased array, 412 sequence–spread spectrum, 261 to home (service delivery), 389 Directv, 7, 119, 441, 444 receiving antenna, 446 Discrete cosine transform, 208 Dishman, 507 Dishnetwork, 442 antenna, 445 Distance angular (orbit), 30 Hamming (coding), 280 measuring equipment (navigation), 462 minimum (coding), 281 Distortion, group delay, 78 Domestic (satellites), 125 Doppler shift, 48, 462, 471 Double-hop links, 348 Double conversion see superheterodyne Double sideband suppressed carrier, 166 Downlink, 100 CN, 131 CN budget, PCS, 145 design, 112 Drift, orbital, 39, 42, 57 Duration, solar eclipse (satellite), 71 Dwell time, 394

E EIRP see effective isotropic radiated power ECEF coordinate system, 464 ERFC (erfc) complementary error function function, 192 table, 504 EIRP see effective isotropic radiated power ELV see expendable launch vehicle ENST, 292 EOC see edge of coverage EODL see end of design life EOML see end of maneuvering life ESA see European Space Agency Early Bird, 1, 389, 424 Earth average radius, 20 core, 400 mantle, 400 station, 15 station, simplified, 107 station, for satellite systems, small, 117 East-West maneuver, 41 station keeping, 63, 67 Eccentric anomaly, 25 Eccentricity, 22, 24, 394 Echo I and II, 3 Echostar, 6, 7, 443 Eclipse, solar, 52, 71 Ecliptic, 40 Edge of coverage loss, 117, 416 Effective isotropic radiated power, 101, 296 uplink, 123 Effective pathlength, 301, 507, 508 Efficiency, aperture, 102 Electrical noise, 105 Elements orbital, 20, 29 radiating, antenna, 411 Elevation angle, 31 considerations of, 408 minimum, 37, 408 scaling of attenuation with, 325 Ellipso, 425

Elliptical orbit, 22, 394 Embratel, 9 Emergency locator transmitter, 463 Emissions, off-axis (antenna), 360 Emphasis De-, 161, 163 Pre-, 161, 163 End of design life, 38, 390 life, 71 maneuvering life, 38, 390 Energy, solar radiation, 71 European Space Agency, 44 Equalizer, 78 training sequence, 182 transversal, 182 Equation, waveform, FM, 158 Equator geographic, 400 geomagnetic, 400 Equatorial bulges, 39 orbit, 391 plane, 2 Equiprobable values (statistical), 319 Equivalent noise source, 105, 109 output, 110 Error burst, 281 function, complementary, 192 quantization, 202 sampling, 201 Error control, 273 DBS-TV, 451 Error correction, 273 DBS-TV, 452 forward, 187 Error detection, 273 DBS-TV, 452 Error rate bit, 188 BPSK, 194 QPSK, 194 symbol, 188 Example, NGSO system design, 432 Expendable launch vehicle (ELV), 43 Explorer 1, 3 Explorer, Mars, 405

ind.qxd 28/08/02 19:33 Page 527

INDEX

Eutelsat, 10 Examples, system design, 132 Exceedance curves, 312 F FAW see frame alignment word FCC see Federal Communications Commission fD ratio, antenna, 415 FDM see multiplexing, frequency division FDM/FM, 491 FDMA, 77, 124, 139, 157, 222, 232, 355 FDMA-SCPC-DA, 139, 250, 252 FDM-FM-FDMA, 223, 225 FEC see forward error correction FFSK see fast frequency shift keying FM-FDM see Appendix B FM see frequency modulation FM Carson’s rule, 159 companding, 169, 204 demodulator, 160 improvement, 157, 161 satellite TV, 441 threshold example, 168 waveform equation, 158 FS see fixed service FSK see frequency shift keying FSS see fixed satellite service Fade margin, 296 Fading, low angle, 309 Failure rates (bath tub curve), 89 Faraday rotation, 297, 310 Fast frequency shift keying, 200 Federal Communications Commission 74, 120, 444 Feed loss, 122 matrix, antenna, 411 phased array, 83 Fiber, optical, 156 Field, gravitational, 38 Figure, noise, 111 Filter bandpass, 75 Butterworth, 178 Chebychev, 178

elliptic function, 178 matched, 178 roll-off (RRC), 177 First point of Aries, 27 Fixed access, 223 assignment, 224 earth cell, 430 power sharing, 231 satellite service, 409 service, 412 Flag (start of frame, etc), 256 Flux density, 101, 120, 125 Focus, prime, 305 Force, 17 centrifugal, 18 centripetal, 18 gravitational, 18 in-plane, 39 out-of-plane, 40 Formula, reciprocal CN, 128 Forward error correction, 122, 187, 273, 296 margin, PCS, 147 Fourier transform, Nyquist ISI filter, 176 Fractional transmission coefficient, 332 Frame alignment word (TDM), 209 basic TDMA, 362 reframing, 210 relay, 351 TDMA, Iridium, 421 Free fall, 18 Frequency allocations, 97 band, 406 deviation, FM, 158 deviation, peak (Carson’s rule), 159 deviation, rms multicarrier, 497 highest (Carson’s rule), 159 intermediate (IF), 367 L1, 460 L2, 460 modulation, 15, 157 modulation threshold, 118 (see also modulation) radio (RF), 16, 367 reuse, 221, 303, 410 scaling of attenuation, 325 shift keying, 157

527

Frequency Division Multiple Access see FDMA Frequency Division Multiplexing see FDM Front-fed antenna, 367 Function, transfer, FM pre/de-emphasis, 163 G GaAsFET see gallium arsenide field effect transistor GEO see geostationary earth orbit GEO/LMDS, 15 GEO orbit requirements, 25 GPS see global positioning system GT, 105, 112, 116, 349 Gain antenna, 101, 297, 432 coding, 146 Galactic background, 24 Galaxy -5 -6 -9 -1RR, 8 Galileo, 461 Gallium arsenide field effect transistor, 106 Garuda 1, 9, 11 Gateway (earth station, hub), 138, 139, 143 Gaussian distribution, 191 GE Americom, 7 Gemini, 394 Generation of QPSK signals, 198 Geographic equator, 400 Geomagnetic equator, 400 Geostationary earth orbit, 1, 4, 35 transfer orbit, 46, 48 Geosynchronous, 35, 37 graveyard orbit, 39 Global beam, 80 transmit horn (antenna), 58 receive horn (antenna), 58 Global positioning system, 3, 6, 264, 389, 458 differential, 461, 466, 482–484 kinematic, 483 time, 464, 466 Globalstar, 5, 13, 223, 258, 411, 424, 426 Glonass, 460

ind.qxd 28/08/02 19:33 Page 528

528

INDEX

Gold code see code and CDMA Grazing angle, minimum, 419 Graveyard, geosynchronous orbit, 39 Gravitational constant, 19 field, 38 force, 18 Gravity gradient boom (attitude control), 67 Greenwich meridian, 31 observatory, 28 Gregorian antenna, 367 Grid reference, 30 Ground track, 395 Group delay distortion, 78 Growth, incremental, 424 Gstar 4, 7 Guard bands, 355, 493 Guiana space center (Kourou), 44 H HDLC see high level data link control HDTV see high definition TV HEO see highly elliptical orbit HPA see high power amplifier Hale cycle (see also sunspot cycle), 400 Hamming distance (coding), 280 Handset radiation safety limits, 420 PCS, 138 Height adjustment factor, 301 Hemi-beam, satellite antenna, 80 High definition TV, 16, 156 Highly elliptical orbit, 4, 455 High level data link control, 353 High power amplifier, 78, 367 equalization, 230 linearization, 226 nonlinearity, 226 quasi-linear, 230 Home satellite TV, 440 Hopping beam (antenna), 410 Horizontal adjustment factor, 301 Horn antenna (scalar), 366 corrugated, 82

Hot bird (DBS-TV), 10 Hub station, 99, 138 Hughes, 48 Hughes Electronics Corporation, 441 Hybrid multiple access, 223 Hydrazine, 62 Mono-methyl, 63 Hydrometeors, 297 Hypergolic (propellant), 63 Hypothetical reference circuit, 128 I ICO global, 390 ICO, New, 390, 428 IDU see Indoor Unit IF see intermediate frequency IFL see inter-facility link ILS see instrument landing system IM see intermodulation ISI see interference, intersymbol ISL see inter-satellite link ISO-OSI model, 255 ISS see international space station ITU see international telecommunications union Ice crystal depolarization, 332 Impact, micrometeor, 71 Implementation margin, 121, 141, 186, 197 Improvement FM, 157, 161 subjective, 205 Indoor unit (VSAT), 368 Information (see also Shannon), 275 theory, 275 throughput, 420 Inbound (inroute) signal, 138, 356 Inclination, 28 moon’s orbit, 40 earth’s spin axis, 40 sun’s equatorial plane, 40 Inclined orbit operation, 37, 66, 390 Incremental growth, 424 Inertial space, 25 Infra red sensor (attitude control), 65

Injection, low side (mixing), 107 Inmarsat see international maritime satellite organization Inner code, 363 In-route signal, 356 Instantaneous coverage, 410, 415 Instrument landing system, 461 Intelsat see international telecommunications satellite organization Integrity monitoring, 466, 472, 483 INTELSAT K, 5 INTELSAT I, 4 INTELSAT V, 5 INTELSAT VI, 5, 58 INTELSAT VII, 5, 424 INTELSAT VIII, 5 INTELSAT IX, 5 Inter-facility link, 368 Interference level, 125 inter-symbol (ISI), 129, 170, 172 intersymbol, zero ISI, 175 off-axis, 126, 364 Interim operations, 424 Interleaved code, 296 Interleaving, 290 DBS-TV, 451 Intermediate frequency, 106, 107, 180 Intermodulation, 75 products, 77, 124, 128 third order, 226–230, 357 International maritime satellite organization, 5, 10, 84 International telecommunications satellite organization, 3, 10, 73, 425 International space station, 17, 389 International telecommunications union, 3, 125 Internet satellites, 431 traffic centers, 432 Interpolation, 291, 451

ind.qxd 28/08/02 19:33 Page 529

INDEX

Inter-satellite link, 394, 422, 426 across plane, 422 Inter-symbol interference, 129 Ionosphere, 480 Ionospheric scintillation, 297, 312 Iridium, 5, 13, 389, 394, 411, 418, 428 Isotropic antenna, 101 J JCSat, 10 JPEG see joint picture experts group Jets, arc, 63 Joint picture expert group, 208 Julian day, 28 Julian date, 28 Juno I, 3 K Ka band, 5, 79 Kasami code see code, Kasami Kazakhstan, 42, 392 Kennedy space flight center, 44 Keplerian orbit, 38 Kepler’s constant, 19 Kepler’s laws, 22 Kopernicus -1 -2, 9 Koreasat, 11 Korea telecom, 45 Kourou (Guiana space center), 44 krad(Si) (space radiation level), 402 Ku band, 4 downlink, system design example, 134 rain attenuation, 120, 147 rain effects, 135 uplink, system design, example, 133 L LEO see low earth orbit LEOs, big, 411 LMS see land mobile service LNA see low noise amplifier LNB see low noise block (converter) LP see linear polarization

LPC see linear predictive encoding L band, 7 L-star, 9 Land mobile service, 428 Latch up (permanent bit flip), 400 Latitude, 30 dip angle (magnetic), 400 Launch direct insertion into orbit, 48 rockets, next generation, 46 vehicle price, 46 vehicle selection factors, 47 Law A-, 169 of cosines, 34 -, 169 Laws and Parsons, 316 Laws of motion, 17 Length, constraint (convolutional code), 284 Level, interference, 125 side lobe, 125 Life, beginning of, 71 end of, 71 Lifetime, design, 38 maneuvering, 38 operational, 37 Limited, power, 72 Linear energy transfer (radiation), 403 polarization, 297, 310, 326 predictive encoding, 207 Linearity (of transponder), 77 Link availability, 295 budget, 113, 295 budget equation, 103 budget example, C-band, earth coverage, 115 budget, example, Ku-band DBS TV, 121 CN ratios, VSAT star network, 375 combining CN and CI ratios, 127 design, satellite, 96 equation, 100, 103

529

inbound, PCS, 141 margin, 116, 295 margin calculation, VSAT star network, 370 margin, with FEC, PCS, 146 outbound, PCS, 144 performance, 295 Lobe, main, antenna, 414 Local oscillator (LO), 78, 107 Location, primary, Indian, 31 Long March rocket, 47, 425 Longitude, 30 Look angle determination, 30 Look angles, 31 Loral Skynet, 8 Loss edge of coverage, 416 feed, 122 mechanisms, 298 miscellaneous, 116, 143 path, 103, 125, 297 propagation, 116, 297 scan, antenna, 416 waveguide, 111 Low angle fading, 309 earth orbit, 4, 5, 99, 388 noise amplifier, 104, 107, 122, 367 noise block, 107, 369 noise block converter, 446 side injection (mixer), 107 Luminance, TV, 165 M MEO see medium earth orbit m-sequence see PN sequence MF-TDMA see multi-frequency TDMA MPEG-2, 16, 117, 165, 208, 447, 451 MSAT 1, 12 MSK see minimum shift keying, 200 MSS see Mobile Satellite Service MTBF see mean time between failures Maneuvers, station keeping, 60 Maps, rain climatic, 314 rainfall exceedance contour, 316 Mabuhay Philippines, 11

ind.qxd 28/08/02 19:33 Page 530

530

INDEX

Maneuver, orbital, east-west and north-south, 41 Maneuvering life, end of, 38 Margin fade, 75, 296 implementation, 121, 141, 186, 197 link, 295 link, with FEC, 146, 147 system, 103 Mars Explorer, 405 Mascons see mass concentrations Mass concentrations, 39 Master control station, 392 DBS-TV, 452 Master group, 493 Mean time between failure, 90 Mean anomaly, 26 solar day, 2 Medium earth orbit, 4, 5, 16, 99, 388 Melting layer, 298, 300 Mercury (capsule), 388, 394 Mesh network, 348 Microburst, rain, 302 Micrometeor (impacts), 71 Microwave source, hot (sun), 53 Minimum elevation angle, 37 grazing angle, 419 shift keying, 200 Misalignment, polarization, 142 Mixer, 107 Mixing, turbulent, 308 Mobile satellite service, 409, 425 terminal, 140 Modulation, 156 cross product (FM), 159 differential, 190 frequency, 491 16-QAM, 247 sine wave, 160 Modulo-8, 354 Modulo-128, 354 Molniya, 4, 67, 396 orbit, 396 Momentum angular, 20 wheels, 60 wheel, unload (dump energy), 64

Monitoring system (on board), 68 Monochrome (TV), 165 Morse code, 15 Motion, laws of, 17 planetary, 22 Mu-law (-law), 169 Multi-frequency TDMA, 359 Multipath, atmospheric, 310 Multiple access, 221 demand assigned, 139 hybrid, 223 Multiplexing, 156, 221, 493–495 analog, 156 frequency, 156 frequency division, 156, 223, 493 N NACK see negative acknowledgement NASA, 88 NGSO see non-geostationary satellite orbit NGSO system design, 432 N2H2 see hydrazine NRZ see non return to zero NS 513, 11 NTSC (TV standard, US), 165, 168 Nadir, 31 Nahuel 1, 11 NahuelSat, 11 Narrowcasting, 347 Navigation message, 460, 470, 472–473 Navstar GPS, 14 Negative acknowledgement signal, 286, 350 Neutral atmosphere, 298 Next generation launchers, 46 New ICO, 390, 411, 428 New Skies company, 5, 11 Newton, 17, 18, 23 Newtonian equations, 17 Nickel-hydrogen batteries, 72 Node, ascending, 28 descending, 28 Noise bandwidth, 115 bandwidth and symbol rate, RRC filter, 121

contributions, 128 electrical, 105 figure, 111 figure, standard temperature, 111 power, baseband, FM, 160 power budget, DBS-TV, 450 power spectral density, 105, 160, 161 source, equivalent, 106, 109 source, equivalent output, 110 suppression, FM demodulator, 160 temperature, 105 temperature increase (due to rain), 116, 122 temperature, sky, 116 temperature, system, 105 thermal, 129 white, 161 Noiseless amplifier, 105 Non-GEO rain attenuation prediction, 324 Non-geostationary satellite orbit (NGSO) systems, 427 Non return to zero (line code), 172, 174 QPSK channel, 186 North-south maneuver, 41 station keeping, 63 Number of NGSO satellites per plane, 433 total, 433 Nutation, 60 damper, 58 Nyquist criterion, 174, 179 O OBP see on board processing ODU see outdoor unit OMT see orthogonal (ortho) mode transducer Observatory, Greenwich, 28 Occupied bandwidth, 180 Off-axis emission, 360 interference, 364 scanning, 412 Offset-fed antenna, 367 Omnidirectional antenna, 80 On board processing, 76, 247–248, 370

ind.qxd 28/08/02 19:33 Page 531

INDEX

Operational lifetime, 37 Operations, interim, 424 Optical fiber, 156 Options, replenishment, 424 Optimizing system performance, PCS, 146 Optimum orbital altitude, 418 Orbcomm, 13, 393, 407, 424, 429 Orbit apogee, 22, 23 determination, 42 eccentricity, 22 elliptical, 22, 30 equatorial, 391 geostationary transfer, 46 graveyard, geosynchronous, 39 highly elliptical, 455 inclined, 37 osculating, 38 perigee, 22, 23 perturbations, 38 precession, 40 stable, 17 Orbit raising, slow, 48 Orbital altitude, optimum, 418 drift, 39 elements, 20, 38 elements (determination of), 68 maneuvers, 41 period, 23 period, apparent, 391 plane, 23 radius, 19 slot spacing, 74 slots, 84 stability, spinner, 60 stability, three axis, 60 velocity, 19 Orthogonal polarization, 221 Ortho-mode transducer (OMT), 305, 444 Oscillator, local (LO), 78 Osculating orbit, 38 Orthogonal mode transducer, (OMT), 305, 444 polarizations, 297 Outage, 100, 113, 120 sun transit, 53 DBS-TV, 447

Outbound (outroute) signal, 138, 357 Outdoor unit (VSAT), 368 Outer code, 296, 363 Output power back-off, 77, 115, 124, 129, 357 saturated, 115, 129 Out-route signal, 356 Over deviation (FM TV), 166 Overall CN ratio, 113, 123 P P code, 460, 467, 472 PAM see pulse amplitude modulation PAS 1, 8 PCM see pulse code modulation PCS see personal communication service PLL see phase lock loop PM-AM conversion, 124 PN see pseudo noise PSK see phase shift keying PSTN see public switched telephone network PT Pasifik Satelit, 11 Packet radio, 254 header, 256 Palapa C1, 11 PanAmSat, 8 Parity, 277 single, 278 Passive device, 411 phased array, 412 Path attenuation, total, 122 blockage, 149 loss, 103, 125, 297 Path length, effective, 298, 301, 507, 508 physical, 319 Pay per view, 447 Peenemunde, 404 Pegasus, 424, 425 Performance degraded, 296 design to, specific, 131 link, 295

531

optimizing, system performance, PCS, 146 threshold, 296 Perigee, 22 time of, 26 west, argument of, 28 Perihelion, 25 Period anomalistic, 38 GEO satellite, 24 orbital, 23 symbol, 187 Personal communication service, 137 design example, LEO system, 137 inbound (inroute) link, 141 outbound (outroute) link, 144 Perturbations, orbit, 38 Phase channel, in, 183 continuous, 171 equalizers, 182 front, 412 lock loop, 239 quadrature, 183 Phase shift keying, 157, 173 binary, 180, 187, 189, 362 continuous, 171 generation, QPSK, 198 minimum, 171 QPSK variants, 199 quaternary, 116, 121, 187, 198, 296 Phased array antenna, 411 direct radiating, 412 scan angle, 412 Pitch axis (satellite), 63 Plane, equatorial, 27 across seam, ISL, 422 orbital, 28 number of (NGSO) satellites per, 433 Planetary motion, 22 Points, stable and unstable, 39 Polarization circular, 297, 326, 444 linear, 297, 310, 326 misalignment, 142 orthogonal, 72

ind.qxd 28/08/02 19:33 Page 532

532

INDEX

Power carrier, uplink, 123 control, uplink, 126, 148, 371 limited, 72, 275 noise, spectral density, 105 output, back-off, 77, 115, 124, 129, 357 output, saturated, 115, 129 system, 59, 71 thermonuclear, 71 Precession, orbit, 40, 405 Precise code see P code Prediction rain attenuation, GEO, 317 rain attenuation, NGSO satellites, 324 site diversity gain, 336 XPD, 326 Preflight testing, satellite, 88 Pressure, solar, 60 Price, launch vehicles, 46 Prime focus, 305 Primestar, 441, 442 Probability of symbol error, 191 Products, intermodulation, 77, 124, 128 Prograde orbit, 391 Project Score, 3, 388 Propagation impairment countermeasures attenuation, 333 depolarization, 337 diversity, 335 power control, 334 signal processing, 335 site diversity, 336 Propagation loss, 116, 297 Propellant tanks, 58 Protocol, 255 spoofing, 286, 352 stack, 350 window, 351 X.25, 256 Protocols, access control, 349 Proton rocket, 42, 425 Proton M, 46 Primary location, Indian Ocean, 31 Atlantic Ocean, 32 Project Score, 3 Psuedorange (GPS), 465 Pseudonoise (sequence), 467 Public switched telephone network, 426, 428

Pulse amplitude modulation, 201 code modulation, 201, 356 Pyrotechnic (deployment), 86 Q Q factor see subjective improvement factor Q function (Gaussian), 192 tables, 505 QAM see quadrature amplitude modulation QPSK see quaternary phase shift keying Quadrature amplitude modulation, 173, 235 Qualification, space, 87 Quality assurance, 87 control, 87 Quantization error, 202 Quantizing, 201 Quaternary phase shift keying, 116, 121, 198, 296 spectrum of, 184 R RF see radio frequency RHI see range height indicator (radar) RLV see reusable launch vehicle RRC see root raised cosine (filter) RZ see return to zero (line code) Rad-hard see radiation hardened Radar, S-band, 303 Radiating elements, antenna, 411 Radiation belts (Van Allen), 398, 402, 426 black body, 105 dose, 403 effects, 398 hardened, 403 hazards, EM, 124 linear energy transfer, 403 safety, handset, 420 Radio frequency (RF), 16, 107, 367 Radius, average earth, 20 orbital, 19, 22

Rain accumulation, 312 added noise temperature, 111 attenuation, 15, 116, 297, 317, 507–510 attenuation at Ku-band, 147 attenuation margin, 120 attenuation prediction, 317 attenuation statistics, DBS-TV, 449 climatic maps, 314 convective, 298 effects, Ku-band, 135 exceedance contour maps, 316 height, 508 microburst, 302 stratiform, 298 streamer, 302 Raindrops absorption, 297 distribution, 315 scatter, 297 shape, 327 size, 302 Rainfall rate, 312 Rain gauge, tipping bucket, 312 Random access, 254 Range height indicator, 303 Range ambiguity, 483 Ranging (orbital ephemeris), 59 tones (range determination), 69 Ratios CN and CI, combining, 127 receive power, 102 Receiver command, 58 DBS-TV, 446 GPS, 476–480 Reciprocal CN formula, 128 Recovery, carrier (TDMA), 362 Redundancy (spare), 75, 87, 90 parallel, 91 ring, 91 series, 91 Redundant bits (coding), 274 Reed-Solomon code, 296, 363 DBS-TV, 450

ind.qxd 28/08/02 19:33 Page 533

INDEX

Reference circuit, hypothetical, 128 grid, 30 station, 482 Reflector shaped, 367 offset parabolic, 445 Refractive effects, 308 Reframing (TDM), 210 Regression coefficients, 318 Reliability, 87, 88 Repeater, 15 Replenishment options, satellite, 424 Resonant absorption, 307 Retrograde orbit, 391 Return to zero (line code), 172 Reusable launch vehicle, 43 Reuse frequency, polarization, 72 frequency, spatial, 72 Right ascension, 27 Roll axis (satellite), 63 Root raised cosine filter, 121, 173, 355 roll-off factor, 177 Rotation, Faraday, 297, 310 Rules of thumb, CN, 128 S SA see selective availability SAM (simple attenuation model), 507 SARSAT, 463 SBS-4, 5, 6, 8 SCPC see single channel per carrier SCPC-FDMA, 226 SDARS see satellite digital radio service SES see Societe Europenne des Satellites SISO (soft input soft output) see decoding SN see signal-to-noise SSHPA see solid state high power amplifier SSPA see solid state power amplifier SSTO see single stage to orbit STK see satellite tool kit STS see space transportation system Safety, radiation handset, 420

Sampling, 201 Satellite antenna gain, 432 axes, defined, 63 digital radio service, 445 domestic, 125 electrical model, 87 GPS, 458 link design, 96 mechanical model, 97 number, per plane, 433 prototypes, 87 subsystems, 57 systems using small earth stations, 117 telephone, 140 thermal model, 87 tool kit, (STK), 35 virtual packet, 384 Satellite orbital velocity, 19 Saturated output power (HPA), 115, 129 Scalar antenna horn, 366 Scaling attenuation, 126, 127, 325 Scan angle, phased array, 412 total, 414, 415 Scan loss, antenna, 416 Scanning beam (antenna), 78, 410 Scanning, off-axis, 412 Scintillation, ionospheric, 297, 312 tropospheric, 297 Score see Project Score Seam, ISL, across plane, 422 Selection factors, launch vehicles, 47 Selective availability, 460, 466, 476 Semilatus rectum, 22 Semimajor axis, 23 Semiminor axis, 23 Service, customer, base, 420 Shake and bake tests, 88 Shaped reflector, 367 Shannon, 275 bound, 275 -Hartley law, 275 Shear, wind, 302 Shift, Doppler, 49 Shuttle, space, 29, 394 Sidebands, FM, 159

533

Sidereal day, 25 Signal polar, 172 bipolar, 172 Signal-to-noise, 15, 100 in digital voice systems, 206 Simple attenuation model see SAM Sinc function see (sin x)x Sine wave modulation, 160 Single channel per carrier, 169, 225, 359 event upset (radiation), 398 parity, 278 stage to orbit, 43 Sino satellite, 11 SinoSat 1, 11 (Sin x)x, 173 SIRIO satellite, 313 Sirius 1, 12 Sirius Satellite Radio Inc, 455 Sky noise temperature, 332 increase in rain, 116, 122, 136 Skybridge, 394, 429, 431 Slots, orbital, GEO satellite, 84 Small earth stations, satellite systems, for, 117 Smart card, 447 Societe Europenne des Satellites, 5, 11, 442 Solar cells, 59 day, mean, 25 eclipse, 71 eclipse duration, 71 pressure, 60 sail, 70 sunspot cycle, 400 Solid state high power amplifier, 78, 226 Source, hot microwave (sun), 53 Soyuz rocket, 425 SS-Loral see space systemsloral Spacecom satellite communication, 12 Spacelab, 44 Space flight center, Kennedy, 44 center, Guiana (Kourou), 44 communications corp., 12

ind.qxd 28/08/02 19:33 Page 534

534

INDEX

Space (Continued) frontier, 17 inertial, 25 qualification, 87 shuttle, 29 transportation system, 43 wings, 17 Space systems-loral, 6 Sparklies see Demodulator, FM threshold Specific attenuation, 297, 318 performance, design to, 131 Spectrum, QPSK, 184 Speech compression, 207 Spin (spun) up, 62 Spin stabilized, 71 Split two-way (split-IP), 348 Spoofing, protocol, 286, 352 Spot beam, 80 antenna, 412 DBS-TV, 448 multiple, 80 SPOT satellite, 388 Spread spectrum see CDMA Spreading codes see CDMA Sputnik, 1 Sputnik 1, 388 Stable orbit, 17 point (in the orbit), 39 Stabilized Spin(ner), 60 Three-axis, 60 Staging (rockets), 42 Standard atmosphere, 207 noise temperature, 111 Star network (VSAT), 343, 348 Stationary coverage (NGSO), 430 Station-keeping box, 41 maneuvers, 60 thrusters, 60 Statistics, long-term, 313 Store-and-forward, 171 Stratified layers, 308 Stratiform rain, 298, 409 Streamer, rain, 302 Stutzman, 507 Subjective improvement, 205 (weighting) factor, 167

Sub-reflector, 344 Sub-satellite point, 31 Subsystems, satellite, 57 Sunset-sunrise orbit, 404 Sunspot (cycle), 401 Sun synchronous orbit, 390, 403 Sun transit outage, 53 Superbird A, B, 11 Super group, 493 Superheterodyne, 107, 108 Swedish space corporation, 12 Symbol, 235 bits per, 187 error probability, 191 error rate, 188 period, 187 rate, 172, 235 recovery circuit, 189 Synchronous orbit, sun, 390, 403 Synchronization, channel (TDM), 212 Syncom (first GEO satellite), 48 System, design examples, 131 design, NGSO constellation, 432 design, procedure, 131 design, to specific performance, 131 noise temperature, 116 performance, optimizing, PCS, 145 Systematic block code, 280 Switch beam, 78 matrix transponder, 75 T T carrier standards, USA, 211 TCP/IP, 287 TDD see time division duplexing TDM see time division multiplexing TDM-FDMA, 222 TDM-SCPC-FDMA, 226 TDMA see time division multiple access TDRSS satellite, 7, 44, 394 TIROS satellite, 388, 404 TMI communications, 12

TNC see terminal node controller TTC&M see telemetry, tracking, control and monitoring TV chrominance, 165 color, 166 color subcarrier, 167 DBS, 118 DSBSC, 166 hue, 166 monochrome, 165 saturation, color, 166 set-top boxes, 167 VSB, 166 TWTA see traveling wave tube amplifier Taurus rocket, 425 Teledesic, 394, 423, 430, 431 Telemetry, tracking, control, and monitoring, 42, 68 Telenor satellite, 12 Telesat Canada, 12 Telurometer (distance measuring device), 69 Telstar I and II, 3, 388 Temperature sky noise, 332 increase due to rain, 116, 122, 131 Tempo 2, 7 Terminal node controller, 171 Test point, 492 preflight (satellite), 88 shake and bake, 88 tone, 492 tone (FM), 158 visibility, 36 Thermal noise, 129 noise, sun, 53 problems, 87 Thin-route traffic, 355 Third order intermodulation, 226, 357 Thor 1, 2, 12 Three axis stabilized (satellite), 63 Threshold FM, 118 FM, extension, 169 Throughput, information, 421 Throw weight/mass, 47, 388

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INDEX

Thrusters arc jet, 63 ion, 63 orbit maneuvering, 63 Tilt angle, 327 Time universal, 28 zulu, 28 GMT, 467 UTC, 467 Time division duplexing, 421 multiplexing, 209, 222, 226, 357 Time of perigee, 26 Time division multiple access, 77, 223–246, 355 burst duration, 243, 245 burst transmission, 234 efficiency, 237 frame, 234 frame length, 234 guard bands, 225 guard times, 235 multifrequency, 223 preamble, 236 reference burst, 236 satellite switched, 246, 248 synchronization, 222 transmitter power, 243 Tipping bucket rain gauge, 312 Titan III, 45 Titan III, IV, 46 Total dosage (radiation), 398 path attenuation, 122 Tracking (of satellite), 68 Traffic centers, Internet, 432 Training sequence (equalizer), 182 Transducer, orthomode, 305 Transfer function, FM, pre/deemphasis, 163 orbit, geostationary, 46 TRANSIT (satellite), 388, 461 Transmission coefficient, fractional, 332 data, using analog FM channels, 170 digital, 172 digital, of analog channels, 201

digital, through bandlimited channel, 182 error free, 157 level, 491 (link) equation, 100 theory, basic, 100 Transponder, 59, 72, 75 backoff, 357 bandwidth, 221, 226 baseband processing, 247 bent pipe, 226 channelization, 76 DBS-TV, 119 HPA, 221 linear, 130 linearity, 77 receive noise temperature change, 130 switch matrix, 75 types, 129 Transversal equalizer (ISI), 182 Traveling wave tube amplifier, 78, 124, 226, 229 Trellis code, 285 Trilateration, 460 Troposphere, 480 Tropospheric scintillation, 297, 308 True anomaly, 25 Turbo code, 275, 285, 292 Turbulent mixing, 308 Turksat 1B, 1C, 12 Turk Telekom, 12 Tycho Brahe, 22 Tyuratam, 42 U ULPC see uplink power control UPC see ULPC USAT see ultra small aperture terminal UT see universal time, 28 UW see unique word Ultra small aperture terminal, 344 Unique word (TDMA), 191, 239, 362 correlator, 240 Universal time, 28 Unload, momentum wheel energy, 64 Unstable point (in the orbit), 39

535

Uplink CN, 131 CN budget, PCS, 144 carrier power, 123 design, 123 Ku-band, uplink, system design example, 133 power control, 126, 135, 148, 371 V V2 rocket, 404 VOR beacons, 461 VOW see voice order wire VSAT see very small aperture terminal VSAT hub station, 343 VSAT star network, 343 VSAT/WLL, 15, 346 VSB see vestigial sideband Van Allen radiation belts , 398, 402, 426 Vandenburg air force base, 44 Velocity angular, 25 satellite, 19 of light, 460 Very small aperture terminal, 107, 124, 298, 343 Vestigial sideband, 166 Virginia Tech, 439 Virtual Geosatellite, 431 Visibility test, 36 Viterbi decoding algorithm, 284, 451 Voice order wire, 240 Vostok, 388 W WAAS see wide area augmentation system WAC see world administrative conference (formerly WARC) WARC see world administrative radio conference WBFM see wide band FM WESTAR I, 4 WLL see wireless local loop Water vapor absorption, 308 Waveform, equation, FM, 158 Waves, characteristic, 310

ind.qxd 28/08/02 19:33 Page 536

536

INDEX

Weighting psophometric, 497 subjective improvement factor, 167 Wide area augmentation system, 466, 483 Wideband FM, 15, 157 Wind shear, 302 Window, protocol, 351 Wireless local loop, 15, 345 Word frame alignment, 209 unique, 191 World Administrative Radio Conference, 74

WorldSpace corporation, 8 World radio conferences, 15 X XPD see cross polarization discrimination XPD prediction, 326 XPI see cross polarization isolation X.25, X.75, 256, 351 X-33, X-34 (SSTO), 43 Y Y code, 460 Yaw axis, 63

Z z-axis intercept (satellite, antenna reference), 63 Zenit rocket, 425 Zenith, 32 Zenith attenuation, 307 Zero-ISI waveform, 170 Zero ISI filter, 175 non return to (line code), 172, 174 return to (line code), 172 Zone beam, 80

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