Radio frequency micromachined switches, switching networks, and phase shifters 9780815361435, 0815361432, 9781351021340, 1351021346

Table of contents :
Content: Behavioral study of micromachined contact switch --
Single-pole multi-throw (SPMT) MEMS switching networks --
Lateral MEMS switch and switching networks --
Micromachined microwave phase shifters --
Digital MEM switched line phase shifters --
DMTL phase shifter design using MAM capacitors and MEMS bridges --
Push-pull type micromachined phase shifter --
Reconfigurable micromachined phase shifter using push-pull actuators --
Multi-frequency MEMS phase shifters --
Reliability and power handling of MEMS switches and phase shifters --
MEMS 3-and 4-bit phase shifters using two back-to-back switching networks --
Digital MEMS phase shifters using combination of switched-line and DMTL topologies --
Packaging and integrated technologies.

Citation preview

Radio Frequency Micromachined Switches, Switching Networks, and Phase Shifters

Radio Frequency Micromachined Switches, Switching Networks, and Phase Shifters

Shiban Kishen Koul Sukomal Dey

CRC Press Taylor & Francis Group 52 Vanderbilt Avenue, New York, NY 10017 © 2019 by Taylor & Francis Group, LLC CRC Press is an imprint of Taylor & Francis Group, an Informa business No claim to original U.S. Government works Printed on acid-free paper International Standard Book Number-13: 978-0-8153-6143-5 (Hardback) This book contains information obtained from authentic and highly regarded sources. Reasonable efforts have been made to publish reliable data and information, but the author and publisher cannot assume responsibility for the validity of all materials or the consequences of their use. The authors and publishers have attempted to trace the copyright holders of all material reproduced in this publication and apologize to copyright holders if permission to publish in this form has not been obtained. If any copyright material has not been acknowledged please write and let us know so we may rectify in any future reprint. Except as permitted under U.S. Copyright Law, no part of this book may be reprinted, reproduced, transmitted, or utilized in any form by any electronic, mechanical, or other means, now known or hereafter invented, including photocopying, microfilming, and recording, or in any information storage or retrieval system, without written permission from the publishers. For permission to photocopy or use material electronically from this work, please access www.copyright.com (http:// www.copyright.com/) or contact the Copyright Clearance Center, Inc. (CCC), 222 Rosewood Drive, Danvers, MA 01923, 978-750-8400. CCC is a not-for-profit organization that provides licenses and registration for a variety of users. For organizations that have been granted a photocopy license by the CCC, a separate system of payment has been arranged. Trademark Notice: Product or corporate names may be trademarks or registered trademarks, and are used only for identification and explanation without intent to infringe.

Library of Congress CataloginginPublication Data Names: Koul, Shiban K., author. | Dey, Sukomal, author. Title: Radio frequency micromachined switches, switching networks, and phase shifters / Shiban Kishen Koul, Sukomal Dey. Description: Boca Raton, FL : CRC Press, Taylor & Francis Group, [2019] | Includes bibliographical references and index. Identifiers: LCCN 2019011927| ISBN 9780815361435 (hardback ; alk. paper) | ISBN 0815361432 (hardback ; alk. paper) | ISBN 9781351021340 (ebook) | ISBN 1351021346 (ebook) Subjects: LCSH: Radio frequency microelectromechanical systems. | Phase shifters. Classification: LCC TK7875 .K68 2019 | DDC 621.3815--dc23 LC record available at https://lccn.loc.gov/2019011927 Visit the Taylor & Francis Web site at http://www.taylorandfrancis.com and the CRC Press Web site at http://www.crcpress.com

To Mansi Narain, Grace Yoo Koul, Amendra Koul, Anant Koul and Swapan Ranjan Dey, Pranati Dey, Nibedita Dey, Nairick Dey

Contents Preface............................................................................................................................................ xiii Authors............................................................................................................................................xv 1. Introduction to Microelectromechanical Systems............................................................1 1.1 MEMS Overview............................................................................................................ 1 1.2 Transmission Line.......................................................................................................... 2 1.3 RF MEMS Switch........................................................................................................... 5 1.4 RF MEMS Phase Shifters..............................................................................................8 1.5 Major Applications of RF MEMS Phase Shifters..................................................... 10 1.6 Book Organization....................................................................................................... 11 References................................................................................................................................ 12 2. Behavioral Studies of Micromachined Contact Switches............................................. 15 2.1 MEMS Switch Design.................................................................................................. 15 2.2 Switch Model................................................................................................................ 15 2.3 Switch Design, Simulation and Measurements....................................................... 17 2.3.1 Switch Profile Analysis and Mechanical Resonance................................. 17 2.3.1.1 Switch Mechanical Measurements............................................... 20 2.3.2 Switch Pull-In Behavior Basics..................................................................... 20 2.3.3 Dynamic Behavior of the Switch.................................................................. 23 2.3.3.1 Actuation/Release Voltage Measurements................................. 28 2.3.3.2 Switching/Release Time Measurements..................................... 28 2.3.4 Switch Contact Resistance Basics................................................................. 29 2.4 Thermal Behavior of the Switch................................................................................ 31 2.4.1 Thermal Measurement...................................................................................34 2.5 RF Power Handling.....................................................................................................34 2.5.1 RF Power Handling Measurements............................................................. 38 2.6 Switch S-Parameter Analysis and Measurements.................................................. 39 2.7 Switch Linearity and Its Measurement..................................................................... 41 2.8 Conclusion.....................................................................................................................43 References................................................................................................................................43 3. Single-Pole Multithrow MEMS Switching Networks................................................... 47 3.1 High Isolation Single-Pole Double-Throw (SPDT) Switch..................................... 47 3.1.1 Series–Shunt Switch Design and Measurement......................................... 47 3.1.2 SPDT Switch Design and Measurement...................................................... 48 3.2 Single-Pole Four-Throw (SP4T) Switch Design and Measurement....................... 50 3.3 Compact SPMT Switching Networks........................................................................ 53 3.3.1 Single-Pole Single-Throw (SPST) Switch Design and Measurement....... 53 3.3.2 Single-Pole Three-Throw (SP3T) Switch Design and Measurement....... 55 3.3.3 Single-Pole Six-Throw (SP6T) Switch Design and Measurement............. 56 3.3.4 Single-Pole Seven-Throw (SP7T) Switch Design and Measurement....... 57 3.3.5 Single-Pole Eight-Throw (SP8T) Switch Design and Measurement........ 57 3.3.6 Single-Pole Ten-Throw (SP10T) Switch Design and Measurement......... 59 vii

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3.3.7 3.3.8 3.3.9

Single-Pole Eleven-Throw (SP11T) Switch Design and Measurement........59 Single-Pole Twelve-Throw (SP12T) Switch Design and Measurement.......59 Single-Pole Fourteen-Throw (SP14T) Switch Design and Measurement................................................................................................... 61 3.4 Equivalent Circuit Model of the SPMT Switch........................................................ 61 3.5 Key Design Features of MEMS SPMT Switches...................................................... 62 3.6 Third-Order Intermodulation Intercept Point (IIP3) Measurements of MEMS SPMT Switches................................................................................................ 62 3.7 Conclusion.....................................................................................................................64 References................................................................................................................................64 4. Lateral MEMS Switches and Switching Networks........................................................ 67 4.1 Design, Simulation and Measurement of the SPST Lateral Switch ..................... 67 4.1.1 Design of Lateral MEMS Switch................................................................... 67 4.1.2 Lateral Switch Characterization................................................................... 69 4.1.2.1 SPST Lateral Switch Characterization.......................................... 69 4.1.2.2 SPDT Lateral Switch Characterization......................................... 70 4.1.2.3 SP3T Lateral Switch Characterization.......................................... 70 4.1.2.4 SP4T Lateral Switch Characterization.......................................... 71 4.1.2.5 SP6T and SP7T Switch Characterizations.................................... 72 4.2 Conclusion..................................................................................................................... 72 References................................................................................................................................ 74 5. Micromachined Microwave Phase Shifters.....................................................................77 5.1 Introduction..................................................................................................................77 5.2 Applications.................................................................................................................. 79 5.3 Technologies................................................................................................................. 79 5.4 Theoretical Background..............................................................................................80 5.5 Classifications............................................................................................................... 82 5.5.1 Reflection-Type Phase Shifter....................................................................... 82 5.5.2 Switched-Line Phase Shifter......................................................................... 87 5.5.3 Loaded-Line Phase Shifters........................................................................... 89 5.5.4 Low-Pass/High-Pass Network Phase Shifter............................................. 92 5.5.5 Distributed MEMS Transmission Line (DMTL) Phase Shifter................ 93 5.6 Conclusion..................................................................................................................... 95 References................................................................................................................................ 97 6. Digital MEMS Switched-Line Phase Shifters............................................................... 101 6.1 Introduction................................................................................................................ 101 6.2 Switched-Line Phase Shifter Design....................................................................... 101 6.3 MEMS Switch Design................................................................................................ 102 6.4 MEMS Switch Performance Analysis..................................................................... 102 6.5 MEMS Primary Bit Phase Shifter Fabrication and Measurements..................... 105 6.6 5-Bit Switched-Line Phase Shifters Fabrication and Measurements.................. 107 6.6.1 5-Bit Phase Shifter Using SW1: Phase 1..................................................... 107 6.6.2 5-Bit Phase Shifter Using SW2: Phase 2..................................................... 110 6.7 Life Cycle of 5-Bit Phase Shifters............................................................................. 112 6.8 Conclusion................................................................................................................... 114 References.............................................................................................................................. 114

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7. DMTL Phase Shifter Design Using MAM Capacitors and a MEMS Bridge.......... 117 7.1 Introduction................................................................................................................ 117 7.2 Unit Cell Phase Shifter Design and Modeling....................................................... 117 7.3 Fabrication and Measurements................................................................................ 121 7.3.1 Unit Cell Phase Shifter Measurements and Results................................ 121 7.3.2 Primary Phase Bits Measurement Results................................................ 124 7.3.3 Complete 5-Bit Phase Shifter Measurements and Results...................... 125 7.4 Reliability Measurements and Results................................................................... 127 7.4.1 Reliability Measurements of the MEMS Bridge....................................... 127 7.4.2 Reliability Measurements of the 5-Bit Phase Shifter............................... 129 7.5 Conclusion................................................................................................................... 130 References.............................................................................................................................. 130 8. Push–Pull Type of Micromachined Phase Shifters..................................................... 131 8.1 Introduction................................................................................................................ 131 8.2 Operating Principle of the Push–Pull Actuator.................................................... 131 8.2.1 Analysis of Bridge Pull-In Voltage Using Quasistatic Approximation.............................................................................................. 132 8.2.2 Analysis of the Push–Pull Actuator under Step and Modulated Voltage Responses........................................................................................ 137 8.3 Modeling of the Push–Pull Bridge.......................................................................... 139 8.4 DMTL Unit Cell Phase Shifter Design and Modeling.......................................... 141 8.5 Fabrication................................................................................................................... 143 8.6 Measurements............................................................................................................ 144 8.6.1 Mechanical Measurements.......................................................................... 144 8.6.2 Electrical Measurements.............................................................................. 147 8.6.3 S-Parameter Measurements of the Unit Cell Phase Shifter.................... 148 8.6.4 Measurements of the Complete Push–Pull Analog Phase Shifter........ 148 8.7 Discussion of Push–Pull Bridge Performances..................................................... 151 8.8 Conclusion................................................................................................................... 151 References.............................................................................................................................. 152 9. Reconfigurable Micromachined Phase Shifters Using Push–Pull Actuators......... 155 9.1 Introduction................................................................................................................ 155 9.2 Design and Analysis of a Phase Shifter Using a Push–Pull Actuator............... 156 9.3 Design and Analysis of the Push–Pull Voltages and a Travel Range................ 157 9.4 Design of Primary Cell Phase Shifters................................................................... 160 9.5 Design of Complete 5-Bit Phase Shifter.................................................................. 162 9.6 Fabrication Process Details....................................................................................... 164 9.7 Measurements of the Push–Pull Actuator and Primary Phase Bits................... 166 9.7.1 Mechanical Measurement of the Push–Pull Actuator............................ 166 9.7.2 Vpull, Vpush and Capacitance Measurements............................................... 168 9.7.3 Response Time Measurements of the Push–Pull Actuator.................... 169 9.7.4 S-Parameter Measurements........................................................................ 169 9.8 Measurements of the Complete 5-Bit Phase Shifter.............................................. 174 9.9 Power Handling and Reliability Measurements................................................... 175 9.10 Reliability Measurements on the 5-Bit Phase Shifter........................................... 178 9.11 Conclusion................................................................................................................... 181 References.............................................................................................................................. 181

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10. Multifrequency MEMS Phase Shifter Bank.................................................................. 183 10.1 Introduction................................................................................................................ 183 10.2 Proposed Topology for the Wideband Digital Phase Shifter.............................. 184 10.3 Wideband Digital Phase Shifter Design and Characterization of the MEMS Switches: Phase 1.......................................................................................... 184 10.3.1 Design, Analysis and Measurements of the SPST Switch...................... 184 10.3.2 Design, Analysis and RF Measurements of the SP4T Switch................. 186 10.3.3 RF Performance Improvement on the SP4T Switch................................. 189 10.4 Design and Characterization of the 4-Bit Phase Shifter Bank: Phase 2............. 190 10.4.1 Design, Analysis and Measurements of the Four 4-Bit DMTL Phase Shifters................................................................................................ 190 10.4.2 Design and Measurements of the Complete 4-Bit Phase Shifter Bank................................................................................................... 192 10.5 Conclusion................................................................................................................... 193 References.............................................................................................................................. 195 11. Reliability Analysis of MEMS Switches and Phase Shifters.................................... 197 11.1 Introduction................................................................................................................ 197 11.2 Phase Shifter Design Topology................................................................................ 197 11.3 2-Bit and 1-Bit Phase Shifter Design and Analysis: Phase 1................................ 199 11.4 2-Bit and 1-Bit Phase Shifter Measurements: Phase 1........................................... 200 11.5 Design and Measurements of the 5-Bit Phase Shifter: Phase 1........................... 203 11.6 Reliability Measurements of the SPST and SP4T Switches: Phase 1................... 204 11.6.1 Temperature Stability Measurements on the MEMS Switch................. 204 11.6.2 Power Handling Measurements on the MEMS Switch........................... 207 11.6.3 Reliability Measurements of MEMS Switches.......................................... 209 11.6.4 Creep Measurements on the MEMS Switch............................................. 210 11.7 Reliability Measurements of the Phase Shifter: Phase 1...................................... 211 11.7.1 Phase Shifter Testing on a Chip and within a Module........................... 211 11.7.2 Phase Shifter Reliability Measurements under Different Temperatures................................................................................................. 212 11.7.3 Phase Shifter Reliability Measurements with RF Power........................ 212 11.8 Design Modification and Measurements of the Phase Shifter: Phase 2............ 214 11.9 Reliability Measurements on Switches: Phase 2.................................................... 217 11.9.1 Temperature Measurements of the MEMS Switch.................................. 219 11.9.2 Power Handling Measurements on the MEMS Switches....................... 219 11.9.3 Reliability Measurements of MEMS Switches with RF Power.............. 219 11.9.4 Reliability Measurements of MEMS Switches with RF Power and Temperature........................................................................................... 220 11.10 Reliability Measurements of the Phase Shifter: Phase 2...................................... 220 11.10.1 Phase Shifter Reliability Measurements with RF Power........................222 11.10.2 Phase Shifter Reliability Measurements with RF Power and Temperature...................................................................................................222 11.10.3 Phase Shifter Testing under Prolonged Actuation.................................. 223 11.11 Qualification Testing of the Phase Shifter: Three-Axis Vibration...................... 224 11.12 Failure Analysis of the 5-Bit MEMS Phase Shifter................................................ 224 11.13 Design Guidelines for a Reliable MEMS 5-Bit Phase Shifter with Alternative Topology................................................................................................. 226 11.14 Conclusion................................................................................................................... 227 References.............................................................................................................................. 227

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12. MEMS 3-Bit and 4-Bit Phase Shifters Using Two Back-to-Back Switching Networks............................................................................................................................... 229 12.1 Introduction................................................................................................................ 229 12.2 3-Bit Phase Shifter Design and Measurements..................................................... 230 12.2.1 Design and Modeling of the 3-Bit Phase Shifter...................................... 230 12.2.2 RF Measurements of the 3-Bit Phase Shifter............................................. 231 12.2.3 Reliability Measurements of the 3-Bit Phase Shifter............................... 232 12.2.4 Device Testing under Prolonged Actuation..............................................234 12.3 Design and Measurements of the 4-Bit Phase Shifter Using SP16T Switching Networks..................................................................................................234 12.3.1 Design of the SP16T Switch.........................................................................234 12.3.2 Measurements of the SPST and SP16T Switches...................................... 236 12.3.3 Design and Modeling of the 4-Bit Phase Shifter...................................... 238 12.3.4 RF Measurements of the 4-Bit Phase Shifter............................................. 240 12.4 Conclusion................................................................................................................... 241 References.............................................................................................................................. 243 13. Digital MEMS Phase Shifters Using Combinations of Switched-Line and DMTL Topologies........................................................................................................ 245 13.1 Introduction................................................................................................................ 245 13.2 Proposed Design Topology of the Phase Shifter................................................... 245 13.3 MEMS Switch Design and Measurements............................................................. 247 13.3.1 Single MEMS Switch Design and Measurements.................................... 247 13.3.2 SP4T MEMS Switch Design and Measurements...................................... 249 13.3.3 SP8T MEMS Switch Design and Measurements...................................... 249 13.4 Reliability Measurements of MEMS Switches....................................................... 252 13.4.1 Reliability Measurements with 0.1–1 W of RF Power.............................. 252 13.4.2 Reliability Measurements at 50°C–85°C with 0.1 W of Power..............................................................................................................254 13.5 Phase Shifters Design and Measurements............................................................. 255 13.5.1 Design and Simulation of 3-Bit and 4-Bit Phase Shifters........................ 255 13.5.2 RF Measurements of the 3-Bit and 4-Bit Phase Shifters.......................... 258 13.5.3 Hot- and Cold-Switching Reliability Measurements............................... 258 13.5.4 Phase Shifter Testing at 50°C–85°C with 0.1–0.5 W of Power.............................................................................................................. 259 13.6 Device Responses within Low-Cost Packaging.................................................... 260 13.7 Design Guidelines of the Proposed Device............................................................ 260 13.8 Conclusion................................................................................................................... 264 References.............................................................................................................................. 265 14. Packaging and Integration Technologies....................................................................... 269 14.1 Introduction................................................................................................................ 269 14.2 1-Level Packaging...................................................................................................... 269 14.3 0-Level Packaging...................................................................................................... 270 14.3.1 Thin-Film Packaging.................................................................................... 270 14.3.2 Chip/Die Capping........................................................................................ 272 14.3.2.1 Fabrication of GaAs Microcaps................................................... 273 14.3.2.2 Fabrication of Glass Microcaps................................................... 273 14.3.2.3 Cap Attachment............................................................................. 273 14.3.2.4 Results and Discussion................................................................. 275

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14.4 LTCC Packaging and Integrated Modules............................................................. 278 14.5 Other Bonding Mechanisms ................................................................................... 281 14.6 Conclusion................................................................................................................... 282 References.............................................................................................................................. 283 Index��������������������������������������������������������������������������������������������������������������������������������������������� 285

Preface Radio frequency microelectromechanical systems (RF MEMS) switches, switching networks and phase shifters are widely employed in electronic beam-steering systems that are based on phased antenna arrays. MEMS technology offers much lower insertion loss, good matching, excellent phase accuracy, high linearity over a larger bandwidth and lower power consumption compared to other solid-state technologies. Higher IIP3 and P1dB of MEMS switches demonstrate higher signal power levels in the transmit/receive (T/R) system. Moreover, one more primary characteristic of this technology is the compatibility with traditional MMIC/CMOS circuits for system-level applications. CMOS-based phase shifters are compact in size, but to compensate for the loss and noise, these active phase shifters require a T/R module at each antenna element. This greatly increases the cost of such phased arrays. Ferrite-based phase shifters have good performance in terms of power handling and lower insertion loss, but these are bulky and not easy to integrate with other circuit elements, and are more expensive to fabricate than MEMS counterparts. Nevertheless, in terms of batch production MEMS technology is superior to others. In this book, an attempt has been made to discuss different micromachined switching devices considering the present state-of-the-art utilizing the coplanar waveguide (CPW) transmission line. Different topologies of MEMS switches have been discussed considering reliability for long-run applications. The book provides major design guidelines for the development of MEMS-based digital phase shifters with extensive measurement data. It takes multiple iterations and extensive characterizations to conclude with a reliable MEMS digital phase shifter, and these aspects are given prime attention in this book. The devices reported in the book are primarily based on alumina substrate (εr = 9.8) using the surface micromachining process. Devices are made on a 635 µm alumina substrate after RCA cleaning of the wafer. In brief, after sputtering and patterning (using liftoff) of 70 nm titanium tungsten (TiW), a 0.5 µm SiO2 layer is deposited and patterned on the TiW. Then, 1 to 2 μm of gold is electroplated to form a fixed electrode and CPW lines. A 0.5 to 0.7 µm of SiO2 is deposited as a dielectric layer on the fixed electrode. Later, 2.5 μm of spin-coated polyimide (PI) is coated, and anchor holes and dimple openings of 1 µm are made on the PI layer. A gold seed layer for electroplating is sputtered on the PI layer, and a 2–4 µm beam is electroplated. Finally, the sacrificial layer is released using the CO2 criticalpoint drying (CPD) process at 350°C in the oven and etched using EKC 265. The yield is limited by the yield of the CPD process as well as the type of geometries. Generally, the average yield is more than ~80% in the fabrication phase. The book is divided into 14 chapters covering different aspects of MEMS-based switching networks and phase shifters. The main focus is on reliability or life-cycle improvement techniques that do not compromise other performances such as microwave (loss, matching, linearity, phase error, group delay), microelectromechanical (actuation voltage, switching time, mechanical resonance) and qualification (power handling, temperature stability). Different digital phase shifters operating in single frequency and multifrequencies (reconfigurable) are also discussed in detail. Chapter 1 presents a basic introduction to RF MEMS. A brief description of CPW transmission lines is given in this chapter. Different types of MEMS switches, switching networks and phase shifters are discussed followed by applications in phased array systems. Chapter 2 starts with the design, development and complete characterization of xiii

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a metal-contact MEMS switch. The motivation is to observe the switch functionalities in terms of mechanical behavior, electrical behavior, transient analysis, linearity, power handling, temperature, S-parameter and cold-switching reliability. Chapter 3 discusses the design and implementation of different MEMS single-pole multithrow (SPMT) switching networks starting from single-pole double-throw (SPDT) to single-pole fourteen-throw (SP14T) configurations. Chapter 4 presents the design and development of dielectric-less lateral MEMS switches and switching networks. Chapter 5 presents the theory, design and analysis of different types of microwave MEMS phase shifters. Chapter 6 describes Ku-band conventional switched-line phase shifters using MEMS in-line metal-contact switches. The complete design strategy, fabrication of the individual primary phase bits and performance optimization of two 5-bit phase shifters are discussed in this chapter. Chapter 7 gives detailed analysis of a distributed transmission line-based 5-bit phase shifter using MEMS bridges and metal–air–metal (MAM) capacitors. Chapter 8 presents the design, analysis and measurement of a push–pull MEMS actuator. In addition to this, the utility of the push–pull configuration is validated and tested on an analog phase shifter configuration. Chapter 9 attempts to address the design and development of a reconfigurable 5-bit DMTL phase shifter using the push–pull topology, with detailed analysis and experimental investigation. This 5-bit phase shifter is capable of providing 32 phase states with constant resolution over 10–25 GHz. Initially, to validate frequency reconfiguration, all primary unit cells are fabricated and tested. Later, a complete 5-bit phase shifter is developed, tested and discussed in detail keeping in view the microwave and reliability performances. Chapter 10 describes multifrequency phase shifters utilizing MEMS switching networks and DMTL phase shifters. A complete phase shifter bank is reported here that operates from 17 GHz to 60 GHz. Chapter 11 reports the design and development of a 5-bit true-time-delay (TTD) phase shifter with low loss and desirable phase shift within a smallest possible area. This phase shifter drastically reduces the number of switches and thus improves the reliability of the overall device. Two phases of design, fabrication and characterization are performed and discussed, which substantially improve the overall performance of the 5-bit phase shifter compared to one another. Chapter 12 discusses the design and implementation of a MEMS-based digital phase shifter using two standalone back-to-back SPMT switching networks. Chapter 13 discusses a novel topology of digital phase shifter design using a combination of switched-line and DMTL phase shifters. The complete design methodology and characterization are discussed in this chapter. Chapter 14 describes the packaging and integrated technologies of MEMS-based devices and components.

Authors Shiban Kishen Koul earned a BE degree in electrical engineering from the Regional Engineering College, Srinagar, Jammu and Kashmir, India, in 1977; an MTech in radar and communication engineering in 1979; and a PhD in microwave engineering in 1983 from the Indian Institute of Technology (IIT), Delhi. He is the Dr. R.P. Shenoy Astra Microwave Chair Professor at IIT Delhi. He served as deputy director (Strategy & Planning) at IIT Delhi from 2012 to 2016, and presently he holds the position of deputy director (Strategy & Planning, International Affairs and R&D) at the Indian Institute of Technology Jammu. He is also the chairman of M/S Astra Microwave Products Limited, Hyderabad, a major company involved in the development of RF and microwave systems in India. His research interests include RF MEMS, high frequency wireless communication, microwave engineering, microwave passive and active circuits, device modeling, millimeter wave integrated circuit design, and reconfigurable microwave circuits including antennas. Koul has successfully completed 34 major sponsored projects, 52 consultancy projects and 57 technology development projects. He is author/coauthor of 403 research papers, 8 state-ofthe-art books, and 3 book chapters. He holds 10 patents and 6 copyrights. Koul is a Fellow of the Institution of Electrical and Electronics Engineers, USA (IEEE), Fellow of the Indian National Academy of Engineering (INAE), and Fellow of the Institution of Electronics and Telecommunication Engineers (IETE). He is the chief editor of the IETE Journal of Research and associate editor of the International Journal of Microwave and Wireless Technologies. He has delivered more than 266 invited technical talks at various international symposia and workshops. He is currently a serving MTT-S ADCOM member and a member of IEEE MTT Society’s Technical Committees on Microwave and Millimetre Wave Integrated Circuits (MTT-6) and RF MEMS (MTT-21); a member of the India Initiative team of IEEE MTT-S, Advisor Education Committee; Membership Services Regional co-coordinator Region-10; member of Sight Ad Hoc Committee MTT-S; and MTT-S Speaker bureau lecturer. He served as a distinguished microwave lecturer of IEEE MTT-S for 2012–2014 and distinguished microwave lecturer-emeritus of IEEE MTT-S in 2015. Koul is recipient of the Gold Medal by the Institution of Electrical and Electronics Engineers Calcutta (1977); S.K. Mitra Research Award (1986) from the IETE for the best research paper; Indian National Science Academy (INSA) Young Scientist Award (1986); International Union of Radio Science (URSI) Young Scientist Award (1987); the top Invention Award (1991) of the National Research Development Council for his contributions to the indigenous development of ferrite phase shifter technology; VASVIK Award (1994) for the development of Ka-band components and phase shifters; Ram Lal Wadhwa Gold Medal (1995) from the IETE; Academic Excellence award (1998) from the Indian government for his pioneering contributions to phase control modules for Rajendra Radar; Shri Om Prakash Bhasin Award (2009) in the field of electronics and information technology; teaching excellence award (2012) from IIT Delhi; award for contributions made to the growth of smart material technology (2012) by the ISSS, Bangalore; VASVIK Award (2012) for the contributions made to the area of information, communication technology (ICT); M.N. Saha Memorial Award (2013) from the IETE for the best application-oriented research paper; and IEEE MTT Society Distinguished Educator Award (2014).

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Sukomal Dey earned a BTech degree in electronics and communication engineering from the West Bengal University of Technology, Kolkata, India, in 2006; an MTech degree in mechatronics engineering from the Indian Institute of Engineering Science and Technology (IIEST), India; an MTech dissertation (one year) from the Central Electronics Engineering Research Institute, Pilani, India, in 2009; and a PhD in microwave engineering in 2015 from the Indian Institute of Technology (IIT), Delhi. From August 2015 to July 2016 Dey worked as a project scientist at Industrial Research and Development (IRD) Centre, IIT Delhi. He also worked on a collaborative research project supported by Synergy Microwave Corp., New Jersey, during the same period. From August 2016 to June 2018, he worked at the Radio Frequency Microsystem Lab (RFML), Institute of Nano Engineering and Micro Systems, National Tsing Hua University, Taiwan, as a postdoctorate research fellow. Since June 2018 he has worked as an assistant professor in the Department of Electrical Engineering, Indian Institute of Technology, Palakkad. He is the recipient of the postgraduate student award from the Institute of Smart Structure and System (2012), Bangalore; first prize for best poster on MEMS switches and phase shifters in Science Day Celebrations at IIT Delhi (2015); Best Industry Relevant PhD Thesis Award from the Foundation for Innovation in Technology Transfer (2016); and Distinction in Doctoral Thesis from the PhD thesis review committee, IIT Delhi (2016); and the Postdoctoral Research Fellow scholarship from Ministry of Science and Technology, Taiwan (2016). He has been awarded an Early Career Research (ECR) grant from the Science and Engineering Research Board (SERB), India (2019). His research interests include development of RF MEMS devices and components for future RF front ends.

1 Introduction to Microelectromechanical Systems

1.1 MEMS Overview Radio frequency (RF), microwave and millimeter wave circuits have been conceptualized using microelectromechanical systems (MEMS) as early as the 1990s. MEMS devices can be fabricated in large numbers on a single chip (batch production) using conventional integrated circuit (IC) and semiconductor processing technologies. The IC processing technique has been well established using surface and bulk micromachining processes since the 1980s. Advances in these processes over the last decade have accelerated the development of MEMS devices in almost all areas of engineering. The key advantages are smaller size, low cost and improved performance compared to other conventional alternatives [2]. The journey of MEMS structures started with relatively simple acceleration, pressure and temperature sensors. Since then, MEMS technology has been proven to be applicable in many areas of engineering such as microfluidics, pneumatic, RF, optical and biomedical. Figure 1.1 shows some examples of MEMS devices [3,27]. MEMS devices that mechanically respond to external electromagnetic signals are broadly called radio frequency MEMS (RF MEMS). RF MEMS devices are usually fabricated employing low-temperature micromachining processes, and, thus, it makes them compatible with postprocessing semiconductor circuit integration such as CMOS, GaAs and SiGe. Different types of microwave components like micromechanical resonators and filters, phase shifters, antennas, micromachined electromagnetic structures, RF MEMS

FIGURE 1.1 SEM images of MEMS (a) micromirror by Texas Tech University and (b) RF switch by UCSD. (From K.-J. Koh, and G. M. Rebeiz, A 6-18 GHz 5-bit active phase shifter, in IEEE MTT-S International Microwave Symposium Digest, Anaheim, CA, May 2010, pp. 792–795; and M. J. W. Rodwell et al., IEEE Trans. Microw. Theory Tech., vol. 39, pp. 1194–1204, July 1991.)

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RF Micromachined Switches, Switching Networks, and Phase Shifters

switches and varactors are fabricated employing the mechanical movement of the structure. Moreover, all the RF MEMS components have the ability to replace the conventional, costly and large off-chip discrete components by enabling integrated solutions that can be batch fabricated. This technology is rapidly growing due to its impressive figure of merit performance and a promising commercial market. Presently, research in RF MEMS is focused toward the development of novel devices for commercial applications such as 5G, microsatellites, reconfigurable integrated circuits and lightweight integrated subsystems. Low-cost packaging, reliability of the fabricated circuits, integration with existing technologies and state-of-the-art performance are some of the key challenges.

1.2 Transmission Line Miniaturization of microwave and millimeter wave transmission lines and their implementation in radio frequency micromachining technologies based on silicon or quartz is accepted as a quite promising research domain in terms of low loss and compactness compared to traditional techniques [4]. Among the various families of transmission line configurations available, and well known for decades [5], microfabrication technologies are particularly suited for planar devices. Most RF MEMS devices are implemented on the coplanar waveguide (CPW) because it is uniplanar in nature, which leads to one of the best choices for any device due to fabrication simplicity and easier integration with other components. The microstrip transmission line also attracts attention in micromachining for high-performance passive circuits. Figure 1.2a and b show 3D schematic views of CPW and microstrip transmission lines, respectively. In the former, a central metallization acts as the RF signal line, while two wider metalized patches are meant to be reference ground planes for the traveling RF signal. The central line and ground planes are separated by a gap, and all the metal layers lie on the same side of the substrate. As the RF signal propagates along the waveguide, the electromagnetic field is confined between the central line and the ground planes, partially through the dielectric material underneath the metal layers and partially through the air above them. Very often, a thin insulating layer is deposited on the substrate prior to the electroplating/evaporation of the CPW itself. This helps to reduce dielectric losses due to the substrate. In the microstrip configuration, however, the RF signal line is placed on top of the substrate, while a unique reference ground plane is metalized on the opposite face of the wafer. In this case, as

RF Ground

Signal

RF Ground

Signal Substrate

Substrate

(a)

Ground

(b)

FIGURE 1.2 Schematic 3D view of (a) coplanar waveguide (CPW) transmission line and (b) microstrip transmission line.

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the RF signal propagates along the waveguide, the electromagnetic field is mainly confined within the substrate, between the two metal layers (signal line and ground plane). A basic lumped element network description useful to model the RF behavior of CPW and microstrip transmission lines is proposed by Mahmoud [5] and the same is depicted in Figure 1.3. Across the waveguide input and output ports P1 and P2, a resistance and inductance are inserted, representing the series resistive (Rline) and inductive (Lline) contributions of the metal RF line, respectively. On the other hand, the shunt-to-ground capacitance and resistance model the capacitive coupling (Cgnd) and the resistive losses (Rgnd) between the RF line and the ground plane(s), respectively, through the substrate material and through air. The value of the resistive and reactive components in Figure 1.3 are correlated to the physical properties of the transmission line, and can be parameterized in quite a straightforward fashion, in order to account for the most relevant geometrical features of the CPW or microstrip transmission line such as the length, gap and substrate thickness [7]. The RF behavior of a typical CPW/microstrip transmission line in terms of scattering parameters (S-parameters) versus frequency [6] is shown in Figure 1.4. The curves result from the simulation of a CPW by means of a finite element method (FEM)

-20

0

-30

-0.2

-40

-0.4

-50

-0.6

-60

-0.8

-70

S21 (db)

S11 (db)

FIGURE 1.3 Equivalent lumped element network of a CPW/microstrip transmission line. P1 and P2 are the input/output ports, respectively.

-1 0

10

20

30

40

50

60

Frequency (GHz) FIGURE 1.4 Typical S-parameters versus frequency characteristic of a CPW/microstrip line concerning reflection (S11 at P1) and transmission losses through the line (S21 from P1 to P2).

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RF Micromachined Switches, Switching Networks, and Phase Shifters

software tool, in the frequency range from DC to 60 GHz. The S11 parameter indicates the fraction of RF signal reflected at the input port of the CPW; and as it is small over the whole frequency range (better than –22 dB), most of the RF signal flows into the waveguide. The S11 and S22 (reflection at the input and output ports, respectively) are particularly suited to provide an indication of matching between the characteristic impedance of the RF source and of the transmission line. Low values of S11/S22 mean good impedance matching. The S21 parameter indicates the amount of RF power reaching the output port of the CPW. Since its worst value (~–0.9 dB) is quite close to 0 dB (i.e., ideal zero losses), the attenuation of the RF signal introduced across the waveguide is limited overall across the analyzed frequency span. RF MEMS devices can be fabricated using two different micromachining processes: bulk micromachining and surface micromachining. Bulk micromachining defines structures by selectively etching inside a substrate. Whereas surface micromachining creates structures on top of a substrate, bulk micromachining produces structures inside a substrate. Bulk micromachining starts with a silicon wafer or other substrates that are selectively etched, using photolithography to transfer a pattern from a mask to the surface. Surface micromachining builds microstructures by the deposition and etching of different structural layers on top of the substrate. Besides the exploitation of a typical surface micromachining step like selective deposition of thin metal films, additional techniques started to be explored with the aim of improving the RF characteristics of miniaturized CPWs and microstrip transmission lines. For instance, shallow tranches were etched in the gap between the RF signal line and the reference ground planes in order to reduce the losses due to penetration of the electromagnetic field through the substrate, as reported by Yang et al. [8]. In other examples, bulk micromachining was used to remove most parts of the silicon substrate, yielding CPWs suspended above a thin membrane, resulting in a significant reduction of losses and parasitic coupling effects, as discussed by Farrington and Iezekiel, and Shi et al. [9,10]. There is considerable interest in developing reconfigurable and tunable integrated RF MEMS front ends using high dielectric constant substrates such as Si, alumina and GaAs [1–10]. In addition, due to the high dielectric constant of the substrate, there is a possibility of excitation of surface waves. These problems can be overcome by introducing a small air gap between the dielectric substrate and the ground plane, as shown in Figure 1.5. The air gap can be easily created using the bulk micromachining technique. The advantages of creating a small air gap beneath the signal conductor are (a) a lower effective dielectric constant, hence wider circuit dimensions; (b) ease of fabrication and relaxed dimensional tolerances; (c) lower attenuation; (d) enhanced radiation efficiency in

FIGURE 1.5 Cross section of a micromachined microstrip line.

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the case of antennas; and (e) elimination of surface waves. Using the micromachined lines, it is possible to design and develop high-performance passive circuits and printed micromachined antennas at microwave and millimeter wave frequencies.

1.3 RF MEMS Switch The most important MEMS device used at RF, microwave and millimeter wave frequencies is the switch. The MEMS switch is a miniature device that uses mechanical movements to achieve ON or OFF states in a transmission line. There are several ways to do these mechanical movements or actuations such as electrostatic, thermal, magnetostatic or piezoelectric. Electrostatic actuation is widely used due to its simplicity among all these actuation mechanisms. Although RF MEMS switches have disadvantages such as medium switching speed (1–100 μs) and lower power handling capability (40 0.05–0.2 1–6 W 8–20 >108

±3–5 3–20 3–20 1–100 ns >35 0.3–1.2 109

3–5 ~0 0.05–0.1 1–100 ns 15–25 0.4–2.5 109

MEMS Bridge

Bridge height

Ground

Ground

G

W

G

(a)

L

C R

(b) FIGURE 1.6 (a) Suspended MEMS bridge in shunt configuration over a CPW transmission line and (b) its equivalent circuit model.

metal-contact switches, including Omron [12,13], Radant MEMS [14,15], Teravicta [16], RFMD [17], Hitachi/Michigan [18,19], Rockwell Scientific [20,21], Motorola [22], NEC [23] and from other research groups [24,25]. To date, the Omron switch and the Radant MEMS switch are the only two successful commercial products satisfying the customer needs for large cycles of operations. Reliability or life cycle becomes more critical when multiple switches are actuating simultaneously in a multibit digital phase shifter (>3 bit) or switching networks. A few successfully developed RF MEMS capacitive and metal-contact switches are shown in Figure 1.8.

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Signal

(a) L

R C

(b) FIGURE 1.7 (a) Suspended MEMS bridge in series configuration over a CPW transmission line and (b) its equivalent circuit model.

FIGURE 1.8 Examples of RF MEMS devices. Clockwise from top left: Raytheon, University of Michigan, Analog Devices, and Lincoln Laboratories. (From A. Q. Liu, RF MEMS Switches and Integrated Switching Circuits, Springer, March 2010.)

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RF Micromachined Switches, Switching Networks, and Phase Shifters

An RF MEMS-based metal-contact switch covers a wide area of applications. Automatic test equipment (ATE) and wideband electronic instrumentation require an RF MEMS metal-contact switch because of its broadband performance and low loss with good isolation characteristics. A satellite system requires a metal-contact switch for its compactness and weight saving. A metal-contact switch and its single-pole N-throw (SPNT) switching networks are also very useful in base stations and microwave communication equipment for telecommunications networks. RF MEMS metal-contact switches show a great promise in defense applications like wideband transceivers and phased array systems. The list is not very exhaustive, as there are many other application areas where RF MEMS metalcontact switches can prove their usefulness to improve overall system functionalities, such as in low-power switching networks, switched capacitors in voltage-controlled oscillators (VCOs), switched inductors, and broadband tunable filters and filter banks.

1.4 RF MEMS Phase Shifters Phase shifters are widely employed in electronic beam-steering systems that are based on phased antenna arrays. MEMS technology offers much lower insertion loss, good matching, excellent phase accuracy, high linearity over a larger bandwidth and lower power consumption compared to other solid-state technologies. Moreover, one more very primary characteristic of this technology is the compatibility with traditional MMIC/CMOS circuits for system level applications. Ferrite-based phase shifters have good performance in terms of power handling, but cannot be easily integrated and are more expensive to fabricate than MEMS technology. Nevertheless, in terms of batch production, MEMS technology is superior to others. Phase shifters are primarily categorized as analog or digital type. Phase shift can be changed continuously with the bias in an analog phase shifter that provides larger phase shift with higher switching speeds and good phase resolution. The digital phase shifter exhibits excellent phase accuracy in relatively wider bandwidth with high input power levels [26–28]. To improve the performance of a microwave phase shifter over the band of interest, most relevant parameters are operating bandwidth, switching speed, differential phase difference (ΔΦ), insertion loss, return loss and group delay. Selection of transmission line in a phase shifter is an important requirement for its optimum performance. For minimum cost, several phase shifters use microstrip transmission line; however, a lowloss stripline or a suspended stripline can also be used. CPW is uniplanar in nature, which leads to one of the best choices in phase shifter for fabrication simplicity and thus is easier to integrate with other components [29]. Different types of digital phase shifters have been implemented during the past decade using MMIC and CMOS technology [30–35]. A 5-bit 5–20 GHz CMOS phase shifter was recently reported with an insertion loss of 27 dB with an input/output return loss of better than 12 dB [34]. Although CMOS-based active phase shifters are compact in size, the array will require a transmit/receive (T/R) module (with power amplifier and low-noise amplifier) at each antenna element to compensate for the phase shifter loss and noise. This will greatly increase the cost of such phased arrays. On the other hand, one T/R module connected to multiple low-loss phase shifters in phased arrays will result in lower component count and thus reduction in overall cost of the array. The most widely discussed MEMS phase shifters are true-time-delay (TTD) and distributed MEMS transmission line (DMTL) phase shifters. Different MEMS single-pole

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FIGURE 1.9 Photomicrographs of (a) a MEMS SPDT switch and (b) a one-bit Ka-band MEMS TTD phase shifter. (From S. P. Sah et al., IEEE Trans. Microw. Theory Tech., vol. 68, no. 8, pp. 3024–3033, August 2013.)

FIGURE 1.10 Photographs of (a) SP4T switch and (b) 2-bit phase shifter. (From M. C. Scardelletti, G. E. Ponchak, and N. C. Varaljay, Ka-band, MEMS switched-line phase shifters implemented in finite ground coplanar waveguide, in European Microwave Conference, Milan, Italy, October 2002, pp. 1–4.)

double-throw (SPDT) to single-pole four-throw (SP4T) switching networks followed by different delay line lengths are used to develop a TTD phase shifter. Figure 1.9 shows a conventional one-bit Ka-band MEMS switched-line TTD phase shifters based on the SPDT switches [36]. Figure 1.10 shows a 4-bit MEMS phase shifter with four SP4T switches with 21 mm2 area reported by Tan et al. that provides an average insertion loss of 1.1 dB and phase accuracy of +2.3° at 10 GHz [37]. The concept of DMTL originated about 1960 in which a transmission line loaded with millimeter-wave Schottky diodes was used to achieve the desired phase shift. Later, this periodically loaded-line concept was used in developing microwave phase shifters using MEMS technology utilizing MEMS varactors as the capacitive loading, and the resulting structure shown in Figure 1.11 was named a DMTL phase shifter. DMTL-based X- and

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RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 1.11 Top view of a CPW line periodically loaded by shunt MEMS bridges, where W and S are the CPW line width and spacing, w and l are the MEMS bridge width and length, respectively, and s is the spacing between two consecutive MEMS bridges.

Ka-band 2- bit phase shifters were reported by Rebeiz et al. in 2002 [38] with worst-case insertion loss of 1.6 dB at 13.6 GHz using metal–air–metal (MAM) capacitors. In this configuration, the shunt capacitance associated with the MEMS bridges is in parallel with the distributed capacitance of the transmission line. The height of the MEMS bridge varies with a single analog control voltage. It changes the propagation characteristics on the transmission line and produces differential phase shift. An RF MEMS-based digital phase shifter has advanced significantly over the past decade. These phase shifters have been reported up to 6 bits using switched line, reflect line, low-pass/high-pass type and DMTL topology in different frequency bands [39–43]. A switched-line-based 5-bit phase shifter provides an average insertion loss of 3.6 dB at 10 GHz within a 28 mm2 area [39]. A 6-bit phase shifter was demonstrated with 2.8 dB loss at 18 GHz over a 40 mm2 area [40]. A packaged X-band 5-bit low-pass/high-pass phase shifter provides an average insertion loss of 4.5 dB at 12 GHz in a 9.2 mm2 area [41]. A DMTLbased 4-bit phase shifter is reported with 1.7 dB of average loss at 15 GHz with 7° of average phase error [42]. A triple stub topology-based DMTL phase shifter recently reported an average loss of 3.6 dB over 15–22.5 GHz band in a 63.7 mm2 area [43]. Major challenges of these phase shifters are to achieve low loss with desirable phase shift with good repeatability within a small area. These challenges become very critical with higher bit configurations at lower microwave frequency. Almost all the higher-bit (>4-bit) phase shifters reported so far have experienced the challenge to achieve low loss and good matching simultaneously with good phase accuracy over large cycles of operations in terms of power handling and life cycle.

1.5 Major Applications of RF MEMS Phase Shifters Electronic beam steering is a primary issue in modern phased arrays systems and is extensively used in military and commercial applications such as wireless communications and radar systems. Many studies and experiments have been performed on the electronic beam steering principle. The most popular and useful method is to use phase shifters in combination with an antenna array [44,45], as shown in Figure 1.12. The method of beam steering is broadly classified as time delay, frequency scanning and phase scanning. The phase shifters produce phase shifts between the elements of an antenna array and can steer the

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FIGURE 1.12 Schematic of a beam-steering, front-end-based phased-array antenna.

beam to the desired direction. Phase shifters are very critical and essential components in the beam steering, because they are the general devices used in phased arrays and they account for nearly half of the cost of an entire electronically scanned phased array.

1.6 Book Organization The book contains 14 chapters with this first chapter serving as an introduction to microelectromechanical systems. Chapter 2 starts with the design, development and complete characterization of a metal-contact MEMS switch. The motivation is to observe the switch functionalities in terms of mechanical behavior, electrical behavior, transient analysis, linearity, power handling, temperature behavior, S-parameters and cold-switched reliability. Chapter 3 discusses the design and implementation of different MEMS single-pole multithrow (SPMT) switching networks starting from single-pole double-throw (SPDT) to single-pole fourteen-throw (SP14T) configuration. Chapter 4 presents the design and development of dielectric-less lateral MEMS switches and switching networks. Chapter 5 reports on the theory, design and analysis of different types of microwave MEMS phase shifters. Chapter 6 describes Ku-band conventional switched-line phase shifters using MEMS in-line metal-contact switches. Complete design strategy, fabrication of the individual primary phase bits and performance optimization of two 5-bit phase shifters are discussed in detail in this chapter. Chapter 7 gives detailed analysis of a distributed transmission line-based 5-bit phase shifter using MEMS bridge and metal–air–metal (MAM) capacitors. Chapter 8 presents the design, analysis and measurement of a push–pull type MEMS actuator. In addition to this, the utility of the push–pull configuration is validated and tested on an analog phase shifter configuration. Chapter 9 attempts to address the design and development of a reconfigurable 5-bit DMTL phase shifter using the push– pull topology with thorough detailed analysis and experimental investigations. This 5-bit phase shifter is capable of providing 32 phase states with constant resolution over the 10–25 GHz frequency band. Initially, to validate frequency reconfiguration, all primary unit cells are fabricated and tested. Later, a complete 5-bit phase shifter is developed, tested and

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discussed in detail keeping in view the microwave and reliability performances. Chapter 10 describes a multifrequency phase shifter utilizing MEMS switching networks and DMTL phase shifters. A complete phase shifter bank that operates from 17 GHz to 60 GHz is discussed in detail in this chapter. Reliability and power handling of MEMS switches and phase shifters are described in detail in Chapter 11. The design and development of a 5-bit TTD phase shifter with low loss and desirable phase shift within a smallest possible area is reported in this chapter. This phase shifter drastically reduces the number of switch count and thus improves the reliability of the overall device. Two phases of design, fabrication and characterization are performed and discussed that substantially improve the overall performance of the 5-bit phase shifter compared to one another. Chapter 12 reports the design and development of 3-bit and 4-bit phase shifters using back-to-back switching networks. Chapter 13 discusses a novel topology of a digital phase shifter design using a combination of switched-line and DMTL phase shifters. Complete design methodology and characterization are discussed in this chapter. Chapter 14 describes the packaging methods and integration techniques for MEMS-based devices and components.

References 1. G. M. Rebeiz, and J. B. Muldavin, RF MEMS switches and switch circuits, IEEE Microw. Mag., vol. 2, no. 4, pp. 59–71, December 2001. 2. S. Lucyszyn, Advanced RF MEMS, Cambridge University Press, August 2010. 3. S. Oak, G. F. Edmiston, G. Sivakumar, and T. Dallas, Rotating out-of-plane micromirror, J. Microelectromech. Syst., vol. 19, no. 3, pp. 632–639, June 2010. 4. L. P. B. Katehi, G. M. Rebeiz, T. M. Weller, R. F. Drayton, H. J. Cheng, and J. F. Whitaker, Micromachined circuits for millimeter- and sub-millimeter-wave applications, IEEE Antennas Propag. Mag., vol. 35, no. 5, pp. 9–17, October 1993. 5. S. F. Mahmoud, Electromagnetic Waveguides: Theory and Applications, 1st ed., Institution of Engineering and Technology, June 1991. 6. D. M. Pozar, Microwave Engineering, 3rd ed., John Wiley & Sons, 2007. 7. B. C. Wadell, Transmission Line Design Handbook, 1st ed., Artech House, 1991. 8. S.Yang, Z. Hu, N. B. Buchanan, V. F. Fusco, J. A. C. Stewart, Y. Wu, B. M. Armstrong, G. A. Armstrong, and H. S. Gamble, Characteristics of trenched coplanar waveguide for high-resistivity Si MMIC applications, IEEE Trans. Microw. Theory Tech., vol. 46, no. 5, pp. 623–631, May 1998. 9. N. E. S. Farrington, and S. Iezekiel, Design and simulation of membrane supported transmission lines for interconnects in a MM-wave multichip module, Prog. Electromagn. Res. B, vol. 27, pp. 165–186, 2011. 10. Y. Shi, Z. Lai, P. Xin, L. Shao, and Z. Zhu, Design and fabrication of micromachined microwave transmission lines, Proc. SPIE, 2001, vol. 4557. 11. A. Q. Liu, RF MEMS Switches and Integrated Switching Circuits, Springer, 2010. 12. M. Sakata, Y. Komura, T. Seki, K. Kobayashi, K. Sano, and S. Horike, Micromachined relay which utilizes single crystal silicon electrostatic actuator, in IEEE International Conference on Microelectromechanical Systems, Orlando, FL, 1999, pp. 21–24. 13. Y. Uno, K. Narise, T. Masuda, K. Inoue, Y. Adachi, K. Hosoya, T. Seki, and F. Sato, Development of SPDT structured RF MEMS switch, in International Conference on Transducers, Solid-State Sensors, Actuators and Microsystems, Denver, CO, 2009, pp. 541–544. 14. S. Majumder, J. Lampen, R. Morrison, and J. Maciel, A packaged, high-lifetime ohmic MEMS RF switch, in IEEE MTT-S International Microwave Symposium Digest, Philadelphia, PA, June 2003, pp. 1935–1938.

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15. H. S. Newman, J. L. Ebel, D. Judy, and J. Maciel, Lifetime measurements on a high-reliability RF-MEMS contact switch, IEEE Microw. Wirel. Compon. Lett., vol. 18, no. 2, pp. 100–102, February 2008. 16. D. A. Goins, R. D. Nelson, and J. S. McKillop, Design of a 20 GHz low loss ohmic contact RF MEMS switch, in IEEE MTT-S International Microwave Symposium Digest, Honolulu, HI, June 2007, pp. 371–374. 17. J. Costa, T. Ivanov, J. Hammond, J. Gering, E. Glass, J. Jorgenson, D. Dening, et al., Integrated MEMS switch technology on SOI-CMOS, in IEEE Solid-State Sensors, Actuators, and Microsystems Workshop, Hilton Head, SC, June 2008, pp. 18–21. 18. N. Nishijima, J.-J. Hung, and G. M. Rebeiz, Parallel-contact metal-contact RF MEMS switches for high power applications, in IEEE International Conference on Microelectromechanical Systems, Maastricht, Netherlands, January 2004, pp. 781–784. 19. N. Nishijima, J.-J.Hung, and G. M. Rebeiz, A low-voltage, high contact force RF-MEMS switch, in IEEE MTT-S International Microwave Symposium Digest, Fort Worth, TX, June 2004, pp. 577–580. 20. J. Yao, and M. Chang, A surface micromachined miniature switch for telecommunication applications with signal frequencies from Dc up to 4 Ghz, in 8th International Conference on Solid-State Sensors and Actuators and Eurosensors IX. Transducers, vol. 2, June 1995, pp. 384–387. 21. R. E. Mihailovich, M. Kim, J. B. Hacker, E. A. Sovero, J. Studer, J. A. Higgins, and J. F. DeNatale, MEM relay for reconfigurable RF circuits, IEEE Microw. Wirel. Compon. Lett., vol. 11, no. 2, pp. 53–55, February 2001. 22. A. De Silva, C. Vaughan, D. Frear, L. Liu, S. Kuo, J. Foerstner, J. Drye, et al., Motorola MEMS switch technology for high frequency applications, in Microelectromechanical Systems Conference, Berkeley, CA, 2001. 23. K. Suzuki, S. Chen, T. Marumoto, Y. Ara, and R. Iwata, A micromachined RF microswitch applicable to phased array antennas, in IEEE MTT-S International Microwave Symposium Digest, Anaheim, CA, June 1999, pp. 1923–1926. 24. H. Sedaghat-Pisheh, J. Kim, and G. M. Rebeiz, A novel stress-gradient-robust metal-contact switch, in IEEE International Conference on Microelectromechanical Systems, Sorrento, Italy, January 2009, pp. 27–30. 25. H. Sedaghat-Pisheh, and G. M. Rebeiz, Variable spring constant, high contact force RF MEMS switch, in IEEE MTT-S International Microwave Symposium Digest, Anaheim, CA, May 2010, pp. 304–307. 26. C. D. Patel, and G. M. Rebeiz, A high-reliability high-linearity high-power RF MEMS metalcontact switch for DC-40-GHz applications, IEEE Trans. Microw. Theory Tech., vol. 60, no. 10, pp. 3096–3112, October 2012. 27. M. J. W. Rodwell, M. Kamegawa, R. Yu, M. Case, E. Carman, and K. S. Giboney, GaAs nonlinear transmission lines for picosecond pulse generation and millimeter-wave sampling, IEEE Trans. Microw. Theory Tech., vol. 39, pp. 1194–1204, July 1991. 28. A. S. Nagra, and R. A. York, Monolithic GaAs phase shifter with low insertion loss and continuous 0°–360° phase shift at 20 GHz, IEEE Microw. Guided Wave Lett., vol. 9, pp. 31–33, January 1999. 29. R. N. Simons, Coplanar Waveguide Circuits, Components, and Systems, John Wiley & Sons, 2001. 30. D.-W. Kang, H. D. Lee, C.-H. Kim, and S. Hong, Ku-band MMIC phase shifter using a parallel resonator with 0.18-μm CMOS technology, IEEE Trans. Microw. Theory Tech., vol. 54, no. 1, pp. 294–301, January 2006. 31. K. Kwang-Jin, and G. M. Rebeiz, 0.13-μm CMOS phase shifters for X-, Ku-, and K-band phased arrays, IEEE J. Solid-State Circuits, vol. 42, no. 11, pp. 2535–2546, November 2007. 32. B. Min, and G. M. Rebeiz, Single-ended and differential-band BiCMOS phased array frontends, IEEE J. Solid-State Circuits, vol. 43, no. 10, pp. 2239–2250, October 2008. 33. K.-J. Koh, and G. M. Rebeiz, A 6-18 GHz 5-bit active phase shifter, in IEEE MTT-S International Microwave Symposium Digest, Anaheim, CA, May 2010, pp. 792–795. 34. Jae Young Choi, Moon-Kyu Cho, Donghyun Baek, and Jeong-Geun Kim, A 5-20 GHz 5-bit true time delay circuit in 0.18 μm CMOS technology, J. Semicond. Tech. Sci., vol. 13, no. 3, pp. 193–197, June 2013.

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35. S. P. Sah, X. Yu, and D. Heo, Design and analysis of a wideband 15-35 GHz quadrature phaseshifter with inductive loading, IEEE Trans. Microw. Theory Tech., vol. 68, no. 8, pp. 3024–3033, August 2013. 36. M. C. Scardelletti, G. E. Ponchak, and N. C. Varaljay, Ka-band, MEMS switched line phase shifters implemented in finite ground coplanar waveguide, in European Microwave Conference, Milan, Italy, October 2002, pp. 1–4. 37. G.-L. Tan, R. Mihailovich, J. Hacker, J. DeNatale, and G. Rebeiz, Low-loss 2- and 4-bit TTD MEMS phase shifters based on SP4T switches, in IEEE Trans. Microw. Theory Tech., vol. 51, no. 1, pp. 297–304, January 2003. 38. G. M. Rebeiz, G. L. Tan, and J. S. Hayden, RF MEMS phase shifters: Design and application, IEEE Microw. Mag., vol. 3, no. 2, pp. 72–81, June 2002. 39. Zhu Jian, Yu-Yuan Weil, Chen Chen, Zhang Yong, and Lu Le, A compact 5-bit switched-line digital MEMS phase shifter, in IEEE International Conference on Nano/Micro Engineered and Molecular Systems, Zhuhai, China, January 2006, pp. 623–626. 40. C. D. Nordquist, C. W. Dyck, G. M. Kraus, C. T. Sullivan, F. Austin IV, P. S. Finnegan, and M. H. Ballance, Ku-band six-bit RF MEMS time delay network, in IEEE Compound Semiconductor Integrated Circuits Symposium, Monterey, CA, October 2008, pp. 1–4. 41. M. A. Mortonand, and J. Papapolymerou, A packaged MEMS-based 5-bit X-band high-pass/ low-pass phase shifter, IEEE Trans. Microw. Theory Tech., vol. 56, no. 9, pp. 2025–2031, August 2008. 42. B. Pillans, L. Coryell, A. Malczewski, C. Moody, F. Morris, and A. Brown, Advances in RF MEMS phase shifters from 15 GHz to 35 GHz, in IEEE MTT-S International Microwave Symposium Digest, Montreal, June 2012, pp. 1–3. 43. Mehmet Unlu, Simsek Demir, and Tayfun Akin, A 15–40-GHz frequency reconfigurable RF MEMS phase shifter, IEEE Trans. Microw. Theory Tech., vol. 61, no. 8, pp. 2397–2402, August 2013. 44. M. Albani, T. Cadili, F. Di Maggio, R. Gardelli, A. Incorvaia, C. Mollura, I. Pomona, et al., A 2-D electronic beam steering phased array for point-multipoint communication applications, in European Microwave Conference, Munich, October 2007, pp. 1629–1632. 45. M. Y.-W. Chia, L. Teck-Hwee, Y. Jee-Khoi, C. Piew-Yong, L. Siew-Weng, and S. Chan-Kuen, Electronic beam-steering design for UWB phased array, IEEE Trans. Microw. Theory Tech., vol. 54, no. 6, pp. 2431–2438, June 2006.

2 Behavioral Studies of Micromachined Contact Switches The objective of this chapter is to investigate the behavior of a microelectromechanical systems (MEMS) DC-contact switch and its characterization. The switch is designed, fabricated and tested using a simple gold-based surface micromachining process. The switch contacts or disconnects with an electrostatic actuation by an out-of-plane motion of the cantilever beam. The switch is implemented on a coplanar waveguide (CPW) transmission line. The performance of the metal-contact switch is analyzed and critically evaluated using different levels of characterizations. Comprehensive modeling and design of the switch are verified using different experimental investigations that include beam profile analysis, mechanical, electrical, switching and release time, linearity, temperature stability, power handling capability, and reliability (cold and hot switching), and the results are discussed in detail in this chapter.

2.1 MEMS Switch Design Figure 2.1 presents the top view and cross section of a simple MEMS metal-contact switch with all relevant structural parameters marked. The main purpose is to study the effect of the switch design parameters and to realize switch performances. The switch is fabricated using a 2 μm thick electroplated gold beam and a gap of 2.5 μm on 635 μm alumina substrate. Two 5.5 μm radius gold dimples with 1 μm depth are used at the tip of the beam to overcome a direct contact with the DC-isolation dielectric, and this reduces the associated dielectric charging effect. The switch is designed on a CPW transmission line with an inline configuration and actuated by electrostatic force. Titanium tungsten is used as a high resistive bias line that is covered by SiO2, as marked in Figure 2.1.

2.2 Switch Model The switch consists of a rectangular cantilever beam that is anchored with two metal posts with two symmetrical springs (l1). The stiffness of the beam (k) is primarily dependent on the beam thickness (tb) and length (L) as kα(tb/L)3. The Young’s modulus (E) of the electroplated gold is 45 GPa. The total length of the beam was restricted to 200 μm for a robust mechanical design. The width (W) of the cantilever beam was also limited to 100 μm in order to prevent the effect of vertical stress gradient, leading to curling at the corner of the beam tip. Furthermore, the spring constant and tip deflection versus process-induced residual stress (70 to 180 MPa) were optimized in CoventorWare [18]. A switch model including different model parameters is developed on the CoventorWare Saber platform, as shown in Figure 2.2. 15

16

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 2.1 Top view and cross section of the MEMS SPST switch.

FIGURE 2.2 Lumped representation of MEMS switch in Saber Architect. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

Behavioral Studies of Micromachined Contact Switches

17

The switch is designed using lumped beam, beam electrode, rigid plate, mechanical bus connector and gap generic squeeze damper modules [18] in the Saber schematic. Later, the measured damping coefficient (b) and mechanical quality factors (Qm) are introduced in the model to validate the switch performance in terms of actuation voltage and switching time.

2.3 Switch Design, Simulation and Measurements In this section, mechanical study and modeling of the cantilever-based DC-contact switch are discussed. The main focus is to study the effect of the switch design parameter in terms of mechanical behavior. The switch is fabricated on a 635 μm alumina substrate by using the well-established 7-mask process. Microscopic image of the switch is shown in Figure 2.3. The in-line MEMS switch is integrated in a 50Ω CPW line with 50 μm gap (G) and 110 μm line width (W). The 2 μm thick electroplated gold membrane is anchored to CPW transmission line by means of two 46 μm beams. 2.3.1 Switch Profile Analysis and Mechanical Resonance Initially, switch out-of-plane deformation is measured to observe the stress distribution on the cantilever beam. To measure the out-of-plane deformation, surface topographies are acquired using the Taylor Hobson Optical Profilometer. Interferometric profilometry would be ideally suited in terms of height resolution. However, it cannot deal with the large local surface angles associated with the relatively high surface roughness of the metallic MEMS. The out-of-plane deformation of the movable beam and top surface profile on different layers is experimentally observed as shown in Figure 2.4. Deflection at the tip is more in the switch as shown in the 3D profile. The switch has an air gap of 3.4 μm, which indicates that it has experienced 0.9 μm extra deformation than its initial gap height of 2.5 μm after release.

FIGURE 2.3 Microscopic image of the SPST MEMS switch. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

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RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 2.4 Measured interferometry profile of the cantilever switch after release. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

The tip deflection (Δz) of the cantilever beam can also be found analytically and is given by [1] ∆z =



∆ σz L2 (2.1) 2Eeff

Here, Eeff is the effective Young’s modulus and it is replaced by E/(1–υ2). Δσz is the residual stress gradient of the cantilever beam. For a bilayer beam, the tip deflection is represented by 3l 2  σ2 σ1  (1 −m ) (2.2) −   t  E2 E1  1 / mn + m3 n + 4m2 + 6m + 4 2



∆z =

(

)

where m = t1/t2 is the thickness ratio, a residual stress of σ1 and σ2 and a biaxial Young’s modulus ratio of n = E1/E2. Due to the nonuniform profile of the effective residual stress (σeff), the cantilever beam either deflects upward or downward. A positive stress gradient lifts the beam upward, whereas a negative stress gradient bends the beam downward. Figure 2.5 presents a positive stress gradient variation versus tip deflection (Δz) as a function of three different stress levels (150, 170 and 180 MPa). Process-induced residual stress varies from 150–190 MPa at room temperature (25°C) and maximum tip deflection is found to be 2.04 μm for a 200 μm long and 100 μm wide cantilever beam and it is positive in nature. The observed out-of-the plane deformation is not uncommon in MEMS switches. Residual gradient stress causes undesirable out-of-plane deformation [19]. When a thin film is deposited on a sacrificial layer at a temperature lower than its flow temperature, the intrinsic stresses develop in the film-sacrificial layer system [20]. A detailed study has already been reported that theoretically explains the mechanisms of these stresses [21,22] and experimentally validated their effects [23,24]. Furthermore, in-plane residual stress primarily increases the switch spring constant and it is therefore essential to control it within a reasonable limit. It was experimentally found that the induced tensile stress during the fabrication process was of the order of 170 MPa

Behavioral Studies of Micromachined Contact Switches

19

FIGURE 2.5 Stress gradient versus tip deflection for different stress values.

[24]. A first-order approximation that shows the impact of this stress on the device parameter is analyzed by Roark and Young [25] where the deflection of a guided-end cantilever is also calculated under simultaneous axial tension and concentrated transverse loading. The maximum deflection of the tip of the cantilever is given by Equation 2.3 as follows [26]:



zmax =

2   Wtl  ( cosh ( γ l ) −1) −( sinh ( γ l ) −γ l )  (2.3) γ P  sinh ( γ l )   

where P is the axial tensile load, Wtl is the transverse concentrated load at the tip of the beam, l is the total length of the beam, and γ is as given in Equation 2.4 [26]:

γ=

P (2.4) EI x

Here, Ix is the moment of inertia along the x-axis and is given by Equation 2.5:

Ix =

wt 3 (2.5) 12

where w and t are the width and thickness of the cantilever beam, respectively. The mechanical resonance frequency of the cantilever beam depends on the dimension of the beam and material properties, such as the Young’s modulus, residual stress and material density. The fundamental resonance frequency can be found from [27]

f 0 = 0.16154

tb L2

Ee (2.6) ρ

where ρ is the gold density and this value is 19300 kg m–3.

20

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 2.6 Variations of mechanical resonance frequencies ( f0) with positive stress gradients at different stress values.

The change in mechanical resonance frequency with +1 MPa/μm to +5 MPa/μm stress gradient variation is depicted in Figure 2.6. Simulations show a minimum f0 = 4.4 kHz at 190 MPa (+5 MPa/μm) and maximum f0 = 14.3 kHz at 150 MPa (+1 MPa/μm). In the present study resonant frequency of the beam is much higher than the squeeze film cut-off frequency (ωc) and it can be explained by Equation 2.7 [1]:

ωc =

σc g02Pa (2.7) 12µr 2

where Pa is the ambient pressure, μ is the viscosity of the gas and r is the characteristic length of the beam. The σc is the cut-off squeeze number and its value is 19.74. For this design, at standard temperature and pressure, ωc is 3.18 kHz. 2.3.1.1 Switch Mechanical Measurements Mechanical behavior of the MEMS switch can be characterized using a laser Doppler vibrometer (LDV) where electrical connection is established between the bridge and the fixed electrode by a wafer prober [28]. A chirp voltage with a frequency sweep of 0 to 170 kHz is applied at the cantilever. The switch is excited with 3 volt actuation voltage, which is much less than its pull-in. LDV is made up of a MSA 400 Micro-motion Analyser, OFV 511 Laser Interferometer, an optical microscope with a CCD camera and OFV 3001 Vibrometer Controller [28–32]. The vibration spectrum of the switch shows the magnitude of deflection (pm) with respect to the frequency (kHz) as depicted in Figure 2.7. The switch shows natural frequency (wn) at 9.7 kHz with the excitation. The inset of the figure shows the shape of the fundamental mode of vibration under excitation, which confirms that switches are successfully released after fabrication. The quality factor and damping of switches can be extracted from the mechanical measurement. 2.3.2 Switch Pull-In Behavior Basics The voltage driving scheme of the cantilever-based contact switch is similar to the condition of the parallel plate actuator under electrostatic actuation. Figure 2.8 shows a schematic

Behavioral Studies of Micromachined Contact Switches

21

FIGURE 2.7 Vibration spectrum of the switch showing 9.8 KHz resonance. Inset shows fundamental mode of vibration. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

k

x

FIGURE 2.8 Schematic of an electrostatic actuator. The plate is attached to a spring k.

of an electrostatic actuator that could be used, for example, as a tunable radio frequency (RF) capacitor for the present study. The mass of the beam is used as a movable electrode and the fixed electrode is under the mass with a gap height of g. When a potential difference is created between two parallel plates, it pulls the beam toward the fixed electrode and reduces the plate separation g – x. Elastic recovery force tends to pull the mass of the beam toward the initial position once voltage is removed. The balance position of the beam is simply obtained by the condition of force balance with electrostatic (Fe) and restoring (Fk = kx) forces. As the electrostatic force is inversely proportional to the square of the g, a nonlinear behavior is observed that causes the instability problem in the beam actuation. To derive the expression of point of instability or pull-in voltage (Vpi), we start by writing the total potential energy in the system:

1 1 E = - CV 2 + kx 2 2 2

22



RF Micromachined Switches, Switching Networks, and Phase Shifters

E=-

1 eA 2 1 2 V + kx (2.8) 2 g-x 2

where A is the parallel plate actuation area, C is the deformable capacitor and V is the external bias voltage. The first term of Equation 2.8 represents the electrostatic potential of the deformable beam and of the voltage source. The second term represents mechanical energy stored in the spring. The force (F) acting on the movable electrode is obtained by taking the derivative of Equation 2.8 with respect to x and it is

F=-

¶E 1 eA 1 2 2 = kx (2.9) 2 V ¶x 2 ( g - x ) 2

At the point of equilibrium, electrostatic and restoring forces are equal and cancel (F = 0), and Equation 2.9 gives

kx =

1 eA V 2 (2.10) 2 ( g - x )2

Equation 2.10 has no solution above the point of displacement or the pull-in voltage (Vpi) and it becomes unstable. The expression for the pull-in point is obtained by taking the partial derivative of Equation 2.9 to obtain the stiffness of the beam:

¶F eA = V 2 - k (2.11) ¶x ( g - x )3

Substituting Equation 2.10 in Equation 2.11 gives the stiffness at the equilibrium point and it represents

¶F 2kx = - k (2.12) ¶x g - x

¶F At zero-bias condition, Equation 2.12 is simplified to = -k. The point of instability is ¶x ∂F obtained by keeping = 0, and it results in the expression ∂x

x=

1 g (2.13) 3

Beyond this point stiffness of the beam becomes positive and the system is unstable. Finally, the pull-in voltage of the beam is obtained by substituting Equation 2.13 into Equation 2.10, and it results in the expression given by Equation 2.14:

Vp =

8kg 3 (2.14) 27 Aε

Equation 2.14 describes the pull-in behavior of an electrostatic actuator. Note that when mass of the beam is pulled in, it would not be released until the electrostatic force is taken

Behavioral Studies of Micromachined Contact Switches

23

away completely. Moreover, the pull-in effect takes place from one-third of the gap from the beam top or two-thirds of the gap from the bottom fixed electrode. The pull-in effect of the beam is completely eliminated using a series of capacitors and related descriptions are available from Nishijima et al. [1]. 2.3.3 Dynamic Behavior of the Switch Two different spring constants play an important role in the switch actuation. One is the natural spring constant (kn), which is defined by the tip displacement due to distributed force over the beam. Another one is the actuation spring constant (ka) that is defined as the tip deflection due to distributed force over the actuation electrode. Dimple height determines the switch deflection at zero-bias state with kn and leading to the release force. The point-to-point force applied at the dimple area is divided by the displacement as a function of ka. The pull-in (Vpi) and release voltage (Vr) as functions of Δσ vary from 1 to 5 MPa/μm as shown in Figure 2.9. It is seen that Δσ variation of +2 to 5 MPa/μm results in a 7–12 V (5–8 V) shift in the Vpi (Vr). Figure 2.10 shows the contact force (Fc) and release force (Fr) variation at different bias voltages (30–70 V) for Δσ = 0 and 4 MPa/μm. It is observed from the simulation that the switch can operate satisfactorily under both conditions with a shift in applied voltage. The pull-in and release behavior of the MEMS switch is defined with an applied electrostatic force and zero external force, respectively, from the dynamic 1–D equation and it can be expressed as given in Equations 2.15 to 2.19 [1]:



me

1 e0 AV (t)2 d2z dz + ( ) + = + FLJ (Pull-in) (2.15) b z k z a 2 ( g 0 - z )2 dt 2 dt FLJ =

c1 A c2 A (2.16) ( g0 - z)3 ( g0 - z)10

FIGURE 2.9 Pull-in (Vpi) and release (Vr) voltage versus stress gradient.

24

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 2.10 Simulated contact (Fc) and release (Fr) forces versus voltage for different stress gradients.



me

d2z dz + b( z) + k n z = 0 (Release) (2.17) 2 dt dt b( z) =



Qn =

3 m A2 (2.18) 2 p ( g 0 - z )3

kn ka (2.19) , Qa = 2πf 0b( z) 2πf 0b( z)

where me is the effective switching mass that can be determined from the principle of equivalence of kinetic energy. Equation 2.15 represents the Lennard-Jones (FLJ) force, which is a combination of the attractive van der Waals force (first term) and the repulsive nuclear forces (second term). c1 = 10−80 Nm and c2 = 10−75 Nm8 are considered to improve the convergence of the numerical Equation 2.15. The damping b(z) is a function of beam deflection, where μ is the air viscosity (μ = 1.076 × 10−5 Pa. s). The natural (Qn) and actuation (Qa) dependent quality factors depend on the damping. Here, the switch does not snap down on the actuation electrode, but it is stopped when the dimple touches the bottom transmission line. The pull-in voltage of the switch can also be found analytically using Equation 2.20 [33]:

Vpi = ( g0 - z)

2 ( k1 + k 2 ) z (2.20) e0 A

where (g0 – z) is the gap as the switch closes, ε0 is the permittivity of free space, A is the electrostatic overlap area, and z is the pull-in distance, which increases with positive stress gradients. Figure 2.11 shows the change in pull-in voltages with Δz for different stress values. It shows that the tip deflection of 1–1.3 μm results in 2 V shift in pull-in voltage (Vpi). Stress gradient on the beam can significantly impact the capacitance–voltage (C–V) characteristics. A higher positive stress gradient shifts the C–V curve toward the right with

Behavioral Studies of Micromachined Contact Switches

25

FIGURE 2.11 Simulated pull-in voltage versus tip deflection for different stress.

FIGURE 2.12 Simulated C–V curves with hysteresis at different stress gradients.

higher down-state capacitances with good linear tuning at 150 MPa stress, as depicted in Figure 2.12. Stress gradients of +3 and +5 MPa/μm show pull-in voltage at 34.5 V and 45 V, respectively. Simultaneously, release voltages are 24 V and 35 V, respectively. At zero stress the gradient beam gives pull-in at 20 V bias voltages. The critical voltage is the voltage at which the cantilever beam comes in physical contact with the bottom fixed electrode or drive electrode and the device shorts out. This critical or collapse voltage can be found analytically using Equation 2.21:

æ g - d ö 2kdc (2.21) Vcr = ç 0 c ÷ è a ø e0 A

26

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 2.13 Simulated switching time versus voltage for different gap heights.

where dc is the collapse distance and a is the applied load position on the cantilever beam. The calculated Vcr of the switch is 93 V, which is 2.65 times the switch pull-in voltage (35 V). The operating voltage range of the switch always lies between the switch initial pull-in and collapse voltage. Equation 2.15 is solved analytically for switching time (ts) versus voltage for z = 0 to z = x μm, where x varies from 2.5 μm to 3.5 μm with 0.5 μm step, as shown in Figure 2.13. The velocity of the switch has been calculated using Equations 2.15 and 2.18 at three different voltages (35 V, 40 V, 45 V) and it varies from 10 to 21 m/s. The shorter switching (ts) time can also be found using Equation 2.22 [1]:

ts = 3.67

Vp Vsω 0

where Vs = 1.5Vp (2.22)

Impact force (Fi) is also an important parameter in switch reliability. Fi is a function of the induced hardening during the contacting period and dislocation in the crystalline structure of the gold–gold contact and it is given by (2.23):



Fi = 0.4653 ( me )

3/5

( νi )

−6/5

(

 Fc 1 −υ2  r3 

)  

−1/5

E−13/5 (2.23)

where me is the effective switching mass, vi is the impact velocity at the contact and r is the radius of the contact spot under the contact force Fc. The contact force is the reaction force and it can be found by assuming zero tip deflection, as given in Equation 2.24:

Fc =

ε0 AeVs2 a 2 (3L −a) (2.24) 4L3 ( g0 −z)2

Figure 2.14 presents the impact force versus applied bias voltage for different Δz values for 150 MPa stress and +3 MPa/μm stress gradient. It is observed from the calculated impact force that it drastically increases for dimple depth of less than 1 μm, and between 1 μm and 1.5 μm, Fi variations are much less. In this study, dimple depth has been chosen to be 1 μm

Behavioral Studies of Micromachined Contact Switches

27

FIGURE 2.14 Calculated impact force versus actuation voltage assuming purely elastic contact for different tip deflections.

to improve the switch reliability with minimum contact bounce. The contact bounce was calculated using Equation 2.15 with initial conditions dz/dt = –cvi and z = g0 – d at t = 0, where d is the dimple depth and c is the rebound coefficient, which is considered to be 1 for purely plastic contact (for worst case behavior). Equation 2.18 can also be used for the bounce analysis with b(z) = 0 when dz/dt  1 (2.25)

where bn, mn and ωn are the damping, effective mass (mn = ½ ρwLtb) and angular resonant frequency, respectively, of the nth mode. 2.3.3.1 Actuation/Release Voltage Measurements Electrical measurement of the DC-contact MEMS switch is performed with an Agilent 4284A LCR meter with a probe station. A brief description of the setup can be found in Dey and Koul [30]. To observe the C–V profile from the fabricated MEMS switches, a voltage sweep from 0 to 50 V has been imposed to the top cantilever beam using the probes, whereas fixed electrodes are connected to the ground. A small AC signal of 5 MHz is given on the DC actuation voltage to measure the capacitance value. Finally, open circuit offset measurement corrections were ensured before recording the change in capacitance values with applied bias. The C–V measurement shows the pull-in voltage at ~40 V as shown in Figure 2.17. This is basically the capacitance variations between the beam and controlled electrode. Measured release voltage (Vr) was found to be at ~26.4 V after which the switch bounces back to its initial position. The spring constant of the cantilever switch is ~20 N/m at 150 MPa stress. 2.3.3.2 Switching/Release Time Measurements Switching/release time measurement was carried out using the Agilent-made Infiniium DSO-X 92504, 25 GHz high frequency digital storage oscilloscope (DSO). A square wave of 0 to 60 volt with 1 KHz frequency was imposed on the actuation pad and the corresponding effects on switching (ts) and release (tr) times were recorded from the DSO. The

Behavioral Studies of Micromachined Contact Switches

29

FIGURE 2.17 C–V measurements of MEMS switch showing pull-in at 40 V and pull-out at 26.4 V.

FIGURE 2.18 Measured switching and release times are 78 μs and 118 μs. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

measured ts and tr times are 78 μs and 118 μs, respectively, as shown in Figure 2.18. The switch bounces back to its initial position at 84 μs and it takes additional 34 μs to completely settle down. The switching time is measured to be 97 μs to 78 μs for actuation voltage of 40–60 V (1.5 times of Vpi). The switching and release time measurements closely follow our prior assumption of tip deflection (1 μm) with 3–4 MPa/μm of Δσ and its effect on switching time response. The effect of 3–4 MPa/μm deflects the tip by about 0.5 to 1 μm, and it shifts the switching time from 55 μs to 73 μs (Fig. 2.18). The small deviation is mostly attributed to the environmental damping. Switching time measurement setup is available in Dey and Koul [36]. 2.3.4 Switch Contact Resistance Basics The dimples of the cantilever beam and bottom transmission line are making a finite number of asperity contacts during electrostatic actuations. These contact spots are different in size based on the roughness, which limits the contact area during the diffusive transport of electrons and increasing the overall ohmic resistance. More electrons are ballistically

30

RF Micromachined Switches, Switching Networks, and Phase Shifters

transported and the boundary scattering of electrons to the total constriction resistance will increase when the radius of the asperity contact is of the order of the electron mean free path (50 nm in Au) [37]. This boundary scattering effect can drastically reduce the switch lifetime during a long cycle of operations. The constriction resistance (Rc) can be defined by Equation 2.26, after considering both ohmic and boundary scattering effects [38,39]:

4ρλ  λ  λ ρ Rc = f   RM + Rs = f   + (2.26)  r  r  2r 3πr 2

where λ is the mean free path, RM is the Maxwell resistance due to the lattice scattering mechanism [40] and Rs is the Sharvin resistance from the boundary scattering of electrons. In Equation 2.26 f(λ/r) is an interpolation function which is defined by Equations 2.27 to 2.29:

 λ  1 + 0.83K n f =  r  1 + 1.33K n

where, K n =

λ (2.27) r



 λ f   = 0 when K n ≅ 0 for r > > λ (2.28)  r



 λ f   = 0.624 when K n ≅ α for r < < L (2.29)  r

where Kn is the Knudsen number, which depends on the transition between two resistance regimes. The radius of the contact spot (r) for two independent contact regimes used in this work can be defined by Equation 2.30:

r=

Fc (2.30) 2πH

where H is the Meyer indentation hardness of the contact material (1.6 GPa for Au) [33]. To get the lower limit of the contact resistance (Rlc) for two contacts, it is assumed that contact spots do not interact with each other and are in parallel. Hence, it can be approximated as in Equation 2.31:

Rlc =

Rc1 + Rc 2 (2.31) Rc1Rc 2

where Rc1 and Rc2 are two contact resistances from two dimples. An upper limit of the contact resistance (Ruc) can also be defined by considering all asperity contacts with single asperity where all contact area are equal to the total effective contact area with an effective radius of reff and an average resistivity (ρav). The Ruc is defined by Equation 2.32:

Ruc = f

ρav 4ρavλ + (2.32) 2 2reff 3πreff

The change in contact resistance versus applied bias at different Δσ values is depicted in Figure 2.19. The contact resistance changes from 5.3 Ω (5 MPa/μm) to 2.65 Ω (0 MPa/μm) from 20 to 100 V.

Behavioral Studies of Micromachined Contact Switches

31

FIGURE 2.19 Simulated contact resistance versus applied bias for different stress gradient.

2.4 Thermal Behavior of the Switch The temperature variation results in a change in the residual stress in the beam due to the difference in the thermal expansion coefficients (α) between the alumina substrate (αalumina = 7.2 × 10 –6/°C) and gold beam (αgold = 14.2 × 10−6/°C). The change in the residual stress (σ*) is given by Equation 2.33 [2]:

s* = sres - DaEDT , Da = a gold - a alumina (2.33)

where σres is the residual stress in the bridge (150 MPa) and ΔT is the bridge and substrate temperature change due to thermally induced stress. At high temperature, the bridge tensile stress turns into the compressive stress and it affects the spring constant of the switch. In the cantilever structure, the stress-dependent part of the spring constant (k2) is less compared to the fixed–fixed beam structure. The change in spring constant versus temperature is shown in Figure 2.20. The spring constant of the MEMS beam is 37 to 13 N/m as the temperature changes from –30°C to +70°C for different Δσ values. This also shifts the pull-in voltage (Vpi) from 12 to 14 V. Hardening of the gold–gold contact material increases the resistivity during continuous contacting period even at room temperature (25°C). As a result, the contact area increases due to extensively deformed contact asperities. This process is more prominent with surface roughness. After a few cycles of operations, resistivity is mostly affected by the contact temperature (Tc), which can soften the contact material and give rise to the hardening effect of inelastic deformations. Equation 2.34 describes a power law formulation on strain hardening and softening effects on resistivity [28,29]: q



ep ö æ æ æ Qa ö ö r = r* ç 1 + ÷ ç 1 - exp ç - kT ÷ ÷ (2.34) e c øø ref ø è è è

32

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 2.20 Simulated temperature versus spring constant and pull-in voltage at different Δσ.

where ρ* is the average resistivity of a contact spot at temperature Tc; εref and εp are the reference and plastic strain, respectively; q is the material dependent parameter; k is the Boltzmann’s constant; and Qa is the activation energy. When the size of the contact spots are close to the electron mean free path of the contact material, the contact resistance increases due to the boundary scattering effect, which does not contribute to the Joule heating of the contact spot [38]. The temperature of the contact is given by Equation 2.35 [37]:

Tc =

γ RM 2 Vc + T0 (2.35) 4LRc

where L is the Lorenz number (L = 2.45 × 10−8 W Ω K−2); Vc is the contact voltage; γ is a scaling parameter, which is based on surface flatness and frequency distribution density; and T0 is the ambient temperature. Equation 2.35 is valid for clean metal contacts with an assumption that the electric and thermal current follows the same paths [37]. The presence of a resistive film and circular contact area generates heat uniformly over the contact interface and it is given by Equation 2.36:

Tc =

I 2Rc 2πK

2πH + T0 (2.36) Fc

where I is the root mean square (rms) value of RF current and K is the thermal conductivity of gold [318 W/(mK)]. Equation 2.36 assumes heat is dissipated evenly between the top and bottom contacts, which are approximated as semi-infinite surfaces. The initial gap height of the cantilever beam is a function of the stress gradient (Δσ), which also changes with temperature. The simulated tip deflection versus temperature for 0, +3 and +5 MPa/μm (assuming 150 MPa stress) is depicted in Figure 2.21. It shows the beam deflects downward with temperature rise, which is also the reason for the reduction of Vpi (Figure 2.11). The thermal time constant (τ) of the contact area can be analytically obtained using Equation 2.37 [1]:

τ=

ρcC pl 2 (2.37) K

Behavioral Studies of Micromachined Contact Switches

33

FIGURE 2.21 Simulated tip deflection versus temperature for different Δσ.

where ρc and Cp are the density and the specific heat capacity of the contact material, respectively. Due to the small thermal mass of the contact, localized temperature increases up to the order of a few nanoseconds. The thermal time constant in this switch is 57 µs. The increase in contact temperature also results in an increase in the contact resistance given by Equation 2.38 [1]:

∆Rc =

R (T )  2  =  1 + αT  (2.38)  R0 3 

where ΔRc is the change in contact resistance, R0 is the resistance at ambient temperature and R(T) is the resistance at temperature (T) above the ambient at the contact point. Furthermore, conduction and convection heat transfer are the main sources of temperature distribution in the gold beam. This can be explained using Equations 2.39 and 2.40 [1]:

Qcond = KAt

∆T ∆T L = = > Rcond = (2.39) L Rcond At K

where Qcond is the rate of conduction heat transfer, At is the surface area subjected to conduction and Rcond is the thermal resistance of conduction heat transfer process. Convection heat transfer can also be defined by Equation 2.40:

Qconv = KAt∆ T =

∆T 1 = > Rcond = (2.40) Rcond At h

where h is the convection heat transfer coefficient (W/m2K). Qcond is defined by Fourier’s law of heat conduction and Qconv is defined by Newton’s law of cooling. For the case of a 200 μm length cantilever beam, Rcond = 3.94 × 103 K/W and Rconv = 0.696 × 106 K/W. The effect of convection heat transfer is negligible in this case. Moreover, radiation heat transfer between the beam and surrounding surface is also negligible. Therefore, the primary mode of heat transfer in the present study is conduction.

34

RF Micromachined Switches, Switching Networks, and Phase Shifters

2.4.1 Thermal Measurement The temperature stability of the MEMS switch was observed by measuring the change in Vpi and Vr voltages as a function of temperature. A temperature controller (Temptronic Corporation, Mansfield, Massachusetts) was attached to the chuck of the probe station. It was used to set a stable operating temperature during the measurement. The chuck temperature was increased from –30°C to 70°C and then again decreased to room temperature (25°C). During this temperature rise, beam tensile stress turns to compressive stress. Initially, the cantilever beam deflection profile was observed without applying any bias. The maximum deflection of –0.36 μm (downward) was observed over 100°C chuck temperature range (−30°C to +70°C). This deflection was measured from the S21 response (isolation) of the (200 μm length) cantilever switch. Finally, temperature measurement was carried out with 0.1 W of RF power. Measurements show Vpi and Vr are decreased by ~5 V and ~8.2 V, respectively, when temperature is changed from −30 °C to +70°C, as depicted in Figure 2.22. This is most likely due to the effect of spring softening at higher temperature that leads to decrease in spring constant (softening temperature of gold is 370°K). Beam in-plane tensile stress turns into compressive with high temperature due to the different thermal coefficient of expansions between gold (αm) and alumina substrate (αs) [1]. It deflects the beam downward and decreases the Vpi and Vr, respectively.

2.5 RF Power Handling The RF power handling capability for a DC-contact MEMS switch is mostly limited by a localized temperature rise at the contact spot (Tc). Low thermal resistance of the beam generates low temperature rise on the membrane. Nonuniform heating on the cantilever beam should be minimized to reduce the effect on spring constant and tip deflection. The temperature rise due to the incident RF power can be expressed as Equation 2.41:

FIGURE 2.22 Measured change in Vpi and Vr over –25°C to +70°C temperature.

Behavioral Studies of Micromachined Contact Switches



Tc =

35

γ RM Rc Pinc + T0 (2.41) 4LZ0

where Pinc is the incident RF power and Z0 is the transmission line impedance (50 Ω). Contact temperature versus RF power for clean metal contacts for various contact forces is depicted in Figure 2.26. For incident RF power above 1.6–3 W, all contact forces result in temperature beyond the softening temperature of the gold (370 K) [40], which is not desirable. This analysis was carried out with Δz = 1 μm and Δσ = +3 MPa/μm. In this analysis 15 μN force is used for the upper limit, which requires 50 V bias (Figure 2.23). Furthermore, the switch gives 35 V pull-in at Δσ = +3A MPa/μm (Figure 2.12), which corresponds to 6 μN contact force (Figure 2.23). Pure gold–gold contact can sustain up to 3 W of RF power with 6 μN force. Contact temperature variation with power for a contaminated surface can be obtained using Equation 2.42:

Tc =

Rc PRF + T0 (2.42) 2πrZ0

Figure 2.24 presents the contact temperature versus RF power (PRF), assuming 3.5 Ω of the contact resistance in each case with an alumina substrate. It shows that switch RF power handling capability reduces with resistive contaminations. Compared to a clean surface, RF power reduces to 500 to 700 mW with 6–11 μN of contact force. All the analysis is based on the limit of gold softening temperature (370 K or 97°C). When the switch is in the up-state, then virtually all the incident RF power is reflected back to the source. The required power to actuate the switch or self-actuation at high RF power can be written as Equation 2.43:

Pact =

kz( g0 −z)2 (2.43) 2ε0 AZ0

The minimum required RF power to cause self-actuation as a function of different stress gradients for various values of tip deflection is depicted in Figure 2.25. It shows that

FIGURE 2.23 Contact temperature versus RF power for clean metal contact.

36

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 2.24 Contact temperature versus RF power for contaminated metal contact with a resistive film of 3.5 Ω.

FIGURE 2.25 Calculated minimum self-actuation RF power versus stress gradient as a function of tip deflection.

pull-down RF power varies from 2.5 W (Δz = 0) to 4.1 W (Δz = 1) within 0 to 5 MPa/μm stress gradient. The self-actuation occurs only when rms voltage VRF is greater than the switch pull-in voltage (Vpi). In the down-state position, open circuit voltage across the series switch is given by Equation 2.44 [1]:

Vsw =

2Vpk 2

2 RF

2 0

1 + 4ω C Z

≈

Vpk 2ω CRF Z0

for ω Cu Z0 > > 1 (2.44)

where Vpk is the peak-to-peak RF voltage, CRF is the RF capacitance over the actuation electrode and the MEMS beam and ω is the operating frequency. The switch electrostatic force (Fe) must be larger than the switch mechanical restoring force, for a switch stays at the down-state position. This is defined by the hold-down voltage as given by Equation 2.45:

37

Behavioral Studies of Micromachined Contact Switches

FIGURE 2.26 Calculated hold-down power versus spring constant at three different frequencies.



Vh =

2k( g0 −z)  t  g0 + d  (2.45) ε0 A  εr 

The incident power that results in an RF hold-down (Ph) of the switch can be obtained by combining Equations 2.32 and 2.33, as given in Equation 2.46: 2



t   4ω CRF k( g0 −z)  g0 + d   εr  (2.46) Ph = ε0 A

The variation of Ph with spring stiffness at three different frequencies (2, 10 and 20 GHz) is shown in Figure 2.26. Spring constant variation from 6 to 30 N/m was also observed over the temperature range (Figure 2.18) and it reflects on hold-down power. The Ph is high at higher frequency, which is attributed to the short circuit at high RF power. Beam tip deflection and stress gradient are assumed to be zero for the analysis. The RF power dissipation from the MEMS switch was examined using full-wave simulation in a High Frequency Structure Simulator (HFSS) platform, as shown in Figure 2.27. Symmetrical boundary condition was used during the simulation and S-parameter results are obtained. Power dissipated (Pdiss) by the MEMS switch can be calculated using Equation 2.47 [1]:

2

2

Pdiss = Pi (1 − S11 − S21 ) (2.47) 2

2

where Pi is the incident power and Loss = 1 − S11 − S21 . Power dissipated in the entire switch is first determined from the S-parameter simulation after including the loss of the transmission line, dielectric, substrate and MEMS beam. To extract the loss of the switch alone, all other metal and dielectrics are considered to be lossless in the simulations. The conductivity of the gold beam is considered to be 4.7 × 107 S/m. These two processes were repeated in up-state and down-state of the switch. The simulations show that Pdiss = 0.0633 Pi (S11 = 28 dB, S21 = 0.15 dB) at down-state and Pdiss = 0.0083 Pi (S11 = 0.147 dB, S21 = 16.2 dB) at

38

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 2.27 Simulated current distributions in (a) up-state and (b) down-state positions at 20 GHz.

up-state at 20 GHz frequency. Switch power dissipation is more in the down-state compared to the up-state. It also attributes to the high current density in down-state compared to the up-state at 20 GHz, as depicted in Figure 2.27. The figure also shows that temperature rise in the down-state is much higher than in the up-state over the frequency of interest. 2.5.1 RF Power Handling Measurements Temperature sensitivity and current density are the primary sources of RF power handling capability of DC-contact MEMS switch. The beam temperature is a function of the incident power and temperature is higher at down-state compared to up-state in metalcontact switches. All cable losses were calibrated out in the measurement. Figure 2.28 shows the measured pull-in (Vpi) and release (Vr) voltages of MEMS switch as a function of the incident power. The switch can handle a maximum ~1.5 W of RF power before failing to release. The Vpi (Vr) was reduced by ~38% (36%) up to 1.5 W with a shift in voltage from 40 to 29 V (26 to 18 V) at 2 GHz. It is mostly attributed to the dielectric charging with incident power due to temperature increase in the switch and also due to contact point degradation and contaminations with additional attractive force from the RF power. In addition to this, power handling is also dominated by the microwelding. It depends on the

FIGURE 2.28 Measured power versus pull-in and release voltages.

39

Behavioral Studies of Micromachined Contact Switches

material hardness (1.6 GPa for gold), surface condition (roughness and asperities), softening temperature (370 K for gold) and applied force. The equivalent Vrms voltage on the switch at 1.5 W is 8.94 V and has the effect of reducing Vpi from 40 V to 38.98 V (402 = Vpi2 + Vrms2). The additional Vpi drop is attributed to the dielectric charging effect and gold softening under high RF power level [2]. Measured power dissipated (Pdiss) by the MEMS beam in the down-state (0.1379 W) is 1.5 times higher than in the up-state (0.0915 W) at 20 GHz with 1.6 W of incident power. It is also observed from the measurements that self-actuation (VRFrms > Vpi) or latching (VRFrms > Vr) starts from ~1.6 W of RF power.

2.6 Switch S-Parameter Analysis and Measurements Figure 2.29 presents the equivalent circuit model of the SPST in-line series switch. The elements Ls, Cu and Rs are the MEMS bridge inductance, capacitance (OFF-state) and resistance (ON-state). The Cb is the parallel plate capacitance between the bridge and the bottom fixed electrode. The Rb1 and Rb2 are the internal and external bias resistances. The Q factor loss is dominated by bias resistances. Since Rb1 and Rb2 are very large (>25 kΩ), the complete bias network has a very negligible effect on the RF performance of the switch. The series capacitance of the MEMS bridge (Cu) is composed of coupled capacitance (Cc). The Cc is the capacitance from the signal feed through the MEMS beam that is coupled through the alumina substrate. The Cc value of the alumina substrate is 11.8 fF. Two identical air bridges are placed on the CPW transmission line to equalize the ground potential. The electromagnetic simulation is carried out in HFSS platform and all lumped parameters are extracted from the simulation. The FEM simulations are also verified with the circuit model-based ADS simulator platform. The switch inductance, capacitance (up-state) and resistance (down-state) can be found analytically from Equations 2.48 to 2.50 [1]. The switch up-state capacitance (Coff) can be found from the isolation response (S21), as given in Equation 2.48: 2

S21 ≈ 4ω 2Cu2Z02 (2.48)



L = 150 µm, 50 W Ls

L = 200 µm, 54 W

OFF-state

Rt

Cs

Ls

L = 150 µm, 50 W

Cb Rb1

Cp

Rs

ON-state

Vb Rb2 Bias network FIGURE 2.29 Equivalent circuit model of the MEMS switch. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

40

RF Micromachined Switches, Switching Networks, and Phase Shifters

The ON-state resistance of the switch (Rs) is the summation of the transmission line resistance and contact resistance (Rc). It can be found from the switch down-state return loss (S11), as given in Equation 2.49:

S11

2

2

 R  =  s  (2.49)  2Z0 

The inductance (Ls) of the switch can also be obtained from the down-state return loss (S11), as given in Equation 2.50: 2



 ωL  2 S11 =  s  (2.50)  2Z0 

The lumped parameter values from the equivalent circuit model are Ls = 85 nH, Cs = 9.3 fF, Rt = 0.03 Ω, Rs = 3.8 Ω, Rb1 = 25 kΩ and Rb2 = 30 kΩ, respectively. The switch is implemented on a CPW transmission line configuration (Z0 = 50 Ω, CPW length = 500 μm, transmission line width = 110 μm, CPW gap = 50 μm). The SPST switch insertion, return and isolation loss are shown in Figure 2.30. The simulated switch exhibits 28 dB of return loss, 0.25 dB of insertion loss and 16.6 dB of isolation from DC to 20 GHz frequency. The variation of isolation under different Δz is 16.5 dB to 18 dB from 0 to 1 μm of Δz under positive residual stress gradient. S-parameter measurements are carried out on the unpackaged device at room temperature and in a standard laboratory environment. The measurement has been performed using an Agilent 8510C vector network analyzer with GSG RF probes and a Cascade DC probe. The short-open-load-through (SOLT) method is used for the calibration. The switch is biased at 40–46 V for the ON-state measurement. The measurement was taken from 0.1 to 20 GHz, as shown in Figure 2.31. In the open-state, measured isolation is equivalent to 11 fF capacitance resulting in >17.5 dB of isolation at 20 GHz. In the down-state, the fitted switch resistance is 4.2 Ω and the switch is very well matched up to 50 GHz. The SPST

FIGURE 2.30 Simulated S-parameter performance of the MEMS switch. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

41

Behavioral Studies of Micromachined Contact Switches

FIGURE 2.31 Measured S-parameter response of MEMS switch. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

switch has a measured insertion loss of 21 dB up to 12 GHz. Now, different SPMT switch configurations are developed using this SPST switch and discussed in detail in the subsequent sections. RF measurement of all SPMT switches is carried out using the Agilent PNA series E8361C Vector Network Analyzer using Cascade DC probes. Calibration is done using the short-open-load-through (SOLT) method. The conditions are (a) ON-wafer and using a probe station; (b) room air; (c) standard laboratory pressure, temperature and humidity; (d) nonpackaged; and (e) no nitrogen blowing in the device. 3.3.2 Single-Pole Three-Throw (SP3T) Switch Design and Measurement The microscopic image of the single-pole three-throw (SP3T) switching network is shown in Figure 3.11a. All the three switches are placed at 90° and they are in-line in nature. Measured isolation characteristics of the SP3T shows better than 26 dB of isolation up to

56

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 3.11 (a) Microscopic image and (b) average S-parameter responses of the fabricated SP3T switch. (Used with permission of Institute of Physics and IOP Publishing Limited.)

20 GHz with 18 fF of capacitance, as depicted in Figure 3.11b. The switch demonstrates measured return loss of better than 22 dB and worst-case insertion loss of 0.35 dB up to 12 GHz with 62 V of bias voltage, as shown in Figure 3.11b. Total area of the SP3T switch is 0.43 mm2. 3.3.3 Single-Pole Six-Throw (SP6T) Switch Design and Measurement The microscopic image of the fabricated single-pole six-throw (SP6T) switch is shown in Figure 3.12a. All switches are placed here with an angle of 51.42°. All six switches are anchored at the center, and the center anchor radius is optimized to 18 µm for better matching without causing difficulties in fabrication. A junction capacitance (Cj) is introduced at the input line with a fixed to fixed beam structure. Cj is optimized using an FEM solver to improve the matching performance of the switches. The SP6T switch demonstrates measured return loss of >21 dB, worst-case loss of 18 dB up to 12 GHz, as depicted in Figure 3.12b. Total area of the SP6T switch is 0.58 mm2.

FIGURE 3.12 (a) Microscopic image and (b) S-parameter responses of the fabricated SP6T switch. (Used with permission of Institute of Physics and IOP Publishing Limited.)

Single-Pole Multithrow MEMS Switching Networks

57

FIGURE 3.13 (a) Microscopic image and (b) S-parameter responses of the fabricated SP7T switch. (Used with permission of Institute of Physics and IOP Publishing Limited.)

3.3.4 Single-Pole Seven-Throw (SP7T) Switch Design and Measurement The microscopic image of the fabricated single-pole seven-throw (SP7T) switch is shown in Figure 3.13a. The central anchor radius is optimized to 29 µm. Total area of the SP7T switch is 0.64 mm2. The switch shows measured return loss of better than 20 dB and insertion loss of ~14 dB up to 40 GHz, as depicted in Figure 3.15d. The switch isolation result is obtained when all eight switches are in the OFF-state that is 2–2.6 dB worse than the isolation with one switch in the ON-state condition, as depicted in Figure 3.15c.

Single-Pole Multithrow MEMS Switching Networks

59

FIGURE 3.15 (a) Microscopic image of the fabricated SP8T switch. Measured (b) return loss, (c) insertion loss and (d) isolation performances of the switch from 26 to 40 GHz. (Used with permission of Institute of Physics and IOP Publishing Limited.)

3.3.6 Single-Pole Ten-Throw (SP10T) Switch Design and Measurement Microscopic image of the single-pole ten-throw (SP10T) switch is shown in Figure 3.16a. Total area of the switch is 0.83 mm2. The switch demonstrates return loss of >15 dB, worst-case insertion loss of ~1.5 dB and isolation of 16.3 dB up to 12 GHz, as depicted in Figure 3.16b. 3.3.7 Single-Pole Eleven-Throw (SP11T) Switch Design and Measurement The fabricated image of the single-pole eleven-throw (SP11T) switch is shown in Figure 3.17a. The center anchor radius is optimized to 120 µm and switches are placed at an angle of 30°. The switch demonstrates measured return loss of better than 15 dB, worst-case insertion loss of 1.67 dB and isolation of >17 dB up to 12 GHz, as depicted in Figure 3.17b. Total area of the SP11T switch is 0.92 mm2. 3.3.8 Single-Pole Twelve-Throw (SP12T) Switch Design and Measurement Microscopic image of the single-pole twelve-throw (SP12T) switch is shown in Figure 3.18a. The central anchor radius of the switch is 144 µm and total area of the SP12T switch is 1.03 mm2. All 12 switches are placed at an angle of 27.7°. The switch demonstrates measured

60

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 3.16 (a) Microscopic image and (b) S-parameter responses of the fabricated SP10T switch. (Used with permission of Institute of Physics and IOP Publishing Limited.)

FIGURE 3.17 (a) Microscopic image and (b) S-parameter responses of the fabricated SP11T switch. (Used with permission of Institute of Physics and IOP Publishing Limited.)

FIGURE 3.18 (a) Microscopic image and (b) S-parameter responses of the fabricated SP12T switch. (Used with permission of Institute of Physics and IOP Publishing Limited.)

Single-Pole Multithrow MEMS Switching Networks

61

return loss of better than 15 dB, worst-case loss of 1.67 dB and isolation of >17 dB up to 12 GHz, as depicted in Figure 3.18b. 3.3.9 Single-Pole Fourteen-Throw (SP14T) Switch Design and Measurement The microscopic image of the fabricated single-pole fourteen-throw (SP14T) switch is shown in Figure 3.19a. Total area of the switch is 1.2 mm2. All 14 switches are anchored at the center, and the anchor radius is optimized to 178 µm. All SPST switches are placed at an angle of 24° to build the complete SP14T network. The switch shows return loss of >14 dB, insertion loss of 14.5 dB up to 12 GHz, as depicted in Figure 3.19b.

3.4 Equivalent Circuit Model of the SPMT Switch The equivalent circuit model of the SPMT switch is shown in Figure 3.20 and detailed parameters are listed in Table 3.2 for completeness. In this circuit, Lb represents inductance due to the cantilever beam, Cb is the parallel plate capacitance between the beam and the

FIGURE 3.19 (a) Microscopic image and (b) S-parameter responses of the fabricated SP14T switch. (Used with permission of Institute of Physics and IOP Publishing Limited.)

FIGURE 3.20 Equivalent circuit model of the SPMT switches. (Used with permission of Institute of Physics and IOP Publishing Limited.)

62

RF Micromachined Switches, Switching Networks, and Phase Shifters

TABLE 3.2 Circuit Parameters of the SPMT Switches (M = 3–14) Parameter

SPST

SP3T

SP4T

SP6T

SP7T

SP10T

SP11T

SP12T

SP14T

l1 (µm) l2 (µm) Cj (fF) Lb (pH) Rc (Ω) Coff (fF) Rbias (kΩ) Cb (fF) Vpi (V) FOM (THz)

156 156 NA 23 0.42 23 9.8 13.6 63 16.48

168 172 NA 26 0.68 22.4 9.4 13.2 63 14.45

190 196 3.4 ~27 0.87 23.4 9.7 14.4 64 7.82

212 218 3.66 27.8 1.02 22 ~10 12.8 64 7.09

216 228 3.86 28.6 1.18 22.8 9.8 13.77 64 5.91

244 250 4.06 32 1.48 21.8 9.2 ~14 67 4.93

264 278 4.13 37 1.6 ~23 9.6 13.96 65–68 4.32

272 293 4.32 41 1.77 22.7 ~10 14.12 67–70 3.96

286 302 ~4.4 46 1.82 24.12 10.2 14.08 67–70 3.62

fixed electrode, Coff is the switch OFF-state capacitance at zero-bias state, Rc is the contact resistance of the beam, Rbias is the bias resistance and Cp represents the parasitic capacitance.

3.5 Key Design Features of MEMS SPMT Switches After extensive measurements of the SPST switch, all identical switches are used to develop different switching networks. Each switch is actuated separately using an isolated pull-down electrode that makes DC contact with the output line. To compensate for the effect of impedance mismatch from the input to the output line, a junction capacitance (Cj) is introduced on the input line and optimized for all switching configurations. The SPMT switch parameters that vary with number of ports are mostly limited to (a) angle of switch separation, (b) central anchor radius, and (c) parasitic inductive effect from the central anchor and switches. In addition, nonuniform tip deflections (192 nm to 176 nm) were observed between switches due to different electroplated thickness (3.4 to 3.72 µm) of the gold cantilever beam. It also affects the switch performances with different values of Coff at the nonactuated state and different values of Rc with the same bias voltage at the ON-state. Note that demonstrated isolations were measured when all switches were in up-state, that is 2.7–3.2 dB worse than the isolation with one switch ON-state condition. The figure of merit (FOM) performances of the SPMT switch are obtained using Equation 3.3 and the same are listed in Table 3.2 for completeness:

FOM = f c =

1 (3.3) 2πRcCoff

3.6 Third-Order Intermodulation Intercept Point (IIP3) Measurements of MEMS SPMT Switches The third-order intermodulation intercept point (IIP3) of SPMT switches is measured using a two-tone test with f1 = 1.94 GHz and f2 = 1.98 GHz. The IIP3 measurement setup is shown in Figure 3.21, where PT is the transmitted power, PIN is the power at the input port of the

63

Single-Pole Multithrow MEMS Switching Networks

FIGURE 3.21 IIP3 measurement setup. (Used with permission of Institute of Physics and IOP Publishing Limited.)

DUT, the power coming out from the output port of the DUT is POUT and PR is the received power at the spectrum analyzer. PIN is defined by the power difference between PT and the loss between the VNA and the input port of DUT, and the value was ~8 dBm. POUT is the total loss between output port of the DUT and spectrum analyzer (~5 dB) and PR. The IIP3 can be calculated using Equation 3.4:

IIP3 =

∆P + PIN (3.4) 2

IIP3 is the fictitious input power for which the power of the sideband would be equivalent to the power of the input signal if all the input power is transmitted to the output, and Equation 3.4 is helpful to calculate the IIP3. In the in-line DC-contact switch, OFF- and ON-states capacitance and resistance values are very weak functions of nonlinearity. The change in capacitances and resistances are very less even when the RF voltage is applied to the electrode. The change in switch resistance by ohmic heating and mechanical motion of the beam are the primary reasons for the intermodulation distortion (IMD) for the switch. The input IIP3 of the MEMS switch can be calculated analytically using Equation 3.5:



1+ α 2 2  2 ( 2R + R0′ )   R     IIP3 =    2R + RR0′   R  CRR0′   

2

α

(3.5)

where R is the load and the source resistance is 50 Ω, R0′ is the average resistance variation, C is a constant that is independent of power but depends on signal frequency, and α is equal to 2.11. The single switch demonstrates measured IIP3 of 53 dBm at up-state, as shown in Figure 3.22a. Later, IIP3 measurement is carried out on different SPMT switching networks using the same setup. Measured average IIP3 values of the SPMT switch are 50 dBm and 14 dBm at up- and down-states, respectively, as shown in Figure 3.22b. A variation of IIP3 is found between 0.44–0.92 dBm from all SPMT switches, as depicted in Figure 3.22b. IIP3 values plotted in Figure 3.22b are the average values obtained from each port in up- and downstate conditions. Down-state performance is entirely driven by Rc. In summary, reasons for this deviation are mostly due to (a) angle of switch separation, (b) central anchor radius, and (c) parasitic inductive effect from the central anchor and switches. In addition, nonuniform tip deflections (192 nm to 176 nm) were observed between switches due to different electroplated thickness (3.4 µm to 3.72 µm) of the gold cantilever beam. The IIP3 was limited to the input/output transmission lines, contact resistance of the CPW probe tip and a different center anchor radius.

64

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 3.22 IIP3 measurements of the SPMT switches: (a) SPST switch and (b) SPMT switch with different RF output ports. (Used with permission of Institute of Physics and IOP Publishing Limited.)

3.7 Conclusion In this chapter, the design, development and characterization of different SPMT switching networks are presented. The maximum return loss of >14 dB, worst-case insertion loss of 1.76 dB and maximum isolation of >14.5 dB have been obtained from the SP14T switching configurations up to 12 GHz. The maximum area of the SPMT switching network is 1.2 mm2. It is thought that the switch performance could be further improved in hermetic condition to overcome the effect of surface charges trapped due to some residual humidity [21] and stiction due to the unclean environment.

References 1. S. Lucyszyn, Advanced RF MEMS, Cambridge University Press, August 2010. 2. A. Botula, A. Joseph, J. Slinkman, R. Wolf, Z.-X. He, D. Ioannou, L. Wagner, et al., A thinfilm SOI 180 nm CMOS RF switch technology, in Proceedings of IEEE Topical Meeting Silicon Monolithic Integrated Circuits in RF Systems, January 2009, pp. 1–4. 3. 4G Americas, 5G spectrum recommendations, August 2015. www.4​gamer​icas.​org/f​iles/​6514/​ 3930/​9262/​4G_Am​erica​s_5G_​Spect​rum_R​ecomm​endat​ions_​White​_Pape​r.pdf​. 4. S. Dey, and S. K. Koul, Reliability analysis of Ku-band 5-bit phase shifters using MEMS SP4T and SPDT switches, IEEE Trans. Microw. Theory Tech., vol. 63, no. 12, pp. 3997–4012, December 2015. 5. G. M. Rebeiz, RF MEMS Theory, Design, and Technology, John Wiley & Sons, 2003. 6. H. Zareie, and G. M. Rebeiz, Compact high-power SPST and SP4T RF MEMS metal-contact switches, IEEE Trans. Microw. Theory Tech., vol. 61, no. 8, pp. 2397–2402, January 2014. 7. A. Q. Liu, W. Palei, M. Tang, and A. Alphones, Single-pole-four-throw switch using highaspect-ratio lateral switches, Electron. Lett., vol. 40, no. 18, pp. 1281–1282, September 2008. 8. C. D. Patel, and G. M. Rebeiz, A high-reliability high-linearity high-power RF MEMS metalcontact switch for DC-40-GHz applications, IEEE Trans. Microw. Theory Tech., vol. 60, no. 10, pp. 3096–3112, October 2012.

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9. S. Dey, and S. K. Koul, Systematic measurement of high isolation DC–20 GHz miniature MEMS SPDT switch, Microw. Opt. Technol. Lett., vol. 58, no. 5, pp. 1154–1159, May 2016. 10. J. Lee, C. H. Je, S. Kang, and C. A. Choi, A low-loss single-pole six-throw switch based on compact RF MEMS switches, IEEE Trans. Microw. Theory Tech., vol. 53, no. 11, pp. 3335–3344, November 2005. 11. H.-H. Yang, A. Yahiaoui, H. Zareie, P. Blondy, and G. M. Rebeiz, Symmetric and compact single-pole multiple-throw (SP7T, SP11T) RF MEMS Switches, J. Microelectromech. Syst., vol. 24, no. 3, pp. 685–695, June 2013. 12. www.rfmd.com/product-category/switches. 13. Q. Chaudhry, R. Bayruns, B. Arnold, and P. Sheehy, A linear CMOS SOI SP14T antenna switch for cellular applications, in IEEE Radio Frequency Integrated Circuits Symposium, Montreal, 2012, pp. 155–158. 14. S. Dey, and Shiban K. Koul, Design and development of a CPW-based 5-bit switched-line phase shifter using inline metal contact MEMS series switches for 17.25 GHz transmit/receive module application, J. Micromech. Microeng., vol. 24, no. 1, November 2013. 15. S. K. Koul, S. Dey, A. K. Poddar, and U. L. Rodhe, Ka-band reliable and compact 3-bit TTD phase shifter using MEMS single-pole-eight-throw switching networks, J. Micromech. Microeng., vol. 26, June 2016. 16. H. S. San, X. Y. Chen, P. Xu, G. Li, and L. X. Zhan, Using metal insulator-semiconductor capacitor to investigate the charge accumulation in capacitive RF MEMS switches, Appl. Phys. Lett., vol. 93, no. 6, pp. 063506-1–063506-3, August 2008. 17. G. Li, H. S. San, and X. Y. Chen, Charging and discharging in ion implanted dielectric films used for capacitive radio frequency microelectromechanical systems switch, J. Appl. Phys., vol. 105, no. 12, pp. 124503-1–124503-6, June 2009.

4 Lateral MEMS Switches and Switching Networks The objective of this chapter is to study lateral types of radio frequency microelectromechanical systems (RF MEMS) series switches. Lateral switches have attracted considerable attention as the strongest alternatives to the conventional vertically driven MEMS switches owing to their high broadband isolation and no dielectric charging properties. Lateral switches contact or release with the RF circuits by in-plane motion of the beam. These switches are mostly operated by electrostatic actuation because of their large actuation forces and suitability for high-volume wafer-scale standard manufacturing processes [1]. The performance of the contact-type lateral MEMS switch is investigated by different research groups and most variants are implemented in the coplanar waveguide (CPW) transmission line [2–10]. All switches demonstrate in-plane design flexibility, reliable mechanical stability and high reliability. Lateral switches reported by Liu et al. [2] demonstrate return loss of >15 dB, insertion loss of 16 dB at 20 GHz within a 0.4 × 0.7 mm2 area using a cantilever beam. This switch is fabricated on a silicon-on-insulator substrate process based on the deep reactive-ion etching (DRIE) [2]. A non-contacttype lateral capacitive RF MEMS switch is reported using an air gap and it works over a specific frequency range [3]. Lateral switches based on the high-aspect-ratio electroplating nickel process have been reported by researchers [4,5]. In addition to these, lateral switches are also developed for high Q tunable capacitors, two-port and three-port switches, and different single-pole multithrow (SPMT) switching networks, [6–18]. All these switches are mechanically monostable as they remain typically OFF-state without any external bias. The organization of this chapter is as follows. First, RF design, circuit modeling, simulation and measurements of the single-pole single-throw (SPST) switch are presented. Later, different types of SPMT switching networks are designed using SPST switches utilizing the CPW transmission line. These include single-pole double-throw (SP2T), single-pole three-throw (SP3T), single-pole four-throw (SP4T), single-pole six-throw (SP6T) and singlepole seven-throw (SP7T) switching circuits. The advantages of switching networks are low loss, good matching and excellent isolation within a small footprint.

4.1 Design, Simulation and Measurement of the SPST Lateral Switch 4.1.1 Design of Lateral MEMS Switch Figure 4.1 illustrates the schematic of a basic SPST lateral RF MEMS switch. As shown in Figure 4.1, the lateral switch includes a coplanar waveguide, a cantilever beam extending between the first and second ports of the coplanar waveguide, and an electrostatic actuator for actuating the cantilever beam. The actuator is configured to apply a DC bias voltage between the cantilever and the ground line of the coplanar waveguide, thereby causing the free end of the cantilever beam to deflect in the direction of a fixed electrode. When 67

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Port - 1

Cantilever beam

Ground

Signal

Ground

Port - 2 FIGURE 4.1 Schematic of a lateral switch. (From S. Dey, and S. K. Koul, CARE IIT Delhi: RF MEMS 1 Internal Report, 2018.)

sufficient DC bias is applied, the cantilever beam deflects enough to contact a mechanical stopper of the second port, resulting in the closing (ON-state) of the switch. When the DC bias is lowered or removed, the beam returns to its atrest state, thereby opening the switch (OFF-state). One drawback of the lateral switch design is that it is prone to electromechanical failure after several switching cycles, especially under hot-switching conditions. For instance, the switch may fail due to static friction (or stiction) buildup between the cantilever beam and the mechanical stopper of the waveguide port. Furthermore, the spring constant of the cantilever beam is often too small to overcome the stiction. Therefore, there is a need for an improved design of the RF MEMS lateral switch that achieves improved wideband performance with improved repeatability (e.g., lifetime in the order of millions of switches) at lower microwave frequencies. More specifically, it is necessary to design an improved RF MEMS switch that is capable of switching a large number of ports in a small chip area, since area is directly proportional to cost in large-volume manufacturing processes. Figure 4.2 shows an example RF MEMS lateral switch. This lateral switch includes a coplanar waveguide, input and output ports, and a cantilever beam between the input and output ports. A mechanical spring is attached at the center of the cantilever beam. The mechanical spring has a semitriangular shape and is positioned between the cantilever beam and the ground of the waveguide. The mechanical force of the spring provides an additional force to move the cantilever beam back to its atrest position when the switch is in an OFF-state. The amount of mechanical force is selected so as to overcome any potential failure of the switch due to stiction while taking into consideration the effect of the electrostatic force induced when a bias voltage is applied. As in other in-line DC-contact cantilever switches, electrostatic actuation between the center line and ground causes the cantilever to move in a lateral direction toward the mechanical stopper of the second port. When the cantilever

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FIGURE 4.2 Schematic of a lateral switch with additional mechanical spring. (From S. Dey, and S. K. Koul, CARE IIT Delhi: RF MEMS 1 Internal Report, 2018.)

moves, it is necessary that the cantilever contact the second port of the center line without the mechanical spring contacting the ground line, since contacting the ground line would result in a short circuit of the switch. Therefore, a design constraint of the present design, and particularly of the mechanical spring, is that the atrest distance between the cantilever beam and mechanical stopper of the second port (a in Figure 4.2) should be significantly less than the distance between the mechanical spring and ground line (b in Figure 4.2), so that when a DC bias is applied, the cantilever beam contacts the mechanical stopper without the mechanical spring contacting the ground line. Figure 4.3 shows the equivalent circuit of the SPST lateral series switch. This model consists of two input and output CPW transmission lines with characteristic impedance (Z0) of 50 Ω and cantilever beam of resistance (R1); a cantilever beam line inductor (L) and switch series capacitor (Cs) at OFF-state or a ON-state switch contact resistor (Rc); and a shunt coupling capacitor (Cg). All equivalent circuit parameters can be extracted from S-parameters using Equations 3.48 to 3.50 (see Chapter 3). The lateral switch is based on an electrostatic actuator with suspended cantilever beam serving as a movable electrode, an anchor on the substrate to support the cantilever beam, a fixed electrode opposite to the cantilever beam and a contact bump. The thickness of the movable beam is ~3.5 µm and the material is gold. The electromechanical modeling and dynamic analysis of a lateral MEMS switch are given by Liu [19]. 4.1.2 Lateral Switch Characterization 4.1.2.1 SPST Lateral Switch Characterization The S-parameter performances of lateral switches are measured using a vector network analyzer (VNA) with Cascade Microtech ground–signal–ground coplanar probes with tungsten tip of 150 μm pitch. The system is calibrated using short-open-load-through (SOLT) on-wafer calibration technique. The SEM image of the SPST switch is shown in Figure 4.4a. A separate post is made for DC actuation. The SPST switch demonstrates

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FIGURE 4.3 Equivalent circuit of the lateral switch. (From S. Dey, and S. K. Koul, CARE IIT Delhi: RF MEMS 1 Internal Report, 2018.)

FIGURE 4.4 (a) SEM image and (b) S-parameter performance of the switch. (From S. Dey, and S. K. Koul, CARE IIT Delhi: RF MEMS 1 Internal Report, 2018.)

measured return loss of better than 17 dB, isolation greater than 30 dB and worst-case insertion loss of 2 dB up to 20 GHz, as depicted in Figure 4.4b. 4.1.2.2 SPDT Lateral Switch Characterization The SEM image of the SPDT switch is shown in Figure 4.5a. The SPDT switch includes a single cantilever beam with two triangular shape mechanical springs laterally attached to opposing sides of the cantilever beam. The free end of the cantilever beam is positioned to be able to deflect in either lateral direction so as to come in contact with a contact bump of either port 1 or port 2, depending on the direction in which the cantilever beam deflects. Deflection is determined based on the bias voltage applied to the actuators from each of the bias pads. At a given time, one of the actuators may be ON, while the other is OFF. Actuation and release of the cantilever beam may be aided by the mechanical spring on the side of the beam to which the beam deflects. The SPDT switch gives a measured return loss of better than 15 dB, isolation greater than 25 dB and worst-case insertion loss of 2.23 dB up to 20 GHz, as shown in Figure 4.5b. 4.1.2.3 SP3T Lateral Switch Characterization The SEM image of the SP3T switch is shown in Figure 4.6a. The input port of the SP3T lateral switch includes a central junction from which three separate cantilever beams extend.

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FIGURE 4.5 (a) SEM image and (b) S-parameter performance of the SPDT switch. (From S. Dey, and S. K. Koul, CARE IIT Delhi: RF MEMS 1 Internal Report, 2018.)

FIGURE 4.6 (a) SEM image and (b) S-parameter performance of the SP3T switch. (From S. Dey, and S. K. Koul, CARE IIT Delhi: RF MEMS 1 Internal Report, 2018.)

Each cantilever beam includes the same triangular-shape mechanical spring that is actuated by a separate actuator. Each actuator is biased by a separate bias pad. Note that, at a given time, one of the actuators may be biased, such that the cantilever beam associated with that actuator is deflected and contacts its corresponding output port. The input port and cantilever beams are uniformly distributed around the central junction. The SP3T switch demonstrates measured return loss of better than 15 dB, isolation greater than 32 dB and worst-case insertion loss of 2.6 dB up to 20 GHz, as shown in Figure 4.6b. 4.1.2.4 SP4T Lateral Switch Characterization The SEM image of the SP4T switch is shown in Figure 4.7a. The SP4T switch is similar in design to the SP3T switch in that each output port of the switch is connectable to a separate cantilever beam, the input port and cantilever beams are evenly distributed around a central junction, and each cantilever beam has its own mechanical spring, actuator and biasing pad to affect deflection of the beam. One drawback of the lateral switch design is that, with a large number of output ports, they do not achieve a wideband performance. In this switch configuration, all CPW ground planes are connected using gold bond wire

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FIGURE 4.7 (a) SEM image and (b) S-parameter performance of the SP4T switch. (From S. Dey, and S. K. Koul, CARE IIT Delhi: RF MEMS 1 Internal Report, 2018.)

just to equalize the ground potential as depicted in Figure 4.7a. Finally, the SP4T switch demonstrates a measured return loss of better than 27 dB, isolation greater than 35 dB and worst-case insertion loss of 3 dB up to 20 GHz, as shown in Figure 4.7b. 4.1.2.5 SP6T and SP7T Switch Characterizations The SEM images of the SP6T and SP7T switches are shown in Figure 4.8. Both switches are similar in design to the SP3T and SP4T switches in that each output port of the switch is connectable to a separate cantilever beam; the input port and cantilever beams are evenly distributed around a central junction; and each cantilever beam has its own mechanical spring, actuator and biasing pad to affect deflection of the beam. All adjacent ground planes are shorted with bond wires. Finally, both switches provide measured return loss of better than 13 dB, isolation greater than 18 dB and worst-case insertion loss of 3.67 dB up to 20 GHz, as shown in Figure 4.9.

4.2 Conclusion In this chapter, design, development and characterization of different lateral MEMS switching networks are presented. Matching and loss of MEMS lateral switching networks can be further improved by reducing the parasitic inductive effects caused by switches. These effects largely occur between the central junctions of adjacent switches. Parameters such as central junction length (as well as switch footprint and parasitic inductive effects) may be characterized using a full-wave simulation. The results of the full-wave simulation may then be utilized to modify the switch parameters, thereby improving or optimizing overall performance. In addition, CPW discontinuities (e.g., between adjacent switches) may include inductive bends. The purpose of these bends is to eliminate higher-order modes. The bias pads of the switches may also be routed in a manner that avoids signal leakage and other parasitic effects without affecting the switch performance. The bias pads and lines may themselves be made of a conductive material (e.g., titanium tungsten), and a film or layer of dielectric material (e.g., silicon dioxide) may be positioned between the bias lines and the CPW to prevent shorting. Another beneficial property of the configuration of the aforementioned example switches is their symmetry (e.g., equal angle between each

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FIGURE 4.8 SEM images of (a) SP6T and (b) SP7T switches. (From S. Dey, and S. K. Koul, CARE IIT Delhi: RF MEMS 1 Internal Report, 2018.)

FIGURE 4.9 Average S-parameter performances of the (a) SP6T and (b) SP7T switches. (From S. Dey, and S. K. Koul, CARE IIT Delhi: RF MEMS 1 Internal Report, 2018.)

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throw of a given switch). This configuration of the example switches permits them to be placed closer together (in designs that accommodate multiple switches). This means that a device with multiple MEMS RF lateral switches (e.g., a phase shifter) may be designed with greater compactness without any fabrication difficulties. The symmetry is especially beneficial for improving the compactness of the design. All switch configurations may lead to reduction of overall area of a device including these switches on the order of square microns or even square millimeters, as compared to other conventional topologies. The maximum return loss of >13 dB, worst-case insertion loss of ~3.7 dB and maximum isolation of >8 dB have been obtained from the SP7T switching configurations up to 20 GHz. Switch performances could be further improved in hermetic condition to overcome the effect of trapped surface charges due to some residual humidity and stiction due to the unclean environment.

References 1. S. Lucyszyn, Advanced RF MEMS, Cambridge University Press, August 2010. 2. A. Q. Liu, W. Palei, M. Tang, and A. Alphones, Single-pole-four-throw switch using highaspect-ratio lateral switches, Electron. Lett., vol. 40, no. 18, pp. 1281–1282, September 2008. 3. M. Tang, A. Q. Liu, A. Agarwal, Z. S. Liu, and C. Lu, A single-pole double-throw (SPDT) circuit using lateral metal-contact micromachined switches, Sens. Actuators Phys. A, vol. 121, pp. 187–196, 2005. 4. M. Tang, A. Q. Liu, and J. Oberhammer, A silicon-on-glass single-pole double-throw (SPDT) switching circuit integrated with silicon-core metal-coated transmission line, J. Micromech. Microeng., vol. 18, 9 pages, 2008. 5. A. Q. Liu, A. B. Yu, M. F. Karim, and M. Tang, RF MEMS switches and integrated switching circuits, J. Semicond. Technol. Sci., vol. 7, no. 3, pp. 166–176, 2007. 6. M. Tang, A. Q. Liu, and A. Agarwal, Lateral single-pole-double-throw (SPDT) MEMS switches for 50 MHz-20 GHz wideband applications, in Asia-Pacific Conference of Transducers and MicroNano Technology (APCOT) 2006, Singapore, June 25–28, 2006, Paper no. A-26. 7. M. Tang, A. Q. Liu, A. Agarwal, and Q. X. Zhang, A new approach of lateral RF MEMS switch, Analog Integr. Circuits Signal Process, vol. 40, pp. 165–173, 2004. 8. A. Q. Liu, W. Palei, M. Tang, and A. Alphones, Microstrip lateral RF MEMS switch integrated with multi-step CPW transition, Microw. Opt. Technol. Lett., vol. 44, no. 1, pp. 93–95, 2005. 9. M. Tang, A. Q. Liu, W. Palei, and A. Agarwal, A single-pole-double-throw (SPDT) circuit using deep etching lateral metal-contact switches, in IEEE MTT-S International Microwave Symposium Digest (IMS), Fort Worth, TX, June 6–11, 2004, pp. 581–584. 10. M. Tang, A. Agarwal, and A. Q. Liu, A low loss lateral micromachined relay using substrate transfer fabrication process, in International Conference on Material for Advanced Technologies (ICMAT), 2005. 11. M. Tang, A. Agarwal, Z. X. Shen, and A. Q. Liu, A high reliability and low loss lateral RF micromachined relay, in Asia-Pacific Conference of Transducers and Micro-Nano Technology (APCOT), July 4–7, 2004, pp. 1042–1047. 12. W. Palei, M. Tang, and A. Q. Liu, RF MEMS switch integrated with an enhanced back to back microstrip to CPW transition, in Asia-Pacific Conference of Transducers and Micro-Nano Technology (APCOT), July 4–7, 2004, pp. 479–482. 13. M. Tang, A. Agarwal, J. Li, Q. X. Zhang, P. Win, J. M. Huang, and A. Q. Liu, An approach of lateral RF MEMS switch for high performance, in Design, Test, Integration, and Packaging of MEMS/ MOEMS (DTIP), Cannes, 5–7 May 2003, pp. 99–102.

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14. M. Tang, A. Agarwal, P. Win, Q. X. Zhang, J. Li, and A. Q. Liu, A new lateral RF switch using SOI-deep-etching fabrication process, Int. J. Comput. Eng. Sci., vol. 4, no. 2, pp. 369–372, 2003. 15. X. M. Zhang, F. S. Chau, C. Quan, and A. Q. Liu, A study of the static characteristics of a torsional micromirror, Sens. Actuators Phys. A, vol. 1–2, pp. 73–81, 2001. 16. J. Oberhammer, M. Tang, A. Q. Liu, and G. Stemme, Mechanically tri-stable, true single-poledouble-throw (SPDT) switches, J. Micromech. Microeng., vol. 16, pp. 2251–2258, 2006. 17. J. Oberhammer, M. Tang, A. Q. Liu, and G. Stemme, Mechanically tri-stable in-line single-poledouble-throw all-metal switch, in 19th IEEE International Conference on Micro-electromechanical Systems (MEMS), Istanbul, Turkey, January 22–26, 2006, pp. 898–901. 18. A. Q. Liu, W. Palei, M. Tang, and A. Alphones, Single-pole-four-throw switch using highaspect-ratio lateral switches, Electron. Lett., vol. 40, no. 18, pp. 1125–1126, 2004. 19. A. Q. Liu, RF MEMS Switches and Integrated Switching Circuits, Springer, March 2010. 20. S. Dey, and S. K. Koul, CARE IIT Delhi: RF MEMS 1 Internal Report, 2018.

5 Micromachined Microwave Phase Shifters

5.1 Introduction A phase shifter is one of the most essential microwave components in modern phased arrays and provides a controllable phase shift of the radio frequency (RF) signal. The primary job of a phase shifter is to change the transmission phase of the input microwave signals with the electrical control signal applied to the component to get the phase shifted output signal. Based on the applications, phase shifters are broadly classified into two categories: active and passive. Besides providing controlled phase shift, active phase shifters provide gain, whereas passive phase shifters are lossy. The gain of the active phase shifter is defined when the phase shifter amplifies while phase shifting, whereas in passive phase shifters the loss is defined when the phase shifter attenuates while phase shifting. The active phase shifters are generally nonreciprocal in nature, whereas passive phase shifters can be both reciprocal as well as nonreciprocal. Phase shifters are widely employed in active or passive electronically scanned arrays that are based on phased antenna arrays. In addition, RF and microwave phase shifters find applications in subsystems such as phase discriminators, beam-forming networks, variable power dividers and linearization of power amplifiers. Phase shifters are also categorized into analog and digital formats based on their applications. Analog phase shifters provide a continuously variable phase shift or time delay. The most common semiconductor control elements used in analog phase shifters are varactor diodes and Schottky diodes. Varactor diode-based analog phase shifters provide larger phase shift and high speed and require fewer diodes than digital phase shifters, but at the cost of decreased accuracy, relatively narrow bandwidth and low input power levels (typically −3.5 dB for [( f0 − ∆f ) < f < ( f0 + ∆f )] for ∆f = 0.25 f0 (5.9b)

For such a splitter, the theoretical maximum coupling is –3 dB in each channel that is half power. The coupling characteristics are most sensitive to the gap width (s) and the metal thickness (t) for a given number of fingers (N) required for the coupler. The characteristics of the substrate are also an important factor for this coupler design. The requirements for return loss (S11) and transmission to the isolated port (S13) are commonly chosen less than some threshold in a frequency range (Δf) about f0 and these are defined as

S11 < −15 dB for [( f0 − ∆f ) < f < ( f0 + ∆f )] for ∆f = 0.25 f0 (5.10a)



S 31 < −15 dB for [( f0 − ∆f ) < f < ( f0 + ∆f )] for ∆f = 0.25 f0 (5.10b)

Note that the phase of the isolated port and input port are approximately 0 and –180°, respectively. The other two important parameters are the width of the bridge wire (W2) and the conductor width at the port (W1), as defined in Figure 5.4a. The width at the port is usually set by the associated circuit. The conductor width can be selected corresponding to the characteristic impedance near 50 Ω. The bridge wire configuration is determined by manufacturing considerations. The coupler bandwidth is assessed from its characteristics in terms of power split, phase relationship between the output ports, input VSWR and isolation as a function of frequency. When all these factors are taken into account, the useful bandwidth of the branch line coupler, the rat race coupler and parallel coupled backward wave coupler as 3 dB

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FIGURE 5.5 (a) Equivalent-circuit schematic of an analog reflection-type phase shifter. (b) 3D schematic and detailed dimension of offset air-gap overlay coupler used in the analog reflection-type phase shifter. (From S. Lee et al., IEEE Trans. Microw. Theory Tech., vol. 52, no. 1, pp. 211–219, January 2004.)

FIGURE 5.6 Schematic view of the two-bit reflection-type phase shifter with air-gap overlay CPW coupler and series MEMS switches. (From J.-H. Park et al., IEEE Trans. Microw. Theory Tech., vol. 11, no. 6, pp. 808–814, December 2002.)

hybrids is approximately 10%, 20% and 35%, respectively. It may be noted that the branch line and parallel coupled backward wave couplers are inherently 90° hybrids, whereas the rat race coupler is a 180° hybrid. Some researchers have reported low-loss couplers operating at 40 and 60 GHz with overlapping transmission line using a micromachining process, as shown in Figures 5.5 and 5.6, respectively [19,20]. In the overlay coupler, the air-gap offset broadside coupling

Micromachined Microwave Phase Shifters

87

between the two vertically separated lines offers tight coupling and reduced conductor loss by redistributing the currents over broad surfaces. This results in improved bandwidth and low-loss characteristics when compared with the Lange couplers. The threedimensional (3D) schematic and the detailed dimensions of the air-gap overlay coupler are shown in Figure 5.5b. This coupler provides losses of the order of 0.38 dB at 40 GHz and 0.66 dB at 60 GHz. Raytheon has reported 2-bit and 4-bit reflection-type phase shifters at X-band on a 500 µm thick silicon substrate using microstrip line and MEMS shunt switches [21]. A Lange coupler is used here that limits the phase shifter bandwidth between 7 and 11 GHz. This 4-bit phase shifter demonstrates –1.5 dB of loss in the frequency band 8–10 GHz and FOM is 0.28 dB/bit at 8 GHz. Total area of this phase shifter is 100 mm2 including Lange couplers and transmission line connections, as shown in Figure 5.7. An MEMS SP3T switch and reflect stub were used to develop the 5-bit X-band phase shifter by HRL Laboratories [22]. It was fabricated on low dielectric constant substrate and the area of the device is large. The 11.25°, 22.5° and 45° delay bits are designed and fabricated using a Lange coupler and reflect stubs whose electrical length is just half of the desired delay bits at the design frequency. The 90° and 180° delay bits are built using another Lange coupler. Later, HRL MEMS series switches are placed at the appropriate location with silver epoxy and bond wires. The average insertion loss of this phase shifter is 1–1.5 dB in the frequency band 7–10 GHz with return loss of better than 10 dB. The FOM of this phase shifter is 0.2–0.3 dB/bit. This design is very useful for X-band, but difficult to implement at K-band and higher frequencies. 5.5.2 Switched-Line Phase Shifter A switched-line phase shifter is the easiest way to implement a microwave phase shifter. It employs a delay line and two back-to-back SPDT switches as shown in Figure 5.8. The working principle of the switched-line phase shifter is dependent only on two lengths of

FIGURE 5.7 Photograph of 4-bit reflection type phase shifter developed by Raytheon. (From A. Malczewski et al., IEEE Microw. Guided Wave Lett., vol. 9, no. 12, pp. 517–519, December 1999.)

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FIGURE 5.8 Design schematic of a switched-line phase shifter.

transmission lines (reference and delay lines) that are being switched. The differential path length between the two line segments determines the amount of phase shift that can be achieved. An important advantage of the switched-line phase shifter configuration is that its phase frequency response is nearly linear. Phase shift is primarily dependent on transmission line lengths based on the switching action and making the phase shifter very stable over time and temperature . MEMS switches are used to route the input RF signal into the appropriate length of matched transmission lines. The phase shift (Δϕ) of each bit is given by [1]

∆φ =

2π (ld − lr ) 2πf ε eff (ld − lr ) = (5.11) λ c

where ld and lr are the lengths of delay lines and reference lines, respectively; f is the operating frequency; c is the velocity of light; and εeff is the effective permittivity. The phase shift value deviates linearly from the intended value as the frequency of the signal deviates in either direction from the center (nominal) frequency. Note that switched-line phase shifters are generally used for higher phase bits like 180° and 90°. When path ld is a half wavelength (λ/2) longer than path lr, switching from path lr to path ld introduces an increased phase delay of 180°. So the required physical length difference should be Δl = λ/2 to obtain a 180° phase shift. In a practical design resonances could appear in the OFF-state (in spite of the types of control elements) when the line length is a multiple of λ/2, and the phases will interfere in a way to reflect much of the incoming power back to the input port. Thus, both lengths (ld and lr) must not be multiples of λ/2. The resonant frequency will be slightly shifted due to the series junction capacitances of the reversed biased diodes (for PIN diodes) or of the parasitic capacitances of the SPDT MEMS switches. So to get optimum performance of the phase shifter, the lengths ld and lr must be carefully selected to avoid phase errors, high return loss and high insertion loss. The Schiffman phase shifter is a good choice for flat wideband phase shift response. This phase shifter was invented by Bernard Schiffman in 1958 using a λ/4 wavelength coupler section. Figure 5.9 shows a schematic of a Schiffman phase shifter that uses a switched line with two SPDT switches, one reference regular line of length 3l, and two other parallel coupled line of equal length l = λ/4, directly connected to each other at one end. The phase shift function is determined by the phase difference of signals transmitted through the coupled section of length l and the reference line of length 3l. By proper selection of the

Micromachined Microwave Phase Shifters

89

FIGURE 5.9 Schematic of a Schiffman phase shifter.

length of these lines and the degree of coupling, the phase difference between them can be made to be almost constant over a broad bandwidth. An N-bit phase shifter is made easily by cascading each 1-bit phase shifter with different delay line lengths. The phase delay is obtained by activating the required number of bits and it provides true variable time delay. A switched-line-type phase shifter can be fabricated using MEMS series or shunt switches. Conventional switched-line topology generally used SPDT switch, and bandwidth of the structure is inherently wider. The size of the structure can also be reduced by adding multiple delay lines using a single-pole multithrow (SPMT) switching network. It drastically reduces the overall phase shifter size because delay lines and switches are closely packed together. It also improves the overall insertion loss in the device as the RF signal travels through fewer switches. Figure 5.10 shows two different 2-bit phase shifters using SPDT and SP4T switching networks. Conventionally, a 5-bit switched-line phase shifter uses 20 DC-contact MEMS switches, and the total area of the phase shifter is 36 mm2 [23]. Total loss of this phase shifter is 5.4 dB with a phase error of 1.4° at 17 GHz. Later, an improved version of the 5-bit phase shifter is developed using four SP4T and two SPDT MEMS switches [24]. Compared with the conventional switched-line phase shifter, in which a minimum of ten switches are actuated at a time, this design requires only six switches to be actuated at a time to activate one phase state for 5-bit operation. It drastically reduces insertion loss of the overall phase shifter by 20%–40%. The loss of the overall phase shifter is 2.65 dB and the total area of the phase shifter is 13 mm2, which is almost one-third of the conventional switched-line phase shifter [24]. A 4-bit MEMS phase shifter with four SP4T switches with 21 mm 2 area reported by Rebeiz provides an average insertion loss of 1.1 dB and phase accuracy of 2.3° at 10 GHz using microstrip transmission line on silicon substrate [25]. A 6-bit phase shifter was demonstrated [26] with 2.8 dB loss at 18 GHz over 40 mm2 area using this switched-line topology. All these switch configurations are very much favorable for switched-line phase shifter design because they permit the switches to be placed very close to one another without any fabrication difficulty and they can be operated up to Ka-band frequency. 5.5.3 Loaded-Line Phase Shifters A loaded-line phase shifter is another suitable topology for digital phase shifter design. A loaded-line phase shifter is the most preferred approach when a small phase change (typically up to 45°) between two states (defined by the control signals) is required. The basic

90

RF Micromachined Switches, Switching Networks, and Phase Shifters

180º

270º 90º 90º

0º 180º

(a)

(b)

FIGURE 5.10 2-Bit switched-line phase shifters based on (a) 2-line sections using SPDT switches and (b) 4-line sections using SP4T switches.

BT,

B1 3

B1

B2

B2

FIGURE 5.11 Concept of a loaded-line phase shifter.

circuit is shown in Figure 5.11. The basic form of the loaded-line phase shifter is to load a transmission line with two different impedances and use a midsection matching network, which ensures that the phase shifter is matched to Z0 for both loading conditions. MEMS series or shunt reactive loads are connected at both ends of the load. The loading impedance of the circuit is controlled by the phase difference between two different loads. If one can assume that the phase shifter is lossless and is matched at the design frequency for both loading conditions (assuming pure reactive loads), the following results are obtained for the case of shunt loading:



 cos θ  B1, 2 = Y0  ± tan ( ∆θ / 2 ) (5.12)  cos ( ∆θ / 2 )  ZT = Z0

cos ( ∆φ / 2 ) (5.13) sin θ

91

Micromachined Microwave Phase Shifters

where Y0 is the admittance of the input/output ports, B1,2 are the switch susceptances, ZT and θ are the impedance and electrical delay of the midsection transmission line, and Δϕ is the required phase shift. Similar equations are also applicable for the series switch, but these are rarely used in this kind of phase shifter. The reason for this is that switchable inductive loads are difficult to implement in terms of biasing and providing heat sinks [27]. Three different classes of loaded line phase shifters were identified by Opp and Hoffman in 1974 [28]: Class I—It is a general case where two values of Bi are used where B1 ≠ B2 and B1,2 ≠ 0; it corresponds to phase displacements that are nonzero and nonequal. Class II—B1 or B2 is zero. With B1 = 0, θ = (π ± Δϕ)/2, YT = Y0, and B2 = ±Y0 tan (Δϕ/2); this class is called the load/unload case, because it can be easily obtained using a load that is applied and removed from the transmission line. Class III—B1 = −B2, which is complex conjugate condition, and θ = 90°. The loading changes the phase by Δϕ/2 around θ = 90°, and this condition is easy to implement using MEMS switches. A standard implementation of Class I and Class III loaded-line phase shifters is shown in Figure 5.12. Class I and Class III designs are widely implemented for the loaded-line phase shifter. The phase response obtained from the Class III design is more constant, which leads to larger phase shifter bandwidth. Moreover, Class III provides minimal and constant insertion loss performance and it mostly depends on the load Q factor. The relative phase shift changes very slowly with lower Q factor values due to lossless load components. As θ in Class III design decreases, the average insertion loss and its variation increases, which creates problems in some applications. Due to the larger line lengths in the Class III case, the average difference in loss between Class I and Class III may be smaller if lossy transmission lines are used. In comparison, Class II operation is not good to implement with FET switches but can be implemented with RF MEMS switches due to their low off-state capacitance and ease of design with reasonably good performance. Bandwidth obtained from Class II is pretty low for phase bits larger than 22.5°, whereas Class III provides superior performance in terms of insertion loss, return loss bandwidth and phase bandwidth. A loaded-line phase shifter is more favorable for operation in digital mode due to large load variations that are usually needed and this approach is limited to Δϕ ≤90°. Note that it is believed that no loaded-line phase shifter has been implemented using RF MEMS technology. Also, note that the restrictions in the susceptance values dictated by the matching BT,

BT, Capacitive

1

Zs

2

Zs

or DC-contact MEMS switch

Zs,

(a)

1

(b)

FIGURE 5.12 Standard implementation of (a) Class I and (b) Class III phase shifters.

92

RF Micromachined Switches, Switching Networks, and Phase Shifters

conditions over the bandwidth enable phase shifts only up to 45° by loaded-line phase shifters. Furthermore, if the circuit size should be small, then the minimum frequency of operation is limited since the line length is generally of the order of λ/8. 5.5.4 Low-Pass/High-Pass Network Phase Shifter Low-pass/high-pass phase shifters are composed of low-pass and high-pass filter structures. Shunt inductors and series capacitors form high-pass filters, whereas series inductors and shunt capacitors form low-pass filters. Phase shift is obtained by switching between the two filter circuits, which are shown in Figure 5.13. Low-pass/high-pass phase shifter topology is ideal for broad bandwidth and the small size required for monolithic transmit/receive (T/R) modules. The low-pass LC filter results in insertion phase delay, while a high-pass filter results in phase advance. The input/output impedance of both filters is chosen to be Z0 at the design frequency. The reactive elements Xn and Bn of the T-network (Figure 5.13) are determined to obtain a specific value of Δϕ. For this LC T-network implementation, the primary design equations are given as [29]

 ∆φ   ∆φ  Bn = sin   ; Xn = tan   (5.14)  2   4 

where Xn = ωL/Z0, Bn = ωCZ0 for the low-pass network, and Xn = 1 / ωCZ0, Bn = Z0/ωL for the highpass network. Bn can also be written in terms of Xn as given next:

Bn =

2X n (5.15) Xn2 + 1

The same relations can also be applied to the π-network except that Bn is replaced by Xn and vice versa. The insertion phase of these circuits varies from 0 50 million, >15 million and ~1 million cycles of operations with 0.1, 0.5 and 1 W of RF power, respectively. The switch shows reversible or elastic deformation over these cold-switched reliability processes. Later, the MEMS switch was subjected to a 5 h stress relaxation or creep process under prolonged actuation conditions at room temperature. Results show a 4–6 V shift in Vpi over the time. Plastic deformation was also observed over the time during the

113

Digital MEMS Switched-Line Phase Shifters

VNA PC Phase shifter

5 V supply

GUI Driver circuit FIGURE 6.14 Schematic of the reliability measurement setup for the 5-bit phase shifter. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

continuous pitting and hardening process. Contact surface asperities followed by material deformation are major sources of failure in the switch. Furthermore, the contact resistance (Rc) variation of 3.34% was observed during this process due to thermal effect and hardening. Finally, 5-bit phase shifter reliability was checked under the same operating condition and with the hot-switched method. The phase shifter was diced in the form of a chip and encapsulated within a module or a test jig, and extensive reliability measurement was carried out. The reliability measurement setup is shown in Figure 6.14. The bias (30–40 V) is given to the appropriate bias location using graphical user interface (GUI) from a local PC and corresponding changes in phase shifter average loss and phase error are recorded after every 30 min over the 17–17.5 GHz band. It takes 4 min to complete one cycle (32 states) of the 5-bit phase shifter where a 5 s delay is introduced between two consecutive phase states from the GUI. This operation is repeated with 0.1, 0.5 and 1 W of RF power levels. Results show the phase shifter works satisfactorily up to ~8 K cycles under 0.1 W of RF power with ~7 dB average loss and ~±4° average phase errors, respectively. Moreover, the phase shifter demonstrates ~1.5 K cycles and ~0.7 K cycles with 0.5 W and 1 W of RF power levels, respectively, as shown in Figure 6.15. Finally, current density at the contact region followed by localized high-temperature spots during contact bouncing are major

FIGURE 6.15 Measured cold-switched reliability results of the 5-bit MEMS phase shifter with 0.1–1 W of incident RF power at 17.25 GHz frequency. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

114

RF Micromachined Switches, Switching Networks, and Phase Shifters

sources of failure in this metal-contact switch-based 5-bit phase shifter. To overcome these effects, well-suited contact materials like rhenium, rhodium or gold–palladium alloys can be used [10]. Furthermore, a controlled bipolar voltage waveform can be applied to the actuation electrodes to overcome the contact heating. There are a few aspects where more care can be taken to improve the reliability of the switched-line digital phase shifter. The first is definitely the asymmetric switch actuation due to the nonuniform gap profile. In this 5-bit phase shifter, 10 switches are actuated simultaneously. So, the nonuniform gap profile (0.6–0.91 µm) affects the device performance. The actuation voltage of 11 states out of 32 states was more (~42 V) due to added capacitances the between delay or reference line and the ground plane, and nonuniform overall height (g0). A stiffer beam can be used to overcome the variations in actuation voltage, but that inevitably leads to higher actuation voltage. Undesirable deformation can be observed and removed by a tensile–compressive stress converter structure. The stress gradient can also be minimized using a low temperature release process.

6.8 Conclusion In this chapter, design, fabrication and characterization of the 5-bit CPW-based switchedline phase shifter was demonstrated. The phase shifter uses 20 in-line metal-contact MEMS switches and 10 switches are activated at a time. All functionalities of an in-line MEMS metal-contact switch were presented. Two 5-bit switched-line MEMS phase shifters were proposed and implemented at 17.25 GHz. The performance comparison between the two phase shifters was extensively discussed, leading to a 25% length reduction between them. The design achieves 14 dB of return loss, 5.57 dB of insertion loss and ±1.25° of phase error with 30–40 V of bias voltage. A switch and phase shifter reliability study was discussed under the cold-switched condition and the device has been shown to be reliable up to 8 K (1.5 K) cycles with 0.1 W (0.5 W) of incident RF power level. The device reliability and power handling can be extended with packaging in a clean environment.

References

1. W. M. Zhang, R. P. Hsia, C. Liang, G. Song, C. W. Domier, and N. C. Luhmann Jr., Novel lowloss delay line for broadband phased antenna array applications, IEEE Microw. Guided Wave Lett., vol. 6, pp. 395–397, November 1996. 2. N. S. Barker, and G. M. Rebeiz, Optimization of distributed MEMS phase shifters, in IEEE MTT-S International Microwave Symposium Digest, 1999, pp. 299–302. 3. A. Borgioli, Y. liu, A. S. Nagra, and R. A. York, Low-loss distributed MEMS phase shifter, IEEE Microw. Guided Wave Lett., vol. 10, pp. 7–9, January 2000. 4. Jae-Hyoung Park, Hong-Teuk Kim, Wooyeol Choi, Youngwoo Kwon, and Yong-Kweon Kim, V-band reflection-type phase shifters using micromachined CPW coupler and RF switches, J. Microelectromech. Syst., vol. 11, no. 6, December 2002. 5. H. Zhang, A. Laws, K. C. Gupta, Y. C. Lee, and V. M. Bright, MEMS variable-capacitor phase shifters part I: Loaded-line phase shifter, Int. J. RF Microw. Comput.-Aided Eng., vol. 13, no. 4, May 2003.

Digital MEMS Switched-Line Phase Shifters



115

6. J. S. Hayden, and G. M. Rebeiz, Low-loss cascadable MEMS distributed X-band phase shifters, IEEE Microw. Guided Wave Lett., vol. 10, pp. 142–144, April 2000. 7. K. Topalli, O. A. Civi, S. Demir, S. Koc, and T. Akin, A monolithic phased array using 3-bit distributed RF MEMS phase shifters, IEEE Trans. Microw. Theory Tech., vol. 56, no. 2, pp. 270–277, February 2008. 8. G. M. Rebeiz, RF MEMS: Theory, Design, and Technology, John Wiley & Sons, 2003. 9. S. Lucyszyn, Advanced RF MEMS, Cambridge University Press, August 2010. 10. G. McFeetors, and M. Okoniewski, Distributed MEMS analog phase shifter with enhanced tuning, IEEE Microw. Wirel. Compon. Lett., vol. 16, no. 1, January 2006.

7 DMTL Phase Shifter Design Using MAM Capacitors and a MEMS Bridge

7.1 Introduction In this chapter, design, analysis, fabrication and measurement of another version of a 5-bit phase shifter using distributed MEMS transmission line (DMTL) topology is presented. The device that is fabricated using the same surface micromachining process demonstrates good repeatability with excellent phase accuracy. To improve the loss of the overall phase shifter in higher-bit configurations, metal–air–metal (MAM) or metal–insulator– metal (MIM) capacitors are used in the literature [1,2]. The overall high quality factor of MAM capacitance can provide better phase shift per decibel noise (degree/dB) compared to MIM capacitance over the band of interest. Furthermore, MAM capacitance is easier to realize compared to MIM capacitance [2]. In this work, a phase shifter is built with three fixed-to-fixed beams; one is switchable with electrostatic actuation and the other two are fixed for a MAM capacitor. A simple stiffer fixed-to-fixed beam is used for switching, which substantially improves the overall reliability of the 5-bit digital phase shifter. An overall performance evaluation of the switching beam—including mechanical, electrical, transient, power handling, linearity, loss and reliability measurements—is extensively carried out and reported in this chapter. Later, individual primary phase bits (11.25°, 22.5°, 45°, 90°, 180°), which are fundamental building blocks of a complete 5-bit phase shifter, are fabricated separately, tested and presented followed by the complete 5-bit phase shifter. Reliability performance of the MEMS 5-bit phase shifter is also demonstrated.

7.2 Unit Cell Phase Shifter Design and Modeling The DMTL phase shifter works on the principle of dispersion where an unloaded line is loaded with multiple distributed capacitances in a periodic manner. The changes in loaded capacitance on the line with applied bias contribute to a change in phase velocity, which in turn produces differential phase shift. In order to achieve acceptable matching over a wideband, it is always recommended not to model a DMTL structure with high downstate capacitances. A unit section of the DMTL phase shifter has been optimized with different structural parameters. Figure 7.1a shows the microscopic view of the unit cell phase shifter and Figure 7.1b shows its equivalent circuit model. A MEMS bridge is placed on 117

118

RF Micromachined Switches, Switching Networks, and Phase Shifters

Z0,

eff,

l = s/2

Z0,

eff,

l = s/2

C b/C bd

MEMS Bridge

Lb Self resistance of MAM capacitors

Rbu/Rbd MAM capacitor

(a)

C s/2

(b)

C s/2

Rp Rb

External bias resistors

Rb

Re

Internal bias resistors, one is used for unitcell and 0 two used for 11.25 cell

FIGURE 7.1 (a) SEM image and (b) equivalent circuit model of the unit cell DMTL phase shifter. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

the signal line, which varies the loaded capacitance on the transmission line based on the actuation voltage. The aim of the design is to achieve desired phase shift with minimum insertion loss and good matching (>10 dB) over the X-band. In the proposed design, two MAM capacitors are in series with a MEMS bridge. The MAM capacitor exhibits low capacitance per unit area and high quality factor compared to MIM capacitor. In the proposed scheme, bias can be applied to the MEMS bridge without affecting the MAM capacitor. The total loaded capacitance (Cl) seen by the line is the series combinations of MAM capacitances (Cs) and bridge capacitances (Cb) at zero-bias condition and it is given by Equation 7.1: Cl =



CsCb (7.1) Cs + Cb

At zero-bias state, Cb is much lower than Cs, and the effective loaded capacitance (Clu) seen by the line is Cb. When the MEMS bridge is in the down-state position, the bridge down state capacitance becomes much larger than Cs, thereby the effective loaded capacitance (Cld) is given by Cs. The loaded transmission line (t-line) up-state (Zlu) and down-state (Zld) impedances are given by Equation 7.2:

Zlu =

sLt Ω , Zld = sCt + Cb

sLt Ω (7.2) sCt + Cs

where s is the unit section length, and sLt and sCt are the per unit line inductance and capacitance. Furthermore, s can be calculated by Equation 7.3 [1]:

s=

Zld c meter (7.3) πfB Z0 ε r ,eff

119

DMTL Phase Shifter Design Using MAM Capacitors and a MEMS Bridge

Here, Z0 and c / ε r ,eff are the characteristic impedance and guided velocity of the unloaded high impedance CPW line. The Bragg frequency ( f B) is selected to be 3 times the operating frequency (10 GHz). The f B decides the highest operating limit of a phase shifter after which no power will transfer and impedance will become zero. Equations 7.4 and 7.5 are used to calculate Cb and Cs and are summarized here for the sake of completeness [1]:

2 0

Z

Cs =



( (Z

2 2 Zld Z02 − Zld

Cb = Cs



2 lu

2 ld

−Z

) farads (7.4) )

2 Z02 − Zld farads (7.5) 2 πfB Z0 Zld

The phase shift per unit section can be obtained with the change in phase velocity due to change in loaded characteristic impedance with applied bias and is calculated by Equation 7.6 [1]:

∆φ =

sωZ0 ε r ,eff  1 1  − rad/sec (7.6)   Zlu Zld  c

where ω is the frequency in radians, c is the free space velocity and εr,eff is the relative effective dielectric constant of the unloaded transmission line. Two air bridges are placed at the input and output where 50 Ω impedance changes to the high impedance line for DMTL operation. These air bridges are able to equalize the ground potential and to overcome the effect of any unwanted modes for proper phase shifter operation. Two long air bridges are also placed at the unit section to improve the matching performance (~7 dB) of the phase shifter. The relevant design parameters of the unit cell phase shifter are optimized with S-parameter response using HFSS v13. Here, in this design, all bias lines are covered with dielectric. It substantially improves the design simplicity where the bias line can be routed anywhere without cutting the ground plane. So, the effect of added extra parasitic and RF leakages will be less on phase shifting operation. Table 7.1 summarizes the dimensions of all designed parts used to build a unit cell phase shifter. The total loaded capacitances seen by the line between normal state (Cn) to actuated state (Ca) can also be defined as

Cn = Ct +

CsCb CC , Ca = Ct + s db (7.7) Cs + Cb Cs + Cdb

TABLE 7.1 Dimensions of the Unit Cell Phase Shifter Parameters Signal line width (W) CPW gap (G) MEMS bridge length (lb) MEMS bridge width (wb) MEMS bridge thickness (tb) MAM capacitor length (lm)

Value (µm) 100 140 200  65   2  92

Parameters MAM capacitor width (wm) Air bridge length (la) Air bridge width (wa) Fixed electrode width (we) Length of the unit cell (s) Electrode thickness (te)

Value (µm)  20 370  15  35 300   2

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RF Micromachined Switches, Switching Networks, and Phase Shifters

where Cb and Cdb are MEMS bridge up-state and down-state capacitances, respectively. The Cb and Cs capacitances are optimized to be 8 fF and 24 fF for acceptable impedance matching over the band of interest. The total inductance (Lb), including of the MEMS bridge and MAM capacitors, is 90 pH. Although the unit cell demonstrates a capacitance ratio (Ca/Cn) of 13, the effecting Cs/Cb of 3 plays a role in achieving the desired phase shift from the cell. The series resistance of the bridge (Rb) and the Q factor of Cs play a significant role in the phase shifter operation. The nonmovable static capacitor (MAM capacitor) exhibits high Q factor (>2000 at 10 GHz). The loss of the structure also depends on MEMS bridge resistance in the up-state (Rbu = 1.1 Ω) and down-state (Rbd = 0.6 Ω). The Q factor of the MAM capacitor and loss of the phase shifter are also affected by the bias resistances (Rb). The Q factor of the MAM capacitor is found to be 2123 with a self-resistance (Rp) of 25 kΩ. Figure 7.1a shows a TiW bias line that is placed inside the gap of the CPW and extended outside to connect with the bias pad. An internal TiW bias line is connected in shunt with Rp of MAM capacitor. Hence, the overall Q factor is reduced to 473 from 2123 due to parallel combinations of Rp (25 kΩ) and Rb (20 kΩ) as shown in Figure 7.1b. When two unit cells are connected with bias lines, then the equivalent resistor of the MAM capacitor reduces to Rp||(Rb/2) Ω and the Q factor again reduces to 367 from 473. The resistivity of the TiW bias line is optimized to be 380 Ωmeter, which is an essential parameter to determine the overall loss of the phase shifter. The down-state characteristics change primarily with bias lines. The insertion loss of the unit cell is 0.045 dB without any bias line at 10 GHz, which increases to 0.07 dB due to the reduced Q factor of Cs. The external bias resistor (Re) that is routed underneath the CPW ground plane is only used to excite individual bits separately. The effective Re is very small (0.9 KΩ) due to stronger coupling between the ground plane and external bias resistor underneath. So, the Q factor of the MAM capacitor reduces accordingly to 58 from 367 for those five unit cells that are connected externally with the bias pads for the 5-bit operation of the phase shifter. The insertion loss is also increased to 0.21 dB due to the effect of Re. Return loss is not significantly affected by the bias lines. It depends on Cb and Cs along with unloaded t-line parameters and can be determined from the loaded line impedances Zld and Zlu using Equation 7.2. In this work, the ranges of the loaded characteristic impedances were designed to be 64 Ω (Zlu) and 49 Ω (Zld) for optimum phase shifter operation. The unit cell demonstrates simulated return loss of better than 33.9 dB and insertion loss of ~0.07 dB at low impedance state (49 Ω) and with bias lines over the 8–12 GHz band, as shown in Figure 7.2a.

FIGURE 7.2 Simulated (a) S-parameter and (b) phase shift versus frequency response of the unit cell. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

DMTL Phase Shifter Design Using MAM Capacitors and a MEMS Bridge

121

FIGURE 7.3 Complete schematic of 5-bit DMTL phase shifter where bias lines are indicated with red lines. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

Furthermore, the unit cell gives a simulated differential phase shift of 7° at 10 GHz, as shown in Figure 7.2b. The adverse effect of surface roughness, nonuniform deformation of the beam due to residual stress and any unwanted resonance can degrade the performance of the phase shifter after fabrication. Due to this, differential phase shift can be reduced at the desired frequency point of interest. This is the reason why designed phase shift has been kept intentionally higher than the actual value (5.625°). After the design of a unit cell phase shifter, 62 unit cells were placed with a periodic placement of 300 μm to build a complete 5-bit phase shifter. The schematic of the complete phase shifter including all bias lines and bias pads is shown in Figure 7.3. All individual primary phase bits (11.25°, 22.5°, 45°, 90° and 180°) were designed, fabricated and tested individually using 2, 4, 8, 16 and 32 unit cells, respectively.

7.3 Fabrication and Measurements The unit cell phase shifter, individual primary phase bits and complete 5-bit phase shifter were fabricated using the same surface micromachining process. Initially, unit cell performance of the phase shifter was extensively characterized including electromechanical performances of the MEMS switching beam, electromagnetic, power handling and reliability analysis. Electromagnetic performance of the primary cells and the complete 5-bit phase shifter were systematically measured and these are discussed under the following sections. 7.3.1 Unit Cell Phase Shifter Measurements and Results The microscopic image of the unit cell is shown in Figure 7.1. The optical profilometer shows that MAM capacitors and the MEMS bridge undergo extra out-of-plane deformation of 0.11 μm and 0.4 μm (initial air gap was 2.5 μm), respectively. It was mostly due to the residual stress, which was developed on the film sacrificial layer when deposition temperature was lower than the flow temperature. The pull-in and release voltages of the MEMS bridge were measured to be 53 V and 34 V, respectively, as shown in Figure 7.4a. Figure 7.4b presents the measured mechanical resonance frequency as 47 kHz. For actuation voltage of 66–75 V, the switching time (ts) was measured to be 28–24 μs, respectively, and the release time (tr) was measured to be ~30 μs. Power handling capability of the single MEMS bridge was obtained experimentally. Measurement shows the bridge can handle incident RF power of ~3.5 W within a safety

122

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 7.4 Capacitance versus voltage measurement shows pull-in and pull-out at 54 V and 34 V, respectively. (b) Vibration spectrum of the MEMS bridge shows a resonance at 47 kHz; the inset shows the fundamental mode shape of the mechanical vibration. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

FIGURE 7.5 Measured Vpi and Vr versus incident RF power at 10 GHz at room temperature. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

limit without any self-actuation (VRFrms > Vpi) or latching (VRFrms > Vr), as shown in Figure 7.5. The MEMS bridge demonstrates a very small change in the voltages (Vpi and Vr) up to ~3.5 W of RF power. After that, the Vpi (Vr) was redsuced by ~60% (~40%) up to ~6 W of RF power at the room temperature. It was mostly due to the effect of dielectric charging under the high RF power level. Nevertheless, one more reason could be the effect of gold softening due to temperature rise with high power, which gradually reduces the spring stiffness (k) of the bridge [3–6]. The measured and simulated S-parameter response of the unit cell phase shifter is shown in Figure 7.6. The cell is well matched with return loss (S11) of 41 dB and insertion loss (S21) of 0.07 dB at zero-bias state over 8–12 GHz, as shown in Figure 7.6a. Later, the unit cell gives a return loss of 31 dB and worst-case insertion loss of ~0.09–1.1 dB with 53–56 V bias voltage, respectively, as depicted in Figure 7.6a. The measured differential phase shift of 5.92° was obtained between zero-bias to the actuated state at 10 GHz frequency, as shown in Figure 7.6b. All measured responses were validated using an electromagnetic

DMTL Phase Shifter Design Using MAM Capacitors and a MEMS Bridge

123

FIGURE 7.6 Measured versus simulated S-parameter response of the unit cell: (a) return and insertion loss and (b) phase versus frequency response. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

TABLE 7.2 Simulated and Measured Circuit Parameters of the Unit Cell Phase Shifter Parameter

Simulated

Measured

Z0 Cb Cs Rc Rbd/Rbu εeff Rp Rb Re α0

70 Ω 8 fF 18.23 µN 0.13 Ω 0.6 Ω/1.1 Ω 2.32 25 kΩ 20 kΩ 0.9 kΩ 11 dB/m

70 Ω 6.8 fF 12.6 µN 0.21 Ω 0.6Ω/1.1 Ω 2.32 25 kΩ 20 kΩ 0.9 kΩ 16 dB/m

Parameter Cbd Zld Zlu Q factor Mechanical f0 Spring constant ts, tr (µs) S11 (actuated) Loss Phase shift

Simulated

Measured

56 fF 64 Ω 49 Ω 58 44.6 kHz 11 Nm–1 14.6, 20.4 36 dB 0.12 dB 6.7° (10 GHz)

41.3 fF 66.7 Ω 52.8 Ω 49 47 kHz 13.4 Nm–1 18, 24 32.6 dB 0.13 dB 5.88° (10 GHz)

solver HFSS v13. MEMS bridge initial deformation and bias material resistivity were also considered in the simulations. Table 7.2 shows the simulated and measured circuit parameters for the unit cell phase shifter. Simulated Cb is very close to the measured value, whereas Cbd is degraded more compared to the simulated value due to dielectric surface roughness. A dielectric surface roughness of 14.8 nm was seen from the optical profilometer. However, the MAM capacitor (Cs) shows a deviation (27%) compared to the simulated value. The measured active capacitance ratio (Cs/Cb = 2.75) is very close to the simulated value (Cs/Cb = 3). The decrease of Cb and Cs capacitance values changes the loaded line impedances Zlu and Zld. The deviation between measured and simulated loss is attributed to the influence of the fringing capacitance due to beam movement relative to the signal line. The unit cell initially designed for 7° phase shift actually gives 5.88° when measured, which is much closure to the desired value of 5.625°. The overall Q factor of the unit cell is also reduced to 49 from the simulated value of 58. It is analytically found that if a resistivity of 500 Ωmeter is achieved for TiW bias lines, then insertion loss could be reduced to 0.9 dB and Q factor could be increased to 77 simultaneously from the unit cell. The resistivity of the TiW bias line mostly depends on the gas flow

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RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 7.7 Microscopic images of individual phase bits (a) 11.25°, (b) 22.5°, (c) 45°, (d) 90° and (e) 180°. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

during processing and the chamber condition during the sputtering process. The MAM capacitor Q factor and static resistance (Rs) can also be found using Equation 7.8 [1]: 2

Zp (1 + M ) - 4 MZp2 - X ( M - 1)2 2 Im(S11 ) (7.8) , Rs = Q= 1- M 1- M



2

where M = S11 , X =

1 and Zp = 50 Ω. wC s

7.3.2 Primary Phase Bits Measurement Results A 5-bit DMTL phase shifter consists of five primary bits: 11.25°, 22.5°, 45°, 90° and 180°. All phase bits were fabricated separately to get the desired phase response. The microscopic images of different primary bit phase shifters are shown in Figure 7.7. When two unit cells are cascaded together, then the internal bias resistance (Rb) increases with additional internal bias resistance between two cells. The overall simulated Q factor of MAM capacitors reduces from 333 to 251. The loss and phase shift performances of individual phase bits are shown in Figure 7.8. Moreover, individual phase bits give maximum phase error of 2° at 10 GHz. Insertion loss of the phase shifter increases with the high impedance CPW line, line loss of the high impedance CPW line, line loss of the high impedance bias lines and Q loss of the MAM capacitors. Loaded line loss is directly related to the unloaded line loss with a multiplicative factor of Z0/Zld. Simulated and measured Q factor loss are listed in Table 7.3, where Rbd = 0 Ω, Rp = 25 kΩ, and Rb = Re = infinity. In all individual bits, one cell gives a measured

DMTL Phase Shifter Design Using MAM Capacitors and a MEMS Bridge

125

FIGURE 7.8 (a) Measured loss of the individual phase bits when structure is tuned to 53 Ω. Measured versus simulated phase shift of the individual phase bits: (b) 11.25°, (c) 22.5°, (d) 45°, (e) 90° and (f) 180°. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

TABLE 7.3 Simulated and Measured Q Factor Loss Component at 10 GHz for Individual Phase Bits in the Actuated State Parameters 2 cells Q factor loss (11.25°) 4 cells Q factor loss (22.5°) 8 cells Q factor loss (45°) 16 cells Q factor loss (90°) 32 cells Q factor loss (180°) Bridge resistance (Rbd) loss

Simulated

Measured

0.13 dB 0.24 dB 0.49 dB 0.79 dB 1.1 dB 0.32 dB

0.21 dB 0.39 dB 0.6 dB 0.9 dB 1.46 dB 0.19 dB

Q factor of 49, which is connected with the external bias, and the rest of the cell gives a Q factor of 177. The individual phase bit performances are tabulated in Table 7.4. 7.3.3 Complete 5-Bit Phase Shifter Measurements and Results Finally, all individual primary phase bits are cascaded together to build the complete 5-bit DMTL phase shifter. The complete schematic of the phase shifter is shown in Figure 7.3, and Figure 7.9 shows the SEM close image of part of the fabricated 5-bit phase shifter. The complete phase shifter demonstrates an average measured return loss of ~12 dB and average insertion loss of 4.72 over the 8–12 GHz from 32 phase states as shown in Figure 7.10. The maximum phase error of ±3.2° (average phase error ~1.84°) was obtained at 10 GHz from the 5-bit phase shifter, as depicted in Figure 7.11. Total 32 air bridges are placed on the high impedance line after each 11.25° cell to improve the matching. The total area of the 5-bit phase shifter is 19.4 mm2. The maximum

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RF Micromachined Switches, Switching Networks, and Phase Shifters

TABLE 7.4 Measured Individual Primary Phase Bits Performance over 8–12 GHz Phase State 11.25° (ref) 11.25° (actuated) 22.5° (ref) 22.5° (actuated) 45° (ref) 45° (actuated) 90° (ref) 90° (actuated) 180° (ref) 180° (actuated)

S11 (dB)

S21 (dB)

∆Φ (deg) at 10 GHz

∆ΦE (deg) at 10 GHz

28.75 26 26.2 24.4 21 18.5 19 16.3 15 13.7

0.088 0.13 0.15 0.38 0.35 0.6 0.54 0.85 0.8 1.48

— 11.4° — 23.6° — 45.89° — 90.73° — 178.77°

— +0.15° — +1.1° — +0.89° — +0.73° — −1.23°

FIGURE 7.9 SEM image of fabricated phase shifter structure where all functional blocks are marked. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

FIGURE 7.10 Measured S-parameter response of the 5-bit phase shifter (a) return loss (b) insertion loss is verified with simulated average loss. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

DMTL Phase Shifter Design Using MAM Capacitors and a MEMS Bridge

127

FIGURE 7.11 Measured phase versus frequency characteristics of the complete cells. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

initial deformation on the MEMS bridge was found to be 0.72 µm from the fabricated 62 bridges on the complete 5-bit phase shifter. The switching voltage was varied from 54 V to 62 V for the 5-bit operation. Maximum MAM capacitance deformation of 0.17 µm (upward) was also captured from the surface profile data. The decrease in MAM capacitors (Cs) and MEMS bridge capacitors (Cb) deviates the Zlu and Zld values, which in turn change the desired phase shift from the 5-bit phase shifter. The maximum variation in Zlu and Zld was found to be ~±2 Ω from the unit cell data (66.7 Ω and 52.8 Ω). The increase of the attenuation constant of the unloaded line increases the loaded line loss. The measured MAM Q factor of 57 cells is 126 (simulated value ~157) and the remaining 5 cells that are connected to the external bias resistors contribute a Q factor of 38 (simulated value ~57). The performance summary of the complete 5-bit DMTL phase shifter is listed in Table 7.5.

7.4 Reliability Measurements and Results 7.4.1 Reliability Measurements of the MEMS Bridge The power handling capability of the MEMS bridge was described in Figure 7.5. The MEMS bridge was tested under a prolonged actuation condition with 55 V bias and with 0.1 W of RF power at 10 GHz in a standard laboratory environment. The corresponding changes in Vpi and Vr voltages were recorded after every 5 min, as shown in Figure 7.12a. Results show a very negligible change in voltages up until ~4 hours of actuation. The degradations on Vpi and Vr were more prominent after 4.5 h up to 6 h. This change in voltage is due to the gold spring softening and dielectric charging over 6 h of continuous actuations. Although SiO2 is hydrophilic in nature and it has lower trap density, the surface charge on the dielectric film can be significantly impacted by residual humidity and it takes a longer time to discharge completely. The stiction due to the unclean environment can also impact the switching performance and degrade the reliability. Three identical MEMS bridges were

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RF Micromachined Switches, Switching Networks, and Phase Shifters

TABLE 7.5 Performance Summary of the 5-Bit Phase Shifter over 8–12 GHz Phase State   0°  11.25°  22.5°  33.75°  45°  56.25°  67.5°  78.75°  90° 101.25° 112.5° 123.75° 135° 146.25° 157.5° 168.75°

S21 (dB)

∆Φ (deg)

∆ΦE (deg)

Phase State

S21 (dB)

∆Φ (deg)

∆ΦE (deg)

3.77 3.85 4.06 4.13 4.2 4.28 4.37 4.44 4.54 4.6 4.49 4.53 4.67 4.71 4.77 4.8

0 10.15° 23.7° 35.1° 46.58° 57.62° 68.8° 80.12° 88.58° 103.04° 110.82° 120.87° 132.8° 144.37° 155.97° 166.9°

0 −1.1° +1.2° +1.35° +1.58° +1.37° +1.3° +1.37° −1.42° +1.79° −1.68° −2.88° −2.2° −1.88° −1.53° −1.85°

180.0° 191.25° 202.50° 213.75° 225.00° 236.25° 247.50° 258.75° 270.00° 281.25° 292.50° 303.75° 315.00° 326.25° 337.50° 348.75°

4.9 4.92 4.95 4.93 4.96 5.02 4.97 5.1 5.3 5.22 5.16 5.21 5.1 5.11 5.24 5.4

177.3° 192.8° 203.22° 215.6° 227.4° 237.4° 245.76° 260.45° 271.73° 278.32° 294.35° 305.49° 316.82° 327.95° 340.38° ~352°

−2.7° +1.55° +1.72° +1.85° +2.4° +1.15° −1.74° +1.7° +1.73° −2.93° +1.85° +1.79° +1.82° +1.7° +2.88° +3.25°

FIGURE 7.12 Measured Vpi and Vr versus (a) extended continuous actuation and (b) temperature at 0.1 W of RF power at 10 GHz. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

actuated with the same operating conditions and results show the same trend. The reliability performance could be easily accelerated further using packaging in a clean environment. Extended reliability tests under high RF power with prolonged actuations were not performed. The Vpi and Vr voltages were measured at –30°C to 70°C temperature variation and the results are presented in Figure 7.12b. A temperature controller (Temptronic Corporation) was connected to the chuck of the probe station. A controller was used in order to set operating temperatures during the measurements. The Vpi changes from 58 V to 47.5 V, and Vr shifts from 33 V to 25.4 V over 100°C temperature duration at 10 GHz. It represents an average variation of 0.125 V/°C (Vpi) and 0.075 V/°C (Vr). It is mostly attributed to the strain caused by the different thermal expansion coefficient (α) between the gold bridge (α = 14 × 10 –6 °C–1) and the alumina substrate (α = 7.2 × 10 –6 °C–1) [6]. Bridge tensile stress turns

DMTL Phase Shifter Design Using MAM Capacitors and a MEMS Bridge

129

into compressive stress at high temperature. As a result, Vpi and Vr voltages change due to the effect of spring softening. The measurement result indicates that the device can be packaged under high temperature conditions. 7.4.2 Reliability Measurements of the 5-Bit Phase Shifter After successful fabrication and bare die measurement of the DMTL 5-bit phase shifter, extensive reliability measurements were carried out. The phase shifter is diced in the form of a chip and mounted onto a test jig for measurement. The module is made with gold-coated brass material. Reliability measurement on the 5-bit phase shifter was carried out under cold-switched conditions and with different temperature variation. Phase shifter reliability performance was continued with the same setup shown in Figure 6.14 (see Chapter 6), and corresponding changes in phase shift and loss performances were recorded after every 10 min with 0.1–1 W of RF power at 10 GHz, as shown in Figure 7.13a. Results show maximum average phase error and loss variation of 8.8° and 11 dB, respectively, with 1 W of RF power and with 55–70 V of bias voltage after 1.5 K cycles of operation. Here, one cycle contains 32 states of operation. The added bond wire parasitic and cable losses are the primary reasons for higher phase error and loss between bare die and the module form. The added parasitic changes the overall Q factor of the phase shifter, which attributes to the higher loss on the phase shifter in a test jig. Nevertheless, the dielectric charging at high RF power is also a reason for these high values. Moreover, 0.5–0.84 µm overall deformation on the MEMS bridge (0.15–0.26 µm for the MAM capacitor) was observed from the optical profilometer. It also deviated the device response from the desired range followed by the changes in Zlu and Zld values. All these issues could be reduced with a hermetic packaging. Phase shifter performance was observed under 100°C temperature variation and corresponding changes in average phase errors (from 32 states) and insertion loss were recorded, as shown in Figure 7.13b. This process was performed with 55–65 V actuation voltage under 0.1 W of RF power at 10 GHz. The phase shifter shows an average loss of 6.6 dB and average phase error of 8° up to 55°C. The performance started degrading after ~55°C and most of the failures occur between 60°C to 70°C, as depicted in Figure 7.13b. During these cycles (60°C and 70°C), the phase shifter works well up to ~0.8 K cycles with average 8.7 dB loss and with an average phase error of 10°, respectively.

FIGURE 7.13 Reliability measurements of the 5-bit phase shifter under (a) 1 W of RF power and (b) 40°C–70°C temperature variation with 0.1 W of RF power. (Used with permission from Institute of Physics (IOP) Publishing Limited.)

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RF Micromachined Switches, Switching Networks, and Phase Shifters

Three identical phase shifters were tested during these cycles and in most cases failure occurs at 56.25°, 78.75°, 101.25°, 191.25°, 168.75° and 236.25° to 348.75° (except 270° and 315°) phase bits. It is most likely due to the reduction of thermal conductivity of the gold beam (317 W/mK) at high temperature during the conduction heat transfer process [6]. The failure due to convection heat transfer and radiation heat transfer are fairly negligible in this case. The probability of the failure can be improved with a higher metal thickness of the gold beam.

7.5 Conclusion In this chapter, design, fabrication and characterization of the 5-bit DMTL phase shifter using a MEMS bridge and MAM capacitors have been demonstrated. Phase shifter uses 62 in-line MEMS bridges and 124 MAM capacitors. The working functionalities of the phase shifter have been discussed and validated. Functional behaviors of MEMS switching elements have been thoroughly observed and characterized. Extensive design approach and measurements of the individual phase bits of the 5-bit phase shifter were discussed. Different issues that influence the MEMS phase shifter performance in a nonhermetic environment were also discussed in detail. The 5-bit phase shifter exhibits average return loss of 12 dB, average insertion loss of 4.72 dB and average phase error of ±3.25° at 10 GHz. It has been observed that the phase shifter is reliable with 1 W of RF power up to 1.5 k cycles. Moreover, phase shifter reliability has been measured under 40°C to 70°C temperature variation and the results are discussed. The reliability of the 5-bit phase shifter could be improved with packaging and using fewer switching beams.

References

1. J. S. Hayden, and G. M. Rebeiz, Low-loss cascadable MEMS distributed X-band phase shifters, IEEE Microw. Guided Wave Lett., vol. 10, pp. 142–144, April 2000. 2. K. Topalli, O. A. Civi, S. Demir, S. Koc, and T. Akin, A monolithic phased array using 3-bit distributed RF MEMS phase shifters, IEEE Trans. Microw. Theory Tech., vol. 56, no. 2, pp. 270–277, February 2008. 3. D. Peroulis, S. P. Pacheco, K. Sarabandi, and L. P. B. Katehi, Electromechanical considerations in developing low-voltage RF MEMS switches, IEEE Trans. Microw. Theory Tech., vol. 51, no. 1, pp. 259–270, January 2003. 4. S. Pamidighantam, R. Puers, K. Baert, and H. A. C. Tilmans, Pull-in voltage analysis of electrostatically actuated beam structures with fixed-fixed and fixed-free end conditions, J. Micromech. Microeng., vol. 12, no. 12, pp. 458–464, June 2002. 5. H. A. C. Tillman, and R. Legtenberg, Electrostatically driven vacuum-encapsulated polysilicon resonators. Part II. Theory and performance, Sens. Actuator A, 45, pp. 67–84, 1994. 6. C. L. Goldsmith, and D. I. Forehand, Temperature variation of actuation voltage in capacitive MEMS switches, IEEE Microw. Guided Wave Lett., vol. 15, no. 10, pp. 718–720, October 2005.

8 Push–Pull Type of Micromachined Phase Shifters

8.1 Introduction The distributed MEMS transmission line (DMTL) phase shifter was implemented and demonstrated from X- to W-band by many researchers using different topologies. Barker and Rebeiz [1] were among the first who proposed and studied the microelectromechanical systems (MEMS) distributed phase shifter. Later, further improvements in phase shifters were presented [2–12]. The symmetric toggle microelectromechanical systems (MEMS) varactor, designed by K. J. Rangra et al. [12], was used to obtain 360° phase shift at 35 GHz with a return loss of 15 dB. However, all of these phase shifters used high actuation voltage. To use the phase shifter in transmit/receive (T/R) module of a phased array, a charge pump can be used. However, charge pumps do not adapt well to the particularity high load of radio frequency (RF) MEMS, for their limited slew rates and require longer settling time. Furthermore, the charge pump has limited choice for Vout versus Vin, and it also generates noise in the T/R module. This chapter concentrates on the design, development and complete characterization of a push–pull type of MEMS actuator. A systematic analytical design methodology of the push–pull actuator is comprehensively worked out and discussed. The behavior of the push–pull actuator is analyzed, supported with the system level simulation approach and validated with different stages of measurement setup. The last phase of this chapter concentrates on the utility of the push–pull bridge. The concept of the push–pull actuator is validated using an analog-type DMTL phase shifter topology, which gives us the ability to achieve a complete differential phase shifter with a very low actuation voltage at 40 GHz frequency.

8.2 Operating Principle of the Push–Pull Actuator The microelectromechanical push–pull actuator works on the principle of electrostatic actuation. The schematic image of the push–pull actuator is shown in Figure 8.1. There are four fixed electrodes on the substrate: two are the push electrodes and the other two are the pull electrodes. When the pull voltage is zero and the push voltage is applied, as shown in Figure 8.2a, the central part is lifted upward. In the other state, when voltage is applied to the pull electrode, as shown in Figure 8.2b, the central part of the beam moves down. Such a membrane is composed of different parts, with the central part acting as the mobile plate 131

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RF Micromachined Switches, Switching Networks, and Phase Shifters

Torsion spring

Anchor Fixed electrode

Transmission line

Contact beam Vpull

Vpush Lever

Mobile electrode

FIGURE 8.1 Schematic of the top surface of the push–pull type MEMS bridge. (Used with permission of Institute of Physics and IOP Publishing Limited.)

of the variable capacitor, two mobile electrodes implementing the toggling performance, and two levers connected with the central parts and mobile electrodes. The bridge can be pushed up or down by applying a voltage on two couples of fixed electrodes, which are symmetric with respect to the central transmission line underneath the contact beam as shown in Figure 8.1. Actuation is carried out with the electrostatic torque (Te) and restoring torque (Tr) as a function of rotation angle (θ). The electrodynamic analysis of the push–pull actuator is carried out using the moment balance equation with electrostatic torque (Te) and restoring torque (Tr) as a function of rotation angle (θ) as shown in Figure 8.2. The simplified two degrees of freedom (2-DOF) model is used to study the actuation mechanism of the push–pull bridge with the influence of van der Waals (vdW) and Casimir torques followed by the electrostatic and restoring torques. The influence of torques on critical tilting angle is comprehensively worked out to obtain the accurate analytic model of actuation voltage. The dynamic study of this electrostatic torsional push–pull bridge is discussed in detail in the next section. 8.2.1 Analysis of Bridge Pull-In Voltage Using Quasistatic Approximation The analytic model of this 2-DOF approximation is carried out under infinite spring bending stiffness with pure rotational movement for one half of this symmetric model. The balance of moments affecting the membrane rotation with respect to the anchor points is only considered in the present case. Membrane deflection is defined in terms of rotation by θ. When the Vpush voltage is applied between electrodes, an electrostatic attracting force between them will produce an electrostatic torque (Te), and under the action of

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Push–Pull Type of Micromachined Phase Shifters

fr fcas

fvdw Vpush

g0 fe

fvdw

Line of symmetry

fcas

(a) Line of symmetry

fr

fcas

fvdw

f g0 fcas

fvdw

fe

Vpull

(b) FIGURE 8.2 Schematic of (a) beam lifted upward due to Vpush and (b) beam snaps down due to Vpull under different applied forces. (Used with permission of Institute of Physics and IOP Publishing Limited.)

torque the mobile plate will tilt. For a given bias voltage (V bias), the distributed electrostatic force (Fe) is

dFe ( x) Vbias = e0 wme (8.1) dx ( g0 - qx)

where wme is the mobile electrode width, g0 is the effective gap between the mobile electrode and fixed electrode, and x  is the distance from the anchor points according to the reference system used in Figure 8.1. The following assumptions are made to simplify the analysis:

1. Residual stress in the mobile plate is ignored. 2. The error derived from nonhomogeneous distribution of charge, after the mobile electrode deflects, is ignored. 3. The fringing field effect is ignored. 4. Only small deflection is assumed.

The electrostatic torque (Tr) is expressed using a small angle approximation ( lme  g0 , sin q ~ q ) in Equation 8.2 as follows [13]:

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RF Micromachined Switches, Switching Networks, and Phase Shifters

Te (q) =

e0Vbias 2wme 2q2

é t æ g0 - lmeq + d ê ç lmeq ed ê + ln ç ê g0 - lmeq + td ç g0 - ( lme - lfe ) q + td ç ê ed ed è ê ê ( lme - lfe ) q êê g0 - ( lme - lfe ) q + td êë ed Te (θ ) =



öù ÷ú ÷ú ÷ú ÷ú ø ú ú ú ú úû

2 ε 0Vbias wme ( m ) (8.2) 2 2θ

where lme is the mobile electrode length, lfe is the length of the fixed electrode, and td and εd are the thickness and dielectric constant of the insulator, respectively, as shown in Figure 8.3. When voltage is applied to the push electrodes, interaction between the tip of the mobile electrode and fixed electrode uses a fairly general Lennard-Jones force, whereas McCarthy et al. [14] consider it as a linear springlike force. This new force has an attractive component, called vdW force (long-range force), and one repulsive component, called Casimir force (short-range force) that were proved to be of fundamental importance in MEMS devices [15]. These forces allow predicting quickly and with good approximation the dynamic response of the system, including the effect of bouncing. Bouncing can introduce severe damage to the mobile electrode and fixed electrode interface by mechanical hammering and electrical arching affecting the durability of the system, and possibly leading to the permanent adhesion or stiction in the MEMS bridge [16]. When the mobile electrode rotates counterclockwise with an angle of θ, the vdW differential forces acting on parallel differential plates with width wme and infinitesimal length dr at both sides of the torsional beam is given by Equation 8.3 as follows [17]:

L dFvdW =

Awme dr Awme dr R = , dFvdW (8.3) 6 π ( g0 + rθ )3 6 π ( g0 + rθ )3

FIGURE 8.3 3D model of the push–pull MEMS bridge. (Used with permission of Institute of Physics and IOP Publishing Limited.)

Push–Pull Type of Micromachined Phase Shifters

135

where A = π2αρ2 is the Hamaker constant (assumed to be 2 × 10−19 J), ρ is the volume density of material and α is the constant character in the interaction between two atoms. The torque of these vdW differential forces to the torsional beam is defined by Equation 8.4 [17]: lme



MvdW =

∫ r (dF

L vdW

)

R − dFvdW =

0

− g0 + 2lmeθ  Awme 1  g0 + 2lmeθ   + 2 2 12π θ  ( g0 + lmeθ ) ( − g0 + lmeθ)2  

MvdW =



Awme ( n ) (8.4) 12πθ 2

Casimir torque is defined by

Mcasimir =



π 2 cwme 1  − g0 + 3lmeθ g0 + 3lmeθ    + 1440 θ 2  ( g0 − lmeθ )3 ( g0 + lmeθ )3    Mcasimir =

π 2 cwme ( p) (8.5) 1440 θ 2

where  is the Planck constant divided by 2π, which is equal to 1.055 × 10−34 Js; and c is the speed of light and equals to 3 × 108 ms−1. When mobile electrodes rotate around the torsional spring, a restoring torque will be produced due to elastic restoring torque of the beam, which can be simplified as a flexure spring with torsional stiffness K. Under the application of electric bias, the corresponding reaction moment M A acts at the anchor, and reaction forces FA and FB are applied at the anchor and lever, respectively. Restoring moments of the torsional bridge is given by

Mr = M A + MB = M A + FB lfe (8.6)

where lfe is the fixed electrode length. The reaction moment at the anchor is given by [14]

M A = 2Kt q (8.7)

where Kt is defined by

Kt = C1

E wts tts 3 (8.8) 2 (1 − ν) lts

where wts, tts and lts are width, thickness and length of torsional spring, respectively. E and υ are the effective Young’s modulus and Poisson ratio of membrane material. The constant C1 depends on the wts/tts ratio and it is called the torsion coefficient. For a rectangularshaped beam, wts/tts > 10, and C1 is equal to 0.33. For wts/tts ~ 5, C1 is equal to 0.25 [18]. The reaction force due to a lever can be demonstrated by Equation 8.9 [19]:

FB =

2 EIlθ (8.9) ll2

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RF Micromachined Switches, Switching Networks, and Phase Shifters

where Il = wltl3/12, and wl, tl and ll are width, thickness and length of the lever beam, respectively. Rearranging Equations 8.6, 8.7, 8.8 and 8.9, spring stiffness of the beam is determined and it is as follows [18]: K=



2 EIl lfe + 2kt (8.10) ll2

In a push–pull type of torsional MEMS actuator, the vdW and Casimir torques depend on the separation between two surfaces and the rotation angle. Differences of two dimensionless torques exist when the separation gap is in nanoscale, and the effects of these torques becomes larger with decrement of the gap in presence of actuation bias. For a small dimensionless gap, the difference is not so sensitive to the rotation angle. It is negligible unless the rotation angle is large enough. So, the influence of all kinds of torques on pull-in effect on push–pull torsional MEMS bridges is considered. From this model, the following equation of motion is derived by the classical second-order linear differential equation using a small angle approximation,

 + bθ + kθ = Me + MvdW + Mcasimir (8.11) Iθ

where I is the mass moment of inertia about the center of rotation and b is the damping coefficient of the mobile plate. Now using the quasistatic assumption (  q = q = 0), the following moment balance equation is formed:

Me + MvdW + Mcasimir − kθ = 0 (8.12)

By comparing Equations 8.2, 8.4, 8.5 and 8.9 with 8.12, actuation voltage of the push–pull bridge is defined by

Vbias =

2θ 3  K    −S ε 0 wme  m 

where

S=

A πε 0

2  n  π c   + m 720ε 0

 p  m  (8.13)

Expressions for m, n and p are given in Equations 8.2, 8.4 and 8.5, respectively. The values of constant critical rotation angle and pull-in voltage are dependent on the size of the structure. The constant value of the critical rotation angle of this kind of actuator is defined in [19] and its value is 0.4404. The critical angle is not constant but tends to be constant with an increase in the gap. An effect due to vdW and Casimir torques on critical angle is negligible. The point of instability in terms of the critical angle is numerically found from the maxima of the Equation 8.13, and it is

θ* ≈ 0.4404 θ0 (8.14)

where θ0 is the maximum rotation angle (θ0 ~ g0/lme).

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Push–Pull Type of Micromachined Phase Shifters

Finally, pull-in voltage (Vpi) of the push–pull bridge is obtained using the condition of the maximum rotation angle and the same is expressed by

0.1708 g03  K  −S 3   m  ε 0 wme lme

Vpi =

where 0.0279 A 27.978c (8.15) + g0 ε 0 m ε 0 g02 m

S=



When bias voltage is small, there exists a solution. However, above a threshold voltage, Equation 8.15 cannot be solved. It means that the mobile electrode is abruptly pulled down to touch the fixed electrode at threshold voltage. Maximum deflection (∆g) from the zero-bias position where the mobile plate can move up and down without snapping is defined by l g  ∆ g =  lme + l  0 (8.16)  3  lfe



This Δg is also defined as a maximum travel range of this kind of push–pull actuator. 8.2.2 Analysis of the Push–Pull Actuator under Step and Modulated Voltage Responses Electromechanical modeling of a push–pull bridge and its lumped model representation is presented in the last section. In this study, pull-in voltage (Vpi) is quasistatic in nature and it gives a lower bound on the pull-in voltage. Under the step voltage, the push–pull actuator tends to overshoot the equilibrium position due to an underdamped condition. Pull-in could happen at a lower voltage if the overshoot is large. Overshoot attains its maximum value (θmax ~ 0.65 θ0) at zero damping condition. This overshoot corresponds to a step voltage and that can be found from the time–energy balance equation, as given in Equation 8.17: θmax



Einjected =

∫ (T (θ) + T e

vdw

(θ) + Tcasimir (θ))dθ (8.17)

0

where Te, Tvdw and Tcasimir are the electrostatic torque, van der Waals torque and Casimir torque, respectively, and they are defined by Equations 8.2 to 8.6. All torques are a function of the rotation angle (θ) of the push–pull bridge. The potential energy stored in the device is defined by

Epotential =

1 2 Kθ max (8.18) 2

The energy injected is always equal to the energy stored at a zero damping condition. Solving Equations 8.17 and 8.18, the corresponding pull-in voltage under step voltage is

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RF Micromachined Switches, Switching Networks, and Phase Shifters

Vspi =

 K  − − U 1  wme    0.355 g0 ε 0 log  0.645 g0 lfe  0.355 g0 +  lme

    

(8.19)

where U1 =



3 πc lme A lme 1 . 47 − g + ( ) 90 g03 1.75π g04

At zero damping condition and with the same device geometry, the ratio between step pull-in voltage (Vspi) to the quasistatic pull-in voltage (Vpi) is less than unity. It indicates that low pull-in can be achieved at voltage lower than the level predicted by quasistatic analysis. Similarly, the input energy stored in the mechanical system can be considered under modulated voltage where energy is lost in a form of damping during each cycle over a number of mechanical oscillations. The same energy balance equation (energy injected is equal to energy dissipated) is used to compute the pull-in voltage under the excitation of a modulated voltage. The injected energy (Einjected) over one oscillation at the limit cycle is defined by θmax



Einjected =

∫

(Te (θ) + Tvdw (θ) + Tcasimir (θ))dθ (8.20)

− θmax

Tvdw and Tcasimir terms become zero under the limits of integration in Equation 8.20, which indicates that under modulated voltage the Lennard-Jones force (FLJ) plays no role in Einjected. It improves the device reliability where linear springlike forces have negligible contribution in bridge dynamics under modulated voltage. The pull-in point at maximum rotation angle (F *max ~ 0.73F 0 ) under modulated voltage is found to be Vmpi ≈ 3.34



Kg03 ; (8.21) 3 ε 0 wmeQlme log( x)

where

x=

0.467 g02 2

l g2 l  0.467 g + 1.06 fe 0 + 0.467  fe  g02  lme  lme



2 0

If all the device design parameters are taken to be a constant (C), then the ratio of the modulated pull-in voltage to the quasistatic pull-in voltage can be written as

Vmpi 1 ≈C (8.22) Vpi Q

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Push–Pull Type of Micromachined Phase Shifters

FIGURE 8.4 Calculated rotation angle versus under quasistatic, step and modulated applied voltages. (Used with permission of Institute of Physics and IOP Publishing Limited.)

The analysis of the modulated voltage pull-in indicates that for given device design parameters, Vpi is equal to the square root of Q times Vmpi. Thus, modulated voltage leads to significant reduction in the pull-in of the push–pull bridge. A 2–3 V improvement in pull-in voltage is observed under Vmpi compared to Vspi with the same set of device parameters. Beam nonuniform deformation under residual stress or stress gradients and skin effects are not considered in the analysis. Figure 8.4 shows the calculated rotation angle of the push–pull actuator under quasistatic (Vpi), step (Vspi) and modulated (Vmpi) pull-in voltages. Results show pull-in is achieved at 15.4 V, 14.2 V and 13.6 V with Vpi, Vspi and Vmpi voltages. Moreover, it is also interesting to note that pull-in achieved at θ = ~0.73° in all voltage conditions with the same physical dimensions of the push–pull actuator. The initial design parameters and material properties of the MEMS push–pull bridge are tabulated in Tables 8.1 and 8.2, respectively (the asterisk in Table 8.2 shows fabrication process limited).

8.3 Modeling of the Push–Pull Bridge The lumped model representation of the push–pull bridge is shown in Figure 8.5. A prototype process is developed in the Process Editor of CoventorWare along with material TABLE 8.1 Designed Dimensions of the Push–Pull Bridge Parameter

Value

Fixed electrode length (lfe) Fixed electrode width (wfe) Mobile electrode length (lme) Mobile electrode width (lme) Fixed plate thickness (tm)

90 µm 230 µm 200 µm 230 µm 1 µm

Parameter Lever width (wl) Lever length (ll) Torsional spring length (lt) Torsional spring width (wts) Mobile plate thickness (tm)

Value 20 µm 85 µm 190 µm 15 µm 1.25 µm

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TABLE 8.2 Material Properties of the MEMS Push–Pull Bridge Parameter Plate material Conductivity of plate material Density of plate material (1D70C) Young’s modulus (E)

Value Gold 4.1 e7 S/m* 19300 kgm−3 45 GPa*

Parameter Poisson’s ratio (υ) Sheet resistance of fixed plate Sheet resistance of mobile plate Dielectric thickness (td)

Value 0.4 0.025 Ω/sq* 0.02 Ω/sq* 0.5 µm

FIGURE 8.5 Lumped representation of the push–pull MEMS bridge in CoventorWare Saber Architect. (Used with permission of Institute of Physics and IOP Publishing Limited.)

properties, as given in Table 8.2. The push–pull bridge was designed using lumped Beam, Beam electrode, rigid plate and rigid plate electrode modules [19] in the Saber schematic CoventorWare. The initial gap height (g0) of the push–pull bridge is 2.5 µm. Pull-in analysis was performed on the torsional beam actuator in two states. The pull-in point under Vpull is obtained and shown in Figure 8.6a. A point demonstrates two points: Vpull and gap. Point P1 (4.91 V, 0.75 µm) shows the Vpull value required to actuate the beam up to 0.75 µm gap from the top and point P2 (5 V, 0.83 µm) is the pull-in point after that beam will snap down and make contact with a bottom transmission line. It almost validates the theoretical estimation of pull-in that occurs at one-third of g0, which is 0.83 µm. When voltage is applied to the push electrodes, the mobile plate is lifted upward and it increases the effective gap height from the center portion of the plate, as shown in Figure 8.6b. The gap height is linearly increasing from the zero-bias state under the application of Vpush, as shown by point P3 (7 V, 0.96 µm). Point P4 (7.98 V, 3.81 µm) shows the optimum limit of the beam where the mobile electrode touches the bottom fixed electrode. Effective gap height or maximum displacement at this point is 6.34 µm (g0 + 3.81 µm), which validates the theoretical estimation up to a reasonable extent if designed parameters from Table 8.1 are put in Equation 8.16. Small signal ac analysis shows fundamental frequency at 4.86 kHz, as given in Figure 8.7.

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Push–Pull Type of Micromachined Phase Shifters

FIGURE 8.6 Variation of gap height with applied (a) Vpull and (b) Vpush voltages. Inset shows beam upward and downward deflections. (Used with permission of Institute of Physics and IOP Publishing Limited.)

FIGURE 8.7 Vibration spectrum of the beam shows the fundamental frequency at 4.86 kHz. (Used with permission of Institute of Physics and IOP Publishing Limited.)

8.4 DMTL Unit Cell Phase Shifter Design and Modeling The DMTL unit cell works with three operating stages: push, pull and normal. In the pull stage, the bridge is pulling down closer to the center conductor, which in turn increases the loading capacitance (Cpull) on the distributed line, in addition to varying the propagation characteristics and decreasing the phase velocity of the DMTL. It leads to a decrease in loaded impedance (Zpull). In the push stage, the bridge moves in an upward direction and it decreases the loading capacitance (Cpush) on the line. It leads to an increase in loaded impedance (Zpush). The variation of loaded impedances is a function of actuation voltages, which in turn is a function of the tuning ratio (Cpull/Cpush). The resulting change in the phase velocity between two stages of the DMTL produces the differential phase shifts over the operating band. The impedance for each stage of the unit cell is given by

Zup =

sLt , Zpush = sCt + Cup

sLt , Zpull = sCt + Cpush

sLt (8.23) sCt + Cpull

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RF Micromachined Switches, Switching Networks, and Phase Shifters

where Lt and Ct are the per unit length inductance and capacitance, and these are defined by Ct =



ε eff , Lt = Ct Z02 (8.24) cZ0

where εeff is the effective dielectric constant of the transmission line, Z0 is the unloaded line impedance, c is the free space velocity and s is the unit cell length. Cpush, Cup and Cpull can be obtained by parallel plate approximation, as given by

Cpush =

ε 0 wcbW1 ε w W ε w W + C f , Cup = 0 cb 1 + C f , Cpull = 0 cb 1 + C f (8.25) td td t g1 + g0 + g2 + d εr εr εr

The DMTL phase shifter has a cut-off frequency called Bragg frequency ( f B). It is defined by the point where almost total reflection occurs and impedance becomes zero. The Bragg frequency for the unit cell is defined by

−1

fB =  π sLt ( sCt + Cb )  (8.26)  

where Cb is the bridge capacitance. In this work, the fringing capacitance (Cf ) was chosen at 0.34 times the parallel plate capacitance. The spacing (s) and bridge capacitance (Cb) of the shunt beam are defined by rearranging Equation 8.23 and is given by Equations 8.27 and 8.28, respectively, as follows:

(

s = c πfB ε eff



)

−1

(8.27)

 L  L   L   Cpush = s  2 t − Ct  , Cup = s  2t − Ct  , Cpull = s  2 t − Ct  (8.28)  Zpush   Zup   Zpull 

It is seen from Equation 8.27 that s is inversely proportional to f B and the εeff of the substrate. Using Equations 8.23 to 8.28, the phase constants in each state (βpull, βpush) and the net phase shift (Δϕ) are derived in Equation 8.29, as follows:

β push =

Cpush   360   sω  LtCt  1 +   2π Ct    



β pull =

Cpull   360   sω  LtCt  1 +   2π Ct    



∆φ = β push − β pull Cpush   Cpull  360    sω  LtCt  1 +   − LtCt  1 + C  (8.29)  2π C t t  

Push–Pull Type of Micromachined Phase Shifters

143

In the push stage, the center portion of the beam is lifted upward so the phase constant (βpush) decreases due to smaller loading capacitance (Cpush), leading to high loaded impedance (Zpush). The center beam gets downward deflection in the pull stage, leading to higher phase constant (βpull) due to higher loading capacitance (Cpull) with lower loaded impedance (Zpull). As a result, difference in phase constants (∆ϕ) between the two stages is increased. Total travel range (TTR) of the push–pull phase shifter is given by

TTR (mm) = g1 + g 2 = 3.81 + 0.75 = 4.56 (8.30)

8.5 Fabrication The push–pull actuator and a complete phase shifter is fabricated using a simple surface micromachining process on 635 µm alumina substrate. A 70 nm chromium (Cr) is sputtered using the lift-off technique to form the bias lines. Later, 0.5 µm of silicon dioxide (SiO2) is patterned to cover the last Cr layer. Next, 1 μm thick gold is electroplated to form fixed and bias electrodes and coplanar waveguide (CPW) transmission lines. Again, 0.7 µm of SiO2 is deposited as a dc isolation layer on the bottom electrode. Next, spin-coated polyimide (PI) is coated to a thickness of 2.5 µm to get the sacrificial layer. Then, anchor holes and dimple openings (with 1 µm) are created on the sacrificial layer. A 1.25 µm gold layer is electroplated to form the structural layer. Finally, the PI layer is released using an oxygen plasma dry etch process. The SEM (scanning electron microscope) image of a unit cell phase shifter is shown in Figure 8.8. Nonuniform distribution of gap height is captured at different stages of the bridge under the SEM. It is observed that the mobile plate deforms downward over the pull stage and is lifted upward over the push stage. It is most likely due to the asymmetric nature of the push–pull bridge configuration, where the pull stage is always fixed between two levers with the other side of the pull stage, so that it can release the stress. The push stage is buckled up as it cannot release the stress. The process-induced residual stress (σ0) was 110 MPa. Furthermore, this residual stress is tensile (σ0 > 0) in nature, while a compressive stress (σ0 4.5 V) it completely snaps down, as shown

FIGURE 8.12 Capacitance versus voltage variations at (a) pull and (b) push stages. (Used with permission of Institute of Physics and IOP Publishing Limited.)

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RF Micromachined Switches, Switching Networks, and Phase Shifters

in step 2. Nonuniform distribution of gap height over the mobile plate along with the effect of stress-stiffing leads to two steps in pull-in as defined by step 2 and step 3, respectively. The nonuniformity is a function of the dielectric-membrane gap throughout the push–pull bridge. One explanation of this voltage shift (step 2 to step 3) could be the effect of dielectric charging during step 3, as the bottom electrode was insulated with dielectric. Due to the high electric field, laterally inhomogeneous distribution of the charge can be injected in the dielectric. The trap charges change the electric field and lead to a shift in the C–V curve to step 3 with a built-in voltage. This built-in voltage is proportional to the amount of charge, and to the distance between the trapped charge and the fixed bottom pull electrodes. The primary effect of such a trapped charge is to shift the whole C–V characteristic toward positive of the voltage axis due to the negative charge induction into the dielectric layer. In addition to this, another possible explanation of shifting the C–V curve from step 2 to step 3 is the reduction of the gap due to mechanical degradation of the springs when the bridge is stressed. So, the narrowing effect of the C–V curve at step 3 can be avoided with controlled actuation voltage or stiffer mobile plate design. Finally, in this work, the pull-in voltage of the bridge was considered to be at 4.3 V at the pull stage. Measured pull-in voltage of the push stage is shown in Figure 8.12b. A nonuniform profile of the beam introduces a very slow change in capacitance variation with a voltage variation up to 8.1 V, as shown by step 1 in Figure 8.12b. Once it reaches 8.2 V (point A), the beam encounters the point of instability with a sudden decrement of capacitance (step 2). The bridge moves up completely (point B) with 15 V of actuation bias followed by constant capacitance variation up to 17 V bias. The point of instability (pull-in) of the push–pull structure is considered to be 8.1 V at the push stage over bridge actuation and to avoid the beam encounter with the dielectric charging effect and other related effects. 8.6.3 S-Parameter Measurements of the Unit Cell Phase Shifter A CPW line is used as the base transmission line with 100 µm wide center conductor (W) and 140 µm gaps (G) on the alumina substrate. This line is loaded with one MEMS bridge with a line length of 780 µm to make a unit cell, which is the fundamental building block of a complete distributed cell. An extra 200 µm length of a 50 Ω (W = 100 µm, G = 45 µm) line is kept in either side of the unit cell to start with the RF measurement. In order to maintain acceptable matching over a wideband, it is always recommended not to overload the transmission line with an excessively large MEMS capacitance. In this circuit, a sawshaped center conductor is used at the place where the MEMS bridge is built to reduce the overlapping area and it leads to the reduction of the down-state impedance. It introduces an inductance of 4.02 nH on the line on either side of the bridge. Finally, the unit cell demonstrates measured return loss of better than 12 dB and worst-case insertion loss of 0.28 dB from 1 to 40 GHz, as shown in Figure 8.13. All measured responses are validated using the full-wave simulator HFSS v13. The discrepancy between measured and simulation results in S-parameters is attributed to the overall height (g0) nonuniformities in the MEMS bridges that lead to the asymmetric distribution of loaded line capacitances. The measured phase shift of 32.12° is obtained from the unit cell. The measured figure of merit (FOM) of the unit cell phase shifter is ~114.64°/dB at 40 GHz. 8.6.4 Measurements of the Complete Push–Pull Analog Phase Shifter The complete analog phase shifter is designed using 11 MEMS push–pull bridges. The spacing between each unit cell is 780 µm. The schematic image of the complete phase

Push–Pull Type of Micromachined Phase Shifters

149

FIGURE 8.13 Measured versus simulated S-parameter response of a unit cell: (a) return loss and (b) insertion loss performance. (Used with permission of Institute of Physics and IOP Publishing Limited.)

FIGURE 8.14 Layout of phase shifter. Inset shows the saw-shaped CPW. Complete area of the phase shifter is 8.5 mm2. (Used with permission of Institute of Physics and IOP Publishing Limited.)

shifter is shown in Figure 8.14. A close SEM image of the distributed cell is shown in Figure 8.15. A measured return loss of better than 11.5 dB is achieved up to 40 GHz from the distributed cell, as shown in Figure 8.16a. Maximum insertion loss of 3.75 dB is noticed up to 20 GHz. Later, it goes down to 5.7 dB at 40 GHz in pull states, as shown in Figure 8.16b. The agreement between measured and simulated insertion loss is found to be within the 15% tolerance limit due to the asymmetric distribution of gap profile throughout the phase shifter. Typical height nonuniformities variations are 0.43–0.7 μm over the 11 MEMS bridges. Furthermore, the increase in insertion loss in the complete cell compared to the unit cell loss (0.28 dB) is likely due to the signal leakage via the Cr bias lines. The 36% reduction of bias resistance is found from 34.3 kΩ to 21.7 kΩ from the distributed cell to the unit cell. The nature of normal and push state transmission losses closely followed each other, whereas push state loss degrades at high frequency (>25 GHz), which is closely validated with the nature of the unit cell performance. Measured variations of phase shift with applied bias for pull and push states with reference to the normal state are recorded from the measurement and these are shown in Figure 8.17. The maximum differential phase shift of 317.15°/cm is obtained at 40 GHz with 4.1 Vpull voltage, as shown in Figure 8.17a. A sudden change in phase shift from 56° to 152° is observed with a voltage variation from 1.9 to 2.1 V. It is mostly due to asymmetric gap distribution over the pull electrode leading to overall change in the loaded line capacitance, which also validates the C–V profile (Figure 8.12a) where a noticeable change in

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RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 8.15 (a) Top and (b) cross section SEM images of the distributed MEMS phase shifter. (Used with permission of Institute of Physics and IOP Publishing Limited.)

FIGURE 8.16 Measured versus simulated S-parameter responses of the distributed cell: (a) return and (b) insertion loss. (Used with permission of Institute of Physics and IOP Publishing Limited.)

FIGURE 8.17 Measured variation of phase shift with applied bias at (a) pull and (b) push states, respectively, at 40 GHz frequency. (Used with permission of Institute of Physics and IOP Publishing Limited.)

Push–Pull Type of Micromachined Phase Shifters

151

capacitance is captured with the same voltage transition. In the push state, the maximum 44.1°/cm phase shift is obtained with reference to the normal state at 40 GHz with 8.1 V bias, as shown in Figure 8.17b. A continuous phase shift of ~0–360° is obtained from the fabricated device from the push to pull state. Measured phase shift per dB or FOM performance of 63.25°/dB is obtained from the distributed cell at 40 GHz.

8.7 Discussion of Push–Pull Bridge Performances The effects of bridge nonlinearity on mechanical, electrical and RF performances have been discussed thoroughly in this chapter. The results suggest that the structure can be used as a micromechanical varactor for phase shifter applications. However, there are a few more aspects where extra care needs to be taken for obtaining better performance. The first is definitely the asymmetric bridge actuation. The nonuniformity of different layers is an indeterminate parameter that determines the functionality of the device once they are pulled in. The stiffer torsional springs can reduce the probabilities of stiction-induced failures with dielectric charging. The thickness or the length of the mobile plate can be modified to increase the stiffness. The width can also be a parameter to adjust for higher stiffness but at the cost of increased damping. Therefore the design has to be an optimal one with respect to switching voltage, damping and chances of stiction. Another issue that may lead to degradation of the performance of the device is the effect of intrinsic residual stress on the mobile plate. Under this stress, the structure experiences undesirable deformation at the free end over the push electrode after removal of the sacrificial layer. Residual gradient stress causes undesirable deformation that tends to make the suspended beam extremely warped. This results in structure actuating at higher voltage and higher upstate capacitances than anticipated. A tensile–compressive stress converter structure can be used for this purpose. An experimental low temperature release step must be performed at the end in order to guarantee minimal stress gradient and good planarity on the mobile plate. The effect of dielectric charging also affects the reliability on the device performance due to low stiffness and inhomogeneous distribution of trapped charges. A thinner (0.1–0.2 μm) dielectric layer or high dielectric constant materials can be used to overcome the effect of charging. Furthermore, mechanical stoppers can be introduced with thickness higher than the underpass layer to overcome the effect of charging. However, the bridge is not performing any switching action, so the region of instability due to charging can be avoided with controlled voltage waveform. To ensure optimum performance of the phase shifter, a shortterm reliability measurement is performed where a long pulse waveform is applied to the pull electrode, sweeping from 0 to 4 V. Immediately after, 0 to 8 V is imposed on the push electrode, keeping the pull electrodes at 0 V. This process is then repeated for a longer time (~4 h) and almost constant bridge actuation behavior is noticed over the period.

8.8 Conclusion In this work, design, development and characterization of a push–pull type MEMS bridge were discussed and presented with experimental justifications. Extensive analytical studies for design optimization of the push–pull MEMS bridge is carried out and it has been

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RF Micromachined Switches, Switching Networks, and Phase Shifters

consolidated using simulation results. Later, 11 push–pull bridges are cascaded to build an analog DMTL phase shifter. Finally, the complete phase shifter demonstrates maximum return loss of better than 11 dB and worst-case insertion loss of 5.7 dB from DC–40 GHz. The phase shifter also gives complete phase shift (~360°) at 40 GHz with maximum actuation voltage of 8 V. The performance of the reported phase shifter can be improved significantly with hermetic sealing.

References 1. N. S. Barker, and G. M. Rebeiz, Optimization of distributed MEMS phase shifters, in IEEE MTT-S International Digest, 1999, pp. 299–302. 2. Y. Borgioli, Y. Liu, A. S. Nagra, and R. A. York, Low-loss distributed MEMS phase shifter, IEEE Microw. Guided Wave Lett., vol. 10, pp. 7–9, January 2000. 3. J.-H. Park, H.-T. Kim, W. Choi, Y. Kwon, and Y.-K. Kim, V-band reflection-type phase shifters using micromachined CPW coupler and RF switches, J. Microelectromech., vol. 11, no. 6, pp. 808–814, December 2002. 4. H. Zhang, A. Laws, K. C. Gupta, Y. C. Lee, and V. M. Bright, MEMS variable-capacitor phase shifters. Part I: Loaded-line phase shifter, Int. J. RF Microw. Comput.-Aided Eng., vol. 13, no. 4, pp. 321–337, May 2003. 5. J. S. Hayden, and G. M. Rebeiz, Low-loss cascadable MEMS distributed X-band phase shifters, IEEE Microw. Guided Wave Lett., vol. 10, pp. 142–144, April 2000. 6. K. Topalli, O. A. Civi, S. Demir, S. Koc, and T. Akin, A monolithic phased array using 3-bit distributed RF MEMS phase shifters, IEEE Trans. Microw. Theory Tech., vol. 56, no. 2, pp. 270–277, February 2008. 7. B. Lakshminarayanan, and T. Weller, Distributed MEMS phase shifters on silicon using tapered impedance unit cells, in Proceedings IEEE MTT-S International Microwave Symposium Digest, Seattle, WA, June 2002, pp. 1237–1240. 8. G. McFeetors, and M. Okoniewski, Distributed MEMS analog phase shifter with enhanced tuning, IEEE Microw. Wirel. Compon. Lett., vol. 16, no.1, pp. 34–36, January 2006. 9. Q. Wu, K. Tang, Z.-R. Feng, F.-L. Sun, and L.-W. Li, A DMTL phase shifter using insulation layer and saw-shaped CPW, in IEEE Proceedings of Asia Pacific Microwave Conference, Bangkok, 2007, pp. 1–4. 10. K. V. Caekenberghe, and T. Vaha-Heikkila, An Analog RF MEMS slotline true-time-delay phase shifter, IEEE Trans. Microw. Theory Tech., vol. 56, no. 9, pp. 2151–2159, September 2008. 11. F. Solazzi, P. Farinelli, and R. Sorrentino, Design of distributed MEMS transmission line based on toggle MEMS varactors, International Doctorate School in Information and Communication Technologies, University of Trento, Italy, 2010. 12. K. J. Rangra, B. Margesin, L. Lorenzelli, F. Giacomozzi, C. Collini, M. Zen, G. Soncini, L. D. Tin, and R. Gaddi, Symmetric toggle switch—A new type of RF MEMS switch for telecommunication applications: Design and fabrication, Sens. Actuators A, vol. 123–124, pp. 505–514, September 2005. 13. D. Hah, E. Yoon, and S. Hong, A low-voltage actuated micromachined microwave switch using torsion springs and leverage, IEEE Trans. Microw. Theory Tech., vol. 48, pp. 157–160, December 2000. 14. B. McCarthy, G. Adams, N. McGruer, and D. Potter, A dynamic model, including contact bounce, of an electrostatically actuated microswitch, J. Microelectromech. Syst., vol. 11, pp. 276– 283, June 2002. 15. F. W. Delrio, M. P. De Boer, J. A. Knapp, E. D. Reedy Jr., P. J. Clews, and M. L. Dunn, The role of van der Waals forces in adhesion micromachined surfaces, Nat. Mater., vol. 4. pp. 629–634, July 2005.

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16. J. M. Huang, K. M. Liew, C. H. Wong, S. Rajendran, M. J. Tan, and A. Q. Liu, Mechanical design and optimization of capacitance micromachined switches, Sens. Actuators A, vol. 93, pp. 273–285, May 2001. 17. J.-G. Guo, and Y.-P. Zhao, Influence of van der Waals and Casimir forces on electrostatic torsional actuators, J. Microelectromech. Syst., vol. 13, pp. 1027–1035, December 2004. 18. F. Solazzi, Novel design solutions for high reliability RF-MEMS switch, PhD dissertation, International Doctorate School in Information and Communication Technologies, University of Trento, Italy, 2010. 19. K. J. Rangra, Electrostatic low actuation voltage RF-MEMS switches for telecommunications, PhD dissertation, Department of Information and Communication Technology, University of Trento, Italy, 2005. 20. CoventorWare ARCHITECT version 2008, MEMS and Microsystems System-Level Design, Coventor, Inc., March 2008, www.coventor.com.

9 Reconfigurable Micromachined Phase Shifters Using Push–Pull Actuators

9.1 Introduction Many research groups worldwide have demonstrated frequency reconfiguration in phase shifters with liquid crystal [1], photonic [2] and ferroelectric technologies [3,4]. Furthermore, radio frequency microelectromechanical systems (RF MEMS) phase shifters have been reported that can operate over a wide frequency band [5–14]. A reflection-type phase shifter that was reported by Lee in 2004 [15] with an average loss of 3.5 dB and average phase error of 4.9° operates over 15–45 GHz. A CMOS phase shifter used by Kwang-Jin in 2007 [16] offers 3.8 dB average insertion loss and 9.7° average phase error in 15–26 GHz band. A linear phase–type distributed MEMS transmission line (DMTL) phase shifter is one of the best choices reported by researchers for high frequency application (>30 GHz) with excellent phase setting resolution as compared to the switched-line true-time-delay (TTD) networks. A DMTL phase shifter was demonstrated by Pillans in 2012 [17], with 1.7 dB average insertion loss and 7° of average phase error in the 15–35 GHz band. All these phase shifters [1–17] reported so far operate at a single fixed frequency over the band of interest. A DMTL-type frequency reconfigurable phase shifter using triple-stub topology was reported by Unlu in 2013 [18], which can operate at any given frequency within a targeted band of 15–40 GHz with adjustable phase steps. This phase shifter can work in wideband (8–26 GHz) but with different resolution. However, there are no reconfigurable digital phase shifters reported so far that can operate with constant resolution at each band over a wide frequency band of spectrum. This chapter presents a novel configuration of band-tunable reconfigurable phase shifters using DMTL topology. Detailed analysis behind the development of a 5-bit TTD phase shifter using push–pull actuation with minimum insertion loss and good phase accuracy at different bands from 10 to 25 GHz is reported. The theory behind the actuation mechanism of the push–pull bridge is discussed in Chapter 8, where the operating principle with analytical solution and experimental results for a 0°–360° analog phase shifter was presented as proof of concept. In this chapter, the emphasis is given to a more real-life example with a 5-bit phase shifter that is capable of providing 32 phase states with constant resolution (11.25°) at each band between 10–25 GHz. This chapter provides more detailed design, analysis and measurement results and emphasizes the reliability performances with a large number of cycles over a 10–25 GHz band.

155

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9.2 Design and Analysis of a Phase Shifter Using a Push–Pull Actuator The top view and actuation states of a push–pull bridge are shown in Figure 9.1. Two stages of actuations contribute capacitance variation along the transmission line that gives differential phase shift through impedance variation using a torsional electrostatic actuator. Figure 9.1a and b show the top view and equivalent circuit model, respectively, of the unit cell phase shifter utilizing a push–pull actuator and connecting coplanar waveguide (CPW) transmission lines. The bridge is fabricated using 2 µm thick electroplated gold with an air gap of 2.5 µm using 635 µm thick alumina substrate. Each structural parameter of the push–pull bridge is optimized and a travel range of the push–pull bridge is developed that gives the ability to achieve constant resolution (11.25°) over different operating bands. Matching, loss and phase accuracy are thoroughly observed with different gap height variations between the push and pull stage at each phase state. The working functionalities of the push–pull actuator for phase shifting applications were discussed in the previous chapter. The proposed reconfigurable phase shifter is designed for a given frequency over the 10–25 GHz frequency band. However, the electrical length of the complete phase shifter is chosen to be at 17 GHz. The electrical lengths at any given frequency more than 17 GHz are automatically larger. The operating frequency below 17 GHz (10–16 GHz) is controlled with the appropriate tuning from the push–pull actuator. The price paid at low (10–15 GHz) and very high (22–25 GHz) frequency bands within the band (10–25 GHz) is an increase in the phase error and a decrease in the matching. So, the proposed band is chosen with moderate matching (>12 dB) and acceptable phase error (28 dB over the band of interest as shown in Figure 10.10a. Measured average return loss of >17 dB and insertion loss of 10 dB, average insertion loss of 14 >14 >13 >12 >12 >11 >10 >10

2.32 2.38 2.74 2.97 3.42 3.8 ~5 6

2.8° 1.76° 2.86° ~2° 2.93° 3.28° 3.8° 3.7°

63–66 60–63 64–66 61–64 59–62 64–66 66 58–61

PS-1 PS-1 PS-2 PS-2 PS-3 PS-3 PS-4 PS-4

phase shifter demonstrates measured average insertion of 6 dB, average return loss of 10 dB and maximum average phase error of 3.8° up to 60 GHz. The complete area of the proposed wideband phase shifter is comparable to other reported MEMS phase shifters. The performance of the reported phase shifter can be significantly improved with a hermetic condition to overcome the effect from the surface charges trapped due to some residual humidity and stiction due to the unclean environment.

References 1. S. Lucyszyn, Advanced RF MEMS, Cambridge University Press, August 2010. 2. S. Muller, P. Scheele, C. Weil, M. Wittek, C. Hock, and R. Jakoby, Tunable passive phase shifter for microwave applications using highly anisotropic liquid crystals, in IEEE MTT-S International Microwave Symposium Digest, Fort Worth, TX, June 2004, pp. 1153–1156. 3. C. Qingjiang, Q. Li, Z. Ziyang, Q. Min, Y. Tong, and S. Yikai, A tunable broadband photonic RF phase shifter based on a silicon microring resonator, IEEE Photonics Technol. Lett., vol. 21, no. 1, pp. 60–62, January 2003. 4. G. E. Erker, S. A. Nagra, L. Yu, P. Periaswamy, R. T. Taylor, J. Speck, and R. A. York, Monolithic Ka-band phase shifter using voltage tunable BaSrTiO3 parallel plate capacitors, IEEE Microw. Guided Wave Lett., vol. 10, no. 1, pp. 10–12, January 2000. 5. G. M. Rebeiz, RF MEMS Theory, Design, and Technology, John Wiley & Sons, 2003. 6. S. Lee, J.-H. Park, H.-T. Kim, J.-M. Kim, Y.-K. Kim, and Y. Kwon, Low-loss analog and digital reflection-type MEMS phase shifters with 1:3 band width, IEEE Trans. Microw. Theory Tech., vol. 52, no. 1, pp. 211–219, January 2004. 7. D.-W. Kang, H. D. Lee, C.-H. Kim, and S. Hong, Ku-band MMIC phase shifter using a parallel resonator with 0.18-µm CMOS technology, IEEE Trans. Microw. Theory Tech., vol. 54, no. 1, pp. 294–301, January 2006. 8. K. J. Kwang , and G. M. Rebeiz, 0.13-µm CMOS phase shifters for X-, Ku-, and K-band phased arrays, IEEE J. Solid-State Circuits, vol. 42, no. 11, pp. 2535–2546, November 2007. 9. B. Min, and G. M. Rebeiz, Single-ended and differential-band BiCMOS phased array frontends, IEEE J. Solid-State Circuits, vol. 43, no. 10, pp. 2239–2250, October 2008. 10. K.-J. Koh, and G. M. Rebeiz, A 6–18 GHz 5-bit active phase shifter, in IEEE MTT-S International Microwave Symposium Digest, Anaheim, CA, May 2010, pp. 792–795. 11. J. Y. Choi, M.-K. Cho, D. Baek, and J.-G. Kim, A 5–20 GHz 5-bit true time delay circuit in 0.18 μm CMOS technology, J. Semicond. Tech. Sci., vol. 13, no. 3, pp. 193–197, June 2013.

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12. C. D. Nordquist, C. W. Dyck, G. M. Kraus, C. T. Sullivan, F. Austin, P. S. Finnegan, M. H. Ballance, Ku-band six-bit RF MEMS time delay network, in 2008 IEEE Compound Semiconductor Integrated Circuits Symposium, Monterey, CA, October 2008, pp. 1–4. 13. M. A. Morton, and J. Papapolymerou, A packaged MEMS-based 5-bit X-band high-pass/lowpass phase shifter, IEEE Trans. Microw. Theory Tech., vol. 56, no. 9, pp. 2025–2031, August 2008. 14. B. Pillans, L. Coryell, A. Malczewski, C. Moody, F. Morris, and A. Brown, Advances in RF MEMS phase shifters from 15 GHz to 35 GHz, in IEEE MTT-S International Microwave Symposium Digest, Montreal, June 2012, pp. 1–3. 15. M. Unlu, S. Demir, and T. Akin, A 15–40-GHz frequency reconfigurable RF MEMS phase shifter, IEEE Trans. Microw. Theory Tech., vol. 61, no. 8, pp. 2397–2402, August 2013. 16. S. Dey, and Shiban K. Koul, Design and development of a CPW-based 5-bit switched-line phase shifter using inline metal contact MEMS series switches for 17.25 GHz transmit/receive module application, J. Micromech. Microeng., vol. 24, no. 1, 24 pages, November 2013. 17. S. Dey, and Shiban K. Koul, Design, development and characterization of an X-band 5 bit DMTL phase shifter using an inline MEMS bridge and MAM capacitors, J. Micromech. Microeng., vol. 24, no. 9, 095007, June 2014. 18. S. Dey, and Shiban K. Koul, 10–25-GHz frequency reconfigurable MEMS 5-bit phase shifter using push-pull actuator based toggle mechanism, J. Micromech. Microeng., vol. 25, no. 6, pp. 1–20, May 2015. 19. S. Dey, and Shiban K. Koul, Reliability analysis of Ku-band 5-bit phase shifters using MEMS SP4T and SPDT switches, IEEE Trans. Microw. Theory Tech., vol. 60, no. 9, pp. 2863–2874, 2012. 20. G.-L. Tan, R. Mihailovich, J. Hacker, J. DeNatale, and G. M. Rebeiz, Low-loss 2- and 4-bit TTD MEMS phase shifters based on SP4T switches, IEEE Trans. Microw. Theory Tech., vol. 51, no. 1, pp. 297–304, January 2003. 21. S. Gong, H. Shen, and N. S. Barker, A 60-GHz 2-bit switched-line phase shifter using SP4T RF-MEMS switches, IEEE Trans. Microw. Theory Tech., vol. 59, no. 4, pp. 894–900, April 2011. 22. H. S. San, X. Y. Chen, P. Xu, G. Li, and L. X. Zhan, Using metal insulator-semiconductor capacitor to investigate the charge accumulation in capacitive RF MEMS switches, Appl. Phys. Lett., vol. 93, no. 6, pp. 063506-1–063506-3, August 2008. 23. S. Dey, and Shiban K. Koul, Systematic measurement of high isolation DC–20 GHz miniature MEMS SPDT switch, Microw. Opt. Technol. Lett., vol. 58, no. 5, pp. 1154–1159, May 2016.

11 Reliability Analysis of MEMS Switches and Phase Shifters

11.1 Introduction The two 5-bit phase shifters that are discussed in Chapter 7 and Chapter 9 are challenged to achieve low loss and good matching simultaneously within a small area and with good repeatability. These challenges become even more critical with higher-bit configurations at lower microwave frequency (typically 4 bits) configurations at lower microwave frequency. In this chapter, to further improve the phase shifter performance in terms of loss, matching, size and reliability, a new topology is proposed that drastically reduces the number of switch counts in a 5-bit digital phase shifter at 17 GHz frequency. This work is primarily focused on the development of a 5-bit switched-line phase shifter using four MEMS single-pole four-throw (SP4T) and two single-pole double-throw (SPDT) switches. The phase shifter is primarily required to be operated over the frequency band of 16.75–17.25 GHz (500 MHz bandwidth); however, its performance in the entire Ku-band is discussed. Two phases of design, fabrication and characterization are performed to ensure optimum phase shifter performance. It leads to the overall performance improvement compared to each other in terms of loss, matching, phase accuracy and reliability.

11.2 Phase Shifter Design Topology The proposed design schematic of the 5-bit true-time-delay (TTD) phase shifter model is shown in Figure 11.1. Compared with the conventional switched-line phase shifter discussed in Chapter 6, where a minimum of 10 switches are actuated at a time, the 197

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FIGURE 11.1 Schematic diagram of the 5-bit phase shifter using four SP4T and two SPDT switches. (Used with permission of IEEE.)

present design requires only 6 switches to be actuated at a time to activate a phase state for 5-bit operation. The proposed topology contains fine-bit (0°, 11.25°, 22.5°, 33.75°), coarse-bit (0°, 45°, 90°, 135°) and 1-bit (0°, 180°) sections. Results of the SPST (singlepole single-throw) and SP4T switches used in the proposed topology are discussed in Chapter 3, Section 3.2. To improve simplicity and reliability from the proposed topology, a small footprint (130 × 40 µm2) of the cantilever beams is used for switching. It introduces the following salient features in the design: 1. The switch is less sensitive to stress due to its small size with a fast switching time. 2. A single-contact cantilever switch is less sensitive to planarity and stress, which significantly improves the overall contact force and equal division of electrostatic force over all different paths in the phase shifter. 3. A multicontact cantilever switch is very prone to single-contact failure (one contact permanently stuck down) or an actuator failure (permanently up). Singleswitch failure can completely damage the overall phase shifter performance for long-range operations. 4. Multicontact, long and complex designs of cantilever switches are sensitive to stress gradient. The residual stress effects the uneven distribution of tip deflection between all identical structures. Hence, in most of the cases different blocks need different voltages to actuate. It decreases overall yield of the device where at a time six switches are actuating. 5. A simple cantilever beam can easily be placed on the coplanar waveguide (CPW) line, which substantially improves the compactness in the SP4T and SPDT structures. 6. Also, due to its small thickness (2 µm), it can be easily packaged using a thin-film package, which is very compatible with the CMOS process. Individual primary fine-bit (2-bit), coarse-bit (2-bit) and 1-bit sections are fabricated and tested individually to ensure the complete phase shifter performance. Later, all of them are cascaded together to build the final 5-bit phase shifter. The same fabrication process discussed in Chapter 3, Section 3.2 is adopted here.

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11.3 2-Bit and 1-Bit Phase Shifter Design and Analysis: Phase 1 The microscopic images of fine-bit, coarse-bit and 1-bit phase shifters fabricated under phase 1 are shown in Figure 11.2. The overall size of the fine-bit section is 3.84 mm2 and the coarse-bit section is 6.45 mm2. Two 2-bit phase shifters (fine and coarse bits) are designed using four delay lines and two SP4T switches connected to the input and output transmission lines. Electrical lengths of different bits are designed with respect to the reference line at 17 GHz. The design scheme of the reported phase shifter is motivated by the work reported in Tan et al. [2] and Gong et al. [10]. The electrical length of the reference line is optimized to be 429 µm (electrical length: 210 at 17 GHz) for both the sections. All three delay lines from the fine- and coarse-bit sections contain a section equal to the reference line plus additional delay line to obtain the desired phase shift. Hence, line length of the delay lines is 690 µm (11.25°), 945 µm (22.5°), 1180 µm (33.75°), 1560 µm (45°), 2664 µm (90°) and 3662 µm (135°), respectively. All different delay lines are optimized using a fullwave simulator (HFSS) to achieve the desired phase response. A 90° CPW bend is used at each corner of the line and optimized accordingly to achieve little transmission distortion caused by intracoupling in the line. Inductive bends are also used on the line to overcome the excitation of coupled slotline modes at CPW discontinuities [11]. To check the effect of coupling between various delays lines, an eight-port simulation is done in HFSS. The isolation is better than 32 dB in all cases over the entire Ku-band. Because 11.25°/22.5° and 45°/135° lines are closely packed parallel with each other, the coupling observed is 6–9 dB more (S73) than other combinations of lines, as depicted in Figure 11.3. The high impedance (30–50 KΩ) bias lines (made by titanium tungsten) are 12 µm wide and covered with SiO2 (0.7 μm). So, it can route very easily underneath the t-line and this improves the simplicity of the device with negligible effect on the insertion loss and low leakage in RF signal. The individual 2-bit design is verified using HFSS simulation with return loss better than 28 dB and insertion loss less than 0.5 dB and no off-path resonance is observed over the band of interest. A sanity check of two complete 2-bit phase shifters is carried out in ADS with all bend (90° bends and inductive bends) data taken from HFSS and with measured SP4T response for the completeness.

FIGURE 11.2 Microscopic images of (a) fine- and (b) coarse-bit sections of the 5-bit phase shifter. (Used with permission of IEEE.)

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FIGURE 11.3 Simulated coupling among the delay lines: (a) fine-bit section and (b) coarse-bit section. (Used with permission of IEEE.)

FIGURE 11.4 Microscopic image of a 1-bit section of the 5-bit phase shifter. (Used with permission of IEEE.)

Microscopic image of fabricated 1-bit section is shown in Figure 11.4 and it uses four switches. The total area of the phase shifter is 3.723 mm2 (1.7 mm × 2.19 mm). The length of the reference (0°) and delay (180°) lines are 388 µm and 4078 µm, respectively. Also, 900 CPW bends are used at each corner to overcome the intracoupling effect. The optimized dimensions of the 90° bends used all over the device are also depicted in Figure 11.4. The 1-bit phase shifter performance analysis is carried out using HFSS and with lumped model in ADS.

11.4 2-Bit and 1-Bit Phase Shifter Measurements: Phase 1 After successful fabrication of individual primary phase bit blocks, switch initial deformations are observed under an optical profilometer. It shows 0.4 µm to 0.57 µm variation in tip deflection of the switch along the length of the cantilever. It leads to 3.97–5.3 fF

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201

FIGURE 11.5 Measured versus simulated S-parameters of the individual fine-bit section. (a) Return loss and (b) insertion loss over the entire Ku-band. (Used with permission of IEEE.)

FIGURE 11.6 Measured versus simulated S-parameters of the individual coarse-bit section. (a) Return loss and (b) insertion loss over the entire Ku-band. (Used with permission of IEEE.)

variation in Coff (at zero bias) and 2.8–3.7 Ω variation in Rc (applied bias) throughout different delay lines with 53 V actuation voltage. The equivalent circuit model of the SPST and SP4T switches are presented in Figures 3.1 and 3.5, respectively (see Chapter 3). The measured return loss of better than 21 dB and less than 0.82 dB of insertion loss are obtained between 13 and 18 GHz from the fine-bit section as shown in Figure 11.5. The coarse-bit section provides return loss of better than 24 dB (20 dB between 13 and 15 GHz) and worst-case loss of 0.92 dB (at 135°) over 16–18 GHz, as shown in Figure 11.6. Figure 11.7 shows the measured versus simulated S-parameters of the individual 1-bit section. Phase versus frequency response of the fine and coarse bit sections are shown in Figure 11.8a and b. A maximum phase error of 0.7° is observed experimentally at 17 GHz from both sections. Moreover, the 1-bit section exhibited measured return loss of better than 27 dB, maximum insertion loss of 1.16 dB and phase error of 0.58° at 17 GHz (see Figure 11.7 and Figure 11.8c). It may be noted that the dotted lines in Figures 11.5 through 11.8 represent the simulated results.

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FIGURE 11.7 Measured versus simulated S-parameters of the individual 1-bit section. (a) Return loss and (b) insertion loss over the entire Ku-band. (Used with permission of IEEE.)

FIGURE 11.8 Measured versus simulated phase versus frequency response of (a) fine-bit, (b) coarse-bit and (c) 1-bit sections over the entire Ku-band. (Used with permission of IEEE.)

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11.5 Design and Measurements of the 5-Bit Phase Shifter: Phase 1 The microscopic image of the complete 5-bit phase shifter is shown in Figure 11.9. The complete area of the phase shifter is 5.17 × 3.19 mm2, including all bias lines and bias pads. All three (two 2-bit and 1-bit) sections are cascaded together and simulated using HFSS and verified in ADS for completeness. An unwanted resonance is observed in full-wave simulation at 11 GHz over 8–18 GHz. Although the dip in the simulated insertion-loss response is out of the band, after fabrication the dip can shift toward the operating band (13–18 GHz) due to added parasitics and line capacitances. To overcome this off-path resonance from different phase bits, the connecting line length between the fine- to coarse-bit (L1) and coarse- to 1-bit sections (L2) are optimized to values of 383 µm and 344 µm, respectively. To ensure optimum phase shifter performance with good phase accuracy, multiple line length optimizations are carried out and the results are shown in Figure 11.10. As a result, the effect of off-path resonance is completely eliminated with almost flat response in insertion loss in all the phase states over the band. Moreover, one inductive section is optimized and introduced at the middle of the connecting lines (L1 and L2) between two sections. Simulated average return loss of 22 dB and insertion loss of 3.3 dB is obtained from 13 to 18 GHz with maximum phase error of 0.38° at 17 GHz.

FIGURE 11.9 Microscopic images of the Ku-band 5-bit TTD phase shifter fabricated under phase 1. (Used with permission of IEEE.)

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FIGURE 11.10 Effect of different connecting line lengths on S-parameters. (Used with permission of IEEE.)

The S-parameters and group delay performance of the 5-bit phase shifter is measured systematically over the band. The results show that matching (S11 and S22) is better than 19 dB over 0.1–18 GHz and average insertion loss is 3.89 dB within 13–18 GHz, as shown in Figure 11.11a and b. The maximum phase error is 1.14° at 17 GHz in 258.75° phase state. The primary phase versus frequency response is shown in Figure 11.11c. The Rc is maximum (4.3 Ω) in the 258.75° state and minimum (2.4 Ω) at the 0° state. The maximum group delay of ~182 ps with the delay step of ~4.67 ps is obtained at 17 GHz, as shown in Figure 11.11d. Finally, the present phase shifter demonstrates figure of merit (dB/bit) of 0.9 at 17 GHz. Nonuniform tip deflections (0.4 µm to 0.54 µm) in all 20 MEMS switches are observed which can be attributed to contact resistance variation with the same bias voltage (53 V).

11.6 Reliability Measurements of the SPST and SP4T Switches: Phase 1 To observe the phase shifter reliability over a practical environment, a few more tests are carried out. Reliability and power measurements are initially performed on the SPST and SP4T switches at 2 GHz with various levels of RF power. As a matter of fact, RF frequency is not a primary parameter to measure DC-contact switch reliability or power handling capability. To ensure optimum phase shifter performance, where all six switches are performing at a time, an extensive measurement process is carried out. Initially, the switch actuation voltage is measured with different incident power levels. Later, switch-contact resistance (Rc) variations are extensively observed at different power and temperature scales. 11.6.1 Temperature Stability Measurements on the MEMS Switch The temperature stability of the MEMS switch is observed by measuring the change in Vpi and Vr voltages as a function of temperature. The 3D measured profile of the switch shows ~270 nm downward deformation from a 130 µm long cantilever beam between 25°C and 85°C. It indicates the presence of thermomechanical behavior. A temperature controller (Temptronic Corporation, Mansfield, Massachusetts) is attached to the chuck of the probe

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station. It is used to set a stable operating temperature during the measurement. The chuck temperature is increased from 25°C up to 85°C and then again decreased to room temperature. The switch voltage changes  15 V from 25°C to 70°C. During this process, maximum Vpi changes from 43–27 V, but abrupt change in Vr is observed. It is mostly due to the dielectric charging with incident power due to increase in temperature and also due to contact point degradation and contaminations with additional attractive force from the RF power [14]. For clean metal contact, the incident RF power (Pinc) and contact voltage (Vc) are represented by Equations 11.5 and 11.6, respectively, as a function of temperature, where (Rc 100 M cycles without any self-actuations or failure. The maximum power handling limitation (Pmax) of this kind of phase shifter can be defined by Equations 11.11 to 11.13 [13]: Vopen Iclosed (11.11) 2



Pmax =



Vopen = 2 Z0 Pmax (11.12)



Iclosed =

Pmax (11.13) Z0

where Vopen is the open-state voltage of the DC-contact switch, Iclosed is the closed-state current of the switch and Z0 is the characteristic impedance of the transmission line. In the reported phase shifter, the contact switch can withstand high open-state voltage (~40 V), and therefore the main power handling limitation comes from the current density in the closed state. For Pmax = 1 W and Z0 = 50 Ω, the Vopen and Iclosed values are 14.14 V and 0.14 A, respectively. Note that the switch can withstand 14 V of open-state voltage (switch pull-in at 43 V), but it fails with 0.14 A of closed-state current over large cycles in the phase shifter where six switches are performing at a time. Moreover, large ohmic heat dissipation takes place during the contacting period and that also affects the linearity performance of the switch [9].

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Latching (VRFrms > Vr) and self-actuations (VRFrms > Vpi) are also observed in some of the cases even at 1 W of RF power after a few million cycles. To improve the reliability of the reported phase shifter, one solution could be the reduction of Iclosed with high Z0 (>50 Ω), but it will increase losses and degrade input/output matching. Low sensitivity of stress with temperature rise and nonuniform heating on the beam can be improved with a thicker beam (thickness >4 µm) or stiffer cantilever beam structure. Efficient heat transfer through anchors and medium value spring constant with high release voltage are very essential facts to improve the MEMS device reliability for high power applications [13]. All these aspects are taken care of in the next phase (phase 2) to improve the overall phase shifter reliability up to a reasonable extent.

11.8 Design Modification and Measurements of the Phase Shifter: Phase 2 To improve the performance and reliability of the overall 5-bit phase shifter, a few design modifications are carried out during phase 2. These are listed next. 1. In this work, the DC-contact switch has dimensions of 90 µm × 30 µm with a dimple dimension of 12 µm × 12 µm × 1 µm. It results in negligible tip deflection (140 nm upward) after the O2 plasma dry release process. It is mostly due to the miniature switch profile that is less sensitive to stress gradient and residual stress [55]. The main purpose of using the miniature switch profile in phase 2 is to improve the yield of the switch and that reflects on the overall phase shifter performance. 2. The switch is implemented using an 18/40/18 µm (50 Ω) CPW line. It makes a more symmetric and compact SP4T switch. The SP4T switch fabricated in Phase 2 has a total area of 0.55 mm2, which is 24% more compact than the phase 1 SP4T switch. It also leads to the reduction in the overall phase shifter area and hence cost, since area is directly proportional to the cost in large volume manufacturing processes. 3. Small in-line cantilever switches and lower CPW dimensions (G/W/G) are more favorable in phase 2 because they permit the switches to be placed very close to one another. It also improves the parasitic inductive effect of the CPW lines between the central junction and the switches. It leads to the improvement on matching over the entire Ku-band. 4. A few more design parameters like junction capacitance (Cj), spoke length, inductive bends and bias line resistivity are modified during this phase to accelerate the phase shifter performance. Note that the fabrication process steps of phase 2 are similar to those of phase 1.

The actuated switch has measured pull-in voltage (Vpi) of 62 V and a stable contact appears to be at 70 V. Measured switching time is 13 to 9 µs for 62–80 V with overall settling time of ~16 µsec due to the switch bounce. The release time is 5 µs with a final settling time of 7.6 µs. The present switch gives 52 kHz of mechanical resonance frequency with a Q factor of 8.7. The switch demonstrates measured return loss of better than 31 dB, worst-case insertion loss of 0.16 dB and isolation of 30 dB up to 18 GHz. The port definition of the SP4T switch is similar as in phase 1. To eliminate the unwanted off-path resonance and to improve the matching, a 4.4 fF of Cj is introduced on the input

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line on the SP4T switch. Furthermore, spoke length was also reduced to 22 µm, which leads to the significant improvement in matching. The line length of the input signal line and the other four arms are also optimized to achieve good wideband matching. As a result, the SP4T switch demonstrates an excellent performance with maximum 0.23 dB loss and return losses better than 27 dB at all ports up to 18 GHz as shown in Figure 11.21a. An average measured isolation of 30.2 dB is obtained over the band with all OFF-port conditions, which is 1.8 dB worse than the isolation with one port (P2) in the ON condition (Figure 11.21a shows the average isolation in all OFF-port conditions). The measured return loss better than 24 dB up to 18 GHz and insertion loss of 0.7 dB is obtained with 70 V bias voltage between 13–18 GHz from the fine-bit section, as shown in Figure 11.21b,c. The coarse-bit section gives return loss better than 25 dB and less than 0.86 dB of insertion loss, respectively (Figure 11.21b and c). Phase error less than 0.48° (33.75°) and –0.33° (45°) are obtained from fine- and coarse-bit sections, respectively, at 17 GHz as shown in Figure 11.21d. The 1-bit section demonstrates return loss of better than 28 dB, and worst-case insertion loss of 0.98 dB from both states with 0.26° of phase error over the band of interest. Microscopic images of the fine-bit and coarse-bit section of 5-bit phase shifter are shown in Figure 11.22. The total area of the fine-bit section is 2.6 mm2 and the coarse-bit section is 5.6 mm2. The optical profilometer shows very negligible variation (0.092 µm–0.17 µm) in tip deflection on switch profile of the overall 2-bit phase shifters (both fine- and coarse-bit sections). As a result, variation in Coff and Ron are much less throughout different states on the phase shifters for the same bias voltage. Inductive sections and 90° CPW bends are also introduced here to overcome the intracoupling effect and any unwanted resonance. As in

FIGURE 11.21 Measured S-parameter response of the (a) SP4T switch. Response of individual sections of 5-bit phase shifter: (b) return loss, (c) insertion loss and (d) phase versus frequency response. (Used with permission of IEEE.)

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FIGURE 11.22 Microphotograph images of (a) fine- and (b) coarse-bit sections of the 5-bit phase shifter. (Used with permission of IEEE.)

FIGURE 11.23 Microscopic image of the Ku-band 5-bit phase shifter fabricated from phase 2. (Used with permission of IEEE.)

phase 1, line lengths of different reference and delay lines of two 2-bit phase shifters are also optimized using HFSS. Microscopic images of the 5-bit phase shifter and complete schematic are shown in Figures 11.23 and 11.24, respectively. The total size of the phase shifter is 13 mm2 (including bias pads and bias lines). All three sections are cascaded together and connecting line lengths (L1 and L2) are also optimized to be 333 µm and 298 µm, respectively. Return loss is better than 22 dB up to 18 GHz and average insertion loss is less than 2.65 dB from all the states over 13–18 GHz with 70 V, as shown in Figure 11.25. The maximum average phase error is 0.68° at 17 GHz, which shows ~40% improvement compared to phase 1. Furthermore, the maximum measured group delay of ~159 ps with the delay step of ~4.06 ps is achieved at 17 GHz. The phase shifter also demonstrates 0.7dB/bit figure of merit (FOM) at 17 GHz. Performance comparison between phase 1 and phase 2 phase shifters is listed in Table 11.1. Detailed phase shift and loss data are tabulated in Table 11.2.

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FIGURE 11.24 Schematic and the equivalent circuit of the complete 5-bit phase shifter at 0° (reference) state. (Used with permission of IEEE.)

11.9 Reliability Measurements on Switches: Phase 2 Similar to phase 1, here switch reliability is systematically characterized under different temperatures and RF power levels. Hot- and cold-switched reliability are also measured and presented in the subsequent sections.

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FIGURE 11.25 (a) Measured return loss performance of 5-bit phase shifter up to 18 GHz. (b) Measured versus simulated average insertion loss over 13–18 GHz. (Used with permission of IEEE.)

TABLE 11.1 Performance Comparison between Phase 1 and Phase 2 Phase Shifters at 17 GHz Phase Shifter

S11 (dB) (avg)

S21 (dB) (avg)

Phase Error (avg)

Group Delay (ps)

Size (mm 2)

19 22

4.2 3.65

1.14° 0.35°

182 159

16.4 13

Phase 1 Phase 2

TABLE 11.2 Performance Summary of the 5-Bit Phase Shifter over 8–12 GHz Phase State   0°  11.25°  22.5°  33.75°  45°  56.25°  67.5°  78.75°  90° 101.25° 112.5° 123.75° 135° 146.25° 157.5° 168.75°

S21 (dB)

∆Φ (deg)

∆ΦE (deg)

Phase State

S21 (dB)

∆Φ (deg)

∆ΦE (deg)

1.65 1.71 1.79 1.82 1.86 1.96 2.05 2.09 2.15 2.26 2.32 2.37 2.41 2.5 2.6 2.63

0   11.28°   22.78°  34.1°  44.9°   56.33°   67.72°  79.2°  89.78  101.28° 112.7° 124.2°  134.87°  146.93° 157.8° 169.1°

0 0.03° 0.28° 0.35° −0.1° 0.08° 0.22° 0.45° −0.22° −0.03° 0.2° 0.45° −0.13° 0.68° 0.3° 0.35°

180.00° 191.25° 202.50° 213.75° 225.00° 236.25° 247.50° 258.75° 270.00° 281.25° 292.50° 303.75° 315.00° 326.25° 337.50° 348.75°

2.66 2.75 2.83 2.87 2.91 2.97 3.15 3.55 3.19 3.27 3.34 3.4 3.47 3.53 3.58 3.72

179.7° 191.04° 202.4° 213.81° 224.5° 235.8° 246.8° 257.87° 269.3° 280.76° 292.2° 303.53° 314.37° 325.74° 336.72° 349.3°

−0.3° −0.21° −0.1° 0.06° −0.5° −0.45° −0.7° −0.88° −0.7° −0.49° −0.3° −0.22° −0.36° −0.51° −0.78° 0.55°

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FIGURE 11.26 (a) Measured pull-in and release voltages of SPST switch versus temperatures. (b) Switch power handling response at different temperature scales. (Used with permission of IEEE.)

11.9.1 Temperature Measurements of the MEMS Switch The temperature stability is observed on the SPST switch by measuring the change in Vpi and Vr as a function of temperature, as depicted in Figure 11.26a. The measured 3D profile shows negligible difference (114 nm) from 25°C to 85°C on the switch profile, which indicates a robust thermomechanical performance than the phase 1 switch performances. The switch shows a variation of 30 V is quite good enough for longtime operation. The switch performance started to degrade between 3 and 4 W after 50°C with a 23.6 V reduction in voltage. Again, it is mostly due to the contact point degradation and contaminations at high RF power and with an elevated temperature [14]. 11.9.3 Reliability Measurements of MEMS Switches with RF Power Switch Rc is measured periodically using four-point probes to ensure the switch performance at different RF powers. During this process, both SPST and SP4T switches are cycled up to 10 M cycles and corresponding changes in Rc are recorded at five different RF powers (0.5 W, 1 W, 2 W, 3 W and 4 W) at 2 GHz. All measurements are carried out at room temperature with 70 V, as shown in Figure 11.27. As similar to phase 1, Rc is recorded after each 100 cycles to ensure the stiction-free condition. Results show that the SPST switch can handle >10 M cycles up to 3 W with no contact failure or stiction. But, reliability started degrading

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RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 11.27 Cold-switched reliability of the SPST and SP4T switches at 25°C. (Used with permission of IEEE.)

for 3–4 W of RF power with an abrupt change in Rc after ~105 cycles and failed after ~1 M cycles. Finally, the SPST switch can handle 2.5–3 W of RF power up to 10 M cycles at 25°C. The SP4T switch is also characterized during this process. Each arm of the switch is actuated independently with 0.8–1.1 mN of contact force. During this cold-switched reliability process, the reported SP4T switch can withstand up to 1.5–2 W of incident power up to 10 M cycles and 3 W for 105 cycles without failure of any of the contacts (see Figure 11.27). Again, to ensure the phase shifter performance, five identical SP4T switches are tested up to 100 M cycles with 0.5–1 W of power and the test is completed without failure. 11.9.4 Reliability Measurements of MEMS Switches with RF Power and Temperature Figure 11.28a and b present the reliability results of the SPST and SP4T switches with 0.1–1 W at 50°C and 70°C, respectively. The SPST switch shows that for an incident power of 0.1–0.5 W, the reliability is >50 M cycles at 50°C and >20 M cycles at 70°C. At 1 W, the reliability measured is >1 M cycles at both temperatures, as depicted in Figure 11.28a. The SP4T switch reliability is >1 M cycles with 0.5 W of power at both temperatures, but it quickly fails (~105 cycles) with 1 W power at 70°C, as shown in Figure 11.28b. Microwelding typically limited the switching lifetime under hot-switching conditions and especially for the gold–gold contact [13]. Finally, the switch is actuated with 70 V bias voltage and corresponding changes in switch actuation voltages are recorded under prolonged hold-down conditions. The Vpi drifts 2.4–3.8 V after 6 h during this stress relaxation process. This drift in switch actuation voltage is primarily due to the dislocation glide in the material, which depends on material properties, temperature stress and activation energy.

11.10 Reliability Measurements of the Phase Shifter: Phase 2 Phase shifter reliability is observed on a chip with 0.1 W of power at 25°C. Initially, reliability performances are observed on individual sections (fine-bit, coarse-bit and higher-bit), prior to starting with the complete 5-bit. All sections are measured up to 100 M cycles to

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FIGURE 11.28 Reliability of (a) SPST and (b) SP4T switches for 0.1–1 W of RF power at 50°C (top) and 70°C (bottom) with 70 V actuation voltage. (Used with permission of IEEE.)

FIGURE 11.29 (a) Reliability performances of individual sections of the 5-bit phase shifter. (b) Performance comparison of the 5-bit phase shifter on a chip and in a package, at 17 GHz with 0.1 W of RF power. (Used with permission of IEEE.)

ensure the optimum phase shifter performance. The reliability performance of this 5-bit phase shifter is entirely driven by upper limit of higher-bit section reliability, as is also observed in phase 1. No failure is observed up to 100 M cycles of operation in all fundamental blocks of the 5-bit phase shifter. Results show maximum average loss and phase error of 1.64 dB and ~1.1°, respectively, at 17 GHz, as shown in Figure 11.29a.

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FIGURE 11.30 Simulated performance variation between chip and module with (a) return and insertion loss and (b) phase shift at reference state of the phase shifter. (Used with permission of IEEE.)

Finally, phase shifter reliability is measured on a chip and within a module (module area is 12 × 10 mm2) and shown in Figure 11.29b. Results show ~20% degradation in average loss and phase errors compared to the bare die results with 0.1 W of RF power at room temperature and at 17 GHz frequency. These changes are mostly due to the added bond wire parasitic and input and output transitions. Note that all cable losses are calibrated out before the measurements in the package form. Moreover, variation in loss and phase shift from bare die to package is also validated using a full-wave simulation in the reference state and results show 15%–20% tolerances due to added parasitic, as shown in Figure 11.30a and b, respectively. 11.10.1 Phase Shifter Reliability Measurements with RF Power Figure 11.31a shows phase shifter reliability results at four power levels (0.1 W, 0.5 W, 1 W and 2 W) at 25°C. Results show the phase shifter works satisfactorily up to >30 M cycles with maximum average loss and phase error of 5.34 dB and ~2.8°, respectively, with 0.5 W–1 W of power levels. Furthermore, failure in the phase shifter is observed after 30 and 5 M cycles at 1 W and 2 W of power level, respectively. In most of the cases, failure occurs in the higher-bit section. At 1–2 W, the total rms current in the switch is 0.14–0.2 Arms and each contact is handling 35–50 mArms. Thus, reliability of the phase shifter at high power level is limited by the current density in closed state and due to the effect of electromigration like Joule heating [13]. 11.10.2 Phase Shifter Reliability Measurements with RF Power and Temperature Phase shifter reliability is checked at two different temperatures (50°C and 70°C) with 0.5 W and 1 W of RF power. The phase shifter performs satisfactorily (>10 M cycles) at 5 M cycles at both temperatures. At 1 W, reliability is >5 M at 50°C and >1 M at 70°C. Maximum average loss and phase error variation during this process are 5.8 dB and 3.78°, respectively. Note that all power handling and reliability measurements of the reported phase shifter are carried out under the CW mode of operation. Phase shifter performance may change under the pulse mode conditions because of different physical failure phenomena. A quick simulation shows the phase shifter can sustain up to 380°C and average 20%–30% improvement in loss with a 4 µm thicker beam and rhodium as a contact material.

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FIGURE 11.31 Reliability of the phase shifter with (a) 0.5–2 W of power under cold-switched conditions and (b) for 0.5–1 W of power and at two different temperatures (50°C and 70°C) under hot-switched conditions. (Used with permission of IEEE.)

FIGURE 11.32 Measured change in loss and phase error at (a) 25°C and (b) 50°C over 6 h of prolonged actuation (ON-state) with 0.1 W of power at 17 GHz. (Used with permission of IEEE.)

11.10.3 Phase Shifter Testing under Prolonged Actuation To observe the phase shifter performance under prolonged actuation conditions, one more on-wafer test is done here. The phase shifter is measured under the ON-state condition at 25°C and 50°C temperatures with 0.1 W of RF power and with 65–70 V bias. The process is continued up to 6 h, and corresponding changes in loss and phase error are recorded after every 10 min. Although Figure 11.32 shows the measured responses at three different phase states, all 31 states are observed with respect to the “ref”-state at 17 GHz. During this stress relaxation process, the phase bits show ~1.36 dB (3.55–4.91 dB) of loss variation from the initial value and maximum ~1.24° (0.87°–2.11°) of phase error at 17 GHz and at 25°C. This variation is greater at 50°C with an extra 1.88 dB (3.57–5.44 dB) of loss and ~1.8° (0.87°–2.68°) of phase error. These variations can be justified by (11.14)

(

2

Pin 1 − S11 − S21

2

) × t = ms∆T (11.14)

where s is the specific heat of the material (0.129 J/g°C at 25°C for gold), m is the mass and ∆T is the change in temperature.

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RF Micromachined Switches, Switching Networks, and Phase Shifters

Equation 11.14 expresses that the maximum power loss as heat in a given time (t = 6 h is this case) is proportional to the heat dissipation from the DUT (m and s are constant here) over a prolonged ON-state condition. It leads to an increase in temperature by ∆T and affects the sensitivity of the S-parameter. This measurement is entirely limited by the time where beam curvature decreases with time. Moreover, the sensitivity of the S-parameter can change further with an increase in power and with longer time of operation (t). This effect can be improved further with the appropriate choice of beam material like aluminum alloy and well-suited contact material like rhodium or gold–palladium alloys.

11.11 Qualification Testing of the Phase Shifter: Three-Axis Vibration Three-axis vibration testing is performed in phase 2 for the qualification testing. The phase shifter is encapsulated within a module to measure its performance with externally applied vibration and shock. An electrodynamics shaker table made by Unholtz-Dickie is used during testing. Vibration modes from the shaker table are controlled by a power amplifier and a DC power source. The phase shifter is accelerated over 0.04 g2 Hz−1 of power spectral density (PSD) and responses are recorded from 20 Hz to 2 kHz in all three axes. The measured response is taken with 20 Hz to 80 Hz under +3 dB/octave, 80 Hz to 350 Hz under PSD, and 350 Hz to 2 kHz under –3 dB/octave. Each axis measurement is carried out for 5 min durations. The phase shifter is sustained within a safety limit of 6.47 grms at 0.04 g2 Hz−1 PSD. The PSD value is higher the in z-axis compared to other axes as shown in Figure 11.33. The g-level rms value can be found from PSD × Bandwidth (15,16) and the value is 48.79 g. Phase shifter performance was observed after the vibration test and it worked satisfactorily with no abnormal change in the behavior.

11.12 Failure Analysis of the 5-Bit MEMS Phase Shifter The reliability operation of this phase shifter brings a few interesting facts. Here, one switch cycle is defined by only one actuation state (ON and OFF), but in the case of the phase shifter, one cycle counts 32 states of operation where fine- and coarse-bit switches

FIGURE 11.33 Power spectral density versus frequency for shaker table testing under 0.04 g2 Hz−1 PSD. (Used with permission of IEEE.)

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225

are individually actuated 8 times, whereas in the higher-bit section they are actuated 16 times. So, the probability of failure is always higher at the higher-bit section compared to other sections (fine and coarse bits) over the continuous reliability cycles. So, in this proposed 5-bit topology, a nonuniform switch actuation is found on the device throughout the reliability operation. Figure 11.34 clearly shows switches S9 and S10 (total of four switches) are actuated 16 times compared to other switches on the phase shifter over one complete cycle. This nonuniform switch actuation is the primary reason for the device failure after tens of millions of cycles of operation and it will definitely not be the common case for an even-bit phase shifter such as 4-bit (two SP4T switches) or 6-bit (two SP8T switches). The figure of merit for this kind of RF MEMS device is defined by mean time to failure (MTTF) or mean time to the first failure. Here, in this 5-bit phase shifter, the lower limit of the reliability is defined by MTTF. The reason of MTTF of the proposed phase shifter is not limited to one; multiple reasons of failures are encountered during the reliability test and primarily it is due to the effect from electromigration for contact-type switches and it is defined by Equation 11.15 [13]:

MTTF =

E

A - kTa e (11.15) Jn

where A is the cross section area dependent constant, J is the conduction current density, k is the Boltzmann constant, Ea is the effective activation energy and n is the scaling factor (usually, n is set to 2). The primary reason of MTTF in the proposed 5-bit phase shifter is due to the effect of current crowding, which gives rise to the contact heating with larger J at the operating frequency. Note that mean time to first failure of the device is limited by the factor of 1/J2 over the operation.

FIGURE 11.34 Schematics of individual switch actuations from the 5-bit phase shifter over one complete cycle. (Used with permission of IEEE.)

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11.13 Design Guidelines for a Reliable MEMS 5-Bit Phase Shifter with Alternative Topology









1. Reliability of the proposed 5-bit phase shifter is primarily limited by the number of switch counts per phase state. Moreover, the nonuniform actuation of switch per cycle also leads to an early failure after a few cycles. To further improve the 5-bit phase shifter reliability a new topology is proposed and shown in Figure 11.35. This topology contains two SP8T and two SP4T switches and connecting lines. This design requires only four switches to be actuated at a time to activate one phase state and it leads to a uniform actuation over the cycle. 2. The loss and matching of the phase shifter is entirely limited by the loss of the SP8T and SP4T switching network. Circular-type configuration is very much useful for this kind with a 40° and 72° angle between two in-line series switches for the SP8T and SP4T switching network, respectively. 3. This configuration also permits switches to be placed close together with more compactness without any fabrication difficulties. It leads to the reduction of the overall area of the device up to a few microns or millimeters squared. 4. Matching and loss of the overall phase shifter can be improved by reducing the parasitic inductive effect between the central junction and switches. 5. A few more design parameters like junction capacitance, spoke length and inductive bends are to be placed at each CPW discontinuity to eliminate the higherorder modes. 6. High-resistive bias lines should be critically optimized and routed accordingly without affecting device performance with signal leakage and added parasitic. 7. The connecting line length (lc) between the fine- and coarse-bit sections needs to be optimized using full-wave simulation to nullify the effect of any off-path resonance over the band.

FIGURE 11.35 Schematic of a 5-bit MEMS phase shifter using two SP8T and two SP4T switches. (Used with permission of IEEE.)

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8. The in-line MEMS switch is one of the most favorable options for better matching. In addition, the thicker cantilever beam can significantly improve the reliability of the overall device. 9. Power handling and temperature stability of the reported phase shifter can be drastically improved by proper selection of the contact material with higher softening temperature. 10. Hermetic packaging of the device can significantly improve the reliability with lower contact contaminants. Note that this kind of topology is very much useful at lower microwave frequency (100 M cycles over 0.1–1 W of incident RF power. The

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FIGURE 12.5 Measured (a) phase versus frequency and (b) group delay performances of the 3-bit MEMS phase shifter over 500 MHz. (Used with permission of IOP.)

TABLE 12.1 Phase Shift and Loss Data of the 3-Bit Phase Shifter at 35 GHz Phase State

0°

45°

90°

135°

180°

225°

270°

315°

Measured Phase error Loss (dB)

0° 0° 3.7

46.07° 1.07° 3.94

90.8° 0.8° 4.11

136.19° 1.19° 4.38

180.58° 0.58° 4.59

226.2° 1.2° 4.78

270.9° 0.9° 4.89

316.78 1.78° 5.07

FIGURE 12.6 Measured (a) reliability results at different RF power and (b) change in insertion loss and phase error at 25°C over 6 h of continuous actuation bias with 0.1 W of RF power at 35 GHz. (Used with permission of IOP.)

proposed 3-bit phase shifter works satisfactorily up to >100 M cycles (test is performed up to 100 M cycles) at 0.1–1 W and >30 M cycles at 2 W of incident RF power, respectively. Note that this measurement is performed at room temperature. The maximum average insertion loss and phase error change to 3.67–6.8 dB and 2.16°–5.1°, respectively, at 35 GHz during this cycling process. Although 10 identical phase shifters are measured during this phase, the results show win-win case performances of the phase shifters. The measurement is

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RF Micromachined Switches, Switching Networks, and Phase Shifters

controlled by the current density over the contacting region at higher incident power level and also limited to the charge injection problem after ~103 cycles of operations. 12.2.4 Device Testing under Prolonged Actuation The phase shifter is tested under continuous actuation conditions at 25°C and 50°C with 70–80 V bias voltage. The incident RF power level is set to 0.1 W, and corresponding changes in measured loss and phase error are captured up to 6 h in the ON-state, as shown in Figure 12.6b. Note that results shown here are at three phase states (135°, 225° and 315°), but all eight states are actually measured with reference to the 0° state at 35 GHz with the same setup and same operating conditions. Insertion loss variation of ~1.44 dB (4.23–5.6 dB) and phase error variations of ~1.8° (3.77°–5.03°) are observed at 35 GHz during this process. Device performance is entirely a function of the operation time when the beam curvature gradually decreases with time due to the effect from spring softening. This effect could be improved with an appropriate choice of contact material like rhodium or gold–palladium alloys or a thicker beam layer.

12.3 Design and Measurements of the 4-Bit Phase Shifter Using SP16T Switching Networks Different topologies of a 4-bit phase shifter are designed and optimized using two SP16T switches with different delay and reference line combinations. The final schematic of the proposed 4-bit phase shifter model is depicted in Figure 12.1b. Compared with other 4-bit phase shifter configurations, in which a minimum of eight [5] or four [3] switches are actuated at a time, the present design requires only two switches to be actuated to activate one phase state with the same phase resolution (22.5°). This 4-bit phase shifter uses a total of 32 switching elements out of which only 2 of them are actuated at a time. In addition, a single SPST switch is actuated only one time over one cycle during complete phase shifter operation. It leads to a uniform actuation cycle where one switch cycle is equivalent to one device cycle. It greatly improves the simplicity of the device and thus its reliability. Note that the total number of switches (32) is more in the proposed structure compared to 16 switches used in other studies [3,5]. Hence, reliability is still a concern in the proposed design from a broader perspective, such as a higher chance of switch failure over the cycle. The fabrication process yield plays a very crucial role in this work, and multiple process iterations were performed to ensure optimum performance. Small and simple cantilever beam structures used for switching are less sensitive to planarity and stress, and it improves overall yield of the device. Moreover, the switch can easily be placed on the t-line, which substantially improves the compactness in the SP16T structure. 12.3.1 Design of the SP16T Switch The SP16T switch is a stand-alone component in the modern switched-beam satellite antenna system. A SP16T switch was recently reported with five SP4T switches in which two switches need to be actuated to activate one arm [6]. To improve the design complexity and performance, a new topology of the SP16T switch is presented in this work.

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The SP16T switch is implemented on a CPW line made with 2 µm thick electroplated gold on the 635 µm alumina substrate. This CPW (16 µm/34 µm/16 µm = 50 Ω) line is routed in a circular configuration where it utilizes 16 cantilever-type in-line MEMS switches. The switch is made with 3.5 µm thick electroplated gold with an electrode area of 34 µm × 50 µm, and the total area of the switch is 34 × 110 µm2. The switch is in-line in nature, and electrostatic voltage (V) is applied between fixed electrode and cantilever beam. When the switch crosses the pull-in limit (V > Vpi), it snaps down, connects with the output RF transmission line, and gives insertion (S21) and return (S11) loss. At V = 0, the switch gives isolation (S21). The single switch gives simulated Vpi at 61 V, switching time of 9.74 µs and resonance frequency ( f0) at 45.7 kHz. Figure 12.7 presents schematic diagram and equivalent circuit representation of the proposed symmetric and circular-type MEMS SP16T switching network. A total of 16 MEMS switches are placed in-line and the angle between two consecutive switches is 21.17°. The radius of the dimple is ~5.5 µm with a depth of 1 µm. All CPW ground lines are connected together with air bridges for better matching. As shown in Figure 12.7, RON (3.08 Ω), Coff (19.3 fF) and t-line elements are used to represent different sections. The distance between two consecutive switches is optimized to 138 µm. Near- and far-port performances of the SP16T switch are seen in a finite element solver to ensure better coupling between different lines across different ports. Note that all bias lines (24 kΩ bias resistance) and DC pads are considered in the simulation. In this configuration, port 1 (P1) port 16 (P16), port 2 (P2) port 15 (P15) and port 3 (P3) port 14 (P14) are considered near ports. The remaining ports are far ports as they are 552 µm away from the center point of the common port. The maximum far port distance is 1104 µm. A metal– air–metal capacitance (Cj) is introduced with 20 µm beam width to compensate for the impedance mismatch at the central junction. The effect of its input/output matching (S11 and S22) is investigated up to 26 GHz using a full-wave simulator (HFSS v15), as depicted in Figure 12.8a. Results show that 3.54 fF of capacitance gives better response in terms of matching. The placement of the Cj is an important factor, and it is optimized to be 60 µm from the central point of the T-junction. Moreover, Cj also reduces reflection coming back from the T-junction up to a reasonable extent. Figure 12.8b shows that SP16T switch delivers good matching characteristics up to 26 GHz with return loss of better than 17.6 dB at 20 GHz and 15.4 dB at 26 GHz. Simulated worst-case insertion loss is 1.63 dB and 1.77 dB at 20

FIGURE 12.7 Schematic diagram and equivalent circuit model of the SP16T switch. Note all lumped parameters are obtained from S-parameter measurements. (Used with permission of IEEE.)

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RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 12.8 (a) Simulated input and output return loss of the SP16T switch with different junction capacitances (Cj). (b) Simulated return and insertion loss of SP16T switch using equivalent circuit model and HFSS simulation. (Used with permission of IEEE.)

GHz and 26 GHz, respectively. Matching between central junctions to the near port (P1) is ~3.4 dB better than the far port (P8) response due to the strong electromagnetic coupling associated between common to near port. Note that the far port shows narrower return loss than the near port. Performances of the SP16T switch are mostly dominated by input and output RF lines, length of the switch, Cj and bias line resistivity. 12.3.2 Measurements of the SPST and SP16T Switches Prior to starting with the SP16T switch, single-switch performances were critically evaluated. Initially, an optical profilometer showed initial tip deflection of ~130 nm (upward) with ~2.1 MPa/µm of stress gradient along the length of the cantilever. Switch resonance frequency was tested using laser Doppler vibrometer (LDV) and the measured value is 46.8 kHz with a Q factor of 7.9. Pull-in (Vpi) and release voltages (Vr) of the switch are 72 V and 52.8 V, respectively, and it is measured using an LCR meter. Measured ON-switching time (ton) of 11.2–8.03 µs and OFF-switching time (toff) of 4.78 µs are obtained with an actuation voltage of 72–85 V. RF measurement is done using the Agilent PNA series E8361C Vector Network Analyzer with Cascade DC probes. Measurements are calibrated using short-open-load-through (SOLT) standard to the probe tips. Figure 12.9 presents measured and modeled S-parameters of the single switch with Coff = 19.3 fF, Ron = 3.08 Ω and Lb = 540 pH, respectively. The switch is well matched with return loss >18 dB up to 26 GHz. The measured insertion loss is 0.46–0.39 dB up to 26 GHz with an applied bias of 72–80 V. Isolation of the switch is >18 dB up to 26 GHz. Note that no package is placed on top of the measured switch. The microscopic image of the fabricated SP16T switch is shown in Figure 12.10. The SP16T switch demonstrates 1.47–1.89 dB of insertion loss from 18–26 GHz with Ron = 2.8–3.08 Ω, as shown in Figure 12.11a and b. The device is well matched with >14 dB of return loss up to 26 GHz with Lb = 540 pH and Cj = 3.54 fF, as shown in Figure 12.11a and b. These results are also verified by full-wave simulations. The measured S-parameter responses for all odd ports (P1, P3, P5, P7, P9, P11, P13 and P15) are closely followed with even ports (P2, P4, P6, P8, P10, P12, P14 and P16) responses, as predicted from the simulation response. Moreover, it is also observed that loss and matching are degraded from near port to far port. The maximum measured deviation of return loss between near port

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FIGURE 12.9 Measured versus simulated S-parameter performances of the SPST switch. (Used with permission of IEEE.)

FIGURE 12.10 Microscopic images of the MEMS SP16T switching network. (Used with permission of IEEE.)

(P1 or P16) to far port (P7 or P9) is ~3 dB at 20 GHz and ~3.8 dB at 26 GHz, which successfully validates simulated return loss deviation of ~3.3 dB. Measured loss variation from near to far port is 1.5 dB at 20 GHz and 1.78 dB at 26 GHz with 80 V bias. The isolation of the SP16T switch is >15.8 dB at 20 GHz and >~14 dB at 26 GHz with Coff = 19.3 fF, as depicted in Figure 12.11c. Coff change of 19.3–20.1 fF was observed between all 16 MEMS switches in SP16T configuration due to 0.13–0.163 µm variation in beam tip deflection (upward). Note that isolation is measured when all switches were in the up-state, that is, 1.8–2.8 dB worse than the isolation with one switch in the ON-state condition (in port 8).

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FIGURE 12.11 Measured return loss and insertion loss performances of the SP16T switch at all (a) odd ports and (b) even ports over 18–26 GHz. (c) Measured and simulated isolation characteristics of the SP16T switch. (Used with permission of IEEE.)

12.3.3 Design and Modeling of the 4-Bit Phase Shifter A K-band 4-bit phase shifter is designed using 16 delay lines and 2 back-to-back SP16T switching networks as shown in Figure 12.12. The primary design aim is to achieve low insertion loss and compact size. In the phase shifter, all delay lines are made closer in order to make a compact structure. A 32-port simulation is done in HFSS to check the effect of coupling between 16 lines. Separations between adjacent paths are 50–90 µm, which is optimized enough to keep the coupling below 15 dB at 20 GHz. As a result, simulated average isolation is better than 16 dB at 20 GHz and 12 dB at 26 GHz, as shown in Figure 12.12. Although results show isolation responses with respect to port 1, all variants are checked using full-wave simulation. Coupling is maximum in S211 (15–20 dB more) than other combinations in the lines, as depicted in Figure 12.12. The fabricated microscopic image of the device is shown in Figure 12.13. Figure 12.14 shows the complete schematic model of the 4-bit phase shifter. All 16 CPW lines are placed between two back-to-back SP16T networks and phase shift is obtained by actuating MEMS switches. It substantially reduces the area and makes it more compact in design. The insertion loss of the phase shifter is mainly governed by the metal and switch losses. There are only two switches at each path, in contrast to any other conventional design based on eight SPDT or four SP4T switches, which has eight or four switches in each path.

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FIGURE 12.12 Simulated coupling among the delay lines with respect to port 1. (Used with permission of IEEE.)

FIGURE 12.13 Microscopic images of the fabricated MEMS 4-bit phase shifter using two SP16T switches. Total area is 3.62 × 4.07 mm2 (including bias lines and DC pads). (Used with permission of IEEE.)

In order to obtain optimum device performance, the reference line of the phase shifter is optimized to 481 µm (electrical length: 21° at 20 GHz). All 15 delay lines of the phase shifter contain a section equal to the reference line plus additional delay line as per the desired phase shift. All delay lines are routed with 90° CPW bends at each corner and inductive bends are used in all delay lines to overcome the excitation of any higher-order modes (see Figure 12.13). High impedance (20–30 kΩ) bias lines are also considered in the design. Bias lines are made with 10 µm width TiW and covered with SiO2 (0.7 μm). It can route very easily underneath the t-line and improves the simplicity of the device with negligible effect in signal leakage. Finally, the matching and loss performances of the complete 4-bit phase shifter carried out in HFSS are shown in Figure 12.15. No OFF-path resonance was observed in the band of interest. Average simulated return loss of better than ~16 dB and maximum average

240

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 12.14 Schematic of the complete 4-bit phase shifter where Block A represents the SP16T switching network (see Figure 12.7). Block C was also used on the lines. (Used with permission of IEEE.)

FIGURE 12.15 Measured (a) return loss and (b) insertion loss performances of the 4-bit phase shifter from 18 to 26 GHz. (Used with permission of IEEE.)

insertion loss of ~3.1 dB are obtained from the 18–26 GHz band. The maximum simulated phase error is optimized at ~1.8° at 20 GHz over the 19.75–20.25 GHz (500 MHz) bandwidth. 12.3.4 RF Measurements of the 4-Bit Phase Shifter The S-parameter performance of the 4-bit phase shifter is measured systematically over the band of interest. The total area of the fabricated device is 13.5 mm2 including all bias lines and bias pads. The optical profilometer shows negligible variation (133–165 nm) in beam tip deflection over all 32 MEMS switches used in the phase shifter. It ensures less variation in Coff and Ron on the device. Finally, measured and simulated average insertion and return loss performances of the reported phase shifter is shown in Figure 12.15. The associated average return loss is better than 14 dB and average loss of the device changes from 3 to 4.62 dB over 18–26 GHz (see Figure 12.15). The phase shifter demonstrates an average loss of ~3.47 dB over 19.75–20.25 GHz (500 MHz) (see Figure 12.15). The measured

Phase Shifters Using Two Back-to-Back Switching Networks

241

FIGURE 12.16 Measured (a) phase versus frequency response and (b) group delay performances of the 4-bit phase shifter over 19.75–20.25 GHz. (Used with permission of IEEE.)

loss is more (4.23 dB) at 337.5° phase state as line length is more here compared to the other phase states. Although results show multiple dips in the insertion loss response due to OFF-path resonance, most of them occur between 22 and 26 GHz, which is out of the band of interest. The proposed phase shifter can be readily adapted to a microstrip design to further reduce the line loss. Figure 12.16a shows measured phase versus frequency responses of the 4-bit phase shifter over 18–26 GHz. The average phase error of the device is ~1.77° at 20 GHz. The Rc was maximum (4.1 Ω) in the 292.5° state and minimum (~3 Ω) at the 0° state. The maximum measured differential time delays of ~169.47 ps with the delay step of ~9.06 ps is obtained over 19.75–20.25 GHz (see Figure 12.16b). The results also show no unwanted resonance over the concern band of interest. Measured phase shift/length, phase shift/dB loss and maximum loss/length of the proposed phase shifter are 118°/mm, 78.9°/dB and 1.49 dB/mm, respectively. In addition, the figure of merit (loss/bit) of this 4-bit phase shifter is 0.86 dB. The area/bit of the proposed circuit is 3.72 mm2. Detailed measured phase shift and loss data of the device are tabulated in Table 12.2.

12.4 Conclusion In this chapter, design, fabrication and measured responses of a 3-bit MEMS phase shifter is presented using two back-to-back SP8T switches. The phase shifter demonstrates better than 13 dB of average return loss, average insertion loss of 4.4 dB and phase error of 0.98° over the band of 34.25–35.25 GHz. The phase shifter satisfactorily works >100 M cycles with 0.1–1 W of power. The area of the 3-bit phase shifter is also comparable with other reported MEMS phase shifters and the value is 5.95 mm2. The performance of the reported phase shifter can be significantly improved with a hermetic condition. Later, a 4-bit MEMS phase shifter was presented using two SP16T switches. The SP16T switch is the primary functional block in the proposed phase shifter. Extensive investigations on the design, fabrication and characterizations have been performed to improve the performance of the SP16T switch. The SP16T switch demonstrates average return loss of

Measured Phase error Loss (dB)

Phase State

22.5°

23.18° 0.68°

3.21

0°

0° 0°

~3

3.3

45.57° 0.57°

45°

3.44

68.4° 0.9°

67.5°

3.5

88.63° −1.37°

90°

3.61

111.8° −0.7°

112.5°

~3.68

133.4° −1.6°

135°

Phase Shift and Loss Data of the 4-Bit Phase Shifter at 20 GHz

TABLE 12.2

3.71

155.7° −1.8°

157.5°

180°

3.82

178.89 −1.11° 3.83

200.3° −2.2°

202.5°

225°

3.9

223.7° −1.3°

247.5°

3.94

245.22° −2.28°

270°

3.96

268.8° −2.2°

292.5°

4.1

288.73° −3.77°

315°

4.16

312.08° −2.92°

4.23

334.1° −3.4°

337.5°

242 RF Micromachined Switches, Switching Networks, and Phase Shifters

Phase Shifters Using Two Back-to-Back Switching Networks

243

better than 14 dB and average insertion loss of 1.89 dB over the band of interest. The 4-bit phase shifter demonstrates better than 13 dB of average return loss, average insertion loss of 3.47 dB and average phase error of 1.77° over the band of 19.25–20.25 GHz. The proposed device demonstrates low loss, moderate matching, good phase accuracy and excellent reliability performances over the band. The area of the phase shifter is fairly comparable with the present state-of-the-art MEMS phase shifters.

References

1. B. R. Norvell, R. J. Hancock, J. K. Smith, M. L. Pugh, S. W. Theis, and J. Kviatkofsky, Micro electro mechanical switch (MEMS) technology applied to electronically scanned arrays for spaced based radar, in IEEE Aerospace Conference Proceedings, Aspen, CO, 1999, pp. 239–247. 2. G. M. Rebaiz, RF MEMS Theory, Design, and Technology, John Wiley & Sons, 2003. 3. G.-L. Tan, R. Mihailovich, J. Hacker, J. De Natale, and G. M. Rebeiz, Low-loss 2- and 4-bit TTD MEMS phase shifters based on SP4T switches, IEEE Trans. Microw. Theory Tech., vol. 51, no. 1, pp. 297–304, January 2003. 4. S. Dey, and S. K. Koul, Reliability analysis of Ku-band 5-bit phase shifters using MEMS SP4T and SPDT switches, IEEE Trans. Microw. Theory Tech., vol. 63, no. 12, pp. 3997–4012, December 2015. 5. N. Kingsley, P. Kirby, G. Ponchak, and J. Papapolymerou, 14  GHz MEMS 4-bit phase shifter on silicon, in Topical Meeting on Silicon Monolithic Integrated Circuits in RF Systems, Atlanta, 2004, pp. 326–328. 6. P. Farinelli, H. El Ghannudi, A. Cazzorla, R. Sorrentino, and L. Capponi, A 0-10 GHz SP16T MEMS switch for switched beam satellite antenna systems, in Proceedings of the 44th European Microwave Conference, Rome, Italy, October 6–9, 2014, pp. 195–198.

13 Digital MEMS Phase Shifters Using Combinations of Switched-Line and DMTL Topologies

13.1 Introduction To increase the number of cycle counts for a microelectromechanical systems (MEMS)based digital phase shifter, the number of switches plays a very important role for higher reliability. Rebeiz reported a 4-bit phase shifter where four MEMS switches are actuated at the same time to activate one phase state [1]. A 5-bit phase shifter has been reported where six switches are actuated to activate one phase state [2]. Nonuniform switch actuation over the complete device cycle is the primary reason for the early failure. Different MEMS-based digital phase shifters are reported in the literature utilizing different topologies such as switched line, reflect line, low pass/high pass and distributed MEMS transmission line (DMTL) at microwave to millimeter wave frequencies [3–21]. A DMTL phase shifter is reported with loss of 3.6 dB with 10° step over the 15–40 GHz band in 63.7 mm2 area [18]. A 5-bit switched line phase shifter is reported with 5.4 dB of loss at 17.2 GHz over 36 mm2 area [19]. A 5-bit phase shifter is also reported with 3.76–4.84 dB of loss over 10–25 GHz within 15.6 mm2 area [20]. A switched-line type 5-bit phase shifter is reported with average loss of 2.65 dB within 13 mm2 area at 17 GHz [21]. The work presented here in this chapter greatly improves the overall performance of the phase shifter within a compact size. A new phase shifter topology is proposed here that combines the advantages of DMTL and switched-line topologies. This work broadly focuses on the design, development and characterization of 3-bit and 4-bit phase shifters using two back-to-back single-pole four-throw (SP4T) and single-pole eight-throw (SP8T) switches, respectively. The proposed phase shifter operation is optimized over the frequency band 34.75–35.25 GHz. In addition to the RF performance, the reliability of the reported switches and phase shifters are also tested up to a reasonable extent.

13.2 Proposed Design Topology of the Phase Shifter Single switch cycle count (SW) plays a very crucial role in overall device performance. Reliability for a digital phase shifter can be defined using a simple logic as given in Equation 13.1:

Rc [PS( N −bit )] = 2 N × [SW (C )] (13.1) 245

246

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 13.1 Schematic diagram of the proposed MEMS phase shifter topology. (Used with permission of IEEE.)

where Rc[PS(N – bit)] defines one complete cycle for a digital phase shifter, N is the number of phase shifter bits and C is the number of switch counts per phase state. For example, in a conventional switched-line 4-bit phase shifter, 48 switch actuations over a complete cycle are needed [12], where C plays a very crucial role for better repeatability. Figure 13.1 shows the design schematic of the proposed digital phase shifter. The salient features of the proposed phase shifter topology are as follows: 1. It is a combination of switched-line and DMTL phase shifter topology. A 3-bit (4-bit) phase shifter uses two back-to-back SP4T (SP8T) switches and they are connected with four (eight) tunable delay lines. 2. All four (for SP4T) and eight (for SP8T) delay lines are loaded with MEMS varactors for desired phase state. Note that all varactors work within the pull-in limit, thus are highly reliable without crossing the point of instability. 3. Each connecting line generates two phase states; one from the delay line length and other from the DMTL topology. 4. The proposed design topology requires 16 single-switch actuations over one complete cycle for a 4-bit phase shifter operation. 5. The design drastically reduces the number of switch counts (C) per phase state and thus improves the reliability of the device. 6. The switch and varactor are modeled and designed such that they operate at the same bias voltages. This means the actuation voltage of the DC-contact switch is equal to the varactor control voltage. 7. The proposed design results in substantial size reduction for a 4-bit MEMS phase shifter that operates at 35 GHz. The total area of the 4-bit phase shifter is ~7.5 mm2. Phase shift (Δϕ) of the proposed device is obtained using the following two steps [31]: Step 1: ∆φ =



ω ε r ,eff ( ld −lr ) (13.2) c

Step 2:

∆φ =

sω Z0 ε r ,eff  1 1   Z − Z  (13.3) c lu ld

Phase Shifters Using Switched-Line and DMTL Topologies

247

Here, ld and lr are the lengths of delay lines and reference lines, respectively; ω is the operating frequency; εr,eff is the effective dielectric constant; c is the free space velocity; and Zlu and Zld are loaded and actuated state impedances. These two steps are followed at each connecting line between two switches. Step 1 is used for odd number (0°, 90°, 180° and 270°) and step 2 is used for even number (45°, 135°, 225° and 315°) phase states for 3-bit phase shifter. A 4-bit phase shifter also follows the same logic.

13.3 MEMS Switch Design and Measurements To ensure optimum device performance, all individual SPST, SP4T and SP8T switches are designed, fabricated and tested extensively. All electromechanical and S-parameter performances are checked systematically on the fabricated switches and discussed in the subsequent sections. 13.3.1 Single MEMS Switch Design and Measurements Figure 13.2 shows top and side views of the single MEMS switch, with all structural dimensions marked, that were used to develop SP4T and SP8T switches. More detailed information on the SPST and SP4T switches can be found in Dey et al. [41]. This study is greatly expanded upon with more measurement results and their utility in practical phase shifter operations. A similar spring-mass model of this kind of switch is presented in Dey et al. [41]. To improve the stability of the structure, the proposed switch is designed like a push-up exercise with two hands (k1) in front and legs (k2) in back that are supported by anchors. An equivalent circuit model of the SPST switch and their descriptions are also reported in Dey et al. [41]. Simulation results of the switch show pull-in voltage (Vpi) at 108 V and mechanical resonance frequency ( f0) at 85.6 kHz, as shown in Figure 13.3a and b, respectively. A fabricated image of the switch is shown in Figure 13.4 and line dimensions are marked with the spring model. Measured f0 of the switch was 80 kHz in room air and measured Vpi , Vr voltages are in the range of 82–109 V [41]. The switch voltage decreases by ~6 V after 0.5 W and 11 V after 1 W of RF power up to 85°C due to the RF latch [41]. Measured ON-time of 9.5–12.6 µs with OFF-time of 6.8 µs is obtained at 108–120 V bias (see Figure 13.5a).

FIGURE 13.2 Schematic diagram of top and side view of a single MEMS switch. (Used with permission of IEEE.)

248

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 13.3 Simulated (a) pull-in voltage and (b) mechanical resonance frequency of the switch. (Used with permission of IEEE.)

FIGURE 13.4 Microscopic image of the single MEMS switch. (Used with permission of IEEE.)

FIGURE 13.5 Measured (a) mechanical switching time and (b) S-parameter performances of the SPST switch up to 40 GHz. (Used with permission of IEEE.)

Phase Shifters Using Switched-Line and DMTL Topologies

249

RF measurement is done using the Agilent Vector Network Analyzer (E8361C) using Cascade DC probes and calibrated using short-open-load-thru (SOLT) standard to the probe tips. Figure 13.5b presents measured and modeled S-parameters of the single switch with Coff = ~16 fF, Rc = 1.14 Ω and Lb = 36 pH. The switch is well matched with return loss >28 dB up to 40 GHz. The measured insertion loss is 1.07 dB up to 40 GHz with 112 V. Isolation of the switch is >17.4 dB up to 40 GHz. No package is placed on top of the measured switch. The switch shows measured IIP3 of 49 dBm and it was limited by RF probe contact resistance and CPW transmission lines [41]. 13.3.2 SP4T MEMS Switch Design and Measurements Microscopic image of the SP4T switch is shown in Figure 13.6. Identical switches are used at four outputs to receive the input RF signal. Total area of the switch is 0.98 mm2. SP4T switch descriptions and its equivalent circuit model are reported in Dey et al. [41]. Switch design parameters are optimized using full-wave simulations up to 40 GHz. To improve the impedance matching at the central junction, a junction capacitance (Cj) of 4.3 fF is added on the input line and it is optimized for different values of Cj up to 40 GHz, as shown in Figure 13.7a. Moreover, near port (P2 and P3) and far port (P1 and P4) performances of the SP4T switch are checked and optimized using full-wave simulations. The near port (P2/P3) shows better matching (~3.7 dB) than the far port (P1/P4) responses due to better coupling with input port, as depicted in Figure 13.7b. RF performance of the SP4T switch demonstrates measured average return loss of >17 dB and insertion loss of 17 dB up to 40 GHz and it is measured at one port in the ON-condition (see Figure 13.8b). 13.3.3 SP8T MEMS Switch Design and Measurements Fabricated image of the SP8T switch is shown in Figure 13.9a. Total area of the switch is 1.66 mm2. Equivalent circuit model of the switch is shown in Figure 13.9b. All single

FIGURE 13.6 Microscopic image of the fabricated SP4T switch. (Used with permission of IEEE.)

250

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 13.7 (a) Input/output matching of the SP4T switch for different junction capacitance values. (b) S-parameters of the SP4T switch at near port and far port up to 40 GHz. (Used with permission of IEEE.)

FIGURE 13.8 Measured versus fitted (a) S-parameters. (b) Isolation of the SP4T switch up to 40 GHz. (Used with permission of IEEE.)

switches are actuated separately using an isolated pull-down electrode that makes dccontact with the output line. Figure 13.10a shows 3.48 fF of Cj capacitance gives optimum response in terms of matching in the SP8T switch. Figure 13.10b shows good simulated impedance matching characteristics up to 40 GHz with return loss of >15.4 dB up to 40 GHz. Simulated worst-case insertion loss is 1 B cycles with maximum Rc variation from 2.4 to 6.3 Ω. Performance of the SP8T switch degrades with an abrupt change in Rc from 5.8 to 6.3 Ω after 600 M cycles. It was mostly due to (a) contact point degradations due to additional attractive force from RF power at high temperature, (b)

Phase Shifters Using Switched-Line and DMTL Topologies

255

different thermal coefficients of expansion between the substrate and gold beam [10] and (c) a charge injection problem after a few cycles when the beam curvature decreases with time. From the analysis, it is evident that beam Vpi is reduced with rising of contact temperature, and it increases the Vc periodically. It leads to the softening at higher incident RF power level and at higher temperature. All these issues could be improved with (a) doping the dielectric materials [27] or having no dielectric layer structures [28], (b) phase change material [29] and (c) well-suited contact materials [30].

13.5 Phase Shifters Design and Measurements The 3-bit and 4-bit phase shifters are designed, fabricated and tested systematically. An extensive list of measurements is adopted on the devices including RF, power handling and temperature stability, and discussed in the subsequent sections. 13.5.1 Design and Simulation of 3-Bit and 4-Bit Phase Shifters Fabricated images of phase shifters are shown in Figure 13.15. The conventional switchedline phase shifter uses two SP4T switches for the 2-bit, but the present topology provides 3-bit operation using the same number of switches. It leads to a substantial reduction in the overall area of the device. The total area of the 3-bit and 4-bit phase shifters is 4.3 mm2 and 7.5 mm2, respectively. The insertion loss of the phase shifter is mainly governed by the metal and switch losses. Matching of the structure is mostly dominated by the design of the DMTL. The buckle-type beam is used as a varactor for DMTL. The design of the beam is inspired by Mahameed and Rebeiz [37] and this beam design is less prone to temperature and stress. The stiffness

FIGURE 13.15 Microscopic images of the (a) 3-bit and (b) 4-bit phase shifters. Total area of the 3-bit and 4-bit phase shifter is 4.3 mm2 and 7.5 mm2, respectively. (Used with permission of IEEE.)

256

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 13.16 Simulated (a) in-plane displacement versus in-plane stress and (b) vertical displacement with beam distance with σ = 100 MPa. (Used with permission of IEEE.)

of the beam (kz) is a combination of structural stiffness (ks) and stiffness due to the biaxial residual stress (kb). For this clamped–clamped structure, temperature change will effect kb and thus the pull-in voltage. The kz (kz = ks + kb) is a function of the Young’s modulus of gold (45 GPa) and structure geometry. The pull-in voltage of the beam (Vpb) is defined by

Vpb =

8 ( k s + kb ) g 2 (13.4) 27 ( Cu + C f )

where Cu is the up-state capacitance of the beam and it is defined by ε0A/g and Cf is the fringing capacitance between two plates. In the present work, a thicker bottom electrode thickness (2 µm) is used that increases the vertical deflection of the beam due to bending moments associated at the step. It also creates nonplanarized topography underneath the beam plane. The angle (φ) of the tilted spring is optimized to be 45° with 3.5 µm thickness and 10 µm of spring width. It substantially reduces this deflection due to in-plane displacement effects (δσx) from tensile stress in –x direction, and this is irrelevant from an RF MEMS varactor perspective. COMSOL simulation shows tensile φ = 45° gives higher stiffness compared to other φ (Figure 13.16a). Simulation also shows vertical deflection of 13.7 dB up to 40 GHz, as shown in Figure 13.18. The results here show isolation responses with respect to port 1, but all variants are checked using full-wave simulations. Isolation is poor in the 4-bit phase shifter because all delay lines are closely packed parallel to one another as compared to the 3-bit. Unit cells of the phase shifters are designed for 45° (3-bit) and 22.5° (4-bit) phase bits using two beams. The Bragg frequency ( fB) of the circuit is

Phase Shifters Using Switched-Line and DMTL Topologies

257

FIGURE 13.17 FEM simulation of bridge shows pull-in at 114 V. (Used with permission of IEEE.)

FIGURE 13.18 Simulated coupling among the delay lines with respect to the port 1 in (a) 3-bit and (b) 4-bit phase shifters over a 500 MHz band. (Used with permission of IEEE.)

258

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 13.19 Measured S-parameter responses of the (a) 3-bit and (b) 4-bit phase shifters. (Used with permission of IEEE.)

optimized at 87 GHz and 80 GHz for 3-bit and 4-bit, respectively. Finally, the phase shifters demonstrate average simulated return loss of >13 dB and average loss of 25 dB and insertion loss of 14 dB, insertion loss of 12 dB, insertion loss 1 B cycles (test stopped at 1 B) over 0.1–1 W of power at 25°C. Maximum average loss and phase error variations are 5.28–9.12 dB and 1.67°–5.6°, respectively, at 35 GHz. Furthermore, the reliability of the MEMS bridges of each DMTL is checked under the same power and bias voltage (112 V) (Figure 13.22b). Results show maximum Cup variations of 62.1–64.88 fF up to 1 B cycle with 0.1–1 W power at 25°C. 13.5.4 Phase Shifter Testing at 50°C–85°C with 0.1–0.5 W of Power Next, phase shifter reliability is measured at two different temperatures (50°C and 85°C) and with 0.1 and 0.5 W of power. The phase shifter performed well (>400 M cycles) at 0.5 W power over 50°C–85°C, as depicted in Figure 13.23a. Maximum average loss and phase error variation during this process are ~9.72 dB and 6.35°, respectively (Figure 13.23b). Performance degraded fast at 1 W over 50°C–85°C and devices sustained up to >10 M cycles at 85°C. Performance degradation at higher temperatures is limited due to thermal

260

RF Micromachined Switches, Switching Networks, and Phase Shifters

TABLE 13.2 Phase Error and Loss Data of the 3-Bit Phase Shifter at 35 GHz Phase State

0º

45º

90º

135º

180º

225º

270º

315º

Phase Error Loss (dB)

0º 3.16

0.38º 3.37

1.1º 3.57

0.58º 3.7

~ 1º 3.88

1.7º 4.1

1.58º 4.38

2.3º 4.7

conductivity reduction of the gold beam (317 W/mK) over the conduction heat transfer process. As a matter of fact, DMTL (step 2) worked perfectly well here because all bridges actuated within the point of stability.

13.6 Device Responses within Low-Cost Packaging To observe device performances for end users, 3-bit and 4-bit phase shifters are diced as a chip and mounted onto a carrier, as depicted in Figure 13.24. Phase shifters are encapsulated within a module of gold-coated brass material. Full-wave simulations confirm average loss and phase error tolerances of 15%–25% from bare die to package due to added parasitic. Initially, both devices are actuated with the same bias and they worked well up to ~1 B cycles with 0.1 W of power at 25°C. Average loss and phase error variations are 1.3 dB and 2.28°, respectively, from bare die to its package form. At 50°C–85°C, phase shifter performances are degraded to 3.8 dB and 4.7°, respectively, between 0.1 and 0.5 W power compared to its bare die results, as depicted in Figures 13.25. Added bond-wire parasitic and cable losses are the primary reasons for this deviation. The total size of the 4-bit device within a package is 1.7 × 1.5 cm2.

13.7 Design Guidelines of the Proposed Device The primary design goal of the proposed device is highly reliable with a long life cycle and compactness without compromising the RF performances. Key notes of the proposed structure are:

1. Switching network—Proposed phase shifters are designed combining DMTL with switched-line techniques. DMTL is normally suitable for high frequency (>K-band) due to its distributed capacitive manner, while switched lines work much better at lower frequencies. Hence, the designing of the switching network (SP4T and SP8T) in terms of loss and matching at high frequency plays a very important role. More attention should be given to (a) junction capacitance, (b) contact resistance and (c) radius of the central junction. A circular-type switch is useful here and it permits switches to be placed close together without any fabrication difficulties. 2. Robust switch—To perform two steps systematically, two back-to-back switches are biased at the ON-state for a longer time here in contrast to any other switched line-based topology. For this, the switch has to be robust and a 3.5–4 µm thicker

22.5º

0.38º 4.49

0º

0º 4.32

Phase State

Phase Error Loss (dB)

0.87º 4.58

45º 1.8º 4.77

67.5º 1.07º 4.86

90º 1.7º ~5.08

112.5º

Phase Error and Loss Data of the 4-Bit Phase Shifter at 35 GHz

TABLE 13.3

0.9º 5.14

135º 2.43 5.348

157.5º 1.3º 5.47

180º 1.8º 5.62

202.5º 1.6º 5.78

225º

1.78º 5.89

247.5º

2.1º 6.02

270º

2.27º 6.38

292.5º

2.7º 6.27

315º

3.8º ~6.6

337.5º

Phase Shifters Using Switched-Line and DMTL Topologies 261

262

RF Micromachined Switches, Switching Networks, and Phase Shifters

FIGURE 13.22 Measured cold- and hot-switching reliability of the (a) 3-bit and 4-bit phase shifters. (b) MEMS bridge. (Used with permission of IEEE.)

FIGURE 13.23 Measured phase shifter responses at 50°C and 85°C with (a) cold (0.1 W) and (b) hot (0.5 W) switching conditions. (Used with permission of IEEE.)

FIGURE 13.24 Phase shifter within a package. (Used with permission of IEEE.)

Phase Shifters Using Switched-Line and DMTL Topologies

263

FIGURE 13.25 Measured phase shifter responses at 50°C and 85°C with (a) 0.1 W and (b) 0.5 W of incident RF powers. (Used with permission of IEEE.)

FIGURE 13.26 Measured creep results of (a) 3-bit and (b) 4-bit phase shifters. (Used with permission of IEEE.)

beam is used and two additional arms provide extra robustness to the structure. Switch actuation voltage changes over time and shows ~9 V variations (mostly on Vr) over 10 h and it is mostly due to stiction and mechanical contact deformation. Finally, phase shifter performances are tested under the prolonged ON-state condition, and loss and phase error variations are recorded up to 6 h, as shown in Figure 13.26. Although responses are shown at three selected phase states, all states are tested with respect to the 0°-state at 35 GHz. Loss variation of ~1.37 dB (4.23–5.6 dB) and phase error variations of ~1.26° (3.77°–5.03°) are observed at 35 GHz during the process, as shown in Figure 13.26. 3. Varactor design—One of the design aims is to achieve the same actuation voltage between the switch and varactor. It means varactors will produce distributed capacitance per phase state with the same bias voltage as in the switch. It also improves the bias network simplicity. This process needs multiple iterations in terms of design and process yield.

264

RF Micromachined Switches, Switching Networks, and Phase Shifters

TABLE 13.4 Performance Comparison of State-of-the-Art MEMS Phase Shifters Type

Frequency (GHz)

Phase Shift (º)

Resolution (º)

Avg. IL* (dB)

Avg. RL* (dB)

Avg. Phase Error (º)

Size (mm 2)

MEMS, 2003, [34] MEMS, 2006, [13] MEMS, 2008, [15] DMTL, 2012, [16] DMTL, 2013, [17] MEMS, 2013, [18] DMTL, 2014, [19] DMTL, 2015, [20] MEMS, 2015, [21] This work: 3-bit This work: 4-bit

35 10 10 35 15–25 17.25 10 24.75–25.25 16.75–17.25 35 35

0–360 0–360 0–360 0–360 0–180 0–360 0–360 0–360 0–360 0–315 0–337.5

45º 11.25 11.25 22.5 10 11.25 11.25 11.25 11.25 45 22.5

2.2 3.6 4.5 2.7 3.6 5.4 4.72 4.84 2.65 3.85 5.4

1 B cycle at 0.5 W

This work 1 B cycles at 1W This work >1 B cycles at 1W

This work >1 B cycles with 1W

1 B cycles with 0.1 W of power. Nevertheless, the reported phase shifters worked for ~400 M cycles with 0.5 W of power at 85°C. Device performances were also checked with low-cost packaging and the results are presented in this chapter. Total area of the 4-bit phase shifter is 7.5 mm2 and it is the smallest reported phase shifter to date operating over the Ka-band (at 35 GHz). To the best of the authors’ knowledge, these are the first reported 3-bit and 4-bit MEMS phase shifters in the literature that have undergone extensive measurement stages with a minimum number of switching elements per phase state.

References

1. B. R. Norvell, R. J. Hancock, J. K. Smith, M. L. Pugh, S. W. Theis, and J. Kviatkofsky, Micro electro mechanical switch (MEMS) technology applied to electronically scanned arrays for spaced based radar, in Proceedings Aerospace Conference, Aspen, CO, 1999, pp. 239–247. 2. Shiban K. Koul, and B. Bhat, Microwave and Millimeter Wave Phase Shifter, Vol. 2, Artech House, 1991. 3. D. Parker, and D. Zimmermann, Phased arrays—Part I: Theory and architectures, IEEE Trans. Microw. Theory Tech., vol. 50, no. 3, pp. 678–687, March 2002. 4. D. W. Kang, H. D. Lee, C. H. Kim, and S. Hong, Ku-band MMIC phase shifter using a parallel resonator with 0.18-µm CMOS technology, IEEE Trans. Microw. Theory Tech., vol. 54, no. 1, pp. 294–301, January 2006. 5. K. Kwang-Jin, and G. M. Rebeiz, 0.13-µm CMOS phase shifters for X-, Ku-, and K-band phased arrays, IEEE J. Solid-State Circuits, vol. 42, no. 11, pp. 2535–2546, November 2007. 6. B. Min, and G. M. Rebeiz, Single-ended and differential-band BiCMOS phased array frontends, IEEE J. Solid-State Circuits, vol. 43, no. 10, pp. 2239–2250, October 2008. 7. K.-J. Koh, and G. M. Rebeiz, A 6-18 GHz 5-bit active phase shifter, in IEEE MTT-S International Microwave Symposium Digest, Anaheim, CA, May 2010, pp. 792–795. 8. J. Y. Choi, M.-K. Cho, D. Baek, and J.-G. Kim, A 5-20 GHz 5-bit true time delay circuit in 0.18 μm CMOS technology, J. Semicond. Tech. Science, vol. 13, no. 3, pp. 193–197, June 2013. 9. C. Liang, C. Xinyu, Z. Youtao, L. Zhiqun, and Y. Lei, A high linearity X-band SOI CMOS digitally-controlled phase shifter, J. Semicond., vol. 36, no. 6, pp. 1–8, June 2015. 10. S. Lucyszyn, Advanced RF MEMS, Cambridge University Press, August 2010.

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11. A. Q. Liu, RF MEMS Switches and Integrated Switching Circuits, Springer, March 2010. 12. G.-L. Tan, R. Mihailovich, J. Hacker, J. DeNatale, and G. M. Rebeiz, Low-loss 2- and 4-bit TTD MEMS phase shifters based on SP4T switches, IEEE Trans. Microw. Theory Tech. vol. 51, no. 1, pp. 297–304, January 2003. 13. Z. Jian, Y.-Y. Weil, C. Chen, Z. Yong, and L. Le, A compact 5-bit switched-line digital MEMS phase shifter, in IEEE International Conference on Nano/Micro Engineered and Molecular Systems, Zhuhai, China, January 2006, pp. 623–626. 14. C. D. Nordquist, C. W. Dyck, G. M. Kraus, C. T. Sullivan, F. Austin, P. S. Finnegan, and M. H. Ballance, Ku-band six-bit RF MEMS time delay network, in 2008 IEEE Compound Semiconductor Integrated Circuits Symposium, Monterey, CA, October 2008. 15. M. A. Mortonand, and J. Papapolymerou, A packaged MEMS-based 5-bit X-band high-pass/ low-pass phase shifter, IEEE Trans. Microw. Theory Tech., vol. 56, no. 9, pp. 2025–2031, August 2008. 16. B. Pillans, L. Coryell, A. Malczewski, C. Moody, F. Morris, and A. Brown, Advances in RF MEMS phase shifters from 15 GHz to 35 GHz, in IEEE MTT-S International Microwave Symposium Digest, Montreal, June 2012, pp. 1–3. 17. M. Unlu, S. Demir, and T. Akin, A 15–40-GHz frequency reconfigurable RF MEMS phase shifter, IEEE Trans. Microw. Theory Tech., vol. 61, no. 8, pp. 2397–2402, August 2013. 18. S. Dey, and Shiban K. Koul, Design and development of a CPW-based 5-bit switched-line phase shifter using inline metal contact MEMS series switches for 17.25 GHz transmit/receive module application, J. Micromech. Microeng., vol. 24, no. 1, 015005, November 2013. 19. S. Dey, and Shiban K. Koul, Design, development and characterization of an X-band 5 bit DMTL phase shifter using an inline MEMS bridge and MAM capacitors, J. Micromech. Microeng., vol. 24, no. 1, 095007, June 2014. 20. S. Dey, and Shiban K. Koul, 10–25 GHz frequency reconfigurable MEMS 5-bit phase shifter using push–pull actuator based toggle mechanism, J. Micromech. Microeng., vol. 25, no. 6, 065011, May 2015. 21. S. Dey, and Shiban K. Koul, Reliability analysis of Ku-band 5-bit phase shifters using MEMS SP4T and SPDT switches, IEEE Trans. Microw. Theory Tech., vol. 63, no. 12, pp. 3997–4012, December 2015. 22. S. Gong, H. Shen, and N. S. Barker, A 60-GHz 2-bit switched-line phase shifter using SP4T RF-MEMS switches, IEEE Trans. Microw. Theory Tech., vol. 59, no. 4, pp. 894–900, April 2011. 23. H. Zareie, and G. M. Rebeiz, Compact high-power SPST and SP4T RF MEMS metal-contact switches, IEEE Trans. Microw. Theory Tech., vol. 61, no. 8, pp. 2397–2402, January 2014. 24. A. Q. Liu, W. Palei, M. Tang, and A. Alphones, Single-pole-four-throw switch using highaspect-ratio lateral switches, Electron. Lett., vol. 40, no. 18, pp. 1281–1282, September 2008. 25. S. Kang, H. Cheol Kim, and K. Chun, A single-pole, eight-throw, radio-frequency, microelectromechanical systems switch for multi-band/multi-mode front-end module, J. Sens. Sci. Technol., vol. 20, no. 2, pp. 78–81, 2011. 26. S. Pranonsatit, A. S. Holmes, and S. Lucyszyn, Microwave modelling of radio frequency microelectromechanical rotary switches, IET Microw. Antennas Propag., vol. 5, no. 3, pp. 255–261, March 2011. 27. G. Li, H. S. San, and X. Y. Chen, Charging and discharging in ion implanted dielectric films used for capacitive radio frequency microelectromechanical systems switch, J. Appl. Phys., vol. 105, no. 12, pp. 124503-1–124503-6, June 2009. 28. D. Mardivirin, A. Pothier, A. Crunteanu, B. Vialle, and P. Blondy, Charging in dielectricless capacitive RF-MEMS switches, IEEE Trans. Microw. Theory Tech., vol. 57, no. 1, pp. 231–236, January 2009. 29. M. Wang, Y. Shim, and M. Rais-Zadeh, A low-loss directly heated two-port RF phase change switch, IEEE Electron. Device Lett., vol. 35, no. 4, pp. 491–493, April 2014. 30. F. Ke, J. Miao, and J. Oberhammer, A ruthenium-based multimetal-contact RF MEMS switch with a corrugated diaphragm, J. Microelectromech. Syst., vol. 17, no. 6, pp. 1447–1459, December 2008.

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31. G. M. Rebeiz, RF MEMS Theory, Design, and Technology, John Wiley & Sons, 2003. 32. S. Dey, Shiban K. Koul, A. K. Poddar, and U. L. Rohde, Extensive performance evaluations of RFMEMS single-pole-multi-throw (SP3T to SP14T) switches up to X-band frequency, J. Micromech. Microeng., vol. 27, no. 1, 10 pages, November 2016. 33. Shiban K. Koul, S. Dey A. K. Poddar, and U. L. Rohde, Ka-band reliable and compact 3-bit true-time-delay phase shifter using MEMS single-pole-eight-throw switching networks, J. Micromech. Microeng., vol. 26, no. 10, 9 pages, August 2016. 34. J. B. Hacker, R. E. Mihailovich, M. Kim, and J. F. DeNatale, A Ka-band 3-bit RF MEMS truetime-delay network, IEEE Trans. Microw. Theory Tech., vol. 51, no. 1, pp. 305–308, January 2003. 35. C. D. Patel, and G. M. Rebeiz, A high-reliability high-linearity high-power RF MEMS metalcontact switch for DC-40-GHz applications, IEEE Trans. Microw. Theory Tech., vol. 60, no. 10, pp. 3096–3112, October 2012. 36. R. Stefanini, M. Chatras, P. Blondy, and G. M. Rebeiz, Miniature MEMS switches for RF applications, J. Microelectromech. Syst., vol. 20, no. 6, pp. 1324–1335, December 2013. 37. R. Mahameed, and G. M. Rebeiz, A high-power temperature stable electrostatic RF MEMS capacitive switch based on thermal buckle-beam design, J. Microelectromech. Syst., vol. 19, no. 4, pp. 816–826, August 2010. 38. www.r​adant​mems.​com/r​adant​mems.​data/​Libra​ry/Ra​dant-​Datas​heet2​21_2.​0.pdf​. 39. www.a​stram​t l.co​m/pdf​/data​%20sh​e ets/​Advan​ce%20​I nfor​m atio​n/Swi​tches​/DC_2​0 GHz_​ MEMS_​SP4T_​Modul​e.pdf​. 40. www.r​adant​mems.​com/r​adant​mems.​data/​libra​ry/ra​dant-​datas​heet2​40_1.​1.pdf​. 41. S. Dey, Shiban K. Koul, A. K. Poddar, and U. L. Rohde, Ku-V-band 4-bit MEMS phase shifters using high isolation SP4T switches and DMTL structures, J. Micromech. Microeng., vol. 27, no. 10, pp. 10, September 2017.

14 Packaging and Integration Technologies

14.1 Introduction Radio frequency microelectromechanical systems (RF MEMS) device packaging at the wafer level is one of the critical areas for its application, as the MEMS elements are delicate and can easily get damaged during wafer scribing or the packaging process. In addition, a packaging engineer always tries to reduce the parasitic to a large extent so that overall device performance is not drastically affected. In the packaging stage, almost all RF MEMS devices (e.g., switches, varactors, and inductors) are housed in an inert ambient (preferably dry nitrogen). Hermetic or near-hermetic packaging ensures the best case stability performance in MEMS devices without being affected by moisture. In general, 1-level and 0-level packaging are widely used by the MEMS industry. The 1-level ceramic packaging is done using a chip capsule followed by interconnecting leads for outside connection [1], whereas 0-level packaging is done on-wafer in device scale with batch fabrication. At the beginning of this chapter, different packaging stages are briefly discussed. Then, a novel approach for wafer-level encapsulation of GaAs-based RF MEMS switches is discussed in detail.

14.2 1-Level Packaging The 1-level ceramic packaging is made with low temperature cofired ceramic (LTCC) in most cases. A chip capsule or interconnecting leads are used to make 1-level packaging. In LTCC 1-level packaging, encapsulation is done by soldering a ceramic cap with a metal sealing ring on the substrate to make the cavity [1–4]. This cavity formation allows flexibility to the sealing gas composition. The main drawback of this packaging level is that ceramic packages are expensive, hence it is difficult to implement in high volume applications like as in mobile phone handset applications. It is capable of working in satellite, defense and telecommunication base station applications. Note that other complications of 1-level packaging come after the release of the MEMS structure. For example, the injection molding process of plastic packages sometimes contaminates the MEMS device. It adversely affects the switch behavior with early failure or with a higher value of switchcontact resistance (for ohmic-contact switches). Hence, cleaning is extremely important in the packaging stage to improve the device performance for the long run. Plastic molded packaging is applicable for frequencies below a few gigahertz (~10 GHz). The packaging engineer prefers to package the die during wafer processing prior to dicing. For this 269

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reason, a new class of packaging level is widely preferred and it is referred to as wafer-, zero- or 0-level packaging. The most modern commercially available MEMS devices use 0-level packaging [5–10].

14.3 0-Level Packaging To implement the 0-level packaging, two approaches are widely used and they are (1) thin-film encapsulations and (2) chip/die capping. The 0-level packaging demonstrates few advantages to a MEMS device, including (a) low-cost fabrication process with physical protection, (b) low loss or less RF signal feed through, and (c) hermetic-preventing moisture. Note that after 0-level packaging, the wafer can be diced without affecting the MEMS device. The height of the MEMS device after packaging is also important for application in the mobile handset. Later, individual chips can be mounted using wire bonding or flip-chip solder bumping into a low-cost plastic molded 1-level package. For example, a small-outline integrated circuit, 8-lead (SOIC-8) [6] or ball grid array (BGA) [11] is used. Zero-level packaged MEMS devices can be joined or interfaced easily in a printed circuit board (PCB) as a chip-scale package (CSP). Readers may note that multilayer LTCC or high temperature cofired ceramic (HTCC) technology with low loss flip-chip assembly for short interconnects exhibit potential application for MEMS devices up to millimeter wave frequencies [12]. A brief overview of thinfilm and chip-capping packaging is given in the subsequent sections. 14.3.1 Thin-Film Packaging The thin-film encapsulation demonstrates much low-profile wafer-level integrated-circuit compatibility. In this packaging level, sacrificial material is deposited over the unreleased MEMS structure followed by deposition of a dielectric shell. The sacrificial material is removed via channels that are subsequently sealed. Phosphosilicate glass is used as a sacrificial layer for a polysilicon resonator and it is deposited using low-pressure chemical vapor deposition (LPCVD) silicon nitride at 800°C [13]. But unfortunately, this process cannot be adopted for RF MEMS switches because this temperature range can cause undesirable switch deformation and actuation failures [13]. Hence, plasma-enhanced chemical vapor deposition (PECVD) and sputter deposition are alternative techniques in RF MEMS switches [14]. Note that the sealing quality has to be good enough in this packaging level for optimum device performance. PECVD silicon nitride needs sealing temperature at 30°C, whereas LPCVD-based polysilicon and silicon nitride used to seal empty cavities need much higher temperatures (>600°C) to achieve the same sealing quality [15]. Encapsulation of MEMS switches is also reported using a dielectric shell and plasma etch removal of sacrificial photoresist [8]. The major drawback of CVD techniques is that the cavity pressure and gas species are dictated by the deposition conditions. Kevin D. Leedy et al. reported a thin-film four-mask packaging process with three additional steps for the encapsulation [16]. Figure 14.1a shows a schematic cross section of a sealed switch, and Figure 14.1b shows the scanning electron microscope (SEM) image of the capacitive RF MEMS switch. In this process, a 2–3 µm sacrificial photoresist layer is deposited. The shell sacrificial layer is formed over the switch using 3 µm of sacrificial photoresist (Figure. 14.1a). A 1.67 µm silicon nitride dielectric shell layer is then deposited as dielectric shell-6

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FIGURE 14.1 (a) Schematic cross section of a packaged switch and (b) SEM image of a capacitive RF MEMS switch. (Used with permission of IEEE.)

FIGURE 14.2 Measured isolation and insertion loss of switches (a) without a shell and within an unsealed shell, and (b) within an unsealed shell and a sealed shell. (Used with permission of IEEE.)

using low temperature (