Printed MIMO Antenna Engineering [1 ed.] 9781608076826, 9781608076819

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Printed MIMO Antenna Engineering

For a complete listing of titles in the Artech House Antennas and Propagation Library, turn to the back of this book.

Printed MIMO Antenna Engineering Mohammad S. Sharawi

Library of Congress Cataloging-in-Publication Data A catalog record for this book is available from the U.S. Library of Congress. British Library Cataloguing in Publication Data A catalogue record for this book is available from the British Library. Cover design by Igor Valdman

ISBN 13: 978-1-60807-681-9

© 2014 ARTECH HOUSE 685 Canton Street Norwood, MA 02062

All rights reserved. Printed and bound in the United States of America. No part of this book may be reproduced or utilized in any form or by any means, electronic or mechanical, including photocopying, recording, or by any information storage and retrieval system, without permission in writing from the publisher.   All terms mentioned in this book that are known to be trademarks or service marks have been appropriately capitalized. Artech House cannot attest to the accuracy of this information. Use of a term in this book should not be regarded as affecting the validity of any trademark or service mark.

10 9 8 7 6 5 4 3 2 1

To my lovely wife Rana and two adorable little angels, Basima and Rand

Contents

Preface

xi



Acknowledgments

xv

1

Introduction

1

1.1

Wireless Technology Evolution

1

1.2

Multiple-Input-Multiple-Output (MIMO) Technology

5

1.3

Market Forecasts

8

1.4

Conclusions

8

References

9

2

Antenna Fundamentals

11

2.1

Radiation Mechanism

11

2.2 2.2.1 2.2.2 2.2.3 2.2.4 2.2.5 2.2.6

Single Antenna Parameters Resonance Radiation Patterns Directivity Efficiency Gain Polarization

15 15 18 18 19 21 21

vii

viii

Printed MIMO Antenna Engineering

2.3 2.3.1 2.3.2 2.3.3 2.3.4 2.3.5 2.3.6 2.3.7

MIMO Antenna Parameters Total Active Reflection Coefficient Isolation Correlation Coefficient Mean Effective Gain Diversity Gain Branch Power Ratio System Capacity

23 23 25 25 27 30 32 32

2.4 2.4.1 2.4.2 2.4.3 2.4.4 2.4.5 2.4.6

Printed Antenna Types Printed Dipole Antennas Printed Monopole Antennas Printed Loop Antennas Printed Patch Antennas Printed Inverted-F Antenna Other Printed Antenna Structures

36 37 42 42 46 53 57

2.5 2.5.1 2.5.2 2.5.3

Antenna Integration Effects Effect of PCB Ground Plane Effect of Mobile Terminal Casing Effect of the Presence of User’s Hand and Head

58 58 60 60

2.6

Antenna Arrays

63

2.7

Conclusions

69

References

69

3

Electrically Small Printed Antennas

77

3.1 3.1.1 3.1.2 3.1.3 3.1.4

Features of Electrically Small Antennas The Quality Factor (Q) Maximum Gain (Gmax) Efficiency Radiation Patterns

77 78 82 84 85

3.2 3.2.1 3.2.2 3.2.3

ESA Examples Meander Line ESAs Loop and Spiral ESAs Other Printed Geometries for ESAs

85 85 89 91

3.3

Conclusions

96

References

100



Contents

ix

4

Printed Single-Band MIMO Antenna Systems

103

4.1

MIMO Antennas for Access Points and General Applications

103

4.2

MIMO Antennas for Cellular and Smartphones

111

4.3

MIMO Antennas for Large PCs and Tablets

122

4.4

MIMO Antennas for USB Dongles

134

4.5

Conclusions

140

References

144

5

Multiband Printed MIMO Antenna Systems

151

5.1

Printed Multiband MIMO Antenna Systems for Wireless Access Points and Generic Applications

152

5.2

Printed Multiband MIMO Antennas for Mobile Phones

163

5.3

Multiband Printed MIMO Antennas for Portable Computers

185

5.4

Multiband MIMO Antennas for USB Dongle Applications

191

5.5

Design Guidelines for Multiband Printed MIMO Antennas

196

5.6

Conclusions

205

References

207

6

Isolation Enhancement Techniques for Printed MIMO Antenna Systems

215

6.1

Antenna Placement and Orientation

216

6.2

Decoupling Networks

224

6.3

Parasitic Elements

234

6.4

Defected Ground Structures

243

6.5

Neutralization Lines

253

x

Printed MIMO Antenna Engineering

6.6

Metamaterial-Based Isolation Enhancement Structures 259

6.7

Conclusions References

266 270

7

MIMO Antenna Performance Measurements

273

7.1 7.1.1 7.1.2

Conventional Antenna Measurement Methods S-Parameter Measurements Antenna Radiation Pattern Measurements

273 274 274

7.2 7.2.1 7.2.2 7.2.3

MIMO Antenna Performance Measurement Technique The Anechoic Chamber Multiprobe OTA Method The Two-Stage OTA Method Reverberation Chamber OTA Method

277 278 279 281

7.3

Remarks on MIMO OTA Testing

283

7.4

Conclusions

285

References

285



List of Abbreviations

287



About the Author

291



Index

293

Preface Wireless communications have made a huge leap during the past two decades. The mobile phone that was invented in the 1970s was bulky, expensive, had a very short battery life, and was meant only for voice communications. Nowadays, your mobile phone is like a small computer. You can pretty much do everything with it except for high-end simulations and data analysis. But you can be always connected to the Internet, with full connectivity to your office and home. The list of technological advancements that affected this revolution is long. The need for higher data rates and user multimedia experiences has pushed the limits on available bandwidth and power transmissions. In a single antenna system, the amount of data that can be transmitted given a certain frequency bandwidth and power levels is limited because both of these parameters are usually restricted. The frequency bandwidth is limited to certain bands assigned by regulatory agencies around the world, and the power levels are also restricted to avoid interfering with other applications and wireless standards. Thus a new technology was required to overcome these limitations and provide higher data limits. The multiple-input-multiple-output (MIMO) technology was proposed in the 1990s as a viable solution that can overcome the data rate limit experienced by single-input-single-output (SISO) systems (i.e., single antenna systems). MIMO requires multiple antenna elements at the transmitter and at the receiver to achieve a linear increase in the data rates with an increase in the number of antennas. This has led to the current high data rates provided by the fourth generation of wireless standards. These MIMO antenna systems are an essential part of current wireless terminals, and there design procedures and performance metrics go beyond those of the single antenna systems that were used in third-generation mobile terminals and prior systems. The small form xi

xii

Printed MIMO Antenna Engineering

factor of current mobile terminals such as cellular phones and USB dongles poses a new challenge for antenna designers due to the space restrictions of such devices, and thus special techniques need to be devised to maintain an acceptable amount of coupling between closely spaced antennas. The focus of this book is on printed MIMO antenna system design. Printed antennas are widely used in mobile and handheld terminals due to their conformity with the device, low cost, good integration within the device elements and mechanical parts, and ease of fabrication. The book starts by reviewing generic antenna performance metrics and then introduces the metrics required for evaluating the performance of MIMO antenna systems. Chapter 2 focuses on MIMO antenna performance metrics and provides a detailed list of available printed antenna geometries that can be used as a starting point for a printed MIMO antenna design. The features of each of these widely used printed geometries are provided as well as some miniaturization techniques. Chapter 3 covers the basic theory of electrically small antennas (ESAs). Due to the excessive miniaturization that is required as a result of the limited space in some mobile terminals, the antenna must become electrically small, and thus care should be taken when assessing its performance and checking its efficiency and operating bandwidth. The maximum gain and bandwidth that can be achieved for a specific ESA is also limited based on the amount of volume the ESA occupies within the radian sphere. ESAs are designed for various applications, and if utilized properly, they can be integrated within mobile terminals. Several examples from literature as well as design examples are given in this chapter to provide the student/researcher/designer with a starting point for modeling and checking the performance of such antennas. Chapters 4 and 5 focus on single-band and multiple-band printed MIMO antenna systems and their design. The chapters classify antennas by their application size as well as their types. The application size is considered based on the final product size or system ground plane size. Four categories are given in this classification: (1) MIMO antennas for generic applications and wireless access points with medium-size ground planes, (2) printed MIMO antennas for cellular phones, (3) printed MIMO antennas for large mobile terminals such as laptops and tablets, and (4) MIMO antennas for USB dongles. Within each category, several antenna types are given as examples for the reader to check and consider as a starting point for his or her design, followed by the MIMO performance metrics that characterize the system behavior. A procedure is devised to provide the designer with some guidelines when starting to design a printed MIMO antenna at the end of Chapter 5. These two chapters include most of the references that have appeared in literature in past 5 years or so. Thus they also provide a comprehensive literature review in this area. The placement of several antennas in close vicinity within the user mobile terminal increases the amount of coupling and lowers the radiation efficiency



Preface

xiii

of the antenna system if not designed properly. Chapter 6 discusses in detail the various methods that have appeared in the literature to enhance the isolation between adjacent printed MIMO antennas. The chapter provides several design options based on the application and type of antennas considered in the design. All options can be integrated within the same printed backplane of the antennas. The operating principles for all methods are presented in detail in addition to their advantages and disadvantages. This chapter also presents the latest work and methods in the area of isolation enhancement of printed MIMO antenna systems. The characterization of MIMO antenna systems and over-the-air (OTA) testing procedures are presented in Chapter 7. Although OTA testing has not yet been standardized, three methods have been used by industry and academia to evaluate end-to-end MIMO system performance. These three methods are described in this chapter with a focus on their operating principles and setups, as well as their advantages and disadvantages. A comparison between the anechoic chamber methods versus the reverberant chamber ones is also provided. This book is a complete reference book on printed MIMO antenna system design. It can be used in an advanced antenna design course or as a design companion for practicing engineers. The book provides full design examples from the literature with detailed illustrations for the various antenna geometries used; it also covers the various applications that currently depend on printed MIMO antennas. Design guidelines and remarks are provided throughout the book to give more guidance to the antenna designer. In addition, the book covers almost all of the work in this area that has appeared since 2008 until the time of the book’s completion.

Acknowledgments Writing a book is not like writing a scientific paper or a review article; it requires more time and effort to make complicated things easy to grasp and understood. It is a project that requires dedication, organization, and a lot of time to finish. But towards the end, you feel the joy of accomplishment by how much has been achieved after all those sleepless nights and long (or actually short!) weekends. I have to say that I owe my family a great deal for the time they have given me to finish this book. I have taken from my kids’ time as well as my spouse’s time, and I thank all of them for their understanding and patience. I would like to thank my graduate students, in particular Mr. A. B. Numan and Mr. M. U. Khan because parts of their research work are presented in various chapters within this book. I would also like to thank Mr. S. S. Deif for helping with the art generation in Chapter 7. Many thanks go to the book reviewers for their valuable comments and prompt replies. Their feedback and observations improved the content and flow of the ideas in various chapters. Thanks also go to my book commissioning editor, Aileen Storry from Artech House, for her patience, motivation, and quick follow-up during this project. She has been a great motivator. Her comments and expertise were very helpful to finishing the book on time. Finally, I would like to thank King Fahd University of Petroleum and Minerals (KFUPM) and the Deanship of Scientific Research for providing laboratory access, a suitable environment, and research funding opportunities from which several results in this book were accomplished.

xv

1 Introduction Wireless communications have evolved very rapidly in the past two decades. Nowadays we are dealing with small form factor wireless devices with powerful computing and application capabilities. Your cell phone can make calls, browse the Internet, play movies and music, and provide you with navigation assistance while on the road. Your portable computer now provides much faster wireless Internet connectivity than ever before, and you can even access the Internet while riding in your car or on a train. The introduction of several new wireless technologies and standards is behind this rapid proliferation of highperformance devices. In this chapter, we briefly present an overview of the wireless evolution of mobile phones and portable devices, followed by the need for multiband antennas within such devices. The new multiple-input-multiple-output (MIMO) technology is then introduced with an emphasis on the importance of multiantenna systems for such a technology. The effect of using multiple antennas within a MIMO system on the channel capacity and data rate is discussed. Finally, we take a look at the market forecast for mobile terminals with MIMO technology.

1.1  Wireless Technology Evolution In the early 1980s, the first analog-based mobile phone was commercialized (although it was introduced and prototyped in the late 1970s). It was intended for voice calls only and supported limited data transfer in the form of text as well. This was considered the first generation (1G) in cellular and mobile terminals and was based on the advanced mobile phone system (AMPS). A few years later, the technology matured a little more with the introduction of the 1

2

Printed MIMO Antenna Engineering

Global System Mobile (GSM) standard and the second generation (2G) of such phones was introduced with voice and data capabilities and a more than 200 times increase in the data rate compared to 1G. While mobile phone technology was evolving, wireless local-area networks (WLANs) started showing up in the late 1990s. Wireless standards such as Bluetooth and also ZigBee in the early 2000s became widely used in small wireless terminals and mobile devices. The cellular phone industry moved toward 3G around 2006 and to 4G around 2011. With each generation, severalfold increases in the data rates were achieved through the use of new technology and standards. This allowed data transfers in excess of 100 Mbps in cellular phones, which allowed for real-time video transfers and high-definition broadcasting. Figure 1.1 is a chart showing the wireless evolution and its capabilities in terms of data rates as well as applications. The x-axis shows increases in the data rate as the standards evolved, whereas the y-axis shows increases in the mobility of service. For example, 4G devices can support users moving at much higher speeds (i.e., riding trains moving at more than 100 km/h) with high data rates, whereas 1G devices were mainly for pedestrian or slow moving vehicle use. The new WLAN standards such as 802.11ad can provide multigigabit transfers at short distances for stationary users. Advancements in antenna design and microelectronics led to a dramatic reduction in mobile phone sizes, as is evident in Figure 1.2. First-generation phones were bulky, heavy, with limited capabilities (mainly for voice calls) and a short battery life. They used large antennas that were clearly seen outside the phone package (see the largest phone toward the left of Figure 1.2). Then

Figure 1.1  The evolution of wireless standards and their capabilities.



Introduction

3

Figure 1.2  The evolution of mobile phones.

the size started to be reduced, and the functionality and complexity started to increase due to the increased integration of microelectronic chips and compact packaging. We see that 2G and 3G phones were very small in size, with the antennas embedded within the phone casing (see the mobile phones at the center of Figure 1.2). Printed-type antennas have been used since 2G and beyond in user terminals. Then with 3.5G and 4G phones, we started to notice an increase in the phone size again (phones toward the far right side of Figure 1.2), but without that much increase in weight, and with an increase in all of its capabilities as well as battery life. These smartphones have multiple antenna systems that are part of the phone package to support multiple standards and high data rates. They also have large screens to provide good video and application interfaces to the user. Smartphones are moving toward being a small PC that fits in your pocket. In 3G and 4G phones, more capabilities were integrated and standards supported within a single portable platform compared to earlier generations. Thus several antennas were required to cover the various frequency bands of the various standards as well as within common bands to allow for diversity gain (multiantenna systems). Multiband antennas became an integral part of these portable devices. The various wireless standards cover different frequency bands, and these bands need to be supported by the antenna system to allow for proper reception by the mobile terminal. Table 1.1 shows the frequency bands for the various generations of cellular devices [1]. In addition to multiple standard coverage within cellular bands (i.e., a 3G phone should be backward compatible with 2G ones), WLAN connectivity at the 2.4- or 5-GHz band as well as global positioning system (GPS) support at 1.5 GHz are usually needed within a smartphone platform. Figure 1.3(a) shows an example of multiband antennas within the iPhone 4 mobile phone from Apple Inc. The first multiband antenna covers

IMT Advanced

LTE (Rel. 8) (2×2 MIMO) WiMax IEEE 802.16e LTE Advanced 70 Mbps >500 Mbps

10 MHz

Var. up to 100 MHz Var. up to 100 MHz 270 Mbps 675 Mbps

75 Mbps

Var. up to 20 MHz

1 Gbps IMT

IMT

2600/3500

900/1800 2100/2600

DL: > 30 UL: > 15 DL: > 15 UL: > 6.75

3.7

16.32

2.88 12.5

Peak Spectral Efficiency (bit/s/Hz) 0.17 0.13 EDGE 0.51

GSM: Global System Mobile; GPRS: General Packet Radio Service; EDGE: Enhanced Data Rates for GSM Evolution; HSPA: high-speed downlink packet access; IMT: International Mobile Telecommunication; LTE: long-term evolution; WCDMA: wideband code-division multiple access; WiMAX: Worldwide Interoperability for Microwave Access.

From [1].

4G

5 MHz

Carrier BW 200 kHz

HSPA 5 MHz HSPA+ (16 QAM) 5 MHz (64 QAM + Dual)

Generation Technology 2G GSM/GPRS EDGE 3G W-CDMA

Table 1.1 Mobile Phone Generation Bands and Features UL Peak DL Peak Latency Frequency Bands Data Rate Data Rate (ms) (MHz) 56 kbps 114 kbps 500 900/ 118 kbps 236 kbps 300 1800 384 kbps 384 kbps (2 250 900/1800 Mbps) 2100/2600 5.7 Mbps 14 Mbps ~70 900/2100/2600 11.5 Mbps ~30 900/ -28 Mbps (42 2100/2600 Mbps)

4 Printed MIMO Antenna Engineering



Introduction

5

Figure 1.3  Multiple antenna elements within an iPhone. (Source: Apple Inc.)

the Bluetooth (2.4-GHz), Wi-Fi (2.5- and 5-GHz), and GPS (1.575-GHz) frequency bands, while the second antenna covers the GSM/EDGE (800-, 900-, 1800-, 1900-MHz) and UMTS/HSDPA (850-, 900-, 1900-, 2100MHz) bands (UMTS is the Universal Mobile Telecommunication Standard). An example of multiple antennas with the same band coverage is shown in Figure 1.3(b). In this model, two identical antennas are used to improve the data rates of the system, as discussed in the following section, where multiple antennas covering the same bands are used to enhance the system capacity performance (or data rates). These antenna elements are also multiband ones in general so that they have a practical use in covering multiple wireless standards.

1.2  Multiple-Input-Multiple-Output (MIMO) Technology Fourth-generation wireless communication standards rely on three major technologies to achieve the high data rates they promise to deliver: adaptive modulation and coding (AMC), the use of orthogonal frequency-division multiple access (OFDMA) and the use of multiple-input-multiple-output (MIMO) technology. The former two technologies deal with the way the data is coded and modulated in an adoptive way to improve the transmission over changing wireless channels, the latter technology (MIMO) is the one that is concerned with the need for multiple antennas. MIMO technology combats multipath fading in wireless channels by sending the data from multiple antennas at the transmitter to multiple receivers at the receiver. The data from multiple antennas go through different paths in the wireless environment, and thus the chance of receiving a good representation of the transmitted data in such fading envi-

6

Printed MIMO Antenna Engineering

ronments increases, achieving better data rates when sending parallel streams of different data from each transmitting antenna. This is illustrated in Figure 1.4. Based on the channel conditions, the use of MIMO systems will yield an increase in the channel capacity (in bps) according to the modified Shannon equation given by [2]

N   C ≈ MB log 2 1 + SNR   M 

(1.1)

where M is the number of antennas at the transmitter side, N is the number of antennas at the receiver side, B is the bandwidth (in Hz), C is the channel capacity (in bps), and SNR is the signal-to-noise ratio. It is evident that a linear increase in the channel capacity is achieved when the number of antenna elements is increased. This is based on the assumption that the multipath channels are not correlated (i.e., an ideal environment). Thus any radiation pattern correlation or coupling between the antenna elements will degrade this performance enhancement, and designing MIMO antenna systems with low correlation between their elements is very critical in order to achieve the high data rates anticipated. Because MIMO systems rely on the use of multiple antenna elements at the user mobile terminal as well as the base station, the research in this area began picking up about 5 years ago. Having multiple antennas on the base station tower is usually not a problem because space is abundant there, and multiple antennas can be spaced a quarter- or half-wavelength apart to achieve low correlation and low coupling between them. This spacing issue becomes critical when the antennas are placed on the user terminal, because most of the times such spacing between the antennas is not possible due to the small form factor of the mobile terminal (i.e., a cell phone). This will make field correlations and mutual coupling increase and thus degrade the diversity performance of the MIMO antenna system. Most current mobile terminals incorporate printed antennas. Printed antennas make use of the system ground plane of the device, can be conformed to the device’s needs, are low cost, and have a mature enough technology that they can be fabricated with ease. The research community has been able to provide printed MIMO antenna solutions with attractive properties such as low mutual coupling and low correlation even when the antennas are very closely spaced within the mobile device. In the past few years, the amount of published research work in this area has been almost doubling every year [3]. Printed MIMO antennas with multiband operation and operating frequencies lower than 1 GHz are an active area of research that is still growing. The importance

Introduction

Figure 1.4  MIMO system block diagram.

7

8

Printed MIMO Antenna Engineering Table 1.2 Worldwide Devices Shipments by Segment (in thousands of units) Device type 2012 2013 2014 2017 PC (desk-based and 341,263 315,229 302,315 271,612 notebooks) Ultramobile 9,822 23,592 38,687 96,350 Tablet 116,113 197,202 265,731 467,951 Mobile phone 1,746,176 1,875,774 1,949,722 2,128,871 Total 2,213,373 2,411,796 2,556,455 2,964,783 Source: www.gartner.com.

of this area and the need for novel MIMO antenna solutions that are of practical use for next-generation mobile terminals are increasing on a daily basis.

1.3  Market Forecasts The forecasts for smart mobile phone and other portable device usage are shown in Table 1.2. The forecasts show that by the year 2017, smartphone proliferation will increase by approximately 15% compared to its 2014 forecast, thus dominating the market share. Such smartphones will include multiband and MIMO-based antenna systems. This shows the importance of these antenna systems in real practical applications. In addition, the data traffic from these smartphones will dominate the world mobile data traffic starting in 2013 according to Figure 1.5, exceeding traffic from laptops and tablet PCs. More than 65% of the world data traffic will be dominated by smartphones in 2017, and a compound annual growth rate (CAGR) of 66% in the data traffic from mobile devices between 2012 and 2017 is forecasted [4]. Thus efficient and reliable antenna systems are a major design requirement within all of these portable and small form factor devices. This book focuses specifically on the design of printed MIMO antennas for different types of wireless terminals, including large devices such as laptop computers and access points as well as small devices such as mobile phones and USB dongles.

1.4  Conclusions MIMO-based wireless communication devices will rely on multiple antennas that are integrated within the mobile device. The use of multiple antennas can increase data rates and thus provide users with more multimedia and real-time video connectivity and data transfer capabilities. The market forecasts show that MIMO-based devices will grow exponentially in the years to come, and MIMO



Introduction

9

Figure 1.5  Data transmission forecast for mobile wireless terminals. (Source: Cisco Inc.)

technology will be used in all future wireless generations. This means that the antenna portion of the system should be carefully designed and characterized to allow the system to achieve its anticipated high data rates. The following chapters in this book present the performance metrics, design, characterization of, and procedures used in the design of printed MIMO antenna systems for mobile wireless devices and terminals.

References [1] “Recognizing the Promise of Mobile Broadband,” White Paper, UMTS Forum, July 2010. [2] V. Garg, Wireless Communications and Networking, San Francisco: Morgan Kaufmann, 2007. [3] M. S. Sharawi, “Printed Multi-band MIMO Antenna Systems and Their Performance Metrics,” Antennas and Propagation Magazine, Vol. 55, No. 5, pp. 218–232, 2013. [4] “Cisco Visual Networking Index: Global Mobile Data Traffic Forecast Update, 2012– 2017,” Cisco, 2013.

2 Antenna Fundamentals In this chapter, we start with a brief overview of the various antenna parameters that characterize the performance of single antennas in free space. Such parameters include the radiation pattern, efficiency, directivity, and gain. Then we present details for another set of parameters that become extremely important when we design MIMO antenna systems. These parameters include the capacity increase due to the use of multiple antenna elements, total active reflection coefficient, isolation, correlation coefficient, mean effective gain, and diversity gain. Following the introduction of the parameters needed to characterize MIMO antenna system performance, we present some of the widely used printed antenna elements that can be used in printed MIMO antenna systems: printed dipoles, patches, and others. Then we conclude the chapter with a brief section about linear antenna arrays. In this chapter, we consider MIMO antenna parameters as fundamental quantities that need to be analyzed and optimized early in the design process to ensure a successful MIMO antenna system that will provide the anticipated channel capacity increase for 4G communication systems and beyond.

2.1  Radiation Mechanism Radiation of electromagnetic fields occurs whenever there is an acceleration/ deceleration of charge or time-varying currents within a conductor [1]. This is a direct consequence of Maxwell’s equation. Time-varying currents are the ones used in wireless communications to carry information, and once they pass through a conducting element that is terminated with a discontinuity such as free space, portions of them get radiated. 11

12

Printed MIMO Antenna Engineering

An example of such radiation due to a time-varying current exciting a discontinuous conductor is shown in Figure 2.1. The time-varying voltage source generates a wave through a transmission line or waveguide. The wave gets guided until it reaches the radiating element (antenna). Upon reaching the conducting radiating element (antenna), the discontinuity at its ends causes radiation by allowing the electric field lines to detach after half a period of the incoming wave. An antenna is a device or element that radiates (transmits) and captures (receives) electromagnetic fields. Antennas can be metallic or dielectric in nature [2–4]. Metal-based antennas are the most common type used in current wireless communication devices and terminals. Figure 2.2 shows the mechanisms of radiation from a simple two-wire antenna (a dipole). Figure 2.2(a) shows the time-varying excitation (a sine wave). During the first quarter of its cycle, positive charges accumulate on its upper side. Electric field lines form from top to bottom as shown in Figure 2.2(b) until the excitation reaches its maximum after λ/4. The excitation starts decreasing in value, which can be interpreted as a change in charge polarity between the top and the bottom arms of the antenna. This causes a reversal in the field line direction as shown in Figure 2.2(c). After half a cycle (λ/2), the total excitation is zero, and thus there should be no net charge on the two-wire antenna arms. This indicates that the electric field line closes on itself and detaches from the antenna. The same process is repeated in the opposite direction. We can obtain some closed-form expressions for the radiated fields within some metallic-type antennas such as a wire dipole. A detailed treatment can be found in [5–7] and, therefore, is not discussed in this section. The radiation phenomena also take place if a constant current is used but within a path that has discontinuity within it. Printed circuit boards (PCBs)

Figure 2.1  A radiating antenna system diagram showing the source, transmission line, and dipole antenna along with the current direction.



Antenna Fundamentals

13

Figure 2.2  The creation of electric field lines in the positive and negative signal cycles and the detachment of the wave from the antenna into free space: (a) sinusoidal signal, (b) one quarter of the cycle, and (c) second quarter of a cycle.

should also be checked for unwanted radiation, especially at high digital clocking frequencies of operation. Although radiation can be obtained or created by satisfying simple conditions, to achieve an efficient radiator (antenna), we usually have to satisfy more conditions to maximize the radiation and obtain good performance. The task of designing an efficient antenna is on the shoulders of the design engineers who need to satisfy the radiation conditions and come up with a structure that radiates efficiently given the desired design constraints. The presence of a charge density and current flow (current density) give rise to vector and scalar magnetic potential functions from which the radiated and fields can be found. Figure 2.3 shows the electric field lines around a dipole antenna.

14

Printed MIMO Antenna Engineering

Figure 2.3  Electric field lines around a dipole antenna from a 2D MATLAB simulation result for a dipole antenna.

The space around a radiating antenna is divided into three regions depending on how far we go from the antenna element (Figure 2.4): 1. Reactive field region (Zone 1): This region lies between the antenna and D3 , where D is the largest dimension λ of the antenna and λ is the wavelength of operation. Stored energy within the antenna resides in this region. 2. Radiating near-field region (Zone 2): This region starts from the rethe sphere of radius R1 < 0.62

active near-field region and ends at the far-field boundary of 3

2

2D2

l

.

D 2D . In this region, < R2 < λ λ radiating fields dominate, and the general shape of the pattern is formed. This region might not exist if the antenna’s largest dimension is much smaller than a wavelength. 3. Far-field region (Zone 3): This is where we want to measure the radiating fields of an antenna. The parameters of the antennas, such as the gain and directivity, that are used in communication systems link budgets are assumed to be operating in this region, which is far enough 2D 2 from the radiating antenna with a minimum distance R3 > . λ Thus, this region lies between 0.62



Antenna Fundamentals

15

Figure 2.4  The radiation zones around an antenna as a function of distance.

2.2  Single Antenna Parameters In this section, the basic metrics and parameters that are used to characterize single antennas are presented. These parameters include the resonance that specifies the band of operation within which the antenna can operate, the radiation pattern that specifies the spatial characteristics of its radiated fields, and the antenna’s directivity, efficiency, gain, and polarization. 2.2.1  Resonance

Every antenna structure has at least one resonant frequency at which its reactive part approaches zero (or a minimum value) and its input impedance looks real. Because most antennas can be modeled as RLC circuits, at resonance the real part dominates. The radiation resistance is a function of the antenna structure and has specific formulas for various well-known antennas [5, 6]. Each antenna type will resonate if its dimensions satisfy a certain relationship with the wavelength of the exciting source. For example, a thin wire antenna in a dipole configuration will resonate and radiate if its length (total

16

Printed MIMO Antenna Engineering

λ (see Figure 2.5). So, if you are interested in 2 designing a simple dipole to operate at 1800 MHz, your arm length should be length) is approximately equal to







l=

c f

(2.1)

λ (1800 MHz) = 0.1667 m

→ λarm =

λ = 4.167 cm 4

In real designs, one needs to tune to the length of the arms using a computer-aided design (CAD) tool because the preceding formulas are based on simple approximations about the wire used. Usually the single arm length is less than 0.25 and lies between 0.21λ and 0.24λ. CAD tools should be used to verify the design before prototyping it. The resonance behavior of an antenna is identified using the reflection coefficient curves. This curve will allow the designer to identify the center resonance frequency of the antenna as well as its operating bandwidth (BW). Figure 2.6 shows a sample printed dipole antenna geometry and its corresponding reflection coefficient curve (magnitude of |S11|). The antenna was modeled on an FR-4 material substrate with a 4.4 dielectric constant and 0.8-mm thickness. The metal sheet width was 1.1 mm. The antenna has a center resonance frequency of 2.37 GHz and a –10-dB bandwidth of 260 MHz. The center

Figure 2.5  Dipole antenna arm lengths with respect to operating wavelength.



Antenna Fundamentals

17

Figure 2.6  Printed dipole modeled using a 3D EM field solver (HFSS): (a) geometry and (b) reflection coefficient curve.

resonant frequency is the point of minimum reflection coefficient or best matching condition. The –10-dB BW corresponds to the 2:1 voltage standing-wave ratio (VSWR) of the antenna, which is also a measure of matching. Matching the input frequency dependent impedance of the antenna to that of the feeding line is an important design aspect as well. In recent years, the antenna design community accepted a more relaxed matching condition, and a –6-dB operating BW is usually used for electrically small antennas [8–10] and closely packed antennas such as in MIMO antenna systems [11, 12]. The –6-dB BW corresponds to the 3:1 VSWR ratio. The relationships that tie the BW, |S11|, and VSWR for microstrip radiators are given by the following equations:

VSWR =

1 + S11

1 - S11

(2.2)

18

Printed MIMO Antenna Engineering





VSWR - 1 VSWR + 1

(2.3)

VSWR - 1 QT VSWR

(2.4)

S11 =

BW =

where QT is the total quality factor of the radiator. 2.2.2  Radiation Patterns

The radiation pattern of an antenna shows the spatial distribution of its radiated energy. These patterns are important when characterizing an antenna because they show its region of coverage and its coverage shape. These patterns are most of the times measured or calculated in the far field of the antenna. The direction of maximum radiation, the half-power beamwidth (HPBW), and the sidelobe levels (SLLs) among other parameters can be deduced from radiation patterns. Radiation patterns are sometimes described by 2D cuts taken at specific angles to simplify the visualization process. Most of the literature either shows the gain patterns as the radiation patterns of the antenna under test [13, 14] or the electric fields radiated as the radiation patterns [15, 16]. Figure 2.7 shows a 2D rectangular plot (cut) of the E-field pattern radiated from a printed patch antenna on a finite ground plane substrate operating at 1.575 GHz at f = 90° and simulated using the 3D electromagnetic field solver HFSS [17]. From the figure, the direction of the maximum E-field along this cut as well as the HPBW are easy to identify. The maximum radiation occurred at q = 0° and the HPBW was 83°. The SLL with respect to the main lobe was approximately –19 dB. 2.2.3  Directivity

All realizable antennas are directive, which means that they have concentrated energy radiations toward some specific regions surrounding the antenna. The directivity of the antenna is defined as the ratio between the radiation intensity of an antenna in a given direction to its total radiated power divided over the complete sphere around it. This can be expressed as [5, 6]



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19

Figure 2.7  The E-field as a function of elevation angle for a GPS patch antenna.

D ( q, f) =

4 pU ( q, f) = Prad

4 pU ( q, f)

4 pU ( q, f)

 ∫∫ U ( q, f) dΩ  ∫∫ U ( q, f) dΩ Ω

U ( q, f) =

=

Ω

(2.5)

2 r2 r2  2 2 E (r , q, f) ≅ E q (r , q, f) + E f (r , q, f)    2η 2η 

where D(θ) is the directivity as a function of the spatial coordinates; U(θ, φ) is the radiation intensity in free space and is proportional to the electric field magnitude |E|, the distance r, and the medium characteristic impedance η; and Ω is the solid angle. The directivity is unitless, and sometimes is expressed in decibels. Directivity patterns are also considered to be a form of radiation pattern. Figure 2.8 shows the 3D directivity pattern in decibels of the patch considered in Figure 2.7. Notice that the maximum directivity is toward (θ = 0°) with a value of 6.7 dB. 2.2.4  Efficiency

The efficiency of an antenna is defined as a percentage that shows the amount of radiated power to the amount of power supplied to an antenna. It is a measure of the losses within the antenna field structure and radiating structure. Thus, it includes losses due to any impedance mismatch at the input terminal

20

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Figure 2.8  Directivity pattern of a GPS patch antenna operating at 1.575 GHz.

of the antenna as well as any losses within the radiating structure of an antenna (i.e., conductive or dielectric losses). The total radiation efficiency (η) can be expressed as

η = ηcd ηr

(2.6)

where ηcd is the factor due to the conductive and dielectric losses, and ηr is the factor due to any impedance mismatches. The former is usually not easily controlled, because it depends on the material properties. Thus, choosing a substrate of lower dielectric constant can maximize this factor (since it has lower losses). The later factor can be maximized by proper impedance matching between the transmission line feeding the antenna and the antenna input point. This is given by

(

ηr = 1 - Γ

2

)

(2.7)

where Γ is the reflection coefficient at the input terminal of the antenna. Although ηr is easy to calculate if the impedances are known, ηcd is usually difficult to calculate, so for this calculation we either rely on computer software to estimate this value or measure it in the laboratory. In literature, the radiation efficiency usually does not include the impedance mismatches, whereas total radiation efficiency does.



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21

2.2.5  Gain

The gain is another quantity or parameter that describes the behavior of an antenna. In most practical cases we use the relative gain, which is defined as the ratio between the power gain in a certain radiation direction and the power gain of the reference antenna in the same direction. The 3D gain pattern is related to that of directivity by

G ( q, f) = ηcd D ( q, f)

(2.8)

which shows that the gain is equal to the directivity times the efficiency of the radiator excluding mismatch and polarization effects. The gain pattern of the GPS patch we introduced in Figures 2.7 and 2.8 will have a similar pattern shape but with a scaled value according to (2.8). The efficiency of the antenna was approximately 50% as obtained from the field solver (using an FR4 substrate). 2.2.6  Polarization

The polarization of the electromagnetic wave is defined as “that property of a radiated electromagnetic wave describing the time varying direction and relative magnitude of the electric field vector; specifically the figure traced as a function of time by the extremity of the vector at a fixed location in space, and the sense in which it is traced, as observed along the direction of propagation” [1, 18]. The three types of electromagnetic wave polarization are shown in Figure 2.9. The free tool EMANIM1 was used to generate the three cases to illustrate the phenomena. It can be downloaded freely from the Internet and can show real-time animation so that the user gets a better feel for the phenomena. Figure 2.9(a) shows a linearly polarized wave where the vector describing the field at a point in space as a function of time traces a line. Figure 2.9(b) shows a right-hand circularly polarized wave. The point in space traces a circle in a clockwise direction (a left-hand circularly polarized would trace in a counterclockwise direction). Finally, Figure 2.9(c) shows the general case of elliptically polarized wave propagation. Thus, the point in space as a function of time traces an ellipse. This type of polarization can be generated when summing two linearly polarized waves with different amplitudes and phases. Check the vectors of the two components of the elliptically polarized wave in Figure 2.9(c) to observe this condition.  The necessary conditions to obtain any of the three polarizations of the E or H field components can be found in [5, 6], and there is no need to repeat them here. For linear polarization, we have vertically and horizontally polarized 1. http://www.enzim.hu/~szia/emanim/emanim.htm

22

Printed MIMO Antenna Engineering

Figure 2.9  The three main polarizations of an EM wave: (a) linear (vertically polarized), (b) circular (left-hand circularly polarized), and (c) elliptical.

waves based on their direction with respect to the ground. For circular, we have left-hand and right-hand circularly polarized waves. To check the polarization of an antenna, we need to have a look at the Eθ and components of the radiation pattern. This can be obtained from the antenna modeling tool or via measurements. Most practical printed antennas for mobile terminals have elliptical polarization because they operate indoors where polarization reversal can be easily encountered, thus they can work properly and receive a linearly transmitted signal (whether vertically or horizontally polarized one).



Antenna Fundamentals

23

2.3  MIMO Antenna Parameters In the previous section, we gave an overview of the basic single antenna parameters that an antenna designer needs to check to characterize antenna performance. These parameters are enough for single antenna designs. When we consider the design of multiple antennas, however, especially for MIMO systems, several other parameters needs to be evaluated to characterize the true antenna system performance. In this section, we review the parameters that need to be characterized for MIMO antenna systems: the total active reflection coefficient, isolation, correlation coefficient, mean effective gain, diversity gain, branch power ratio, and channel capacity. 2.3.1  Total Active Reflection Coefficient

To properly characterize the efficiency and bandwidth of the MIMO antenna system, the scattering matrix is not enough [19]. To realize better characterization, the total active reflection coefficient (TARC) is used. TARC is defined as the ratio of the square root of the total reflected power divided by the square root of the total incident power [20] in a multiport antenna system. TARC can be computed using the scattering parameters of the MIMO antenna. For an N-element antenna, TARC is given by

∑ i =1 bi N



Γta

=

∑

2

2 N a i =1 i



(2.9)

where ai and bi are the incident signals and reflected signals, respectively. These can be computed from the measured S-parameters. The relationship between the incident and reflected waves in a multiport network with similar characteristic impedances at all ports is given by [21]

b = Sa

(2.10)

where S is the S-parameter matrix. The boldface in (2.10) indicates vector quantities with magnitude and phase. Thus, for an N-port MIMO antenna system, the S-parameter matrix will become N × N. TARC accounts for coupling as well as random signal combinations between ports. TARC has a value between 0 and 1, where a 0 means all power was radiated, whereas a 1 means all incident power was reflected and nothing was radiated. The available power is the sum of powers available on all ports of the antenna system. TARC is presented in decibels.

24

Printed MIMO Antenna Engineering

S-parameter matrixes grow exponentially with the increase in the number of antennas. For a two-antenna system, the S-parameters matrix is 2 × 2, and for a three-element antenna system the matrix size grows to 3 × 3. It is very difficult to track all of the curves for a large number of radiating elements in an antenna system. TARC is a method for manipulating all of the S-parameters for an N-port network and displaying a single curve that has all the information of the S-parameters. In addition to compressing the information from many curves into a single curve, TARC also includes the effect of a feeding phase to the antenna port. Hence, a single curve of TARC can be used to determine the resonance frequency and impedance bandwidth of the whole antenna system for a specified phase excitation between the ports [22]. For a two-port MIMO antenna system, TARC can be evaluated using [23]



Γta =

2   jq 2   + S + S e j q    S11 + S12e   21 22 

(2.11)

2

where θ is the input feeding phase, Sxx is the reflection coefficient of the port, and Sxy is the isolation between the two ports associated with the antenna structure. Once the S-parameters of a two-port network have been found, the random phase is swept between 0 and 180 degrees to investigate the effect of the phase variation between the two ports on the resonance behavior of the antenna and thus create the corresponding TARC curves. Figure 2.10(a) shows the TARC curves created for phase steps of 30 degrees starting from 0 to 180 degrees for the two-port antenna system presented in [23] and using (2.11). Figure 2.10(b) shows the S-parameters for the same two-element MIMO antenna system. It is

Figure 2.10  TARC curves (a) compared to the two single-port reflection coefficient curves (b). Note the narrower bandwidth and the effect of the phase difference between the two antenna ports. (From: [23]. © 2012 IEEE. Reprinted with permission.)



Antenna Fundamentals

25

evident that the S-parameters do not present the effective BW of the antennas, while TARC does (i.e., the TARC –10-dB BW is lower). If the TARC for a multiport antenna system is needed for more than two ports, (2.10) is used in conjunction with (2.9). Port 1 is excited with a signal having unity amplitude and zero phase (i.e., 1e  j0), while the other ports are excited with similar amplitudes but different phases. Then, after applying (2.10) to get the reflected signal values (bi ) from the incident (excitation) values (ai ), (2.9) is used to find the TARC values. Because the S-parameters are recorded as a function of frequency, the TARC curves will cover the same frequency range as well. 2.3.2  Isolation

Isolation measures how much power is coupled between adjacent radiators within the multiantenna system structure. It does not represent the coupling through the radiation patterns, only that within the structure of the antenna system (i.e., the substrate and ground plane). Isolation is measured using the S-parameters. The transmission coefficient (Sxy) between the two radiators’ feeding ports (radiator “x” and radiator “y”) measures this quantity. Figure 2.11 shows the isolation curve between four printed MIMO antennas that fit within a handset size for 4G wireless applications [24]. Note that the isolation is considered good between elements 1 and 2, and 1 and 3, whereas between 1 and 4, the isolation is low (approximately 7 dB) around the operating frequency of 2.45 GHz. The isolation can be compromised by a number of factors. The radiating elements can couple with each other through electric and/or magnetic fields within the antenna structure. Ground plane currents can also be a major factor in coupling the radiating elements, especially for small printed antennas that have a small shared ground plane. So, it is important to determine the exact cause of the coupling before using a technique to lower it and enhance the isolation of the antenna system. A large number of techniques are available in the literature that addresses this issue (i.e., isolation enhancement). These techniques will be discussed in detail in Chapter 6. 2.3.3  Correlation Coefficient

The correlation coefficient ( ρ) is a measure that describes how much the communication channels are isolated or correlated with each other. This metric considers the radiation pattern of the antenna system and how much the patterns affect one another when operated simultaneously (which is the case in a MIMO antenna system). The square of the correlation coefficient is known as the envelope correlation coefficient. The envelope correlation coefficient ( ρe) can be calculated using the following formula [25]:

26

Printed MIMO Antenna Engineering

Figure 2.11  Isolation curves between four printed MIMO antenna elements operating around 2.45 GHz for a 4G wireless handset application. (From: [24]. © 2012 IET. Reprinted with permission.)



   q , f * F F ) ( 2 ( q, f)d Ω ∫∫  1  4p

ρe =



∫∫ F1 (q, f)

4p

2



2

2



(2.12)

d Ω ∫∫ F2 ( q, f) d Ω 4p

 where Fi ( q, f) is the 3D field radiation pattern of the antenna when the ith port is excited, and Ω is the solid angle. The expression in (2.12) is valid when a uniform multipath environment of balanced polarization is considered (i.e., an isotropic environment). It is a complicated expression that requires 3D radiation pattern measurements and numerical integration. A simple derivation in [25] shows that the correlation coefficient can be calculated using the S-parameters and the radiation efficiency of the antenna system. This expression is given by



Antenna Fundamentals

27 2



ρij

2

= ρeij =

(

1 - Sii

Sii* Sij + S ji* S jj 2

- S ji

2

)(

1 - S jj

2

- Sij

2

)

1/2



(2.13)

where ρij is correlation coefficient between elements i and j, ρeij is the envelope correlation coefficient, and Sij is the S-parameter between the i and j elements. In this formula we need to know only the S-parameters that can be evaluated easily compared to the 3D radiation patterns required by (2.12). A correlation coefficient value of 0.3 has been set as an acceptable value for 4G wireless systems [26]; the upper value for the envelope correlation coefficient is 0.5. It should be noted that (2.13) is applicable for antennas with high efficiencies. We should mention that the isolation and correlation coefficient are two different quantities. High isolation does not guarantee a low correlation coefficient and vice versa. A high isolation and low correlation coefficient are required for a MIMO antenna system with good diversity performance. Figure 2.12 shows the envelope correlation coefficient curves for the four-element printed MIMO antenna system presented in [24]. 2.3.4  Mean Effective Gain

The mean effective gain (MEG) is a measure of the antenna performance in a predefined wireless environment where the effect of the environment is taken into account in the gain performance of the antenna. The stand-alone antenna gain (i.e., the one calculated for single antenna elements as defined in Section 2.2.5) is not a good measure of antenna performance because the antenna is not used in an anechoic chamber in practical applications. Antennas are used in certain environments for specific applications. So the study of the effect of the environment on an antenna’s radiation characteristics is important to evaluate its true performance. One way to do this is to fabricate an antenna, operate it under the specific conditions along with another standard antenna with known characteristics, and determine the antenna performance. We have to fabricate a prototype, test it to get the results, tune the antenna, and repeat the process to get the desired design. This procedure is very time consuming and costly. The practical method of calculating MEG is described in [27]. A solution to this problem was proposed in [28]. In this work, a probabilistic model for the environment was proposed and by using the 3D radiation

28

Printed MIMO Antenna Engineering

Figure 2.12  Envelope correlation coefficient curves between four printed MIMO antenna elements operating at 2.45 GHz for a 4G wireless handset application. (From: [24]. © 2012 IET. Reprinted with permission.)

patterns along with the proposed statistical model we can obtain MEG numerically by solving a mathematical expression that combines the two quantities. This numerical method allows us to determine the MEG using the simulated/ measured gain patterns in an ideal environment (i.e., the simulation tool or an anechoic chamber) and a model of the environment suitable for the application in which antenna is being designed. The mathematical expression for the MEG calculation is shown in (2.14) and (2.15):

MEG =

2 pp

∫∫

0 0

{

}

XPD 1 G q ( q, f) Pq ( q, f) + G f ( q, f) Pf ( q, f) (2.14) 1 + XPD 1 + XPD

which satisfies the conditions



Antenna Fundamentals

29

2 pp

∫ ∫ {G q (q, f) + G f (q, f)} sin qd qd f = 4 p



0 0 2 pp

2 pp

0 0

0 0

∫

∫Pq (q, f) sin qd qd f =

XPD =

∫ ∫Pf (q, f) sin qd qd f = 1



(2.15)

PV PH

where XPD is the cross-polarization power ratio (or cross-polarization discrimination) that represents the distribution of the incoming power (the ratio between the vertical mean incident power to the horizontal mean incident power); Gθ(θ, φ) and Gφ(θ, φ) are antenna gain components; and Pθ(θ, φ) and Pφ(θ, φ) represent the channel model. The equations in (2.15) represent the conditions needed for evaluating (2.14). A number of channel models are available in the literature. A channel model suits a particular environment: urban, rural, and so forth. A general channel model was given in [28]. This model assumes a uniform distribution for the signals in the azimuth direction and a Gaussian distribution in the elevation direction. This represents a regular Rayleigh fading channel for cellular communications. Mathematically it can be written as follows:



2    p     q -  - mv         2  Pq ( q, f) = Aq exp  -   , (0 ≤ q ≤ p ) 2 σV2     2    p    q -  - mH       2     Pf ( q, f) = Af exp  -  2  , (0 ≤ q ≤ p) 2σH    

(2.16)

where Ai is the amplitude in the vertical or horizontal directions, mi is the horizontal or vertical mean, and σi is the standard variation for a horizontal or vertical statistical model. For this model, the average XPD values of 0 and 6 dB are usually used for indoor and urban environments, respectively [29]. MEG is usually evaluated at these two XPD values in MIMO antenna systems.

30

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This model requires 3D radiation pattern measurement and processing. To simplify the process, the incoming waves are assumed to be concentrated in the horizontal plane only. This assumption significantly reduces the complexity of the MEG calculation, which becomes



MEG =

 XPD  G ( p 2, f) Pq ( p 2, f)  1 + XPD q    1 + G f ( p 2, f) Pf ( p 2, f)  1 + XPD 

2p 

∫

0

(2.17)

Furthermore, if we assume a uniform distribution in the horizontal plane, then the channel model becomes Pq ( p 2, f) = Pf ( p 2, f) =



1 2p

(2.18)

In [29], MEG is calculated for a mobile antenna under an urban environment model. A statistical model for a multipath line-of-sight urban environment is proposed and an experimental MEG was calculated to compare the results with the theoretical model. In [30], MEG was analyzed in a Ricean channel. Interpretation of MEG under Rayleigh fading is also provided in this reference. The models provide a good representation based on the behavior of the antenna in these channels. 2.3.5  Diversity Gain

Diversity is usually achieved when the transmitter receives multiple versions of the transmitted stream through different channel paths (since we have multiple antennas). If the signals are uncorrelated, the combined signals at the receiver will provide higher SNR levels and thus better signal reception. Diversity gain (DG) is a measure of the effect of diversity on the communication system. Diversity gain is defined as the difference between the time-averaged SNR of the combined signals within the diversity antenna system and that of a single antenna system in one diversity channel, provided the SNR is above a reference level. Mathematically the diversity gain is defined as follows [31]:

 g g  Diversity gain =  c - 1   SNR c SNR 1 P ( gc < gs /SNR )

(2.19)



Antenna Fundamentals

31

where gc and SNRc are the instantaneous and mean SNR for the diversity system, respectively, and g1 and SNR1 are the instantaneous and mean SNR for the single branch with maximum values in the diversity system; gs is the SNR reference level. Assuming uncorrelated signals with Rayleigh distribution, the probability that the instantaneous mean SNR of the diversity system is less than the reference level [P ( gc < gs /SNR )] can be approximated as

g     g  P  gc < s  = 1 - exp  - s     SNR   SNR  

M



(2.20)

where M is the number of antennas. The increase in the number of antennas will increase the received combined power in a diversity system (MIMO system). Figure 2.13 shows the obtained diversity gain as a function of the number of antenna branches according to (2.20). The 1% point on the curves corresponds to 99% reliability and is the mark where diversity gain is accessed. As can be seen, the more antenna elements, the larger the obtained gain up until 6 or higher where the amount of gain obtained offers little improvement. The diversity gain and correlation coefficient are related. The lower the correlation coefficient, the higher the diversity gain.

Figure 2.13  Cumulative distribution function of (2.20) showing the diversity gain obtained as a function of the increase in the number of antenna elements.

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2.3.6  Branch Power Ratio

Another factor that affects the performance of a MIMO antenna system is the relative power levels coming from the different antenna branches. The power levels of the various branches should be very close. To represent these power level differences, we introduce the two branch power level ratio (k) as k=



Pmin Pmax

(2.21)

where Pmin is the power level of the antenna with the lower power, and Pmax is the power from the higher power antenna. The power level ratio will affect the diversity gain, as the overall obtained diversity gain is multiplied with the inverse of k to obtain the actual diversity gain. This revises (2.20) to

g  1   g  P  gc < s  ≈ 1 - exp  - s      SNR    k SNR

M



(2.22)

The branch power ratio (k) can also be found from the MEG values calculated from a MIMO antenna system. For a two-element MIMO antenna system,

 MEG2 MEG1  k = min  ,  MEG1 MEG2 

(2.23)

where MEG1 and MEG2 are the mean effective gains of antennas 1 and 2, respectively, found from Section 2.3.4. To obtain the maximum diversity gain and avoid significant loss in the diversity performance of the MIMO antenna system, k should be more than –3 dB (between 0 and –3 dB). Some examples will be provided when we discuss the performance of printed MIMO antenna systems in Chapters 5, 6, and 7. 2.3.7  System Capacity

The main advantage of a MIMO antenna system is that it provides an improved channel capacity in a multipath environment as compared to a SISO system. Therefore, the upper bound of the channel capacity of a MIMO antenna system is also a performance metric. The channel capacity of a MIMO antenna system depends on the channel matrix, which is a function of the radiation characteristics of the antenna elements and the channel environment. In case of an N-element MIMO antenna, when the channel conditions are not known to



Antenna Fundamentals

33

the transmitter, equal power is allocated to each element of the MIMO antenna system. The channel capacity under these circumstances is given as

ρ    C = log 2 det  I N + HH    N  

(2.24)

where ρ is the average SNR, H is the normalized channel coefficient matrix, IN is an N × N identity matrix, and N is the number of antenna elements at the receiver as well as the transmitter side. If all channels are totally uncorrelated and the antenna elements have a zero correlation coefficient, then HHT becomes an identity matrix and a linear increase in channel capacity is obtained as compared to a SISO system when number of antenna elements increase. Thus, (2.24) becomes

ρ  C = N × log 2 1 +   N

(2.25)

This is an ideal channel capacity limit for a MIMO system. This limit cannot be achieved because there is always some correlation between the channels so the correlation coefficient between the antenna elements is never zero. Thus, the mutual coupling between the antenna elements and the correlation in the channel results in degradation in the performance of a MIMO antenna. Note that in a line-of-sight (LOS) environment, when all of the channels are totally correlated, the performance of a MIMO antenna is nothing but that of a SISO one with an increase in the effective aperture of the antenna due to multiple elements. Thus, the real use of MIMO antenna systems is in the multipath environments that exist in practical wireless mobile communication environments. In such environments, the H matrix contains information about the correlation between different channels due to the antenna as well as the propagation environment. Thus, the calculation of the channel coefficient matrix H of a MIMO antenna system operating in a particular environment is important as it relates to the channel capacity of the antenna in that environment. Different experimental methods are employed to find the channel coefficient matrix, which includes the channel sounding technique. In [32], a theoretical model is derived to find the channel coefficient matrix for a particular MIMO antenna system. For the propagation environment, the assumptions are that the fading envelope is Rayleigh distributed, the incoming wave arrives in the horizontal plane only, and the time-averaged power density per steradian is constant. The channel coefficient matrix is defined by the Kronecker model which defines H as

34

Printed MIMO Antenna Engineering

H = ψ1R 2G ψT1 2



(2.26)

where G is a random matrix with i.i.d. complex Gaussian entries and ψR and ψT are the receive and transmit correlation matrixes, respectively. When the receiver and transmitter antennas are the same, ψR and ψT become equal. The entries of the ψ matrix depend on the field pattern of the antenna elements of the MIMO antenna system as well as the characteristics of the propagation environment. For the assumption made in [32], the (i,j) entries of the ψ matrix are given as follows ψi , j =

µij µii µ jj

(2.27)

p   *p  , ϕ A j q  , ϕ  2 2  d ϕ   p   *p  + Ai ϕ  2 , ϕ A j ϕ  2 , ϕ   

(2.28)

where 



µij =

2 p (XPD)Ai q

∫

0

In the above equation, XPD is the cross-polarization discrimination of the incoming signal. It is the ratio of the horizontally polarized component to the vertically polarized component of the incident signal. In an urban fading environment, which has many multipaths, this ratio is almost equal to 1 (0 dB). Using these equations, the channel coefficient matrix of a particular antenna operating in an urban environment can be found and then used to find the cumulative distributive function (CDF) of the channel capacity of the antenna as well as its mean channel capacity versus SNR. This is an important model that can be used to compare a novel antenna’s channel capacity characteristics with that of a standard dipole antenna or a hypothetical isotropic antenna. It can also be used to find the loss in channel capacity due to mutual coupling between the antenna elements. For a two-element printed patch-based MIMO antenna system operating at 2.45 GHz with a separation of 10 mm between the patch edges, the CDF of the channel capacity was computed at an SNR of 20 dB using the abovementioned model and equations. The correlation between the radiated fields from the two antennas was computed. Figure 2.14 shows these CDF curves for the two-element patch MIMO antenna, a SISO one and the ideal curve based

Antenna Fundamentals

Figure 2.14  CDF of channel capacity of a two-element MIMO antenna system compared to ideal two-element antenna and SISO antenna systems.

35

36

Printed MIMO Antenna Engineering

on (2.24) with zero correlation between the radiated fields. The CDF curve of the two-element patch antenna shows that the channel capacity is better than the SISO antenna system. However, the decrease in the channel capacity of the designed antenna compared to the ideal one is due to the correlation between the field patterns of the two antenna elements. Because it is difficult to achieve zero correlation in practical designs, the channel capacity of a practical system in rich multipath environments is always less than the theoretical channel capacity of a MIMO antenna system whose correlation coefficient between the antenna elements is assumed to be zero. Figure 2.15 shows the average channel capacity of a two-element printed patch-based MIMO antenna operating at 2.45 GHz compared to a SISO antenna system and an ideal two-element MIMO antenna system operating in an urban fading environment. The improvement in the capacity of the designed antenna with the increase in SNR is much more than that of a SISO antenna system. At an SNR of 30 dB, the average channel capacity of the designed antenna is 4 bps/Hz higher than that of a SISO antenna system.

2.4  Printed Antenna Types Printed antennas are widely used in mobile devices because of their small size, ease of integration, and low manufacturing cost and mechanical support. In this section, we discuss the most widely used printed antenna geometries/structures and give some examples of their features and applications.

SISO Ideal

Figure 2.15  Average channel capacity of the designed MIMO antenna system compared to ideal antenna and SISO antenna systems.



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2.4.1  Printed Dipole Antennas

The dipole antenna is very well known and was among one of the first antennas to be realized. The basic radiation mechanism that was presented in Section 2.1 was based on the two-wire antenna, which represents a dipole. To design a dipole antenna, the length of its arms should usually be close λ to , where λ is the operating wavelength of the antenna. The radiation pat2 terns that we get from dipole antennas are omni directional (doughnut shape) around the radiating structure. Dipole antenna basics have been treated extensively in literature and all governing mathematics can be found in [5–7]. When designing a dipole antenna, full wave analysis is followed to predict its characteristics. The method of moments (MoM), finite-difference time domain (FDTD), or finite element method (FEM) is used in the design and modeling of printed dipole antennas. As an example, we would like to design a printed dipole antenna to operate at a 2.45-GHz center frequency for WLAN applications on a commercial duroid dielectric substrate with εr = 2.2 and 1.56-mm thickness. One-halfounce copper is used with a 17.5-µm thickness. In this example, we will use HFSS as the modeling and simulation tool. HFSS is a FEM-based full wave field solver that is widely used in industry and academia. The wavelength of such an antenna in free space is λ2.45GHz = 12.25 cm, and its effective waveλ λ = 8.259 cm. For a 2 2.2 λ λ dipole, the length of each arm will be approximately (i.e., for both arms). 4 2 In addition, the microstrip sheet should have a width. We will use a width of 1.23 mm, and a small gap between the two arms of also 1.23 mm. The signal source will be applied within this gap (i.e., the feeding point from a transmission line or balun). Figure 2.16 shows the geometry of the printed antenna designed. After defining some rough number, the model is created in HFSS and the length of the antenna is tuned so that it will resonate at the desired frequency. λ A parametric sweep is run in the vicinity of the expected length of the 2 dipole to identify the best antenna length given the provided structure and microstrip wire parameters. Figure 2.17 shows the reflection coefficient curves (|S11|) for arm lengths between 45 and 52 mm. It is obvious that the best resonance at the desired frequency will be obtained when the dipole length (including the gap) is 49.16 mm. The figure also shows that the longer the dipole length, the lower the resonant frequency becomes, and vice versa. length due to the presence of the substrate is λeff =

38

Printed MIMO Antenna Engineering

Figure 2.16  Geometry of the printed dipole.

Figure 2.17  Reflection coefficient curves as a function of the dipole length.



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39

The obtained 3D gain pattern is shown in Figure 2.18(a), and the 2D gain pattern in the elevation direction at an azimuth of 90° (y-z plane) is shown in Figure 2.18(b). The omnidirectional shape is obvious for this WLAN dipole. Before deciding that this antenna is ready for fabrication, other parameters need to be investigated, particularly its radiating efficiency. For this designed antenna, the radiating efficiency was almost 96% (remember that there are no impedance mismatches). Dipole antennas for wireless and handheld devices are widely used. Several types of dipole antennas have been created to miniaturize the size [33–36], increase the BW [37, 38], and provide adjacent resonances and the like [33, 36, 39, 40]. In [33] the miniaturization of a dual-band printed dipole antenna was accomplished using a spiral structure for the arms. The number of turns in the spiral determines the resonance frequencies. This is also a dual-band structure covering the two WLAN standard bands of 2.4 and 5.8 GHz. Figure 2.19 shows the structure of this antenna. Notice that the back side is also constructed

Figure 2.18  Gain patterns of the designed printed dipole antenna at 2.45 GHz: (a) 3D gain pattern and (b) elevation pattern for φ = 90°.

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Printed MIMO Antenna Engineering

Figure 2.19  Miniaturized printed dipole with spiral arms. (From: [33], © 2005 IEEE. Reprinted with permission.)

as a spiral (i.e., the bottom spiral is one arm and the top one is the other). The spacing between the top and bottom spirals can be used to control the operating BW. The operating BW in the lower band was 400 MHz; in the higher one it was 600 MHz. The maximum gain obtained at 5.8 GHz was 2.7 dBi. Meandering the two dipole arms is considered a good miniaturization technique for printed antennas. A printed meandered dipole for radio-frequency identifier (RFID) readers operating at 892 to 990 MHz was proposed in [34]. The board size was 90 × 9 mm2 on an FR-4 substrate material. The maximum gain obtained was 4.7 dBi. In [35] a compact wideband printed dipole antenna with multiple loops was proposed for digital TV applications that covered the bands from 430 to 1180 MHz. The maximum gain in the UHF band was approximately 2.7 dBi. A new printed dipole miniaturization method that is based on metamaterial (MTM) loading is presented in [36]. A negative refractive index transmission line (NRI-TL) metamaterial is used to load the printed dipole antenna structure for antenna miniaturization and multiple resonances. The antenna covers the bands centered at 1.15, 2.88, and 3.72 GHz, with a BW of 37 and 1150 MHz, for the first and second and third resonances combined. The antenna size was 0.19λ° × 0.04λ° × 0.003λ° at 1.15 GHz. Figure 2.20 shows the proposed miniaturized dipole. The maximum gain and efficiency at 2.88 GHz were 3.26 dBi and 95.6%, respectively. Two broadband techniques are presented in [37, 38]. The first [37] utilized a tapered slot feed along with a parasitic element for broadbanding. The dipole, the slot, and the parasitic element are printed on the top layer, while the tapered feed is printed on the bottom layer. The antenna covers a range from 3.1 to 10.6 GHz and is 71 × 86 mm2. In [38] a three-layer printed dipole is presented that has both wideband operation and low cross-polarization radiation.



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41

Figure 2.20  Miniaturized printed dipole loaded with NRI-TL metamaterial loading with multiband operation: (a) top view, (b) bottom view, and (c) reflection coefficient curves. (From: [36]. © 2012 IEEE. Reprinted with permission.)

Aperture feeding is used within the middle layer. The antenna covers the band from 2.7 to 4.7 GHz. The maximum measured gain was approximately 7 dBi. Multiple resonances can be achieved using several methods such as NRITL as in [36] or the spiral arms method from [33]. Other techniques include introducing slots within the arms of the printed dipole as in [39] or using metamaterial unit cell loading as in [40]. A printed dipole with a single arm length of 23 mm and width of 4 mm was loaded with a U-shaped slot so that it would resonate at 2.4 and 5.2 GHz for WLAN applications in [39]. The maximum

42

Printed MIMO Antenna Engineering

gain was achieved at around 2.44 GHz with 4.1 dBi loading the dipole arms, with a MTM unit cell that allowed the antenna to resonate at 0.94 to 1.7 GHz as shown in [40]. The maximum gain was 1 dBi. The total size of the printed antenna was 28 × 85 mm2. The MTM unit cell size was 23.54 × 15.55 mm2. The methods presented in this section for printed dipole antenna miniaturization, broadbanding, and multiresonances are just examples of some current, widely used methods. These are not the only methods; others exist but we just wanted to list some of the methods from the literature. 2.4.2  Printed Monopole Antennas

A monopole antenna is constructed by considering only one arm of a dipole and ground plane beneath it. The signal source is connected to the monopole antenna and the return path is the ground plane. The length of the monopole (a wire base) should be around 0.25l to resonate at the desired frequency. Printed monopoles can be easily fabricated on a two-layer PCB. Several shapes have been proposed and investigated in the literature: meander lines [41–43], patches and squares [44, 45], and triangular [46], elliptical [47], e-shaped [48], and slot-based [49] shapes. Other configurations exist, but the previous list is just to show the reader the magnitude of the different geometries used to construct printed monopole antennas. Meander line monopoles are widely used because they provide miniaturization. Broadband and multiband operation can be achieved as shown in [41, 42] and [43–48], respectively. In [41], the meander line–based antenna was merged with a loop to provide a broadband of 845 MHz covering the standards from GPS 1.57 GHz up to WiBro at 2.4 GHz. The antenna size was 18.5 × 10 × 1.6 mm3 with a ground plane size of 40 × 50 mm2. The maximum measured gain was 3.66 dBi. Multiband operation was achieved in [43] by meandering a folded monopole. The antenna covered the bands from 919 to 1225 MHz and 1582 to 2313 MHz. The maximum achieved gain was 1.54 dBi. The total antenna size was 34 × 5.1 × 4 mm3. Printed monopole antennas are widely used in UWB applications. Slot antennas or slot loading was used to achieve UWB operation in [49] and [44], respectively. Also, a quasi-square patch was used to provide an operating BW from 3.9 to 21.4 GHz in [45]. The total antenna size was 11 × 16 mm2, and it provided a maximum gain of 5.85 dBi. Figure 2.21 shows the fabricated antenna and its VSWR and gain curves. 2.4.3  Printed Loop Antennas

Printed loop antennas are currently considered to be viable substitutes for printed inverted-F (PIFA) antennas in handheld devices due to their relatively small size and mainly because they do not cause excessive surface currents on



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43

Figure 2.21  Miniaturized printed UWB monopole antenna: (a) fabricated antenna and (b) its VSWR and gain curves. (From: [45]. © 2010 IEEE. Reprinted with permission.)

the ground plane beneath them, thus causing less interference with adjacent electronic components. A loop antenna resonates when its circumference length is around 1λ of the operating center frequency. But several techniques have been devised that provide loops resonating at 0.5λ [50, 51] or even 0.25l [52]. Wideband loop structures have been also proposed in [53–56]. In [50], a 0.5λ printed loop antenna for mobile handheld terminals was proposed. The antenna size occupied 50 × 14.5 mm2 from the 50 × 00 mm2 antenna board. An inner inverted-L strip is used to provide a second loop path for dual-band operation. The antenna covers the bands from 870 to 1150 MHz and 1700 to 1900 MHz when considering the 2.5:1 VSWR points (i.e., –7-dB BW). The outer loop strip has a length of 128 mm, thus supporting the 0.5λ mode at 900 MHz and 1λ mode at approximately 1100 MHz.

44

Printed MIMO Antenna Engineering

The maximum gain obtained at the 900-MHz band was 0.8 dBi and in the 1800-MHz band, it was approximately 4 dBi. Efficiencies of 50% and 78% were obtained for the lower and upper bands of operation, respectively. Figure 2.22 shows the configuration of this antenna as well as its resonant response. Note that the two adjacent bands of 900 and 1100 MHz form a wide band of about 280 MHz in width.  λ A quarter-wave   printed loop antenna was proposed in [52]. The  4 0.25λ printed loop was designed with an 85-mm metal strip length at 900 MHz and occupying 10 × 35 mm2 out of the 100- × 40-mm2 board. A microstrip matching circuit was used to reduce the inductive input reactance of this antenna configuration and provided impedance broadening at the higher frequency of operation. This antenna covers the 890–1050-MHz and 1650– 2250-MHz bands when considering the –6-dB BW (i.e., VSWR of 3:1). The measured efficiency was between 57% and 75% on the GSM and DCS/PCS/ UMTS bands. The maximum gain of the lower band was 0.8 dBi, and at the

Figure 2.22  Miniaturized printed loop antenna for mobile terminals: (a) antenna geometry and (b) measured and simulated return loss. (From: [50]. © 2007 IEEE. Reprinted with permission.)



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45

higher band, approximately 3.3 dBi. Figure 2.23 shows the antenna structure λ of this printed antenna. Figure 2.24 shows the return loss curves of the pro4 posed antenna. To gain a better understanding of the operation of the antenna, Figure 2.25 shows the current distribution on the surface of the antenna structure at three frequencies: 925, 1750, and 2100 MHz. Note the length of the current path in each configuration, which relates to the mode and resonant λ frequency under investigation. For example, the current path length at 925 4 MHz (D) is approximately 85 mm, corresponding to the mode. To obtain wideband operation in printed loop antennas, parasitic microstrip element loading [53] and periodic capacitive loading [55] techniques can be adopted.

Figure 2.23  Miniaturized printed loop antenna for mobile terminals: (a) complete antenna and board sizes and (b) close-up view on the antenna portion. (From: [52]. © 2009 IEEE. Reprinted with permission.)

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Figure 2.24  Return loss curves for the multiband printed antenna shown in Figure 2.23. (From: [52]. © 2009 IEEE. Reprinted with permission.)

2.4.4  Printed Patch Antennas

Patch antennas were among the first printed antennas. They consist of a rectangular (or circular) metallic patch on top of a dielectric material of very small thickness compared to the operating wavelength and a ground plane on the other side of the dielectric substrate. Figure 2.26 shows the geometry of a microstrip patch antenna with a microstrip line feed. Patch antennas are easy to design and integrate within planar circuits, are very reliable, and are inexpensive to manufacture. These antennas also conform to the surface on which they are printed. Patch antennas can be designed by finding the appropriate width and length sizes (for a rectangular one) as well as the location of the feed point. Some closed-form expressions for finding the dimensions of rectangular and circular patch antennas are presented in [5, 6]. These formulas are derived from transmission line and cavity models, and should be used to come up with rough design dimensions based on the material used and the designed frequency of operation. The feed location can affect the input impedance as well as the polarization of the antenna, and should be chosen carefully. The basic design equations for a rectangular patch antenna are given by (2.29) through (2.34):

εreff =

εr + 1 εr - 1  W + 1 + 12   2 2  h 

-1 2

, W h >1

(2.29)



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Figure 2.25  Current distribution at the various bands of operation and the current path for the loop antenna. (From: [52]. © 2009 IEEE. Reprinted with permission.)

(ε ∆L = 0.412 h (ε

reff



reff



)

W  + 0.3  + 0.264  h  W  - 0.258  + 0.8  h 

)

Leff = L + 2 ∆L

(2.30)

(2.31)

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Printed MIMO Antenna Engineering

Figure 2.26  Geometry of a microstrip-fed rectangular patch antenna.



f c TM 010 =



W =



L=

v0 2 (L + 2 ∆L ) εreff

(2.32)

v0 2 fr

(2.33)

2 εr + 1

v0 - 2 ∆L 2 f r εreff

(2.34)

where εr is the dielectric constant of the substrate, εreff is the effective dielectric constant due to the air on top of the patch, h is the thickness (height) of the substrate used, W is the width of the patch, L is the length of the patch, ∆L is the change in the length due to the fringing of the fields at the patch edges, and v0 is the speed of light. Note that if the dimensions of the patch are known, the resonant frequency of the dominant mode (i.e., TM010) can be found by applying (2.29) through (2.32). If the frequency of operation is known along with the material properties (i.e., εr and h), then (2.33) and (2.34) are used to come up with the patch dimensions. Once the patch dimensions are known, they are fine-tuned within the computer tool to resonate at the desired band. In addition, the feed location is optimized for proper matching. The radiation pattern of a patch antenna has a maximum at broadside, that is, normal to the patch. The BW of patch antennas is usually narrow and it is directly proportional to the thickness of the dielectric substrate and inversely proportional to the dielectric constant of the substrate. The efficiency of a patch



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49

antenna is inversely proportional to the dielectric constant of the substrate and is inversely proportional with the substrate height [57, 58]. Figure 2.27 shows a typical gain pattern from a patch antenna designed and optimized using a 3D field solver (HFSS). The patch dimensions were L = 40.5 mm, W = 48.4 mm, and the microstrip feed was used in the middle of the patch edge with an inset feed length of 12.37 mm and inset feed gap of 2.4 mm (for impedance matching). The feed line width was 4.85 mm and the substrate

Figure 2.27  3D gain pattern (a) and reflection coefficient (b) of a rectangular microstrip-fed patch antenna with L = 40.5 mm and W = 48.4 mm on a duroid substrate with εr = 2.2 and h = 1.56 mm.

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size (as well as the ground [GND]) was 121 × 82 mm2. The resonant frequency of this WLAN patch was 2.45 GHz with a maximum gain of approximately 7.7 dBi. The antenna efficiency was 95.5%. A summary of the effect of the substrate material and thickness on the patch size, BW, and efficiency is shown in Figure 2.28. The effect of the substrate material on the size of the patch is shown in Figure 2.28(b). It is clear that higher dielectric materials can be used to miniaturize (shrink) the size of patch antennas. But this will cause degradation in the antenna efficiency as shown in Figure 2.28(c). Thick substrates can be used to increase the operating BW of patches, but at the expense of lower efficiency. Thus, the trade-off should be carefully investigated when choosing design parameters. A large amount of work has appeared in literature that addressed the trade-offs between efficiency, BW, and miniaturization. Also, different feeding mechanisms were proposed to enhance the performance of one or more metrics of interest. The two widely used feeding mechanisms for printed patch

Figure 2.28  Patch size as a function of (a) dielectric constant (b) operating frequency and the path efficiency as a function of substrate height (c) and its bandwidth (d). (From: [58]. © 2006 IEEE. Reprinted with permission.)



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antennas are the probe feed and the microstrip feed [59]. These feeding structures provide narrower operating BW compared to proximity and slot coupled feeds [60, 61], but are easier to manufacture and fabricate. Patch antennas can be miniaturized by introducing shorting posts at one of the radiating edges, yielding an approximately 50% reduction in size in that dimension [62–64]. The performance of an 1800-MHz shorted patch for mobile communication handsets was presented in [63]. The peak gain of the shorted patch was approximately 3.24 dBi and its operating bandwidth was 180 MHz, which corresponds to about 10%. In [64], another miniaturization technique called folding was introduced along with shorting edges to reduce the λ overall size of the patch to approximately 0 (see Figure 2.29). The prototyped 8 antenna operated at 2.4 GHz with 4% operating bandwidth. The antenna efficiency was 90% with a maximum gain of approximately 2 dBi. Another miniaturization technique utilizes magneto-dielectric and ceramic substrates [65–66]. It was derived in [65] and showed that to enhance the operating BW of patch antennas, one can use a magneto-electric substrate (ferrite loaded or film laminates) to lower the quality factor of the resonating patch and thus widen the bandwidth with minor degradation in efficiency, contrary to using purely high dielectric constant substrates that yield a smaller antenna but give rise to surface waves that degrade the efficiency and complicate the matching to the antenna. This comes at a higher cost when using such magnetic substrates. Metamaterial loading can be used for patch antenna miniaturization as demonstrated in [67]. A miniaturization factor of 6.45 was achieved for a patch antenna resonating at 250 MHz. MTM loading provides good miniaturization performance but significantly degrades the antenna efficiency. The reported ef-

Figure 2.29  Development of a folded-shorted patch: (a) conventional patch, (b) conventional shorted patch, (c) folding of a shorting patch, and (d) folded-shorted patch. (From: [64]. © 2004 IEEE. Reprinted with permission.)

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ficiency of the miniaturized patch was 19.8%, with a maximum gain of –4 dBi and a bandwidth of only 0.83%. The use of fractal geometrics provides another way to miniaturize as discussed in [68]. Giuseppe-Peano fractal geometry was used, which increased the patch perimeter. This in turn reduced the resonance frequency of the patch without any increase in its area size. The frequency bandwidth was also increased using the miniaturization method when using a slit or the patch surface and an air gap between. A minimum gain of 4.5 dBi was achieved at 2.5 GHz with 73% efficiency. Although the size of the antenna needs to be miniaturized to fit within user handheld terminals, its single resonance for the primary excitation mode and its relatively narrow bandwidth make it unattractive for mobile terminals that need to support wide bandwidths and multiple standard frequency bands. Thus bandwidth enhancement methods and multiple resonance techniques have attracted a lot of attention for practical use of antennas in wireless terminals and access points. Some dual-band and wideband antennas are presented in [69–74]. The introduction of slots within the patch is one common technique for multiband and wideband operation as shown in [69, 71, 73]. In [71], for example, the effect of multiple U-slots or the multiresonance behaviors of a rectangular patch were investigated. For each U-slot introduced, a new band is introduced with slight effect on the gain of the patch. Table 2.1 summarizes some of the miniaturization techniques and their effects on patch antenna performance. These techniques can also be applied to other microstrip and printed antenna types.

Table 2.1 Microstrip Patch Antenna Miniaturization Techniques and Their Effect on Antenna Performance Technique Effect 1. Shorting posts Approximately 50% reduction in antenna size, with increase in BW and decrease in efficiency. 2. Folding 50% reduction in antenna size, narrower BW, and good efficiency. 3. Using magneto-dielectric materials Size reduction with enhanced BW, minor efficiency (substrates) degradation with high manufacturing cost. 4. Using metamaterials Miniaturization factors of almost 6 times less, very low efficiencies, and narrow BW. 5. Using slits within the patch structure Increase BW, multiband operation, and minor efficiency degradation. 6. Stacked patch geometries Wider BW, higher cost, and good efficiency. 7. Parasitic element loading Wider BW and an increase in gain if the separation exceeds 0.3λ.



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2.4.5  Printed Inverted-F Antenna

Printed inverted-F (PIFA) antennas are widely used in mobile terminals nowadays due to their multiband operation, ease of design and fabrication within a mobile terminal housing, and reduced back lobe radiation towards the user’s hand. A typical PIFA antenna is shown in Figure 2.30. It consists of a large ground plane on one side of a dielectric medium, and a patch with a feed and a short. In a side view of the antenna, the shape look like an inverted F. PIFA antennas are usually used for multiband operation and coverage. This can be accomplished by devising two or more resonant paths for multiband coverage or using different resonant modes for a single resonant path. This can be observed from the PIFA presented in Figure 2.31. This PIFA resonates at 950 and 1790 MHz with an operating bandwidth of 25 and 123 MHz in the two bands, respectively [75]. The introduction of the slit in the radiating patch allowed for two current paths at the two resonant frequencies as shown in Figure 2.31(b). The starting equation for designing a PIFA with a single resonance is given as

L +W =

λ 4 εr

(2.35)

where L and W are the length and width of the patch, respectively; λ is the operating wavelength; and εr is the dielectric constant of the substrate. Equation (2.35) shows that the sum of the lengths and width should approximately equal to quarter of the effective wavelength. The design parameters should be optimized to tune for the desired operating bands.

Figure 2.30  Geometry of a printed/planar inverted-F antenna (PIFA).

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Figure 2.31  Dual-band PIFA antenna: (a) geometry and (b) current distribution/path at the two bands. (From: [75]. © 2002 Wiley. Reprinted with permission.)

Several techniques have been proposed for multiband operation and broadbanding of PIFAs such as those shown in [76–81]. A multiband printed PIFA utilizing ground plane parasitic resonators was proposed in [76]. The antenna covers the GSM 900 and the DCS 1800 bands. The antenna size was 15 × 49.5 × 9.5 mm3. The gain over the operating bands had a variation of 3.3 dBi and was always above –1 dBi. In [77], a complete vehicular antenna that covered the GPS band as well mobile telephone bands of 900 and 1800–2100 MHz was proposed. The antenna consisted of a patch and a dual-band PIFA. The radome was designed to cover both antennas and to be placed on the roof of a car. Figure 2.32 shows a fabricated photo of this antenna. The PIFA board was 65 × 20 × 15 mm3. A –0.2-dBi gain was measured at 900 MHz, and a



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Figure 2.32  Vehicular multiband PIFA antenna with case. (From: [77]. © 2005 IEEE. Reprinted with permission.)

–3.6-dBi gain was measured at 1795 MHz. The measured radiation patterns on the roof of the car are shown in Figure 2.33 and they show good azimuthal coverage. A triband PIFA using dual shorting posts was proposed in [79]. The antenna covered the 2-, 3.7-, and 5-GHz bands with a total size of 40 × 40 × 3.75 mm3. The achieved percentage bandwidths for the three bands were 11%,

Figure 2.33  Measured azimuthal gain patterns of the vehicular multiband PIFA on a car roof top. (From: [77]. © 2005 IEEE. Reprinted with permission.)

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8.8%, and 10%, respectively. The measured gains in the three bands were 2.05, 2.32, and 3.47 dBi, respectively, with a minimum efficiency of 60%. A coupled-fed wideband PIFA was shown in [80] with 305 and 1040 MHz for a –6-dB BW for the lower band at 800 MHz and the upper band at 2.2 GHz, respectively. The antenna consisted of a driven T-shape monopole and two coupled folded strips. The minimum efficiency was 45% at 750 MHz. The total size of the antenna was 15 × 60 × 80.8 mm3. Bandwidth broadening techniques for PIFA antennas can be achieved by these means: 1. Reduce the ground plane size and introduce slits to reduce the quality factor and thus provide a wider bandwidth. 2. Use thick substrates. 3. Use parasitic resonators near the resonance of the antenna to widen its bandwidth. 4. Excite the multimodes close to one another. Note that most PIFAs are elliptically polarized to allow for receiving both vertically and horizontally polarized signals indoors when depolarization is common. PIFA antenna miniaturization is possible via spiraling the radiating arm as shown in [82]. The spiraling of the PIFA is shown in Figure 2.34. The spiraling was shown to even lower the resonance of the higher band of operation from around 6 to 5 GHz, while the lower band was not changed at 2.5 GHz.

Figure 2.34  Spiraling the PIFA radiating arm for antenna miniaturization. (From: [82]. © 2007 IEEE. Reprinted with permission.)



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The efficiency was degraded by 5% from 71% to approximately 67% at 2.5 GHz. The reduction in antenna size was 50% using this method. The antenna occupied 9.5 × 6.5 mm2 on an 0.8-mm FR-4 substrate. The ground plane was 46 × 45 mm2. The maximum gains were 3 and 0 dBi for the two bands, respectively. An isolation technique from adjacent elements using parasitic strips was proposed in [83] for a PIFA antenna. The presence of conducting objects near the radiating element will degrade its return loss, and thus this work shows that two small shielded strips placed on both sides of the radiating PIFA can minimize and almost isolate its return loss from closely placed conducting elements. The antenna was operating in the wireless LAN band of 2.4 GHz with a 145-MHz bandwidth. Some practical examples of multiband PIFA antennas are shown in Figure 2.35 [84]. Three Nokia mobile phone models and their respective PIFA elements are shown. Two antennas are shown in each model. Several slots were used in the lower band GSM antenna (larger one) to allow for multiple resonances [Figures 2.35(a) and (b)] and also allowed for closer separation between the GSM and 3G antennas compared to Figure 2.35(c) where only one slot was introduced. 2.4.6  Other Printed Antenna Structures

In the previous sections, we discussed the most common printed antenna structures that are being utilized in mobile and portable wireless devices. These antenna structures are also used in printed MIMO antenna systems as will be shown in the coming chapters.

Figure 2.35  Mobile phone PIFA antennas: (a) Nokia 6680, (b) Nokia 6630, and (c) Nokia E60. (From: [84]. © 2012 IEEE. Reprinted with permission.)

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Some derivatives of the previously presented structures or other novel printed geometries also exist in the literature. F-shaped printed structures are presented in [85] and T-shaped structures are shown in [86–87]. An L-shaped geometry is presented in [88]. All of these geometries are presented as singleband structures resonating above 1 GHz, except for [87] where a dual resonance behavior is shown. A 4-shaped antenna geometry with dual-band operation covering the 800-MHz band and 2.4 to 2.6 GHz of the LTE standard is presented in [24, 89] for 2 × 2 and 4 × 4 MIMO configurations. A Z-shaped printed antenna is presented in [90] for single-band operation at 2.48 GHz for WLAN applications. All of the aforementioned geometries can be good candidates for different wireless standards based on their respective features. They are compact in size and printed in nature. But the designer needs to investigate the properties of each to determine if it satisfies the design requirements. Reconfigurable printed antennas are becoming an attractive pick for software defined radio (SDR) applications and cognitive radios. Some printed reconfigurable antennas were proposed in [91, 92]. In [91] a single reconfigurable printed antenna is presented covering four bands in the 540–890-MHz frequency range. The four bands are chosen according to solid state switch (PIN diodes) combinations that vary the electrical length of the antenna and thus its resonance frequency. Another reconfigurable dual-band antenna that covered two bands simultaneously was presented in [92]. The lower band covered between 1.1 and 1.5 GHz according to the bias voltage value, while the second band covered from 1.7 up to approximately 2.8 GHz. The voltage tuning values ranged from 0V to 30V, which limits the use of some bands/configurations in a real design. The gain patterns at the different bands were consistent.

2.5  Antenna Integration Effects In all wireless devices, the printed antennas are surrounded by the casing of the device, which will interact with the radiated fields and have a noticeable effect on antenna performance. In addition, the presence of the hand and head of the user has a direct effect on the antenna efficiency and radiated power. In this section, we discuss such effects in detail so that antenna designers are aware of these effects on the antennas they are designing. 2.5.1  Effect of PCB Ground Plane

Printed antennas are either placed on the top portion of a dielectric back plane with the system GND plane below it, or are placed on top of the GND plane as shown in Figure 2.36. Depending on the structure and the radiation mechanism of the printed antenna, the size of the GND plane can affect the resonance



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Figure 2.36  Antenna locations with respect to the mobile handset GND plane: (a) above the GND plane (no metal on the bottom side of the PCB under the antenna) and (b) over the GND plane (GND place covers the bottom layer of the antenna).

and impedance bandwidth of the printed antenna in addition to its radiation characteristics. Formulas describing the behavior of patch antennas are based on an infinite ground plane assumption. In practical applications, finite GND planes are used, and their dimensions are a few multiples of the antenna size itself. Thus, the effect of the GND plane size is important to characterize for any printed antenna design. Reference [93] showed that a finite GND plane size affects the input impedance as well as the gain/radiation performance of a circular patch. Increasing the GND plane size reduced the input impedance of the antenna and provided higher gain values with a maximum when the GND plane radius approached 0.63λ0. The effect of a finite ground plane was also seen to have provided narrower HPBW as prescribed in [94]. A 35o narrower HPBW was observed when the width of the PCB used for a square patch was 1λ0 wide. Smaller GND planes lower the impedance bandwidth of printed antennas, especially at the lower frequency bands (i.e., 900 MHz) [95]. In addition, certain antenna types, such as printed monopoles and dipoles, are more affected by GND plane sizes compared to other printed antenna types due to the large surface cements that are induced, which makes the GND planes with typical sizes of λ/4 at the lower frequency bands of 800/900 MHz become radiating elements (complementary to the radiating elements used), thus becoming part of the antenna structure. Thus, careful design and analysis are required to balance these factors for a certain design.

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The efficiency of an antenna is also affected by the size of the GND plane. The efficiency of printed antennas increases as the width of the GND plane is increased for an UWB monopole antenna [96]. The resonance frequency at the lower band was also decreased with the increase in the GND plane length. The coupling between the antenna and the GND plane usually occurs through electric fields (i.e., capacitive). Increasing the coupling will increase the impedance BW and the resonant frequency [97, 98]. For spiral-type antennas, the coupling is magnetic (i.e., inductive). To reduce the effect of the GND plane on antenna performance, antenna loading and introducing slots in the GND plane are used [96, 98]. 2.5.2  Effect of Mobile Terminal Casing

Printed antennas in mobile terminals are covered with the device package, or case, which is usually a plastic material. Structures covering antennas are usually called radomes (from radar dome) [99]. These radomes affect the radiation as well as the resonant behavior of antennas when they are placed in proximity to or touching the antennas. Loading an antenna with a plastic material will shift its resonance frequency toward lower bands and will affect its impedance BW. Figure 2.37 shows the frequency shifts due to the plastic carrier and plastic covers of iPhone 3G handsets [84]. The efficiency of the antenna can be slightly improved (by 2% to 10%) if plastic casing or rings are used. The use of metal- or chrome-plated covers or casings degrades the efficiency of antennas significantly. 2.5.3  Effect of the Presence of User’s Hand and Head

The effect of the user’s hand and head, which are usually in proximity to the user terminal (especially for mobile phones), has been investigated for various types of antennas in several works in the literature [84, 100–106]. The hand and head of the user are high dielectric structures that directly affect antenna performance when in close contact or placement. The presence of a hand or head near the radiating antenna will load it with different impedance values. The impedances will dissipate part of the radiated power and lower the antenna efficiency. The total radiated power (TRP) is a measure of the RF transmitting performance of the mobile terminal. It can be found from [84]:

TRP ≅

1 p 2p EiRPq ( q, f) + EiRPf ( q, f) sin qd qd f 4 p q∫= 0 f∫= 0 

(2.36)



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Figure 2.37  Effect of phone cover on the resonant frequency of an iPhone 3G cellular phone. (From: [84]. © 2012 IEEE. Reprinted with permission.)

where EiRP is the time arranged radiated power across the spherical surface enclosing the antenna. The presence of the user hand close to the antenna (while holding the mobile phone) will significantly degrade its TRP as well as its efficiency. The closer the hand gets to the antenna (i.e., covering it and touching the plastic case), the higher the power loss and, hence, the lower the efficiency.

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Table 2.2 shows the measurement results of the TRP of an iPhone for the four different hand gripping positions shown in Figure 2.38 compared to a nohand scenario (position 0, P0). Position 4 (P4) provided the worst TRP because the hand is close to the antenna located in the lower portion of the phone. The efficiency can be degraded by up to 40% to 50% if the hand touches the antenna or covers it completely as shown in [103–105]. Plots for the MEG and the efficiency curves of a diversity antenna system consisting of a whip antenna and a PIFA as a function of the antenna length and the presence of a human shoulder plots are shown in Figure 2.39 [104]. A large increase in the MEG values due to the presence of the human shoulder are observed. This is because at 900 MHz, the shoulder acts as a reflector that is λ/4 to λ/2 away from the antenna. Also, the higher the elevation angle of the incident waves (mv and mH), the larger the MEG values. In Figure 2.39, σV and σH are the variances of the vertical and horizontal wave components, D is the distance between the antenna and the head, and α is the tilt angle of the phone with respect to θ = 0. The efficiency on the other hand was not affected as much. It is clear that MEG is highly dependent on the antenna type. The correlation coefficient is increased due to the presence of the hand as it affects the relative phases of the patterns from the different antennas in a diversity system, thus degrading its performance. The head effect is less severe when it comes to detuning, power loss, or efficiency. Its effect has to be considered though because it will load the antenna when in proximity, which will degrade its performance. A specific metric that is usually evaluated in the presence of the head is the specific absorption rate (SAR). SAR is a fundamental parameter that is used to describe the health risks of electromagnetic radiation on human tissues. It measures the amount of power absorbed per unit mass of tissue [106]. SAR is calculated from the tissue characteristics and the incident E-field according to

Table 2.2 Effect of Hand Gripping Positions on the TRP and Efficiency of an iPhone 3G Handset TRP-GSM P0 P1 P2 P3 P4 iPhone 2 27.5 dB –8.7 dB –9.0 dB –4.2 dB –8.3 dB iPhone 3G 27.3 dB –8.2 dB –9.8 dB –3.5 dB –8.1 dB iPhone 4 28.5 dB –6.9 dB –8.8 dB –4.1 dB –11.6 dB TRP-3G P0 P1 P2 P3 P4 iPhone 3G 19.0 dB –7.6 dB –9.7 dB –3.4 dB –8.5 dB iPhone 4 20.6 dB –7.2 dB –10.2 dB –4.0 dB –11.1 dB From: [84]. © 2012 IEEE. Reprinted with permission.



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Figure 2.38  Effect of hand gripping positions on TRP. (From: [84]. © 2012 IEEE. Reprinted with permission.)



SAR =

σ 2 E 2ρ

(2.37)

where σ is the tissue conductivity, and ρ is the material density. SAR values averaged over 30-min intervals for a 1g mass of tissue should be less than 1.6 mW/g according to the ANSI and IEEE standards. Most current mobile phone designs require more characterization and careful design to pass the various requirements and to account for the various parameters that directly affect their performance.

2.6  Antenna Arrays Antenna arrays are groups of identical antenna elements arranged in a certain geometry to provide an enhanced gain performance and a more directive one. Antenna arrays are used in cell phone base stations, radar systems, airplanes, jet fighters, and so forth. They are widely used to:

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Figure 2.39  (a) Radiation efficiency and (b, c) MEG for shoulder and nonshoulder models at α = 60° and D = 2 cm. (From: [104]. © 2001 IEEE. Reprinted with permission.)

1. 2. 3. 4.

Increase the directivity and gain of antennas; Narrow the HPBW; Combat noise and increase the SNR; Change the maximum value of the radiation pattern location via beam steering; 5. Eliminate signals coming from predefined locations via null steering. The gain value and the direction of the radiation beam from an antenna array can be controlled by one of the following methods: 1. Geometric configuration of an array (linear, circular, planar, etc.);



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2. Relative displacement between elements (interelement spacing); 3. Excitation amplitude of different elements (adopting various amplitude distributions such as binomial, Dolph-Tschebyscheff, etc.); 4. Excitation phase of different elements (interelement phase excitation for beam steering, null steering, etc); 5. Type (relative pattern) of individual elements. Several reference books handle the topic of antenna arrays extensively such as [5–7]. In this section, we present some formulations for the linear array pattern expressions as a function of the number of elements, interelement spacing, and phase excitation. This will give the reader a good feel for how the field is found, and can supply and understanding of the more complex geometries accordingly. Consider the linear array of n-isotropic elements shown in Figure 2.40. The resulting pattern for this array will be the resultant sum of the individual patterns from each radiator. Taking into account the relative delays in the received/transmitted waves between the various elements with respect to a reference one, the sum of the various fields can be written as

E = 1 + e j ψ +  + e j (n -1) ψ

(2.38)

where



ψ=

2 pd cos q + δ λ

Figure 2.40  A linear array of isotropic radiators.

(2.39)

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and where d is the spacing between adjacent elements, θ is the angle between the axis of the linear array and broadside, δ is the phase difference in the excitation between two adjacent elements, and n is the number of elements in the array. We can rewrite ψ as ψ = d r cos q + δ



(2.40)

If we multiply (2.38) by e jψ, we get Ee j ψ = e j ψ + e j 2 ψ + e j 3 ψ +  + e j (n ) ψ



(2.41)

Subtract (2.38) from (2.41) and divide by (1 – e jψ) to get

jn ψ

jn ψ 2

1-e e = jψ jn ψ 1-e e

  ψ  sin  n   e jn ψ 2 - e - jn ψ 2   2 jζ  = e   (2.42)  e jψ 2 -e -jψ 2     sin  ψ    2  



E=

where ζ =

(n - 1) ψ . The expression in (2.42) is called the array factor of a linear

2

2 antenna array. The normalized E of this linear array is given by



En =

E E max



(2.43)

For this linear array structure, the beam pattern can be controlled by varying the spacing between the adjacent elements (d) or the phase of the driving current (δ). The spacing between adjacent elements is not to exceed λ/2 in all designs, because such spacing would increase the sidelobe levels and generate grating lobes. Figure 2.41 shows the change in the beam direction as a function of the excitation phase δ for three cases. The first case, shown in Figure 2.41(a), shows that an excitation phase of 104.4° between two adjacent elements (progressive phase) will move the beam maximum toward 45°. The figure shows the elevation cut at any azimuth angle. Figure 2.41(b) shows that a progressive phase of 120° directs the beam towards 90° in the elevation plane. Figure 2.41(c) shows that a progressive phase of 138.6° directs the beam towards 135°. The array presented in these figures consists of eight elements spaced λ/2 from one another.



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Figure 2.41  Effect of the excitation phase on the radiation pattern of a linear antenna array of eight-element isotropic radiators: (a)�q0 = 45°, (b)�q0 = 90°, and (c)�q0 = 135°.

To steer the beam in a certain θ direction, you should find the corresponding δ by solving the following equation:

0 = d r cos q0 + δ

(2.44)

In Figure 2.42, the spacing between adjacent elements was varied from 0.2λ to 0.6λ in 0.2λ steps. Observe the broadening of the beam and also the lower directivity as the elements come closer to each other as in Figures 2.42(a) and (b). Also, observe the grating lobe (toward 180o) when the spacing exceeds 0.5λ as shown in Figure 2.42(c). In this figure, the progressive phase was set to direct the beam toward θ0 = 45°. The examples presented in Figures 2.41 and 2.42 were based on isotropic radiators (radiators that emit equal energy everywhere, i.e., with a sphere

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Figure 2.42  Effect of interelement spacing d on the radiation pattern of a linear antenna array of eight-element isotropic radiators: (a) d = 0.2λ , (b) d = 0.4λ, and (c) d = 0.6λ.

radiation pattern; these radiators do not exist, but are used as a reference). If we have a certain type of antenna—dipole, patch, and so on—we can easily obtain the beam pattern for that array by multiplying the antenna pattern of a single element, that is, the dipole radiation pattern, with the array pattern obtained from a geometry and configuration based on isotropic elements to get the total array response. This procedure is called the principle of array multiplication. This principle assumes that the mutual coupling between the adjacent elements is very low. But to get a more accurate array pattern, one should use a full wave field solver that takes all effects into account and creates the expected patterns for an actual design geometry of an antenna array. The unit excitation active element method can be used, and it incorporates the mutual coupling between adjacent elements [107].



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2.7  Conclusions In this chapter, we covered all of the fundamental parameters that need to be evaluated for a MIMO antenna system. We started with the conventional S-parameters and the radiation patterns, and then moved on to the specific MIMO antenna performance parameters such as the TARC, correlation coefficient, MEG, diversity gain, and channel capacity. The governing equations and the methods of finding these quantities were addressed with some actual examples from the literature. We also covered several widely used and recently proposed printed antenna element structures that can be good candidates for printed MIMO antenna systems. System-level integration effects such as the effect of the mobile terminal case and the effect of the hands and head on the performance of printed antennas were presented to give the designer and student a better feel for such effects, which should not be ignored when designing a real product. This chapter covered the fundamentals that a MIMO antenna system designer must be aware of in order to create a successful design.

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[58] M. S. Sharawi, “Use of Low-Cost Patch Antennas in Modern Wireless Technology,” IEEE Potentials Magazine, pp. 35–38, 47, July/August 2006. [59] A. K. Bhattaacharyya, “Long Rectangular Patch Antenna with a Single Feed,” IEEE Trans. on Antennas and Propagation, Vol. 38, No. 7, pp. 987–993, 1990. [60] C. C. Huang and T. H. Chu, “Radiating and Scattering Analyses of a Slot-Coupled Patch Antenna Loaded with a MESFET Oscillator,” IEEE Trans. on Antennas and Propagation, Vol. 43, No. 3, pp. 291–298, 1995. [61] M. Veysi, M. Kamyab, and A. Jafargholi, “Single-Feed Dual-Band Dual-Linearly-Polarized Proximity-Coupled Patch Antenna,” IEEE Antennas and Propagation Magazine, Vol. 53, No. 1, pp. 90–96, 2011. [62] R. Waterhouse, “Small Microstrip Patch Antenna,” Electronics Letters, Vol. 31, No. 8, pp. 604–605, April 13, 1995. [63] J. T. Rowley and R. B. Waterhouse, “Performance of Shorted Microstrip Patch Antennas for Mobile Communications Handsets at 1800 MHz,” IEEE Trans. on Antennas and Propagation, Vol. 47, No. 5, pp. 815–822, 1999. [64] R. Li et al., “Development and Analysis of a Folded Shorted-Patch Antenna with Reduced Size,” IEEE Trans. on Antennas and Propagation, Vol. 52, No. 2, pp. 555–562, 2004. [65] P. M. T. Ikonen et al., “Magnetodielectric Substrates in Antenna Miniaturization: Potential and Limitations,” IEEE Trans. on Antennas and Propagation, Vol. 54, No. 11, pp. 3391– 3399, 2006. [66] J. S. Kula et al., “Patch-Antenna Miniaturization Using Recently Available Ceramic Substrates,” IEEE Antennas and Propagation Magazine, Vol. 48, No. 6, pp. 13–20, 2006. [67] K. Buell, H. Mosallaei, and K. Sarabandi, “A Substrate for Small Patch Antennas Providing Tunable Miniaturization Factors,” IEEE Trans. on Microwave Theory and Techniques” Vol. 54, No. 1, pp. 135–146, 2006. [68] H. Oraizi and S. Hedayati ,”Miniaturization of Microstrip Antennas by the Novel Application of the Giuseppe Peano Fractal Geometries,” IEEE Trans. on Antennas and Propagation, Vol. 60, No. 8, pp. 3559–3567, 2012. [69] D. Sanchez-Hernandez and I. D. Robertson, “Analysis and Design of a Dual-Band Circularly Polarized Microstrip Patch Antenna,” IEEE Trans. on Antennas and Propagation, Vol. 43, No. 2, pp. 201–205, 1995. [70] K. L. Lau and K. M. Luk, “A Wide-Band Circularly Polarized L-Probe Coupled Patch Antenna for Dual-Band Operation,” IEEE Trans. on Antennas and Propagation, Vol. 53, No. 8, pp. 2636–2644, 2005. [71] K .F. Lee et al., “On the Use of U-Slots in the Design of Dual-and Triple-Band Patch Antennas,” IEEE Antennas and Propagation Magazine, Vol. 53, No. 3, pp. 60–74, 2011. [72] P. H. Rao, V. F. Fusco, and R. Cahill, “Wide-Band Linear Circularly Polarized Patch Antenna Using a Printed Stepped T-Feed,” IEEE Trans. on Antennas and Propagation, Vol. 50, No. 3, pp. 356–361, 2002. [73] S. J. Lin and J. S. Row, “Monopolar Patch Antenna with Dual-Band and Wideband Operations,” IEEE Trans. on Antennas and Propagation, Vol. 56, No. 3, pp. 900–903, 2008.

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[74] Y. Sung, “Bandwidth Enhancement of a Microstrip Line-Fed Printed Wide-Slot Antenna with a Parasitic Centre Patch,” IEEE Trans. on Antennas and Propagation, Vol. 60, No. 4, pp. 1712–1716, 2012. [75] F. Hsiao et al., “A Dual-Band Planar Inverted-F Patch Antenna with a Branch-Line Slit,” Microwave and Optical Technology Letters, Vol. 32, No. 4, pp. 310–312, February 20, 2002. [76] B. S. Izquierdo, J. Batchelor, and R. Langley, “Multiband Printed PIFA Antenna with Ground Plane Capacitive Resonator,” Electronics Letters, Vol. 40, No. 22, pp. 1–2, October 28, 2004. [77] R. Leelaratne and R. Langley, “Multiband PIFA Vehicle Telematic Antennas,” IEEE Trans. on Vehicular Technology, Vol. 54, No. 2, pp. 477–485, 2005. [78] K. Wong et al., “A Small-Size Penta-Band WWAN Antenna Integrated with USB Connector for Mobile Phone Applications,” Proc. Int. Conf. on Applications of Electromagnetics and Student Innovation, pp. 147–151, 2010. [79] H. F. AbuTarboush et al., “Multiband Inverted-F Antenna with Independent Bands for Small and Slim Cellular Mobile Handsets,” IEEE Trans. on Antennas and Propagation, Vol. 59, No. 7, pp. 2636–2645, 2011. [80] L. Ying, Y. Ban, and J. Chen, “Low-Profile Coupled-Fed Printed PIFA for Internal SevenBand LTE/GSM/UMTS Mobile Phone Antenna,” Proc. Cross Strait Quad-Regional Radio Science and Wireless Technology Conference, pp. 418–421, 2011. [81] M. Kim et al., “A Multi-Band Internal Antenna for All Commercial Mobile Communication Bands and 802.11 a/b/g/n WLAN,” Proc. Asia-Pacific Microwave Conference, pp. 171– 174, 2011. [82] Y. Wang, M. Lee, and S Chung, “Two PIFA-Related Miniaturized Dual-Band Antennas,” IEEE Trans. on Antennas and Propagation, Vol. 55, No. 3, pp. 805–811, 2007. [83] K. Wong, C. Chang, and Y. Lin, “Printed PIFA EM Compatible with Nearby Conducting Elements,” IEEE Trans. on Antennas and Propagation, Vol. 55, No. 10, pp. 2919–2922, 2007. [84] C. Rowell and E. Y. Lam, “Mobile-Phone Antenna Design,” IEEE Antennas and Propagation Magazine, Vol. 54, No. 4, pp. 14–34, 2012. [85] H. Y. D. Yang, “Miniaturized Printed Wire Antenna for Wireless Communications,” IEEE Antennas and Wireless Propagation Letters, Vol. 4, pp. 358–361, 2005. [86] P. C. Ooi and K. T. Selvan, “Compact T-Shaped CPW-Fed Printed Antenna for 3.5 GHz WiMAX Applications,” Proc. Int. Symp. on Antennas, Propagation and EMC Technologies, pp. 262–264, 2009. [87] S. Zhong and X. Liang, “Dual-Band Printed T-Shaped Slot Antenna for WLAN Application,” Proc. of Asia-Pacific Microwave Conference, pp. 1–3, 2006. [88] P. Mousavi, B. Miners, and O. Basir, “Wideband L-Shaped Circular Polarized Monopole Slot Antenna,” IEEE Antennas and Wireless Propagation Letters, Vol. 9, pp. 822–825, 2010. [89] M. S. Sharawi et al., “A Dual-Element Dual-Band MIMO Antenna System with Enhanced Isolation for Mobile Terminals,” IEEE Antennas and Wireless Propagation Letters, Vol. 11, pp. 1006–1009, 2012.



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[90] C. C. Lin, L. C. Kuo, and H. R. Chuang, “A Horizontally Polarized Omnidirectional Printed Antenna for WLAN Applications,” IEEE Trans. on Antennas and Propagation, Vol. 54, No. 11, pp. 3551–3556, , 2006. [91] D. Peroulis, K. Sarabandi, and L. P. B. Katehi, “Design of Reconfigurable Slot Antennas,” IEEE Trans. on Antennas and Propagation, Vol. 53, No. 2, pp. 645–654, 2005. [92] N. Behdad and K. Sarabandi, “Dual-Band Reconfigurable Antenna with a Very Wide Tunability Range,” IEEE Trans. on Antennas and Propagation, Vol. 54, No. 2, pp. 409– 416, 2006. [93] A. K. Bhattacharyya, “Effects of Finite Ground Plane on the Radiation Characteristics of a Circular Patch Antenna,” IEEE Trans. on Antennas and Propagation, Vol. 38, No. 2, pp. 152–159, 1990. [94] F. B. Oyibo, D. Smith, and S. J. Foti, “The Effects of a Finite Ground Plane on the Characteristics of Printed Patch Antennas with and without a Suspended Patch,” Proc. Int. Symp. on Communication Systems, Networks and Signal Processing, pp. 111–114, 2010. [95] D. Manteuffel et al., “Design Considerations for Integrated Mobile Phone Antennas,” Proc. 11th Int. Conf. on Antennas and Propagation, pp. 252–256, 2001. [96] Y. F. Weng, S. W. Cheung, and T. I. Yuk, “Effects of Ground-Plane Size on Planar UWB Monopole Antenna,” Proc. IEEE Region 10 Conference, pp. 422–425, 2010. [97] A. T. Arkko, “Effect of Ground Plane Size on the Free-Space Performance of a Mobile Handset PIFA Antenna,” Proc. Int. Conf. on Antennas and Propagation, pp. 316–319, 2003. [98] P. Vanikainen et al., “Resonator-Based Analysis of the Combination of Mobile Handset Antenna and Chassis,” IEEE Trans. on Antennas and Propagation, Vol. 50, No. 10, pp. 1433–1444, 2002. [99] D. J. Kozakoff, Analysis of Radome-Enclosed Antennas, 2nd ed., Norwood, MA: Artech House, 2010. [100] J. Krogerus, C. Icheln, and P. Vainikainen, “Effect of Human Body on 3-D Radiation Pattern and Efficiency of Mobile Handsets,” Proc. Instrumentation and Measurement Technology Conference, pp. 271–276, 2005. [101] W. Yu et al., “Accurate Simulation of the Radiation Performance of a Mobile Slide Phone in a Hand-Head Position,” IEEE Antennas and Propagation Magazine, Vol. 52, No. 2, pp. 168–177, 2010. [102] A. Sarolic et al., “Influence of Human Head and Hand on PIFA Antenna Matching Properties and SAR,” Proc. Int. Conf. on Software, Telecommunications and Computer Networks, pp. 1–5, 2011. [103] K. L. Wong, Y. C. Lin, and B. Chen, “Internal Patch Antenna with a Thin Air-Layer Substrate for GSM/DCS Operation in a PDA Phone,” IEEE Trans. on Antennas and Propagation, Vol. 55, No. 4, pp. 1165–1172, 2007. [104] K. Ogawa, T. Matsuyoshi, and K. Monma, “An Analysis of the Performance of a Handset Diversity Antenna Influenced by Head, Hand, and Shoulder Effects at 900 MHz: Part I—Effective Gain Characteristics,” IEEE Trans. on Vehicular Technology, Vol. 50, No. 3, pp. 830–844, 2001.

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[105] K. Ogawa, T. Matsuyoshi, and K. Monma, “An Analysis of the Performance of a Handset Diversity Antenna Influenced by Head, Hand, and Shoulder Effects at 900 MHz: Part II—Correlation Characteristics,” IEEE Trans. on Vehicular Technology, Vol. 50, No. 3, pp. 845–853, 2001. [106] M. A. Jensen and Y. R. Samii, “EM Interaction of Handset Antennas and a Human in Personal Communications,” Proc. IEEE, Vol. 83, No. 1, pp. 7–17, 1995. [107] D. Kelly and W. L. Stutzman, “Array Antenna Pattern Modeling Methods that Include Mutual Coupling Effects,” IEEE Transactions on Antennas and Propagation, Vol. 41, No. 12, pp. 1625–1632, 1993.

3 Electrically Small Printed Antennas Electrically small antennas (ESAs) are antennas that have overall sizes that are much smaller than their operating wavelengths. An antenna whose maximum dimension is less than a radian length (or can be enclosed within a radian sphere) is considered an ESA. A radian length is (l/2p). The first work that focused on ESAs and their fundamental features was conducted by H. Wheeler back in 1948 [1]. Since then, a significant amount of work has appeared in the literature to explain more and come up with more analytic expressions for the various parameters that describe the behavior of ESAs, mainly their radiation quality factor (Q) and gain (G). The treatment and understanding of the features of ESAs are of great importance for antenna designers and professionals because current mobile terminals require (because of limited space) small in size printed antennas that are usually considered electrically small. Such antennas suffer from various gain and bandwidth limitations due to their small size. In this chapter, we review the features of ESAs and show the various bounds on the radiation Q and the maximum gains that ESAs can provide, followed by several practical examples that appeared in the literature for some real applications. Elaborate derivations are omitted and only the final results and their implications are stressed. The reader is encouraged to check the detailed derivations in the original references.

3.1  Features of Electrically Small Antennas Since the early work on ESAs presented in [1], several follow-up works have tried to generalize and use different assumptions in the finding of lower bounds for the quality factor (Q) and maximum achievable gain (G) for ESAs [2–21]. 77

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In addition to Q and G, the efficiency (η) and the radiation characteristics of ESAs are discussed in the following sections. ESAs are generally defined as antennas that fit inside a sphere of radius a =  2p  1/k, where k is the wavenumber   associated with the electromagnetic field  λ at the frequency of operation [1]. Thus, an ESA is an antenna that can satisfy

ka < 1

(3.1)

Several recent references tighten this condition and consider an antenna to be an ESA if the antenna satisfies this condition [15–18, 21]:

ka < 0.5

(3.2)

Figure 3.1 shows the radius a (the radian length) of a sphere that encloses an ESA. This sphere is called the radian sphere. It contains the stored energy in the antenna’s electric and magnetic fields and serves as the boundary between the near fields and far fields radiated from the structure. 3.1.1  The Quality Factor (Q)

ESAs have high reactive input impedances and thus require sharp, narrow band tuning. This results in a narrow operating bandwidth for such antennas. To avoid confusion with different bandwidth definitions (e.g., 3-dB bandwidth, –10-dB bandwidth), the quality factor is usually used to describe the bandwidth of ESAs because they are inversely proportional to one another. The quality factor of an antenna can be written as follows [2]:

Figure 3.1  The radian sphere of radius a enclosing an antenna of maximum dimension D.



Electrically Small Printed Antennas



 2 ωW e  P , We > Wm  r Q =  2 ωWm , Wm > We  Pr

79

(3.3)

where We is the time-averaged, nonpropagating, stored electric energy; Wm is the time-averaged, nonpropagating, stored magnetic energy; ω is the radian frequency; and Pr is the radiated power. An exact derivation of the radiation Q based on spherical fields and considering either a TM10 or TE10 mode (i.e., linear polarization) is given in [8], and gives the following expression for Q :

Qmin -linear =

1 1 + ka (ka )3

(3.4)

where a is the radian sphere radius and k is the wavenumber. If the two TE and TM modes are combined/available (i.e., circular polarization), then the lower bound on the quality factor becomes [8]

Qmin -circular =

1 2 1   +  2  ka (ka )3 

(3.5)

Thus, for small ka values, the circularly polarized antenna Q is approximately one-half that of its linearly polarized counterpart, meaning the obtained frequency bandwidth is almost doubled. For a crossed dipole configuration, [10, 11] showed that the minimum Q can go down to one-third of (3.4) when the relative phases between the two antennas are considered. This was not considered in [2–4, 8], since these works were based on a frequency-domain approach, whereas [10, 11] were based on the time-dependent Poynting theorem approach. It is worth mentioning that (3.4) and (3.5) were derived based on lossless antennas that are residing in free space, with no ground plane around and without taking the matching network into effect while ensuring that the antenna occupies the complete sphere surrounding it (fills the radian sphere) [1–8]. When considering the matching network effect, the total Q (QT) of the antenna becomes [13]

1 1 1 = + QT Q A Qm

(3.6)

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where QA is the minimum antenna Q (for linearly or circularly polarized) and Qm is the matching network Q. Also, for nonperfect efficiency (i.e., η < 100%), the efficiency value lowers the total Q in a direct way, because its value is multiplied by (3.4) and (3.5). Figure 3.2 shows the lower bound Q curves as a function of the antenna size filling the radian sphere based on (3.4) and (3.5) and the effect of nonideal matching networks for both cases with Qm = 10, 50. Also, the effect of the efficiency of the antenna on its total Q is shown (η = 50%). Note the effect of these factors compared to the ideal cases of (3.4) and (3.5). Solid lines are for linearly polarized antennas, while dashed curves are for circularly polarized ones. The cases are symbol coded for ease of comparison. The matching network will degrade the Q factor of the antenna, and thus will broaden its BW. The efficiency has a similar effect.

Figure 3.2  The minimum radiation Q values for linearly polarized (solid curves) and circularly polarized (dashed curves) antennas according to (3.4) and (3.5). The effect of the matching network with different quality factors values (10 = circle; 50 = square) as well as the 50% efficiency (vertical dash) on the Q values as a function of the ESA size are shown.



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Best in [15–18] gave an exact expression for the Q of an ESA based on its impedance characteristics. This Q is valid over a very wide bandwidth and is given by  X (wo )  wo 2 Q (wo ) = R ′ (wo ) +  X ′ (wo ) +  2R (wo ) wo   2



(3.7)

where R(w) and X(w) are the frequency-dependent antenna feed point resistance and reactance, respectively; R  ′(w) and X   ′(w) are the frequency derivatives of the frequency-dependent resistance and reactance at the feed point, respectively. Equation (3.7) is applicable to small and large values of Q and to ESAs and normal size antennas. It has been found that it provides good agreement with measurements performed on different antenna structures as shown in [15–18]. Although (3.7) is not directly related to the antenna feature size, the antenna size effect will reflect and show through its resistance and reactance characteristics. It has been also shown that antennas occupying as much volume within the radian sphere as possible provide lower Q values and approach the Chu limit in (3.4) for linearly polarized antennas. A more recent work took the effect of a small antenna’s geometry into the calculation of the Q for linearly polarized antennas [19, 20]. The new bound depends on the volume occupied by the antenna relative to a complete sphere. In addition, the gain/directivity and Q are shown to be two interdependent quantities [21]. Under polarization matching conditions (i.e., between the antenna and the incoming wave), the bound between the directivity (D) and Q is given by [16, 20]



D k3 ≤ ( g1 + g2 ) Q 2p

(3.8)

Figure 3.3  The value of the eigenvalue γ1 normalized to 4πa3, where a is the radius of the Chu sphere. The prolate spheroid, the circular ring, and the circular cylinder correspond to the generalized semiaxis ratio of 10–3. (From: [19]. © 2009 IEEE. Reprinted with permission.)

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where g1 ≥ g 2 ≥ g3 are the eigenvalues of the high polarizability dyadic γ∞. The eigenvalues depend on the long-wavelength properties of the antenna and can be numerically calculated. The value γ2 vanishes if the antenna is nonmagnetic. In (3.8) k is the wavenumber. Figure 3.3 shows the multiplier values for some common antenna shapes needed to calculate the γ1 values according to [19]

g1 = 4ka 3 p

(3.9)

Figure 3.4 shows the eigenvalues γ1, γ2, and γ3 for a rectangular parallelepiped resembling a mobile device (phone, laptop, and so forth) as a function of its edge length radius. Solid curves resemble a1/a3 = 5 and dotted curves resemble a1/a3 = 10 [19]. Several expressions have been derived in the literature to estimate the minimum Q that can be obtained for an ESA. Each of the expressions was based on assumptions that might not be applicable to your design geometry especially if you are focusing on printed antenna structures. Thus, numerical results from full wave field solvers complemented with measurements are still considered the most accurate. For MIMO antennas, TARC curves are the ones that should be used to obtain the effective bandwidth for such antennas. 3.1.2  Maximum Gain (Gmax)

The gain and bandwidth are inversely proportional in a way that if you want more gain, your operating BW is reduced, and if you want more bandwidth, your gain is reduced. In addition, the efficiency of an ESA is much lower than

Figure 3.4  Eigenvalues as a function of the ratio a2/a1 for a rectangular volume of edge lengths a1, a2, and a3. Solid curves ��represent a1/a3 = 5 and dotted lines represent a1/a3 = 10. (From: [19]. © 2009 IEEE. Reprinted with permission.)



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its larger size counterpart because it stores more energy that it radiates, thus it has large Q values [1–4, 7]. A maximum gain value for the ESA was defined in [3, 4] as a function of the antenna size and operating frequency, and was given by

2 Gmax = 2 (ka ) + (ka )

(3.10)

 2p  where k is the wavenumber   , and a is the radius of the radian sphere  λ enclosing the antenna. Equation (3.10) was derived based on spherical wave functions radiated from an ESA enclosed by (filling) the radian sphere (Chu sphere). This gain estimate usually cannot be obtained because (3.10) does not account for losses and assumes the antenna fills the Chu sphere. In [9] a printed microstrip resonating antenna operating at 1.89 GHz was shown with a 30-mm diameter and 1.3-mm thickness for the high dielectric substrate. Applying (3.10), one would expect a maximum gain of 1.8 dBi. The actual measured gain was –3 dBi. The discrepancy is due to the fact that (3.10) does not account for losses and assumes that the antenna fills the sphere around it, while in reality, printed antennas only fill a disk out of the sphere. Other factors that might affect the antenna gain values would be the way the current distribution on the antenna looks like [13] and its efficiency. As stated earlier, the gain and bandwidth (gain bandwidth product) are tightly related, and usually one needs to optimize the product or the ratio between gain and the quality factor (GB or G/Q or D/Q) as shown in (3.8). Exact expressions without considering approximations (as was done in [2] for the maximum gain to Q values) were derived in [14] as a function of the antenna size and operating frequency (ka). It was concluded that the maximum gain expected from a linearly polarized, omnidirectional antenna would approach 1.5 (1.76 dBi), and for directional ones, 3 (4.77 dBi). These claims were rejected by [19, 20] recently when it was proved that it is impossible to obtain a gain of 3 at the minimum Q values for circularly polarized antennas (with TM and TE mode excitations). The new gain formula was given as [21]



Gmax

3   ka ) (   = 1.5 1 + 6  1 + (ka ) 

(3.11)

It is worth noting that (3.10) and (3.11) assume that the radian sphere encloses the antenna and the antenna fills most of the sphere. The effect of the antenna shape on the gain and quality factor of an ESA was extensively studied in [19, 20]. The gain bandwidth product for a polarization matched antenna is given by

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G max B ≤

4 p3 ( g1 + g2 ) λo3

(3.12)

where γ1 and γ2 are the eigenvalues of the high polarizability dyadic γ∞, and can be found numerically as shown in Section 3.1.1. 3.1.3  Efficiency

ESAs suffer from low efficiencies due to their low radiation resistance and high reactance, which stores most of the field thus providing poor radiation. The introduction of a matching network increases the losses (a lossy network) and thus degrades the efficiency even more. The antenna radiation efficiency (ηr) is given by [13]

ηr =

Rr Rr + Rl

(3.12a)

where Rr is the radiation resistance of the antenna and Rl is the resistance due to losses (i.e., conductive, dielectric). The impedance matching network will affect the system (antenna plus matching network) efficiency (ηr) according to [22]:



ηs = ηr ηm =

ηr Q 1+ A Qm

(3.13)

where ηm is the matching network efficiency, QA is the quality factor of the antenna, and Qm is the quality factor of the matching network. When Qm is much larger than QA , the efficiency of the system approaches that of the antenna alone. Equation (3.13) is valid when the matching network does not store energy the same way as the antenna (i.e., electric or magnetic). The radiation efficiency of multiturn and left-handed antennas were investigated in [23–24]. The efficiency of the antenna also should be accounted for when considering the lower bound of its radiation Q, such as [18]

 1 1  Q lb = ηr  + 3  ka (ka ) 

(3.14)

Most of the work on ESAs before [7] assumed efficient radiators, and thus the efficiency effect should not be neglected. The assumption was that



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85

the antenna occupies the radian sphere for efficient radiation, but most if not all printed antennas do not, so efficiency is a major quantity that needs to be quantified for a printed ESA. Because printed antennas cover a wide range of geometries, the evaluation of their efficiency is made using full wave analysis or field solvers. 3.1.4  Radiation Patterns

A wire-based ESA will have omnidirectional patterns (gain patterns) similar to those obtained from small dipoles when placed in free space [15, 16, 18]. In the case of printed antennas, the radiation pattern will be altered and affected by the adjacent ground plane and nearby electronics. These effects need to be carefully investigated via actual measurements of such printed antennas.

3.2  ESA Examples In this section, we provide several printed ESA examples and present their characteristics and radiation patterns. Almost all of the ESA examples that have appeared in the literature that are designed to result in minimum Q and gain expressions are based on antennas in free space. Thus, the focus in this section is on printed antennas with practical applications in wireless communication systems and their characteristics. 3.2.1  Meander Line ESAs

A printed meander line ESA is modeled in Figure 3.5. The antenna is designed to operate in the 1.8-GHz band (this can be changed by altering the number of arms). An FR-4 substrate with εr = 4.0 is used. The dimensions of the antenna are (all in mm) L = 30, L1 = 19, L2 = 1, Wa = 0.8, gap = 0.4, Wm1 = 13, Wm2 = 17, W = 18, Wf = 3.5. The antenna has a –10-dB BW of 32 MHz and a –6-dB BW of 68 MHz as shown in Figure 3.6. The antenna dimensions were 10 × 18 × 0.8 mm3, which corresponds to ka < 0.37 at 1.8 GHz. Note the surface current on the antenna structure as well as the feeding arm in Figure 3.7. This shows that the feeding structure of an ESA contributes to the radiation mechanism. This was also discussed in [9]. Even after considering the GND plane and the feed line as part of the antenna structure, the antenna is still considered electrically small (ka < 0.68) The maximum simulated gain was –0.7 dB as shown in Figure 3.8. The value expected from (3.10) using ka = 0.37 is –0.57 dB, while using ka = 0.68 gives a maximum gain of 2.61 dBi. The radiation pattern shape of this ESA is an omnidirectional one (a doughnut shape). The efficiency of this antenna at 1.8 GHz was 67%.

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Figure 3.5  Electrically small meander line antenna model in HFSS.

Figure 3.6  Simulated reflection coefficient curve for the meander line antenna shown in Figure 3.5.



Electrically Small Printed Antennas

Figure 3.7  Simulated surface current distribution of the 1.8-GHz meander line ESA.

Figure 3.8  Simulated 3D gain pattern of the 1.8-GHz meander line ESA.

87

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Meander line antennas or their derivatives are among the widely used printed ESA structures. In [25] a printed compact Peano line antenna is proposed and its performance is compared against a similarly sized meander line antenna. The antenna size was 6.1 × 3.2 mm and fabricated on a 50-µm polyamide substrate of εr = 3.5. Figure 3.9 shows the antenna geometry and Figure 3.10 shows the simulated and measured S-parameters of this ESA. This antenna resonates at approximately 963 MHz. Checking the condition of an ESA, ka = 0.235, which validates the ESA condition. It is clear that the input impedance is not very well matched to a 50Ω line as shown in Figure 3.10(a). The obtained –6-dB BW was more than 200 MHz around the 1-GHz point. The measured gain was approximately –8 dBi. The expected gain using (3.10) would be –2.8 dBi. It is worth noting that the 5-dBi gain difference is due to the fact that (3.10) is derived in free space with the assumption that the antenna fills the sphere and has a high efficiency, not to mention that the GND plane becomes part of the ESA as shown in the meander line example. In [26], a wideband printed T-loaded meander line antenna is presented that covers the bands from 1.57 to 2.5 GHz with more than 50% BW. The antenna size was 18 × 7.2 × 0.245 mm3 fabricated on an RT/duroid 5880 substrate with dielectric constant of 2.2. This is an ESA since ka = 0.11. The antenna structure is shown in Figure 3.11. The measured reflection coefficient

Figure 3.9  Geometry of a Peano line antenna (a) and its dimensions (b). (From: [25]. © 2009 Wiley. Reprinted with permission.)



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Figure 3.10  Measured S-parameters (a) and gain characteristics (b) of the Peano line antenna. (From: [25]. © 2009 Wiley. Reprinted with permission.)

is shown in Figure 3.12 (note the mistake in the paper; return loss is a positive quantity, while the reflection coefficient is a negative one). The radiation pattern obtained is omnidirectional across all bands of operation as shown in Figure 3.13. The maximum gain obtained was 1.53 dBi at 2 GHz. Comparing this value with that obtained from (3.10) reveals a gain that is 3.6 dB lower than expected, which is due to the presence of the GND plane and the fact that the antenna does not occupy the complete radian sphere. The efficiency of this antenna is 90%. 3.2.2  Loop and Spiral ESAs

A printed ESA based on loop and spiral antennas can be realized. In [27] an efficient printed loop antenna is proposed. The antenna loop acts as an inductor that resonates with a split-ring resonator (SRR) in the ground plane whose gap acts as the resonating capacitor. The antenna resonates at 2.045 GHz with an

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Figure 3.11  Geometry of the wideband antenna (all in mm): H = 7.2, W = 18, WT = 15.6, ws = 0.6, wt = 1.8, wf = 0.75, t = 0.254, and Lf = 15. (From: [26]. © 2007 IEEE. Reprinted with permission.)

impedance bandwidth of 100 MHz. The fabricated antenna is shown in Figure 3.14 and has the dimensions of 40 × 40 × 0.8 mm3 including the GND plane. A Teflon substrate with εr = 2.17 was used and ka < 1 for this antenna, thus it is considered to be an ESA. The measured maximum gain was 3.6 dBi with an efficiency of 81%. The actual antenna dimensions without the ground plane are 14.5 × 6.7 × 0.8 mm3. Thus, according to (3.10), a maximum gain of 0.9 dBi is expected. Figure 3.15 shows the measured radiation patterns as a function of the antenna orientation. The increase in the gain is due to the fact that the antenna is embedded within the GND plane, and thus the GND plane is acting as part of the complete antenna. A spiral-like, folded, self-complementary antenna is presented in [28]. Folded symmetry is used to increase the impedance bandwidth. The proposed antenna is showed in Figure 3.16 along with its matching network. The antenna is fed by a coplanar waveguide line. The antenna resonates at 340 MHz with an impedance bandwidth of 3.5 MHz (i.e., 1.1%), which shows 25% improvement over other similar structures without complementary structure.



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Figure 3.12  Measured and simulated return loss of the wideband antenna in Figure 3.11. (From: [26]. © 2007 IEEE. Reprinted with permission.)

The dimensions of the antenna were 160 × 80 × 0.787 mm3 with a substrate having εr = 2.2. This is an ESA because ka < 0.5. The efficiency of this antenna was only 19% and it had a maximum gain of –4.5 dB. This value is higher than what would be expected from (3.10) due to the ground plane structure and the ground of the measured setup as indicated in the reference. The gain pattern was omnidirectional in the θ plane. 3.2.3  Other Printed Geometries for ESAs

A 4-shaped printed ESA geometry was proposed in [29]. The antenna operated at two bands covering 780 to 840 MHz and 2642 to 2888 MHz. Two antennas were placed on a single substrate to build a 2 × 1 MIMO antenna system for portable and handheld devices. A single antenna occupied 33.5 × 25 × 1.56 mm3 of board area and was fabricated on an FR-4 commercial substrate with εr = 4.0. Figure 3.17 shows the fabricated 2 × 1 MIMO ESA based on 4-shaped elements. This is ESA because ka = 0.25. The maximum measured gain at 800 MHz as well as 2750 MHz was 0 dBi. This is higher than the expected maximum gain predicted by (3.10) because of the GND plane shared between the two antennas. A nonplanar ESA with wideband tuning was proposed in [30]. The antenna was based on patterning helical lines wrapped around an FR-4 substrate with εr = 4.5. The total size of the radiator was 10 × 30 × 3 mm3. Due to its electrically small nature (ka = 0.03), the input resistance is low and its input capacitance was high. A lumped component impedance matching network was designed to provide tuning selectivity and proper matching over a 40-MHz

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Figure 3.13  Measured radiation patterns of the wideband antenna at (a) 1.6 GHz, (b) 2 GHz, and (c) 2.5 GHz. (From: [26]. © 2007 IEEE. Reprinted with permission.)



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Figure 3.13  (continued)

range. The antenna operated between 180 and 220 MHz with 6-MHz channels for terrestrial digital multimedia broadcasting (T-DMB). Figure 3.18 shows the geometry of this ESA. Gain patterns and other performance metrics were not included for this ESA. An UWB printed ESA covering from 520 to 2400 MHz was proposed in [31]. The antenna size was 106 × 64.6 × 1.56 mm3 and it was fabricated on an FR-4 substrate with εr = 4.4. The antenna is an electrically small antenna because its total size including its ground plane gives ka < 0.58 with a CPW feed. The antenna is based on a tapered patch antenna as shown in Figure 3.19, and its dimensions were (all in mm) W = 64.4, L = 45.6, Ln = 15.2, Ln1 = 3, Wn = 11.4, Wn1 = 14.9, Wn2 = 2, G = 0.32, and Wf = 1.36. The measured gain patterns are shown in Figure 3.20 and they exhibit an omnidirectional pattern with a maximum gain of 3.6 dB. Metamaterial (MTM) based ESAs have also appeared in the literature [32, 33]. In [32] a double-layered composite right left hand (CRLH) MTM printed antenna was proposed with a zeroth-order resonance at 3.44 GHz. The antenna is shown in Figure 3.21. The ESA (ka < 0.3) efficiency was improved to 77% by using the second layer. The maximum antenna gain was 0.6 dB. The overall dimensions of the MTM ESA were 30 × 35 × 1.575 mm3 including the ground plane and the feed line.

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Figure 3.14  Fabricated small loop antenna. (From: [27]. © 2008 Wiley. Reprinted with permission.)

Several printed MTM-inspired antennas were proposed in [33]. Magnetic- and electric-based easy to design, fabricate, and measure antennas are proposed (thus called EZ antennas). All of the proposed antennas were electrically small with ka < 0.5. We present one design here that is denoted as design 17 in the original reference. This is a planar, electrically small, MTM-inspired antenna constructed on a two-layer duroid 5880 substrate with a 0.787-mm thickness and εr = 2.2. On one side of the substrate lies a printed monopole that is 8.3 mm long and 1.5 mm wide. The other side of the substrate has a meander line that covers the complete substrate and that is connected to the bottom finite GND plane. The size of the substrate was 14.7 × 18 mm2. Figure 3.22 shows the geometry of this MTM-inspired ESA.



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Figure 3.15  Measured gain patterns as a function of antenna orientation: (a) x-z plane, (b) y-z plane, and (c) x-y plane. (From: [27]. © 2008 Wiley. Reprinted with permission.)

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Figure 3.16  Geometry of the miniaturized self-complementary folded antenna with a distributed-element realization of the matching network. (From: [28]. © 2007 IEEE. Reprinted with permission.)

Figure 3.17  Fabricated dual-element 4-shaped MIMO antenna system. (From: [29], © Wiley 2012. Reprinted with permission.)

The antenna resonated at 1.373 GHz with ka < 0.497 and 55 MHz of impedance bandwidth. The efficiency of the antenna was higher than 94%. A picture of the fabricated antenna is shown in Figure 3.23.

3.3  Conclusions In this chapter we covered the theory of electrically small antennas (ESAs) and stated the major closed-form equations that describe their performance metrics



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Figure 3.18  Antenna configuration and tuning circuit. (From: [30]. © 2008 Wiley. Reprinted with permission.)

Figure 3.19  Geometry of the UWB antenna. (From: [31]. © 2007 Wiley. Reprinted with permission.)

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Figure 3.20  Measured gain patterns for the UWB antenna: (a) x-y plane and (b) y-z plane at four different frequencies. Dashed lines represent the simulation and solid lines the measurements. (From: [31]. © 2007 Wiley. Reprinted with permission.)



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Figure 3.21  Fabricated double-sided MTM antenna. (From: [32]. © 2012 Wiley. Reprinted with permission.)

Figure 3.22  Geometry of the MTM-inspired antenna with a monopole feed and a meander line on the other side of the board. (From: [33]. © 2008 IEEE. Reprinted with permission.)

such as the quality factor (Q), maximum gain (Gmax), efficiency (η), and radiation patterns. The effect of the antenna geometry on the radiation Q and maximum achievable gain are discussed based on the latest literature. We know that ESAs suffer from narrow bandwidth, highly reactive input impedance, and low efficiency. Several examples of ESAs that have appeared in the literature were discussed and their performance metrics were compared against those predicted from theory, especially their maximum achievable gain. Printed antennas have several effects that are not considered in the closed-form expressions such as the effect of the nearby GND plane and the fact that they

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Figure 3.23  Fabricated MTM-inspired antenna operating at 1.373 GHz over a GND plane. (From: [33]. © 2008 IEEE. Reprinted with permission.)

are not very efficient. These effects will alter their behavior compared to that predicted in the closed-form expressions for ESAs.

References [1] H. A. Wheeler, “Fundamental Limitations of Small Antennas,” Proc IRE, pp. 1479–1484, 1947. [2] L. J. Chu, “Physical Limitations of Omnidirectional Antennas,” Technical Report No. 64, Research Laboratory of Electronics, Massachusetts Institute of Technology, pp. 1–21, 1948. [3] R. F. Harrington, “On the Gain and Beam Width of Directional Antennas,” IRE Trans. on Antennas and Propagation, pp. 219– 225, 1958. [4] R. F. Harrington, “Effect of Antenna Size on Gain, Bandwidth, and Efficiency,” J. of Research of the National Bureau of Standards–D. Radio Propagation, Vol. 64D, No. 1, pp. 1–12, 1960. [5] H. A. Wheeler, “Small Antennas,” IEEE Trans. on Antennas and Propagation, Vol. AP-23, No. 4, pp. 462–469, 1975. [6] R. C. Hansen, “Fundamental Limitations in Antennas,” Proc. IEEE, Vol. 69, No. 2, pp. 170–182, 1981. [7] J. S. Mclean, “The Radiative Properties of Electrically-Small Antennas,” Proc. IEEE Int. Symp. on Electromagnetic Compatibility, pp. 320– 324, 1994. [8] J. S. McLean, “A Re-Examination of the Fundamental Limits on the Radiation Q of Electrically Small Antennas,” IEEE Trans. on Antennas and Propagation, Vol. 44, No. 5, pp. 672–676, 1996.



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[9] Q. Staub, J. F. Zurcher, and A. Skrivervik, “Some Considerations on the Correct Measurement of the Gain and Bandwidth of Electrically Small Antennas,” Microwave and Optical Technology Letters, Vol. 17, No. 3, pp. 156–160, 1998. [10] C. A. Grimes et al., “Time-Domain Measurement of Antenna Q,” Microwave and Optical Technology Letters, Vol. 25, No. 2, pp. 95–100, 2000. [11] D. M. Grimes and C. A. Grimes, “Minimum Q of Electrically Small Antennas: A Critical Review,” Microwave and Optical Technology Letters, Vol. 28, No. 3, pp. 172–177, 2001. [12] J. C.-E. Sten, A. Hujanen, and P. K. Koivisto, “Quality Factor of an Electrically Small Antenna Radiating Close to a Conducting Plane,” IEEE Trans. on Antennas and Propagation, Vol. 49, No. 5, pp. 829–837, 2001. [13] G. A. Thiele, P. L. Detweiler, and R. P. Penno, “On the Lower Bound of the Radiation Q for Electrically Small Antennas,” IEEE Trans. on Antennas and Propagation, Vol. 51, No. 6, pp. 1263–1269, 2003. [14] W. Geyi, “Physical Limitations of Antenna,” IEEE Trans. on Antennas and Propagation, Vol. 51, No. 8, pp. 2116–2123, 2003. [15] S. R. Best, “The Radiation Properties of Electrically Small Folded Spherical Helix Antennas,” IEEE Trans. on Antennas and Propagation, Vol. 52, No. 4, pp. 953–960, 2004. [16] S. R. Best, “A Discussion on the Properties of Electrically Small Self-Resonant Wire Antennas,” IEEE Antennas and Propagation Magazine, Vol. 46, No. 6, pp. 9–22, 2004. [17] A. D. Yaghjian and S. R. Best, “Impedance, Bandwidth, and Q of Antennas,” IEEE Trans. on Antennas and Propagation, Vol. 53, No. 4, pp. 1298–1324, 2005. [18] S. R. Best, “Low Q Electrically Small Linear and Elliptical Polarized Spherical Dipole Antennas,” IEEE Trans. on Antennas and Propagation, Vol. 53, No. 3, pp. 1047–1053, 2005. [19] M. Gustafsson, C. Sohl, and G. Kristensson, “Illustrations of New Physical Bounds on Linearly Polarized Antennas,” IEEE Trans. on Antennas and Propagation, Vol. 57, No. 5, pp. 1319–1327, 2009. [20] M. Gustafsson, C. Sohl, and G. Kristensson, “Physical Limitations on Antennas of Arbitrary Shape,” Proc. Royal Society, A Mathematical, Physical & Engineering Sciences, pp. 2589–2607, 2007. [21] H. L. Thal, “Gain and Q Bounds for Coupled TM-TE Modes,” IEEE Trans. on Antennas and Propagation, Vol. 57, No. 7, pp. 1879–1885, 2009. [22] G. S. Smith, “Efficiency of Electrically Small Antennas Combined with Matching Networks,” IEEE Trans. on Antennas and Propagation, Vol. 25, No. 2, pp. 369–373, 1977. [23] G. S. Smith, “Radiation Efficiency of Electrically Small Multiturn Loop Antennas,” IEEE Trans. on Antennas and Propagation, Vol. 20, No. 5, pp. 656–657, 1972. [24] Q. Liu, P. S. Hall, and A. L. Borja, “Efficiency of Electrically Small Dipole Antennas Loaded With Left-Handed Transmission Lines,” IEEE Trans. on Antennas and Propagation, Vol. 57, No. 10, pp. 3009–3017, 2009. [25] T. Terada et al., “Design of a Small, Low-Profile Print Antenna using a Peano Line,” Microwave and Optical Technology Letters, Vol. 51, No. 8, pp. 1833–1838, 2009.

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[26] R. Li et al., “A Compact Broadband Planar Antenna for GPS, DCS-1800, IMT-2000, and WLAN Applications,” IEEE Antennas and Wireless Propagation Letters, Vol. 6, pp. 25–27, 2007. [27] J. Kim et al., “Efficient Electrically Small Loop Antenna using SRR Structure on the Ground Plane,” Microwave and Optical Technology Letters, Vol. 51, No. 1, pp. 201–204, 2009. [28] R. Azadegan and K. Sarabandi, “Bandwidth Enhancement of Miniaturized Slot Antenna Using Folded, Complementary, and Self-Complementary Realizations,” IEEE Trans. on Antennas and Propagation, Vol. 55, No. 9, pp. 2435–2444, 2007. [29] M. S. Sharawi, “A Dual-Band Dual-Element Compact MIMO Antenna System for Mobile 4G Terminals,” Microwave and Optical Technology Letters, Vol. 55, No. 2, pp. 325–329, 2013. [30] I. Yoon et al., “Electrically Small Antenna with Frequency Tuning Circuit for Wideband Applications,” Microwave and Optical Technology Letters, Vol. 50, No. 1, pp. 244–247, 2008. [31] S. Tourette et al., “Compact UWB Printed Antennas for Low Frequency Applications matched to different Transmission Lines,” Microwave and Optical Technology Letters, Vol. 49, No. 6, pp. 1282–1287, 2007. [32] H. E. A. El-Raouf and S. S. Zaheer, “Design of Small Planar Antennas based on DoubleLayered CRLH Metamaterials,” Microwave and Optical Technology Letters, Vol. 54, No. 10, pp. 2224–2229, 2012. [33] A. Erentok and R. W. Ziolkowski, “Metamaterial-Inspired Efficient Electrically Small Antennas,” IEEE Trans. on Antennas and Propagation, Vol. 56, No. 3, pp. 691–707, 2008.

4 Printed Single-Band MIMO Antenna Systems In this chapter we focus on various single-band MIMO antenna systems that have appeared in the literature along with their geometries, features, and performances. The chapter is organized in a different way from a conventional one: Rather than sorting the systems based on their antenna types, we will sort them according to the application they cover. In addition, the operating frequency band within which the MIMO antenna system operates in is another classification criterion. Four major applications are identified for which printed MIMO antenna systems are classified within: (1) MIMO antennas for access points and general applications, (2) MIMO antennas for cellular and smartphones, (3) MIMO antennas for large PCs and tablet-type devices, and (4) MIMO antennas for USB dongle applications. The bands are classified as being below or above 1 GHz. This is based on the fact that below 1 GHz antennas usually suffer from additional coupling and are more challenging to design and isolate compared to those operating above 1 GHz. For each application area, antennas covering both bands are discussed.

4.1  MIMO Antennas for Access Points and General Applications Single-band MIMO antenna systems for wireless access points have appeared in several works such as [1–11]. All of these designs covered frequency bands higher than 1 GHz. The types of antennas used covered printed monopoles [1], dipoles [4, 11], patches [2, 6, 7, 9, 10], rings [8], and inverted-F antennas (IFAs) [5]. Several generic single-band MIMO antennas have been proposed 103

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[12–24]. All of these works cover frequency bands above 1 GHz except [24]. Some of the proposed systems can fit within small handheld devices [12, 14, 24], while others are bigger [13, 18]. We present four examples for this category covering four different antenna types. The first example is a three-element MIMO antenna system covering the 5–6-GHz band and thus can be used for IEEE 802.11n 5GHz systems [1]. The single antenna element consists of a wideband bowtie-like printed monopole antenna with dimensions of 15 × 16 × 1.58 mm3 on an FR-4 substrate. Broadbanding the antenna response and using an input impedance value near 50Ω was achieved using a three-point feeding technique that reduces the input reactive part as shown in Figure 4.1. The figure compares the response of the bowtie antenna that is fed at three points with that of a standard feed one. The effect of the three-point feeding is clear in the 5–6-GHz band. The three-element MIMO antenna system is shown in Figure 4.2. The three elements are placed such that they provide directional patterns covering three sectors. This minimizes the field correlations among the antenna elements and allows them to achieve good diversity performance. The obtained radiation pattern when one element is activated is shown in Figure 4.3. The measured isolation between the ports was around 10 dB. The performance of the MIMO

Figure 4.1  Input impedance curves for the three-point feed versus a regular feed. (From: [1]. © 2012 Wiley. Reprinted with permission.)



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Figure 4.2  Three-element MIMO antenna array for wireless access points based on the three-input bowtie monopoles. (From: [1]. © 2012 Wiley. Reprinted with permission.)

Figure 4.3  Directional radiation pattern for the three-element MIMO antenna array shown in Figure 4.2. (From: [1]. © 2012 Wiley. Reprinted with permission.)

antenna system compared to a standard uniform linear array (ULA) in terms of bit rates using different modulation schemes was comparable, despite its more compact size (i.e., the MIMO array is smaller than the ULA array). The second example is a printed reconfigurable dipole antenna for a dualelement MIMO antenna system for possible access point applications [11]. The proposed antenna had two possible frequency band configurations via the control of the length of the dipole arms via pin diodes. The two outer frequencies

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covered for the long and short dipole lengths were 2.25 and 2.5 GHz, respectively. The two antenna elements were place a quarter-wavelength from each other. This gave a worst case isolation of 13 dB. Figure 4.4 shows the fabricated single-element antenna of these MIMO antenna systems. The total substrate size for a single element was approximately 100 × 40 × 1.5 mm3 and it was built on an FR-4 material. The measured S-parameters for the two configurations are shown in Figure 4.5. The reported antenna efficiencies for the long and short configurations were 62% and 75%, respectively. The performance of the MIMO antenna system was characterized in an indoor environment (hallways of Drexel University, Philadelphia, PA). A 2 × 2 MIMO OFDM-based test bed was used. To determine the wideband MIMO channel capacity of the established link, the normalized channel matrix was calculated for each subcarrier to remove path loss differences among the number of channel matrices. Thus all channel matrices obtained for each receiver location and for each subcarrier were normalized to

Figure 4.4  Single-element fabricated photograph of the reconfigurable dipole printed antenna of the MIMO antenna system in [11]: (a) top side and (b) bottom side. (From: [11]. © 2008 IEEE. Reprinted with permission.)



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Figure 4.5  Measured S-parameter curves for the two antenna elements for the two configurations of the reconfigurable dipole antenna shown in Figure 4.4. (From: [11]. © 2008 IEEE. Reprinted with permission.)

the case of using the short configuration at that exact location and subcarrier used. Figure 4.6 shows the capacity improvement at five different indoor locations as a function of the SNR. The improvement is considered as the difference between the best obtained capacity out of the 16 possible configurations (2 configurations per antenna, two antennas at the transmit side and two antennas at the receiver side) and that of a fixed, all-short configuration 2 × 2 system. As seen from the figure, location 1 had no improvement at all, because at that location, the all-short configuration was the best performing one. Thus, when the difference is taken relative to itself, no relative improvement is obtained. The advantages of using the MIMO antenna configurations are clear from the figure. A compact three-element MIMO antenna system that occupied a 27 × 30 × 1 mm3 substrate and operating in the 5.2–5.8-GHz band is considered as our third example for access point applications [14]. The MIMO antenna consisted of two printed dipoles on the edges of the substrate with a slot antenna in between. The geometry and fabricated models are shown in Figure 4.7. The presence of the slot antenna between the dipoles not only enhances the isolation but also provides polarization diversity. The measured isolation curves between the three antennas are shown in Figure 4.8. The correlation coefficient was much less than 0.1 throughout the operating band. The measured radiation patterns are shown in Figure 4.9. Figure 4.9(a) shows the three plane cuts of the copol (Eθ) and cross-pol (Ef) patterns for

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Figure 4.6  Measured capacity improvement curves for five indoor locations using a 2 × 2 MIMO antenna system based on the printed reconfigurable dipole antenna shown in Figure 4.4. (From: [11]. © 2008 IEEE. Reprinted with permission.)

Figure 4.7  Three-element printed MIMO antenna [14]: (a) antenna diagram and (b) fabricated prototype. (From: [14]. © 2009 Wiley. Reprinted with permission.)

dipole antenna 1. Figure 4.9(b) shows the three plane cuts for dipole antenna 2, and Figure 4.9(c) shows the pattern of the slot antenna. It is clear that the x-z and x-y planes for the antenna show polarization diversity behavior because the maximum of one antenna does not coincide with the maximum of the other.



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Figure 4.8  Measured S-parameters for the three-element printed MIMO antenna shown in Figure 4.7. (From: [14]. © 2009 Wiley. Reprinted with permission.)

This will also reduce the field correlation coefficient and provide much better diversity performance. The final example is a metamaterial-based compact MIMO antenna system of three elements [23]. The antenna system operates in the ISM band centered at 2.45 GHz. The elements are simple printed monopole antennas with a 14-mm (0.11λo) separation distance. The geometry of a two-element MIMO antenna based on the simple monopoles is shown in Figure 4.10. To enhance the isolation between the two antennas, a split was made in the GND plane and capacitively loaded loops (a metamaterial) were added between the antenna elements. These two techniques improved the isolation and correlation coefficient values as shown in Figure 4.11. The three-element MIMO antenna system occupied approximately 75 × 50 × 1.5 mm3 of an FR-4 substrate. Thus, it can be used in access points as well as small portable devices. It is worth mentioning that having GND splits might not be practical in a real design because the GND of the MIMO antenna system is usually connected to the overall system ground plane. These coupling currents will flow back to the input ports of the proposed monopoles through the system ground, thus reducing the coupling between the antennas and degrading the MIMO performance. Isolation enhancement techniques will be presented in detail in Chapter 6 of this book. But since several enhancement techniques are mentioned in several references in this section, the methods used are briefly discussed here. In [12, 14, 15, 20, 21], polarization diversity was achieved by using the same antenna element [12] and placing elements orthogonal to one another, or different antenna types with different radiation mechanisms were used [14, 15, 20, 21]. This technique lowers the mutual coupling and enhances the diversity performance.

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Figure 4.9  Measured radiation patterns of the three-element MIMO antenna system: (a) antenna 1 is active, (b) antenna 2 is active, and (c) the slot antenna is active. (From: [14]. © 2009 Wiley. Reprinted with permission.)

MTM-like structures were used in [18, 19, 22], and [25] to enhance the isolation as well as provide some sort of miniaturization. The use of MTMs usually reduces the efficiency of the radiating structures and narrows their operating bandwidth.



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Figure 4.9  (continued)

Figure 4.10  Printed monopole antennas with MTM isolation. (From: [23]. © 2013 IEEE. Reprinted with permission.)

4.2  MIMO Antennas for Cellular and Smartphones One of the major challenges in MIMO printed antenna designs is their integration within user handheld terminals and cellular phones. Single-band printed antenna systems for mobile and smartphone terminals have appeared in several publications [25–67]. The classification of this application was based on the use of the standard substrate area size of 100 × 50 mm2, which resembles the size of a standard phone. A wide variety of printed antenna geometries were proposed and verified to satisfy MIMO requirements. While the majority of work tackled the higher bands of wireless standards (i.e., higher than 1 GHz),

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Figure 4.11  S-parameter curves when capacitively loaded loops are used between the twoelement monopole MIMO antenna system. (From: [23]. © 2013 IEEE. Reprinted with permission.)

several covered the lower GSM bands of 900 MHz and the new LTE bands at the 700-MHz band. Table 4.1 shows the antenna types used in the references covered in this section [25–67] for mobile phone applications. It is evident that PIFA-based, single-band MIMO antenna systems for cellular phone applications are the most widely used followed by meander lines, MTM inspired/loaded printed antennas, and printed slots. Out of these 42 references, only 10 of them [25–34] cover operating bands lower than 1 GHz. This is because at such low Table 4.1 Antenna Geometries Used in Single-Band MIMO Antenna Systems for Mobile Phones Antenna Type References Printed dipole/folded dipole [35] Printed loops/spiral [27, 30] Meander line [28, 29, 31, 33, 46, 61] Metamaterial Printed monopole/folded monopole Patches PIFA/IFA Printed slots Other

[52, 53, 62–65] [25, 32, 38, 43] [45, 48] [26, 34, 36, 37, 39–42, 47, 49, 56, 58, 60] [44, 54, 55, 57, 59] Chip antennas [50, 66] and resonators [51]



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frequencies, the system ground plane becomes part of the radiator, and the separation between the two antennas becomes very small, thus higher levels of correlation and very poor isolation are expected. Most of these low-band designs use an isolation enhancement mechanism. Those mechanisms are discussed in Chapter 6 in detail. Three examples are considered in this section. The first example is of a single-band printed MIMO antenna system based on a PIFA element. The second example presents a four-element printed MIMO antenna based on monopole elements. And the third one is based on a metamaterial-inspired fourelement MIMO antenna. The first example is a dual-element MIMO antenna system consisting of two PIFAs operating between 746 and 787 MHz [26]. Each antenna element occupies 18 × 15 × 5 mm3 and both are placed on the top side of a standard ground plane of 46 × 85 × 1 mm3. The geometry of this antenna is shown in Figure 4.12. A photograph of the fabricated antenna as well as its measured Sparameters are shown in Figure 4.13. The efficiency of this antenna was 50%, and its correlation coefficient did not exceed 0.2. The isolation was improved by using a combination of a suspended line and a branch line. The measured radiation patterns were given for a single antenna, and are shown in Figure 4.14 for the x-z and x-y planes. It is worth mentioning that most of the proposed single-band printed MIMO antenna systems operating at bands lower than 1 GHz are of a 3D

Figure 4.12  Two-element printed MIMO antenna with a branch line for isolation enhancement [26]. (From: [26]. © 2010 Wiley. Reprinted with permission.)

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Figure 4.13  (a) Fabricated antenna in [26] and (b) measured S-parameters. (From: [26]. © 2010 Wiley. Reprinted with permission.)



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Figure 4.14  Measured radiation patterns of the two-element MIMO antenna system: (a) x-z plane at 760 MHz and (b) x-y plane at 760 MHz. (From: [26]. © 2010 Wiley. Reprinted with permission.)

nature except those in [28] and [29], where only meander lines were used on the same GND plane substrate. Using a 3D raised antenna such as the one presented in Figure 4.12 might violate the practical allowed volume for the antennas with an actual user terminal, where usually it should not exceed 2 to 3 cm3 of its volume. A four-element printed MIMO antenna system consisting of monopole antennas operating between 1.88 and 2.2 GHz is considered as the second example for mobile phone applications [44]. Two pairs of printed monopoles are

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used for the top and bottom sides of the phone. Each pair consists of identical elements and performance controlling stubs to improve the matching and resonance frequency. The two different pairs were chosen to enhance the isolation due to the different radiation mechanisms and the way they induce ground currents. Also, the spatial separation and high band operation (i.e., above 1 GHz) make the isolation acceptable. Figure 4.15 shows the fabricated antenna. It occupies 95 × 60 × 0.8 mm3 of an FR-4 substrate. The measured S-parameters are shown in Figure 4.16. Figure 4.17 shows the rectangular gain plots of the MIMO antenna system. Three cuts are shown, the azimuth cut in part (a), and two elevation cuts at 0° azimuth and 90° azimuth in parts (b) and (c), respectively. A maximum measured gain of 3.5 dBi for a single antenna was obtained. The maximum envelope correlation obtained was 0.263 (ρe < 0.5), thus good diversity performance is expected. The MEG values of the four antenna elements were within –6.267 and –5.67 dB for an XPD value of 0 dB, and were within –6.3545 and – 5.597 dB for an XPD value of 6 dB. In both environments, the MEG values did not differ by more than 1 dB. (Remember that for good diversity performance, MEGi – MEGj < 3 dB.) The third example in this category is the MTM-inspired, four-element printed MIMO antenna system operating in the ISM band [63]. Complementary split-ring resonators (CSRRs) are the duals of SRRs and provide negative permittivity when used in periodic structure arrangements to form a metamaterial. A single CSRR is used to load each patch in the MIMO antenna system for miniaturization purposes, thus it is called a metamaterial-inspired antenna. The

Figure 4.15  Four-element printed monopole MIMO antenna system. (From: [44]. © 2007 IEEE. Reprinted with permission.)



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Figure 4.16  S-parameter curves. (From: [44]. © 2007 IEEE. Reprinted with permission.)

four-element printed MIMO antenna system resonated at 2.475 GHz with at least a 50-MHz operating bandwidth. The geometry and the fabricated model are shown in Figure 4.18. A single patch (antenna element) occupied 14 × 18 × 0.8 mm3 with an interelement separation of 10 mm from each side. The CSRR was etched out from the GND plane with an outer radius of 6 mm. The four-element part occupied one-half of a standard mobile terminal size of 100 × 50 × 0.8 mm3 board. The substrate used was a commercial FR-4 substrate with εr = 4.4. The simulated and measured S-parameters are shown in Figure 4.19. Very good agreement is observed. The measured minimum isolation was 10 dB. The current distribution when element 1 was excited while others were terminated with 50Ω is shown in Figure 4.20. Note the high current levels at the patch edges as well as around the CSRR on the GND plane.

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Figure 4.17  Measured rectangular gain curves: (a) azimuth cut, (b) elevation cut at 0° azimuth, and (c) elevation cut at 90° azimuth. (From: [44]. © 2007 IEEE. Reprinted with permission.)



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Figure 4.18  Geometry of the four-element, MTM-inspired printed MIMO antenna: (a) top layer, (b) bottom layer, and (c) fabricated prototype. (From: [63]. © 2013 IEEE. Reprinted with permission.)

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Figure 4.19  Simulated and measured S-parameters. (From: [63]. © 2013 IEEE. Reprinted with permission.)

Figure 4.20  Current distribution when element 1 is excited.

The TARC curves obtained for this MIMO antenna system are presented in Figure 4.21. The excitation of element 1 was fixed at 1V and zero phase, while the phases of elements 2, 3, and 4 were varied. Figure 4.21(a) shows the TARC curves obtained when elements 1 and 4 were in phase while the phases of element 2 and 3 were varied. The minimum –6-dB bandwidth obtained was 50 MHz. Figure 4.21(b) shows the TARC curves obtained when the phases of elements 1 and 2 were similar while the phases of elements 3 and 4 were varied.



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Figure 4.21  TARC curves: (a) Elements 1 and 4 are in same phase while elements 2 and 3 vary. (b) Elements 1 and 2 are in same phase while elements 3 and 4 vary.

The minimum –6-dB BW obtained was 60 MHz. The TARC curves show that the effective bandwidth of the multiport MIMO antenna will not change from the single-port measurements (S-parameters shown in Figure 4.19) when different phase excitations are encountered. The correlation coefficient curves between the various elements do not exceed 0.05 in the band of interest. The efficiency of such a MIMO antenna system was approximately 30%. Such efficiencies are expected when high miniaturization levels are obtained (the patch size was 76% smaller than a rectangular one operating at 2.45 GHz). The maximum measured gain value was –0.8 dBi. The measured normalized gain patterns are shown in Figure 4.22 for the copolarized and cross-polarized configuration for all four elements. The MEG values of the four elements were within –5.44 to –8.64 dB for both XPD val-

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Figure 4.22  Measured gain patterns at 2.48 GHz: (a) x-z plane; (b) y-z plane, where circles = copol element 1, solids = copol element 2, dots = cross-pol element 1, dashes = cross-pol element 2; (c) x-z plane; and (d) y-z plane, where circles = copol element 3, solids = copol element 4, dots = cross-pol element 3, dashes = cross-pol element 4 [63].

ues of 0 and 6 dB. The difference between the values was approximately 3 dB, showing acceptable diversity behavior.

4.3  MIMO Antennas for Large PCs and Tablets Most of the aforementioned antennas are suitable for use within larger size devices such as laptop computers and tablet PCs. The presence of the large ground plane and device size allows for spatial separation between the antennas and thus enhancing the isolation. References [68–76] cover some examples of



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printed MIMO antennas with large ground planes for laptop and tablet PC applications. Figure 4.23 shows some printed antennas integrated in laptop display housings. A variety of printed MIMO antennas for large devices have appeared in literature. For example, in [71] a compact four-element MIMO antenna system for WLAN operation was proposed. The MIMO antenna system consisted of two types of antennas to reduce the coupling and enhance isolation, one was a proximity-coupled fed square ring patch and the other was a quarter-wave microstrip slot antenna. The overall MIMO antenna system size was approximately 82 × 62 × 2.6 mm3. To enhance the isolation between the adjacent square rings and slot antennas, a group of ground plane slits were introduced between them. These act as a bandstop filter and reduce the coupling currents on the GND plane. Figure 4.24 shows the geometry of the proposed fourelement MIMO antenna systems. The measured S-parameters are presented in Figure 4.25. The isolation between all antenna elements was better than 25 dB. The envelope correlation coefficient was less than 0.022 in the band covered. The 3D measured gain patterns are shown in Figure 4.26. The maximum measured gain for the slot antennas was 2.8 dBi, whereas for the square rings it was 2.3 dBi. The second example is a two-element MIMO antenna consisting of monopoles placed at the two sides of the ground plane operating in the 2.4GHz WLAN band as presented in [72]. The overall GND plane size was 60 × 60 mm2 and an FR-4 substrate was used. Figure 4.27 shows the geometry of the antenna. A suspended line was introduced between the feed points of the antennas to improve the isolation levels to higher than 20 dB. The envelope

Figure 4.23  Laptop integrated MIMO antennas: (a) patch antennas and (b) E-shaped antennas. (From: [68]. © 2007 IEEE. Reprinted with permission.)

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Figure 4.24  Geometry of the four-element printed MIMO antenna system: (a) side view, (b) top view, and (c) bottom view. Parameters (all in mm): L1 = 37.7, W1 = 25, L2 = 22, W2 = 7, h1 = 1.86, h2 = 0.8, Ls = 18.75, Lc = 30, and Wc = 1. (From: [71]. © 2009 IEEE. Reprinted with permission.)

correlation was lower than 0.1. The apparent diversity gain was calculated to be 9.95 dB, while the actual measured diversity gain in the reverberation chamber was 7.25 dB. The measurement was conducted at 1% probability of the CDF as shown in Figure 4.28. The difference between the two diversity gains is attributed to the efficiency of the antennas. A compact minilaptop printed MIMO antenna was proposed in [73] and is considered here as the third example for this application. The single-element antenna was an inverted U-shaped loop antenna operating at 2.6 GHz. The basic shape of this antenna placed on the back of the screen is shown in Figure 4.29. The size of a single element is 30 × 9 mm2 and the thickness of the FR-4 substrate used was 0.8 mm. The size of the system GND plane was 220 × 170 mm2. Three configurations of the two-element MIMO antenna system were examined.



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Figure 4.25  Measured S-parameters of the fabricated four-element MIMO antenna. (From: [71]. © 2009 IEEE. Reprinted with permission.)

Figure 4.26  Measured radiation patterns of the four elements of the MIMO antenna in [71]: (a) antenna 1, (b) antenna 2, (c) antenna 3, and (d) antenna 4. (From: [71]. © 2009 IEEE. Reprinted with permission.)

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Figure 4.27  Two-element MIMO antenna system for tablet PC applications. Parameters (all in mm): l1 = 24, l2 = 18.2, g = 0.5, h = 1.1, and fg = 2.85. (From: [72]. © 2010 IET. Reprinted with permission.)

Figure 4.28  Calculated and measured diversity gain of the antenna shown in Figure 4.27. (From: [72]. © 2010 IET. Reprinted with permission.)



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Figure 4.29  Inverted U-shaped loop antenna on a PC screen: (a) antenna geometry, (b) front side, and (c) back side. Parameters (all in mm): L = 220, W = 170, Ls = 19.5, Ws = 6, Lg = 30, Wg = 9, S1 = 0, S2 = 5.5, and S3 = 5. (From: [73]. © 2011 IEEE. Reprinted with permission.)

Figure 4.30 shows the frequency response of the three configurations. Note that configuration 2 yields the best isolation performance as shown in Figure 4.30(b). Note also that the separation between the two antenna elements in all configurations was only 1 mm. This makes the type of antenna very appealing because it does not require large spatial separation. The measured gain and radiation patterns are shown in Figure 4.31 for configuration 1. The gain variation in each case was less than 1 dB and between various elements was less than 3 dB, thus showing good performance for MIMO systems. The maximum gain obtained was approximately 3 dB at 2.6 GHz. The MIMO channel capacity performance was characterized in a reverberation chamber. The measured channel capacities for SISO and MIMO configurations are shown in Figure 4.32. The measured capacity was 5.7 bps/Hz for a SISO configuration and was 10.9 bps/Hz for the 2 × 2 MIMO case at the 20-dB SNR level. The last example is a radiation pattern reconfigurable U-slot patch-based printed MIMO antenna system than can be integrated within PC/laptop computer screens [74]. The antenna operated in the 5.3-GHz bands with a measured –10-dB bandwidth of 6.6%. The pattern reconfigurability was achieved by loading the U-slot antenna with PIN diodes that connect the four edges of the U-slot patch to shorting posts. Figure 4.33 shows the geometry of the

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Figure 4.30  Measured S-parameters for three configurations of the inverted U-shaped MIMO antenna on a laptop screen: (a) configuration 1, (b) configuration 2, and (c) configuration 3. (From: [73]. © 2011 IEEE. Reprinted with permission.)



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Figure 4.31  Measured (a) gain patterns for configuration 1 and (b) peak gain of the inverted U-shaped MIMO antenna on a laptop screen in configuration 1. (From: [73]. © 2011 IEEE. Reprinted with permission.)

proposed antenna and the location of the PIN diodes. The RT/duroid 5880 substrate size used was 50 × 50 mm2 with a thickness of 3.175 mm (εr = 2.2, tan d = 0.0009). The antenna size (patch) was 12.6 × 12.6 mm2. Table 4.2 shows the dimensions of the various antenna parameters shown in Figure 4.33. A comprehensive parametric study on the effects of the major parameters on the performance of antennas was conducted. Three operating

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Figure 4.32  Measured capacity curves for the inverted U-shaped MIMO antenna on a laptop screen in configuration 1. (From: [73]. © 2011 IEEE. Reprinted with permission.)

Figure 4.33  Geometry of the pattern reconfigurable U-slot antenna. (From: [74]. © 2012 IEEE. Reprinted with permission.)



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Table 4.2 Reconfigurable U-Slot Antenna Dimensions Parameter L1 L2 L3 L4 L5 Value (mm) 12.6 50 5.5 5.4 4.2 Parameter L6 L7 L8 L9 r Value (mm) 4.6 0.7 1.7 9.8 0.7 From: [74].

Table 4.3 Excitation Table for the States and the Resulting Radiation Pattern Diodes A Diodes B Radiation Pattern State I Zero bias Zero bias Boresight State II Reverse bias Forward bias Conical with maximum power in y-z plane State III Forward bias Reverse bias Conical with maximum power in x-z plane

states were identified to give three radiation patterns based on the excited modes of the patch when various shorting posts were connected to it. Table 4.3 shows these three states/modes and the corresponding patterns generated. The measured S-parameters for the three states are shown in Figure 4.34. The measured normalized radiation patterns for the three states are shown in Figure 4.35 in the y-z plane. The maximum gain curves for the three states are shown in Figure 4.36. The reconfigurable antenna was tested in a 2 × 2 MIMO configuration with 1λ separation between the antennas. The envelope

Figure 4.34  Measured S-parameters for the three states of the reconfigurable antenna. (From: [74]. © 2012 IEEE. Reprinted with permission.)

Figure 4.35  Measured radiation patterns in the y-z plane for the three states: (a) state I, (b) state II, and (c) state III. (From: [74]. © 2012 IEEE. Reprinted with permission.)

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Figure 4.36  Measured maximum gain curves for the three states of the reconfigurable U-slot antenna. (From: [74]. © 2012 IEEE. Reprinted with permission.)

correlation values based on the radiation patterns for the three states of the antenna are listed in Table 4.4. Please note that the worst case envelope correlation was 0.33, which is less than 0.5; thus, good diversity performance is expected. The performance of the MIMO antenna system was evaluated in an indoor environment. A 2 × 2 MIMO antenna system setup was used in the experiments and both line-of-sight (LOS) and non-line-of-sight (NLOS) scenarios were examined for the nine possible configurations (three states for each antenna). The obtained capacity of the wideband channel was the average over all of the OFDM subcarriers at 10 locations. The LOS and NLOS geometries are shown in Figure 4.37. LOS was achieved when Tx1 and Rx were used (same room), whereas NLOS was achieved when Tx2 and Rx were used (two separate rooms separated by concrete and gypsum boards). The obtained capacity improvement when both reconfigurable antennas were at state I and using a channel matrix averaging method that does not include the antenna gain effect is shown in Figure 4.38. Figure 4.38(a) shows the benchmark case of using standard omnidirectional antennas, while Figure 4.38(b) shows the reconfigurable antenna measurements when in state I (see Table 4.4 Envelope Correlation Values for a TwoElement MIMO Antenna for All States State I State II State III State I 1 0.013 0.07 State II 0.013 1 0.33 State III 0.07 0.33 1

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Figure 4.37  MIMO antenna performance measurement setup: (a) indoor room and antenna setup and (b) two antenna separation. (From: [74]. © 2012 IEEE. Reprinted with permission.)

Table 4.3). At an SNR of 10 dB, the system capacity using the reconfigurable antennas shows an 18% and 13% improvement for the LOS and NLOS scenarios, respectively. The observation that the LOS scenario has better capacity improvement than the NLOS one is due to the fact that generally subchannel correlation in an indoor LOS scenario is larger for the LOS case when the correlation between the MIMO subchannels is reduced.

4.4  MIMO Antennas for USB Dongles Universal serial bus (USB) based dongles are widely used nowadays for connectivity to various wireless standards. The standard size of a USB dongle backplane is approximately 30 × 70 mm2. Several USB dongle MIMO antenna



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Figure 4.38  Capacity improvement: (a) standard omnidirectional MIMO antenna system and (b) U-slot reconfigurable MIMO antenna in configuration 1. (From: [74]. © 2012 IEEE. Reprinted with permission.)

systems have been proposed that cover single operating bands [77–94]. The two-element MIMO antenna systems for USB dongle applications proposed in [77–79] cover frequency bands lower than 1 GHz, whereas those in [80–94] cover frequency bands higher than 1 GHz. Three examples are considered covering three antenna types. A two-element, meander line based, printed MIMO antenna system for USB dongle applications was proposed in [79] and is our first example. The antenna covered the 704–746-MHz band of the LTE standard. The two meander lines were fed via capacitive coupling. Figure 4.39 shows the geometry and

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Figure 4.39  Geometry of the two-element MIMO antenna system for USB dongles: (a) geometry, (b) top side of fabricated model, and (c) bottom side of fabricated model. (From: [79]. © 2012 IEEE. Reprinted with permission.)

fabricated antenna. It occupied a volume of 40 × 20 × 4 mm3 on a 40 × 70 × 1-mm3 backplane. The measured S-parameters are shown in Figure 4.40 with a 4-mm gap between the two elements; the achieved isolation was more than 13 dB in the covered band. The measured total efficiency of the antenna was higher than 37% with a maximum average gain of –2.63 dBi. Figure 4.41 shows the envelope correlation coefficient curves from measurements and simulations, with values of less than 0.43. Figure 4.42 shows the measured radiation patterns for MIMO antennas 1 and 2 at 725 MHz. The radiation patterns are diagonally orthogonal to each other even though they are linearly mounted. The second example is a two-element, printed monopole based MIMO antenna for USB dongle applications [92]. The total size of the antenna system was 30 × 65 × 1 mm3. Figure 4.43 shows the geometry of the proposed twoelement MIMO antenna system. A neutralization line was used to enhance the



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Figure 4.40  Measured and simulated S-parameters for the USB dongle MIMO antenna shown in Figure 4.39. (From: [79]. © 2012 IEEE. Reprinted with permission.)

Figure 4.41  Envelope correlation coefficient curves for the USB dongle MIMO antenna shown in Figure 4.39. (From: [79]. © 2012 IEEE. Reprinted with permission.)

isolation performance of the MIMO antenna system. (More on neutralization line isolation enhancement technique will be given in Chapter 6.) The measured S-parameters of the proposed antenna are shown in Figure 4.44. The antenna covers the 2.4-GHz WLAN band with an operating –10-dB bandwidth of approximately 100 MHz. The calculated TARC curves are shown in Figure 4.45 for 30° steps in the relative phase between ports 1 and 2. It is

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Figure 4.42  Measured radiation patterns at 725 MHz for the USB dongle MIMO antenna shown in Figure 4.39: (a) antenna element 1 and (b) antenna element 2. (From: [79]. © 2012 IEEE. Reprinted with permission.)

clear that the effective bandwidth of the system will not be degraded with such phase changes. The measured envelope correlation coefficient obtained from a reverberation chamber was less than 0.06. This low correlation is due to the fact that the radiation patterns of the two antennas are almost orthogonal. Figure 4.46 shows the measured 3D gain patterns of antennas 1 and 2. The maximum measured gain was 2.1 dBi with a maximum efficiency of 75%. The diversity gain CDF based on 3,000 sample measurements is shown in Figure 4.47. A value of 9.8 dB in diversity gain improvement is observed. A PIFA-based printed MIMO antenna system for USB dongles was proposed in [94] and is considered as our third and last example for this category. It occupied a backplane of 20 × 60 × 1.56 mm3 on an FR-4 substrate. The antenna geometry and its fabricated prototype are shown in Figure 4.48. A coplanar waveguide (CPW) feed is used to feed each of the two PIFAs. The antenna system covers the frequency band between 2.5 and 2.7 GHz, making



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Figure 4.43  The two-element USB dongle MIMO antenna from [92]: (a) antenna geometry, (b) a closer look, and (c) a fabricated model. (From: [92]. © 2012 IEEE. Reprinted with permission.)

it suitable for LTE and WiMAX applications. Figure 4.49 shows the simulated and measured S-parameters. Note that a –8-dB isolation is obtained without using an isolation enhancement technique. The envelope correlation calculated from the measured S-parameters was below 0.1 in the operating bandwidth. The total radiation efficiency was 67% and the maximum measured gain was 2 dBi. Figure 4.50 shows the simulated and measured antenna radiation patterns for the principle cuts at f = 0°, f = 90°, and θ = 90°. Table 4.5 shows the various types of printed antennas that have appeared in the literature that represent USB dongle MIMO antenna systems. PIFAs

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Figure 4.44  Measured S-parameters for the USB dongle MIMO antenna shown in Figure 4.43. (From: [92]. © 2012 IEEE. Reprinted with permission.)

Figure 4.45  TARC curves for the USB dongle MIMO antenna shown in Figure 4.43. (From: [92]. © 2012 IEEE. Reprinted with permission.)

and printed monopoles are the two most used printed antenna types for this application.

4.5  Conclusions This chapter discussed single-band printed antenna systems. The antennas were grouped into four categories based on their use in four major applications/ device sizes: 1. Access point and general use; 2. Cellular and smartphones; 3. Large PCs and tablets;



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Figure 4.46  Measured 3D gain patterns for the USB dongle MIMO antenna shown in Figure 4.43: (a) antenna 1 and (b) antenna 2. (From: [92]. © 2012 IEEE. Reprinted with permission.)

Figure 4.47  Measured diversity gain curves for the USB dongle MIMO antenna shown in Figure 4.43 in a reverberant chamber. (From: [92]. © 2012 IEEE. Reprinted with permission.)

4. USB dongles. Within each category, most of the work that appeared in literature since 2006 until the time of this book’s writing (2013) was cited. Several examples covering various antenna types and frequency bands have been demonstrated and discussed. MIMO antenna system performance parameters were highlighted for the examples listed. Note that some antenna geometries can be used for several applications and within various device types, but we tried our best to fit them with the most suitable application/device size.

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Figure 4.48  Geometry and fabricated model of the USB dongle PIFA: (a) top side, (b) bottom side, and (c) a fabricated prototype. (From: [94]. © 2012 IEEE. Reprinted with permission.)

The majority of the work published in the literature for single-band printed MIMO antennas covered cellular phones and USB dongle applications and operated at high frequency bands (i.e., higher than 1GHz); thus, tables describing the types of antennas used in all of the cited work for these two categories were provided to guide the researcher/designer to the widely used antenna geometries for such devices.



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Figure 4.49  Measured S-parameters for the USB dongle PIFA MIMO antenna system. (From: [94]. © 2012 IEEE. Reprinted with permission.)

Figure 4.50  Measured and simulated gain patterns at 2.6 GHz for the USB dongle PIFA MIMO antenna system: (a) elevation cut (φ = 0°), (b) elevation cut (φ = 90°), and (c) azimuth cut (θ = 90°). (From: [94]. © 2012 IEEE. Reprinted with permission.)

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Printed MIMO Antenna Engineering Table 4.5 USB Dongle MIMO Antenna Types Antenna Type References Meander line [76, 79, 87, 91] Printed monopole [78, 82, 85, 86, 89, 92] PIFA [80, 81, 84, 93, 94] Other [77, 90] Metamaterial [83] Printed slot [88]

References [1] F. M. Pigozzo et al., “A Compact Monopole MIMO Array for the 5-6 GHz Band,” Microwave and Optical Technology Letters , Vol. 54, No. 8, pp. 1854–1858, August 2012. [2] K. M. K. H. Leong, C. J. Lee, and T. Itoh, “Compact Metamaterial Based Antennas for MIMO Applications,” Proc. International Workshop on Antenna Technology, pp. 87–90, 2007. [3] H. Farhat and G. E. Zein, “Antenna Arrays Design and Calibration for 4 × 4 Mobile Radio Channels Measurements at 3.5 GHz,” Proc. 7th Int. Conf. on ITS, pp. 1–5, 2007. [4] C. Y. Chiu, J. B. Yan, and R. D. Murch, “Compact Three-Port Orthogonally Polarized MIMO Antennas,” IEEE Antennas and Wireless Propagation Letters, Vol. 6, pp. 619–622, 2007. [5] C. Y. Chiu and R. D. Murch, “Compact Four-Port Antenna Suitable for Portable MIMO Devices,” IEEE Antennas and Wireless Propagation Letters, Vol. 7, pp. 142–144, 2008. [6] J. Sarrazin et al., “Four Co-Located Antennas for MIMO Systems with a Low Mutual Coupling Using Mode Confinement,” Proc. Antennas and Propagation Society Int. Symp., pp. 1–4, 2008. [7] H. Zhang et al., “A Compact MIMO Antenna for Wireless Communication,” IEEE Antennas and Propagation Magazine, Vol. 50, No. 6, pp. 104–107, December 2008. [8] E. A. Daviu et al., “Design of a Multimode MIMO Antenna Using Characteristic Modes,” Proc. 3rd European Conference on Antennas and Propagation, pp. 1840–1844, 2009. [9] A. Araghi and G. Dadashzadeh, “Oriented Design of an Antenna for MIMO Applications Using Theory of Characteristic Modes,” IEEE Antennas and Wireless Propagation Letters, Vol. 11, pp. 1040–1043, 2012. [10] D. Piazza et al., “Two Port Reconfigurable Circular Patch Antenna for MIMO Systems,” The 2nd European Conference on Antennas and Propagation, pp.1–7, 2007. [11] D. Piazza et al., “Design and Evaluation of a Reconfigurable Antenna Array for MIMO Systems,” IEEE Trans. on Antennas and Propagation, Vol. 56, No. 3, pp. 869–881, March 2008.



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[12] S. Dumanli et al., “The Effect of Antenna Position and Environment on MIMO Channel Capacity for a 4 Element Array Mounted on a PDA,” Proc. 9th European Conference on Wireless Technology, pp. 201–204, September 2006. [13] J. S. Bellon et al., “Textile MIMO Antenna for Wireless Body Area Networks,” Proc. 5th European Conference on Antennas and Propagation (EUCAP), pp. 428–432, 2011. [14] S. W. Su, “A Three-in-One Diversity Antenna System for 5 GHz WLAN Applications,” Microwave and Optical Technology Letters, Vol. 51, No. 10, pp. 22477–2481, October 2009. [15] H. T. Hui, S. K. Padhi, and N. Shuley, “A Small Dual-Polarized Receiving Antenna Array for Diversity/MIMO Systems,” Proc. Antennas and Propagation Society Int. Symp., pp. 329–332, 2006. [16] T. I. Lee and Y. E. Wang, “A Planar Multipolar Antenna for MIMO Applications,” Antennas and Propagation Society Int. Symp., pp. 2429–2432, 2007. [17] J. B. Yan, C. Y. Chiu, and R. D. Murch, “Handset 4-Port MIMO Antenna Using Slit Separated PIFA and Quarterwave-Slot Antenna Pair,” Antennas and Propagation Society Int. Symp., pp. 1–4, 2008. [18] P. Mookiah and K. R. Dandekar ,”Performance Analysis of Metamaterial Substrate Based MIMO Antenna Arrays,” Global Telecommunications Conference, pp. 1–4, 2008. [19] S. K. Hampel, O. Schmitz, and I. Rolfes, “MIMO and Diversity Performance of a Planar 2 × 2 Dipole Array Applying Sievenpiper HIS,” Proc. 1st European Wireless Technology Conference, Amsterdam, The Netherlands, pp. 326–329, October 2008. [20] Z. Jianwu, Z. Yangyang, and W. Yuting, “Compact Diversity Antenna Array For Mobile Terminals of MIMO System,” Proc. Asia-Pacific Microwave Conference, pp. 1–4, 2008. [21] S. Baek and S. Lim, “Compact Planar MIMO Antenna Array with Polarization Diversity on Single Layer,” Electronics Letters, Vol. 46, No. 13, pp. 880–882, June 24, 2010. [22] D. H. Margaret et al., “Mutual Coupling Reduction in MIMO Antenna System Using EBG Structures,” Int. Conf. on Signal Processing and Communications, pp. 1–5, 2012. [23] D. A. Ketzaki and T. V. Yioultsis, “Metamaterial-Based Design of Planar Compact MIMO Monopoles,” IEEE Trans. on Antennas and Propagation, Vol. 61, No. 5, pp. 2758–2766, May 2013. [24] M. S. Sharawi, Y. S. Faouri, and S. S. Iqbal, “Design of an Electrically Small Meander Antenna for LTE Mobile Terminals in the 800 MHz Band,” IEEE GCC Conference and Exhibition (GCC), Dubai, UAE , pp. 213–216, February 19–22, 2011. [25] Y. S. Shin and S. O. Park, “A Monopole Antenna with a Magneto-Dielectric Material and its MIMO Application for 700 MHz-LTE-Band,” Microwave and Optical Technology Letters , Vol. 52, No. 10, pp. 2364–2367, October 2010. [26] H. Bae et al., “Compact Mobile Handset MIMO Antenna for LTE 700 Applications,” Microwave and Optical Technology Letters , Vol. 52, No. 11, pp. 2419–2422, November 2010. [27] Y. Yu et al., “A Compact MIMO Antenna with Improved Isolation Bandwidth for Mobile Applications,” Microwave and Optical Technology Letters , Vol. 53, No. 10, pp. 2314–2317, October 2011.

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[28] M. S. Sharawi, Y. S. Faouri, and S. S. Iqbal, “Design and Fabrication of a Dual Electrically Small MIMO Antenna System for 4G Terminals,” Proc. 6th German Microwave Conference, Darmstadt, Germany, pp. 1–4, March 14–16, 2011. [29] J. Y. Chung et al., “Low Correlation MIMO Antenna for LTE 700MHz Band,” Int. Symp. on Antennas and Propagation, pp. 2202–2204, 2011. [30] M. S. Sharawi, S. S. Iqbal, and Y. S. Faouri, “An 800 MHz 2 × 1 Compact MIMO Antenna System for LTE Handsets,” IEEE Trans. on Antennas and Propagation, Vol. 59, No. 8, pp. 3128–3131, August 2011. [31] X. Zhao, Y. Lee, and J. Choi, “Design of a Compact Planar MIMO Antenna for LTE Mobile Application,” Proc. ISAP 2012, Nagoya, Japan, pp. 1365–1368, 2012. [32] D. Ga et al., “A MIMO Antenna with Improved Isolation Using RFC for LTE Mobile Application,” Antennas and Propagation Society Int. Symp., pp. 1–2, 2012. [33] D. Ga et al., “Design of MIMO antenna with Decoupling Network for LTE Mobile Application,” Proc. APMC 2012, Kaohsiung, Taiwan, pp. 705–707, December 4–7, 2012. [34] D. Ga, Y. Lee, and J. Choi, “Design of a Multi-Input-Multi-Output Antenna with improved isolation using RF Choke for Long-Term Evolution Mobile Application,” Microwave and Optical Technology Letters , Vol. 55, No. 7, pp. 1569–1574, July 2013. [35] S. Zhang et al., “Reduction of the Envelope Correlation Coefficient with Improved Total Efficiency for Mobile LTE MIMO Antenna Arrays: Mutual Scattering Mode,” IEEE Trans. on Antennas and Propagation, pp. 1–11, 2013. [36] S. B. Yeap et al., “Low Profile Diversity Antenna for MIMO Applications,” Electronics Letters, Vol. 42, No. 2, pp. 69–71, January 19, 2006. [37] Y. Gao et al., “Study of Dual-Element PIFA Array for MIMO Terminals,” Antennas and Propagation Society Int. Symp., pp. 309–312, 2006. [38] A. Diallo et al., “Study and Reduction of the Mutual Coupling Between Two Mobile Phone PIFAs Operating in the DCS 1800 and UMTS Bands,” IEEE Trans. on Antennas and Propagation, Vol. 54, No. 11, pp. 3063–3074, November 2006. [39] K. Chung and J. H. Yoon, “Integrated MIMO Antenna with High Isolation Characteristics,” Electronic Letters , Vol. 43, No. 4, pp. 199–201, February 15, 2007. [40] S. H. Chae, S. K. Oh, and S. O. Park, “Analysis of Mutual Coupling, Correlations and TARC in WiBro MIMO Array Antenna,” IEEE Antennas and Wireless Propagation Letters, Vol. 6, pp. 122–125, 2007. [41] Y. Gao, X. Chen, and C. G. Parini, “Channel Capacity of Dual-Element Modified PIFA Array on Small Mobile Terminal,” Electronic Letters , Vol. 43, No. 20, pp. 1060–1062, September 27, 2007. [42] S. Vergerio et al., “Design of Multiple Antennas at 5 GHz for Mobile Phone and its MIMO Performances,” Int. Conf. on Electromagnetics in Advanced Applications, pp. 17–20, 2007. [43] Y. Gao et al., “Design and Performance Investigation of a Dual-Element PIFA Array at 2.5 GHz for MIMO Terminal,” IEEE Trans. on Antennas and Propagation, Vol. 55, No. 12, pp. 3433–3441, December 2007.



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[44] Y. Ding et al., “A Four-Element Antenna System for Mobile Phones,” IEEE Antennas and Wireless Propagation letters, Vol. 6, pp. 655–658, 2007. [45] H. Moon and B. Lee, “3-Antenna Handheld MIMO Systems Evaluated with Full Scale EM Simulations,” Proc. Antennas and Propagation Society Int. Symp., pp. 2417–2420, 2007. [46] C. Peixeiro, “Integration Effects of Multiple Microstrip Patch Antennas into Small Handsets,” Proc. AFRICON, pp. 1–7, 2007. [47] D. J. Kim et al., “Compact 2-Channel MIMO Antenna for WiBro Handy Terminal Application,” Proc. Asia-Pacific Microwave Conference, pp. 1593–1596, 2006. [48] S. Vergerio, J. P. Rossi, and P. Sabouroux, “A Two-PIFA Antenna Systems for Mobile Phone at 2 GHz with MIMO Applications,” Proc. First European Conference on Antennas and Propagation, pp. 1–5, 2006. [49] E. R. Iglesias et al., “Performance of MIMO Systems Employing Multiple Compact Patch Antennas with Radiation Pattern Diversity,” Proc. First European Conference on Antennas and Propagation, pp. 1–6, 2006. [50] B. Lindmark et al., “Evaluation of MIMO Arrays using Antenna Patterns, Reverberation Chamber and Channel Measurements,” Proc. First European Conference on Antennas and Propagation, pp. 1–6, 2006. [51] J. H. Choi et al., “Experimental Investigation of 2 × 2 MIMO Array Antenna for the Mobile WiMax by Measuring 3D Radiation Patterns,” Proc. Antennas and Propagation Society Int. Symp., pp. 1–4, 2008. [52] S. K. Chaudhury, H. J. Chalooupka, and A. Ziroff, “Novel MIMO Antennas for Mobile Terminal,” Proc. 1st European Wireless Technology Conference, pp. 330–333, 2008. [53] C. J. Lee, A. Gummalla, and M. Achour, “Compact Dualband Antenna Subsystem for MIMO Application,” Proc. IEEE Int. Workshop on Antenna Technology, pp. 1–4, 2009. [54] C. C. Hsu et al., “Design of MIMO Antennas with Strong Isolation for Portable Applications,” Proc. Antennas and Propagation Society Int. Symp., pp. 1–4, 2009. [55] Y. Lee et al., “Design of a MIMO Antenna with Improved Isolation Using Meta-Material,” Proc. Int. Workshop on Antenna Technology, pp. 231–234, 2011. [56] Y. Li et al., “Dual-Polarised Monopole-Slot Co-Located MIMO Antenna for SmallVolume Terminals,” Electronic Letters , Vol. 47, No. 23, pp. 1259–1260, November 10, 2011. [57] S. J. Lee et al., “MIMO Antenna Using a Ground Plane Protruding for WCDMA Application,” Proc. Asia-Pacific Microwave Conference, pp. 574–577, 2011. [58] Z. Li et al., “Reducing Mutual Coupling of MIMO Antennas with Parasitic Elements for Mobile Terminals,” IEEE Trans. on Antennas and Propagation, Vol. 60, No. 2, pp. 473–481, February 2012. [59] S. W. Su and C. T Lee, “Highly Integrated, Two-In-One, Metal-Plate PIFA for 2.4-GHz Module Applications,” Microwave and Optical Technology Letters , Vol. 54, No. 10, pp. 2433–2438, October 2012.

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[60] I. T. E. Elfergani et al., “Design of a Compact Tuned Antenna System for Mobile MIMO Applications,” presented at Loughborough Antennas 7 Propagation Conference, Loughborough, UK, November 12–13, 2012. [61] W. Tang et al., “A New Compact MIMO Antenna Design with High Isolation for 1710~2700MHz,” Proc. 10th Int. Symp. on Antennas, Propagation & EM Theory, pp. 154– 158 , 2012. [62] L. Liu et al., “Design and Performance Investigation of a 2.6 GHz Dual-Element MIMO Antenna System for LTE Terminal,” Proc. 6th Asia-Pacific Conference on Environmental Electromagnetics, pp. 226–232, 2012. [63] M. S. Sharawi et al., “A CSRR Loaded MIMO Antenna System for ISM Band Operation,” IEEE Trans. on Antennas and Propagation, Vol. 61, No. 8, pp. 4265–4274, 2013. [64] M. U. Khan and M. S. Sharawi, “A 4-Element MIMO Antenna System Loaded with CSRRs and Patch Antenna Elements,” Proc. 7th European Conference on Antennas and Propagation, pp. 1944–1947, 2013. [65] M. U. Khan and M. S. Sharawi, “A Compact 8-Element MIMO Antenna System for 802.11ac WLAN Applications,” Proc. Int. Workshop on Antenna Technology, pp. 95–98, 2013. [66] M. S. Sharawi, “A 5-GHz 4/8-Element MIMO Antenna System for IEEE 802.11 AC Devices,” Microwave and Optical Technology Letters, Vol. 55, No. 7, pp. 1589–1594, July 2013. [67] G. S. Abo et al., “Hexaferrite Slant and Slot MIMO Antenna Element for Mobile Devices,” Microwave and Optical Technology Letters, Vol. 55, No. 3 7, pp. 551–554, March 2013. [68] D. W. Browne et al., “Performance of Integrated Antenna Arrays for MIMO Enabled Laptops,” Proc. Antennas and Propagation Society Int. Symp., pp. 2425–2428, 2007. [69] T. L. Roach, G. H. Huff, and J. T. Bernhard, “On the Applications for a Radiation Reconfigurable Antenna,” Proc. 2nd NASA/ESA Conference on Adaptive Hardware and Systems, pp. 7–13, 2007. [70] Y. S. Shin et al., “A Monopole Antenna with a Magneto-Dielectric Material and MIMO Applications,” IEEE Antennas and Wireless Propagation Letters, Vol. 7, pp. 764–768, 2008. [71] H. Li, J. Xiong, and S. He, “A Compact Planar MIMO Antenna System of Four Elements with Similar Radiation Characteristics and Isolation Structure,” IEEE Antennas and Wireless Propagation Letters, Vol. 8, pp. 1107–1110, 2009. [72] S. Park and C. Jung, “Compact MIMO Antenna with High Isolation Performance,” Electronics Letters, Vol. 46, No. 6, March 18, 2010. [73] I. H. Liu, S. Y. Lin, and Y. C. Lin, “Compact MIMO Antenna for Mini-Laptop,” Proc. Asia-Pacific Microwave Conference, pp. 833–836, 2011. [74] P. Y. Qin et al., “A pattern Reconfigurable U-Slot Antenna and Its Applications in MIMO Systems,” IEEE Trans. on Antennas and Propagation, Vol. 60, No. 2, pp. 516–528, February 2012. [75] A. Ghasemi et al., “A Reconfigurable Printed Monopole Antenna for MIMO Application,” Proc. 6th European Conference on Antennas and Propagation, pp. 1–4, 2012.



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[76] S. Yoo et al., “Compact MIMO Antenna of the Open-Loop and Meandered-Line 1-Layer Radiators with Improved Isolation,” Proc. Antennas and Propagation Society Int. Symp., pp. 1–2, 2012. [77] N. Honma et al., “Proposal of Compact Three-Port MIMO Antenna Employing Modified Inverted F Antenna and Notch Antennas,” Proc. Antennas and Propagation Society Int. Symp., pp. 2613–2616, 2006. [78] M. Han and J. Choi, “MIMO Antenna Using a Decoupling Network for 4G USB Dongle Application,” Microwave and Optical Technology Letters , Vol. 52, No. 11, pp. 2551–2554, November 2010. [79] B. Kim et al., “Isolation Enhancement of USB Dongle MIMO Antenna in LTE 700 Band Applications,” IEEE Antennas and Wireless Propagation Letters, Vol. 11, pp. 961–964, 2012. [80] A. Sulima et al., “A Modified Printed Inverted-F Antenna for Mobile Communications,” Proc. Antennas and Propagation Society Int. Symp., pp. 1–4, 2008. [81] A. C. K. Mak, C. R. Rowell, and R. D. Murch, “Isolation Enhancement Between Two Closely Packed Antennas,” IEEE Trans. on Antennas and Propagation, Vol. 56, No. 11, pp. 3411–3419, November 2008. [82] S. C. Chen, Y. S. Wang, and S. J. Chung , “A Decoupling Technique for Increasing the Port Isolation Between Two Strongly Coupled Antennas,” IEEE Trans. on Antennas and Propagation, Vol. 56, No. 11, pp. 3411–3419, November 2008. [83] C. J. Lee, M. Achour, and A. Gummalla, “Compact Metamaterial High Isolation MIMO Antenna Subsystem,” Proc. Asia-Pacific Microwave Conference, pp. 1–4, 2008. [84] L. C. Chang et al., “ A Polarization Diversity MIMO Antenna Design for WiMAX Dongle Application,” Proc. Asia-Pacific Microwave Conference, pp 762–765, 2010. [85] Z. Li, M. S. Han, X. Zhao, and J. Choi, “MIMO Antenna with Isolation Enhancement for Wireless USB Dongle Application at WLAN Band,” Proc. Asia-Pacific Microwave Conference, pp. 758–761, 2010. [86] S. W. Su and C. T. Lee, “Printed Two Monopole-Antenna System with a Decoupling Neutralization Line for 2.4-GHz MIMO Applications,” Microwave and Optical Technology Letters , Vol. 53, No. 9, pp. 2037–2043, September 2011. [87] V. Ssorin et al., “Compact Bandwidth-Optimized Two Element MIMO Antenna System for 2.5–2.7 GHz Band,” Proc. 5th European Conference on Antennas and Propagation, pp. 319–323, 2011. [88] W. S. Chen, C. H. Lin, and H. T. Chen, “A Compact Monopole Slot MIMO Antenna for Wireless USB Dongle Application at WLAN Band,” Proc. Asia-Pacific Microwave Conference, pp. 570–573, 2011. [89] V. Ssorin et al., “Compact 2.5–2.7 GHz Two Element MIMO Antenna System for Modern USB Dongle,” Proc. 6th European Conference on Antennas and Propagation, pp. 1955–1959, 2012. [90] J. Kwon et al., “Design of a MIMO Antenna for USB Dongle Application Using Common Grounding,” Proc. 13th Int. Conf. on Advanced Communication Technology, pp. 313–316, 2011.

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[91] Y. Liu et al., “Novel Miniaturized and High-Isolated MIMO Antenna,” Microwave and Optical Technology Letters , Vol. 54, No. 2, pp. 511–515, February 2012. [92] S. W. Su, C. T. Lee, and F. S. Chang, “ Printed MIMO-Antenna System Using Neutralization-Line Technique for Wireless USB-Dongle Applications,” IEEE Trans. on Antennas and Propagation, Vol. 60, No. 2, pp. 456–463, February 2012. [93] W. S. Chen, K. M. Lin, and F. S. Chang, “MIMO Antenna with Enhanced Isolation Elements for USB Dongle Applications,” Proc. Cross Strait Quad-Regional Radio Science and Wireless Technology Conference, pp. 48–51, 2012. [94] V. Sorin et al., “Compact Planar Inverted-F Antenna System for MIMO USB Dongle Operating in 2.5–2.7 GHz Band,” Proc. 42nd European Microwave Conference, pp. 408– 411, 2011.

5 Multiband Printed MIMO Antenna Systems Wireless terminals are usually backward compatible with previous generations, thus they are required to cover a wide frequency bandwidth or cover multiple frequency bands. For example, 4G wireless terminals are supposed to cover the new bands introduced for the 4G wireless standards while continuing to cover the 3G and 2G ones. This requires the use of wideband and multiband antennas in wireless devices. The recent 4G wireless standard (LTE) covers several frequency bands ranging between 698–960, 1427–1660, 1710–2200, 2300–2690, and 3400–3800 MHz. Although a single wireless handset will not cover all of the standard bands, it will cover many of them based on the country where the handset is being used and the service provider’s allocated spectrum. Also, within a certain country and service provider, multiple bands are usually supported and thus the need for multiple-band coverage becomes a must for antenna design. In this chapter, we present a wide range of printed multiband MIMO antenna systems for different wireless devices and applications. We will categorize the antennas based on their use into four categories as we did in Chapter 4. Specifically, we start by presenting multiband printed MIMO antenna systems for generic applications and wireless access points (usually ones of medium to large size), then we discuss antennas for cellular and mobile phone terminals. After that we discuss antennas for tablet and laptop computers followed by antennas for USB dongles. We finish the chapter by presenting some design guidelines that will help a designer develop his or her next multiband printed MIMO antenna system.

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5.1  Printed Multiband MIMO Antenna Systems for Wireless Access Points and Generic Applications Research on multiband and wideband printed MIMO antenna systems for wireless access points and other generic wireless applications such as RFID tags can be found in [1–19]. Most of the work has covered multiple bands, with a few demonstrating wideband operation such as those in [2, 3, 5, 8, 14, 19]. Although the design of printed multiband or wideband MIMO antennas is more challenging than that for single-band antennas, the design becomes even more challenging if one of the bands lies below 1 GHz. Out of the list of references for wireless access points mentioned, only two [6, 9] present multiband MIMO antenna systems with one of the covered bands being below 1 GHz. The challenge with such antennas comes from the fact that due to the larger wavelength of operation, the fields and radiation patterns couple strongly within adjacent antennas, so the correlation coefficient and isolation get degraded. Thus, novel antennas and isolation enhancement techniques are required to overcome such issues. Isolation enhancement methods will be discussed in Chapter 6. In this section we present several types of multiband MIMO antenna systems suitable for wireless routers and access points along with their MIMO performance metrics, types, and features. Table 5.1 classifies dual-band or wideband MIMO antenna systems suitable for wireless access point operation based on their antenna types. The table shows that the majority of the work focuses on printed monopole structures. Printed slots and dipoles are also of wide use. In [4], a triband MIMO antenna system with the three antenna elements each based on two printed loops was proposed for WLAN access points. The antenna covered the 2.4-, 5.2-, and 5.8-GHz bands with at least 80 MHz of bandwidth in each. Each antenna occupied a 10 × 20 × 40 mm3. Figure 5.1 shows the geometry and the fabricated model for a three-element WLAN triband MIMO antenna system. The details of the single dual-loop antenna Table 5.1 Dual-Band Wireless Access Point MIMO Antenna Types Antenna Type References Printed dipole [6, 11, 12] Printed loop [4, 9] Meander line [13] Metamaterial [1, 18] Printed monopole [2, 8, 16, 17] PIFA [7] Printed slot [3, 5, 19] Other [10, 14, 15]



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Figure 5.1  High-gain, dual-loop antenna for wireless access points: (a) configuration, (b) top view, and (c) fabricated model. (From: [4]. © 2010 IEEE. Reprinted with permission.)

element are shown in Figure 5.2. The three modes excited at the centers of each band are best analyzed via the surface currents induced on them. Figure 5.3 shows the surface currents on a single antenna element at the three center frequencies of the covered bands. In Figure 5.3(a), the surface currents at 2442 MHz are shown at the center of the 2.4-GHz band. Two current nulls are observed at the edges of the outer loop, thus the excited current corresponds to the 1λ loop mode (outer loop length around 130 mm at 2.44 GHz). In Figure 5.3(b), the currents on the inner loop exhibit similar behavior, indicating a 1λ loop mode at 5.25 GHz (with a loop size of 62 mm at 5.25 GHz). In this figure, the 2λ mode of the 2.4-GHz band is also visible via the presence of the four nulls on the outer loop shown by the mark (x). This mode will slightly affect the input impedance (i.e., the |S11| curve is not as good at 2.4 GHz) and will affect the gain patterns at 5.25 GHz.

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Figure 5.2  Details of the dual-loop antenna. (From: [4]. © 2010 IEEE. Reprinted with permission.)

Figure 5.3  Current distribution on the dual-loop antenna at three bands. (From: [4]. © 2010 IEEE. Reprinted with permission.)

At the upper 5-GHz band (the 5.8-GHz band), the inner loop shows high current (dominant mode) distribution at 1.5λ. This is visible in Figure 5.3(c). At 5.775 GHz, more nulls are also observed at the outer loop (six in total) suggesting a 2.5λ mode for the 2.4-GHz band. This mode slightly degrades the resonance performance and affects the gain patterns at the 5.8-GHz band. The measured S-parameters for the three-element triband MIMO antenna system are presented in Figure 5.4. The isolation was better than 15 dB through all bands. The envelope correlation coefficient curve is shown in Figure 5.5. Low correlation values are obvious. The reader should note that the curve is found from the S-parameter values without the efficiency included. The measured



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Figure 5.4  Measured S-parameters for the dual-loop antenna: (a) reflection coefficient and (b) coupling curves. (From: [4]. © 2010 IEEE. Reprinted with permission.)

Figure 5.5  Calculated envelope correlation coefficient. (From: [4]. © 2010 IEEE. Reprinted with permission.)

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radiation patterns at the three center frequencies are shown in Figure 5.6. The measured peak gains and measured efficiencies are plotted in Figure 5.7. Maximum gains were higher than 5 dBi in the 2.4-GHz band and higher than 6.9 dBi in the 5-GHz bands. The efficiency was better than 80% in all bands.

Figure 5.6  Measured 3D radiation patterns at three difference frequencies: (a) 2.442 GHz, (b) 5.25 GHz, and (c) 5.775 GHz . (From: [4]. © 2010 IEEE. Reprinted with permission.)

Figure 5.7  Measured peak gain and radiation efficiency. (From: [4]. © 2010 IEEE. Reprinted with permission.)



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A dipole-based dual-band MIMO antenna for wireless access points with one of the bands covered below 1 GHz was proposed in [6]. The two bands covered were 698–812 MHz and 2.12–2.84 GHz. The dual printed MIMO antenna system occupied an 80 × 80 mm2 area on a 0.8-mm-thick FR-4 substrate. The two antennas were perpendicular to each other to achieve polarization diversity. Each antenna was printed on one side of a dual-sided substrate with only 0.8-mm separation between them. Figure 5.8 shows the geometry of the dual element printed MIMO antenna. The fabricated model is shown in Figure 5.9. Antenna feeding was done by using a coaxial cable for each antenna (at each side of the two-layer board). The center conductor was connected to one arm of the dipole (i.e., A and C), while the outer conductor (i.e., GND) was connected to the other arm (i.e., B and D). The J-shape with the slot in the dipole is responsible for the lower band LTE operation, whereas the inverted S-shaped one is responsible for the higher band WiMAX operation. The parameters G1, G2, and W2 can be used to shift the higher resonance frequency. Figure 5.10 shows the measured S-parameters. Isolation of more than 20 dB was achieved due to the orientation of the two antennas. The measured radiation patterns are shown in Figure 5.11. The x-z plane and y-z plane cuts are shown at 0.75 and 2.5 GHz for antennas 1 and 2, respectively. Comparing the 0.75-GHz y-z plane cuts for antennas 1 and 2 shows the polarization diversity that can be achieved from this antenna. The maximum .

Figure 5.8  Geometry of the dipole-based, dual-band printed MIMO antenna for wireless access point applications: (a) top side and (b) bottom side. G1 = 2, G2 = 2, and W2= 5 mm. (From: [6]. © 2011 Wiley. Reprinted with permission.)

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Figure 5.9  The fabricated model of the dual-band printed MIMO antenna based on dipole elements. (From: [6]. © 2011 Wiley. Reprinted with permission.)

Figure 5.10  The measured S-parameters of the dual-band printed MIMO antenna based on dipole elements. (From: [6]. © 2011 Wiley. Reprinted with permission.)

measured gain was 1.07 dBi and –1.25 dBi in the low and high bands, respectively, with a minimum efficiency of 70%. Table 5.2 summarizes the features and MIMO metrics of this antenna system. The envelope correlation coefficient (ECC) was calculated using the S-parameters formula for a uniform multipath environment. A penta-band, four-element printed MIMO antenna for wireless routers was proposed in [9]. The geometry of this antenna is shown in Figure 5.12. Antenna elements 1 and 2 are printed meandered loop antennas covering the 699–798-MHz (LTE) and 2.2–2.4-GHz (WLAN/WiMAX) bands; elements 3 and 4 are also printed meandered loop antennas that cover the 1.7–2.0-GHz



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Figure 5.11  The measured radiation patterns of the dual-band printed MIMO antenna based on dipole elements: (a) antenna 1 and (b) antenna 2. (From: [6]. © 2011 Wiley. Reprinted with permission.)

(UMTS), 3.5–4-GHz (WiMAX), and 5–5.8-GHz (WLAN) bands. The multiband MIMO antenna occupied a 150 × 150 mm2 FR-4 board area with 0.8mm thickness. Two shorting strips were added to elements 1 and 2 to improve the matching performance and suppress unwanted resonances. Figure 5.13(a) shows the measured reflection coefficient curves for antennas 1 and 2, and Figure 5.13(b) shows the curves for antennas 3 and 4. The total efficiency and peak gain values in dBi are shown in Figure 5.14 for all of the covered bands. The minimum efficiency of antennas 1 and 2 in the lower band was 70%, while in the higher band, it was 58%. The maximum gain in the lower LTE band was 2.35 dBi, while in the higher WLAN/WiMAX it was 3.25 dBi. For elements 3 and 4, the minimum efficiency in the UMTS band was 74% with a maximum gain of 4.9 dBi. The minimum efficiency in the WiMax band was 61% with a maximum gain of 3.9 dBi. Finally, the minimum efficiency in the 5-GHz WLAN was 71% with a maximum gain of 5.3 dBi.

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Figure 5.11  (continued)

Table 5.2 MIMO Antenna Characteristics for the Antenna Shown in Figure 5.8 Actual Frequency Efficiency Efficiency MEG Diversity (GHz) ECC Antenna 1 Antenna 2 Ratio Gain (dB) 0.75 0.15 76.3 70.8 1.07 4.52 2.5 0.03 88.2 86.4 1.02 5.31 From: [6].

The maximum ECC was less than 0.1 in the bands of interest and was calculated based on the S-parameters without including the efficiency. Figure 5.15 shows the 3D radiation patterns obtained from the simulations for the two MIMO antenna elements. Good coverage around the antennas is achieved. Three examples of printed MIMO antenna systems for wireless access points have been discussed. The first covered printed loop antennas, the second



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Figure 5.12  The geometry of the printed MIMO antenna for wireless routers. (From: [9]. © 2013 IEEE. Reprinted with permission.)

covered printed dipoles, and the third covered meandered pointed loops. The last geometry to be considered in this application category is a printed slot MIMO antenna. A dual-band printed slot antenna suitable for access point applications and other generic applications due to its moderate size was presented in [19]. The antenna consists of a U-shaped path, a T-shaped monopole, and a ground plane with a pentagonal wide slot. Figure 5.16 shows the geometry of a single element of this MIMO antenna system. A single-element antenna occupied an area of 28 × 28 × 1 mm3 on an FR-4 substrate. The two bands covered by this antenna were WLAN (2.4– 2.485 GHz) and UWB (3.1–10.6 GHz). S-parameter curves with and without the T-shaped monopole are shown in Figure 5.17. The T-strip introduction is responsible for the WLAN resonance. To create an extra resonance below UWB, a λ/4 strip is added to the U-shaped patch. To design a resonance strip with a printed slot antenna, we can use:



λg

LTotal ≈ 4

εr + 1 2

(5.1)

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Figure 5.13  Measured and simulated S-parameters for the penta-band printed MIMO antenna: (a) elements 1 and 2 and (b) elements 3 and 4. (From: [9]. © 2013 IEEE. Reprinted with permission.)

where λg is the effective wavelength in the material, εr is the dielectric constant of the material, and LTotal is the total length of the loading strip. Based on (5.1), L 7 the T-shaped monopole length LTotal = L1 + 2 + L3 = 9 + + 4.5 is 17 mm in 2 2 this design. Four dual-element configurations were investigated in this work for MIMO diversity performance. Figure 5.18 shows the configurations investigated, and Figure 5.19 shows the measured isolation between the two antennas in each configuration. It is clear that configuration (d) shows the lowest coupling



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Figure 5.14  Measured efficiencies and maximum gain for the penta-band printed MIMO antenna: (a) elements 1 and 2, (b) elements 3 and 4. (From: [9]. © 2013 IEEE. Reprinted with permission.)

with values lower than –25 dB across the band of operation. The corresponding correlation coefficient value based on the simulated S-parameters and without the efficiency effect was less than 0.1. The measured radiation pattern for one element in configuration (d) is shown in Figure 5.20. The maximum gain measured was 5.5 dBi in UWB band and went to –2 dBi in the WLAN band. The overall efficiency was better than 85%.

5.2  Printed Multiband MIMO Antennas for Mobile Phones In this section, we focus on printed multiband MIMO antenna systems that are suitable for mobile wireless handsets and mobile phones. These phones have a

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Figure 5.15  Simulated 3D gain patterns for the printed dual-element MIMO antenna: (a) 750 MHz and (b) 2.4 GHz. (From: [9]. © 2013 IEEE. Reprinted with permission.)

Figure 5.16  Geometry of the single element for a printed slot MIMO antenna system. Wf = 1.86 mm, all dimensions are in mm [19].

standard backplane size of 100 × 50 mm2 and for smartphone applications it can go up to 120 × 60 mm2. Multiband operation of such antennas is essential, because they are supposed to cover legacy bands such as GSM (800 and 900 MHz) and PCS (1800/1900 MHz) and the new bands such as LTE (700 MHz) in addition to others such as UMTS (2100 MHz) and WLAN (2400 MHz). A major challenge in multiband MIMO antenna designs for mobile phone applications is the limited area designated for the antenna within the



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Figure 5.17  Simulated S-parameters showing the effect of the T-shaped monopole within the slot. (From: [19]. © 2013 Wiley. Reprinted with permission.)

Figure 5.18  Antenna orientations investigated for the dual-element MIMO antenna: (a) side by side, (b) parallel, (c) front to front, and (d) orthogonal. (From: [19]. © 2013 Wiley. Reprinted with permission.)

backplane as well as the inevitable high coupling problem between adjacent antenna elements when operating at bands below 1 GHz. Several antenna geometries have been proposed [20–39] for multiband operation covering one band below 1 GHz. Other multiband MIMO antenna systems that cover bands above 1 GHz have been proposed [40–65]. In these two categories (multiband MIMO antennas that cover bands above and below the 1-GHz band), several antenna geometries have been proposed. Table 5.3 shows the various geometries that have appeared in the literature for mobile

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Figure 5.19  Simulated mutual coupling for the various orientations. (From: [19]. © 2013 Wiley. Reprinted with permission.)

phone applications. PIFAs and printed monopoles or their derivatives are the most widely used type. In this section, we cover four different printed MIMO antenna implementations for mobile phone applications and discuss their operating principles, features, and performance metrics: (1) a PIFA MIMO antenna, (2) a printed monopole with magneto-dielectric material loading, (3) a 4-shaped MIMO antenna, and (4) a printed loop example. The first example is a multiband printed MIMO antenna system based on PIFA elements. Figure 5.21 shows the geometry of this antenna as proposed in [22]. The two PIFA elements are placed on the top and bottom edges of a 108- × 48-mm2 backplane to minimize coupling (have good isolation) and correlation values. A bandstop matching circuit is introduced at the corner of each antenna element to enhance the isolation in the lower band covering LTE standards (by suppressing the surface currents). Two resonances are introduced at 760 and 860 MHz. P1 is connected to the antenna, while P2 and P3 are floating. The folded patch along with the coupling between the feeding and shorting lines through slit 1 of this structure miniaturizes the antenna size at 800 MHz and widens its operating bandwidth. The size of slit 2 controls the higher frequency band. The bandstop inductor value (LI) can be used to tune the center frequency at the lower LTE band. The single antenna size is 36 × 12 × 6.2 mm3 and is placed on an FR-4 material backplane that is 0.8 mm thick. The antenna covers the 740–980-MHz (LTE, GSM) and 1.81–2.69-GHz bands (GSM, WCDMA, M-WiMAX, and WLAN). Figure 5.22 shows the measured S-parameters for the dual-element multiband MIMO antenna. The low and high operating bands are clearly shown. The obtained isolation was better than 9 dB in the lower band, and better than 20 dB in the higher one when the bandstop circuit was added. Figure 5.23 shows the measured radiation patterns of the two antenna elements at two bands, 780 MHz and 2.55 GHz. The ECC was calculated based on (2.13) with



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Figure 5.20  Measured radiation patterns for the orthogonal configuration of the dual-element printed slot MIMO antenna system: (a) 2.4 GHz, (b) 4 GHz, and (c) 7 GHz. (From: [19]. © 2013 Wiley. Reprinted with permission.)

100% efficiency assumption and the MEG based on (2.14). ECC values lower than 0.4 are achieved. The diversity metrics for this antenna are summarized in Table 5.4 for the case with the bandstop circuit. Note the low efficiency in the lower band. The

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Printed MIMO Antenna Engineering Table 5.3 Multiband Printed Antennas for Mobile Phone Applications Antenna Type References Printed dipole [24, 59, 62] Printed loop [53, 58, 64] PIFA [22, 23, 25, 40–44, 51, 63, 65] Meander line [48, 50, 52, 55] Metamaterial [20, 21] Printed monopole [26, 35, 39, 47, 54, 56, 57, 60, 61] Printed slot [36, 45] Other [27–34, 37, 38, 46, 49]

ratio of the two MEG values for the two antennas is almost unity, indicating that the mean power delivered for the two antennas in a uniform propagating environment are almost the same. It is worth noting that although the design provides good bandwidth coverage at the low and high bands, the size of the antenna element is large (especially in its height above the backplane), which limits its practical use in a real product. The second example is a printed monopole-based MIMO antenna system that is loaded with a magneto-dielectric (MD) material. MD materials can be used for antenna miniaturization, but they need to have high permeability and low magnetic loss in the band of interest. A Y-type hexagonal ferrite (Ba2CO2Fe12O22) was used in [26] for miniaturization purposes. The complex relative permittivity (µr) curves covering from 500 to 3000 MHz are shown in Figure 5.24. It is clear that the real part of µr is constant with a value of 2.1, while the real part material permittivity is 12.7. The printed monopole loaded with an MD material geometry is shown in Figure 5.25. The configuration of the MIMO antenna as well as the fabricated model are shown. The size of the cavity loaded with the MD material is 18 × 8 × 3 mm3. The arm lengths L1 and L2 were 5 and 8 mm, respectively. To show the effect of MD material loading on the resonance of the antenna, the same cavity was loaded with a dielectric material with εr = 13. This model was denoted as (ANTDL). Figure 5.26 shows the measured return loss for the MD loaded monopole (ANTMD) and the dielectric loaded one (ANTDL). The lower resonance is obvious in the case of the MD loaded antenna. The dielectrically loaded antenna monopole length L2 was extended to 17 mm (from 8 mm) to have it resonate at the same frequency as the MD one (ANTDR). This is a clear miniaturization effect. The MD antenna covered the 698–751- and 1656–2171-MHz bands. To improve the isolation between the two adjacent elements, a neutralization line (NL) was added [Figures 5.25(b) and (c)]. The line lengths L3, L4, L5, and L6 were 4, 7, 13, and 6 mm, respectively. The effect on the S-parameters is



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Figure 5.21  Dual-element multiband PIFA geometry: (a) 3D view, (b) top view, and (c) antenna element structure details. (From: [22]. © 2010 Wiley. Reprinted with permission.)

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Figure 5.22  Measured S-parameters of the dual-element, PIFA–based, MIMO antenna: (a) without bandstop matching and (b) with bandstop matching. (From: [22]. © 2010 Wiley. Reprinted with permission.)

shown in Figure 5.27. The minimum isolation obtained was 8 dB in the lower band. More than 10-dB isolation enhancement was observed in the higher band with the NL. The introduction of a NL improves the isolation within a single band as will be shown in Chapter 6. The minimum efficiency was recorded in the lower bands due to the MD material loading, and it was 33% with a maximum peak gain of –0.03 dBi. The ECC was worse in the LTE band with values close to the limit (0.48).



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Table 5.4 MIMO Parameters Table for the Dual-Element DualBand PIFA with Bandstop Matching Circuit Frequency Efficiency Efficiency (GHz) Antenna 1 Antenna 2 MEG Ratio 0.78 52.86 52.2 1.01 1.92 80.75 80.04 1.01 2.55 64.43 64.34 1.02 From: [22].

(a)

(b) Figure 5.23  Measured radiation patterns of the dual-element, PIFA-based MIMO antenna: (a) 780 MHz and (b) 2.55 GHz. (From: [22]. © 2010 Wiley. Reprinted with permission.)  

The third example presents a 4-shaped printed MIMO antenna for mobile phone applications. A two element 4-shaped printed MIMO antenna

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Figure 5.24  Measured complex property curves (µr = µr′ – jµr″, εr = εr′ – jεr″) for the MD material. (From: [26]. © 2012 IEEE. Reprinted with permission.)

geometry with enhanced isolation using capacitively loaded loops (CLL) is presented in [34]. The geometry and fabricated model of the MIMO antenna are shown in Figure 5.28. The antenna dimensions were (all in mm): W = 50, L = 100, Wt = 2.2, H = 6.9635, L1 = 35.3, L2 = 26, Ys = 5.5, Xa2 = 4.0716, Lf = 14.7, Xs = 0.3716, Xf = 1.5716, Wf = 2.5, Ws = 1, Wtr = 1.4, W50 = 3, Yf 1 = 13.35, Yf  2 = 13.5, Yf  3 = 28.65, UE_W1 = 5.727, UE_L1 = 5.8, L3 = 35.8, W1 = 12.2, UE_W = 9, UE_L = 8.927, Y1 = 64.127, Y = 44, W2 = 15.9, Y_dist = 0.273, Gap = 0.127, X_dist = 0.2, Wtrace = 0.25, Y_dist1 = 0.2, X_dist1 = 0.127, GAP1 = 0.36, and Wtrace1 = 0.2. The antenna was fabricated on a 100 × 50 × 1.56mm3 FR-4 substrate. Microstrip impedance transformers were used for proper impedance matching at the higher band. CLLs were used to enhance the isolation performance between the two adjacent elements at the lower band and the higher band. Two CLL arrays were used for the dual-band operation, a top layer one for the higher band (with lower CLL sizes) and a complementary CLL one (etched out from the GND plane) on the bottom layer of the PCB (i.e., on the GND layer) that enhances the isolation in the lower band. The two CLL structures are metamaterial ones, and their characteristics were modeled before being applied. The single cell size on the top layer was 5.727 × 5.8 mm2 and on the bottom layer was 9 × 8.927 mm2. The improvement in the isolation was more than 10 dB in the lower band of operation and more than 3 dB improvement in the higher band. The measured S-parameters for the printed MIMO antenna system are shown in Figure 5.29. The lower operating band covered 827 to 853 MHz and the higher band covered 2.3 to 2.98 GHz. Note that the interelement spacing between the two



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Figure 5.25  Geometry of the two-element MIMO antenna system with MD loaded printed monopoles: (a) single antenna dimensions, (b) top view of the two-element MIMO antenna, (c) neutralization line (NL) for isolation improvement, and (d) fabricated antenna within a real populated board. (From: [26]. © 2012 IEEE. Reprinted with permission.)

antennas is less than λ/15 in the lower band; thus, large coupling between the two elements is expected without using the isolation enhancement structure. The measured gain patterns at 840 MHz are shown in Figure 5.30 for both the x-z and y-z plane cuts. The two element patterns are shown with their copolarized and cross-polarized ones. The maximum measured gain in the low band was –2.8 dBi, whereas in the high band it was 5.5 dBi. The efficiency in the low band was 35% due to the use of larger size CLLs and the GND plane defect, while it was approximately 67% in the higher band. The correlation coefficient was evaluated using the S-parameters formula of (2.13) including the effect of efficiencies. The values were 0.11 and 0.18 for the low and high bands, respectively. The TARC curves for different input signal angles were evaluated based on the measured S-parameters and using (2.11).

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Figure 5.26  Measured return loss curves for the MD loaded printed monopole MIMO antenna. (From: [26]. © 2012 IEEE. Reprinted with permission.)

Figure 5.27  Effect of the NL on the return loss and isolation of the MD loaded monopole MIMO antenna system. (From: [26]. © 2012 IEEE. Reprinted with permission.)

Figure 5.31 shows the TARC curves for angle steps of 30 degrees for both the low band and high band. Note the stable behavior in the low band due to the very good isolation obtained (more than 19 dB) compared to the higher band curves that were more affected by the port phase differences due to the lower isolation levels (around 11-dB minimum isolation). The calculated MEG values were with 1 dB from one another in both bands and for XPD values of 0 and 6 dB. A four-element printed MIMO antenna based on 4-shaped elements was presented in [29] for mobile phones and other small wireless devices.



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Figure 5.28  Dual-element printed MIMO antenna system based on 4-shaped antennas: (a) geometry and (b) fabricated prototype. (From: [34]. © 2013 PIER. Reprinted with permission.)

The channel capacity of this two-element printed dual-band MIMO antenna system was measured using a software-defined radio (SDR) platform. The entries of the channel matrix (H) that are required to calculate the channel capacity of the system are complex numbers that contain the gain and phase

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Figure 5.29  Measured S-parameters of the MIMO antenna system based on 4-shaped antennas: (a) lower band and (b) higher band. (From: [34]. © PIER 2013. Reprinted with permission.)

information between the propagation paths from the transmitter to the receiver. The measured H matrix is then substituted in (2.24) to find the channel capacity in the environment under consideration. The H matrix was measured in indoor LOS and NLOS environments in the electrical engineering hallways at King Fahd University of Petroleum and Minerals (KFUPM) Saudi Arabia.. The SDR platform had a Xilinx Vertix 4 FPGA processing unit, eightchannel ADC/DAC, and quad RF modules that operated at the 2.4-GHz band. To measure the channel coefficients, a transmission burst was sent from one



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Figure 5.30  Measured gain patterns at 840 MHz of the MIMO antenna system based on 4-shaped antennas: (a) x-z plane and (b) x-y plane. Dots = copol element 1, circles = copol element 2, solids = cross-pol element 1, and dashes = cross-pol element 2. (From: [34]. © 2013 PIER. Reprinted with permission.)

channel of the transmitter with all other channels silent. From the data received at both antenna inputs of the receiver (since it is a two-element MIMO system), the complex valued entries of the channel matrix were calculated. The process was repeated using the second channel of the transmitter, and a complete H matrix was calculated. An average H matrix was obtained after performing 1,000 such measurements and averaging them. The measurements were carried out in a 10-ft-wide, 8-ft-high corridor with walls of concrete and a conducting sheet for the ceiling. The measurement setup is shown in Figure 5.32. In the LOS scenario, the transmitter and receiver were separated by an average distance of 20 ft. In the NLOS scenario, the receiver was moved around the edge of a wall. The distance of the transmitter from the edge was 20 ft while for the receiver it was 5 ft. The layout of measurement scenarios is shown in Figure 5.33. Six sets of measurements were conducted, three for the LOS case and three for the NLOS case. The effect of the isolation mechanism on the channel capacity was accessed for the antennas that appeared in [28, 30, 34]. For each measurement, the same antenna was attached to the receiver and transmitter. A total of 25 realizations of the H matrix were obtained for each case by moving the transmitter slightly around its position. The H matrix was averaged and normalized to remove the affects of path loss variations. Figure 5.34 shows that in both LOS and NLOS scenarios, at least a 1 bps/ Hz improvement is achieved when isolation is improved by 3 dB between the two antenna elements at 2.45 GHz at an SNR level of 20 dB. In addition, the MIMO advantage is evident if we compare the measured channel capacities at 20 dB SNR that show about 8 bps/Hz with those of a SISO system that gives approximately 6.5 bps/Hz. The channel capacity values for the NLOS were 1 bps/Hz higher than the LOS ones. This is due to higher propagation-induced

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Figure 5.31  TARC curves for the two-element MIMO antenna system based on 4-shaped antennas: (a) low band and (b) high band. (From: [34]. © PIER 2013. Reprinted with permission.)

correlation in the LOS channels compared to their NLOS counterparts. The reason for the difference between the ideal channel capacity performance of a 2 × 2 MIMO antenna system and the one measured can be explained taking several factors in mind. First, the printed MIMO antennas have limited efficiency, which was approximately 65% compared to 100% in the ideal case. Second, the channels are assumed to be totally uncorrelated in the ideal limit, but in the case of our antennas, they seem to be more correlated and thus the MIMO performance was degraded. But even with these two major effects, the system achieves more than the theoretical SISO limit (which in a real system is usually also not met). The final example in this section presents a two-element printed MIMO antenna system based on open loops [58]. The basic antenna geometry and its parameters are shown in Figure 5.35. A single element that is shown in Figure 5.35(a) consists of an outer open loop responsible for the broadband



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Figure 5.32  Channel capacity measurement setup using an SDR platform at KFUPM.

Figure 5.33  Measurement scenario layouts: (a) LOS and (b) NLOS.

behavior in the lower band covering 1.7 to 2.7 GHz. The outer loop acts as a λ/2 dipole and is excited via coupling from the inner loop. The inner open loop is excited via the microstrip line feed and acts as two monopoles, ABCDEF and ABG. The longer monopole is responsible for the lower frequency of the upper band covered, while the shorter one (ABG) is responsible for the resonance at the higher frequency part of the upper band. The combination of

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Figure 5.34  Measured channel capacity curves: (a) LOS and (b) NLOS.

the two monopole resonances along with the higher order modes of the outer loop lead to the wideband coverage at the higher band ranging from 4.7 to 8.5 GHz. The overall size of the two-element printed MIMO antenna systems was 63 × 46 × 0.8 mm3, and was fabricated on a CEM-1S3110 substrate (εr = 4.4). The current distribution at the three frequency points 1.6, 2.0, and 2.6 GHz demonstrates the resonance behavior of the single antenna as shown in Figure 5.36. The outer loop is excited at 2.6-GHz operation as both loops show high current levels. Improved isolation between the two MIMO antenna elements is achieved by using two U-shaped slots in the GND plane as shown in Figure 5.35(b). The inner slot was used for the higher frequency isolation, while the outer one was used for the lower band. The depth of the slot was optimized using numerical



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Figure 5.35  Two-element, dual-broadband printed MIMO antenna system based on open loops: (a) geometry of a single element and the board stack and (b) geometry of the twoelement MIMO antenna. (From: [58]. © 2012 IEEE. Reprinted with permission.)

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Figure 5.36  Current distribution at the resonance frequencies: (a) 1.6 GHz, (b) 2 GHz, and (c) 2.6 GHz. (From: [58]. © 2012 IEEE. Reprinted with permission.)

tools, and was close to λ/4 at the lower band. Figure 5.37 shows the current distribution on the two antenna elements with and without the isolation Ushaped slots. The coupling reduction on element 2 from element 1 is clear in Figure 5.37(b) compared to Figure 5.37(a). The measured S-parameter curves are shown in Figure 5.38. The isolation was better than 15 dB in the lower band and better than 20 dB in the higher one. The measured peak gain and efficiencies are plotted in Figure 5.39. The peak gain was approximately 3 to 4 dBi in the lower band and between 2 and 7 dBi in the higher band. The efficiency was higher than 70% across the complete band of operation. The measured ECC was less than 0.1 across the complete band of operation. The capacity loss is another measure of the improvement in the channel

Figure 5.37  Current distribution with and without the isolation structure at 2.1 GHz: (a) without U-shaped structure and (b) with U-shaped structure. (From: [58]. © 2012 IEEE. Reprinted with permission.)



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Figure 5.38  Measured S-parameter curves for the dual-broadband, two-element printed MIMO antenna based on open loops with the U-shaped isolation structure. (From: [58]. © 2012 IEEE. Reprinted with permission.)

Figure 5.39  Measured peak gain and efficiency of the dual-broadband, two-element printed MIMO antenna based on open loops with the U-shaped isolation structure. (From: [58]. © 2012 IEEE. Reprinted with permission.)

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capacity of a MIMO system. While we want to increase the channel capacity, we want the capacity loss to be almost zero to achieve that. Because there is a correlation between the MIMO channels and antennas, the capacity loss is larger than zero. The capacity loss can be calculated from the measured S-parameters in the case of high SNR values and a uniform multipath environment using [58]

( )

C (loss ) = - log 2 det ΨR

(5.2)

where ΨR is a 2 × 2 correlation matrix with the following entries:



(

Ψii = 1 - Sii

(

2

+ Sij

2

)

Ψij = - Sii * Sij + S ji * S jj

)



(5.3)

The capacity loss curves are shown in Figure 5.40 for the two-element open-loop MIMO antenna system. The obtained capacity loss is less than 0.3 bps/Hz over the whole band, showing good diversity performance. Note that using the S-parameters only for such a calculation is rather optimistic because the ECC calculation based on the S-parameters is valid when the elements are

Figure 5.40  Capacity loss curves of the dual-broadband, two-element printed MIMO antenna based on open loops with the U-shaped isolation structure. (From: [58]. © 2012 IEEE. Reprinted with permission.)



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100% efficient and operating in an isotropic environment, and usually this is not the case. Thus, ECC values should be obtained using the radiation patterns or from a reverberant chamber for more accurate evaluation and benchmarking.

5.3  Multiband Printed MIMO Antennas for Portable Computers Integrated printed antennas are essential parts of mobile computers such as laptops and tablets. Multiband operation is highly desirable because it reduces the number of antennas required to cover several wireless standards. For MIMO system operation on portable computers, the size limitation that is posed in mobile phones is not there anymore and thus lower correlation levels are obtained due to the spatial separation between the MIMO antenna elements when placed on the two corners of a laptop or tablet screen. The amount of work covering printed MIMO antennas for portable computers is not as much as that for mobile phone and smartphone applications. The most recent work in multiband printed MIMO antennas for laptop and tablet PCs is evident in [66–83]. Due to the abundance of space, the use of multiband MIMO antennas that cover frequency bands lower than 1 GHz is common [68–73, 83]. Table 5.5 classifies the antenna types that appeared in the literature for this application. PIFAs and printed monopole and printed loop antennas dominate the antenna types for multiband operation in MIMO antenna systems for portable PCs. In this section we present three different antenna types for this application. The reader is encouraged to check other types as shown in Table 5.5 according to the design requirements. The first example is a quad-band PIFA-based MIMO antenna system covering the DVB-H UHF band (370–870 MHz), the L band (1300–1520 MHz), the WLAN bands (2.4–2.5 GHz), and the 5.15–5.825-GHz band [68]. One antenna element consisting of a self-complementary meandered PIFA covered the DVB-H UHF band and two other PIFA elements were used for the dual-band WLAN MIMO antenna. Figure 5.41 shows the geometry and dimensions of the two antenna structures. Table 5.5 Antenna Types for MIMO PC and Tablet Applications Antenna Type References Printed loop [72, 73, 83] PIFA [68, 69, 75–77] Printed monopole [64, 67, 78, 81, 82] Patch [74, 79] Other [70, 71, 80]

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Figure 5.41  Geometry of the quad-band MIMO antenna for large handheld devices: (a) dualbroadband antenna for DVB-H and (b) two-element dual-band MIMO antenna for WLAN applications, where Fg= 2.85, L5 = 17.15, L6 = 12, W1 = 2 (all dimensions in mm). (From: [68]. © 2010 IEEE. Reprinted with permission.)

The overall size of the DVB-H UHF antenna was 70 × 30 mm2. The meandered structure is used to increase the electrical length and this help in lowering the operating frequency. The self-complementary structure (SCS) is used to broaden the impedance bandwidth. The width of the shorting line width (L3 = 7.5 mm) increases the bandwidth of operation. The size of a single-element WLAN MIMO antenna was approximately 203 × 2.5 mm2. The stub (S = 1.85 mm) along with trapezoidal ground strip connection on the bottom layer improves the antenna matching as shown in Figure 5.41(b). The size of the substrate ground plane was 72 × 101 mm2 made



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of FR-4 material with a 1-mm thickness. Three mechanisms were devised to improve the isolation between the WLAN antennas. First, the two antennas were oriented 90° with respect to each other to have orthogonal radiation and minimize the coupling. Second, a NL was used [called the connecting line in Figure 5.41(b)] to improve the isolation. Finally, the trapezoidal conducting plane on the bottom layer improved the matching as well as the isolation at the 5-GHz band. The two antenna systems were placed on an i-Station U43, Digital Cube mockup, as shown in Figure 5.42 and tested. Figure 5.43 shows the measured S-parameters and highlights the bands of operation. Note that the 4:1 VSWR is used for the DVB-H band. The isolation between the DVB antenna and that of the WLAN was higher than 10 dB across WLAN bands. The measured gain patterns at DVB-H, L, WLAN 2.4 GHz, and WLAN 5 GHz are shown in Figure 5.44. The maximum gain at 660 MHz was –1.98dBi and –5.5 dBi at 1.47 GHz. At 2.45 GHz the measured peak gains for elements 1 and 2 were 0.6 and –0.5 dBi, respectively. The values at 5.45 GHz were 3.75 and 3.9 dBi, respectively. The patterns show some orthogonal behavior toward their peak value locations (i.e., 0°, 180°, and 270°). Other MIMO parameters such as the correlation coefficient and TARC were not presented in this work, but based on the good isolation and assuming efficiencies higher than 60%, the ECC values are expected be lower than 0.5. The second example is an UWB patch-based MIMO antenna system for PDA applications [74]. The two-element printed MIMO antenna system covers the bands from 2.2 to 11.2 GHz. The antennas are placed on the top corners of an 82 × 80 mm2 ground plane. Each square folded patch had a volume of 10 × 12 × 2 mm3. Figure 5.45 shows the geometry of the two-element MIMO

Figure 5.42  Fabricated quad-band antenna with mockup case. (From: [68]. © 2010 IEEE. Reprinted with permission.)

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Figure 5.43  Measured S-parameters for the bands of operation. (From: [68]. © 2010 IEEE. Reprinted with permission.)

Figure 5.44  Measured radiation patterns for the bands of operation: (a) 660 MHz, (b) 1470 MHz, (c) 2.45 GHz, and (d) 5.45 GHz. (From: [68]. © 2010 IEEE. Reprinted with permission.)

antenna system and the single-element dimensions. The bandwidth of the antenna is improved (widened) when using the connecting strip with width (Cw = 2 mm) as shown in Figure 5.45(b).



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Figure 5.45  Geometry of an UWB MIMO antenna for PDA applications: (a) top view and (b) detailed geometry of the folded patch, where ST = 19, LS1 = 15, LS2 = 9, Cw = 2 (all dimensions in mm). (From: [74]. © 2010 Wiley. Reprinted with permission.)

The two T-shaped stubs that are symmetrically placed from the center of the GND plane are used for isolation improvement across the band of operation. Their effect is evident in Figure 5.46 where the current distribution is shown for the cases of no stubs and with stubs at two frequencies, 4 and 8 GHz. Strong current levels are observed on element 2 when element 1 is excited when no stubs are used, while almost no visible current levels are observed when the stubs are used at both bands of operation (i.e., 4 and 8 GHz in Figure 5.46). The measured S-parameter curves are given in Figure 5.47. Isolation levels lower than 25 dB are maintained across the UWB range. The measured radiation patterns are shown in Figure 5.48 for the principal plane cuts at 4 and 8 GHz. The maximum measured gain values across the operating band are shown in Figure 5.49 with values ranging between 3.2 and 4.7 dBi. The calculated ECC values based on the measured S-parameters and without the efficiency were lower than 0.12 over the complete band of interest. The last example in this section presents a reconfigurable printed loopbased MIMO antenna system covering LTE bands 3 (centered at 1.8 GHz) and 7 (centered at 2.6 GHz) [83]. A PIN diode is used to switch between the two LTE bands covered. In addition, a slot antenna covering the DTV bands between 496 and 862 MHz is integrated within the same board structure. Figure 5.50 shows the geometry of the presented triband antenna as well as a close-up view of the reconfigurable printed loop element of the two-element

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Figure 5.46  Effect of the T-shaped stubs on the current distribution: (a) 4 GHz and (b) 8 GHz. (From: [74]. © 2010 Wiley. Reprinted with permission.)

MIMO antenna system. The size of a single loop antenna was 50 × 10 mm2 on a 0.8-mm FR-4 substrate with εr = 4.6. The triband antenna system was built on a 150 × 150 mm2 board. Antenna 1 in Figure 5.50(a) is the DTV antenna; antennas 2 and 3 are the reconfigurable MIMO loop antennas. Figure 5.50(b) shows the two paths responsible for the two band operations of the reconfigurable loop. When the PIN diode is OFF, the current flows in the outer loop (WXYZ), thus resonating at 1.8 GHz (λ loop). When the PIN diode is switched ON, the majority of the current flows within the inner loop (WXQ) and provides resonance at a 2.6-GHz center frequency. An RF choke inductor along with a 47Ω resistor are used to bias the PIN diode. A 20-pF capacitor is used to block the DC from the RF input of the loop.



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Figure 5.47  Measured S-parameters of the UWB two-element MIMO antenna system with T-shaped stubs. (From: [74]. © 2010 Wiley. Reprinted with permission.)

The fabricated model is shown in Figure 5.51. The measured S-parameters are presented in Figure 5.52. For the DTV bands, antenna tuning (matching) was achieved by increasing the width of the shorting strip via a copper strip addition [Figure 5.52(a)]. A frequency shift in the response of the reconfigurable loop was observed with and without the addition of the biasing cables [Figure 5.52(b)]. This is because of the effect of the nonideal components used in addition to the differences in the biasing cable lengths used that were running from the battery to the bias points. The measured and simulated isolation curves for the MIMO antenna system are shown in Figure 5.53. Isolation levels are well below –10 dB in the band of interest. The discrepancies between the measured and simulated values might be attributed to the differences between the simulated and fabricated material properties and component values with tolerances. The measured gain patterns at 1.83 and 2.61 GHz for the MIMO antenna system are shown in Figure 5.54 and 5.55, respectively. The orthogonal behavior of the pattern is clear in the y-z cuts for elements 2 and 3 [Figures 5.54(b) and (d) for 1.83 GHz and Figures 5.55(b) and (d) for 2.61 GHz]. The maximum gain obtained was between 4.5 and 6 dBi in both bands of the antenna system with a minimum efficiency of 55%. The ECC minimum was 0.25 in both bands based on the measured S-parameters and without the efficiency effect incorporated.

5.4  Multiband MIMO Antennas for USB Dongle Applications The widespread of USB dongle wireless Internet connectivity devices operating within the LTE and WiMax standards is an indicator of the importance of coming up with novel multiband printed MIMO antenna systems that can fit with

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Figure 5.48  Measured radiation patterns of the UWB two-element MIMO antenna system with T-shaped stubs: (a) 4 GHz and (b) 8 GHz. (From: [74]. © 2010 Wiley. Reprinted with permission.)

the small size of a USB dongle, which typically has a 70 × 30-mm2 backplane. Due to the very small size compared to a cellular phone backplane, the antenna types that have appeared in the literature for multiband MIMO antenna systems for USB dongles is rather limited [84–91]. PIFAs dominated the antennas used within this application category [84, 85, 88, 91]. Printed monopoles come second [89, 91]. Two other types of antenna are a downside-discone antenna with a planar structure [86] and a new fork-shaped MIMO antenna [87]. Out of the few geometries that appeared, only two [84, 85] that were based on PIFA



Multiband Printed MIMO Antenna Systems

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Figure 5.49  Measured peak gain of the UWB two-element MIMO antenna system with Tshaped stubs across the band of operation. (From: [74]. © 2010 Wiley. Reprinted with permission.)

elements covered a band that is lower than 1 GHz. This indicates that this area needs more innovative ideas and antenna structures to be built. Two examples are presented in this section. One is based on a PIFA element that covers frequency bands lower than 1 GHz [85], while the second is based on an UWB monopole [89]. The geometry of a multiband printed PIFA-based MIMO antenna system for USB dongle applications is shown in Figure 5.56 [85]. The fabricated prototype is shown in Figure 5.57. The overall size of the backplane used was 66 × 55 × 0.5 mm3. The antennas were raised by 5.2 mm above the backplane to a height of 6 mm. The two PIFAs with a symmetric slotted structure were placed at the two corners of the USB dongle. The first band covering 754–787 MHz was obtained due to the main radiating strip with the coupling slot with a total length of approximately 70 mm (0.18λ at 770 MHz). The higher band covering 2.5–2.62 GHz is tuned and controlled by the slot width (WL = 2 mm) and port 1 position. To improve the isolation, a joined neutralization (shorting) line between the two elements is introduced. The symmetric slotted structure also played a role in improving the isolation in the lower band. Figure 5.58 shows the measured S-parameters in the low and high bands. The isolation across the complete operating bands was better than 15 dB. The measured radiation patterns for the two elements are shown in Figure 5.59 for two frequencies, 770 MHz and 2.55 GHz. It is clear that polarization diversity is achieved in the y-z plane because a 90o pattern tilt is observed. This will provide good correlation coefficient levels (i.e., low values).

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Figure 5.50  Geometry of the reconfigurable MIMO antenna for a DTV player: (a) top and side views and (b) detailed antenna geometry. (From: [83]. © 2013 IEEE. Reprinted with permission.)

The maximum measured gain in the lower band was 0.12 dBi and in the higher band was 3.3 dBi. The maximum calculated ECC value was 0.2 and the minimum efficiencies of the two antennas were 33% and 68% in the lower and higher bands, respectively. In addition, MEG and measured diversity gain values were obtained from a reverberant chamber. The MEG ratios for antenna elements 1 and 2 were almost equal in both bands (i.e.,