IEEE MTT-V055-I02 (2007-02B) [55, 2b ed.]

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FEBRUARY 2007

VOLUME 55

NUMBER 2

IETMAB

(ISSN 0018-9480)

PART II OF TWO PARTS

SPECIAL ISSUE ON APPLICATIONS OF FERROELECTRICS IN MICROWAVE TECHNOLOGY Guest Editorial .... ......... ........ ......... ......... ........ ......... ......... ........ ......... .... R. A. York and S. Gevorgian

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PAPERS

Improving Linearity of Ferroelectric-Based Microwave Tunable Circuits ....... ......... ......... ........ ......... ......... .. .. ........ ......... ......... ........ ......... ......... ........ ......... ... J.-S. Fu, X. A. Zhu, J. D. Phillips, and A. Mortazawi A Low-Noise -Band VCO Based on Room-Temperature Ferroelectric Varactors ..... ......... ........ ......... ......... .. .. ........ ......... ......... ........ ......... ......... ........ ...... M. Norling, A. Vorobiev, H. Jacobsson, and S. Gevorgian A Ferroelectric-Capacitor-Based Tunable Matching Network for Quad-Band Cellular Power Amplifiers ... .. A. Tombak Frequency and Bandwidth Agile Millimeter-Wave Filter Using Ferroelectric Capacitors and MEMS Cantilevers ...... .. .. ........ ......... ......... ........ ......... ...... C. Lugo, G. Wang, J. Papapolymerou, Z. Zhao, X. Wang, and A. T. Hunt Modeling and Applications of Ferroelectric-Thick Film Devices With Resistive Electrodes for Linearity Improvement and Tuning-Voltage Reduction ......... ......... ........ ...... P. Scheele, A. Giere, Y. Zheng, F. Goelden, and R. Jakoby Frequency Tuning and Spurious Signal Generation at Microwave Frequencies in Ferroelectric SrTiO Thin-Film Transmission Lines ..... ........ ......... ......... ........ ......... ......... ........ J. Mateu, J. C. Booth, and S. A. Schima Comparison of Techniques for Microwave Characterization of BST Thin Films ......... ......... ........ ......... ......... .. .. ........ ......... ......... ........ ......... ......... ........ ......... .... P. M. Suherman, T. J. Jackson, and M. J. Lancaster Modified Green’s Function and Spectral-Domain Approach for Analyzing Anisotropic and Multidielectric Layer Coplanar Waveguide Ferroelectric Phase Shifters ..... ......... ......... ..... W. Kim, M. F. Iskander, and C. M. Krowne Geometry-Dependent Quality Factors in Ba Sr TiO Parallel-Plate Capacitors ..... ..... N. K. Pervez and R. A. York Investigation of Ferroelectric Thick-Film Varactors for Microwave Phase Shifters ...... ......... ........ ......... ......... .. .. ........ ......... ......... ........ ......... ......... ........ ..... W. Hu, D. Zhang, M. J. Lancaster, T. W. Button, and B. Su Insertion Loss in Reflection-Type Microwave Phase Shifter Based on Ferroelectric Tunable Capacitor .... .. O. G. Vendik Ferroelectric Phase Shifters at 20 and 30 GHz ... ........ . ........ ... Z. Zhao, X. Wang, K. Choi, C. Lugo, and A. T. Hunt A 360 BST Phase Shifter With Moderate Bias Voltage at 30 GHz ...... ........ ......... ......... ........ ......... ......... .. .. ........ ......... ......... ........ .. G. Vélu, K. Blary, L. Burgnies, A. Marteau, G. Houzet, D. Lippens, and J.-C. Carru Information for Authors

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IEEE MICROWAVE THEORY AND TECHNIQUES SOCIETY The Microwave Theory and Techniques Society is an organization, within the framework of the IEEE, of members with principal professional interests in the field of microwave theory and techniques. All members of the IEEE are eligible for membership in the Society upon payment of the annual Society membership fee of $14.00, plus an annual subscription fee of $20.00 per year for electronic media only or $40.00 per year for electronic and print media. For information on joining, write to the IEEE at the address below. Member copies of Transactions/Journals are for personal use only. ADMINISTRATIVE COMMITTEE J. S. KENNEY, President L. BOGLIONI D. HARVEY S. M. EL-GHAZALY J. HAUSNER M. HARRIS K. ITOH

J. MODELSKI, President Elect L. KATEHI T. LEE B. KIM J. LIN N. KOLIAS

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N. KOLIAS, Treasurer K. VARIAN K. WU R. WEIGEL R. YORK

Distinguished Lecturers K. TOMIYASU L. YOUNG

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K. VARIAN (2006) K. C. GUPTA (2005) R. J. TREW (2004)

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North Jersey: K. DIXIT North Queensland: A. TSAKISSIRIS Northern Nevada: B. S. RAWAT Norway: S. E. WHEATLEY Orange County: H. J. DE LOS SANTOS Oregon: T. RUTTAN Orlando: P. WAHID Ottawa: Q. YE Philadelphia: J. NACHAMKIN Phoenix: C. WEITZEL Poland: M. P. MROZOWSKI Portugal: C. A. CARDOSO FERNANDES Princeton/Central Jersey: W. CURTICE/A. KATZ Queensland: A. ROBINSON Rio de Janeiro: J. R. BERGMANN Rochester: S. M. CICCARELLLI/J. VENKATARAMAN Romania: I. SIMA Russia, Nizhny-Novgorod: Y. BELOV Russia, Saint Petersburg: M. SITNIKOVA Russia, Moscow: V. KALOSHIN Russia, Saratov-Penza: N. RYSKIN Saint Louis: D. MACKE San Diego: J. TWOMEY Santa Clara Valley/San Francisco: J. J. SOWERS Seattle: K. POULSON Seoul Council: H.-Y. LEE Siberia, Novosibirsk: V. SHUBALOV Siberia, Tomsk: O. STUKACH Singapore: O. B. LEONG

Editors-In-Chief DYLAN WILLIAMS NIST Boulder, CO 80305 USA Phone: +1 303 497 3138 Fax: +1 303 497 3970 email: [email protected] AMIR MORTAZAWI Univ. of Michigan Ann Arbor, MI 48109-2122 USA Phone: +1 734 936 2597 Fax: +1 734 647 2106 email: [email protected]

South Africa: P. W. VAN DER WALT South Australia: H. HANSEN South Brazil: L. C. KRETLY Southeastern Michigan: L. M. ANNEBERG Southern Alberta: S. BOUMAIZA Spain: L. FE HARO Springfield: P. R. SIQUEIRA Sweden: A. RYDBERG Switzerland: J. HESSELBARTH Syracuse: E. ARVAS Taipei: C.-S. LU Thailand: M. KRAIRIKSH Toronto: G. V. ELEFTHERIADES Tucson: N. BURGESS/S. MORALES Turkey: O. A. CIVI Twin Cities: M. J. GAWRONSKI UK/RI: A. REZAZADEH Ukraine, Central Kiev: Y. POPLAVKO Ukraine, East: A. A. KIRILENKO Ukraine, Rep. of Georgia: R. ZARIDZE Ukraine, Vinnitsya: V. DUBOVOY Ukraine, West: I. ISAYEV Venezuela: M. PETRIZZELLI Victoria: A. MITCHELL Virginia Mountain: D. MILLER Washington DC/Northern Virginia: J. QIU Winnipeg: V. OKHMATOVSKI Yugoslavia: B. MILOVANOVIC

Associate Editors

DANIEL DE ZUTTER ZOYA POPOVIC YOSHIO NIKAWA Universiteit Gent Kokushikan Univ. Univ. of Colorado, Boulder Belgium Japan USA email: [email protected] email: [email protected] email: [email protected] KENJI ITOH JOSÉ PEDRO SANJAY RAMAN Mitsubishi Electronics Univ. of Aveiro Virginia Polytech. Inst. and State Univ. Japan Portugal USA email: [email protected] email: jcp.mtted.av.it.pt email: [email protected] JENSHAN LIN Univ. of Florida USA email: [email protected] M. GOLIO, Editor-in-Chief, IEEE Microwave Magazine G. E. PONCHAK, Editor-in-Chief, IEEE Microwave and Wireless Component Letters

RICHARD SNYDER RS Microwave Company USA email: [email protected] RUEY-BEEI WU National Taiwan Univ. Taiwan, R.O.C. email: [email protected] ALEXANDER YAKOVLEV Univ. of Mississippi USA email: [email protected] T. LEE, Web Master

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Digital Object Identifier 10.1109/TMTT.2007.892057

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Guest Editorial ECHNOLOGIES for reactive tuning and switching functions are increasingly attractive for modern wireless systems to implement reconfigurable or frequency-agile function in the RF front-end, reduce size and parts count, and lower costs. In recent years, microelectromechanical systems (MEMS) devices, piezoelectrical materials, and tunable dielectrics have emerged as candidate technologies for this purpose. Tunable dielectrics, often based on thin-film ferroelectric materials, appear to be a viable technology for analog varactor control circuits, particularly in applications requiring large RF voltage swings and limited control voltages. Such devices can be inexpensive, robust, and have intrinsically fast response times. This TRANSACTIONS’ “Special Issue on Applications of Ferroelectrics in Microwave Technology” presents a snapshot of the latest work in tunable dielectrics technology for RF/microwave applications by a number of highly respected researchers around the world. In selecting and reviewing these papers, a strong preference was given to experimental papers with RF/microwave device or circuit demonstrations or papers that addressed existing

T

Digital Object Identifier 10.1109/TMTT.2007.891149

limitations of the technology with respect to RF losses, measurements, and circuit performance. The selected papers include work on tunable filters, tunable matching networks for cellular amplifiers, phase shifters, and voltage-controlled oscillators. Papers discussing the modeling and characterization of losses and nonlinearities in these devices, a particular concern for cellular applications, are also to be found in this TRANSACTIONS’ Special Issue. It is our sincere hope that this collection will be of interest to the IEEE Microwave Theory and Techniques Society (IEEE MTT-S) community and will inspire further study and additional interest in this exciting area. Happy reading! ROBERT A. YORK, Guest Editor Department of Electrical and Computer Engineering University of California at Santa Barbara Santa Barbara, CA 93106 USA SPARTAK GEVORGIAN, Guest Editor Department of Microtechnology and Nanoscience Chalmers University of Technology Göteborg, SE-41296 Sweden

Robert A. York (S‘86–M‘91–SM‘99) received the B.S. degree in electrical engineering from the University of New Hampshire, Durham, in 1987, and the M.S. and Ph.D. degrees in electrical engineering at Cornell University, Ithaca, NY, in 1989 and 1991, respectively. He is currently a Professor of electrical and computer engineering with the University of California at Santa Barbara (UCSB), where his group is involved with the design and fabrication of novel microwave and millimeter-wave circuits, high-power microwave and millimeter-wave amplifiers using spatial combining and wide-bandgap semiconductor devices, and application of ferroelectric materials to microwave and millimeter-wave circuits and systems. Dr. York was elected to serve on the IEEE Microwave Theory and Techniques Society (IEEE MTT-S) Administrative Committee (AdCom) in 2006. He was the recipient of the 2004 IEEE MTT-S Outstanding Young Engineer Award, and a 1996 Office of Naval Research Young Investigator Award.

Spartak Gevorgian (M’96–SM’97) received the M.S. degree in radioelectronics from the Yerevan Polytechnic Institute, Yerevan, Armenia, in 1972, and the Ph.D. and Dr.Sci. degrees from the Electrotechnical University, St. Petersburg, Russia, in 1977 and 1991, respectively. From 1972 to 1993, he held different research and teaching positions with the Yerevan Polytechnic Institute and the Electrotechnical University. From 1993 to 1998, he held research positions with Chalmers University of Technology, Göteborg, Sweden. Since 1998, he has been a Professor with Chalmers University of Technology. Since 1996, he has also worked part time with Ericsson Microwave Systems AB (currently Ericsson AB), Mölndal, Sweden. He has authored or coauthored over 270 papers and conference presentations. He holds over 30 patents/patent applications. His research interests are physics, design, and experimental investigation of microwave devices and components based on ferroelectrics, silicon RF integrated circuits (RFICs) and monolithic microwave integrated circuits (MMICs), microwave photonic devices (optically controlled components), and modeling of passive coplanar components based on conformal mapping. 0018-9480/$25.00 © 2007 IEEE

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Improving Linearity of Ferroelectric-Based Microwave Tunable Circuits Jia-Shiang Fu, Student Member, IEEE, Xinen Alfred Zhu, Student Member, IEEE, Jamie D. Phillips, Member, IEEE, and Amir Mortazawi, Fellow, IEEE

Abstract—A device architecture for improving the linearity of barium–strontium–titanate (BST) capacitors is proposed, analyzed, and experimentally demonstrated. The basic concept is to reduce the RF swing across the ferroelectric varactors by connecting multiple capacitors in series. With proper biasing networks, the overall device preserves its tunability. In this paper, the proposed architecture is analyzed to derive the equations that relate its quality factor and tuning speed to the design parameters. Parallel-plate BST capacitors are fabricated based on the architecture. Various measurements are performed to demonstrate the efficacy of the technique. The third-order intercept point at input (IIP3 ) is found to be improved by 16 dB. The hot-tuning test shows that the tunability is maintained for up to 20-V peak-to-peak swing. BST capacitors utilizing the proposed technique are used in a phase-shifter design. The 2.4-GHz phase shifter exhibits high IIP3 greater than 40 dBm. Index Terms—Barium–strontium–titanate (BST), ferroelectric capacitor, linearity, phase shifter.

I. INTRODUCTION

V

ARIABLE capacitors (varactors) are essential components in tunable microwave circuits. Ferroelectric materials, which allow for its permittivity being adjusted by the electric field applied, are legitimate candidates for varactors. Among them, barium–strontium–titanate (BST) has been used in various microwave circuits, such as filters, phase shifters, and matching networks [1]–[3], to provide the benefit of tunability. Compared to microelectromechanical systems (MEMS) varactors, thin-film BST capacitors exhibit a higher tuning ratio under relatively low biasing voltages. However, due to the inherent nonlinear nature of the ferroelectric materials, linearity has been a concern, especially for high-power applications and spectrally packed communication systems. Linearity improvement can be achieved through the proper design of the device architecture rather than changing the base material. While previous researchers have proven the effectiveness of architectural approaches for gap capacitors [4], in this study, a simple method to reduce the nonlinearity of parallelplate capacitors is suggested. The same technique has also been described in [5]–[7]. This paper presents an analysis of the tech-

Manuscript received June 1, 2006; revised October 4, 2006. This work was supported in part by the National Science Foundation under Grant 0300421 and Grant ECS-0238108. The authors are with the Electrical Engineering and Computer Science Department, The University of Michigan at Ann Arbor, Ann Arbor, MI 48109 USA (e-mail: [email protected]; [email protected]; [email protected]; [email protected]). Digital Object Identifier 10.1109/TMTT.2006.889323

Fig. 1. Basic concept underlying the proposed technique illustrated by a fivestacked capacitor example (from [6]).

nique for the study of tradeoffs and optimization of BST varactor-based circuits for wireless communication applications. The basic concept underlying the linearity-improving technique is illustrated in Fig. 1. As the RF swing across the varactor increases, the capacitance of the varactor starts to deviate from the value set by the dc-bias voltage, resulting in signal distortion. Connecting a number of identical capacitors in series immediately reduces the RF swing across each capacitor, which, in turns, lessens the variation of the total capacitance, thus, improving the linearity. Using an -stacked capacitor reduces the of the original swing. voltage swing across each element to To maintain the same capacitance as that of a single capacitor, each element should be times as large. As a result, the area of times as large as that of a an -stacked capacitor would be single capacitor having the same capacitance value. Though the quadratic increase of the occupied area seems disadvantageous, the size of multiple-stacked capacitors is still compact because of the high permittivity of the BST thin film. In this study, a device architecture for improving the linearity of parallel-plate BST capacitors based on the aforementioned concept is proposed and experimentally demonstrated. The details of the architecture and the circuit analysis are presented in Section II. Following that, the design considerations are summarized in Section III. Next, in Section IV, the fabrication process is described and measurement results are shown and discussed. Finally, conclusions are provided in Section V. II. DEVICE ARCHITECTURE AND ANALYSIS The device architecture is illustrated in Fig. 2(a) with a fivestacked example. Five capacitors are connected in series through top and bottom electrodes alternatively. Thin-film resistors link together top or bottom electrodes, respectively, to ensure that the bias voltage falls on each capacitor equally. Fig. 2(b) shows the schematic of the device architecture. Adding biasing resistors preserves the tunability, but inevitably degrades the tuning speed and the quality factor. Large

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Fig. 2. Five-stacked example for the proposed device architecture. (a) Simplified layout. (b) Schematic illustration (from [6]). (c) Rearranged schematic for analysis.

resistance increases the time constant, thus, reducing the tuning speed, while small resistance results in higher loss, lowering the quality factor. To analyze the overall performance of the proposed architecture and quantitatively determine the effects of the resistance, the schematic is then rearranged, as shown in Fig. 2(c). As can be seen in Fig. 2(c), a composite capacitor with five elements can be viewed as a single capacitor at the center constages in cascade. In general, this necting to two four-port viewpoint applies to any composite capacitor that the number of is odd, i.e., elements

since the correBy applying the boundary condition that sponding port is open, one can solve (4) and obtain the relations between the voltage and current variables. Among them, we define

(1)

where is the voltage transfer ratio of the voltage across a single capacitive unit to that across the entire composite capacis the overall admittance of the composite capacitor itor and with capacitive units. For and , and are solved and are listed in the Appendix. The results have also been verified by simulations. It will be shown that the expressions for the tuning speed and and , respectively. the quality factor can be derived from Before going through the derivation, the capacitance value of the capacitive element in an -stacked capacitor can first be determined. Since there are capacitors in series, for the overall capacitance being the same as that of the single capacitor, the capacitance of each capacitor must be times as large, i.e.,

In (1), can be any positive integer. In this sense, an -stacked capacitor with an odd number of elements can then be viewed stages. Since the as a single capacitor with cascaded stages are in cascade, it is helpful to describe each of them with a four-port matrix that is similar to the ABCD matrix for a twoport network as in

(2)

and . On the other hand, where, in this case, the center capacitor is considered as a two-port network, where the voltages across and currents flowing through have the following relation: (3) The direction of currents is flowing into the positive terminals. Using (1)–(3), the -stacked capacitor can then be described by

(5) and (6)

(7) where is the capacitance of the single capacitor before scaling. To derive the expression for the tuning speed, the asymptotic at the lower end of the frequency spectrum response of for

(8)

is examined. Note that (8) is the same as the response of a series circuit, which has a time constant

(4)

(9)

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Fig. 4. Proper scaling for preserving quality factor. (a) Original single capacitor. (b) Stacked capacitor. (c) Equivalent array perspective (from [6]). Fig. 3. Tradeoff between tuning speed and quality factor illustrated by potting time constant (9) and degradation of quality factor (12) versus design parameters. R is the biasing resistance, while C is the capacitance of the single capacitor before scaling. In this example, Q = 50 and the frequency is 1.3 GHz.

Next, to obtain the expression of the quality factor, at high frequency, the overall admittance of the composite capacitor can be simplified to for (10) and

In this case,

(11) where is the effective parallel conductance and is the quality factor of the capacitive unit at frequency of interest . is a function of the dielectric loss of the ferroelectric thin film and the conductor loss of the electrodes. From (10), along with (7) and (11), one can show that the quality factor of the composite capacitor with finite biasing resistance is

(12) Equation (9) and (12) quantitatively show the tradeoff between the tuning speed and the quality factor. This tradeoff is illustrated in Fig. 3, where the quality factor degradation and for composite capacithe time constant are plotted versus tors with various ’s. In this example, the quality factor of the single capacitor is 50 and the frequency is 1.3 GHz. III. DESIGN CONSIDERATIONS A. Quality Factor The quality factor can be maintained through proper scaling. For an -stacked capacitor, each capacitive element is times as large as the single capacitor. Scaling the capacitive element by times along the direction perpendicular to the current flow,

as shown in Fig. 4(b), would reduce the loss per element. This reduction would cancel the increase of loss introduced by the series connection of multiple elements. Fig. 4(c) presents an alternative perspective of viewing the proposed architecture as an -by- array, in which each unit is simply the single capacitor before scaling. The array perspective is helpful in explaining the preservation of the quality factor. Since neither series, nor parallel combination of identical capacitors affects the quality factor, an array, which is nothing but the series combination of parallel combinations (or vice versa), would have the same quality factor as that of each building block. Though properly scaling the physical dimensions of the capacitive units could preserve the quality factor for the -stacked capacitor, the biasing resistors would inevitably bring in a certain amount of loss, lowering the quality factor of the overall composite capacitor. In general, as becomes larger, the signal would pass mostly through the capacitors rather than the resistors, which lessens the deleterious effect of the biasing networks on the quality factor. B. Tuning Speed As the number of elements increases and the capacitance of each element is scaled up, the tuning speed would drop if the resistance of the biasing resistors is not scaled down accordingly. As derived from the previous analysis, to maintain the tuning speed, the resistance may be scaled according to (9). Although the tuning speed can be maintained through reducing when is increased, the RF performance, such as quality factor, would be degraded if becomes excessively small. C. Number of Elements The number of elements is a compromise among linearity, tuning speed, quality factor, and device area. The linearity is improved as increases because of the reduced RF swing across each element, while it is beneficial to have lower for applications requiring fast tuning. The device area expands quadratically with , which seems to be a disadvantage. However, on some occasions, it is necessary to connect capacitors in series in order to obtain small capacitance on the high-permittivity BST.

FU et al.: IMPROVING LINEARITY OF FERROELECTRIC-BASED MICROWAVE TUNABLE CIRCUITS

Fig. 5. Five-stacked capacitor. Device area: 350

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2 160 m .

IV. FABRICATION AND MEASUREMENT A. Fabrication Process The BST capacitors were fabricated on 430- m-thick sapphire. The substrate has a dielectric constant of 9.39 as the electric field is perpendicular to its -axis. The bottom electrodes composed of Ti/Au/Pt (20/2000/1000 Å) were formed first. BST thin film was then deposited using a pulse laser deposition (PLD) technique with a 50/50 BST target. The measured film thickness was 400 nm. The top electrodes composed of Pt/Au (1000/2000 Å) were then formed. After that, the BST thin film was etched in the regions other than the active area. Thin-film SiCr lines with a thickness of 100 nm were then sputtered. Following that, the metal contacts composed of Cr/Au (200/500 Å) were formed to connect the biasing resistors to the electrodes and serve as the bottom electrodes of the metal–insulator–metal (MIM) capacitors in other regions. Silicon–nitride Si N thin film, serving as the dielectric layer in the MIM capacitors and also the passivation for the BST capacitors, was deposited using a plasma enhanced chemical vapor deposition (PECVD) technique. The film thickness was measured to be 1300 Å. Finally, the other metal traces were electroplated with gold to around 8 m. A photograph of a five-stacked capacitor is shown in Fig. 5. The device area of this capacitor is 350 160 m , including the biasing resistors and the passivation layer. The capacitance of it is around 1.4 pF at zero bias. B. Capacitance and Quality Factor Small-signal scattering parameters of the capacitors were measured. The scattering parameters were converted to ABCD-parameters for the ease of post-processing. The capaci, while the total quality factor tance was obtained from . was calculated directly by For and , the capacitances and the quality factors at various bias points are plotted in Fig. 6. This set of the capacitors reaches a 2 : 1 tuning ratio at around 10 V. As can be seen, the capacitances for different ’s are almost the ; its capacitance is less than that of same, except for the multiple-stacked capacitors by approximately 7%. This difference could be resulted from the additional capacitance arise from the gap between electrodes, which is present in only multiple-stacked capacitors. The quality factors of the stacked capacitors are less than that of the single capacitor. This is a result

Fig. 6. Measured capacitances and quality factors (at 1.3 GHz) of the BST capacitor with different numbers of element under various bias voltages.

Fig. 7. Measured (solid line) and estimated (dashed line) quality factors for capacitors with N = 3; 5; 7; and 9 at various bias voltages.

of the loss introduced by the biasing resistors, as explained in Section II and estimated by (12). The measured resistance of the biasing resistor is 12 k in all capacitors. Since the resistance is not scaled down, but remains constant for the stacked capacitors measured, the effect of biasing resistors decreases as the capacitance of the capacitive unit increases along with the number of elements. Fig. 7 shows the comparison between the measured quality factors and those predicted by (12) based on the measured quality factor of the single capacitor at various bias voltage. The observed quality factors deviate from the estimated ones by less than 12%. C. Tuning Speed The tuning speed of the -stacked capacitors was measured with a 1-kHz square-wave input with low and high voltages being 0 and 3 V, respectively. The output voltage across the load was monitored using an oscilloscope. The time constant was then calculated from the monitored waveform. Fig. 8 shows the comparison between the measured time constants and the estimated values by (9). The capacitance at 3-V bias is used in the calculation. As can be seen, the difference between the measurement and estimation increases along with , which

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Fig. 8. Measured (solid line) and estimated (dashed line) time constants for capacitors with N = 3; 5; 7; and 9.

Fig. 9. Simulated (dashed line) and measured (solid line) IIP for N = 1; 3; and 5 at various bias voltages.

Fig. 10. Hot tuning measurement. (a) Setup. (b) Capacitance normalized to zero-bias capacitance for N = 1; 3; and 5 under different input power at various bias voltages.

may be due to the parasitic elements, such as the effective resistance resulted from the leaking current in the BST thin film, which are not included in the above presented simple model.

respectively, for nonlinear simulations. The - curve of the BST capacitors is modeled using the expression derived in [8].

D. Intermodulation Distortion

E. Hot Tuning

As a common indicator for linearity, the third-order intercept was measured. The measurement setup was point at input mostly the same as that in [6], except that power amplifiers were added in between the signal generators and the isolators so that the input power level was large enough for the BST capacitors to produce high enough intermodulation products above the noise floor. Two-tone signals with center frequency and frequency spacing of 1.3 GHz and 1 MHz were used. The frequency of 1.3 GHz was chosen because it was the center frequency of the for isolators used in the measurement setup. The measured capacitors with various numbers of elements is shown in Fig. 9. improvement up to 16 dB was obtained. At zero bias, of the five-stacked case at 3-V bias was not measured because the intermodulation products were too small to be observed using the measurement setup. is also shown in Fig. 9 for comparison. The simulated While the results from the analysis in Section IV-C are useful for linear circuits, the time-domain counterparts of (2) and (3) are stage and the center capacitor, used to describe the cascaded

One manifestation of varactor nonlinearity is its variation of capacitance with the RF amplitude across it. Tunability degradation of BST capacitors under large-signal excitation has been presented at low frequency in [9]. In this study, hot tuning measurements were performed at 1.3 GHz on the BST capacitors and . with As shown in Fig. 10(a), the setup for the hot tuning included a power amplifier to increase the input power level, an isolator to present a good matching condition to the amplifier, and a power meter to measure the output power with high resolution. The method for mapping the capacitance was similar to that described in [10]. While in [10], the return loss was measured for capacitance mapping, here, the insertion loss was instead measured. Fig. 10(b) shows the normalized capacitances of the BST caand , biased at different voltages, under pacitors with various input power level. The input power level is swept from 0 to 28 dBm, which correspond to 0.8- and 20-V peak-to-peak swing across a 1.4-pF capacitor at 1.3 GHz, respectively, from circuit simulation. As can be seen, at zero bias, the capacitance

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V. CONCLUSION A technique has been proposed to improve the linearity of thin-film parallel-plate BST capacitors. In this study, it has been analyzed to provide design equations relating to the quality factor and the tuning speed. The linearity improvement provided by this approach has been demonstrated by and hot tuning measurements. A phase shifter employing BST capacitors using the proposed technique has been designed and fabricated. The good linearity presented by the phase shifter demonstrates the usefulness of this technique for tunable microwave circuits in practice. APPENDIX , one can deApplying the boundary condition that and rive the normalized voltage across the center element the overall admittance of the composite capacitor , as defined in (5) and (6), respectively, by solving (4). The closed-form formula can be obtained using mathematical software; however, they are complicated and cumbersome. Moreover, a large number of elements is not practical in real circuits. Therefore, and are listed below. ’s only the solutions for are

Fig. 11. Phase shifter. (a) Photograph. (b) Small-signal response at 2.4 GHz.

of the single capacitor drops by 20% as input power level inand , creases, implying a strong nonlinearity, while, for the capacitance hardly varies with the RF swing, demonstrating the linearity improvement. Also, one can see that the tunability of the stacked capacitors is much less degraded than that of the single capacitor.

and

’s are

F. Phase Shifter With Linearized BST Capacitors A phase shifter was designed based on an all-pass network [11], [12], where BST capacitors utilizing the proposed archiwere employed. The circuit was laid out in tecture with a flipped sense to reduce the area and facilitate cascading with other circuits. The photograph of the fabricated circuit is shown in Fig. 11(a). Due to fabrication errors, the inductances of the spiral inductors were deviated from the expected value. After trimming with bond wires, the phase shifter was measured. Its small-signal response at 2.4 GHz is shown in Fig. 11(b). The insertion loss was less than 2 dB and the input and output return losses were greater than 10 dB. The relative phase shift over 0 to 25-V bias voltages was approximately 90 . The center frequency and frequency spacing of the two-tone signals for measuring the intermodulation distortion of the phase shifter were 2.4 GHz and 1 MHz, respectively. The observed was greater than 40 dBm.

where

. ACKNOWLEDGMENT

The authors would like to thank K. van Caekenberghe, The University of Michigan at Ann Arbor, for his help in fabricating the SiCr thin-film resistors. REFERENCES [1] A. Tombak, J.-P. Maria, F. Ayguavives, Z. Jin, G. T. Stauf, A. I. Kingon, and A. Mortazawi, “Voltage-controlled RF filters employing thin-film barium strontium titanate tunable capacitors,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 2, pp. 462–467, Feb. 2003.

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[2] B. Acikel, T. R. Taylor, P. J. Hansen, J. S. Speck, and R. A. York, “A TiO thin films,” new high performance phase shifter using Baxsr IEEE Microw. Wireless Compon. Lett., vol. 12, no. 7, pp. 237–239, Jul. 2002. [3] L.-Y. V. Chen, R. Frose, D. Chase, and R. A. York, “Analog tunable matching network using integrated thin-film BST capacitors,” in IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2004, vol. 1, pp. 261–264. [4] Y.-K. Yoon, D. Kim, M. G. Allen, and J. S. Kenney, “A reduced intermodulation distortion tunable ferroelectric capacitor: Architecture and demonstration,” in IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2003, vol. 3, pp. 1989–1992. [5] R. A. York, “Circuit configuration for DC-biased capacitors,” U.S. Patent 6 674 321, Jan. 6, 2004. [6] J.-S. Fu, X. A. Zhu, D.-Y. Chen, J. D. Phillips, and A. Mortazawi, “A linearity improvement technique for thin-film barium strontium titanate capacitors,” in IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2006, pp. 560–563. [7] H. Katta, H. Kurioka, and Y. Yashima, “Tunable power amplifier using thin-film BST capacitors,” in IEEE MTT-S Int. Microwave Symp. Dig., Jun. 2006, pp. 564–567. [8] D. R. Chase, L.-Y. Chen, and R. A. York, “Modeling the capacitive nonlinearity in thin-film BST varactors,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 10, pp. 3215–3220, Oct. 2005. [9] A. Tombak, J.-P. Maria, F. T. Ayguavives, Z. Jin, G. T. Stauf, A. I. Kingon, and A. Mortazawi, “Tunable barium strontium titanate thin film capacitors for RF and microwave applications,” IEEE Microw. Wireless Compon. Lett., vol. 12, pp. 3–5, Jan. 2002. [10] Y. Lu, L. P. B. Katehi, and D. Peroulis, “High-power MEMS varactors and impedance tuners for millimeter-wave application,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 11, pp. 3672–3678, Nov. 2005. [11] D. Adler and R. Popovich, “Broadband switched-bit phase shifter using all-pass networks,” in IEEE MTT-S Int. Microw. Symp. Dig., Jun. 1991, vol. 1, pp. 265–268. [12] D. Kim, Y. Choi, M. Ahn, M. G. Allen, J. S. Kenney, and P. Marry, “2.4 GHz continuously variable ferroelectric phase shifters using all-pass networks,” IEEE Microw. Wireless Compon. Lett., vol. 13, no. 10, pp. 434–436, Oct. 2003.

Jia-Shiang Fu (S’06) was born in Taipei, Taiwan, R.O.C., in 1981. He received the B.S. degree in electrical engineering from National Taiwan University, Taipei, Taiwan, R.O.C., in 2003, the M.S. degree in electrical engineering from The University of Michigan at Ann Arbor, in 2005, and is currently working toward the Ph.D. degree at The University of Michigan at Ann Arbor. His research interests include frequency-agile microwave circuits and linearization of power amplifiers.

Xinen Alfred Zhu (S’05) received the B.Eng. (Hons.) degree in electronic and communication engineering from City University of Hong Kong, Hong Kong, in 2003, the M.S. degree from The University of Michigan at Ann Arbor, in

2005, and is currently working toward the Ph.D. degree at The University of Michigan at Ann Arbor. His research interests include ferroelectric thin-film fabrication using pulsed laser deposition, ferroelectric thin-film-based varactors, and tunable microwave circuits.

Jamie D. Phillips (M’01) received the B.S., M.S., and Ph.D. degrees in electrical engineering from The University of Michigan at Ann Arbor, in 1994, 1996, and 1998, respectively. In his doctoral studies, he made key contributions to the epitaxial growth and device applications of self-assembled InGaAs/GaAs quantum dots, including quantum dot infrared photodetectors and quantum dot diode lasers. From 1998 to 1999, he was a Post-Doctoral Researcher with Sandia National Laboratories, where he contributed to the growth of III–V antimonide materials for mid-infrared lasers. From 1999 to 2001, he was a Research Scientist with the Rockwell Science Center, where he conducted research on HgCdTe infrared detectors. Since 2002, he has been an Assistant Professor with the Electrical Engineering and Computer Science Department, The University of Michigan at Ann Arbor. He was a Guest Editor for the Journal of Electronic Materials “Special Issue on Wide Bandgap Materials.” His technical interests and contributions are in the growth, characterization, and device applications of compound semiconductor and oxide-based materials for opto-electronics and electronics. He has authored or coauthored over 60 peer-reviewed papers on these subjects. Prof. Phillips is a member of the American Society of Engineering Education (ASEE) and the American Vacuum Society (AVS). He was the recipient of the 2003 National Science Foundation (NSF) CAREER Award.

Amir Mortazawi (M’90–SM’05–F’05) received the Ph.D. degree in electrical engineering from The University of Texas at Austin, in 1990. He is a currently a Professor of electrical engineering with The University of Michigan at Ann Arbor. His research interests include millimeter-wave phased arrays, power amplifiers, power-combining techniques, and frequency-agile microwave circuits. Prof. Mortazawi is the Editor-in-Chief of the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES. He is a member of the IEEE Microwave Theory and Techniques Society (IEEE MTT-S) Administrative Committee (AdCom). He is the co-chair of the IEEE MTT-16 Committee on Phased Arrays. He was an associate editor for the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION (1998–2001), an associate editor for the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES (2005), and was guest editor for the December 1995 Special Microwave Symposium issue of the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES. He was a secretary to the IEEE MTT-S AdCom.

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A Low-Noise -Band VCO Based on Room-Temperature Ferroelectric Varactors Martin Norling, Andrei Vorobiev, Harald Jacobsson, Member, IEEE, and Spartak Gevorgian, Senior Member, IEEE

Abstract—This paper reports a -band voltage-controlled oscillator based on room-temperature ferroelectric varactors. The circuit is realized as a hybrid module where flip-chip transistors are mounted on a silicon substrate with integrated ferroelectric varactors and passive circuitry. The size of the module is 4.7 2.2 mm2 . The measured center frequency is 16.5 GHz with a linear tunability of 6.7% and an output power of 3 dBm 1 dB over the tuning range. The measured phase noise at center frequency is 95 dBc/Hz at 100-kHz offset. Another version of the oscillator is measured operating at 19.6 GHz with a tunability of 3.3% and a phase noise of 102 dBc/Hz at 100-kHz offset. Index Terms—Ferroelectric capacitors, heterojunction bipolar transistors (HBTs), voltage-controlled oscillators (VCOs).

I. INTRODUCTION HE NOISE performance of a microwave oscillator significantly depends on the quality factor ( factor) of the involved resonator [1]. Voltage-controlled oscillators (VCOs) make use of tunable resonators, typically consisting of lumpedelement inductors and semiconductor varactors. At relatively low frequencies, i.e., below 10 GHz, semiconductor varactors demonstrate low losses, thus the resonator factor is limited by losses associated with integrated inductor coils. However, losses of semiconductor varactors usually increase with frequency, and at frequencies above 10 GHz, they eventually limit the total factor of the tunable resonator, effectively degrading the oscillator noise performance. In pursuit of low-noise VCOs, several technologies for lowloss high-frequency varactors are considered. Microelectromechanical (MEM) varactors offer high factors [2], but they are too slow for some applications. In addition, MEM components are often associated with reliability problems or need expensive hermetic packages. In the past, ferroelectric varactors in combination with high-temperature superconductor resonators have been considered for low-noise VCOs [3]. However, these VCOs

T

Manuscript received May 30, 2006; revised October 3, 2006. This work was supported under the Swedish SSF Project ICTEA and by the European Union under the Nanostar Project. M. Norling and A. Vorobiev are with the Department of Microtechnology and Nanoscience, Chalmers University of Technology, SE-41296 Göteborg, Sweden (e-mail: [email protected]). H. Jacobsson is with the Microwave and High Speed Electronics Research Center, Ericsson Research, Ericsson AB, SE-43184 Mölndal, Sweden. S. Gevorgian is with the Department of Microtechnology and Nanoscience, Chalmers University of Technology, SE-41296 Göteborg, Sweden, and also with the Microwave and High Speed Electronics Research Center, Ericsson Research, Ericsson AB, SE-43184 Mölndal, Sweden. Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TMTT.2006.889332

require cryogenic cooling, which disqualifies the technology for many applications. factor of room-temperature Recent improvements in the ferroelectric varactors [4] have made such varactors promising for applications operating at frequencies above 10 GHz. Besides having factors higher than semiconductor analogs, the leakage current of thin-film ferroelectric varactors is several orders of magnitude smaller, and the tuning speed of these varactors is in the nanosecond range [5]. Naturally, VCOs based on room-temperature ferroelectric varactors are interesting; such oscillators operating at megahertz frequencies have been demonstrated [6]. A GaN high electron-mobility transistor (HEMT) oscillator based on room-temperature ferroelectric varactors, operating at 5 GHz, was also recently published [7]. In this study, room-temperature Ba Sr TiO (BSTO) ferroelectric varactors integrated with silicon substrates are used to design a -band VCO. It is implemented as a balanced Colpitts oscillator and based on SiGe flip-chip transistors from Infineon Technologies AG, Munich, Germany.1 The aim is to demonstrate the potential of ferroelectric varactors in high tunability low phase-noise VCOs with highly linear frequency tuning characteristics. This paper is organized as follows. Sections II and III describe the main components used in the VCO. The circuit topology, layout, and a short description of the simulation are given in Section IV. Measurements and a brief analysis of the results are given in Sections V and VI. Finally, Section VII presents conclusions. II. SUBSTRATE TEMPLATE AND VARACTORS A. Template Fig. 1 depicts the template used in this study. The associated parameters are summarized in Table I. Essentially, the template consists of two patterned metal layers separated by a ferroelectric BSTO film. Parallel-plate varactors are readily formed using these layers. The layers are grown on high-resistivity silk cm. icon (HR-Si) with a bulk resistivity specified to Further, the backside of the substrate is metallized, enabling microstrip design. The use of HR-Si as the substrate ensures low microwave losses and avoids problems associated with the thermal expansion coefficient when using flip-chip mounted Si-based transistors. B. Fabrication Starting with HR-Si substrates, all processing is carried out in-house. First, a metal layer stack (M1) consisting of Ti (adhe1Infineon

Technologies AG. [Online]. Available: http://www.infineon.com

0018-9480/$25.00 © 2007 IEEE

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Fig. 1. Cross section of varactor and substrate.

TABLE I TEMPLATE LAYERS

sion layer), Au, and Pt is deposited using e-beam evaporation and patterned by ion milling. The Pt supports the growth of the ferroelectric BSTO film with the desired crystal orientation. The BSTO film is subsequently deposited by laser ablation. In addition to the high resistivity of the BSTO film, the high work function of Pt ensures extremely low leakage currents. Next, a layer of SiO is introduced in order to prevent dc shorts in large-area decoupling capacitances (via pin holes in the BSTO film). This layer is removed where the small-sized varactors are formed in order to maximize the tunability of these components. Lastly, the second metal layer stack (M2) is deposited and patterned by liftoff. It consists of Ti (improving adhesion on SiO ) and Au. Noteworthy is that all the passive components of the VCO, i.e., transmission lines, balun, decoupling capacitors, and varactors, are fabricated in a single processing sequence. C. Varactors As mentioned above, parallel-plate varactors are formed between metal layers M1 and M2. The parallel-plate configuration was chosen because of its high tunability and high factor. It is also operable with low tuning voltages. Low-frequency measurement of test varactors, sharing the substrate with the VCO of this study, is shown in Fig. 2(a). Similar varactors have also been characterized at microwave frequencies [see Fig. 2(c)]. The tuning characteristics of the ferroelectric varactors are very attractive for wide tuning-range VCOs. As seen in Fig. 2(a), the tunability is large and the factor is high over the full tuning depenrange. Furthermore, the quadratic slope of the dence is desirable for oscillators having an oscillation frequency . Fig. 2(b) illustrates the high dependence degree of linearity in the dependence. This transdependence—important for many lates into a highly linear oscillator applications. For convenience, a capacitive tunability parameter is defined as (1)

Fig. 2. Parallel-plate varactor characteristics. (a) Measured bias dependence of C and Q factor at 1 MHz. (b) Derived 1= C (V ) dependence. (c) Measured frequency dependence of C and Q factor at 0- and 20-V bias.

Over the full tuning range, this tunability is calculated to . The microwave measurements in Fig. 2(c) indicate a frequency-independent capacitance with a constant tunability over a wide frequency range. The factor is also high, e.g., at 16 GHz. In comparison, a typical semiconductor varactor has at the same frequency [4]. While ferroelectric varactors have sufficiently low losses in order not to limit the resonator factor, semiconductor varactors commonly degrade factor lower than the factor of the the resonator with a inductive part. In summary, the ferroelectric varactor has a high factor in addition to a high tunability—desirable features for resonators used in, for instance, high-performance VCOs.

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TABLE II MODEL PARAMETERS

Fig. 3. Photograph of transistor chip (without solder bumps).

Fig. 5. Schematic of a balanced Colpitts oscillator.

Fig. 4. Parasitics model of flip-chipped transistor.

III. TRANSISTOR CHIP

placed onto the silicon substrate, containing prefabricated varactors and other passives, with a vacuum tool. The substrate is then heated up to 300 C. After 10 s at peak temperature, the substrate is rapidly cooled to bond the transistor chips to the contact pads.

A. Transistor Modeling The transistors used in this study are SiGe HBTs from Infineon Technologies AG, Munich, Germany. A photograph of the 0.4 0.4 mm transistor chip is shown in Fig. 3. The transistor core is identical with the Infineon Technologies AG BFP640 packaged transistor. The transistor core is described by a large-signal Gummel–Poon model supplied by Infineon Technologies AG. In designing the VCO, considerable effort has been spent on developing the circuit model of the transistor chip. The parasitic elements associated with on-chip interconnects, bumps, and substrate proximity effects are modeled as equivalent inductors and capacitors, as shown in Fig. 4. The model parameters (see Table II) are extracted from measurements of the device mounted in a specially designed two-port test-fixture fabricated on the template shown in Fig. 1. The details of the modeling are reported in [8]. The model shows good agreement with measurements for frequencies below 10 GHz. However, above 10 GHz, the model is poor and constitutes the main source for simulation inaccuracies. B. Flip-Chip Procedure By using a PP-5 semiautomatic die bonder from JFP Microtechnic, Noeux Les Mines, France, the transistor chips are

IV. CIRCUIT DESIGN A. Topology The topology of a balanced Colpitts common-base oscillator was chosen for this design (see Fig. 5). The Colpitts topology is known to provide low-noise characteristics according to Aparicio and Hajimiri [9]. The balanced design may be considered as two single-ended oscillators operating in antiphase; hence, forming a virtual ground along the symmetry line. This enables efficient isolation of the bias network; no differential-mode currents travel around or off the chip, which otherwise cause voltage fluctuations. This further improves the noise characteristics of the oscillator. The differential output is also naturally advantageous for all-balanced low-noise systems. Basically, the resonator tank, marked in Fig. 5, comprises an , which consists inductance in parallel with capacitance of the series-connected capacitances and [1], i.e., (2) The resonant frequency is given as (3)

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Fig. 6. (a) VCO microstrip layout. (b) Schematic of bias network.

The resistance resonator tank.

represents the output load and losses in the

B. Layout and Implementation Fig. 6(a) shows the layout of passive circuits of the oscillator including ferroelectric varactors implemented in the template described in Section II. To ensure low losses, the passive components shown in Fig. 5 are implemented as distributed elements. For single-ended measurements, the balanced outputs of the VCO are combined using a balun. In addition to its odd-mode operation, the balun also presents a large even-mode impedance, violating even-mode oscillation conditions. Otherwise, parasitic even-mode oscillations may appear because of the bias network, which is connected along the symmetry line of the circuit. Together with the distributed balun, RF-grounded stubs #1 in Fig. 6(a) provide inductive loading on the collector node (cf. in Fig. 5). The factor of the stubs is simulated to be approximately 25 at 16 in Fig. 5) provide dc paths from GHz. Further, stubs #2 (cf. one of the two emitter terminals of the transistors. Since the varactors are connected to the opposite emitter terminals, the length in order not to load the varactor caof stubs #2 is tuned to pacitances.

in Fig. 5 corresponds to the colThe capacitance lector–emitter capacitance of the transistor, which provides enough coupling for oscillation at the desired frequency. Furthermore, in order to isolate the dc-bias network, a large continuous metal plane is formed in the bottom metal (M1), as marked in Fig. 6(a). The equivalent circuit of the bias network is shown in Fig. 6(b). The large decoupling capacitances are formed between M1 and M2 sandwiching the same ferroelectric film as used in the varactors. These large capacitances effectively RF ground the end of stubs #1 and #2 and provide isolation of the bias network. Fig. 7 shows a photograph of a fabricated oscillator module in a measurement test fixture. The size of the module is 4.7 2.2 mm . The test fixture contains nine oscillator modules (see Fig. 8). Fig. 9 shows a micrograph of the varactor used in the VCO. For the 16-GHz VCO, the overlapping area is approximately 45 m . C. Simulation The circuit was simulated in Agilent’s Advanced Design System (ADS). Since no design kit was available, efforts had to be focused on modeling of the various components used in the circuit. Besides the modeling of the transistor, described in Section III-A, the passive components had to be characterized.

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Fig. 7. Photograph of a VCO sample with mounted transistors, placed in test fixture.

Fig. 8. Nine VCO chips mounted in a brass test fixture using silver glue.

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Fig. 10. EM simulation of the distributed balun as a three-port.

expects from a fully balanced circuit. For instance, commonmode noise suppression is reduced, and the signal power of the odd-mode oscillation is decreased. The varactor geometry, as shown in Fig. 9, was similarly modeled in ADS Momentum. The permittivity of the BSTO film was extracted from previously measured test varactors for a number of bias voltages. The extracted permittivity was subsequently used in EM simulations of the varactor geometry, which was characterized as a two-port over a wide frequency range. The simulation captured parasitic effects such as electrode inductances and fringing field capacitances. The resulting varactor model was then used in the simulations of the VCO. Once the models of the various components were established, a small-signal simulation of the circuit was used to analyze the oscillation startup conditions. The circuit was adjusted to oscillate only at the desired frequency. Next, a large-signal harmonic-balance simulation investigated the steady-state oscillation, providing predictions of amplitude, frequency, and phase noise. The circuit was optimized for low phase noise before the layout was generated. Selected parts of the layout was then EM simulated, and the entire VCO was re-simulated. This design procedure was iterated a number of times before reaching the desired performance. V. MEASUREMENT RESULTS

Fig. 9. Photograph of varactors.

Typically, circuit models were deployed for the initial design iterations before gradually verifying the components in electromagnetic (EM) simulations using Agilent’s Momentum. Fig. 10 shows EM simulations of the distributed balun as a three-port, where port 1 corresponds to the unbalanced port. The microstrip design of the balun proved to be hard to optimize towards amplitude and phase balance since the substrate thickness was relatively large compared to the wavelength. Nevertheless, the simulations in Fig. 10 indicate a fair bandwidth of the balun, despite the amplitude and phase imbalance of approximately 1 dB and 10 , respectively. The imbalance obviously makes the circuit asymmetric, somewhat reducing the positive effects one

An HP8562 spectrum analyzer was used to characterize the fabricated VCOs. The measured power spectrum is shown in Fig. 11. The oscillation frequency and output power over the tuning range are shown in Fig. 12(a). The VCO demonstrates good linearity in the output frequency, as anticipated from the dependence of the ferroelectric varactors [see Fig. 2(b)]. MHz/V. The The tuning sensitivity is output power of approximately 3 dBm fluctuates 1 dB over the tuning range, probably because the oscillator is quite strongly affected by the load. Small variations in the output load have a large impact on the oscillation amplitude since no buffer stage is used. Furthermore, the phase noise was measured using a EuropTest PN 9000 test set. This phase noise measurement system is based on the delay line discriminator technique. The oscillator output signal was down-converted by an Agilent E8244A signal generator in combination with a passive mixer before feeding the signal to the PN 9000.

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Fig. 11. Measured output spectrum of 16-GHz VCO. The span is 2 MHz and = 4:5 V, I = 63 mA, V = the resolution bandwidth is 10 kHz. V 13 V.

0

L

0

Fig. 13. (a) Measured phase noise at 16.3 GHz, (100 kHz) = 95 dBc/Hz. (b) Measured phase noise at 19.3 GHz, (100 kHz) = 102 dBc/Hz.

L

0

the VCO demonstrates a low-noise oscillation; measurements of the phase noise is shown in Fig. 13(b). At 100-kHz offset, the phase noise was measured to 102 dBc/Hz. The spectrum slope over a large region of the offset frehad the expected quency, which is in contrast to the measurement of the 16-GHz VCO. At offset frequencies above 1 MHz, the two spectra are similar. VI. DISCUSSION Fig. 12. (a) Output frequency and power over the tuning range for 16-GHz = 4:4 V, I = 63 mA. (b) Output frequency and power over the VCO. V = 2:3 V, I = 6 mA. tuning range for 19-GHz VCO. V

The measured phase noise of the oscillator at 16.3-GHz center frequency is shown in Fig. 13(a). The phase noise is measured to be 95 dBc/Hz at 100-kHz offset, and 125 dBc/Hz at 1 MHz. slope instead of the expected slope This implies a typical for this offset region. This response was repeatable; at the moment, we do not have an explanation of this dependence. An additional version of the VCO, with down-scaled distributed components and smaller varactors, was designed for an oscillation frequency of 23 GHz. However, the transistor model (Fig. 4) failed to correctly predict the involved transistor parasitics at the desired frequency range. As a result, the measured VCO operates at a lower frequency of approximately 19 GHz with a rather poor tunability of 3.3%. Fig. 12(b) shows the measured tuning characteristics of the 19-GHz VCO. Nevertheless,

Table III compares the performance of the VCO design presented in this study with the performance of a number of recently published microwave VCOs: primarily SiGe HBT-based VCOs [10]–[15]. Referred studies are all fully integrated circuits, in contrast to this study, which involves flip-chipped transistors. Further, an InGaP/GaAs VCO [16] with a reported ultra-low phase-noise characteristic and two VCOs based on BSTO varactors [6], [7] are included for comparison. As seen in Table III, room-temperature ferroelectric varactors fabricated on silicon substrates are highly competitive components for microwave VCOs. Both phase noise and tunability compare well with performance reported in other publications. For comparison, both in terms of phase noise and frequency tunability, a figure-of-merit (FOM) is defined as (4) adjusted phase noise

tunability

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TABLE III RECENTLY PUBLISHED VCOs

where is the phase noise at offset frequency , is is the frequency tunability defined the center frequency, and as

TABLE IV SUMMARY OF OSCILLATOR CHARACTERISTICS

(5) The calculated FOM of this study is the highest of the compared SiGe VCOs. In addition, the 16-GHz VCO demonstrate an oscillation frequency highly linear with tuning voltage. Despite the competitive result, the potential of these varactors is not fully exploited in this study. As seen in Fig. 12(a), the varactors are tuned from 18 to 9 V, corresponding to roughly half of the full tuning range provided by the ferroelectric varactors. Hence, if the full tuning range could be used, the frequency tunability should be approximately doubled. In simulations, the VCO is designed to use the full tuning range. Table IV compares simulated and measured results for the 16- and 19-GHz VCO. Indeed, in simulations, the frequency tunability is approximately twice the measured tunability. The explanation is that the capacitance of the fabricated varactors is higher than the capacitance used in simulations. In simulations, the zero-bias F/m , while capacitance was chosen to F/m , as the measured capacitance was seen in Fig. 2(a). Hence, in the experiment, the varactors had to be tuned with a higher voltage to meet oscillation conditions, which reduced the frequency tunability. The variation in capacitance is due to surface distribution and limited reproducibility of the thickness of the ferroelectric film, associated with the specifics of the laser ablation deposition system currently used. This system will be replaced by a magnetron sputtering deposition system, which is believed to improve the predictability of the varactor capacitances. Furthermore, for the 16-GHz VCO, the capacitive tunability over the used tuning range ( 18 to is calculated to 9 V), while the measured frequency tunability is significantly . This is mainly due to the fixed capacitance lower: connected in series with the varactor capacitance (2).

Obviously, the tunability is maximized if , yielding and . The reported VCO demonstrates , which implies . However, this is not readily confirmed since is an implicit element of the transistor model used. Moreover, the varactor is shunted by the parasitic base–emitter capacitance of the transistor, which further reduces the frequency tunability. VII. CONCLUSIONS In summary, a -band ferroelectrically tunable VCO with low phase noise and high tunability is demonstrated for the first time. The phase-noise performance of the VCO is comparable with the performance of recently published VCOs using SiGe transistors. The frequency tuning range of the 16-GHz VCO, where the output power is constant and the voltage dependence of the frequency is linear, is 6.7%. In this tuning range, the output power is approximately 3 dBm. The achieved maximum frequency tuning is 8.9%, which is among the highest reported tunabilities of VCOs using SiGe transistors. In case of the 19-GHz VCO, designed for 23 GHz, the parasitics associated with the flip-chipped transistor clearly degrades the tunability and lowers the oscillation frequency. Still, the measured phase noise is very low and, although a small tunability of 3.3%, the varactor clearly participates in the resonator tank. Regardless of the problems, the oscillators manage to demonstrate the potential of ferroelectric varactors for VCO applications at microwave frequencies requiring high tunability and low

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phase noise. Further improvement of the performance is associated with the development of the ferroelectric film technology with predictable and reproducible varactor capacitances. This will allow fully using the available tunability of the ferroelectric varactor, and substantially increase the frequency tuning range of the VCO. To a certain degree, the factor of the varactors used in the VCO is limited by the thin (0.5 m) gold electrodes and the design of the varactor [17]. Improvement of the varactor design and using thicker electrodes will allow a further increase factor. The modeling of the transistors is a separate in the problem, which was out of task of the current study. However, using transistors with better models and smaller parasitics will certainly support future development of VCOs with even lower phase noise and higher tunability.

[13] Y.-J. E. Chen, W.-M. L. Kuo, Z. Jin, J. Lee, Y. V. Tretiakov, J. D. Cressler, J. Laskar, and G. Freeman, “A low-power Ka-band voltagecontrolled oscillator implemented in 200 GHz SiGe HBT technology,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 9, pp. 1672–1679, Sep. 2005. [14] J.-H. C. Zhan, J. S. Duster, and K. T. Kornegay, “A 25-GHz emitter degenerated LC VCO,” IEEE J. Solid-State Circuits, vol. 39, no. 11, pp. 2062–2064, Nov. 2004. [15] ——, “A high f =f ratio VCO in SiGe BiCMOS technology,” IEEE Microw. Wireless Compon. Lett., vol. 15, no. 3, pp. 156–158, Mar. 2005. [16] H. Zirath, R. Kozhuharov, and M. Ferndahl, “Balanced Colpitt oscillator MMICs designed for ultra-low phase noise,” IEEE J. Solid-State Circuits, vol. 40, no. 10, pp. 2077–2086, Oct. 2005. [17] A. Vorobiev, D. Kuylenstierna, P. Rundqvist, and S. Gevorgian, “Broadband microprobe characterization of the ferroelectric films and varactors,” in Proc. Eur. Microw. Conf., 2006, pp. 843–846.

ACKNOWLEDGMENT The authors would like to thank C. Kärnfelt for advice and support on flip-chip procedure, and D. Kuylenstierna for fruitful discussions on balun design, both with the Department of Microtechnology and Nanoscience, Chalmers University of Technology, Göteborg, Sweden. K. Gnannt and R. Lachner, both with Infineon Technologies AG, Munich, Germany, are also acknowledged for supplying flip-chip transistors.

Martin Norling was born in Värnamo, Sweden, in 1981. He received the M.Sc. degree in electrical engineering from Chalmers University of Technology, Göteborg, Sweden, in 2004, and is currently working toward the Ph.D. degree in microtechnology and nanoscience at Chalmers University of Technology. His main scientific interest is the design of microwave circuits and devices based on ferroelectric materials.

REFERENCES [1] T. H. Lee, The Design of CMOS Radio-Frequency Integrated Circuits. Cambridge, U.K.: Cambridge Univ. Press, 1998. [2] G. M. Rebeiz, RF MEMS: Theory, Design, and Technology. New York: Wiley, 2003. [3] F. A. Miranda, G. Subramanyam, F. W. Van Keuls, R. R. Romanofsky, J. D. Warner, and C. H. Mueller, “Design and development of ferroelectric tunable microwave components for Ku- and K -band satellite communication systems,” IEEE Trans. Microw. Theory Tech., vol. 48, no. 7, pp. 1181–1189, Jul. 2000. [4] A. Vorobiev, P. Rundqvist, K. Khamchane, and S. Gevorgian, “Silicon substrate integrated high-Q factor parallel-plate ferroelectric varactors for microwave/millimeterwave applications,” Appl. Phys. Lett., vol. 83, pp. 3144–3146, 2003. [5] R. York, A. Nagra, E. Erker, T. Taylor, P. Periaswamy, J. Speck, S. Streiffer, and O. Auciello, “Microwave integrated circuits using thin-film BST,” in Proc. IEEE Int. Applicat. Ferroelect. Symp., 2000, pp. 195–200. [6] A. Victor, J. Nath, D. Ghosh, B. Boyette, J.-P. Maria, M. Steer, A. Kingon, and G. Stauf, “A voltage controlled oscillator using barium strontium titanate (BST) thin film varactor,” in IEEE Radio and Wireless Conf., 2004, pp. 91–94. [7] H. Xu, N. K. Pervez, and R. A. York, “Tunable microwave integrated circuits using BST thin film capacitors with device structure optimization,” Integr. Ferroelect., vol. 77, pp. 27–35, 2005. [8] M. Norling and S. Gevorgian, “Circuit model of a SiGe HBT flip-chip mounted onto a silicon carrier,” in Proc. IEEE RFIC Symp., 2006, pp. 377–380. [9] R. Aparicio and A. Hajimiri, “A noise-shifting differential Colpitts VCO,” IEEE J. Solid-State Circuits, vol. 37, no. 12, pp. 1728–1736, Dec. 2002. [10] G. De Astis, D. Cordeau, J.-M. Paillot, and L. Dascalescu, “A 5-GHz fully integrated full pMOS low-phase-noise LC VCO,” IEEE J. SolidState Circuits, vol. 40, no. 10, pp. 2087–2091, Oct. 2005. [11] M. Bao, Y. Li, and H. Jacobsson, “A 21.5/43-GHz dual-frequency balanced Colpitts VCO in SiGe technology,” IEEE J. Solid-State Circuits, vol. 39, no. 8, pp. 1352–1355, Aug. 2004. [12] H. Moon, S. Kang, Y. T. Kim, and K. Lee, “A fully differential LC-VCO using a new varactor control structure,” IEEE Microw. Wireless Compon. Lett., vol. 14, no. 9, pp. 410–412, Sep. 2004.

Andrei Vorobiev received the M.Sc. degree in physics of semiconductors and dielectrics from Gorky State University, Gorky, Russia, in 1986, and the Ph.D. degree in physics and mathematics from the Institute for Physics of Microstructures, Russian Academy of Sciences (IPM RAS), Nizhny Novgorod, Russia, in 2000. From 1986 to 1991, he was an Engineer and then the Head of the Laboratory of Microelectronics, Design Office of Measuring Instruments, Gorky, Russia, where his research interests are in the area of the development of technology of hybrid film microwave integrated circuits. In 1991, he joined the Technology Division, IPM RAS, as Senior Research Associate, where his research interests were in the area of preparation and investigation of high-temperature superconductor films and multilayer structures. Since 2001, he has been with the Department of Microtechnology and Nanoscience, Chalmers University of Technology, Göteborg, Sweden, initially as a Post-Doctoral Fellow and then as a Researcher. His current research interest is in the area of development of fabrication of ferroelectrically tunable devices for microwave applications.

Harald Jacobsson (M’00) received the M.S. degree in engineering physics and the Ph.D. degree in physics from Chalmers University of Technology, Göteborg, Sweden, in 1986 and 1993, respectively. From 1994 to 1996, he was involved with the epitaxy and characterization of SiGe and SiGeC films with Arizona State University. Since 1996, he has been with the Microwave and High Speed Electronics Research Center, Ericsson Research, Ericsson AB, Mölndal, Sweden, where he is involved with the design of Si-based integrated circuits for microwave radio applications. During this time, he has also been the Manager of the MMIC and RFIC Design Group within the microwave link organization of Ericsson Research.

NORLING et al.: LOW-NOISE

-BAND VCO BASED ON ROOM-TEMPERATURE FERROELECTRIC VARACTORS

Spartak Gevorgian (M’96–SM’97) received the M.S. degree in radioelectronics from Yerevan Polytechnic, Yerevan, Armenia, in 1972, and the Ph.D. and Dr.Sci. degrees from the Electrotechnical University, St. Petersburg, Russia, in 1977 and 1991, respectively. From 1972 to 1993, he held different research and teaching positions with the Polytechnic Institute and Electrotechnical University. From 1993 to 1998, he held research positions with Chalmers University of Technology, Göteborg, Sweden. Since 1998, he has been a Professor with Chalmers University of Technology. Since 1996, he has also worked part time with Ericsson Microwave Systems AB, Mölndal, Sweden.

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He has authored or coauthored over 270 papers and conference presentations. He holds over 30 patents/patent applications. His research interests are physics, design, and experimental investigation of microwave devices and components including tunable filters, delay lines, phase shifters, etc., based on bulk and thin-film ferroelectrics integrated with silicon substrate, silicon RF integrated circuits (RFICs) and monolithic microwave integrated circuits (MMICs), optimization of passive components in foundry-based MMICs (VCOs, amplifiers, etc.), microwave photonic devices (optically controlled components based on silicon, photonic generation of microwaves), and modeling of passive coplanar components based on conformal mapping. Dr. Gevorgian was the recipient of scholarships from University College London (1981–1982) and the Electrotechnical University (1988–1991).

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A Ferroelectric-Capacitor-Based Tunable Matching Network for Quad-Band Cellular Power Amplifiers Ali Tombak, Member, IEEE

Abstract—A tunable matching network (TMN) based on ferroelectric capacitors for cellular power-amplifier applications was designed, fabricated, and tested. When the matching network is operated for GSM850/900 frequency bands, the input to impedance varied from with a power gain variation of 1.35 to 1.62 dB. When the circuit is operated for digital communication system/personal communication system/wideband code division multiple access frequency bands, the input impedance varied from to with a power gain variation of 2.05 to 1.79 dB. The nonlinearity of the TMN was also characterized on an on-wafer load–pull system using a 1.95-GHz test signal with CDMA2000 and high-speed downlink packet access modulation strings.

3 44

1 87

3 55

0 02

4 28

4 31 + 0 97

0 06

Index Terms—Barium–strontium–titanate (BST), ferroelectric, output matching network, paraelectric, power amplifier (PA), tunable matching network (TMN), varactor.

I. INTRODUCTION HE RECENT advancement in wireless communications demands an ever increasing improvement in system performance and functionality with a reduced size and cost. For example, with the cellular phones operating in the 850/900- and 1800/1900-MHz bands, the global positioning system (GPS) in the 1.5-GHz band, and the wireless local area networks (WLANs) in the 2.4/5-GHz bands, it is desirable to combine two or more of these standards into a single wireless unit. Dual-band transceivers/power amplifiers (PAs) have recently been introduced to increase the functionality of wireless communication systems by switching between two different bands [1], [2].1 Using conventional design techniques, this is achieved by building multiple independent components for each band such as filters, amplifiers, oscillators, synthesizers (and, therefore, antennas) at the expense of an undesirable increase in the complexity, size, cost, and power dissipation. While there have been efforts to minimize the number of additional components used for dual-band/multiband applications to reduce the complexity, footprint, and cost, researchers in industry and academia are also considering tunable RF/microwave components, which can address these challenges by designing

T

Manuscript received August 22, 2006; revised September 28, 2006. The author is with Corporate Research and Development, RF Micro Devices Inc., Greensboro, NC 27409 USA (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TMTT.2006.889349 1Part Numbers: RF3166, RF3177, RF3178, RF5144, RF6026, RF Micro Devices Inc., Greensboro, NC.

electronically tunable subsystems [3]–[8].2 3 The components can, therefore, function optimally as conditions change (such as the frequency of operation). Depending on the specific circuit where such tunable components are used, the result may be a savings in cost, reduction in size, improvement in battery life, better performance, and/or increased functionality. These components can be designed using various technologies such as microelectromechanical systems (MEMS), ferroelectric materials, solid-state varactor diodes, and ferrites. Among these techniques, electrical tuning of a capacitor is the simplest and most common choice for making tunable circuits. When the value of a capacitor in a circuit is changed, the impedance and phase relationships in the circuit are affected in predictable ways. Therefore, these parameter changes can be exploited to design RF/microwave components with greater functionality, higher performance, and reduced size and cost. A major application of these tunable capacitors is in the area of PAs. Many wireless communication systems, such as cellular phones, wireless modems, and wireless personal digital assistants (PDAs), require RF PAs to boost the signals before transmission through the antenna. In most cases, the PA is the highest power-consuming part of the radio, and occupies a significant amount of die space. High-quality matching network elements are usually used at the output of the PA so that minimal power is lost in the matching network. For quad-band cellular applications such as GSM850, GSM900, digital communication systems (DCSs), and personal communication systems (PCSs), there would usually be two signal paths, i.e., GSM850/900 [low band (LB)] and DCS/PCS [high band (HB)] bands. Using fixed matching networks, the PAs are designed to meet the power requirements of the LB and HB. Having two signal paths increases the die space, module size, and the entire cost of the PA. Using a tunable matching network (TMN), the same active circuits can be used for both bands, which substantially reduces the size and cost of the entire PA module [9]–[13]. Of course, depending on the number of stages in the PA, the interstage matching networks should also be tunable or some sort of interstage-tuning/designing mechanism should be employed. In addition, the TMN helps when there is antenna impedance mismatch due to a nearby object, etc., which directly translates itself into nonoptimal impedance match at the drain/collector of the final stage transistor, and reduced output power and efficiency. By retuning the output matching network, the PA can still be operated at near-optimal performance when there is an 2Tunable filters, phase shifters, TMNs, Agile Materials and Technologies Inc., Goleta, CA. 3Tunable filters, smart antennas, miniaturized RF front-ends, Paratek Microwave Inc., Columbia, MD.

0018-9480/$25.00 © 2007 IEEE

TOMBAK: FERROELECTRIC-CAPACITOR-BASED TMN FOR QUAD-BAND CELLULAR PAs

Fig. 1. C–V curve for a typical ferroelectric capacitor used in this study.

TABLE I MAXIMUM–MINIMUM VALUE AND SIZE OF THE FERROELECTRIC CAPACITORS USED IN THIS STUDY

antenna impedance mismatch. The TMNs can also be used in maintaining high efficiency during reduced output power levels by presenting the optimal impedance at the final stage drain/collector of the PA for that output power level. In this paper, the design, simulations, fabrication, and measurement results for a quad-band TMN based on ferroelectric tunable capacitors for cellular PA applications are given. Ferroelectric varactors offer high tunability, high power-handling capability, fast tuning speeds, and low control voltages. The nonlinearity of the TMN is also quantified by measuring the adjacent channel power ratio (ACPR) degradation using CDMA2000 and high-speed downlink packet access (HSDPA) modulation strings. II. DESIGN AND FABRICATION The ferroelectric-based tunable capacitors usually have a bell-shaped nonlinear C–V relationship, which is symmetric across the -axis, as shown in Fig. 1. With the application of dc bias, the capacitance can be changed nonlinearly. The ferroelectric material used in the tunable capacitors was ParaScan, which is a proprietary form of doped barium–strontium–titanate (BST). Three sizes of ferroelectric capacitors were used in this study, as outlined in Table I. These capacitors are expected to have an effective series resistance (ESR) of 200 m to 1 for bias voltages varying from 0 to 30 V, respectively. In today’s cellular standards, the PA usually delivers approximately 35 dBm at GSM850/900-MHz bands (LB), and 33 dBm at DCS/PCS bands (HB). Depending on the technology utilized (GaAs HBT, Si MOS, pseudomorphic HEMT (pHEMT), etc.) to fabricate the PA, the load line usually varies from 2 to 4 at LB, and from 3 to 5 at HB. Using the available ferroelectric capacitors and a coplanar waveguide (CPW) line, a regular low-pass type output matching network, which would give 3–3.5- load line ( ) at LB and 4–4.5 load line ( ) at HB was designed

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as shown in Fig. 2. The ferroelectric capacitors were connected to the ground pad of the CPW line using conductive epoxy. Gold bondwires were used to connect both ends of the ferroelectric capacitors to the CPW line and ground. Six bondwires were used for the first capacitor, which had a bigger bond pad, whereas four bondwires were used for the second and third capacitors. It is estimated that the six bondwires would have 0.19-nH parasitic inductance and four bondwires would have 0.24-nH parasitic inductance using the Philips/Delft bondwire model, which is available in Agilent’s ADS design software. The CPW line was fabricated on a 5-mil-thick RT/Duroid 6002 substrate with a dielectric constant and loss tangent of 2.94 and 0.012, respectively. The conductor thickness was 2 mil. The width and gap of the CPW line was 9.1 and 4 mil, respectively. The length of mil, mil, the CPW lines were and mil. The ferroelectric capacitors are biased through a bias resistor of approximately 50 k to both suppress RF signals and provide a dc signal path to bias the capacitors. In applications that may require pulsing the dc bias such as global system for mobile communication (GSM) multislot transmission, this resistor should be chosen carefully so that adequate amount of RF suppression is realized while ensuring the RC time constant does not severely affect the rise/fall times of the pulse in the circuit. A single bondwire was also connected from the resistor to a five-line bias line to be probed through a probe card. A photograph of the fabricated TMN is shown in Fig. 3. The circuit of Fig. 2 was simulated in Agilent’s ADS design software. The real and imaginary parts of the simulated input ) for a 50- output impedance and the associimpedance ( ated power gain for both LB and HB are shown in Figs. 4–7. When the circuit is operated for LB by setting the ferroelectric pF, pF, and pF, the capacitors to to input impedance varied from with a power gain variation from 1.15 to 1.34 dB for the GSM850/900 frequency bands. When the circuit is operated pF, for HB by setting the ferroelectric capacitors to pF, and pF, the input impedance ranged to with a power gain varifrom ation of 1.41 to 2.46 dB for the DCS/PCS/wideband code division multiple access (WCDMA) frequency bands. It should be noted that this design was optimized based on the available ferroelectric capacitors and the CPW line. Lower loss TMNs can be designed by optimizing the min/max values of the ferroelectric capacitors and the CPW linewidth and characteristic impedance. III. MEASUREMENT RESULTS The fabricated circuit was tested using an Agilent PNA network analyzer. A line-reflect-match (LRM) calibration using GGB 500- m pitch ground–signal–ground (GSG) probes was made. The bias connection to the circuit was provided through a custom designed probe card. The dc bias was applied using an Agilent semiconductor parameter analyzer. The measured real ) and the assoand imaginary parts of the input impedance ( ciated power gain for both LB and HB are shown in Figs. 8–11. When the circuit is operated for LB by setting the capacitor V, V, and V, the voltages to input impedance varied from to

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Fig. 2. Schematic of the TMN.

Fig. 3. Photograph of a TMN.

Fig. 5. Simulated power gain when the TMN is tuned for LB.

Fig. 4. Simulated real and imaginary parts of the input impedance when the TMN is tuned for LB.

with a power gain variation from 1.35 to 1.62 dB for the GSM850/900 frequency bands. When the circuit is operated for V, HB by setting the capacitor voltages to V, and V, the input impedance varied from to with a power gain variation from 2.05 to 1.79 dB for the DCS/PCS/WCDMA frequency bands. For the LB operation, the real part of the input impedance is predicted very well by the simulations, and the imaginary part is 0.4 more capacitive, which may be related with the probe over-travel distance. For the HB operation, a flatter real part is obtained compared to measurements, and the imaginary part is, again, more capacitive. The measured power gain at LB is approximately 0.3 dB worse compared to simulations, whereas at HB, it varies 0.5 dB compared to simulations. The deviations

Fig. 6. Simulated real and imaginary parts of the input impedance when the TMN is tuned for HB.

in the power gains and partly in the input impedances should be due to the inaccuracies during the fabrication, the lack of an electromagnetic simulation of the entire structure, and the variations in the ESR of the ferroelectric capacitors. Since this TMN utilizes ferroelectric capacitors with a nonlinear C–V relationship, a clean modulated signal with no spectral regrowth at the input of this TMN will be distorted at the output. In order to test the nonlinearity of this TMN, a test signal with CDMA2000 and HSDPA modulation strings at 1.95 GHz was applied on a Maury on-wafer load–pull system. The TMN

TOMBAK: FERROELECTRIC-CAPACITOR-BASED TMN FOR QUAD-BAND CELLULAR PAs

Fig. 7. Simulated power gain when the TMN is tuned for HB.

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Fig. 10. Measured real and imaginary parts of the input impedance when the TMN is tuned for HB.

Fig. 8. Measured real and imaginary parts of the input impedance when the TMN is tuned for LB.

Fig. 11. Measured power gain when the TMN is tuned for HB.

Fig. 9. Measured power gain when the TMN is tuned for LB.

Fig. 12. Measured ACPR and alternate channel power ratio (ACPR1 and ACPR2, respectively) of the system and the TMN as a function of incident power for CDMA2000 modulation string at 1.95 GHz.

was configured for harmonic-balance (HB) operation, and the source impedance of the source tuner was matched to the input impedance of the TMN so that maximum power is transferred into the matching network. The measured ACPR and alternate channel power ratio (ACPR1 and ACPR2, respectively) along with the system’s reference ACPRs on a thru as a function of the incident power are shown in Figs. 12 and 13 for CDMA2000 and HSDPA modulation strings, respectively. For both modulation strings at low input power levels, no significant ACPR1 and

ACPR2 degradation was measured. However, when the input power increases from 10 to 30 dBm, both ACPR1 and ACPR2 were degraded. For CDMA2000, the ACPR1 and ACPR2 were degraded by up to 17 and 20 dB, respectively. For HSDPA, the ACPR1 and ACRP2 were degraded by up to 19 and 23 dB, respectively. Therefore, one should consider this effect while designing a matching network using nonlinear capacitors. In addition, the measured ACPRs can still be improved by carefully

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1.62 dB. At DCS/PCS/WCDMA frequency bands, the input to with a impedance varied from power gain variation from 2.05 to 1.79 dB. A good agreement was obtained between the simulated and measured input impedances. The nonlinearity of the TMN was also tested on an on-wafer load–pull system using a 1.95-GHz input signal with CDMA2000 and HSDPA modulation strings. The ACPR and alternate channel power ratio of the system were degraded up to 19 and 23 dB, respectively, due to the nonlinearity of the ferroelectric capacitors. A discussion on how to reduce this nonlinearity and the loss of the TMN was made.

ACKNOWLEDGMENT Fig. 13. Measured ACPR and alternate channel power ratios (ACPR1 and ACPR2, respectively) of the system and the TMN as a function of incident power for HSDPA modulation string at 1.95 GHz.

designing the circuit such as biasing the capacitors at a more linear region if one has the freedom to choose the capacitor area. Two or more capacitors can be connected in series as well so that the RF voltage, hence, the ACPR, decreases across the unit capacitor. These capacitors can also be utilized in applications that do not require stringent linearity requirements such as in GSM PAs, although the PA has to satisfy AM/AM, AM/PM, and harmonic power specifications. The TMN presented in this paper offers a single circuit solution for LB and HB operation of PAs. The required impedances were successfully realized for GSM/EDGE handset PAs. For these systems, obtaining high power-added efficiency (PAE) for the PA is very crucial. Therefore, the output matching network loss should be minimized. Currently, fixed output matching network losses are usually in the range of 0.4–0.5 dB at LB and 0.6–0.8 dB at HB for these systems. The measured loss for the TMN was higher than these requirements; however, using the same ferroelectric capacitor technology (i.e., the ferroelectric material), the loss of this matching network can certainly be reduced by optimizing the printed circuit board (PCB) properties, capacitor area, and capacitor bond-pad size towards the application. This technology can also be utilized in applications where PAE is not the first priority such as WLAN applications or cellular machine to machine communications applications. The measured loss for the TMN is also compatible with other researchers in this field [12], [13]. For example, in [12], the Chen et al. achieve approximately 0.3-dB loss with an impedance transformation ratio of 3.8 (compared to 50 ), whereas the TMN presented here has an impedance transformation ratio of 14.3 and, hence, higher loss.

IV. CONCLUSION A TMN based on ferroelectric capacitors for cellular PA applications was designed, fabricated, and tested. At GSM850/900 frequency bands, the input impedance varied from to with a power gain variation from 1.35 to

The author wishes to acknowledge R. Baeten, J. Jorgenson, Dr. D. Halchin, and R. Osmani, all with RF Micro Devices Inc., Greensboro, NC, for useful discussions and feedback, C. Lineberry, L. Hill, and S. Dorn, all with RF Micro Devices Inc., for their assistance in the fabrication and testing of the TMN, and Paratek Microwave Inc., Columbia, MD, for supplying the ferroelectric varactors.

REFERENCES [1] S. Wu and B. Razavi, “A 900-MHz/1.8-GHz CMOS receiver for dualband applications,” IEEE J. Solid-State Circuits, vol. 33, no. 12, pp. 2178–2185, Dec. 1998. [2] J. Tham, M. Margrait, B. Pregardier, C. Hull, R. Magoon, and F. Carr, “A 2.7–V 900-MHz dual-band transceiver IC for digital wireless communication,” IEEE J. Solid-State Circuits, vol. 34, no. 3, pp. 286–291, Mar. 1999. [3] A. Tombak, J. P. Maria, F. T. Ayguavives, Z. Jin, G. T. Stauf, A. I. Kingon, and A. Mortazawi, “Voltage controlled RF filters employing thin film barium strontium titanate tunable capacitors,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 2, pp. 462–467, Feb. 2003. [4] A. Tombak, J.-P. Maria, F. T. Ayguavives, Z. Jin, G. T. Stauf, A. I. Kingon, and A. Mortazawi, “Tunable barium strontium titanate thin film capacitors for RF and microwave applications,” IEEE Microw. Wireless Compon. Lett., vol. 12, no. 1, pp. 3–5, Jan. 2002. [5] E. R. Brown, “RF-MEMS switches for reconfigurable integrated circuits,” IEEE Trans. Microw. Theory Tech., vol. 46, no. 11, pp. 1868–1880, Nov. 1998. [6] D. Peroulis, S. Pacheco, K. Sarabandi, and L. Katehi, “Tunable lumped components with applications in reconfigurable MEMS filters,” in IEEE MTT-S Int. Microw. Symp. Dig., 2001, pp. 341–344. [7] Y. Liu, A. Borgioli, A. S. Nagra, and R. A. York, “Distributed MEMS transmission lines for tunable filter applications,” Int. J. RF Microw. Comput.-Aided Eng., vol. 11, pp. 254–260, 2001. [8] A. Abbaspour-Tamijani, L. Dussopt, and G. Rebeiz, “Miniature and tunable filters using MEMS capacitors,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 7, pp. 1878–1885, 2003. [9] H. Zhang, H. Gao, and G.-P. Li, “Broad-band power amplifier with a novel tunable output matching network,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 11, pp. 3606–3614, Nov. 2005. [10] D. Qiao, R. Molfino, S. M. Lardizabal, B. Pillans, P. M. Asbeck, and G. Jerinic, “An intelligently controlled RF power amplifier with a reconfigurable MEMS-varactor tuner,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 3, pp. 1089–1095, Mar. 2005. [11] Q. Shen and N. S. Barker, “Distributed MEMS tunable matching network using minimal-contact RF-MEMS varactors,” IEEE Trans. Microw. Theory Tech., vol. 54, no. 6, pp. 2646–2658, Jun. 2007. [12] L.-Y. Chen, R. Forse, D. Chase, and R. A. York, “Analog tunable matching network using integrated thin-film BST capacitors,” IEEE MTT-S Int. Microw. Symp. Dig., vol. 1, pp. 261–264, Jun. 2004. [13] P. Scheele, F. Goelden, A. Giere, S. Mueller, and R. Jakoby, “Continuously tunable impedance network using ferroelectric varactors,” in IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2005, pp. 603–606.

TOMBAK: FERROELECTRIC-CAPACITOR-BASED TMN FOR QUAD-BAND CELLULAR PAs

Ali Tombak (S’99–M’04) received the B.Sc. degree in electrical engineering from the Middle East Technical University, Ankara, Turkey, in 1999, the M.Sc. degree in electrical engineering from North Carolina State University, Raleigh, in 2000, and the Ph.D. degree in electrical engineering from The University of Michigan at Ann Arbor, in 2004. From 1998 to 1999, he was with ASELSAN Military Electronics Inc., Ankara, Turkey. From 1999 to 2001, he was a Research Assistant with North Carolina State University. From 2001 to 2004, he was a Graduate Student Research Assistant with The University of Michigan at Ann Arbor. He is currently a Senior Design Engineer with the Corporate

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Research and Development Group, RF Micro Devices Inc., Greensboro, NC. His research interests include novel RF/microwave components employing ferroelectric and solid-state tunable capacitors for the design of reconfigurable wireless communication systems, balanced PA architectures for enhanced insensitivity against antenna impedance mismatch, high-efficiency silicon LDMOS PAs, PA linearizers, and integration of PAs with the transceiver. Dr. Tombak is a member of the IEEE Microwave Theory and Techniques Society (IEEE MTT-S) since 1999. He was the recipient of a 2003 Rackham Graduate School Travel Grant. He was the recipient of a 1999 Turkish Scientific and Technical Research Council (TUBITAK) North Atlantic Treaty Organization (NATO) scholarship. He was also the recipient of the First Degree in the National Science Olympiads on physics organized by TUBITAK in 1993.

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Frequency and Bandwidth Agile Millimeter-Wave Filter Using Ferroelectric Capacitors and MEMS Cantilevers Cesar Lugo, Member, IEEE, Guoan Wang, Member, IEEE, John Papapolymerou, Senior Member, IEEE, Zhiyong Zhao, Member, IEEE, Xiaoyan Wang, Member, IEEE, and Andrew T. Hunt

Abstract—This paper demonstrates for the first time a tunable bandpass filter with simultaneous frequency and bandwidth control using a combination of ferroelectric barium–strontium–titanate capacitors and cantilever microelectromechanical systems (MEMS) switches. The center frequency of the filter is tuned in a continuous fashion from 30 to 35 GHz with insertion loss ranging from 10 to 2.7 dB. The fractional bandwidth of the filter can also be independently controlled by a tuning scheme that uses MEMS switches to vary the inter-resonator coupling. The two-pole filter prototypes resulted in fractional bandwidths of 9.6% (wideband configuration) and 4.8% (narrowband configuration) for a tuning ratio of approximately 2 : 1. The third-order filters resulted in bandwidths of 7.8% (wideband configuration) and 3.1% (narrowband configuration) for a passband tunable ratio of approximately 2.5 : 1. Index Terms—Degenerate modes, dual-mode filter, dual-mode resonator, triangular loop resonator, triple-mode resonator.

I. INTRODUCTION

I

N RECENT years, evolving wireless communications have increased the demand for versatile technologies with adaptable frequency behavior. For this reason, components with multiband coverage and multifunctional capabilities are becoming an important technological trend. This trend has led to the constant analysis of circuits with reconfigurable filtering functions [1]–[4]. This is partially due to the fundamental roll of selective filters in RF front-end electronics, and partially due to the vast variety of tunable filter applications. For example, a filter capable of selecting different frequency bands may replace a conventional filter bank reducing size and cost. Radar systems, may employ a bandwidth adjustable filter to eliminate out-of-band jamming spectral components. Communication systems with multiband transceivers may also adapt a filter capable of synchronizing to different information channels. In addition to the increasing need for filters with agile frequency behavior, modern systems place stringent requirements in terms of low loss, low cost, small size, in-band phase delay

Manuscript received May 1, 2006; revised October 16, 2006. This work was supported by the Georgia Electronic Design Center. C. Lugo, G. Wang, and J. Papapolymerou are with the School of Electrical and Computer Engineering, Georgia Institute of Technology, Atlanta, GA 30332 USA (e-mail: [email protected]; [email protected]). Z. Zhao, X. Wang, and A. T. Hunt are with the nGimat Company, Atlanta, GA 30341 USA (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TMTT.2006.889320

Fig. 1. Proposed filter schematic. (a) Two-pole filter. (b) Three-pole filter.

flatness, and dynamic range. Many research efforts are now focused on the development of tunable RF filters using variable reactance elements such as varactor diodes [5], p-i-n diodes [6], and microelectromechanical systems (MEMS) switches [7], [8]. The work presented in [9] shows a lumped-element filter with frequency control using barium–strontium–titanate (BST) films. Past efforts have also demonstrated reconfigurable filter topologies based on dual-mode resonators [10]. This paper reports for the first time a hybrid tunable filter topology in the millimeter-wave range that combines ferroelectric capacitors and MEMS switches. Ferroelectric materials such as the BST capacitors presented here show high controllable characteristics, making them an attractive technology and a major factor in the future generation of tunable components. This paper also demonstrates for the first time a bandwidth tuning scheme that uses MEMS cantilevers to control inter-resonator coupling in a coplanar waveguide (CPW) configuration. II. FILTER DESIGN A. Proposed Topology Schematics of the proposed second- and third-order filters are shown in Fig. 1(a) and (b), respectively. The filters are designed using a CPW end coupled resonator topology with a system

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Fig. 2. Cross section of a BST gap capacitor (not to scale).

impedance . Given the single plane for signal and ground, this topology avoids the need of via-holes and greatly facilitates the introduction of tunable elements such as loading BST capacitors and MEMS switches. The resonator lengths are , where is the guided wavelength at the approximately GHz. The goal is to produce a design center frequency Chebyshev bandpass filter with continuously tunable center frequency and adjustable fractional bandwidth.

Fig. 3. Capacitance versus bias voltage of a BST gap capacitor.

B. Frequency Control Using BST Capacitors Initially, the filter is analyzed with unloaded resonators. The absence of BST loading capacitors produces the conventional resonant poles with a center frequency GHz and res. Frequency control is achieved when onator length the shunt variable BST capacitors are added at the resonator ends. These capacitors effectively increase the electrical length of the resonators causing a frequency shift in the resonant GHz with a zero bias poles to approximately voltage V . Consequently, when the capacitors are biased, the center frequency shifts to higher frequencies GHz until a saturation voltage point at V. Hence, the loading capacitor values are strategically varied V V changes when an applied dc voltage the dielectric constant of the BST film. This tuning procedure changes the center frequency of the filter in a continuous fashion. The BST capacitors were designed using a planar configuration, i.e., metallization was carried out after the BST deposition. Fig. 2 shows a cross section of the BST gap capacitors. The planar configuration requires fewer lithography steps and enables thicker metal to be deposited for lower metal losses. Most importantly, epitaxial BST thin films can be grown on singlecrystal substrates ensuring lower dielectric losses. In order to reduce the dc-bias voltage, the recently developed interdigitated capacitor structures [11] were employed at a 1- m separation. The tuning voltage can be reduced by decreasing the separation. In these structures, the dc-bias line is separated from the RF signal by using a highly resistive thin layer of material [e.g., indium tin oxide (ITO)]. Important characteristics of a thin-film BST capacitor include tunability and dielectric constant of the BST. Studies have shown that these electrical properties are strongly affected by the crystalline structure, microstructure, dopants, composition, and its thickness of the BST films, as well as electrode material and thickness. Numerous groups have attempted different growth techniques to improve BST film quality such as pulsed laser deposition [12], metal–organic chemical vapor deposition [13], sputtering [14], and sol-gel [15]. The nGimat Company,

Fig. 4. Layout of MEMS tuning mechanism for bandwidth control.

Atlanta, GA, has developed its proprietary combustion chemical vapor deposition (CCVD) process [16] for depositing epitaxial BST films on sapphire. The tunability of the capacitor was recorded at 1 MHz using a precision LCR meter, as illustrated in Fig. 3. The capacitance at zero bias is 68 fF, which reduces to 35 fF at 40 V. By varying the capacitor lengths and gapwidths, one can achieve different capacitances and tunability, which will yield filters with various tunable frequency ranges. C. Bandwidth Control Using MEMS Switches The spectral separation of the resonant poles in edge-coupled resonator filters is directly related to the inter-resonator coupling strength. The coupling between adjacent resonators is due almost entirely to the fringing electric field produced at the resonator edges. A mechanism to control the inter-resonator coupling is achieved by a switchable structure that effectively changes the amount of electric field energy coupled from one resonator to the next. This procedure is equivalent to a waveguide adjustable iris and it is demonstrated in a CPW configuration for the first time. The tuning structure consists of a conductive path placed between adjacent resonators. As shown in Fig. 4, this conductive path is connected or isolated to the ground planes using a set of MEMS switch cantilevers. Fig. 5 shows the simulated responses of the two-pole filter in the wideband and narrowband states. GHz corThe tuning mechanism is analyzed here at responding to a pole location in the wideband response and a 10-dB rejection at the narrowband response. A full-wave simulation is conducted using Sonnet 10.52. The simulation emulates the down state of the MEMS switches using a perfect short, and the up state of the MEMS switches using a perfect

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Fig. 7. Geometry of the meander-shaped support.

Fig. 5. Simulated response. Marker at 42.5 GHz corresponds to a pole location in the wideband response and a 10-dB rejection in the narrowband response.

Fig. 8. Fabricated filter.

Fig. 6. Current density distribution at 42.5 GHz (a) Wideband state. (b) Narrowband state.

open. Fig. 6 shows the current densities of the two-pole filter at GHz in both the wideband sate (MEMS switches up) [see Fig. 6(a)] and narrowband state (MEMS switches down) [see Fig. 6(b)]. When the conductive path is isolated, it functions as a floating metal having little to no effect in the electric field coupling between resonators; this can be seen in Fig. 6(a), where the floating path shows a near-zero current density, while the resonators show a high current density. This configuration produces the highest inter-resonator coupling strength, which translates into a wideband state. A wideband filter characteristic is achieved when the resonant poles are placed farther away from each other producing a broader in-band frequency profile. The second configuration of the filter is produced when the MEMS cantilever are in the down state. At this point, the conductive path is grounded, as it is connected to the CPW ground planes. This grounded conductive path effectively contains the amount of electric field energy at the resonator edges reducing the inter-resonator coupling. Fig. 6(b) shows a higher current density at the conductive path and an attenuated current density at the resonators. This configuration produces the narrowband state of the filter by grouping the resonant poles closer together across the passband. A fractional bandwidth of the filters can be adjusted to a tuning ratio of 2 : 1 for second-order filters and 2.5 : 1 for third-order filters.

Fig. 9. Effect of fixed input/output coupling and BST loss on filter response.

The pull-in voltage of the MEMS switch can be calculated from the effective spring constant of the membrane support as

(1) where is the effective spring constant of the membrane, is the initial gap between the switch and the bottom electrode, is the area of the membrane, and is the permittivity of air. When designing for low actuation voltage, the choice of the membrane material and support design is critical. In order to lower the pull-in voltage of the structure, three different ways can be used, which are: 1) increasing the area of membrane; 2) diminishing the gap between the switch and bottom electrode; and 3) designing a structure with low spring constant. In the first case, the area can only be increased by so much before device size becomes a prevailing issue. In the second case, the return loss associated with the RF signal restricts the value of the gap. The

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Fig. 10. Two-pole filter measured wideband state (MEMS switches up). (a) Transmission loss. (b) Return loss.

Fig. 11. Two-pole filter measured narrowband state (MEMS switches down). (a) Transmission loss. (b) Return loss.

third case is the one with the most flexibility since the design of the springs does not considerably impact the size, weight, and/or RF performance of the circuit. To reduce the actuation voltage, MEMS cantilevers in this paper were designed with a meander-shaped support. The effective spring constant of a meander-shaped structure as shown in Fig. 7 is given by

(2)

where and are the Young’s modulus and Poisson’s ratio, respectively. The spring constant for of such structures conand , respecnected in series and parallel are tively. For switches that use gold for the cantilever material, the

expected pull-in voltages are in the range of 20–30 V. In addition, actuation voltage simulation for the designed switch with the meander shape support is also done using FEMLAB, and the result showed that the actuation voltage was around 30 V. III. FILTER FABRICATION The BST films were epitaxially grown on -plane sapphire substrates using the nGimat Company’s proprietary CCVD process [17], [18], After CCVD deposition, the BST films were patterned using a diluted HF solution, following which high resistive bias lines were deposited and patterned by wet etching. Metallizations were then carried out using a liftoff process to form the capacitors, the CPW lines, and the activation pads for the MEMS switches. A Ti/Cu/Au metal stack was used with a total thickness of 2 m for the former two structures and 0.8 m for the latter. The filter was later passivated using benzocyclobutene (BCB) photosensitive polymers. The last step before MEMS fabrication was a metal layer forming the

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Fig. 12. Three-pole filter measured wideband state (MEMS switches up). (a) Transmission loss. (b) Return loss.

Fig. 13. Three-pole filter measured narrowband state (MEMS switches down). (a) Transmission loss. (b) Return loss.

bias pads and lines, which were on top of the BCB layer. For the MEMS switches, a 300-nm-thick Si N layer was first deposited using PECVD, and patterned and etched with reactive ion etching (RIE) between the membrane and the signal line. A 1.8- m-thick photoresist 1813 was then spin coated and patterned to create the air gap. Ti/Au/Ti (300 A/3000 /300 A) seed layer was then evaporated and patterned and electroplated to a thickness of 3 m to form the switch membrane. Finally, after removing the sacrificial photoresist layer with a resist stripper, a critical point drying process was used to release the switches. Fig. 8 shows a picture of the fabricated two-pole filter. IV. RESULTS AND DISCUSSION Conventionally, a capacitive coupled resonator filter would require different critical coupling at the ports. This can be accomplished with a design that includes a variable capacitor connecting the input port to the first resonator and the output port to the th resonator. This option requires an added degree of

complexity and it has been avoided in this design. Since the port couplings are fixed, a simulation has been conducted to model the deterioration of the response due to port mismatch and due to the loss of the BST capacitors. The BST loss tangent varies at V and capacitance of 68 fF from at V and capacitance of 35 fF. to A full-wave simulation was conducted on the two-pole filters including these capacitor characteristics. Fig. 9 shows the passband and return loss degradation for the lower frequency cases. The filters where measured using an Agilent HP 8510 vector network analyzer. An on-wafer short, open, load, thru (SOLT) standard was used for calibration. The wideband and narrowband two-pole filter measured results are shown in Figs. 10 and 11, respectively. The insertion loss of the wideband state ranges from 9 dB V to 2.7 dB V . The midband return loss was 4 dB for biasing voltages below 5 V and 8 dB or better for the remaining cases. The narrowband state yielded an expected higher loss ranging from 16 dB

LUGO et al.: FREQUENCY AND BANDWIDTH AGILE MILLIMETER-WAVE FILTER USING FERROELECTRIC CAPACITORS AND MEMS CANTILEVERS

V to 5 dB V . The return losses ranged from 7 dB for biasing voltages below 5 V and 15 dB or better for higher voltages. The fractional bandwidth of the two-pole filter averaged 9.6% for the wideband state and 4.8% in the narrowband state. The bandwidth of the three-pole filters resulted in 7.8% and 3.1% for the wideband and narrowband, respectively. The third-order filter wideband and narrowband measurement results are shown in Figs. 12 and 13, respectively. The wideV and band state yielded an insertion loss of 16 dB V . The return loss was better than 7 dB for all 6 dB voltage cases. The narrowband state resulted in insertion losses V and 8 dB V . The return of 19 dB loss was 8 dB or better for all biasing voltages. As discussed before, the high insertion and return loss is attributed to conductor losses, fixed input/output port couplings, and the thin-film BST loss at the different biasing voltages. In order to reduce loss and improve performance of tunable devices, efforts have been made to explore novel materials such as bismuth–zinc–niobate [19] and optimize the electrode structure of BST capacitors [20]. The filter loss could also be further reduced if a better return loss is achieved. V. CONCLUSION This paper has demonstrated for the first time a tunable bandpass filter with both frequency and bandwidth control using a combination of ferroelectric (BST) capacitors and cantilever MEMS switches. The center frequency of the filter is tuned in a continuous fashion from 30 to 35 GHz with insertion loss ranging from 10 to 2.7 dB. The fractional bandwidth of the filter can also be independently controlled by a tuning scheme that uses MEMS switches to vary the inter-resonator coupling. The passband tuning ratio achieved by the filter is 2 : 1 for the two-pole filter design and 2.5 : 1 for the three-pole filter design.

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[8] A. Tamijani, L. Dussopt, and G. Rebeiz, “A high performance MEMS miniature tunable bandpass filter,” in IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2003, pp. 1785–1788. [9] D. Kuylenstierna, A. Vorobiev, and S. Gevorgian, “40 GHz lumped element tunable bandpass filters with transmission zeros based on thin Ba Sr TiO (BST) film varactors,” in Silicon Monolithic Integr. Circuits in RF Syst. Top. Meeting, 2006, pp. 342–345. [10] C. Lugo and J. Papapolymerou, “Single switch reconfigurable bandpass filter with variable bandwidth using a dual-mode triangular patch resonator,” in IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2005, pp. 779–782. [11] Y.-K. Yoon, D. Kim, M. G. Allen, J. S. Kenney, and A. T. Hunt, “A reduced intermodulation distortion tunable ferroelectric capacitor architecture and demonstration,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 12, pp. 2568–2576, Dec. 2003. [12] F. A. Miranda, G. Subramanyam, F. W. V. Keuls, R. Romanofsky, J. D. Warner, and C. H. Mueller, “Design and development of ferroelectric - and -band satellite commutunable microwave components for nication systems,” IEEE Trans. Microw. Theory Tech., vol. 48, no. 7, pp. 1181–1189, Jul. 2000. [13] A. Tombak, J. P. Maria, F. T. Ayguavives, Z. Jin, G. T. Stauf, A. I. Kingon, and A. Mortazawi, “Voltage-controlled RF filters employing thin-film barium–strontium–titanate tunable capacitors,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 2, pp. 462–467, Feb. 2003. [14] R. A. York, A. S. Nagra, P. Periaswamy, O. Auciello, S. K. Streiffer, )Ti + O + and J. Im, “Synthesis and characterization of (Ba Sr thin films and integration into microwave varactors and phase shifters,” Integr. Ferroelect., vol. 34, no. 1–4, pp. 1617–1628, 2001. [15] F. D. Flaviis, N. G. Alexopoulos, and O. M. Stafsudd, “Planar microwave integrated phase-shifter design with high purity ferroelectric material,” IEEE Trans. Microw. Theory Tech., vol. 45, no. 6, pp. 963–969, Jun. 1997. [16] A. T. Hunt, W. B. Carter, and J. K. Cochran, Jr., “Combustion chemical vapor deposition: A novel thin-film deposition technique,” Appl. Phys. Lett., vol. 63, no. 2, pp. 266–268, Jul. 1993. [17] ——, “Combustion chemical vapor deposition of films and coatings,” U.S. Patent 5 652 021, Jul. 29, 1997. [18] J. Schmitt, G. G. Cui, H. A. Luten III, F. Yang, F. A. Gladden, S. Flanagan, Y. Jiang, and A. T. Hunt, “Electronic and optical materials,” U.S. Patent 6 986 955, Jan. 17, 2006. [19] J. W. Lu and S. Stemmer, “Low-loss, tunable bismuth zinc niobate films deposited by RF magnetron sputtering,” Appl. Phys. Lett., vol. 83, no. 12, pp. 2411–2413, Sep. 2003. [20] B. Acikel, T. R. Taylor, P. J. Hansen, J. S. Speck, and R. A. York, “A TiO thin films,” new high performance phase shifter using Ba Sr IEEE Microw. Wireless Compon. Lett., vol. 12, no. 7, pp. 237–239, Jul. 2002.

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REFERENCES [1] E. Fourn, A. Pothier, C. Champeaux, P. Tristant, A. Catherinot, P. Blondy, G. Tanne, E. Rius, C. Person, and F. Huret, “MEMS switchable interdigital coplanar filter,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 1, pp. 320–324, Jan. 2003. [2] E. Fourn, C. Quendo, E. Rius, A. Pothier, P. Blondy, C. Champeaux, J. C. Orlianges, A. Catherinot, G. Tanne, C. Person, and F. Huret, “Bandwidth and central frequency control on tunable bandpass filter by using MEMS cantilevers,” in IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2003, pp. 523–526. [3] R. Young, J. Adam, C. Vale, T. Braggins, S. Krishnaswamy, C. Milton, D. Bever, L. Chorosinski, L. Chen, D. Crockett, C. Freidhoff, S. Talisa, E. Cappelle, R. Tranchini, J. Fende, J. Lorthioir, and A. Torres, “Low-loss bandpass RF filter using MEMS capacitance switches to achieve a one octave tuning range and independently variable bandwidth,” in IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2003, pp. 1781–1784. [4] I. C. Hunter and J. D. Rhodes, “Electronically tunable microwave bandpass filters,” IEEE Trans. Microw. Theory Tech., vol. MTT-30, no. 9, pp. 1354–1360, Sep. 1982. [5] A. Brown and G. Rebeiz, “A varactor tune RF filter,” IEEE Trans. Microw. Theory Tech., vol. 48, no. 7, pp. 1157–1160, Jul. 2000. [6] C. Rauscher, “Reconfigurable bandpass filter with a three-to-one switchable passband width,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 2, pp. 573–577, Feb. 2003. [7] B. Lakshminarayanan and T. Weller, “Tunable bandpass filter using distributed MEMS transmission lines,” in IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2003, pp. 1789–1792.

Cesar Lugo (S’01–A’02–M’06) received the B.S. degree and M.S. degree in electrical and computer engineering from the Georgia Institute of Technology, Atlanta, in 2002 and 2003, respectively, and is currently working toward the Ph.D. degree in electrical engineering at the Georgia Institute of Technology. He has developed several synthesis and design techniques for reconfigurable RF/millimeter-wave components such as filters, antennas, couplers, phase shifters, and impedance tuners. He has authored or coauthored over 15 scientific papers in peer-reviewed journals and conferences. His research interests include hybrid semiplanar design of microwave components and adaptive algorithms for electromagnetic simulation.

Guoan Wang (S’05–M’06) received the M.S. degree in semiconductor materials from Zhejiang University, Zhejiang, China, in 1997, the M.S.E. degree in electrical engineering from Arizona State University, Tempe, in 2001, and is currently working toward the Ph.D. degree in electrical engineering at the Georgia Institute of Technology, Atlanta. His doctoral dissertation is entitled “RF MEMS Switches with Novel Materials and Micromachining Technologies for SOC/SOP RF Front-Ends.”

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John Papapolymerou (S’90–M’99–SM’04) received the B.S.E.E. degree from the National Technical University of Athens, Athens, Greece, in 1993, and the M.S.E.E. and Ph.D. degrees from The University of Michigan at Ann Arbor, in 1994 and 1999, respectively. From 1999 to 2001, he was a faculty member with the Department of Electrical and Computer Engineering, University of Arizona, Tucson. During the summers of 2000 and 2003, he was a Visiting Professor with The University of Limoges, Limoges, France. From 2001 to 2005, he was an Assistant Professor with the School of Electrical and Computer Engineering, Georgia Institute of Technology, Atlanta, where he is currently an Associate Professor. He has authored or coauthored over 120 publications in peer-reviewed journals and conferences. His research interests include the implementation of micromachining techniques and microelectromechanical systems (MEMS) devices in microwave, millimeter-wave, and terahertz circuits and the development of both passive and active planar circuits on semiconductor (Si/SiGe, GaAs) and organic substrates [liquid-crystal polymer (LCP), low-temperature co-fired ceramic (LTCC)] for system-on-a-chip (SOC)/system-on-package (SOP) RF front ends. Dr. Papapolymerou currently serves as the vice-chair for Commission D of the U.S. National Committee of URSI and as an associate editor for the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION. During 2004, he was the chair of the IEEE Microwave Theory and Techniques (MTT)/Antennas and Propagation (AP) Atlanta Chapter. He was the recipient of the 2004 Army Research Office (ARO) Young Investigator Award, the 2002 National Science Foundation (NSF) CAREER award, the Best Paper Award presented at the 3rd IEEE International Conference on Microwave and Millimeter-Wave Technology (ICMMT2002), Beijing, China, and the 1997 Outstanding Graduate Student Instructional Assistant Award presented by the American Society for Engineering Education (ASEE), The University of Michigan at Ann Arbor Chapter. His student was also the recipient of the Best Student Paper Award presented at the 2004 IEEE Topical Meeting on Silicon Monolithic Integrated Circuits in RF Systems, Atlanta, GA.

Zhiyong Zhao (S’97–M’00) received the B.S. degree in electrical engineering from Yanshan University, Qinhuangdao, China, in 1991, the M.S. degree in condensed matter physics from Beijing Normal University, Beijing, China, in 1996, and the Ph.D. degree in electrical engineering from the University of South Florida, Tampa, in 2000. In 2000, he joined the nGimat Company, Atlanta, GA, as a Senior Research Scientist, where he has led and participated in numerous aspects of thin-film development using CCVD for applications in microwave sand millimeter waves,

electronics, and optics. His research interests include ferroelectric tunable devices, transparent conductors, and opto-electronics. Dr. Zhao was the recipient of five Small Business Innovation Research Program (SBIR)/Small Business Technology Transfer Program (STTR) Phase I and one SBIR Phase II project awards.

Xiaoyan Wang (S’99–M’03) received the B.S. degree in applied physics from the East China University of Technology, Shanghai, China, in 1991, the M.S. degree in condensed matter physics from Beijing Normal University, Beijing, China, in 1997, and the Ph.D. degree in electrical engineering from the University of Virginia, Blacksburg, in 2003. Since 2004, she has been with the nGimat Company, Atlanta, GA, where she is currently a Senior Research Scientist. Her research interests include integrated chemical sensors, novel photolithographical techniques, and packaging. Dr. Wang was the recipient of two SBIR Phase I awards for the development -band, respectively. of ferroelectric tunable filters at - and

X

Ka

Andrew T. Hunt received the B.S. degree in geology from Auburn University, Auburn, AL, the Masters degree in geology from the Colorado School of Mines, Golden, and the Ph.D. degree in materials science and engineering from the Georgia Institute of Technology, Atlanta. His doctoral thesis concerned CCVD. He founded the nGimat Company, Atlanta, GA, as a “one-man operation,” and has attracted talented professionals to it and has developed commercial relationships with industry-leading companies and research partnerships with key government agencies in nGimat’s target markets. He has authored two book chapters for industry publications on vapor deposition. He has authored or coauthored over 50 scientific papers. He holds over 50 patents, as well as those pending, as a result of his research with the nGimat Company. Dr. Hunt was selected for the National Academy of Engineering (NAE)’s Frontiers of Engineering Symposium in 2002, and the 2005 German–U.S. joint meeting. He currently serves on the National Materials Advisory Board of the National Research Council (NRC) and on several Georgia Institute of Technology external boards. He was the recipient of the Science Applications International Corporation (SAIC) Award for the best Ph.D. dissertation at the Georgia Institute of Technology.

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Modeling and Applications of Ferroelectric-Thick Film Devices With Resistive Electrodes for Linearity Improvement and Tuning-Voltage Reduction Patrick Scheele, Member, IEEE, Andre Giere, Student Member, IEEE, Yuliang Zheng, Felix Goelden, and Rolf Jakoby, Member, IEEE

Abstract—Low-cost planar high- ferroelectric-thick film varactors ( 90) are realized and component architectures using resistive electrodes for dc bias are investigated. A basic model for planar capacitors with resistive electrodes in the gap is developed and verified by finite-difference time-domain simulations and measurements of interdigital capacitors with high-resistivity indium–tin–oxide bias electrodes in the gap. An optimized highcapacitor design based on a series connection of ferroelectric varactors with resistive bias decoupling is presented. The approach allows the increase of device linearity and the reduction of tuning voltages. Based on this technology, a continuously tunable high-power impedance-matching network for 1.875 GHz with tuning voltages below 60 V was developed, realized, and characterized by small- and large-signal measurements with up to 33-dBm input power. The device requires no external dc-block or RF decoupling and features separated RF and dc contacts. The output 3 of up to 47.8 dBm verifies the excellent device linearity.

ages above 60 V are still unacceptably high. A first approach to reduce the tuning voltages of planar gap capacitors by using resistive dc-bias lines in the gaps was described in [7]. In this paper, the implementation of the described component architecture is investigated for low-cost ferroelectric-thick film interdigital capacitors (IDCs) with respect to power-handling capability. Based on this knowledge, an optimized highvaractor design and accompanying model with improved linearity, reduced tuning voltage, and resistive bias decoupling is introduced. Using this technology, a continuously tunable “T”matching network with two varactors, which does not require external components for bias decoupling, has been realized and characterized by small- and large-signal measurements.

Index Terms—Ferroelectric capacitors, ferroelectric devices, impedance matching, linearization, passive circuits, thick film devices, tunable circuits and devices.

The following devices, test-structures, and circuits have all m layer of been realized on and simulated for a Ba Sr TiO , which was screen printed and sintered on top of a m Al O ceramic substrate . This substrate was prepared at the Institute for Materials Research III, Karlsruhe Research Center, Karlsruhe, Germany. The BST layer at 2 GHz and room exhibits a relative permittivity temperature with a loss tangent . The permittivity of this layer can be tuned by application of an external electric field. Metallic contacts are obtained through a two-step lithography process after thermal evaporation of a chrome/gold seed layer. In the first lithography step, structures for the high-resistivity indium–tin–oxide (ITO) electrodes are formed. This is achieved by local etching of the starting metallization and E-beam deposition of a 30-nm indium-oxide/tin-oxide layer, followed by a liftoff step. The ITO is formed by oxidization in air for 1 h at 300 C. After this thermal treatment, the ITO structures turn opk sq . tically transparent with a resistance In the second lithography step, mask structures for gold plating are aligned to the transparent ITO structures. Contacts between ITO and plated gold were found to have ohmic behavior. Device fabrication is completed after removing the photoresist and etching of the starting metallization.

I. INTRODUCTION

L

INEARITY is one of the key issues in the implementation of tunability in communication circuits. Applications in “frequency-agile” radio frontends like tunable multiband filters, tunable duplexers, or impedance-matching networks do not only require low power consumption, high capacitance tuning, and low tuning voltages, but typically also high power-handling capability. Up to now, neither semiconductor varactors, nor RF microelectromechanical systems (MEMS) or other technologies seem to fulfill all of these requirements, especially when it comes to cost-effective device production. Low-cost planar ferroelectric-thick film components have shown sufficiently high tunability and linearity for many applications, but required very high tuning voltages and showed inferior performance in terms of component factors related to substrate losses [4]. Recent progress in material development now allows the realization of high- varactors, but tuning voltManuscript received May 24, 2006; revised September 18, 2006. This work was supported by the German Federal Ministry of Education and Research under Grant 01BU570 MARIO (Multi Access System in Package Radio). The authors are with the Laboratory of Wireless Communications, Darmstadt University of Technology, 64283 Darmstadt, Germany (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TMTT.2006.889351

II. SUBSTRATE PROPERTIES AND PROCESSING

III. PLANAR CAPACITOR ARCHITECTURES A. Capacitor With Resistive Electrodes in the Gap The concept of a planar IDC with a resistive electrode in the gap is shown in Fig. 1. The RF electrodes with conductivity , thickness , width , spacing , and length form the fingers

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Fig. 1. IDC with resistive electrodes in the gap.

of a standard IDC. At least one additional resistive electrode with conductivity is placed in the gap, as proposed by [7]. A tuning voltage applied between this additional electrode and the RF electrodes will lead to a higher electric field strength in the gap and is, therefore, expected to lead to higher dielectric tuning of the BST layer compared to the application of the same voltage between the RF electrodes only. As the electric field strength in across the capacitor is the BST layer due to an RF voltage , and the electric field strength due to a mainly related to dc-tuning voltage is related to , RF linearity can be improved while dc tuning voltages may be reduced at the same time. 1) Analysis, Simulation, and Modeling: For high power-handling capability, not only linearity, but also component losses are of great interest. Typical calculation methods for IDCs based on quasi-static/conformal-mapping approaches are not suitable for the calculation of the additional losses due to a conductor in the gap so a numeric solution based on the FDTD method [3] was applied. The problem was split into 2-D longitudinal homogeneous quasi-TEM sections with a very fine mesh in the high-permittivity dielectric material and in the resistive metal for high accuracy. The capacitance of the sections between two inner fingers was determined by assuming a magnetic wall in the middle of these fingers. In the first approach, metallic losses and the caof the RF electrodes have been omitted pacitance and factor of the capacitor shown in Fig. 1 have been calculated for different frequencies and a given geometry versus conductivity . The results are shown in Fig. 2 with the geometry data used for simulation given in the figure’s caption. Two conditions in Fig. 2(a) can be easily derived: for low con, the gap metal disappears and the IDC can ductivities be calculated with quasi-static methods for a gapwidth . For , the effective gapwidth narrows high conductivities to , leading to a higher capacitance. The conductivity of ITO, which is used as a resistive metal, is typically better than 0.5 10 S/m with a maximum of 10 S/m [8]. In this conductivity range, the quality ( ) factor in Fig. 2(b) is significantly

Fig. 2. Simulated and modeled properties without RF conductor losses. (w = 8 m, g = 16 m, t = 2 m, w = 6 m, t = 20 nm, l = 136 m,  = 3:32). (a) Capacitance versus gap metal conductivity. (b) Q factor versus

gap metal conductivity.

Fig. 3. Equivalent circuit for an IDC with resistive electrodes in the gap.

influenced by the conductor in the gap, dropping from 110 to values below 20. Based on these results, a simple equivalent circuit can be dewith factor in Fig. 3 is a stanrived. The capacitance , dard -finger IDC without electrodes in the gap. For the capacitance with factor is connected in parallel, leading to an increased total capacitance due to the reduction of the effective gapwidth when a conductor is placed in it. The recan be calculated in dependency of the gap metal sistance geometry and conductivity, as given by (1), as follows with an additional factor , which is dependent on the overall device geand : ometry, i.e., all layer permittivities and heights, (1) , which was extracted The modeled response for using finite-difference time-domain (FDTD) data, is shown in

SCHEELE et al.: MODELING AND APPLICATIONS OF FERROELECTRIC-THICK FILM DEVICES WITH RESISTIVE ELECTRODES

Fig. 4. Microscopic images of fabricated devices. A polarizing filter was used to enhance visibility of the transparent ITO strips. (a) Reference IDC C with coplanar ground planes for on-wafer characterization (image size 240 252 m ). (b) IDC C with an additional 30-nm-high and 9.7-m-wide strip of ITO in the 22.6-m gap.

2

Fig. 2. The resulting response shows excellent agreement with the FDTD simulation data and proves the principle validity of the model. 2) Experimental Results and Verification: To verify the simulated and modeled capacitor properties, several IDCs have been fabricated and characterized up to 30 GHz by on-wafer -parameter measurements. For comparison, a reference IDC with gold S/m, m and a typical finger electrodes m, finger spacing m, and an overlapwidth ping finger length m was chosen. Two 50 50 m contact pads were added at each end of the IDC, which was placed between coplanar grounds for calibrated two-port measurements [see Fig. 4(a)]. Fig. 4(b) shows an IDC of the same nm length and comparable gapwidth with an additional m). The ITO is connected ITO strip inside the gap ( to bias pads outside the capacitor structure, which are contacted separately. An evaluation frequency of 1.5 GHz was chosen, which is far away from the parasitic series resonance of the structures around 17 GHz. The comparison of the measured capacitor data with FDTD in the gap in depensimulations for different ITO widths dency of conductivity in Fig. 5 shows good agreement of sim-

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Fig. 5. Comparison of FDTD simulation for different conductor widths and measurements with and without a t = 30-nm-high and w = 9:7-m-wide ITO strip in the gap at 1.5 GHz including RF conductor losses (l = 196 m). (a) Capacitance versus gap metal conductivity for different conductor widths. (b) Q factor versus gap metal conductivity for different conductor widths.

ulation and measurement. Due to the low frequency and thin RF conductors, additional metallic losses had to be accepted, which have also been taken into account in the simulation. The effective conductivity of the ITO film calculated from a dc-resistivity measurement amounts to 170 S/m. The value is lower than expected, which could be a result of conductor thinning due to the substrate roughness with an average grain size of 300 nm. The simulations show a diminishing influence of the resistive metal for very narrow widths , while the absolute minimum shifts to lower conductivities. In terms of a filling factor , only low resistive filling of the gaps leads to acceptable factors. The observation is confirmed by measurements of capacitors with other geometries, listed in Table I. However, all measured capacitors show a significantly reduced factor compared to the . reference capacitor Referring to the initial idea of a tuning voltage reduction, the normalized capacitance versus tuning voltage has been investigated. For comparison, three standard IDCs without ITO and , 7.3 m , and different gapwidths of 20.1 m have been characterized for tuning voltages up 5.2 m to 100 V. While the tuning voltage of a reference capacitor is applied between its fingers, the bias voltage can be applied to the capacitor shown in Fig. 4(b) either between the ITO strip and both capacitor fingers (dc-grounded unipolar bias condition) or between the capacitor fingers with a floating or grounded ITO

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TABLE I MEASURED CAPACITANCES AND Q FACTORS AT 1.5 GHz IN DEPENDENCY OF ITO WIDTH w AND GAPWIDTH g . THE Q-FACTOR REDUCTION CORRELATES WITH THE FILLING OF THE GAP, AS INDICATED BY THE RATIO w =g . COMPARED TO THE REFERENCE CAPACITOR WITHOUT ITO IN THE GAP (w = 0), ALL Q FACTORS ARE SIGNIFICANTLY REDUCED

Fig. 7. Series capacitors with resistive bias decoupling. (a) Component layout. (b) Equivalent circuit.

factor will be significantly reduced and the tuning efficiency is low. B. Capacitor With Resistive Bias Decoupling

Fig. 6. Normalized capacitance versus tuning voltage of three reference capacitors without ITO and different gapwidths and one capacitor with a 9.2-m ITO strip in a 20.9-m gap (C ). In differential bias condition, tuning voltage is applied between the capacitor’s fingers with identical results for floating or grounded ITO bias contact. In unipolar bias condition, the capacitor’s RF contacts are dc grounded and the tuning voltage is applied via the ITO contact to ground.

strip (differential bias condition). Fig. 6 shows the normalized (see Table I) in comparison to the different capacitance of reference IDCs. For differential bias, the tuning of 9% at 100 V is comparable to the reference capacitor with similar gapwidth . Besides the large difference in factors mentioned above, the influence of the ITO strip on the tuning behavior is negligible in this case. In a unipolar bias condition, the gap between the ITO and the capacitor fingers amounts to 5.9 m and the maximum tuning with 5.2- m is doubled to 18% at 100 V. Compared to gaps and 22% tuning at 60 V, the value is much lower than initially expected. The effect can be explained with the different field distribution of the RF electric field between the fingers and the dc electric field for bias between the ITO strip and the fingers. Due to the high permittivity of the BST layer and the nonlinear dependency of permittivity versus electric field strength, tuning efficiency is low as the RF field penetrates much deeper into the BST and Al O than the dc-tuning field [3]. In summary, the influence of an ITO strip in the gap on tunability is much lower than expected and the reduction of the factor is much higher than desired if not very narrow ITO strips can be fabricated ( 1 m). That however requires a high technological effort including very precise alignment and lithographic capabilities, as well as substrates with very smooth surfaces. For less sophisticated (lower cost) processing, an increase of gapwidth for RF linearization combined with an increase in the number of ITO fingers in the gap for tuning voltage reduction is contradictory to power-handling capability as the component

Based on the knowledge about low tuning efficiency and a reduced component factor due to any resistive conductor in an IDC gap, an optimized component concept was developed using a resistive strip of ITO for bias decoupling. The concept, which is based on the series connection of two standard IDCs is shown in Fig. 7. The principle is known from antiseries connections of semiconductor varactors and can also be applied for ferroelectric varactors. 1) Analysis, Simulation, and Modeling: For equal capaciin Fig. 7(b), an RF voltage across the tances device, i.e., between nodes 1 and 2, will be divided while a between node 3 and nodes 1 and 2 is used for dc voltage due to tuning (unipolar bias). A small capacitance change in one of the varacRF large-signal operation tors will then be accompanied by a capacitance change of in the other varactor due to opposite signs of tuning voltage and , leading to additional linearization RF voltage at low RF voltage levels. is realized with an ITO strip of length The resistor and width . The device has three ports and can, therefore, not be compared directly with a capacitor. However, bias port 3 will be typically RF grounded, and a practical “worst case” scenario was chosen with ports 2 and 3 connected to ground. The input impedance at port 1 is then treated as a capacitance to ground and the influence of conductivity versus the factor is investigated. The resulting capacitance and factor for a series connection and a 30-nm resistive of two 0.77-pF capacitors with m and a length of m strip with a width of disappears for low conducis given in Fig. 8. The resistor tivities S/m and the device behaves like a 0.385-pF . is short circuited for very large convaractor with . Compared to the behavior of a resistive ductivities filling in the IDC gap in Fig. 2, the -factor reduction is found at much higher conductivities. Furthermore, the series resistance can be scaled with , offering an additional degree of freedom. 2) Experimental Results and Verification: For verification of the previous simulation, the capacitor structures shown in Fig. 9 have been fabricated and characterized. Again, the results are compared to reference capacitors, as shown in Fig. 4(a). factors at Table II lists the measured capacitances and factor of both capacitors and is 1.5 GHz. The slightly reduced in comparison to , which can be traced

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TABLE II MEASURED CAPACITANCES AND Q FACTORS AT 1.5 GHZ OF TWO DIFFERENT SERIES CAPACITORS WITH ITO BIAS DECOUPLING, AS SHOWN IN FIG. 9. THE ITO LINES OF C ARE CONNECTED TO THE COPLANAR GROUND PLANE ON BOTH SIDES [SEE FIG. 9(a)], WHILE THE OTHER IS CONNECTED TO A SEPARATE BIAS PAD [C , FIG. 9(b)]

Fig. 10. Normalized capacitance versus tuning voltage of three reference capacitors without ITO and different gapwidths and series capacitor C (see Table II). In differential bias condition, tuning voltage is applied between the capacitor’s RF contacts. In unipolar bias condition, the tuning voltage is applied via both capacitor RF contacts to ground.

Fig. 8. Simulated properties of two series capacitors as shown in Fig. 7(b) (ports 2 and 3 connected) with C = 0:77 pF, Q = 110, and R calculated for t = 30 nm, w = 10 m, and l = 150 m versus conductivity  (without RF-conductor losses). (a) Capacitance versus resistive metal conductivity. (b) Q factor versus resistive metal conductivity.



Fig. 9. Microscopic images of fabricated devices with g 10 m gaps. A polarizing filter was used to enhance visibility of the transparent ITO strips (t = 30 nm). (a) Series-IDC C with 2 5 fingers and ITO strips to coplanar ground. (b) Series-IDC C with 2 11 fingers and an ITO strip to a separate bias pad.

2

2

back to a decreased parasitic series resonance at 15 GHz and increased metallic losses due to an increased overall device . Device tunabilities in Fig. 10 were measured length for with tuning voltages up to 100 V. For differential bias, the tuning is only 6.2% with 100 V across the 2 10.2 m gaps, which is lower than the 9.8% tuning of with 20.1- m

gaps ( shows 6% tuning at 70 V). In a unipolar bias condition, the capacitance reduction amounts to 16.3% at 100 V. This value is only slightly lower than the tunability of with 5.9- m gaps in Fig. 6. Summing up the experimental results, the capacitor architecture with resistive bias decoupling shows comparable tunability and higher factors with less sophisticated technological effort (low precision mask alignment, wider conductors, and gaps) in comparison to IDCs with resistive metal in the gaps. Tuning efficiency in the unipolar bias condition is high, as the electric dc-tuning field and the RF-field are superimposed in the capacitors’ gaps. The tunability in the differential bias condition is lower, which gives reason that linearity under large-signal RF operation should be higher. Scaling of linearity and tuning voltage, however, is limited to the aspect ratio of the RF conductors and to cascading of several stages of series capacitors. The latter will have significant influence on the factor due to a reduction of the device’s series resonance frequency. The esin the range timated switching time with series resistances 10 and typical capacitances in the range of of 10 10 F is below 10 s. 10 IV. TUNABLE HIGH-POWER MATCHING NETWORK WITH SEPARATED RF AND DC CONNECTIONS The capacitor architecture with resistive bias decoupling has been used for the design and realization of a continuously tunable impedance-matching network at 1.875 GHz. A T-configuration with two ferroelectric series varactors pF and one discrete surface-mount inductor nH

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Fig. 11. Microscopic image of a matching network varactor pair with coplanar ground planes for on-wafer characterization (picture size 500 660 m ). A polarizing filter was used to enhance visibility of the transparent ITO strips.

2

(Coilcraft 0302CS, at 1.7 GHz) to ground was chosen for circuit realization. This configuration allows a large tuning range with small capacitor values, which can be realized easily in planar technology. Fig. 11 shows a matching network varactor pair with coplanar grounds used for on-wafer characterization. Each varactor has a separate 100 100 m bias pad connected - m-wide and - m-long ITO strip via a nm) to the central IDC fingers. The IDCs have a finger ( m, a gapwidth of m, and an width of overlapping length of m. The thickness of the plated m because of the narrow gold had to be reduced to gaps and the high substrate roughness. Therefore, varactor factors are limited to 56 due to additional conductor losses. Varactor tuning amounts to 15.8% at a bias voltage of 60 V. The upper ground plane with a size of 500 250 m is conand with a nected to the center between the varactors 28 100 m strip of gold. It serves as the contact pad for the surface-mount inductor in the final circuit (see below). The pad is connected via two additional 112- m ITO strips to the RF ports of the network. As the inductor will be connected to ground in the completed circuit, both RF ports are dc-grounded via those resistive lines. Tuning voltages may then be applied via the bias pads to RF ground. As the varactors themselves are used as dc blocks, no additional external components for RF or dc blocking/decoupling for the two independent bias voltages and the two RF ports are needed. A Rogers Corporation RO3006 microwave substrate ( m) is used as a carrier for the complete circuit. A diced varactor pair (without the lower coplanar ground plane) was mounted into a 1.4-mm-diameter hole in the substrate and connected to the 50- RF feed lines and dc-bias lines with dispensed conductive epoxy adhesive, as shown in Fig. 12. Two vias in the carrier substrate connect a top ground pad to microstrip ground. The SMD 0302 inductor is glued on top of the contact pad between the varactor pair and to the top ground pad on the carrier substrate. Fig. 13 shows the completed circuit

Fig. 12. Diced varactor pair mounted on the carrier substrate. The 7.2-nH SMD 0302 inductor placed on top is connected to RF ground, completing the tunable matching network. 50- microstrip lines are connected to the left and right, bias voltages are fed via two separate lines at the bottom. Conductive epoxy adhesive was used for all electrical connections.

Fig. 13. Completed continuously tunable matching network with 50-

SMA connectors and separated control voltage connections. The total size is 30 9 8 mm .

2 2

with the carrier substrate mounted between standard subminiature A (SMA) connectors. The circuit has been characterized using small-signal network and . The meaanalysis in dependency of bias voltages at sured input reflection and transmission for two bias voltage pairs is shown in Fig. 14. The total insertion loss for 50- matching V at ) amounts to 1.2 dB, inat both ports (0 V at cluding the SMA connectors and feed lines. Input reflection is then below 28 dB. From the small-signal measurement, the load impedances for matching at the input of the network have been calculated. It has to be noted that conjugate complex matching in general is not possible on both ports of a lossy matching network [4]. Therefore, conjugate complex matching towards a fixed source at the input of the network in depenimpedance at the output of the network dency of the load impedance was chosen as the matching criterion. This resembles the requirements on a matching network in a mobile handset placed between the power amplifier and an antenna, which changes its

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Fig. 14. Measured transmission and input reflection of the matching network for two different tuning voltage combinations and 50- source and load impedance. The total insertion loss at 1.875 GHz including SMA connectors 28 dB. amounts to 1.2 dB (0 V 60 V) with

=

s 100

Fig. 8. Fabricated 20-GHz phase shifter (

phase shift).

Three stages of such phase shifters were cascaded to realize a 300 phase shift. These phase shifters used a CPW as input and output, facilitating probing for measurements. The dimensions of triply cascaded 20- and 30-GHz phase shifters were 0.95 1.32 0.5 mm and 1 1.2 0.5 mm , respectively. V. MEASURED RESULTS AND DISCUSSION -parameters were measured using an HP8510C vector network analyzer and ground–signal–ground (G–S-G) probes. Offwafer short-open-load-thru (SOLT) calibration was performed setting the reference planes at the probe tips. Phase shifters with both BST gap capacitors (or IDCs) and low-voltage capacitors

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Fig. 9. Measured S -parameters and phase shift of a 20-GHz phase shifter using IDCs (SOLT calibration).

Fig. 10. Measured S -parameters and phase shift of a 30-GHz phase shifter using IDCs (SOLT calibration).

were fabricated. For the ones with gap capacitors, dc biasing was applied through the signal (S) and ground (G) probes, while for the low-voltage phase shifters, there were separate bias pads for dc probing, and both G and S were dc grounded. In both cases, only a single dc bias was required. Fig. 9 shows the measured results of a 20-GHz phase shifter using regular IDCs. The phase shift was 315 at 20 GHz, and

the maximum phase shift was 330 occurred at 21.7 GHz. At 21.7 GHz, the insertion loss was 6.1 dB at 0 V, and reduced to 4 dB at a bias of 20–30 V. The return loss was better than 10 dB for biases 30 V, and reached below 4 dB at a bias of 60 V. Fig. 10 shows the measured results of a 30-GHz phase shifter using regular IDCs. The phase shift was 352 at 30 GHz, and the maximum phase shift was 360 at 32 GHz. At 32 GHz, the

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Fig. 12. Phase shifts at 19–22 GHz of the 20-GHz phase shifter shown in Fig. 11 under different bias voltages.

Fig. 11. Measured S -parameters and phase shift of a 20-GHz phase shifter using low-voltage BST capacitors.

insertion loss was 7.0 dB at 0 V, and reduced to 5 dB under a bias from 10 to 40 V. The return loss was better than 10 dB for biases 30 V, and reached below 4 dB at a bias of 60 V. As can be seen, the measured results agree very well with the simulated ones, indicating that the dielectric constant of the BST used in this study is in the range of 500–600, and the tunability of the BST capacitors is 4 : 1 at a bias of 60 V. Small

deviations are attributed to fabrication tolerances. In the above phase shifters, BST IDCs were used. To reduce bias voltage, the low-voltage phase shifters were designed. Fig. 11 shows the measured results of a 20-GHz low-voltage phase shifter. It is seen that the responses are very similar to those in Fig. 9, but the bias voltage has been reduced from 60 to 30 V. These results bode well with the voltage required for the same tunability of the two capacitor structures (Fig. 4). The capacitors were measured at 1 MHz using an LCR meter, and results showed that their capacitances reduced to 1/4 when biased to 60 V (gap or IDCs) or 30 V (low-voltage capacitors). Even lower control voltages can readily be achieved by reducing the spacing width. Fig. 12 shows the phase shifts of the 20-GHz low-voltage phase shifter shown in Fig. 11 at 19–22 GHz under various bias states. The largest phase-shift variation occurs at 10 V for all four frequencies, which is 30 . For any two adjacent frequencies, the phase-shift variation is within 15 . These variations can be further minimized by designing the cascaded phase shifters, each with a slightly different center frequency. To identify the sources of insertion loss, HFSS simulations have been performed assuming that some or all materials (e.g., metal, BST, sapphire, and BCB) were lossy. Fig. 13 shows simulated insertion loss and return loss of a 30-GHz phase shifter under four conditions, which include that: 1) all materials are lossless; 2) the metal, BCB, and substrate are lossy; 3) all materials are lossy and the BST loss tangent is 0.05; and 4) all materials are lossy and the BST loss tangent is 0.1. Losses contributing to metals and dielectrics (excluding BST) account for approximately 1 dB of insertion loss, and lossy BST adds an) or 3 dB (when ). other 1.5 dB (when Using a CPW test structure [3], we have measured the typical of BST films, which was in the range of 0.05–0.1 at zero bias; the value decreased with increasing bias. Comparing curve (4) in Fig. 13 with the insertion loss at 0 V in Fig. 10, the measured phase shifter exhibits an additional 2 dB of insertion loss. This can be attributed to SOLT calibration, nonideal metal and dielectric (excluding BST) losses, metal/BST interface, and fabrication tolerances. The poor return loss at higher bias voltages is due to the mismatch of inductances and capacitances, which

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REFERENCES

Fig. 13. HFSS simulated results of a 30-GHz phase shifter. (a) Insertion loss. (b) Return loss.

can be improved through design optimization. As this is done, the insertion loss at biased states will also be improved. Two-tone IMD tests have been performed on the BST tunable ) of 30 dBm capacitors, and a third-order intercept power ( was achieved. Since the BST capacitors are the components values are typical for giving nonlinear performance, these phase shifters as well. VI. CONCLUSION This paper demonstrates continuously variable 360 capable ferroelectric phase shifters at 20 and 30 GHz, which were only slightly over 1 mm in size. The fabricated phase shifters had an insertion loss of 6 dB at 0 V, and reduced to 4 dB when biased. The figure-of-merit was better than 50 /dB. These phase shifters are designed with a solder ball attachment so they can be flip-chip mounted onto a printed circuit board for practical applications or can be packaged to be compatible with the end user’s requirements. Due to the small size and little power consumption, these phase shifters are well suited for phased-array applications. ACKNOWLEDGMENT The authors would like to thank Prof. J. S. Kenney and Prof. J. Papapolymerou, both with the Georgia Institute of Technology, Atlanta, for valuable discussions regarding circuit optimization.

[1] A. T. Hunt, W. B. Carter, and J. K. Cochran, Jr., “Combustion chemical vapor deposition: A novel thin-film deposition technique,” Appl. Phys. Lett., vol. 63, no. 2, pp. 266–268, Jul. 1993. [2] J. Schmitt, G. G. Cui, H. A. Luten III, F. Yang, F. A. Gladden, S. Flanagan, Y. Jiang, and A. T. Hunt, “Electronic and optical materials,” U.S. Patent 6 986 955, Jan. 17, 2006. [3] Z. Zhao, Y. Jiang, X. Wang, K. Choi, and A. T. Hunt, “Epitaxial growth of ferroelectric thin films by combustion chemical vapor deposition and their electrical properties,” presented at the 15th IEEE Int.. Applicat. Ferroelect. Symp., Sunset Beach, NC, Jul. 2, 2006, accepted for publication. [4] Y.-K. Yoon, D. Kim, M. G. Allen, J. S. Kenney, and A. T. Hunt, “A reduced intermodulation distortion tunable ferroelectric capacitor-architecture and demonstration,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 12, pp. 2568–2576, Dec. 2003. [5] A. Kozyrev, A. Ivanov, V. Keis, M. Khazov, V. Osadchy, T. Samoilova, O. Soldatenkov, A. Pavlov, G. Koepf, C. Mueller, D. Galt, and T. Rivkin, “Ferroelectric films: Nonlinear properties and applications in microwave devices,” in Proc. IEEE MTT-S Int. Microw. Symp. Dig., 1998, pp. 985–988. [6] D. Kim, Y. Choi, M. G. Allen, J. S. Kenney, and D. Kiesling, “A wideband reflection-type phase shifter at S -band using BST coated substrate,” IEEE Trans. Microw. Theory Tech., vol. 50, no. 12, pp. 2903–2909, Dec. 2002. [7] V. Sherman, K. Astafiev, N. Setter, A. Tagantsev, O. Vendik, I. Vendik, S. Hoffmann, U. Böttger, and R. Waser, “Digital reflection-type phase shifter based on a ferroelectric planar capacitor,” IEEE Microw. Wireless Compon. Lett., vol. 11, no. 10, pp. 407–409, Oct. 2001. [8] Y. Liu, A. S. Nagra, E. G. Erker, P. Periaswamy, T. R. Taylor, J. Speck, and R. A. York, “BaSrTiO interdigitated capacitors for distributed phase shifter applications,” IEEE Microw. Guided Wave Lett., vol. 11, pp. 448–450, Nov. 2000. [9] B. Acikel, T. R. Taylor, P. J. Hansen, J. S. Speck, and R. A. York, “A TiO thin films,” new high performance phase shifter using Ba Sr IEEE Microw. Wireless Compon. Lett., vol. 12, no. 7, pp. 237–239, Jul. 2002. [10] D. Kim, Y. Choi, M. Ahn, M. G. Allen, J. S. Kenney, and P. Marry, “2.4 GHz continuously variable ferroelectric phase shifters using all-pass networks,” IEEE Microw. Wireless Compon. Lett., vol. 13, no. 10, pp. 434–436, Oct. 2003. [11] F. D. Flaviis, N. G. Alexopoulos, and O. M. Stafsudd, “Planar microwave integrated phase-shifter design with high purity ferroelectric material,” IEEE Trans. Microw. Theory Tech., vol. 45, no. 6, pp. 963–969, Jun. 1997. [12] F. W. V. Keuls, C. H. Mueller, F. A. Miranda, R. R. Romanofsky, C. L. Canedy, S. Aggarwal, T. Venkatesan, R. Ramesh, J. S. Horwitz, W. TiO Chang, and W. J. Kim, “Room temperature thin film Ba Sr Ku-band coupled microstrip phase shifters: Effects of film thickness, doping, annealing and substrate choice,” in IEEE MTT-S Int. Microw. Symp. Dig., 1999, pp. 737–740. [13] J. S. Kenney, Y. K. Yoon, M. Ahn, M. G. Allen, Z. Zhao, X. Wang, A. Hunt, and D. Kim, “Low-voltage ferroelectric phase shifters from L- to C -band and their applications,” in IEEE Aerosp. Conf., Big Sky, MT, 2006, pp. 1–9. [14] A. T. Hunt, W. B. Carter, and J. K. Cochran Jr, “Combustion chemical vapor deposition of films and coatings,” U.S. Patent 5 652 021, Jul. 29, 1997.

Zhiyong Zhao (S’97–M’00) received the B.S. degree in electrical engineering from Yanshan University, Qinhuangdao, China, in 1991, the M.S. degree in condensed matter physics from Beijing Normal University, Beijing, China, in 1996, and the Ph.D. degree in electrical engineering from the University of South Florida, Tampa, in 2000. In 2000, he joined the nGimat Company, Atlanta, GA, as a Senior Research Scientist, where he has led and participated in numerous aspects of thin-film development using CCVD for applications in microwaves and millimeter waves, electronics, and optics. His research interests include ferroelectric tunable devices, transparent conductors, and opto-electronics. Dr. Zhao was the recipient of five Small Business Innovation Research Program (SBIR)/Small Business Technology Transfer Program (STTR) Phase I and one SBIR Phase II project awards.

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Xiaoyan Wang (S’99–M’03) received the B.S. degree in applied physics from the East China University of Technology, Shanghai, China, in 1991, the M.S. degree in condensed matter physics from Beijing Normal University, Beijing, China, in 1997, and the Ph.D. degree in electrical engineering from the University of Virginia, Charlottesville, in 2003. Since 2004, she has been with the nGimat Company, Atlanta, GA, where she is currently a Senior Research Scientist. Her research interests include integrated chemical sensors, novel photolithographical techniques, and packaging. Dr. Wang was the recipient of two SBIR Phase I awards for the development of ferroelectric tunable filters at - and -band, respectively.

Cesar Lugo (S’01–A’02–M’06) received the B.S. degree and M.S. degree in electrical and computer engineering from the Georgia Institute of Technology, Atlanta, in 2002 and 2003, respectively, and is currently working toward the Ph.D. degree in electrical engineering at the Georgia Institute of Technology. He has developed several synthesis and design techniques for reconfigurable RF/millimeter-wave components such as filters, antennas, couplers, phase shifters, and impedance tuners. He has authored or coauthored over 15 scientific papers in peer-reviewed journals and conferences. His research interests also include hybrid semiplanar design of microwave components and adaptive algorithms for electromagnetic simulation.

Kwang Choi (S’02) received the B.S., M.S., and Ph.D. degrees in electrical engineering from the Georgia Institute of Technology, Atlanta, in 1995, 1996, and 1999, respectively. His dissertation concerned modeling and simulation techniques for RF embedded passive components in multilayered substrates. He has contributed to nine technical papers on modeling of embedded passives. In January 2000, he joined EMS Technologies, as an RF Engineer, where much of his work was focused on electromagnetic (EM) modeling and simulation. In July 2005, he joined the nGimat Company, Atlanta, GA, as an RF Engineer. His research interests include design, simulation, and testing of tunable RF microwave devices.

Andrew T. Hunt received the B.S. degree in geology from Auburn University, Auburn, AL, the Master degree in geology from the Colorado School of Mines, Golden, and the Ph.D. degree in materials science and engineering from the Georgia Institute of Technology, Atlanta. His doctoral thesis concerned CCVD. He founded the nGimat Company, Atlanta, GA, as a “one-man operation,” and has attracted talented professionals to it and has developed commercial relationships with industry-leading companies and research partnerships with key government agencies in nGimat’s target markets. He has authored two book chapters for industry publications on vapor deposition. He has authored or coauthored over 50 scientific papers. He holds over 50 patents, as well as those pending, as a result of his research with the nGimat Company. Dr. Hunt was selected for the National Academy of Engineering (NAE)’s Frontiers of Engineering Symposium in 2002, and the 2005 German–U.S. joint meeting. He currently serves on the National Materials Advisory Board of the National Research Council (NRC) and on several Georgia Institute of Technology external boards. He was the recipient of the Science Applications International Corporation (SAIC) Award for the best paper at the Georgia Institute of Technology.

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Ka

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A 360 BST Phase Shifter With Moderate Bias Voltage at 30 GHz Gabriel Vélu, Karine Blary, Ludovic Burgnies, Aurélien Marteau, Grégory Houzet, Didier Lippens, and Jean-Claude Carru

Abstract—Paraelectric BaSrTiO3 films deposited by sol-gel have been investigated from 1 kHz to 40 GHz using coplanar waveguides, resonators, and inter-digitated capacitances. At room temperature and without bias, the dielectric constant is around 300 and the loss tangent is 5 10 2 . Tunability is 40% for a 40-V bias voltage. Analog phase-shifter circuits were subsequently fab-band with ricated in order to obtain a 360 phase shift in the a moderate bias voltage. The lowest bias value was 17 V at 40 GHz, whereas 40 V were necessary at 30 GHz. For the latter condition, the insertion loss was 6 dB. The best figure-of-merit of the phase shifters with a loading factor of 1. 4 was 27 /dB. These results show an improvement in terms of voltage-controlled tunability. Index Terms—Analog phase shifters, dielectric characterization, ferroelectric thin films, interdigitated capacitors, tunability.

I. INTRODUCTION ERROELECTRIC thin-film materials are good candidates for voltage-controlled applications at microwave and millimeter waves. The variations of their complex permittivity (real and imaginary parts) with an applied dc field is used to tune the operating frequency of passive components such as filters, delay lines, and phase shifters. One of the most mature ferroelectric material is barium titanate doped with strontium (BST). It combines high dielectric constant, moderate loss tangent, and significant tunability. In the current study, we address the dielectric properties of Ba Sr TiO . This material compound is in the paraelectric state at room temperature with better performance as we recently showed [1] by comparing paraelectric and ferroelectricbased phase shifters. BST films (0.3- m thick) have been deposited by the sol-gel technique [2]. The main interest of this method is a good control of the composition all over the substrate area along its simplicity. In addition, we choose to de-

F

Manuscript received April 25, 2006; revised October 13, 2006. This work was supported in part by the Region Nord-Pas de Calais under the framework of Contract CPER/TACT13 between the Region and the French State. G. Vélu, L. Burgnies, and J.-C. Carru are with the Laboratory of Materials and Components for Electronics, Université du Littoral-Côte d’Opale, 62228 Calais, France (e-mail: [email protected]). K. Blary, A. Marteau, and D. Lippens are with the Institut d’Electonique, de Microélectronique et de Nanotechnologie–Department of High Frequency and Semiconductor, Groupe Quantom Opto and Micro Electronics Devices (DOME), Université de Lille 1, 59652 Villeneuve d’Ascq, France (e-mail: [email protected]). G. Houzet was with the Laboratory of Materials and Components for Electronics, Université du Littoral-Côte d’Opale, 62228 Calais, France. He is now with the Institut d’Electronique de Microélectronique et de Nanotechnologie, Université des Sciences et Technologies de Lille, 59652 Lille, France Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TMTT.2006.889319

velop an inter-digitated capacitance (IDC) technology avoiding several masking stages. In counterpart and to preserve a moderate tuning voltage, electron beam writing was used for fabrication. From the application side, the key objective was to realize a 360 phase shift at millimeter wavelengths with a few tens of dc-bias voltage. Depending on the frequency of operation, the tuning voltage ranges from 17 up to 40 V. To our knowledge, this is a significant improvement while maintaining a high level of performance with respect to state-of-the-art. The losses at millimeter waves for unbiased devices is still a problem with a main contribution stemming from a mismatch at the access regions. However, the overall losses (dielectric and return losses ) are reduced significantly under bias with 6 dB at 40 V and 30 GHz due to a better matching of loaded transmission lines and a decrease of loss tangent. Another originality of this paper is the wide characterization frequency range up to 40 GHz for the phase shifter prototype and for unloaded transmission line integrating thin BST film. This permits us to assess the frequency dependence of the relative permittivity of BST with no relaxation effects pointed out in - and -bands paving the way to use this technology for key millimeter applications notably at 60 GHz. This paper is organized as follows. In Section II, we outline the material deposition method and the microwave characterizations of BST films in a wide frequency range. Section III deals with the optimization of a phase shifter circuit notably in terms of the loading factor and of the dimensions of interdigitated capacitors. Characterization of phase shifters is reported in Section IV with special attention on the differential phase shift and insertion losses up to 40 GHz. Section V is devoted to a transmission line modeling of the phase shifter, assuming lossless condition and full-wave numerical simulations by means of a finite-element method. II. MATERIALS AND CHARACTERIZATIONS We used BST (50/50), as this material presents moderate losses at microwaves and also good tunability. BST films were deposited using a sol-gel technique [2] on the -axis oriented sapphire substrate with 2 2 0.05 cm dimensions. The deposition is carried out by spin coating and by repeating the sequence. Typically, we performed 20 repetitions, which permits us to obtain 300-nm-thick BST thin films. Crystallization of film is obtained by heating the sample at 750 C during 1 h in the air. Crystal quality of epilayers were characterized by X-ray diffraction. The films were polycrystalline, with a perovskite structure, without secondary phases. Surface morphology was

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Fig. 1. SEM of an IDC. Finger width: 1.04 m, finger spacing: 885 nm, pitch: 1.99 m. Fig. 2. Frequency dependence of the real part of permittivity.

characterized by a scanning electron micrograph (SEM) and atomic force microscopy. The BST surface is usually smooth, fairly uniform, and free of cracks, which are good features in view of the realization of tunable components. Electrical characterization of films was carried out first of -band. In all by means of a contactless technique in the short, the sample is inserted within a rectangular resonant cavity slightly changing its eigenmodes [3]. Transmission and reflection measurements were also performed by means of electrooptic sampling techniques whose results will be reported elsewhere. In this study, microwave characterizations were conducted in a wide frequency range, i.e., 1 kHz-40 GHz, using mainly bare coplanar waveguide (CPW) lines and IDCs. From the technological side, IDCs were patterned by electron beam lithography (LECA EBPG 5000). Cr/gold deposition was performed by a standard liftoff process, whereas a gold overlay was deposited by sputtering. Fig. 1 shows an SEM of a typical IDC used as a discrete device or as a load of transmission line. It is worth mentioning that discrete devices, as well as phase shifters are fabricated on the same substrate in order to avoid any dispersion of material characteristic from wafer to wafer. Fig. 2 shows the variations of the real part ( ) of the complex permittivity as a function of frequency measured at room temperature. A slight decrease of with frequency is observed from approximately 330 at 10 kHz to 290 at 40 GHz. This is in agreement with the empirical Curie–Von Schwindler Law used in [4]. There is no dielectric relaxation domain in this investigated frequency range. Measurements up in the terahertz frequency range tends to confirm this trend up to 500 GHz. The is almost constant with frequency with an avloss tangent erage value of 5 10 in the frequency range investigated. Turning now to the tunability, the variations of capacitance as a function of a dc bias are reported in Fig. 3 from 40 to 40 V at a frequency of 22 GHz. This study has been conducted with finger-shaped IDCs having an unbiased capacitance value around 0.1 pF. The topology is identical to that of capacitance shown in Fig. 1 with a finger width and an interspacing of 1 m. No hysteresis effect is noticed, which gives evidence that BST film is in a paraelectric state. Quantitatively, the capacitance value decreases by a factor of 2 between the unbiased condition and a bias of 40 V. The dielectric permittivity was

Fig. 3. Capacitance voltage characteristic measured at 22 GHz.

retrieved by means of the formula published in [5]. The voltage control of is roughly the same as the capacitance one. We have also measured the evolution of the conductance upon bias. From and , can be determined as a is divided by function of dc bias. As in [6], we found that approximately 2 over the voltage range considered in this study: let us mention, however, that the consequence of such a decrease can be of minor concern since, in a differential mode, we have to take into account the worst case, i.e., the losses at 0 V. This tunability was found rather constant (at 40%) from 1 kHz at least up to 40 GHz. This evolution, independent of the frequency, is quite remarkable and shows that paraelectric BST films can be used as voltage-controlled material in microwaves, as well as in RFs. III. OPTIMIZATION OF A PHASE SHIFTER An analog BST phase shifter operating at millimeter waves was first described by Liu et al. [7]. The circuit consists of a CPW line periodically loaded by interdigitated BST capacitors. Any change of capacitance values under a dc bias changes the characteristic impedance so that a taper section is needed at both ends of the CPW. Layout and close-up optical views of the phase shifter, which illustrate the technology we employed, were displayed in [1] with the report of a 310 /3.6-dB -band. In this study, the dimension of the CPW lines are: 1) the width of the central strip 60 m; 2) the width of the slot: 180 m; and 3) the

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Fig. 4. Comparison of tunability at 1 MHz of finger-shaped voltage-controlled capacitances by varying the finger spacing.

metal thickness: 0.4 m. The length of a typical circuit with 29 loading elements is approximately 1.5 cm. In order to improve the performances of the devices, notably in terms of phase shift and of insertion loss, we carried out a parametric study by varying: 1) the number of IDC per unit length; 2) the relevant dimensions of IDCs; 3) the location IDCs; and 4) the shape of the taper. As seen in the following, the results measured experimentally show overall the same trends than those expected by a transmission line model, at least in the low part of the spectrum. Of major concern is the maximum phase shift at moderate voltage, depending mainly on the layout of BST capacitors. To this aim we varied the fingers spacing on a typical 1- m scale. In Fig. 4, we show the capacitance variations against voltage measured here at 1 MHz (but we saw above the frequency constant behavior) for three finger-shaped structures fabricated on the same BST film. The width between fingers are 0.75, 1, and 2 m, respectively, keeping constant the width to spacing ratio. For comparison, the unbiased values of capacitance were normalized to unity. It can be shown that micrometer and submicrometer scale finger-shaped capacitance yields the highest capacitance ratio. Enhanced fringing effects at shorter dimension can explain this optimum in the capacitance dimensions. As a consequence, the phase shifters, which integrate IDCs, were realized using a 1- m technology. Such an issue was recently studied in detail [8] by means of a finite-difference timedomain numerical method. In addition, there exists a tradeoff between the difficulty of fabrication (notably of compatibility with a photolithography setup) and the performances of phase shifters. The second figure-of-merit, which is imperative to take into account, is the insertion losses. To this aim, we varied the transverse dimensions of the coplanar line and the loading factor , which is defined as the ratio per unit length between unbiased IDC and distributed capacitance of the coplanar line. As shown in Fig. 5, the loading factor strongly influences the insertion losses, which are plotted here at 10 GHz for three values of , 1.4, 2, and 3, respectively. It can be noticed that the losses increase drastically with , which has to be chosen close to 1.4 for the range investigated here with a minimum insertion loss of approximately 5 dB. Let us mention that this value is close to by Nagra and York [9], which describe the one found phase shifters based on GaAs varactors.

Fig. 5. Insertion loss at 10 GHz as a function of loading factor.

Fig. 6. Phase versus frequency of scattering parameter S and at 40-V continuous line.

at 0-V dotted line

IV. PHASE-SHIFTER CHARACTERIZATION The characterizations of the phase shifters were carried out at microwave by means of a vector network analyzer (VNA) from Agilent Technologies (E8361A: 10 MHz-67 GHz) and coplanar probes from GGB Industries (40A-GSG-125-DP). The VNA was calibrated from 10 MHz to 40 GHz using a standard calibration technique. Fig. 6 shows the frequency dependence of the phase of for unbiased capacitances and with a 40-V bias. It can be seen that the phase variations at 0 V are steeper than those at 40 V and that the differential phase shift increases upon frequency. (insertion loss) as a function of freThe magnitude of quency is reported in Fig. 7 for 0 and 40 V, respectively. Insertion losses are below 5 dB for a frequency around 30 GHz for the latter and around 15 GHz for the former. A rapid rolloff can be noted above these frequencies, which is explained by the fact that we are approaching Bragg frequency of circuit [9]. The explanation for higher insertion losses at 0 V is twofold. First of all, the characteristic impedance of the periodically loaded transmission line is higher than 50 , typically 75 , whereas it is close to 50 with a 40-V bias. Secondly, dielectric losses of BST at 40 V are less than the one at 0 V. The variation of the differential phase shift as a function of the bias is reported in Fig. 8 for various frequencies. The objective of a 360 value of phase shift is obtained at 30 GHz with a 40-V bias and at 40 GHz with 17 V only. To the best of our knowledge, this result compares favorably in terms of tuning voltage with

VÉLU et al.: 360 BST PHASE SHIFTER WITH MODERATE BIAS VOLTAGE AT 30 GHz

Fig. 7. Magnitude of S

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Fig. 10.

K figure-of-merit as a function of frequency.

as a function of frequency.

For the losses, as the phase shift is differential, we consider the worst case, i.e., the losses at 0 V. as a function of frequency is given in The evolution of Fig. 10 for bias ranging from 10 to 40 V. After a steep increase, shows a slight decrease between 10–40 GHz defining an operating frequency window. The maximum value range from 7 /dB at 10 V to 27 /dB at 40 V. V. CIRCUIT MODELING AND FULL-WAVE ANALYSIS

Fig. 8. Phase shift versus bias voltage by taking frequency as a parameter.

Let us now consider the general trends of circuit analysis, which can be conducted by means of a transmission line model periodically loaded by discrete capacitance in a parallel configuration. For the sake of simplicity, we also assumes a lossless case. The line is described by means of the distributed inducin Henrys/meter and capacitance in Farads/meter. tance (where can be Over a circuit element is defined by defined by the pitch of the periodic structure), the cumulated and capacitance are and inductance . of the unloaded With these notations, the phase constant line is (1) For the loaded line and unbiased condition, the phase

is (2)

Fig. 9. Phase shift versus frequency for various bias voltage.

The corresponding phase with a bias

is (3)

respect to the results published in the literature. For instance, the maximum phase shift obtained by Liu et al. was 310 at 35 GHz with a 100-V bias for a BST film deposited by sputtering. In this study, the highest phase shift is 600 at 40 GHz with a 40-V bias. Fig. 9 shows the variation of the differential phase shift as a function of frequency for different values of the bias voltage. The evolution is linear up to approximately 25 GHz, which is half of the Bragg frequency for our phase shifter. The evolution is then all the more nonlinear as we approach the Bragg frequency. A figure-of-merit is usually defined in degrees/decibels to take into account both the phase shift and insertion losses.

The differential phase shift is (4) In this expression, is the loading index defined previously: and , and characterizes the tunability. Expression (4) is valid for frequencies less than Bragg fredefined without bias by quency (5)

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Fig. 11. Maximum phase shift calculated by a circuit model and compared to experiment. Fig. 12. Frequency variation of

1

with x

= 1:4.

The phase shift achieved by applying a maximum dc voltage is noted and is given by (6) with

As aforementioned, various phase shifters were fabricated with different values of loading index ranging between 1–3. Fig. 11 shows the experimental values of the phase shift measured at calculated with (6). was 10 GHz and the values obtained from the measurement of the BST capacitance with and without bias. Overall, the agreement between experiment and the circuit modeling is good. Moreover, the plots displayed in Fig. 11 show that the phase shift is almost constant for in excess to 1.4. We also previously showed experimentally that , which appears the insertion losses are the lowest for as the optimum loading factor, which gives the highest figure of merit. From (6), the phase shift increases linearly with frequency. up to Fig. 12 thus shows the frequency dependence of 20 GHz and compares these variations to experiment for a phase . A good agreement is observed up to shifter defined by 15 GHz, however, with a growing discrepancy above. In conclusion, the circuit model appears relatively correct up to half of the Bragg frequency, whereas above a refined model taking into account all the losses appears necessary [9]. At higher frequencies, it become more and more difficult to analyze the phase shifter in a circuit formalism and, as a consequence, full-wave electromagnetic modeling was carried out using Ansoft’s commercial software High Frequency Structure Simulator (HFSS) [10]. In a first stage, we performed the numerical simulations of a coplanar line integrating a 0.3- m -thick BST film without taper and by targeting a characteristic . impedance and variations versus Fig. 13 shows the calculated frequency that were compared with experimental results. The best fit was obtained for the following dielectric characteristics and . Let us note that of the BST film

Fig. 13. Comparison between full-wave analysis and experiment for a 50bare CPW fabricated onto a 0.3-m-thick BST layer.



these values are in rather good agreement with those determined on IDCs. In a second stage, we performed simulations of the phase shifter. When we considered the full circuit, however, some discrepancy with the experiment appeared, which was explained by numerical errors. In order to face such a difficulty, it was thus decided to simulate one cell by full-wave analysis and to use a chain matrix technique afterwards in order to cascade the various elemental cells. The results of these calculations are displayed in Fig. 14 for 0 and 35 V, respectively. For the latter, the agreement is very good if some discrepancy is apparent above 20 GHz for the unbiased condition. As expected, a rapid rolloff of transmission is seen at 0 V due to a Bragg frequency around 30 GHz, whereas this limitation is shifted at higher frequency at 35 V due to the decrease in capacitance [see (5)]. VI. CONCLUSION We have reported on microwave and millimeter-wave characterizations of Ba Sr TiO thin films deposited by sol-gel on a -axis sapphire. These films are in a paraelectric state at room temperature. The mean value of the real part of the per, and for loss tangent, . mittivity is The variation of permittivity as a function of a dc bias measured from 40 to 40 V with interdigitated capacitors shows a tunability of 40 almost constant as a function of frequency.

VÉLU et al.: 360 BST PHASE SHIFTER WITH MODERATE BIAS VOLTAGE AT 30 GHz

Fig. 14. Comparison between full-wave analysis and experiment for a phase shifter with x = 1:4 and a Bragg frequency of 30 GHz.

It was shown that loss decreases upon bias voltage, the lowest at 40 V. One way to decrease has value being been proposed by Tagantsev et al. [11] for BST ceramics. It consists of introducing inclusions of dielectric particles with very low loss, typically MgO. A loss tangent of 0.01 at 10 GHz was thus demonstrated by the authors. It is believed that such a technique could be adapted to a sol-gel technique in order to deposit a BST/MgO film on sapphire. It has also been recently reported that small concentrations of acceptor dopants can dramatically modify the properties of perovskite oxide thin-film materials such as BST. To date, the most notable reduction of loss has also been demonstrated in Mg acceptor doped BST thin films [14]–[16]. Analog phase shifters using BST interdigitated capacitors on a micrometer scale periodically loading a high-impedance CPW line were subsequently fabricated. The design was optimized in -band. A 600 phase order to obtain a high phase shift in the shift was obtained at 40 GHz with a 40-V bias and 360 with a 17-V bias. At 40 GHz, the losses of the circuit without bias are relatively high, with a figure-of-merit of 22 /dB @ 40 GHz and 40 V. Work is in progress to fabricate left-handed phase shifters using the same technology, which can also be designed to realize composite left- and right-handed ones. First attempts to realize such phase shifters were published very recently in [12] and [13]. ACKNOWLEDGMENT The authors thank P. Mounaix, Centre de Physique Moléculaire Optique et Hertzienne (CPMOH) Laboratory, Bordeaux University, Bordeaux, France, for the terahertz measurements made on our BST films between 0.2–1.5 THz. REFERENCES [1] G. Vélu, K. Blary, L. Burgnies, J. C. Carru, E. Delos, A. Marteau, and D. Lippens, “A 310 /3.6 dB K -band phase shifter using paraelectric BST thin films,” IEEE Microw. Wireless Compon. Lett., vol. 16, no. 2, pp. 87–99, Feb. 2006. [2] G. Vélu, J. C. Carru, E. Cattan, D. Remiens, X. Mélique, and D. Lippens, “Deposition of ferroelectric BST thin films by sol-gel route in view of electronic applications,” Ferroelectrics, vol. 288, pp. 59–69, 2003.

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[3] D. C. Dube, M. T. Lanagan, J. H. Kim, and S. J. Jang, “Dielectric measurements on substrate materials at microwave frequencies using a cavity perturbation technique,” J. Appl. Phys., vol. 63, no. 7, pp. 2466–2468, Apr. 1988. [4] T. Hamano, D. J. Towner, and B. W. Wessels, “Relative dielectric constant of epitaxial BaTiO thin films in the GHz frequency range,” Appl. Phys. Lett, vol. 83, no. 25, pp. 5274–5276, Dec. 2003. [5] S. S. Gevorgian, T. Martinsson, P. L. J. Linner, and E. L. Kollberg, “CAD models for multilayered substrate interdigital capacitors,” IEEE Trans. Microw. Theory Tech., vol. 44, no. 6, pp. 896–904, Jun. 1996. [6] G. Vendik, S. P. Zubko, and M. A. Nikol’ski, “Microwave loss factor as a function of temperature, biasing field, barium concentration and frequency,” J. Appl. Phys., vol. 92, pp. 7448–7452, Dec. 2002. [7] Y. Liu, S. Nagra, E. Erker, P. Periaswany, T. Taylor, J. Speck, and R. York, “BaSrTiO interdigitated capacitors for distributed phaseshifter applications,” IEEE Microw. Guided Wave Lett., vol. 10, no. 11, pp. 448–450, Nov. 2000. [8] A. Giere, P. Scheele, C. Damm, and R. Jakoby, “Optimization of uniplanar multilayer structures using non linear tunable dielectrics,” in Proc. 35th Eur. Microw. Conf., Paris, France, Oct. 2005, pp. 561–564. [9] A. S. Nagra and R. A. York, “Distributed analog phase shifters with low insertion loss,” IEEE Trans. Microw. Theory Tech., vol. 47, no. 9, pp. 1705–1711, Sep. 1999. [10] HFSS. ver. 9, Ansoft Corporation, Pittsburgh, PA, 2005. [11] A. K. Tagantsev, V. O. Sherman, K. F. Astafiev, J. Venkatesh, and N. Setter, “Ferroelectric materials for microwave tunable applications,” J. Electroceram., vol. 11, pp. 5–66, 2003. [12] D. Kuylenstierna, A. Vorobiev, P. Linner, and S. Gevorgian, “Composite right/left handed transmission line phase shifter using ferroelectric varactors,” IEEE Microw. Wireless Compon. Lett., vol. 16, no. 4, pp. 167–169, Apr. 2006. [13] J. Perruisseau and A. K. Shrivervik, “Composite right/left handed transmission line metamaterial phase shifters (MPS) in MMIC technology,” IEEE Trans. Microw. Theory Tech., vol. 54, no. 4, pp. 1582–1589, Apr. 2006. [14] M. W. Cole, C. Hubbard, E. Ngo, M. Ervin, M. Wood, and R. G. Geyer, “Structure-property relationships in pure and acceptor-doped Ba Sr TiO thin films for tunable microwave device applications,” J. Appl. Phys., vol. 92, pp. 475–483, 2002. [15] P. C. Joshi and M. W. Cole, “Mg-doped Ba Sr TiO thin films for tunable microwave applications,” Appl. Phys. Lett., vol. 77, pp. 289–291, 2000. [16] M. W. Cole, W. D. Nothwang, C. Hubbard, E. Ngo, and M. Ervin, “Low dielectric loss and enhanced tunability of Ba Sr TiO based thin films via material compositional design and optimized film processing methods,” J. Appl. Phys., vol. 93, pp. 9218–9225, 2003.

Gabriel Vélu was born in Béthune, France, in 1969. He received the Ph.D. degree in génie des procédés from the Université de Valenciennes et du Hainaut Cambrésis, Valenciennes, France, in 1998. In 1999, he joined the Université du Littoral-Côte d’Opale, Calais, France, where he is currently an Assistant Professor and a member of the Laboratoire d’Etudes des Matériaux et des Composants pour L’Electronique (LEMCEL). His research interests include ferroelectric thin-films depositions and their tunable applications in the microwave field.

Karine Blary was born in Orléans, France, in 1975. She received the Engineer degree with a speciality in material science from the Ecole Universitaire Des Ingénieurs de Lille (EUDIL), Lille, France, in 1999, and the Ph.D. degree for her work on opto-electronic switching devices with the Institut d’Electonique, de Microélectronique et de Nanotechnologie (IEMN) from the Université de Lille 1, Villeneuve d’Ascq, France, in 2003. She is currently a Research Engineer in charge of process development with the “open lab” service, Université de Lille 1, which is concerned with the collaboration between other academic research centers in France and with the IEMN. She has recently been involved in electronics, opto-electronics, and microelectromechanical systems (MEMS) projects.

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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 55, NO. 2, FEBRUARY 2007

Ludovic Burgnies received the Ph.D. degree in electronics from the Université de Lille 1, Villeneuve d’Ascq, France, in 1997. From 1994 to 1997, he was focused on transport analysis in electronic quantum devices. In 1999, he joined the Université du Littoral–Côte d’Opale, Calais, France, where he is currently an Assistant Professor. His main research is concerned with the design of agile microwave devices based on ferroelectric materials.

Aurélien Marteau was born in Saint Jean d’Angely, France, in 1979. He received the Engineer degree from the Ecole Nationale Supérieure d’Ingénieurs de Limoges (ENSIL), Limoges, France, in 2003, the Master degree from the University of Limoges (DEA Télécommunications Hautes Fréquences et Optiques), Limoges, France, in and is currently working toward the Ph.D. degree within the DOME Group, Institut d’Electronique, de Microélectronique et de Nanotechnologie (IEMN), Université des Sciences et Technologies de Lille (USTL), Villeneuve d’Ascq, France. His current interests concern the modeling of complex electromagnetic structures including double-negative media and their fabrication and characterization.

Grégory Houzet was born in Cambrai, France, on November 21, 1982. He received the Master degree from the Université du Littoral Côte d’Opale (ULCO), Calais, France, in 2006, and is currently working toward the Ph.D. degree at the Institut d’Electronique de Microélectronique et de Nanotechnologie (IEMN), Université des Sciences et Technologies de Lille (USTL), Lille, France. His current interests concern transmission medium with metamaterials and negative-refraction devices.

Didier Lippens received the Master of Science degree in electronics engineering, Ph.D. degree, and Doctor ès Sciences degree from the Université des Sciences et Technologies de Lille, Lille, France, in 1975, 1978, and 1984, respectively. From 1980 to 1981, he was a Research Engineer with Thomson CSF, where he led the Quantum and Terahertz Devices Team until 2001. He is currently the Head of the Quantum Opto and Micro Electronics Devices Group (DOME), Department of High Frequency and Semiconductor, Institut d’Electronique de Microélectronique et de Nanotechnologie (IEMN), Université de Lille 1, Villeneuve d’Ascq, France. He is currently a Professor of electrical engineering with the Université de Lille 1, where he is mainly concerned with nanotechnology and nanosciences. He has been involved with molecular dynamics in liquid crystals and semiconductors physics and is currently more concerned with nonlinear electronics and opto-electronics along with electromagnetism in artificial media. He has undertaken pioneering research on resonant tunnelling devices and, more generally, on heterostructure semiconductor devices. His current interests are terahertz sources, notably quantum cascade lasers (QCLs), photomixers and heterostructure barrier varactors, photonic bandgaps, and metamaterials-based passive and active devices.

Jean-Claude Carru is currently a Professor of electrical engineering with the University of Littoral-Côte d’Opale, Calais, France, where since 1992, he has also been the Director of the Laboratoire d’Etudes des Matériaux et des Composants pour L’Electronique (LEMCEL), Calais, France. He has coauthored over 160 scientific publications. His current research interest concerns the characterization of materials in a large-frequency range from dc to millimeter waves in view of applications in electronics.

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S. Kang P. Kangaslahtii V. S. Kaper B. Karasik N. Karmakar A. Karwowski T. Kashiwa L. Katehi H. Kato K. Katoh A. Katz R. Kaul R. Kaunisto T. Kawai K. Kawakami A. Kawalec T. Kawanishi S. Kawasaki H. Kayano M. Kazimierczuk R. Keam S. Kee L. C. Kempel P. Kenington A. Kerr A. Khalil A. Khanifar A. Khanna F. Kharabi R. Khazaka J. Kiang J. F. Kiang Y. W. Kiang B. Kim C. S. Kim D. I. Kim H. Kim H. T. Kim I. Kim J. H. Kim J. P. Kim M. Kim W. Kim S. Kimura N. Kinayman A. Kirilenko V. Kisel M. Kishihara A. Kishk T. Kitamura K. I. Kitayama T. Kitazawa T. Kitoh M. Kivikoski G. Kiziltas D. M. Klymyshyn R. Knochel L. Knockaert Y. Kogami T. Kolding B. Kolundzija J. Komiak G. Kompa A. Konczykowska H. Kondoh Y. Konishi B. Kopp K. Kornegay T. Kosmanis P. Kosmas Y. Kotsuka A. Kozyrev N. Kriplani K. Krishnamurthy V. Krishnamurthy C. Krowne V. Krozer J. Krupka W. Kruppa D. Kryger R. S. Kshetrimayum H. Ku H. Kubo A. Kucar A. Kucharski W. B. Kuhn T. Kuki A. Kumar M. Kumar C. Kuo J. T. Kuo H. Kurebayashi K. Kuroda D. Kuylenstierna M. Kuzuhara Y. Kwon G. Kyriacou P. Lampariello M. Lancaster L. Langley U. Langmann Z. Lao G. Lapin L. Larson J. Laskar M. Latrach C. L. Lau A. Lauer J. P. Laurent D. Lautru P. Lavrador G. Lazzi B. H. Lee C. H. Lee D. Y. Lee J. Lee J. F. Lee J. H. Lee J. W. Lee R. Lee S. Lee S. G. Lee S. T. Lee S. Y. Lee T. Lee T. C. Lee D. M. Leenaerts Z. Lei G. Leizerovich Y. C. Leong R. Leoni P. Leuchtmann G. Leuzzi A. Leven B. Levitas R. Levy G. I. Lewis H. J. Li L. W. Li X. Li Y. Li H. X. Lian C. K. Liao M. Liberti E. Lier L. Ligthart S. T. Lim E. Limiti C. Lin F. Lin H. H. Lin

J. Lin K. Y. Lin T. H. Lin W. Lin Y. S. Lin E. Lind L. Lind L. F. Lind D. Linkhart P. Linnér D. Linton A. Lipparini D. Lippens V. Litvinov A. S. Liu C. Liu J. Liu J. C. Liu Q. H. Liu S. I. Liu T. Liu T. P. Liu O. Llopis D. Lo J. LoVetri N. Lopez Z. Lou M. Lourdiane G. Lovat D. Lovelace H. C. Lu K. Lu L. H. Lu S. S. Lu Y. Lu V. Lubecke S. Lucyszyn R. Luebbers N. Luhmann A. Lukanen M. Lukic A. D. Lustrac J. F. Luy C. Lyons G. Lyons G. C. M H. Ma J. G. Ma Z. Ma P. Maagt S. Maas G. Macchiarella P. Macchiarella J. Machac M. Madihian A. Madjar V. Madrangeas A. Maestrini G. Magerl S. L. Mageur A. A. Mahmoud S. Mahmoud F. Maiwald A. H. Majedi M. Makimoto S. Makino J. Malherbe G. Manara R. Manas G. Manes T. Maniwa R. Mansour D. Manstretta J. Mao S. G. Mao A. Margomenos R. Marques G. Marrocco J. Martel J. Martens J. Marti G. Martin E. Martinez K. Maruhashi J. E. Marzo H. Masallaei N. Masatoshi D. Masotti G. D. Massa B. Matinpour T. Matsui A. Matsushima S. Matsuzawa H. Matt G. Matthaei L. Maurer J. Mayock J. Mazierska S. Mazumder G. Mazzarella K. McCarthy G. McDonald R. McMillan D. McNamara D. McQuiddy F. Medina C. Melanie A. Á. Melcon F. Mena C. C. Meng H. K. Meng W. Menzel F. Mesa A. C. Metaxas R. Metaxas P. Meyer E. Michielssen A. Mickelson D. Miller P. Miller B. W. Min R. Minasian J. D. Mingo J. Mink B. Minnis F. Miranda D. Mirshekar C. Mishra S. Mitilineos R. Mittra K. Miyaguchi M. Miyakawa H. Miyamoto R. Miyamoto M. Miyashita M. Miyazaki K. Mizuno S. Mizushina J. Modelski W. V. Moer S. Mohammadi H. Moheb J. Mondal M. Mongiardo P. Monteiro C. Monzon A. D. Morcillo J. Morente T. Morf D. R. Morgan M. Morgan

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