Electronics For Microwave Backhaul 1630810150, 9781630810153

Table of contents :
Electronics for Microwave Backhaul......Page 1
Contents......Page 6
Foreword......Page 12
Preface......Page 16
Acknowledgments......Page 18
1.1.1 Services......Page 20
1.1.2 Numbers......Page 22
1.2.1 What Is the Backhaul?......Page 24
1.2.2 How Does the Mobile Evolution Influence the Backhaul?......Page 26
1.2.3 Microwave Backhaul......Page 27
1.3.1 Plesiochronous Digital Hierarchy......Page 30
1.3.2 Synchronous Digital Hierarchy......Page 31
1.3.3 Asynchronous Transfer Mode......Page 34
1.3.4 Ethernet Protocol......Page 35
1.3.5 Synchronization......Page 36
1.4.1 Reliability Objectives......Page 37
1.4.2 Point-to-Point Propagation......Page 39
1.4.3 Fading......Page 42
1.4.4 Interference......Page 44
1.5 Description of Chapters......Page 45
References......Page 46
2.2 Microwave versus Wired......Page 48
2.2.1 Benefits and Disadvantages of Microwave......Page 50
2.3.1 Microwave Pioneers......Page 52
2.3.2 Early Years from the 1920s to the 1940s......Page 53
2.3.3 The Microwave Industry......Page 56
2.3.4 From Analog to Digital Microwave Radio......Page 59
2.4 Microwave Links Nowadays......Page 61
2.5 Microwave Backhaul: Trends and Expected Growth......Page 63
References......Page 65
3.1 Basic Architecture and Options......Page 68
3.1.2 Split Mount......Page 69
3.1.3 Full Outdoor......Page 71
3.2 Building Blocks and Their Role for Present Systems......Page 72
3.2.2 Microwave Module......Page 74
3.2.3 IF Board and Modem......Page 78
3.3 Branching......Page 80
3.4 Capacity Improvement......Page 82
3.4.2 Dual Polarization......Page 83
3.4.3 Line-of-Sight MIMO......Page 84
References......Page 85
4.1 Modulations and Coding......Page 86
4.1.2 QAM Symbol Generation and Detection......Page 87
4.1.3 Filtering, Timing, Carrier and Radio Frame Synchronization,
Equalization......Page 94
4.1.4 Forward Error Correction......Page 102
4.1.5 Interference Cancellation in Case of Frequency Reuse......Page 107
4.2.1 Introduction and Historical Background......Page 110
4.2.2 Nonlinear Distortion......Page 114
4.2.3 Quadrature Modulation/Demodulation Imperfections......Page 117
4.2.4 Phase Noise......Page 124
4.2.5 Other Impairments......Page 127
References......Page 128
5.1.1 Duplexing Techniques......Page 134
5.1.2 Receiver Tasks......Page 136
5.2.1 Noise Considerations: LNA......Page 137
5.2.2 Downconversion......Page 140
5.2.3 Baseband Analog Processing......Page 148
5.3 Technologies......Page 150
5.3.1 GaAs and Other III-V-Based Technologies......Page 151
5.3.3 Si RF-CMOS Technologies......Page 152
5.3.4 Focus on GaN Technologies......Page 153
5.3.5 What Is the Best Choice for Millimeter-Wave Applications?......Page 154
5.4 Low Noise Amplifiers......Page 157
5.4.1 Fundamentals......Page 158
5.4.2 Topologies......Page 166
5.4.3 Cascode Configurations......Page 170
5.5.1 Fundamentals......Page 177
5.5.2 Architectures......Page 183
5.5.3 Topologies......Page 185
5.6 Mixers......Page 192
5.6.1 Fundamentals......Page 193
5.6.2 Basic Architectures......Page 194
5.6.3 Gilbert Cell......Page 198
5.7 Analog-to-Digital Conversion......Page 200
5.7.1 Effective Number of Bits......Page 201
5.7.2 Sampling Frequency......Page 202
5.7.3 Noise Sources......Page 203
5.7.4 Architecture......Page 208
5.8 Real-World Example......Page 213
5.8.2 Baseband Analog Processing......Page 216
References......Page 217
6.1 Introduction......Page 224
6.2 Power Amplifier......Page 225
6.2.1 Power Amplifier Fundamentals......Page 226
6.2.2 Backhaul PA Requirements......Page 235
6.2.3 Tuned Load Class AB......Page 239
6.2.4 PA Design Strategies......Page 250
6.2.5 Power Combining, Drivers, and Stability
......Page 259
6.2.6 Backhaul PA Examples......Page 261
6.3.1 Direct Conversion (Homodyne)......Page 275
6.3.2 Heterodyne......Page 278
6.3.3 Single Side-Band Converters......Page 279
6.3.4 Mixers......Page 280
6.3.5 Voltage Controlled Oscillators......Page 284
6.3.6 Frequency Multipliers......Page 286
6.3.7 Backhaul Upconverter Examples......Page 287
6.4.1 Basis......Page 291
6.4.2 Characteristics and Figures of Merit......Page 293
6.4.3 DACs for Microwave Radio......Page 294
6.5.1 Basis......Page 295
6.5.2 Analog Microwave Predistortion......Page 296
6.5.4 Digital Predistortion......Page 297
References......Page 299
7.1 Basic Concepts......Page 304
7.1.1 Characteristics......Page 306
7.1.2 Requirements for Microwave Backhaul......Page 309
7.2.1 Parabolic Antenna: Basics......Page 310
7.2.2 Feeder......Page 312
7.2.3 Parabolic Antennas in Backhaul......Page 314
7.2.4 Nonparabolic Antennas......Page 316
7.2.5 Printed Antennas......Page 320
7.2.6 Slotted Waveguide Antennas......Page 321
7.2.7 Future......Page 323
References......Page 324
Acronyms......Page 326
About the Editors......Page 334
Index......Page 336

Citation preview

Electronics for Microwave Backhaul

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For a complete listing of the Artech House Microwave Library, turn to the back of this book.

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Electronics for Microwave Backhaul Vittorio Camarchia Roberto Quaglia Marco Pirola Editors

artechhouse.com

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Library of Congress Cataloging-in-Publication Data A catalog record for this book is available from the U.S. Library of Congress. British Library Cataloguing in Publication Data A catalog record for this book is available from the British Library.

ISBN-13: 978-1-63081-015-3 Cover Design by John Gomes © 2016 Artech House 685 Canton Street Norwood, MA 02062 All rights reserved. Printed and bound in the United States of America. No part of this book may be reproduced or utilized in any form or by any means, electronic or mechanical, including photocopying, recording, or by any information storage and retrieval system, without permission in writing from the publisher. All terms mentioned in this book that are known to be trademarks or service marks have been appropriately capitalized. Artech House cannot attest to the accuracy of this information. Use of a term in this book should not be regarded as affecting the validity of any trademark or service mark. 10 9 8 7 6 5 4 3 2 1

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Contents Foreword

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Preface

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Acknowledgments

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1 Introduction

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1.1 The Mobile Evolution 1.1.1 Services 1.1.2 Numbers

1 1 3

1.2 Mobile Networks and Backhaul 1.2.1 What Is the Backhaul? 1.2.2 How the Mobile Evolution Influences the Backhaul? 1.2.3 Microwave Backhaul

5 5 7 8

1.3 Transport Networks 1.3.1 Plesiochronous Digital Hierarchy 1.3.2 Synchronous Digital Hierarchy 1.3.3 Asynchronous Transfer Mode 1.3.4 Ethernet Protocol 1.3.5 Synchronization

11 11 12 15 16 17

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1.4 Link Dimensioning 1.4.1 Reliability Objectives 1.4.2 Point-to-Point Propagation 1.4.3 Fading 1.4.4 Interference

18 18 20 23 25

1.5

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Description of Chapters

References

27

2

29

Evolution of Microwave Radios

2.1 Introduction

29

2.2 Microwave versus Wired 2.2.1 Benefits and Disadvantages of Microwave

29 31

2.3 The Historical Evolution of Radio Links 2.3.1 Microwave Pioneers 2.3.2 Early Years from the 1920s to the 1940s 2.3.3 The Microwave Industry 2.3.4 From Analog to Digital Microwave Radio

33 33 34 37 40

2.4

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Microwave Links Nowadays

2.5 Microwave Backhaul: Trends and Expected Growth

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References

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3 Architecture

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3.1 3.1.1 3.1.2 3.1.3

49 50 50 52

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Basic Architecture and Options Full Indoor Split Mount Full Outdoor

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3.2 Building Blocks and Their Role for Present Systems 3.2.1 Antenna 3.2.2 Microwave Module 3.2.3 IF Board and Modem 3.2.4 Baseband Unit 3.2.5 Power Supply

53 55 55 59 61 61

3.3 Branching

61

3.4 Capacity Improvement 3.4.1 Modulation and Roll-Off 3.4.2 Dual Polarization 3.4.3 Line-of-Sight MIMO 3.4.4 Adaptive Communication Techniques

63 64 64 65 66

References

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4 Modem

67

4.1 Modulations and Coding 4.1.1 Radio Frame Generation 4.1.2 QAM Symbol Generation and Detection 4.1.3 Filtering, Timing, Carrier and Radio Frame Synchronization, Equalization 4.1.4 Forward Error Correction 4.1.5 Interference Cancellation in Case of Frequency Reuse

67 68 68 75 83 88

4.2 Countermeasures Against Imperfections in TX and RX Chains 4.2.1 Introduction and Historical Background 4.2.2 Nonlinear Distortion 4.2.3 Quadrature Modulation/ Demodulation Imperfections 4.2.4 Phase Noise 4.2.5 Other Impairments

98 105 108

References

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5 Receiver

115

5.1 Introduction 5.1.1 Duplexing Techniques 5.1.2 Receiver Tasks

115 115 117

5.2 Architecture 5.2.1 Noise Considerations: LNA 5.2.2 Downconversion 5.2.3 Baseband Analog Processing

118 118 121 129

5.3 Technologies 5.3.1 GaAs and Other III-V-Based Technologies 5.3.2 Si and SiGe BiCMOS Technologies 5.3.3 Si RF-CMOS Technologies 5.3.4 Focus on GaN Technologies 5.3.5 What Is the Best Choice for Millimeter-Wave Applications? 5.3.6 Practical Remarks

131

135 138

5.4 Low Noise Amplifiers 5.4.1 Fundamentals 5.4.2 Topologies 5.4.3 Cascode Configurations

138 139 147 151

5.5 Local Oscillators 5.5.1 Fundamentals 5.5.2 Architectures 5.5.3 Topologies

158 158 164 166

5.6 Mixers 5.6.1 Fundamentals 5.6.2 Basic Architectures 5.6.3 Gilbert Cell

173 174 175 179

5.7 5.7.1 5.7.2

181 182 183

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Analog-to-Digital Conversion Effective Number of Bits Sampling Frequency

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Contents

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5.7.3 Noise Sources 5.7.4 Architecture

184 189

5.8 Real-World Example 5.8.1 RF System in Package 5.8.2 Baseband Analog Processing

194 197 197

References

198

6 Transmitter

205

6.1 Introduction

205

6.2 Power Amplifier 6.2.1 Power Amplifier Fundamentals 6.2.2 Backhaul PA Requirements 6.2.3 Tuned Load Class AB 6.2.4 PA Design Strategies 6.2.5 Power Combining, Drivers, and Stability 6.2.6 Backhaul PA Examples

206 207 216 220 231 240 242

6.3 Upconversion 6.3.1 Direct Conversion (Homodyne) 6.3.2 Heterodyne 6.3.3 Single Side-Band Converters 6.3.4 Mixers 6.3.5 Voltage Controlled Oscillators 6.3.6 Frequency Multipliers 6.3.7 Backhaul Upconverter Examples

256 256 259 260 261 265 267 268

6.4 Digital-to-Analog Conversion 6.4.1 Basis 6.4.2 Characteristics and Figures of Merit 6.4.3 DACs for Microwave Radio

272 272 274 275

6.5 Linearization Techniques 6.5.1 Basis 6.5.2 Analog Microwave Predistortion

276 276 277

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6.5.3 Analog IF Predistortion 6.5.4 Digital Predistortion

278 278

References

280

7 Antenna

285

7.1 Basic Concepts 7.1.1 Characteristics 7.1.2 Requirements for Microwave Backhaul

285 287 290

7.2 Antennas for Microwave Backhaul 7.2.1 Parabolic Antenna: Basics 7.2.2 Feeder 7.2.3 Parabolic Antennas in Backhaul 7.2.4 Nonparabolic Antennas 7.2.5 Printed Antennas 7.2.6 Slotted Waveguide Antennas 7.2.7 Future

291 291 293 295 297 301 302 304

References

305

List of Acronyms About the Editors Index

307 315 317

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Foreword For over 20 years, microwave has been the primary solution for rapidly rolling out cost-effective mobile backhaul infrastructure worldwide. Over 50% of the world’s mobile base stations are connected to the core network using point-topoint and point-to-multipoint microwave technologies. In the recent years, new requirements coming from mobile broadband are challenging microwave technologies to be able to scale up in terms of capacity and to reduce the total cost of ownership in terms of serviceability, power consumption, and product cost itself. Long-term evolution (LTE) deployment, with advanced features like carrier aggregation and multiple-input, multiple-output (MIMO), in addition to new allocations of spectrum in the order of the hundreds of megahertz, requires higher and higher transmission capacity, up to the gigabits per second to the macro tail site and multigigabits per second to the aggregation point. The increase of the number of sites, the network sharing [between site, radio access network (RAN), backhaul], and consolidation of operators, in addition to the penetration of fiber, significantly modify the backhaul network topology: •• Shorter networks, fewer number of hops from the fiber point of pres-

ence to the tails site in star configurations; •• Shorter hops, in order to cover shorter distances among macro sites (the average cell in urban and suburban areas in Europe is on the order of 500 m). The need to increase capacity and quality of experience by users in hot spots and to improve coverage in black spots leads to network densification through xi

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the deployment of small cells in urban environments at street level dictating a complete new set of requirements in terms of visual impacts, power consumption, and installation on nontelecom infrastructures like lampposts or traffic lights. A strong requirement for the LTE, and even stronger for the upcoming 5G, is the need to reduce the whole latency to respectively 10 and 1 milliseconds, with consequent necessity to reduce in the backhaul the delay introduced by baseband processing and coding. The progressive introduction of distributed and centralized RAN with the necessity to front-haul the Common Public Radio Interface leads to at least an order of magnitude increase of capacity and severe requirements in terms of a round-trip delay for the transmission on the order of 100 μs. The mobile backhaul represents today more than 85% of the microwave point-to-point applications and therefore is the main driver of the technical and technological evolution of system and modules. Furthermore, a significant emerging application is the fixed broadband, where wireless transmission can represent a very convenient complement to fiber or copper in many scenarios. Microwave radio has come a long way in the past 30 years, from large, heavy, all-indoor analog systems to the current split-mount/full-outdoor, nodal, high-speed digital Internet Protocol (IP) transport platforms with further evolution being driven by the requirements outlined above. In particular, if we focus on the more characteristic microwave technologies related either to the mo-­ demodulation processing or to the radiofrequency, the most challenging requirement (i.e., the capacity increase) can be achieved either by improving spectrum efficiency or by finding more spectrum. Spectrum efficiency can be improved in several different ways according to the hop length, frequency licensing regime, and other specific conditions: •• Increase the order of the modulation scheme, with products offering up

to 8192 QAM in the near future. Very effective correction coding and adaptive techniques like adaptive coding and modulations and adaptive bandwidth permit an easier planning of link meeting the tough requirements of availability of the connection against severe rain phenomena. •• Use cross-polarization interference cancellation (XPIC) to use the same frequency channel on the other polarization. •• Use line-of-sight MIMO to further double the spectrum efficiency with respect to XPIC. •• Use full duplex, the possibility to simultaneously transmit and receive on the same channel by exploiting both isolation between transmitter and receiver antennas and powerful cancellation algorithms. Finding more spectrum by increasing the channel width in traditional microwave bands is extremely difficult. The most popular bands are already congested,

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Foreword

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even if reusability of the frequency channels is possible up to a certain extent, by improving spectrum efficiency on a geographical basis. Furthermore, some benefit can come by reducing the mutual interference of adjacent links by using higher-quality antennas with lower sidelobes or interference mitigation algorithms. From the above considerations, network densification and network sharing, resulting in hops shorter than the usual lengths of a few years ago, make the use of millimeter-wave bands suitable for the transmission of capacities up to 10 Gbit/s per carrier over distances in the order of 2 or 3 km. As the 5G becomes the next hot topic, the microwave transmission is asked to keep pace with the evolution of the services connecting everything, everywhere in mobility at an unprecedented bandwidth, with many more high-­ capacity links extending to the edge of the network. The electromagnetic spectrum resources are shared by many applications such as broadcasting communication, radio navigation, radio astronomy, satellites, radars, and so on. To avoid the conflicting use and facilitate the management of this resource, the spectrum is divided in bands. For fixed radio services, present channel plans recommended by ITU-R with different channel spacing, frequency spacing for full duplex, are subject to country-specific licensing regimes resulting in operational costs for the operators. So far, frequency bands from 6 to 42 GHz have been the most commonly used for fixed point-to-point services, to the point of being called “traditional bands” by the microwave industry, while millimeter-wave is improperly considered to start at 50 GHz. High interest in millimeter-wave bands has risen in the recent years due to the enormous amount of underutilized bandwidth that lies in this part of the electromagnetic spectrum. The significant advantages offered by the propagation characteristics in terms of frequency reusability and large channel bandwidths make millimeter-wave suitable for transmitting very high capacities of multigigabits per second for the connection of macro base stations. At the same time, it allows for the backhaul of small cells in dense urban environment thanks to very compact antenna size that makes products “blend” in the environment. The microwave industry realized the necessity to cooperate more, in order to facilitate the use of the millimeter-wave bands. In particular, the V-band (57–66 GHz), the E-band (71–76 and 81–86 GHz) and in the future higherfrequency bands above 90 GHz will be adopted for large-volume applications in the backhauling and front-hauling to support all services requiring high-speed wireless transmission. Activities are underway to standardize the W-band (92 to 114.5 GHz) and D-band (141 to 174.8 GHz) posing complex challenges to the evolution of the technology. From the considerations outlined so far, it is relatively easy to evince the importance that the research will play in the future in the many areas that will be necessary to cover in order to develop systems capable to operate at very

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high capacity with spectral efficiency and high performance at frequencies up to 175 GHz. Research and development are being focused on compound semiconductors in order to improve the performance of integrated high-frequency circuits with regard to the phase noise, linearity, power efficiency, and, above all, the move to the highest frequencies mentioned before. It is possible to see, also thanks to other commercial and high-volume applications, for example, automotive radar, emerging applications of silicon-based technology at frequencies not used so far by the microwave industry, permitting a much higher integration level and simplification of the manufacturing processes. The progress of semiconductors for the radio frequency, mainly higher performance and integration, and for digital processing, mainly reduced power consumption, permits the development of entirely new product concepts featuring beam-forming antennas, very wide channels up to 2 GHz with high modulations and coding schemes that were unfeasible just a few years ago. These new features will permit microwave to face the new challenges posed by the evolution of mobile communications. Renato Lombardi Chairman of ETSI ISG mWT Milan Microwave Competence Center Huawei Technologies Italia S.r.l. Milan, Italy

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Preface Twenty-first century engineers are often experts in a specific niche, but their knowledge of the complete framework is only partial. This is particularly true in electronics, where the evolution is tremendously fast and the systems contain many subsystems and specific modules of high complexity. Macro-systems are normally analyzed and designed by top-down approaches and only a few actors retain the vision of the entire scheme. Years ago when we started to work on power amplifier design for microwave backhaul, we directly confronted this problem. The design of the power amplifier had some peculiarities with respect to the classical design for mobile applications, but it is our core business. Furthermore, we come from academia, so by definition we know everything. Unfortunately, when we tried to dig a little deeper to understand the framework when writing the introduction to a scientific article, we got lost very quickly. Which are the working bands, the typical power levels, the protocols employed, the classical design strategies of the different blocks? What is the complete architecture and the regulations? Why are some bands used more, and what are the present and future trends? This information is clearly all available. However, at least in our experience, the sources are sparse and are not always easy to collect and interpret. Books on backhaul are either focused on the software layer, going into details a little too far from our competencies and interest, or very general. This led to the idea of a book focused on the electronics of the backhaul, which would describe with in detail all the subsystems responsible for transforming the information signal that comes from the baseband processing into an electromagnetic wave traveling in the air (and vice versa).

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This book presents an overview of the evolution of the electronics for microwave radios, from their initial development to present implementations and future trends. The seven chapters cover the main building blocks of backhaul radios, introducing the required theoretical foundations and highlighting the key aspects driving the design and implementation steps, keeping an eye on realworld industry products. This book has been written in strong cooperation with experts of the infrastructure coming from leading industries particularly active in the backhaul arena: Ericsson AB, Huawei Technologies, SIAE Microelettronica, and Flann Microwaves. Direct pointers have been added to reference sources that the reader can consult to go more deeply into the details of any item. Each block is correlated by real-world examples and, where possible, the foreseen trends are highlighted. Vittorio Camarchia Torino, Italy November 12, 2015 Roberto Quaglia Torino, Italy and Cardiff, United Kingdom November 12, 2015 Marco Pirola Torino, Italy November 12, 2015

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Acknowledgments The editors would like to thank Piero Coassini of SIAE Microelettronica, Franco Mucchietto of DSPM TELECOMUNICAZIONI and Renato Lombardi of Huawei Technologies for their help and support in the preparation of the book. The editors thank all their colleagues at Politecnico di Torino. As well, they acknowl­edge all the contributing authors, who dedicated their precious time to the writing of this book and contributed greatly to the vast subject of this book. Roberto Quaglia thanks Marco and Vittorio for their help and suggestions, not only in the writing of this book, but in particular in the years they have worked together. Dr. Quaglia thanks all his colleagues at Huawei. He expresses gratitude to Valter Benedetto; despite coming from a tiring working day, Valter still found the strength to teach him some interesting tips about backhaul radios. His friends and family, especially Laura, are sincerely acknowledged for the love, support, and care they give him every day, and he hopes that having enjoyed writing this book may at least in part repay the time he could not spend with them. Vittorio Camarchia and Marco Pirola thank all of their colleagues and friends who shared their time for fruitful discussions and suggestions for the realization of this book.

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1 Introduction Roberto Quaglia, Thomas Emanuelsson, Vittorio Camarchia, and Marco Pirola 1.1 The Mobile Evolution 1.1.1 Services Walking in the crowded streets of a city or standing on a crowded, hot bus, people may be curiously attracted by someone still pressing repetitively on the keyboard of a cellular phone. If the subject of everyone’s interest is observed more closely, it can be seen that the phone in use still has a real keyboard, and that the phone user’s attention is devoted to shrinking the words in strange acronyms, not to exceed the text message character-limit. This person is one of the few people who survived until the advent of smartphones, and still believes that sending a text message is the best you can do with a handled device. Perhaps, he never took a picture with his phone. There is a lot of discussion about the fact that in the future, society will be always connected. However, if the future is not already the present, it is not too far away either. The third and fourth generations of mobile networks have brought to the wide market the combination of the mobility offered by the cellular network and the connectivity of the Internet. To complete the picture, smartphones have made the experience easy, entertaining, and trendy. If you are not in a public place searching for a free wireless connection, you can use the broadband mobile network and still be able to browse the Internet or to chat on the newest social network. Taking some steps backwards, the real explosion of mobile communication began with the introduction of Global System for Mobile Communications 1

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(GSM) in Europe in the 1990s, with the second generation (2G) of mobile, while the previous generation, still based on analog communications, never achieved a large market success. The 2G was based on digital communications, and thanks to a wide penetration in the territory and to the price accessibility of the handsets, it achieved an extraordinary success. The main services were voice calls and the Short Messaging Services (SMS). Another important feature was the use of the Subscriber Identity Module (SIM), which permitted a user to simply transfer his or her phone number from one cellphone to another one. GSM also spread out of Europe, reaching 80% of worldwide subscribers, becoming the most successful ever mobile standard. The 2G counterpart in North America and South Korea was the IS-95 standard, which offered similar services. In the same years, the diffusion of the Internet became capillary worldwide, and the mobile industry started to foresee the potential of handheld devices able to browse the Web and, in general, to access the other services of the worldwide network. For this reason GPRS and, later, EDGE were introduced in the early 2000s allowing mobile phones to browse simplified versions of the Web pages, and to read/send e-mails. The third generation (3G), introduced commercially in the 2000s, was initially thought to enable video-call services, but the main market soon became broadband connection. This success was also helped by the introduction of smartphones, which in turn exploded as the perfect media for this kind of application. A smartphone can be as powerful as a desktop computer, and since it is handled, it makes it possible to use Internet features with high-performance applications from everywhere at any time. For example, Voice over IP (VoIP) services can be run on a smartphone, and despite seeming to be nonsense, it can have cost advantages. One of the most demanding uses of VoIP is the sharing and watching of video that achieves very high peaks during live events (such as concerts or sporting events). The 3G also achieved good subscriber figures since it represents an alternative to fixed broadband connection (e.g., ADSL). In fact, computers can be connected, through USB or wireless, to a 3G modem. CDMA2000 is the 3G standard mainly adopted in North America/South Korea, while UMTS is adopted in Europe, China, and Japan. The fourth generation (4G), represented by LTE and WiMAX-A standards, is under deployment, but its penetration is still limited. An exception is South Korea, where it already reaches 60% of the population, while in Europe it is under 10% also in the best cases. The 4G introduces the possibility of watching high quality (HQ), real-time TV or online programs thanks to its very high speed. What can be observed is that the main service offered by mobile networks changed from voice calls to broadband Internet. However, passing from GPRS to LTE, from a user standpoint, the only difference has been an increasing speed. There is already a lot of talking about the fifth generation (5G), where the

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improvement will be again in terms of speed, but the real change will be on the kind of provided service. In fact, the fully connected society mentioned above will not only refer to connections among people, but also to a wide deployment of machine-to-people and machine-to-machine communications. Billions of potential users going from large grid plants to small sensors in home utilities are expected. Only the general framework of 5G is known, while implementation details are still not clear; in fact, only in 2020 5G standards will be defined. Despite that, some companies are already showing prototypes of equipment connecting wireless at 1 Gbps. However, by 2020, 4G penetration will probably be really capillary, and at that point the vision of what types of services people need and expect from the new mobile generation will be more clear. 1.1.2 Numbers The evolution of mobile standards has been practically driven by an increase of available bit rate for users. At the same time, new standards have involved new frequency bands and advanced methods to share the channel. Figure 1.1 shows the frequency bands adopted by several standards, while Figure 1.2 shows the growth of bit rate from 2G to 4G. The frequency bands used by the 2G standards are 850, 900, 1,800, and 1,900 MHz for the GSM and 800 and 1,900 MHz for the IS-95. GSM assigned to each user two frequency channels, for up and downlink. These channels are shared with other users, using a time division multiplexing (TDM) technique where every user has one of eight slots in a frame. IS-95 uses instead a CDMA approach for channel sharing. GPRS and EDGE adopt the same frequency bands and still use TDM for channel sharing. UMTS works mainly in the 1,885–2,025-MHz band for uplink and the 2,110–2,200-Hz band for downlink. In the United States the 1,700-MHz band is used instead of the 1,900-MHz one. Some native GSM frequencies are also adopted in some countries. The channels have 5-MHz bandwidth, and users share channels using a WCDMA technique. The user bit rate has increased in the several versions: for example, HSDPA allowed up to 7.2 Mbps

Figure 1.1  Frequency bands of worldwide deployed mobile networks.

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Figure 1.2  Peak download rate of user versus mobile technology.

in downlink (peak), while HSPA+ allows more than 40 Mbps in downlink. CDMA2000 has instead evolved from IS-95 on the same frequency bands and uses smaller (1.25 MHz) channel allocation. LTE is based on OFDM, and it supports many different bands allocation, for example, 700, 800, 900, 1,800, and 2,600 MHz in Europe and 1,800 and 2,300 MHz in Oceania. An important feature of 4G networks is the use of multiple input multiple output (MIMO) techniques for increasing the bit rate, that can reach a peak of 300 Mbps in downlink. The number of subscriptions to mobile services is still increasing worldwide, as can be seen in Figure 1.3, where the subscription to the different standards is differentiated. There are, however, regional differences: developed markets as Europe or North America contribute with subscription to broadband standards as LTE, while developing countries are still dominated by 2G subscription, thanks to the increased accessibility to mobile phones. Figure 1.4 shows the penetration of the standards in the different world regions, in terms of percentage of population served. Going towards 2020, the forecasts are stating that there will be a drastic reduction of 2G subscriptions mainly due to the shift of customers to 3G or 4G services.

Figure 1.3  W  orldwide mobile subscriptions: data until 2014 and forecast toward 2020. (Source: Ericsson Mobility Report, June 2015.)

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Figure 1.4  W  orldwide mobile subscriptions: technology share. (Source: Ericsson Mobility Report, June 2015.)

1.2 Mobile Networks and Backhaul 1.2.1 What Is the Backhaul? Mobile networks are distributed by nature, since they must cover a large territory. They are geographically divided into cells, each of them served by a base station that is responsible for providing access to the handsets. A base station transmits to users (downlink) and receives from them (uplink) on defined frequency bands. These bands can be reused by other cells situated at an adequate distance in order to minimize the interference. Cell cover areas of variable dimensions, which depend on the morphology of the territory and on the expected user demand. In order to allow a user to route or exchange packets with the Internet, the cells must be connected together in what it is called the network infrastructure; see the scheme in Figure 1.5. The network core, which is usually at the national

Figure 1.5  Block scheme of a mobile network infrastructure.

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level, stores in a database the position of every user, together with other information about users and network status. The connection between the core and the periphery of the network is the backhaul, composed by nodes and links. The core knows how to route a call or a packet, but it is the backhaul that physically transports the information. A mobile network covers thousands of square kilometers, including both big cities and remote mountain areas, and it must serve millions of users. This gives an idea of the complexity and cost for the deployment of a mobile network and also on how critical the role played by the backhaul in the network can be. The backhaul must transport the information in a reliable and cost-­ efficient way. In a voice call between handsets A and A¢, the handset A communicates through the access air interface with a base station. The base station is connected to the backhaul, which is responsible for aggregating the call from the several users and to route it to the core network. The core registers the call information, searches for the cell where the handset A¢ is positioned, and routes the call through the backhaul to reach it. Finally, handset A¢ will communicate with the base station covering the cell at which is positioned. The network has hence created a connection between the two handsets by assigning network resources to the voice call. From the handset perspective, the way that the network routes the call is transparent, meaning that it will only communicate with the base stations in its vicinity, and that during a call it will “talk” to the assigned base station only. The backhaul can either assign fixed physical resources (circuit switch) or virtual circuit resources (packet switch) to the call or connection. A voice call is an example of high priority traffic with stringent requirements in terms of delay and latency. However, the capacity required by a phone call is small, meaning that it easy to accommodate many users on a single backhaul connection. Intuitively, the best way to manage a phone call seems to be the assignment of fixed resources for the duration of the call. Figure 1.6 shows how the backhaul manages voice calls between A, B, C and A¢, B ¢ C ¢, respectively, by aggregating more voice channels on the same link to connect the core. It is clear

Figure 1.6  Example of backhaul operation in voice calls and data transfer to the Internet.

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that the peripheral backhaul nodes and links have to manage less traffic than the ones close to the network core. In the same figure, user D is connected to the Internet using a data connection. D will communicate with the base station, and the backhaul will transport the data packets to the core, which will interface with the Internet through an access point. Also in this case the mobile network can assign fixed or virtual resources to the Internet connection, depending on the backhaul architecture. This kind of traffic is completely different with respect to the voice call. It is bursty, meaning that the resources are heavily used for short periods, but the capacity requirement can be very high, if the visited Web page contains videos or a large amount of data. In this case, the assignment of fixed resources for the connection does not seem optimum. Finally, the backhaul is also responsible for transporting the network signaling (e.g., for network management, routing, and handovers). This is also lowcapacity, high-priority traffic. Referring to the ISO/OSI model that is intended for a general network of computers, we can state that the backhaul works on the four lower layers of the stack. Most of this book is focused on layer 1, the physical layer, that describes how the raw information is exchanged in terms of electrical signals. Less details are provided on layer 2 (data link), layer 3 (network), and layer 4 (transport), but more details can be found in [1]. 1.2.2 How Does the Mobile Evolution Influence the Backhaul? The decision of operators to provide coverage of an area with an improved generation of service (e.g., passing from 3G HDSPA to 4G LTE) practically leads nowadays to the decision of upgrading the backhaul, both in terms of new links and of improving the existing ones. In fact, the installation of small base stations for microcells does not just require the obvious connection of that base station to the network, but it will also stress the capacity requirements of the backhaul portion that will serve it. However, we must also look at the other side of the issue. In particular, LTE has been regarded as a full packet network also to allow for a simplification of the infrastructure. For the operators, in the long term, it will be convenient to replace the native networks since the new solutions are more efficient. However, the passage cannot be immediate, because it would cost too much, and most of the users still use 2G/3G handsets. With the clear shift towards dominance of the data portion of the traffic, how can the backhaul efficiently manage both voice and data? The answer is not simple, because the technical considerations find their limit when they are in contrast with cost considerations. It is quite accepted by the industry, from a technical point of view, that a fully Ethernet-based backhaul will be the choice of the future. In fact, the technology is ready to guarantee the needed quality for priority services like voice calls on a packet-based network with

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Figure 1.7  E volution of base station capacity for evolving networks. (Source: Ericsson Microwave Towards 2020 Report.)

shared resources like the Ethernet. The advantage in terms of efficiency of the network with respect to fixed circuit-based solutions is really big. However, the operators are finding that the revenues deriving from mobile subscription are not proportional to the services that they provide. For this reason, they are reluctant to start a campaign of complete rethinking of the backhaul network, preferring to substitute or improve the infrastructure more carefully and gradually. An exception is the rapidly growing markets, like India, the Middle East, or Africa, where new networks are being installed starting from scratch. In this case, it is more convenient to develop Ethernet-based backhaul from the beginning. Figure 1.7 (left) shows how operators that decide to introduce 3G in 2013 and 4G in 2019 distribute the capacity on the base station sites. For example, an operator introducing 3G will start with 80% of sites with at least 8 Mbps, evolving to 80% of sites having at least 25 Mbps. Figure 1.7 (right) gives the same share for operators that introduced LTE in 2013 and evolving to LTE-Advanced. 1.2.3 Microwave Backhaul Practically, backhaul can be wired, using copper wires or optical fibers, or wireless, using point-to-point radio links. The latter, also known as microwave backhaul, is the subject of this book. Radio transceivers are used in mobile networks for the access portion, where base stations’ large transmitters and very sensitive receivers work at frequencies below 4 GHz to provide the downlink and uplink connections to the terminal users. However, radio transmissions, in particular at microwave and millimeterwave frequencies, are also widely adopted for backhauling, since digital radios can be very reliable, they can transport high capacity and can be easily deployed without the need of hard-wiring. These features often translate into lower cost for

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the network operators if compared to copper wires or fiber. A detailed discussion about the pros and cons of microwave radio with respect to wired solutions is provided in Chapter 2. The most adopted configuration in backhaul when using microwave radio is ideally a substitute for wire or fiber, in the form of a point-to-point (PtP) link. A PtP link is composed by a pair of radios (radio 1 and radio 2) with the two high gain antennas pointing towards each other, and deployed at a distance called hop length from each other (Figure 1.8). Point-to-multipoint radio has also been proposed for backhauling, in particular, in the 60‑GHz band, but their use is still limited. To form a network, more PtP links can be connected in different configurations; see Figure 1.9. The mobile network core is usually realized with fibers. Microwave can be deployed starting from regional rings and reaching the most peripheral base stations. Rings are employed to protect important traffic from long outages, and can also play a role with protocols such as Ethernet

Figure 1.8  A PtP microwave link using full-duplex communication.

Figure 1.9  An example of possible network implementation with microwave PtP links.

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to optimize the routing. When a single hop is not sufficient to cover a certain distance active repeaters are adopted, and the distance is covered using multiple hops. Passive repeaters and reflectors can be employed to reach hidden locations as mountain valleys. The PtP links use digital communications, where the digital bit streams to be transmitted are modulated into an analog microwave signal that theoretically only occupies a channel (i.e., a limited frequency band around a carrier). The capacity of the link is determined by the product of the channel bandwidth (in megahertz) and the spectral efficiency of the modulation (in bit/s/MHz). The communication between the two radios is in full duplex, meaning that radio 1 transmits and radio 2 receives on channel A around the frequency fa, while radio 2 transmits and radio 1 receives on channel B around f b; see Figure 1.8. Channel A will be the “go” channel for the radio 1 and the “return” channel for radio 2, respectively. The high gain antennas and the geographical planning of the several links permit adopting the same frequency pairs for several different links. The typical bands for microwave radios extend from the so-called sub-6 GHz bands to the millimeter-wave bands, today up to 86 GHz. Every band also defines the possible fa, f b pairs, with the |fa–fb| constant. Figure 1.10 shows a prospect of frequencies in use for microwave backhaul. Lower frequencies are usually adopted for longer hops, but still they can accommodate high capacity thanks to high‑order modulations, while E-band is suitable for short‑distance, high‑capacity links, thanks to the large channel availability. The 60‑GHz band is unlicensed, and is used for dense networks with high‑frequency reuse. Frequency bands below 4 GHz are for very long, low‑capacity hauls. A key feature of a transport network is its reliability, which for microwave radios is affected not only by the equipment design, but also by propagation conditions. The micro­ wave link design thus starts from considerations about the needed capacity and the reliability objectives. All frequency bands are affected by changes in the propagation conditions, meaning that the quality of the communication between the

Figure 1.10   Backhaul frequency bands.

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two radios will change in time. To guarantee on the same link both high capacity and high reliability, the present trend is to use adaptive coding, modulation, and bandwidth. In this way, the maximum capacity is exploited during good propagation periods, while priority traffic can be however guaranteed during adverse propagation times by employing a more robust, but less efficient, communication protocol. This and other methods for improving robustness and capacity will be discussed in Chapter 3. Mobile and backhaul networks evolved together, and different standards have been used requiring different characteristics to the underlying physical layer. In the following section, a brief introduction on the transport networks in use in microwave backhaul is given, followed by a qualitative discussion on the reliability objectives and link dimensioning.

1.3 Transport Networks The goal of a mobile network is to carry the information from the end user to its destination, that can be another user (in a voice call or for a text message) or an Internet access point (when browsing a Web page). The goal of the backhaul network is to transport the information between the base stations and the voice/data switch networks. In 2G networks, dedicated channels (time slots) were assigned to every voice call, using a TDM technique. While the reliability of such a choice is very high, its efficiency when transporting data traffic is far from being optimum. In fact, data traffic is typically noncontinuous, with large amount of data requested in a short period and with long pauses between one request and the successive. For this reason, Ethernet‑based backhaul has been introduced on 3G networks, exploiting the advantages of packet transport. Moreover, voice traffic can also be packetized, opening the possibility of transforming the backhaul into a full packet-based network. This solution is welcomed by the providers for its efficiency, but it needs very careful design to guarantee a carrier‑level service. Either for voice or data, multiplexing is a crucial function of the backhaul. In fact, the single calls or data packets need to be aggregated before they can be efficiently transported over the network. In backhaul nodes, the multiplexing function is carried out by the baseband unit. Since the focus of this book is on the radio transmission, a detailed analysis of multiplexers is not provided; however, the multiplexing techniques covering the backhaul network evolution are briefly overviewed [2]. 1.3.1 Plesiochronous Digital Hierarchy PDH is the acronym for plesiochronous digital hierarchy: plesio stands for nearly, meaning that the several digital streams are combined in order to look synchro-

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Figure 1.11  PDH multiplexing.

nous, even if they are not perfectly synchronous at the origin. This choice is driven by the fact that it is very difficult to provide the same clock reference to the nodes of a distributed network with the required frequency stability and accuracy. The PDH adopts, for every user, a pulse code modulation (PCM). Then it combines 30 user streams that use a common reference clock (i.e., real synchronicity), in a primary line called E1.1 Every E1 line has a nominal clock of 2,048 kbps ±50 ppm, and it is free-running with respect to the other E1 lines. The near-synchronicity is performed from the secondary multiplexing. Four incoming E1 streams are recorded in buffers at clock rates extracted from the streams themselves. Then the multiplexer is forced to operate at the highest of the incoming clocks, in order to do not overflow the input buffer. Moreover, in order to add control bits to the sequence, the multiplexing clock is actually run even faster. As a consequence, some of the slowest streams do not produce any bit to multiplex: through a technique called stuffing the empty spaces are filled with bits. Through control signals, the demultiplexer will recognize the real bits from the stuffing bits. The following multiplexing steps operate in the same way. Figure 1.11 shows the scheme of principle of PDH multiplexing. Scrambling is performed to obtain a pseudo-random bit stream with an even spectrum, from which it easier to extract the clock reference. However, this procedure makes it impossible to assign priority to one E1 line with respect to the others. Moreover, to extract a single channel, all the PDH streams must be demultiplexed. 1.3.2 Synchronous Digital Hierarchy Synchronous digital hierarchy (SDH) was introduced to allow interoperability between equipment based on different PDH standards (E1 or T1), and to

1. E1 is the primary line for International Telecommunication Union (ITU) standard (2,048 kbps), while in North America the standard it is called T1 (1,544 kbps).

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enhance the monitoring of the network performance with respect to PDH networks. In particular: •• Native streams, which can be originated from different standards, are

mapped into containers to form the higher‑level frame, without losing their original characteristics. •• The use of pointers permits synchronizing the container in the overall frame. •• The addition of overheads allows for performance monitoring throughout the network. Figure 1.12 shows how the original PDH streams (E1, T1, and E4, in this case) are mapped into Containers (C-XX  ). In this phase, stuffing is still used to compensate for clock allowed tolerance, so to ensure the correct recovery of

Figure 1.12  SDH multiplexing.

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the information at the demultiplexing end. Virtual containers (VC-XX  ) are created adding overhead, while Tributary Units (TU-XX  ) are formed including the pointer. Administrative Units (AU-XX  ) are composed by TUs and pointer structures used to identify the positions of VCs in the STM stream. Higher‑level containers are then created through multiplexing, using the pointer to maintain the synchronicity. The overheads permit to monitor every branch of the communication. Moreover, thanks to components called add-drop multiplexers (ADMs), tributaries to the SDH frames can be added or dropped without demultiplexing the entire frame. This is helped by the fact that SDH is based on byte interleaving, while PDH is based on bit interleaving. The final administrative unit group (AUG) stream is multiplexed from three AU3s or one AU4, and composes the STM-1 stream, with a 155 Mbps rate. Figure 1.13 shows the organization of an SDH network. The core is composed by cross-connect elements, able to manage big data rates and connected in a frame fashion. The core is connected to several rings or stars (usually at regional and metropolitan levels), composed by ADMs. ADMs receive data from one or more tributaries, and can be used to interlace rings at different hierarchies. On the external part of the SDH network lies the simplest terminal, the customer premise extension (CPE), that is the normal access point to the SDH network. In radio applications, which cover the outside branches of the network, the choice is usually to carry on a single haul N × STM-1 links, while STM-N streams are realized with fiber links. More details on the SDH principle and operation can be found in [2–4]. SDH is compatible with the synchronous optical network (SONET) and other international standards. In a single container more contributors from different native standards can be inserted, while in PDH the contributors have to be homogeneous. Thanks to the possibility to implement

Figure 1.13  Organization of an SDH network.

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rings, SDH permits the realization of highly reliable backhaul networks. The possibility to remotely monitor and control the network status is however the main advantage over PDH, and it also justifies the much lower efficiency of SDH if compared to PDH. In fact, the overhead occupies an important fraction of the band (5 Mbps in an STM-1) that cannot be used to carry user information. Moreover, the added circuitry for the management of pointers has a significant cost. 1.3.3 Asynchronous Transfer Mode The asynchronous transfer mode (ATM) is a packet switching protocol introduced in 1990 and used in UMTS networks. It offers three levels of services: •• Permanent virtual circuits that are similar to leased lines; •• Switched virtual circuits, where the logical link is active for the duration

of the call (usable for voice calls); •• Connectionless service (for data). ATM is connection‑oriented; in fact, its characteristics make it suitable for voice calls, thanks to low latency. The latter is in part ensured by the use of small containers, called cells. Every cell has 53 bytes, of which 5 are headers that contain the necessary information for routing the cell in the network. Another important factor for guaranteeing low delays is that the routing operations are performed via hardware, thanks to labels assigned to the cells that identify the path to be followed. The path is defined in an initial phase with a handshaking protocol. On the contrary, in Ethernet protocols, the packets have variable length and the routing is performed via software through routing tables that guide the packet in subnetworks. Practically, ATM uses multiple PDH links to exchange the information. Inverse multiplexing of ATM (IMA) defines how the cells are multiplexed to be cyclically sent over the different E1 or T1 links (see Figure 1.14). Additional cells are then necessary for instructing the demultiplexer, and dummy cells are also sent to guarantee a continuous data flow on the PDH link.

Figure 1.14  IMA round-robin functionality for transmission over multiple E1 PDH links.

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ATM is then more efficient than SDH, in particular for data transfer, but it does not perform network‑level flow or error control. For this reason, it requires a robust microwave radio link with high reliability characteristics. 1.3.4 Ethernet Protocol Ethernet protocol defines the layers 1 and 2 of the ISO/OSI stack. It is a packet switch protocol without a dedicated channel. Thanks to statistical multiplexing, Ethernet-based networks have the big advantage of using the channel resources in a very efficient way. The main issue with PDH, SDH, and ATM‑based networks is that their cost is practically proportional to their capacity. However, from the mobile operator’s perspective, it is important to note that the revenue does not increase proportionally with the capacity, since the latter is almost all related to data traffic. For this reason, since Ethernet has been successfully exploited in fixed data networks, backhaul designers are also using this technology to accommodate the capacity demand. Another reason for the success of Ethernet in fixed networks, and later in microwave networks, is that it easily supports TCP/IP. However, the big advantage of the Ethernet is also its main limit: by nature, it behaves completely opposite from a circuit-switched network, and thus it is not easy to create a reliable connection needed for priority traffic, as voice calls. The main challenge of Ethernet backhaul is to achieve carrier‑grade performance maintaining its flexibility and cost advantages. Carrier grade is a general term referring to high reliability networks, that in case of backhaul usually means 99.999% availability (i.e., less than 5 minutes of unavailability of the network per year). More details about availability are given in the next sections. A detailed description of the Ethernet protocol can be found in several sources [5,6]. In the following, we limit the description to the functionalities that can enable the development of a carrier‑grade Ethernet. A first consideration for radio point-to-point links is that the communication is in full duplex, so it is not necessary for the devices to implement collision detect techniques that increase the latency of the communication. TDM-links can be emulated in an ethernet network using the scheme of Figure 1.15. The Inter-Working Function (IWF) module provides the transition between the two worlds, dividing the TDM frames into packets at the input and reconstructing the frames at the output.

Figure 1.15  Emulation of TDM links over Ethernet.

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An important service for improving robustness and privacy is given by the Ethernet Virtual Connection (EVC). It connects two or more subscribers, and it prevents the transfer of data to parts that are not members of the EVC. EVC is the base for setting a Virtual Private Network (VPN). Ethernet has also many options for setting the priority of a virtual connection, and they can be exploited to efficiently use adaptive coding and modulation (ACM): •• Port-based queuing assigns a Class of Service (CoS) to every port. •• Tag‑based queuing adds some fields to the frame in order to identify the

assigned CoS, and it can be implemented at different ISO/OSI layers. For example, VLAN tagging tags differently voice and data calls, so they can be sent over different priority channels. •• Flow control stops the overload of the receiving buffers, slowing down the transmission, thanks to pause frames sent to the data source. •• Resource Reservation Protocol (RSVP) operates at the IP level, and it can be used to reserve buffers or bandwidth. By using some of these techniques, it becomes possible to maintain carrier‑grade connections for the core traffic when the ACM is used for lower throughput, sacrificing the other traffic. Of course, the management of priority requires overheads that lower the payload portion of the bandwidth. 1.3.5 Synchronization By definition, a synchronous network is a network where all the clocks run with the same long term accuracy. Synchronization refers, in this case, to the fact that the clock at the input and output points of a network must be aligned in order to avoid overflow or empty running of the output buffer. Thus, it does not refer to the synchronization of several contributors that must be multiplexed (an issue solved by stuffing, for example, in PDH), but to the overall synchronicity. However, also when a single clock is physically distributed throughout the network, it can be affected by jitter and noise, which makes it not aligned enough for correct timing of the frames. Synchronization is a major issue in backhaul networks, since the network performance is heavily affected by the synchronicity. In fact, a classical eye diagram aperture is maximum when the sampling is performed with the exact timing. However, if sampling tends to occur at the limit of the sampling period, the eye tends to close, and the SNR margin decreases. Moreover, if a buffer runs empty, the efficiency is decreased, while data is lost due to buffer overflow.

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In practice, in a network, the clock is distributed starting from very accurate references [primary reference clock, (PRC)]. PDH primary links (E1 or T1) are a very reliable physical mean to distribute the clock. In fact, at the aggregation point, a single clock, faster than the incoming clocks, is used to multiplex the contributors. Thanks to the good spectral symmetry guaranteed by scrambling, the clock extraction is very accurate. In SDH, external synchronization equipment is necessary to ensure that pointers could handle the needed phase correction [7–9]. In GSM, every MSC has its PRC. Alternatively, GPS signal can be used to synchronize cheaper clocks, but with the need to install extra antenna and receiver on site. Then the signal is transferred to the other sites that extract it from the aggregated SDH frames. After 20 network elements a narrowband filter is inserted to clean the clock signal from accumulated noise. Finally, every peripheral backhaul node has an internal clock reference, locked to the incoming signal. The internal reference is not stable enough to ensure the needed accuracy, but it is necessary to hold over the timing when the network clock is momentarily unavailable. ATM is an asynchronous network; thus, it does not require synchronization, and cell stuffing is used to avoid empty buffers. In an Ethernet network, the synchronicity is necessary substantially when a TDM communication must be implemented, at the IWF level (see Figure 1.15). There are several possibilities to synchronize the network. For example, a GPS signal can be used at the cost of an added antenna and GPS receiver. Alternatively, if the Ethernet network is regarded as the expansion of an existing PDH link, it could have some sense to maintain in place a E1/T1 link for clock distribution, that could also be exploited for priority traffic. Finally, there are methods to extract the clock directly from the packets, using differential or adaptive algorithms, in a way similar, but more complex and accurate, to what is done to synchronize personal computers in local area networks [10].

1.4 Link Dimensioning 1.4.1 Reliability Objectives When designing a network, the practical objective is to have an error occurrence exceeding a threshold only for a limited amount of time during a significantly long period, such as a year. The design of the single microwave link is related to the reliability objectives of the whole network. However, it is not easy to relate the single‑hop design to the overall network planning, also considering that a full knowledge of the network is seldom available. Practical rules are then applied for the dimensioning of the hops, according to the desired grade of the link. This grade will depend on the priority of the data the link has to carry. The ITU defines reliability standards for networks

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[11–13], but they refer to transnational reference paths, that usually result quite difficult to relate to the single‑hop design, and for this reason are not directly applied in practice. However, they could be of interest to better understand how the metrics for error measurement are defined. In the following, we give a qualitative overview of the reliability standards adopted for microwave link design. A comprehensive description of the reliability standards can be found in [2]. A first important consideration is about the difference between availability and performance. A link is considered unavailable if it either stops working or the error rate exceeds a threshold for more than 10 consecutive seconds. The availability is measured on annual base. However, the performance of the link is measured, when it is available, with several metrics, all indicating how often a certain bit error rate (BER) is higher than a given threshold (usually 10–3) for a determined time. The 10‑second limit is given by the fact that, for such a long outage, the network will have time to reroute, where possible, the communication on another link. For shorter outages, rerouting would not be possible or convenient. There are several possible causes for unavailability. Propagation in a real‑​ world link, as better described in the next subsection, is influenced by several factors, such as rain, diffraction, and ray bending that can cause outages longer than 10 seconds. The availability objectives due to rain and diffraction are usually set at 99.99% each, on an annual basis. Equipment failure is another cause of unavailability, and the annual occurrence depends on the mean time to failure of the radio and on the mean time to response for repair. For high-grade links, route diversity (like ring protection of SDH) or hot-standby configurations (see Chapter 3) are adopted to ensure the availability objective, while less important links can also be unprotected. Finally, other causes like natural catastrophes or long main supply failures can cause long outages, and in this case only route diversity can guarantee service delivery. The error performance is also influenced by several factors whose effects on the transmission quality last less than 10 seconds. The most important cause for error performance degradation is the multipath fading, discussed in detail in the following subsection. Other causes are the background errors in the equipment and the wind (that can tilt the antenna, lowering the fade margin). The metrics for error performance are quite complex, and they are based on the measurement of wrong data blocks. Usually, the dominant metric is the severely errored second (SES), defined by the Inter­national Telecommunication Union (ITU) as a 1-second period where more than 30% of blocks are with error [13]. In [13], formulas for the calculation of the objectives can be found, based on the link capacity and the hop length, and they will give a maximum SES/month that can be afforded, in the worst month.2 2.

Statistically, the month of the year when propagation conditions are worst.

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Industry tends to resume all the quality objectives in a single annual reliability figure, usually between 99.99% and 99.999%, and formulas and approximations are used to derive it from the availability and performance metrics. 1.4.2 Point-to-Point Propagation The basis for a PtP radio communication hop is simple: one radio and a high gain antenna at each end pointing directly towards the other form the complete communication channel (see Figure 1.16). However, as the properties of the radio path are heavily influenced by meteorological conditions such as humidity, temperature, and barometric pressure, there could be large variations to the conditions for the propagating radio waves when it comes to attenuation and diffraction. The frequency of use is also a key parameter to the behavior of the channel, as there is a substantial difference for frequencies in the lower end of the typical PtP radio spectrum starting around 4 GHz, extending up to the highest band in use today at 86 GHz. To further add complications, the communication is also affected by the terrain surrounding the direct path since it causes reflections. This creates multipath propagation, which at the receiving end yields constructive or destructive interferences of the reflected waves on the directly propagating signal (multipath fading). The attenuation in the channel is the sum of several contributions where the free space loss (FSL) is the main contributor. The free‑space loss (in decibels) is defined as:  4p d FSL = 20log   l

 (1.1)   where d is the hop length, and λ is the wavelength in use. As can be seen from Figure 1.17, the FSL is approximately in the range of 100dB up to 160dB for reasonable hop lengths and standard frequencies in use for PtP. To this, the attenuation due to water, water vapor, and other atmospheric gases needs to be added. Specifically, it should be noted that water and oxygen have resonances at certain frequencies at which the attenuation is significantly higher



1

2

Figure 1.16  A line-of-sight point-to-point radio system.

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Figure 1.17  FSL as a function of hop distance, for different link frequencies.

than average. Figure 1.18 shows the attenuation in dB/km as a function of frequency and for some specific rainfall intensities. The above‑mentioned resonances at approximately 60, 118, 181, and 325 GHz can also be clearly seen. These frequencies are not in use for ordinary high‑quality wireless backhaul systems. Furthermore, as can be seen in Figure 1.16, the curvature of the Earth must not be forgotten when planning a radio link hop, and in particular when deciding the location and the height of the antenna. Even though the distance between two antennas might look very small compared to the Earth radius and would therefore be insignificant to the propagation, it is not. To calculate the curvature of the Earth between two antennas, the following approximate equation can be used (see Figure 1.16):

Figure 1.18  A  tmospheric attenuation for propagating wave versus frequency and rain intensity.

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H =



D1D 2 (1.2) 12.75 k

where D1 and D2 (in kilometers) are the distances from each antenna to the point where H (in meters) is defined, that is, i.e., the sum (D1 + D2) is the total distance at sea level between antennas D, and k is the effective Earth radius factor used to compensate for the diffraction of the microwave beam when propagating through the atmosphere. The value of k for different locations can be found in data given in ITU-R. However, it should be noted that the k is not constant but varies with temperature and humidity and throughout the day, so that each location must ideally be characterized on a per‑hop basis. Standard k-factors vary approximately from 0.5 to 1.5. It should be noted that, after having made the calculation of the Earth’s curvature, this is obviously not sufficient for the tower calculation, as this is only line of sight. For calculating tower height, additional clearance must be made for possible obstacles and surface roughness for having the first Fresnel zone free (see Figure 1.16). Small k-factors will make the corrected curvature very large, as can be depicted from (1.2), and will thus require very high towers even for moderate hop distances. Once the characteristics of the channel are known, other contributing factors from the equipment hardware are added to calculate the possible hop distance. Such contributors are the receiver sensitivity, transmitter output power, and the gains of the two antennas. The transmitter output power is usually in the range of 20 to 30 dBm (100 mW to 1W) at the waveguide flange on the outside of the radio enclosure in the case of mast-mounted equipment or split or alloutdoor architecture. Loss from the feeder waveguide to the antenna reduces the power slightly but not significantly, as this usually is a very short distance. In the case of all indoor equipment (long haul architecture) the output power is defined at the output of the transmitter, but before the branching filter arrangement, so that the actual power at the antenna is reduced by the branching filter losses and the usually quite long feeder waveguide from ground level to the antenna.3 The receiver is defined by the receiver sensitivity level (RSL) measured at the waveguide flange as in the case of the transmitter. The sensitivity is defined as the input level required for a certain BER (typically 1E-6) given the channel arrangement, bandwidth, and modulation. Also in this case the levels are defined in, for example, the ETSI EN 302217-2-2 as classes dependent on spectrum efficiency. Typical RSL ranges from –55 to –75 dBm.

3. The definition of interfaces and levels can be found in equipment standards such as from the European Telecommunications Standards Institute (ETSI), EN302217-2-2 [14].

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High gain antennas are typical of point-to-point communication equipment. The typical antenna for a short haul system (split or all-outdoor) ranges from 0.2 m to 1.8 m in diameter. For a long haul system, it is more characteristic to have larger antennas starting from 0.6 m going all the way up to 4.7 m in diameter for low-frequency and long range installations. As the antenna gain increases proportional to the square of the frequency, antennas at the higherfrequency bands tend to be smaller than at the lower frequencies. Other important parameters for the antenna are the sidelobe level, front-to-back ratio impor­ tant for interference calculation, and, in case of dual-polarized antennas, the isolation between the two polarizations. Further details of antennas are described in Chapter 7. Once having the understanding and facts for the channel and the equipment, the path calculation can be made. The basic objective is to select antennas and output power in order to guarantee an input level to the receiver that is sufficiently high for having bit error-free transmission. This is usually defined as having a fade margin in clear sky conditions to allow for different types of fading due to weather conditions and yearly variations. The fade margin is defined as:

Fade margin = RSL Clear Sky − RSLThreshold



(1.3)

and it is usually in the range of 25 to 30dB. A too-small fade margin will result in poor availability for the hop-meaning frequent outages, and a too-large fade margin will add cost for larger antennas, stiffer towers to allow more wind load, higher power consumption due to higher output power requirement, and other factors. As a consequence, the selection of the fade margin derives from a delicate balance. The planning is normally done using statistics and models stated in the ITU-R for different locations throughout the world. The statistics is implemented in various planning tools that can be found on the market and be used as the basis for hop calculations, selection of antennas, setting of output power, and finally calculation of the availability. 1.4.3 Fading Several phenomena contribute to the general term fading: precipitation in the form of rain or snow and sandstorms or gases in the atmosphere can attenuate the propagating wave severely, as previously shown. Reflections from ground or obstacles can cause a second beam to arrive with a slight delay to the receiver, where the two signals will be added. Depending on the phase at arrival, the several incident beams will add in a constructive or destructive way, see Figure 1.19. Concerning fading due to precipitation, ITU have published statistical data and models to calculate the attenuation at different locations [15]. The Earth, in the

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Figure 1.19  Multipath propagation.

older ITU recommendations [16], was divided into rain zones; see an example in Figure 1.20, and statistical data was given for each zone for planning purposes. Each rain zone has statistical data given for the precipitation as intensity in millimeter per hour. The most common value to use in planning is the occurrence 0.01%/year. Table 1.1 reports the exceeded intensity occurring at 0.01%/ year in the rain zones [16]. For example, in San Diego, California, zone E, the per-year probability to incur on rain intensity higher than 22 mm/h is 0.01%. More recent ITU recommendations [15] give more detailed information on precise locations, allowing a more refined calculation. When planning longer hops, it is also assumed in the statistics that the rain is not continuous for the complete path and thus a correction factor is introduced in the model. It should also be

Figure 1.20  Example of rain zones: North America.

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Introduction Table 1.1 Rainfall intensity in mm/h exceeded for 0.01%/year A

B

C

D

E

F

G

H

J

K

L

M

N

P

Q

8

12

15

19

22

28

30

32

35

42

60

63

95

145

115

noted that there is a significant difference in attenuation due to rain for vertical versus horizontal polarization. In fact, as the raindrops tend to be flat when falling, they will affect the horizontally polarized wave more than the vertically polarized one, and thus higher attenuation for the horizontal wave is shown. Going back to multipath fading, this phenomenon is more pronounced for longer hops that usually use lower frequencies. Long hops mean a very small angle between the direct beam and the reflected beam, so that both beams will fall into the main lobe of the antenna: there will be no selectivity help from the antenna directivity. The delay for the reflected signal compared to the direct path is in the order of 0.5 ns to 5 ns for longer hops, and depending of the phase when added together, the resulting signal can look like the one shown in Figure 1.21, where the original spectrum of a typical QAM-signal is shown beside the distorted signal after multipath. The characteristic notch seen in the spectrum is dependent on the delay difference and the amplitude difference. The receiver equalizer will have to cope with some certain notch depths and also with a notch rapidly moving through the spectrum, which can occur during changing conditions. The minimum requirement for the ability of the receiver to correct for certain notch depths and speed for different modulation classes is defined, for example, in ETSI EN302217-2-1 [17]. 1.4.4 Interference Interference is unavoidable in a microwave radio network, especially in dense networks where a large number of wireless backhaul nodes are deployed. The planning of frequency use is extremely important to not exceed the maximum levels of interfering signals into neighboring receivers in the same frequency band. The receiver interference robustness is specified, for example, in the ETSI EN302217-x-x for maximum levels in adjacent and second adjacent channel. The applied fade margin plays an important role for the overall network planning, since unnecessary high fade margins implies high output power, which will increase the level of interference to surrounding radios. The use of antennas having low sidelobe levels or high front-to-back ratio will improve the situation because the transmit levels in directions outside the main lobe will be significantly reduced. The use of alternate polarizations is another means of reducing interference levels. However, care must be taken as polarization is also a way to improve capacity in a given frequency channel.

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Figure 1.21  Spectrum notch due to multipath propagation.

1.5 Description of Chapters In the following chapters, the different aspects of the electronics for backhaul will be described. In Chapter 2 the advantages and disadvantages of wired versus wireless solutions are highlighted. Then the main milestones of the microwave history from the pioneering era to today are reviewed, envisaging the probable future trends of the wireless backhaul. Chapter 3 will summarize the possible architecture installations and the main building blocks of a backhaul radio system. Chapter 4 to 7 will be focused on these blocks, more specifically Chapter 4 on the digital signal processing operated in the modem, Chapter 5 on the receiver, Chapter 6 on the transmitter, and Chapter 7 on the antenna. The chapters are

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structured to briefly introduce the necessary theoretical background and then to show and comment in details modern implementations and real-world examples.

References   [1] Salmelin, J. and E. Metsala, Mobile Backhaul, New York, Wiley, 2012.   [2] Manning, T., Microwave Radio Transmission Design Guide, 2nd ed., Norwood, MA.  [3] ITU, G.803: Architecture of Transport Networks Based on the Synchronous Digital Hierarchy (SDH), ITU, 2000.  [4] ITU, G.783: Characteristics of Synchronous Digital Hierarchy (SDH) Equipment Functional Blocks, ITU, 2006.   [5] Norris, M., Gigabit Ethernet Technology and Applications, Norwood, MA, Artech House, 2002.   [6] Krzanowski, R., Metro Ethernet Services for LTE Backhaul, Norwood, MA, Artech House, 2013.  [7] ITU, G.811: Timing Characteristics of Primary Reference Clocks, 1997.  [8] ITU, G.812: Timing Requirements of Slave Clocks Suitable for Use as Node Clocks in Synchronization Networks, ITU, 2004.  [9] ITU, G.813: Timing Characteristics of SDH Equipment Slave Clocks (SEC), ITU, 2003. [10] ITU-T, G.8262: Timing Characteristics of a Synchronous Ethernet Equipment Slave Clock, 2015. [11] ITU-R, F.1703-0: Availability Objectives for Real Digital Fixed Wireless Links Used in 27,500 km Hypothetical Reference Paths and Connections, ITU, 2005. [12] ITU-T, G.827: Availability Performance Parameters and Objectives for End-To-End International Constant Bit-Rate Digital Paths, ITU, 2003. [13] ITU-T, G.828: Error Performance Parameters and Objectives for International, Constant Bit Rate Synchronous Digital Paths, ITU, 2000. [14] ETSI, ETS1 EN 302 217-2-2 - Fixed Radio Systems; Characteristics and Requirements for Point-To-Point Equipment and Antennas; Part 2-2: Digital Systems Operating in Frequency Bands Where Frequency Co-Ordination is Applied; Harmonized EN Covering the Essential Requirements of Article 3.2 of the R&TTE Direct, 2012. [15] ITU-R, P.837-6: Characteristics of Precipitation for Propagation Modeling, 2012. [16] ITU-R, P.837-I: Characteristics of Precipitation for Propagation Modelling, 1994. [17] ETSI, ETSI EN 302 217-2-1 - Fixed Radio Systems; Characteristics and Requirements for Point-to-Point Equipment and Antennas; Part 2-1: System-Dependent Requirements for Digital Systems Operating in Frequency Bands Where Frequency Co-Ordination is Applied, ETSI, 2004.

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2 Evolution of Microwave Radios Maurizio Pagani 2.1 Introduction The communication infrastructure has continuously evolved since its introduction more than a century ago, driven by customer and supplier needs, always trying to answer the increasing capacity demand by adapting the hardware capabilities of the moment. An ideal infrastructure has unlimited capacity, is flexible and keen to upgrade when necessary, safely transfers data also in presence of interference (accidental or intentional), it is robust to natural event, and, last but not least, it is inexpensive to install and maintain. The possible implementations are wired or wireless, both presenting advantages and disadvantages. In this chapter the pros and cons of the two solutions are briefly introduced. Then, focusing on the radio link, the history of microwaves from the pioneering era to tomorrow is analyzed, envisaging the probable future trends of the wireless backhaul.

2.2 Microwave versus Wired In a typical telecommunication network, the objective is to transport information, which may be voice or data, from the customer premises or a user terminal through a switch (i.e., a network node) and to the end-user location. The process of sending and receiving information in a communication network can be basically accomplished by two different transmission media: wired and wireless. These two different options have their own advantages and disadvantages and are both commonly employed in modern telecommunication networks. Wired 29

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lines can be made by copper cables or optical fibers, while wireless transmission systems usually utilize radio frequency bands higher than 4 GHz: these systems are commonly called microwave radio links. While traditional microwave radio systems are covering the frequency bands up to 30 GHz, when communication systems utilize frequency bands between 30 and 300 GHz they are better referred to as millimeter-wave systems. Over the years, the needs in telecommunication networks have never stopped increasing. To support this growth, both wired and wireless technologies have evolved in terms of capacity and functionality. This technology progress allowed telecom equipment manufacturers to design, develop, and provide products with continuously improved performance. In such a way, always better fixed and mobile voice and data applications have been supported along with associated new transport protocols. A challenge for engineers that is valid still today is that they need to provide the data capacity required for an exponential bandwidth explosion coming from data services, without equivalent additional subscriber revenue. The user traffic can be combined to be transported over the network in different ways, such as with time division multiplexing (TDM) systems, asynchronous transfer mode (ATM), or Ethernet, they all have pros and cons depending on the type of traffic they are asked to transport (see Chapter 1). Using fibers has many positive aspects: they are reliable against weather conditions, and provide high transmission capacity with excellent service and speed. However, deploying fibers struggles with installation time, maintenance, cause disruption and cost which in the end affects the users. Small, medium, and high‑capacity digital microwave radio links have been developed and used for many years providing telecom operators with flexible, high-quality, and costeffective network connectivity that has allowed them to design network solutions to meet their short- and long-term requirements. The requirement to backhaul mobile radio traffic quickly and reliably led to exponential growth of the microwave radio link industry. It has been often predicted that fiber optic transmission should stunt this growth. However, considering the difficulty to provide last-mile fiber connections to the increasing number of base stations, microwave radio will continue to be an effective solution for the access network and, in some cases, also for the core network. Figure 2.1 shows the expected trend in the technology share for the backhaul. As the access networks grow, fiber is also being pushed deeper into the network requiring last-mile connections, which will further fuel growth of microwave technology. The introduction of the synchronous digital hierarchy (SDH) technology in transmission networks represented a new interesting challenge for microwave radio systems to allow successful integration into advanced synchronous networks. Fiber optics is the preferred medium for long-haul high capacity transmission, but since the introduction of SDH technology, microwave radio has been strongly required by network planners to complement the synchronous

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Figure 2.1  Backhaul technology share: foreseen trend up to 2019 (Ericsson 2014).

Figure 2.2  Q  ualitative cost versus capacity comparison between backhaul technologies (Mobile Backhaul: Fiber vs. Microwave. Ceragon 2009).

fiber network. This is mainly due to its intrinsic benefits in terms of security, speed of deployment, and economics [1,2]. The present trend in terms of transport technology is to shift to a full Ethernet-​ based backhaul that, as depicted in Figure 2.2, is particularly favorable for micro­ wave backhaul since it permits one to increase capacity and lower costs. 2.2.1 Benefits and Disadvantages of Microwave Microwave radio technology has some benefits compared to wired lines in terms of scalability, security, robustness to natural calamities, and cost. In fact, modular solutions from end sites and single hops up to high-capacity aggregation nodes can be realized. Radio link capacity up to 1 Gbps over one hop in one frequency channel is possible adopting cross-polarization with single antenna systems by means of cross polarization interference cancellation (XPIC) (see Chapter 4).

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In the event of earthquakes, floods, or other natural calamities, microwave links are less affected than wired link and less prone to accidental damage. Furthermore, they have a quicker recovery time and for this reason microwave links are often considered as backup solutions to wired cable and fiber optic trunk links. A microwave link can also be quickly established on a temporary structure and being immediately operative on a path with adequate line-of-sight clearance in the event of a natural disaster or humanitarian emergency. The economic investments are more secured by microwave link technology because a smooth migration towards more advanced network solutions is possible. High-capacity links across mountains, rivers, lakes, and seas, when possible, are more economically feasible than wired lines. In cellular networks, backhaul is one of a few primary mobile data connectivity cost drivers and, at the same time, it has become the primary network bottleneck for most mobile operators. When considering backhaul network improvement, one metric to be evaluated by mobile operators is scalability. In fact, to satisfy the always growing capacity and traffic request driven by mobile broadband services, the comparatively poor scalability of leased line approaches is a major issue. The solution of choice should be able to effectively offer scalable capacities to successfully fit changing scenarios that are strongly characterized by differentiated types of traffic and driven by an increasing number of mobile users. As a result, microwave technology can be considered as the preferred solution to this bottleneck providing mobile operators with a compelling economic and performance alternative for mobile network backhaul in every geography, and significantly reducing total cost of ownership (TCO). As for any technological solution, and also for microwave links to the aforementioned advantages, some disadvantages are related, the most important of which is surely the lower capacity in comparison to optical fibers links. Moreover, it is necessary to highlight also some technology issues that should be carefully considered by telecommunication network planners. Microwave radio communication requires a clear line-of-sight (LOS) condition that under normal atmospheric conditions means 30% beyond the optical horizon. Frequencies ranging from 4 to 15 GHz are normally used for highcapacity long-haul transmission while higher frequencies are essentially used for short-haul communication. Millimeter waves have recently become very popular for providing extremely high capacity in a short to medium-distance range. In areas with a lot of rain, high-frequency links are more affected by strong propagation changes with increased path loss attenuation.1 Microwave hops in the proximity of large water surfaces can experience severe multipath fading, and special equipment configurations should be adopted

1.

The issues can be mitigated by using the lowest frequency band allowed for that specific case.

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33

to keep the required quality of service, increasing the cost for the operator. Reflec­ tions may be avoided by carefully selecting the sites where to install the antennas that can provide a better shielding from the reflected rays. The most critical paths are close to the coasts with hot and humid weather.

2.3 The Historical Evolution of Radio Links 2.3.1 Microwave Pioneers For more than a century radio link systems working at microwave frequencies have been considered as viable solutions for building communications infrastructure [3]. This represents a great history of technological advances, shifting market requirements, and entrepreneurship all over the world. The term “microwaves” conventionally covers that portion of the spectrum spanning from 300 MHz to 30 GHz, thus exhibiting wavelengths from 1 meter down to 1 centimeter. At the very beginning of microwave history, in early published works, these types of electromagnetic waves were referred as “short waves” and “quite short waves” [4]. The origin of the term “microwaves” can be found in a paper [5] published in 1931 by the International Telephone and Telegraph (ITT) describing an 18cm wavelength radio system called micro-ray whose electromagnetic radiation was called micro-wave. Subsequently, in 1932 the term “microwaves” was first introduced in a published paper [6] and then commonly used by the scientific literature. Thanks to the successful progresses in the research activities, microwaves have become today a ubiquitous technology that is used not only for communications and radar, but also for the characterization and analysis of materials, cooking and industrial drying, medical diagnosis and treatment, radio astronomy and transmission of power.2 The roots of modern microwave radio links can be perceived in the first experiments carried out by Guglielmo Marconi when, in 1894, he set up his first laboratory at Villa Grifone, about 14 km from his native city of Bologna. There Marconi used very high frequencies, practically in the field of microwaves, for his experiments on wireless communication, and first demonstrated the feasibility of a practical radio link with a much lower frequency [7]. Many scientists before Marconi had devoted their work to study the electric and magnetic phenomena early described by Maxwell with his famous equations: Heinrich R. Hertz, Sir Oliver J. Lodge, Augusto Righi [4,8]. From 1885 to 2. Even if, for frequency exceeding a few gigahertz, wireless power transfer is still at the research level.

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1897, Heinrich Hertz performed many experiments that shown how Maxwell’s equations indeed described the propagation of electromagnetic waves through the atmosphere and space. He developed equipment to generate, radiate and detect microwaves [9]. In 1895, Jagadish Chandra Bose demonstrated in Presidency College, Calcutta, India, transmission and reception of electromagnetic waves at 60 GHz by remotely ringing a bell and detonating some gunpowder. All of these scientists worked with very high frequency to study the kind of propagation of electromagnetic waves, that they called quasi-optical waves, but never demonstrated practical applications. In 1896, encouraged by Sir William Preece of the British Post Office, ­Guglielmo Marconi gave one of the first demonstrations of communication by means of free-space propagation of microwaves. He transmitted telegraphy signals over a 2-mile path using a spark gap generator to produce the microwave signal. The frequency he used was on the order of 1 GHz [10]. Marconi’s basic contribution, for which he deserves the name of inventor of the radio and was awarded a Nobel Prize in 1909, was, first of all, that he modulated by a signal the electromagnetic waves that a spark produced in a Hertz-type oscillator [11]. In his following experiments Marconi tried to increase as much as possible the distance at which the communication was possible, especially between ships. He worked with low, medium, and high frequency, breaking the 100-km barrier of link distance in 1899 and by crossing the Atlantic Ocean in 1901 [4]. Marconi’s predominant interest was not in purely scientific knowledge, but in practical appli­ cation for useful purposes [12]. He was the earliest to recognize and advance the communication technology using electromagnetic propagation in free space and in doing so established the base for the development not only of microwave radio, but also of radar, television and radio broadcasting, long-distance wireless telegraph and telephony, satellite communication links, and wireless access systems. In early 1900s, Guglielmo Marconi developed and commercialized the first transatlantic wireless telegraph communication systems and since then the wireless industry has expanded from point-to-point technologies, to radio broadcast systems, and finally to wireless networks. 2.3.2 Early Years from the 1920s to the 1940s World War I gave a strong push to the investigation of higher frequencies. In order to maintain secrecy in military communications and minimize interceptions, high-directivity point-to-point links were needed. As well known, this requires high gain reflector antennas, which were unfeasible at large wavelengths [7]. In 1919 the Barkausen-Kurz tube was invented with the capability to produce an electromagnetic wave up to 10 GHz (see a simplified schematic in Figure 2.3). Barkhausen and Kurz discovered, whether by accident or design, that, when they

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Figure 2.3  Barkhausen-Kurz tube oscillator.

applied a negative potential to the anode of a triode valve and a high positive potential to the grid, very high frequency oscillations were set up [8]. Hence, experiments with higher frequencies started to be possible. In 1928 the Presidency of Italian Royal Academy was conferred upon Guglielmo Marconi. This made necessary for him to transfer his research activities and staff from England to Italy, although his work continued to be sponsored by British interests. After over 30 years of investigation, during which period he worked on low, medium, and high frequencies, he finally returned to his original band of frequencies, experimenting again on microwave links [10]. In 1931 Marconi started working on what can be called the first true micro­ wave radio link. By using a 600-MHz frequency signal (50 cm of wavelength) that he generated with a Barkhausen tube, he was able to establish a stable radio link over a distance in the order of 35 km, from Santa Margherita Ligure to Levanto, in the Tigullio Gulf in Italy [5]. With that experiment, he was able to show that useful commercial applications based on Point to Point radio links could be achieved over reasonable ranges with pretty low-power transmitters and parabolic reflectors. Of course, Guglielmo Marconi was not alone in this field. In the same year, Clavier and Gallant of ITT built an 18-cm wavelength radio link from Calais to Dover, across the English Channels which separates England from France. This system also used a Barkhausen tube as a signal source and was called the microray. The radiation was called by them “micro-wave” and this term was used, as mentioned above, to name this portion of the spectrum as “microwaves” [5]. Also Uda, Japan, succeeded in establishing a two-way communication link between Otakamori and Sendai, covering a distance of 30 km, operating on frequencies around 600 MHz. All workers in the microwave field in the early 1930s used the humble apparatus. This fact explains the absence of precise measurements. There were literally no radio-frequency devices available for the measurement of power, signal

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intensity, attenuation, or indeed frequency itself, and their pioneering work was carried out under severe difficulties. Nevertheless, Guglielmo Marconi was able to establish in 1932 the world’s first commercial point-to-point terrestrial microwave link between the Vatican City and the summer residence of the Pope at Castel Gandolfo (Italy). This installation was the first microwave telephone in the world and was put into regular service in February 1933 [13]. The transmitting station was placed inside the Vatican Garden Palace, next to St. Peter’s Cathedral. A parabolic reflector antenna was placed on the roof [7]. The transmitting unit was developed with two valves operating in a push-pull configuration. One difficulty was that not every valve could produce the Barkhausen-Kurz continuous-wave oscillation with a sufficient life to be operated in a real transmitter commercial application. Some valves could be working as an oscillator for a few minutes of life and then terminate operating because the grid had melted. Marconi and his team during their experiments were able to develop a reliable valve that had a life of about 40 hours and an output power of about 5 W at a frequency of 600 MHz [10]. The aerial dipole antenna was directly coupled to the grid circuit being capacitanceloaded by circular discs. Four of these transmitting units were used in parallel and kept in phase by an interlinking wire system. They were placed at the focus of a parabolic cylinder reflector. Frequency modulation of the transmitter was obtained by acting on the anode negative potential of the triode valves connected in the Barkhausen-Kurz manner [10]. The high-sensitivity receiver was developed with a radio-frequency detector consisting of a pair of push-pull valves connected in the Barkhausen-Kurz manner in a regenerative condition, followed by a two-stage audio-frequency amplifier [10], The first tests of this system, which was very similar to the one used in the Tigullio Gulf ’s experiments, were carried out establishing a clear line of sight radio link between the Vatican City and Villa Mondragone, in Monte Porzio Catone, close to Rome in Italy, with very good results. A second set of more difficult tests was carried out with the system in his final configuration. The receiver was placed in Castel Gandolfo, at around 20 km of distance, with microwaves propagating happened over a hilly landscape [7]. The first official transmission was done in the presence of Pope Pius XI on April 26, 1932 [14]. The war efforts of World War II allowed a lot of advances in microwave technology. Radar played a decisive role in the war and the efforts on the development of radar were very concentrated and led to major advances in development of microwave technology. The magnetron (see a sketch in Figure 2.4), already invented in 1920 by Hull to produce microwave signals, was greatly developed and improved to increase the frequency of operation. The traveling-wave tube (TWT) and the klystron tube were invented to produce higher-frequency micro­ wave signals.

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Figure 2.4  Sketch of a magnetron.

2.3.3 The Microwave Industry In the years after World War II, the technology developed during the preceding years spread rapidly. New companies selling microwave components, systems, and test equipment were created and many research programs in universities, research institutes, and large companies were funded almost all over the world by national governments, mainly to support new weapon systems developed for the Cold War. All these efforts in microwave technology allowed the development of microwave communications systems, which reached critical mass. This was mainly due to the rapid growth, in many areas of the electronic industry, of a favorable entrepreneurial climate. Thanks to the simultaneous evolution of microwave tube manufacturing companies into the commercial market, several new products with good performance were made available. The following decades saw a rapid industrial growth and many technological advances in terrestrial and satellite microwave communication system development. Some theoretical scientific concepts in the microwave field were transferred to the realization of systems that have profoundly impacted society. These developments have been driven by new commercial applications and the major technical contributions have been those enabling the evolution of the microwave technology market with new industrial applications [4]. One example of those kind of companies that placed a major role in the growth of microwave radio link technology is represented by the Italian Telettra. This company whose name, Telettra, wanted to integrate in one word the union of the words Telefonia (Telephony), Elettronica (electronics), and Radio (radio), was founded in Milan by Virgilio Floriani and Libero Confalonieri in 1946, with a team of 11 people all coming from a previous electronic company of military communication (SAFAR-Industria elettronica di comunicazioni militari). Thanks to a continuous focus on innovation and the high level of its research, together with a strong motivation and spirit of belonging of its employees, this small pioneering company was able to grow during the following decades and develop some of the most advanced microwave equipment (see Figure 2.5) in

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Figure 2.5  The 1967 Telettra the first in the world digital radio link, (13 GHz) [15]. (Courtesy of Dr. Guido Vannucchi.)

the world and establish several world records in microwave communication, as the longest hop for a terrestrial microwave link ever realized and commercially operated (see Figure 2.6). In 1990, after reaching employment of more than 9000 employees, the company was bought by the French giant Alcatel. It has been calculated that from the experience generated by Telettra company during almost 50 years of operation more than 65 different companies have been originated and among them is the big semiconductor company SGS, now part of STMicroelectronics. The first microwave radio link was developed and operated by Telettra in 1949 [15]. In the same years in the United States, particularly in the San Francisco Bay area, companies like AT&T and Lenkurt Electric, which in 1959 became a subsidiary of General Telephone & Electronics (GTE), rapidly grew and established themselves among the most competitive suppliers of systems for use in telecommunication infrastructures. They demonstrated the spirit and successful practice of entrepreneurship that greatly facilitated effective adaptation to rapidly changing market conditions. In 1947 AT&T, which already had developed a 4.5-GHz multichannel micro­ wave radio for the U.S. and U.K. armies during World War II, built a 3.7–4.2-GHz microwave line-of-sight radio system that was based on repeaters. This system was probably the first example of microwave system used as backbone for civil applications and was able to carry 500 voice channels and one black-and-white TV channel from New York to Boston.

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Figure 2.6  1979: Telettra world record radio link with 360 km across the Red Sea [15]. (Courtesy of Dr. Guido Vannucchi.)

During the same years, the 2-GHz pulse position modulated (PPM) “digital” radio had a 24 × 300 baud data channel capacity from General Electrics (GE). This radar-like PPM was used during the World War II and then declassified. Many similar GE and ITT PPM radio hops were deployed in long pipeline, power, and turnpike systems in the 1940s to the 1950s, some up to 75 hops in length, all over the United States and worldwide for the military [21]. The first commercial microwave radio link was deployed in the 1950s. In September 1951, AT&T completed the 107 station 4-GHz Bell Telephone Skyway system and started serving the United States to carry telephone calls and ­television programs from New York to San Francisco. This microwave system running coast-to-coast for over 4000 miles was continually upgraded as micro­ wave technology progressed during the following years. New channels were added in 1953 with an additional 6-GHz system to better support the growing voice and TV services. The performance of analog hops was far more affected than later generation digital radio hops to equipment nonlinearities, interference, thermal noise, multipath distortion, waveguide echoes and molding, and fading. These primitive systems were used for many years, until the backbone technology was changed to optical fiber systems, with the capability to transport much larger traffic [16].

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2.3.4 From Analog to Digital Microwave Radio It has been more than 50 years since microwave technologies developed. The milestone technological advances in this market segment can be summarized as “transistorization” in the 1960s, digital systems in the 1970s, synthesized frequency tuning in the 1980s, and digital signal processing with programmable digital modulation in the 1990s. These technological progresses were accompanied by a continuous development trend toward higher spectral efficiencies, more robust architectures, and the use of even higher-frequency bands. Before 1980, analog microwave systems had been playing a predominant role in communication. Since 1990, digital microwave technologies have been developing rapidly. Apart from the progress of technologies, the characteristic of digital signal of keeping a good signal-to-noise ratio (SNR) has been the key factor ensuring its success and today’s long-haul transmission capability. Traditionally, the microwave radio was used in high-capacity trunk routes by public telecom operators. The primitive radios used analog modulation techniques and, therefore, were classified as analog radios. The analog microwave and coaxial cable carrier transmission system were the two major methods used in the early stage for long haul transmission. Long-haul (up to 25 miles), high data rate, point-to-point radios were largely used in several part of the world for deploying long-distance communications networks. Over the years, microwave radio link networks evolved from primitive systems carrying few simple TV channels and little telephone traffic to more advanced packets solutions with complex data, voice, and video traffic. This transition required continuously increasing bandwidth capacity to accommodate always larger data traffic along with new functionality to maintain the delivery of high quality services to the end user. Canadian Marconi delivered the first pulse code modulated (PCM) digital radios to private microwave users in North America in 1970, some hops remained in service into the millennium, thus triggering the rapid development and deployment of higher capacity (78 Mbps, then 90 Mbps) digital radios for LOS radio-relay hops. This culminated in 1980 when the alarm/network management systems and adaptive equalization were introduced. These trailblazing digital radios were often totally inadequate to accommodate the fragile, bursty characteristics of many high-capacity digital microwave radios. Moreover, these systems exhibited spectral distortion caused by dispersive fading in hops not before seen in frequency modulation (FM)-frequency division multiplexing (FDM) analog radios [21]. As a consequence, in the 1980s these were gradually replaced by fiber optic systems that offered far higher bandwidth. The microwave link industry successfully adapted to the competitive large-scale introduction of fiber optic transmission systems by introducing new generations of digital radio systems for local access applications and mobile infrastructures. Radio networks are more secure than fibers. Fibers are in fact distributed by nature, meaning

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that the whole path is exposed to intentional or unintentional damaging, and their repairing can take a long time. However, the radio link consists of two towers whose location is well known to maintenance crew, and the intentional deflection of the beam is improbable, given the high directivity of the antenna. The continuous evolution of the market scenario has demanded a tremendous development of the transceiver technology, making possible the achievement of extremely high levels of electrical performance, yield, and reliability, while increasing the operating frequency and reducing the cost and form factor. Starting from early 1980s, digital radios (referring to digital modulation techniques) were developed and deployed at 18, 23, 26, and 38 GHz carrier frequencies to provide short-haul communications (10 miles or less) for private networks, and for the interconnection of mobile telephone cell sites. Digital radios offered improved transmission quality compared to analog radios due to their inherent noise immunity. Digital microwave radio systems have for many years provided telecom operators with flexible, high-quality, and cost-effective telecommunications network connectivity. The 1980s thus brought about dramatic improvements in digital microwave modulation efficiencies and, with new adaptive equalization and powerful error correction, robustness to the dispersive (spectrum distorting) fade activity that so degraded digital radio hop performance in the 1970s. The mid-1990s heralded digital signal processor (DSP) equalizers that replaced discrete devices in far more robust advanced asynchronous plesiochronous digital hierarchy (PDH), and 2016/1890 Synchronous Optic Network/ Synchronous Digital Hierarchy (SONET/SDH) point-to-point time-division multiplexing (TDM) digital radios. The Federal Communications Commission (FCC) decided the relocation of analog microwave hops from 2 GHz in the late 1990s to accommodate cellular deployment accelerated this digital migration. These new PDH and SDH digital technologies supported the explosive birth of new high-performance terrestrial fixed wireless systems and fixed wireless access networks in all of their forms (e.g., point-to-point and point-to-multipoint) in synergism with fiber optics and free-space optical (FSO) networks. The requirement to backhaul mobile radio traffic quickly and reliably led to exponential growth of this industry before the turn of the twenty-first century. Due to the increasing competitiveness of the telecom market that was driven by demand of new system development consistently characterized by greater versatility, user friendliness, and cost competitiveness, there was a mix of in-house manufacturing of key active microwave integrated circuit components and of procurement from specialized suppliers. Outsourcing and off-shoring started later and increased over time. The thrust of the research supporting microwave technology has been in establishing the basic elements to sustain the growth and development of the applications and also to extend the usable spectrum to higher and higher frequencies including the millimeter and submillimeter wavelengths and eventually to the optical wavelengths [4].

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The earliest TV program transmission among cities adopted the microwave transmission channels. The small and medium-capacity digital microwave equipment (8.34 Mbps) developed in 1970s began a new era for the digitalization of microwave. In late 1980s, the successful development of SDH digital microwave leads to the emerging of the N × STM-l (see Chapter 1) large-capacity digital micro­wave system. Speaking of analog microwaves, they were no longer used to construct networks in the end of 1980s, and now they are used only in mountain stations owned by the State Administration of Radio, Film and Television of China.

2.4 Microwave Links Nowadays Within the mobile backhaul arena, the majority of cell sites around the globe have been connected within the access domain by point-to-point digital microwave radio systems. As advanced mobile broadband and Voice over IP (VoIP) technologies were deployed, there was no reason why the usage of microwave radio systems could not continue. For many years, microwave bands between 6 and 40 GHz have been used for point-to-point backhauling with the data transmission speed far behind fiber optics. The lower speed is mainly due to the smaller bandwidth that, at the traditional microwave bands, is up to 56 MHz, whereas for fiber it is several gigahertz. The reason for this is that at the time when the microwave bands were defined, transferring voice was the main purpose and channel bandwidth was relatively narrow, but for transferring data with high speed, wide bandwidth or many symbols per hertz are necessary. The narrow bands require the use of high-order modulation schemes to increase the spectral efficiency and offer more throughput. Digital microwave radio is still one of the most viable technologies to transmit data. As the technology has advanced, microwave communication has become pervasive in our world. The remarkable popularity of these technologies caused device makers, infrastructure developers, and manufacturers to continually seek greater radio spectrum for more advanced product offerings. Today, standards for point-to-point systems cover a very large range of traffic capacities, channel separations, modulation formats, and applications over a very wide range of frequency bands that span from 4 GHz to 86 GHz and traffic capacities, from 9.6 kbps to 800 Mbps in the traditional microwave bands, up to 10 Gbps in the highest millimeter-wave bands. Typical applications of microwave point-to-point connections are: •• Rural and urban low, medium, and high-capacity links for backhaul in

mobile infrastructure; •• Long-haul trunk links for transport of information;

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•• Broadband fixed wireless access and wireless local loops in last-mile

application; •• Broadband wireless fiber optic extension; •• Backhaul for wireless local area networks; •• Governmental and military links, utility private fixed networks. Microwave radio technology has played a significant part in the roll-out and expansion of cellular networks since its inception and microwave backhaul is still the dominant technology for backhaul transport in Europe, Latin America, and emerging markets due to lack of fiber or copper availability. It is the most prevalent technology used in the global mobile backhaul market with around 50% of all backhaul connections: the application for the mobile backhaul accounts for approximately 75% of the total microwave radio equipment market. Figure 2.7 shows the present situation and the expected trend in terms of microwave backhaul share for geographical area. In China, Japan, and Korea microwave technology is today considered only as a backup link solution, because they tend to be fiber-rich and nearly all of their data finds its way to the wired network. Within the United States, T1 leased lines have been the backhaul option of choice due to relatively high availability, resulting in over 80% penetration of U.S. cell sites. Until recently, most wireless traffic has been transported, or backhauled, from cell towers to core networks over copper wires or optical fibers. This service was provided using several T-1 wired lines and less expensive Ethernet transport technology.

Figure 2.7  Microwave backhaul penetration: regional differences (Ericsson 2014).

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2.5 Microwave Backhaul: Trends and Expected Growth With the increasing adoption of smart phones and tablets, more users are turning towards mobile broadband as their primary resource for accessing the Internet and content-based applications. As a result, mobile data traffic is growing at a rapid speed and is expected to further increase exponentially in the coming years. 5G will further enhance this demand with capillary deployment of machine-to-machine communication. The emerging mobile communication services are characterized by an extremely higher demand of data traffic for each user. Mobile operators have added more data capacity capabilities to their existing base stations, and they built new towers for new base stations. Because leased-line transport service was priced high, many telecommunication companies have found mobile backhaul to be a fast-growing source of revenue. At the same time, there has been also a growing public resistance to add more radio sites, particularly in public areas, close to schools and hospitals or in highly populated areas. This factor has forced a more pragmatic approach to be adopted in the network planning in order to get microwave links to work reliably also in non-ideal conditions. In the future, to meet the ever-demanding expectations of mobile broadband users, it is very likely that improved and denser macro cells will be complemented by an increasing number of small cell sites, which are less intrusive and with a reduced output power. The availability of sufficient microwave spectrum is a crucial factor for microwave transmission deployments. More bandwidth and new spectrum are just as important for microwave transmission as they are for radio access to pave the way for mobile broadband. Microwave backhaul is an essential part of the network infrastructure but the maximum data-rate is limited today for traditional microwave point-to-point links to around 800 Mbps. Small-cell installations can be considered for indoor and outdoor sites like street lampposts, utility poles, and other urban structures, especially for those areas where the customers can be present more densely. Depending on the installation scenario, each wireless access site has specific requirements for power sourcing, power budget, and backhaul transport. Meanwhile, conventional macro-cell base stations will continue to upgrade network capacity, further driving demand for high-throughput backhaul. Harmonization of the usage of microwave spectrum and how to handle it in terms of the approach to licensing and spectrum fees are topics that have to be prioritized and supported by national regulators, operators, and vendors. The lower part of the microwave bands is generally used for hop length between 10 and 200 km since they are practically insensitive to rain fading. However, the amount of spectrum available in these bands is limited. The upper part of the bands is used for shorter hop lengths, usually lower than 1 km in dense urban areas. The amount of spectrum available in the upper part of the spectrum is

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larger than in the lower part, and millimeter waves seem a very good solution for responding to the capacity need. Millimeter-wave point-to-point digital radios have been manufactured for several years and, due to the high demand of rapid-deploy cellular telephone networks, their market is growing worldwide. This rapidly growing market is challenging the microwave industry with requirements for sophisticated radio front ends manufactured in high volumes, but with prices significantly lower than similar products realized just a few years ago. More recently, some new bands—42 GHz, 60 GHz, and 70/80 GHz—have been made available. However, these are very country or region-specific, and deployments so far are still marginal compared with traditional microwave installations, and their market is still modest in scale at this point. However, the enhanced capacity capabilities delivered by this technology will be invaluable as a backhaul aggregation solution for small cell deployments as they scale up. From an electronic components point of view, the advantages of monolithic integration become increasingly important because the system hardware should have limited size and weight [17]. The power consumption, although not as critical as in consumers’ handsets, needs to be low enough for long-term operation and needs to utilize directional transmission and reception of signals. Instead of mechanically steering the direction of a single antenna, a phased array antenna system can be used for transmission and reception with several advantages: •• For transmitters, the delivered power from all the elements is combined

in the space; thus, the power burden of each element is relieved. •• For receivers, the signal-to-noise ratio (SNR) is improved by summing the correlated signals while the noise is combined in an uncorrelated way. •• The pointing direction is steered electronically by applying incremental phase shifts to the various antenna elements, such that the installation and maintenance processes are simplified. Outdoor small cell backhaul will become the primary long-term market driver for millimeter-wave technology. Utilizing the available bandwidth at millimeter-wave, high-speed wireless transmission (10 Gbps) has been reported at 70/80, 120, and 220 GHz [18–21]. The high-frequency demonstrators use simple modulation and high bandwidth to achieve such high data rate transmission. This is completely the opposite to microwave band radios, where high modulation order and aggressive roll-off are used to improve the spectral efficiency. The transmission properties of very high-frequency, millimeter-wave links enable much simpler frequency coordination, interference mitigation and path planning compared to lower frequency bands. E-band, with two 5-GHz blocks of spectrum allocated at 71–76 GHz and 81–86 GHz, benefits from the large channel

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bandwidth available in this frequency and can give transportation capacity up to 2.5 Gbps per link. The V-band (57–64 GHz) can offer a good solution for the deployment of the small cells that are complementing the traditional macro-cells to improve coverage and mobile user experience. The demand for ever higher data rates is driving the research for broadband wireless communications to data rates up to 100 Gbps. Research works for 100-Gbps wireless communications that uses carrier frequencies beyond 100 GHz with bandwidths of 5 GHz or more and moderate spectral efficiencies have been reported. The large RF data bandwidth available there makes it ideal for short-range, last-mile segments from an optical fiber aggregation point to a nearby building for much less cost than it lay down additional fibers. Various wireless system and component design aspects, including millimeter-wave antenna design, monolithic microwave integrated circuit (MMIC) radio design, broadband analog, and digital baseband signal processing, as well as high-data rate protocol processing, are currently the subject of extensive research in some advanced laboratories worldwide.

References [1]

Pagani, M. et al., “Integrated Modules for SDH.Microwave Radio,” Proceedings of 5th ­International Workshop on GaAs in Telecommunications, April 1995.

[2]

Weizmann, M., “SDH Radio: The Technology of Today and Tomorrow,” Microwave Journal, Vol. 40, No. 1, January 1997.

[3]

Rappaport, T., Wireless Communications: Principles and Practice. Upper Saddle, River, NJ: Prentice Hall, 2002.

[4]

Sobol, H., and K. Tomiyasu, “Milestones of Microwaves,” IEEE Trans. Microw. Theory Techn., Vol. 50, No. 3, Mar 2002, pp. 594–611.

[5]

Clavier, E., “Micro-Ray Radio,” Elec. Commun., July 1931, pp. 20–21.

[6]

Carrara, N., “The Detection of Microwaves,” Proceedings of the Institute of Radio Engineers, Vol. 20, No. 10, October 1932, pp. 1615–1625.

[7]

Bucci, O., G. Pelosi, and S. Selleri, “Marconi and the First Microwave links,” International Symposium on Antennas and Propagation, Vol. 1, July 2001, pp. 18–21.

[8]

Bryant, J., “The First Century of Microwaves - 1886 to 1986,” IEEE Trans. Microw. Theory Techn., Vol. 36. No. 5, May 1988, pp. 830–858.

[9]

Kraus, J., “Heinrich Hertz-Theorist and Experimenter,” IEEE MTT-S International Microwave Symposium Digest, Vol. 1, May 1988, pp. 271–272.

[10] Isted, G., “Guglielmo Marconi and Communication Beyond the Horizon: A­­­­­­Short Historical Note,” Proceedings of the IEE - Part B: Radio and Electronic Engineering, Vol. 105, No. 8, 1958, pp. 79–83. [11] Corazza, G., “Marconi’s History [Radiocommunication],” Proc. IEEE, Vol. 86, No. 7, July 1998, pp. 1307–1311.

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[12] Fleming, A., “Guglielmo Marconi and the Development of Radio-Communication,” Journal of the Royal Society of Arts, Vol. 86, No. 4436, 1937, pp. 41–64. [13] Marconi, G., “First Microwave Radio Telephone Service,” Marconi Review, No. 40, 1933, p. 29. [14] Bea, F., “Mezzo Secolo Della Radio del Papa - [Half a Century of Pope’s Radio],” Città del Vaticano: Edizioni Radio Vaticana, 1981. [15] Vannucchi, G., “La Storia Della Telettra [The story of Telettra],” May 2013. http://www​ .biblioteche.unibo.it/../presentazione-vannucchi. [16] Ivanek, F., “50 Years of Microwave Communications Systems Development in the San Francisco Bay Area,” in IEEE MTT-S International Microwave Symposium Digest, June 2006, pp. 996–999. [17] Sharma, A. et al., “Ka-band I/Q Modulator Multi-Chip Module for High Data Rate Communications,” 2011 IEEE, Aerospace Conference, March 2011, pp. 1–9. [18] Chen, J. et al., “10 Gbps 16QAM Transmission over a 70/80 GHz (E-band) Radio TestBed,” 2012 7th European, Microwave Integrated Circuits Conference (EuMIC), October 2012, pp. 556–559. [19] Hirata, A. et al., “120-GHz-Band Millimeter-Wave Photonic Wireless Link for 10-Gbps Data Transmission,” IEEE Trans. Microw. Theory Techn., Vol. 54, No. 5, May 2006, pp. 1937–1944. [20] Antes, J., “A High Linearity I/Q Mixer for High Data Rate E-Band Wireless Communication Links,” in 2012 7th European, Microwave Integrated Circuits Conference (EuMIC), October 2012, pp. 278–281. [21] Laine, D., “Evolution of Microwave: A History of Wireless Communications,” available at http://www.blog.aviatnetworks.com.

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3 Architecture Roberto Quaglia and Thomas Emanuelsson 3.1 Basic Architecture and Options A backhaul microwave radio is composed by basic building blocks as described in Figure 3.1. The antenna is connected to the microwave front end through a waveguide or coaxial cable. The microwave module is basically analog circuitry, and provides amplification, frequency translations, and filtering at the carrier frequency band. The microwave module communicates with the modem, which aims to prepare the signal containing the information into a signal optimized for the transmission (e.g., through modulation or signal coding). The L3 layer is implemented by the modem. The baseband/control unit takes care of multiplexing/demultiplexing several channels, actually implementing the transport layer. It also acts as interface between the radio and the other parts of the mobile network. If the module is mounted on a base station, the backhaul radio needs in fact to communicate with the base station radio. If the backhaul radio is on the core network side, it will communicate with the Internet and the core network layers. If the radio is installed on a repeater tower, the baseband signal will then be simply transferred to another point-to-point radio module. The power supply generates the needed voltages and distributes the direct current (DC) supply to the several blocks. The physical position of each block can vary: the position of the antenna is on the tower, while all the other blocks can be placed on the tower itself or at the tower’s base, enclosed in a building. Next, the three architectures adopted for microwave radios are described. 49

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Figure 3.1  Block scheme of a backhaul radio.

3.1.1 Full Indoor As stated before, the name full indoor is misunderstanding: the antenna will always be placed on a tower (or on a building wall, if adequate). However, in a full indoor backhaul radio the antenna is actually the only piece of hardware not enclosed in the building at the tower’s base. Figure 3.2 shows a scheme of a full indoor system. The microwave module is connected to the antenna through a coaxial cable or waveguide, according to the working frequency and tower’s height. The full indoor solution is advantageous for very high-power radios that need a microwave module with great size and weight that cannot be placed at the tower’s top. Moreover, the placement in a building permits a thorough control of the environment conditions, as temperature and humidity, that facilitate the design of the electronics. The big limitation of this solution is represented by the long waveguide that connects the microwave module to the antenna, which is expensive and introduces high loss and dispersion. 3.1.2 Split Mount Figure 3.3 shows a block scheme of a split-mount solution. At the tower’s top, the antenna is placed together with the microwave module (outdoor unit), while

Figure 3.2  Full indoor configuration.

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Figure 3.3  Split mount configuration.

all the rest of the system is at the tower’s base (indoor unit). Two possible configurations are possible; see a sketch in Figure 3.4: in one case, the outdoor unit and the antenna are mounted together, while in the other case they are mounted separately and connected by a short piece of soft waveguide. A coaxial cable connects the modem to the microwave module through IF: in this way, the requirements on the quality of the connection between the tower’s ends are relaxed, since the microwave signal will be actually generated and amplified in proximity of the antenna. The power supply of the outdoor unit is provided through the coaxial cable itself. The split mount solution is presently the most adopted for microwave backhaul systems up to Ka-band. This solution is viable thanks to micro­wave modules that are small enough to be placed at the tower’s top. Moreover, they are able to withstand the deeply varying environment changes that occur during the day and the year. On the other hand, the mainteinance of the microwave module by crew is more difficult with respect to a full indoor solution, since the tower’s top must be reached.

Figure 3.4  S  ketch of possible split mount configurations, left: direct mount: right: separate mount.

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3.1.3 Full Outdoor For the highest-frequency bands adopted in microwave backhauling, as V- and E-band, the full outdoor solution is usually exploited. Figure 3.5 shows the block scheme of a full outdoor backhaul radio, while Figure 3.6 shows an example of mounting on a tower. Each block is placed outside, either at the top of a tower (usually an existing one) or on any convenient point like the rooftops, of buildings. The desire of service providers is to be able to install these systems on existing structures like street lights pillars: however, some issues related to the need of accurate and stable positioning and pointing need still to be fully addressed. The connection between the radio and the external interfaces is provided through an Ethernet cable that starts from the tower’s base. An Ethernet access is placed there, together with the power supply that feeds the radio through a power over Ethernet (PoE) solution. For such high bands, since the

Figure 3.5  Full outdoor configuration.

Figure 3.6  Sketch drawing of a full outdoor radio.

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adopted wide bandwidth requires an IF that is actually a microwave frequency, split mount solutions always present the unresolved issue of the long microwave cable or waveguide required.

3.2 Building Blocks and Their Role for Present Systems The scheme of Figure 3.7 reports a detailed block scheme of a microwave radio for backhaul in a split mount structure. The antenna is probably the only static hardware of modern point-to-point radios: in fact, it is a piece of hardware that transduces microwave power into electromagnetic field (or the opposite), and it is not controlled remotely by the baseband units. At the moment of the installation, it is positioned, with respect to the antenna on the other end of the link, for optimum communication, and it is mechanically fixed to prevent/minimize any movement, also in case of a strong wind or adverse weather conditions. All the other components are instead interacting through control signaling for optimizing the effectiveness of the communication. Moreover, some configurations can be controlled remotely by network maintenance crew. The antenna is directly connected to the branching module: this contains circulators and cavity filters. The latter actually define the go and return frequencies of the radio, and

Figure 3.7  Detailed block scheme of a split mount microwave radio.

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they can be substituted by mechanical intervention to change the frequency pair of operation of the link. The branching is connected through waveguides to the microwave module, in particular to the power amplifier (PA) for the go port (or to the power detector), and to the low noise amplifier (LNA) for the return port, respectively. According to the upconversion and downconversion policies, mixers, filters, and amplifying drivers complete the transmitter (TX) and receiver (RX) signal paths. Through a diplexer, a single coaxial cable is used to connect the outdoor unit to the indoor unit using two IF channels. The cable also brings the power supply and some control signals. The indoor unit uses a diplexer for feeding the coaxial cable. On the go path, the signal reaches the diplexer from the IF quadrature upconverter. The in-phase/quadrature (I/Q) input signals are supplied by a digital-to-analog converter (DAC) that converts the digital streams from the modem. On the return path the signal from the diplexer is downconverted from IF, and the obtained I/Q streams are sent to the modem. The modem will mainly implement the modulation/demodulation and the forward error correction functions. It will also execute important countermeasures against analog circuitry imperfections. The modem will pass/receive the digital streams from the baseband unit. This high-speed connection is often a board-to-board connection, and it can be implemented through fibers. Figure 3.8 shows a drawing of a split mount arrangement where the indoor modem unit is connected to a rack of baseband units that can be multiple plesiochronous digital hierarchy (PDH) multiplexers or Ethernet routers. The indoor unit also contains a main control unit that manages all the radios, power supply circuits, and interfaces to the rest of the network. In the following, several building blocks of the radio will be analyzed in detail, indicating those that are in practice the typical realizations in modern microwave radios.

Figure 3.8  Sketch of a split mount radio with multiple baseband units.

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3.2.1 Antenna The antenna represents the interface between the radio and the free-space medium. Its role is to radiate/receive the microwave signal selectively into/from the free space, maximizing the power exchange between the two communicating radios. Its input/output in TX/RX can be resumed as: •• TX. Input: microwave signal from cable or waveguide. Output: propa-

gating wave. •• RX. Input: incident propagating wave. Output: microwave signal to cable or waveguide. By nature, an antenna operates exclusively on L1 of the ISO-OSI stack. The antenna choice/design is guided by working frequency and bandwidth, maximum size and weight, polarization, maximum noise, and cost. The most important figures of merit are gain (or directivity), sidelobe suppression, and polarization crosstalk. Radiation diagrams are usually adopted to describe antenna behavior. In point-to-point radio links, high gain antennas are adopted: this poses strong mechanical requirements of their mounting structures. In fact, high gain corresponds to very directive main lobe, and a misplacing of the antenna of only a few degrees can lead to a considerable reduction of the received power. This could lead to a drastic loss of fade margin with respect to the expected one. The mounting flanges have a pointing system that can be mechanically tuned during the link installation (see an example in Figure 3.9). Practically, almost all microwave link antennas for frequency above 4 GHz are parabolic dishes fed by waveguide horns. They are characterized by high gain, low sidelobes, and the possibility of using dual polarization. The larger the antenna diameter, the higher the antenna gain. The standard diameters range from 30 to 240 cm. It is important to consider the wind load of the tower when selecting the antenna and/or the tower, in order to avoid tower bending that can lead to link gain reduction. The mounting system also changes according to the type of tower (e.g., a trellis metallic tower or a concrete tower). The parabolic dish is often covered by a radome of dielectric material for protection and for lowering the wind load effects. 3.2.2 Microwave Module The microwave module is characterized, in TX/RX, by the following input/output: •• TX. Input: A baseband/IF data stream from the modem. Output: a micro­

wave signal around the carrier frequency, delivered to the antenna with a determined power level.

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Figure 3.9  Mounting structure for a parabolic antenna.

•• RX. Input: Microwave, low-power signal from the antenna. Output:

baseband/IF signal to the modem. The microwave module also needs a frequency reference for proper upconversion/downconversion. According to the selected frequency conversion strategy, this reference can be the carrier frequency (homodyne systems), a frequency close but different to the carrier frequency (heterodyne systems) or a submultiple of the carrier (when on-board multipliers are inserted). The frequency reference is usually generated on-board of the microwave module using a voltage controlled oscillator (VCO), whose frequency can be tuned by adjusting a control voltage. In case of heterodyne systems, as used in split mount architectures, the signal is converted to an intermediate frequency (IF) before being upconverted to the RF carrier frequency. The IF carrier is used to transport the signal on the coaxial cable interconnecting the indoor and outdoor units. Figure 3.10 details

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Figure 3.10  Block scheme of a microwave board in an outdoor unit.

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the scheme of the microwave module. On the TX chain, after the upconversion, the signal is amplified by several gain blocks that come before the power amplifier, whose aim is providing the needed output power. Linearity is the main figure of merit of the transmitter in modern digital communication systems, considering the demanding modulation schemes that are involved. Gain control and power control functionality are exploited alternating gain blocks with voltage variable attenuators. Feedback on power level and transmitted envelope is provided by power and envelope detectors, respectively. The TX is a critical component that requires several controls from the modem to optimize the transmission. For example, currents of all amplifier stages are monitored to compensate for temperature variations. Predistortion is usually adopted to improve linearity, particularly degraded by the power amplifier, and it can be part of the microwave module if it is implemented with analog circuits. Alternatively, it can be carried out digitally at modem level. Envelope detectors or dummy receivers are used to provide a feedback channel for linearity monitoring. In the RX chain, the signal coming from the antenna is amplified by a low noise amplifier (LNA), usually realized by a multistage amplifier that gives an optimum trade-off between noise figure, input return loss, and gain. According to the downconversion policy, image filtering, analog-to-digital conversion, and downconversion blocks can be placed in different order; see details in Chapter 5. The main figure of merit of a receiver is its sensitivity. The microwave module contains also basic analog/digital circuitry for the control of the bias of the several blocks. The microwave module of early radios was realized connecting with cables or waveguides several building blocks, enclosed in robust brass carriers. The modules were realized as hybrid circuits, with filtering, matching and bias on ceramic substrate and active/passive discrete components mounted and connected through bond wires or soldering. The present trend is to have a micro­ wave board, realized on a high-quality multilayer substrate, that connects several blocks in form of packaged microwave monolithic integrated circuits (MMICs). The waveguide launcher is integrated on the board, permitting to mount the branching unit directly on the back of the microwave board. This kind of organization permits to save space and manufacturing costs, but is very demanding in terms of power dissipation management. In fact, many components with important power dissipation are located closely on a single board. As a consequence, particular care must be dedicated to thermal design. A further integration can be thought, especially for millimeter-wave bands, using System in Package (SiP) solutions. In this case, several MMICs share the same high-quality packaging, and are interconnected through small traces of microstrip and bond wires. The waveguide launcher is mounted directly on the

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package. In this case, thermal issues and cross-talking between several blocks are really demanding. 3.2.3 IF Board and Modem The IF board of a microwave radio has the following inputs/outputs: •• TX. Input: byte stream from baseband to be coded, modulated and fil-

tered for proper transmission. Output: IF/baseband I-Q signals to the radio unit. •• RX. Input: IF/baseband I-Q signals from the radio unit. Output: byte stream to the baseband module. The modem (modulator-demodulator) actually takes care of manipulating the signal in its digital form, with ADC and DAC with IF quadrature modulator/­ demodulator transforming the digital representation of the signal to be transmitted in an analog signal at IF. Figure 3.11 shows a block scheme of a backhaul radio modem. In transmission, the main goal of the modem is to transforms a digital stream into a pair of I-Q signals that represent the modulation on a Cartesian plane. The microwave module will then complete the operation when the two signals will be upconverted at the same frequency, but with a quadrature phase. The main function of the modem is to perform the channel coding and decoding. This function is, however, not sufficient for an optimized transmission. The modem also includes other indispensable blocks, as predistortion and digital countermeasures that try to overcome the intrinsic limitations of the analog circuitry; see more detail in Chapter 4. The IF board is usually realized on a multilayer substrate, using highly inte­ grated solutions. IF quadrature modulator and demodulator are usually on two different integrated circuits and sometimes also embed ADC and DAC functionalities. The modem functions are carried out through field programmable gate array (FPGA) or application-specific integrated circuit (ASIC) implementations. Hence, the hardware architectures are rather standard, while firmware implementations is decisive. The IF local oscillators are separated for go and return paths, and they are realized with voltage controlled oscillators (VCOs) connected to phase lock loops (PLLs). Sometimes, they are also integrated in the upconverter/downconverter chips. The board layout and interconnection have a particular impor­tance, given the high sampling frequency adopted and the need for accuracy, meaning that cross-talk and dispersion must be minimized. Also thermal management is important, since FPGA and DAC/ADC can be particularly power-hungry, especially for large-channel bandwidth. As common in digital processing boards, many auxiliary circuits are present, as bias controls and memory storage units.

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Figure 3.11  Block scheme of the modem board in a split mount configuration with dual polarization.

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3.2.4 Baseband Unit The baseband unit is responsible of the switching/multiplexing features of the radio, implementing all the higher levels of the ISO/OSI stack. The baseband unit can be quite simple if placed in an external node of the network, or it can be a really complex multiplexing or routing unit if the radio is mounted on an aggregation node. This book does not cover the baseband unit block: good references about operation of backhaul from a transport network point of view can be found in [1–3]. 3.2.5 Power Supply Most radios operate with 48-V bias, while some low-capacity radios operate on 24 V. The on-board bias circuitry is responsible for sectioning and providing all the different voltage levels needed for several modules. For example, a digital circuit will operate on +5 V or +3.3 V, while analog circuits can require ±15 V or ±5 V, and microwave modules use 6–12 V for drain bias. Gate bias is instead finely controlled by DACs or analog circuits, since it does not require a high-current drive, and it is used to regulate the bias points of amplifiers and mixers for optimizing performance, and for controlling attenuation in voltage variable attenuators. In split mount configurations, the bias is brought to the outdoor unit via the IF coaxial cable. On the outdoor unit, a multiplexer will separate it from the IF signals, and on-board DC/DC converters will derive all the needed bias voltages. In full outdoor configurations, the supply is delivered via the Ethernet cable.

3.3 Branching Branching is a term used to indicate how the transceiver is connected to the antenna. Its function can be as a simple diplexer or more complex connecting different antenna and transceivers [4]. The scope of advanced branching configurations is to improve the reliability of the hop against equipment failure and multipath fading. The hop design must consider branching losses, which for increasing complexity can become significant. In the simplest configuration, a single antenna is connected to a single transceiver. This branching arrangement allows the single antenna to be used for both TX and RX, and it is in practice a diplexer. Microwave PtP radios do not use TDD, so a switch would not be useful: the TX/RX separation is carried out using a circulator. Moreover, bandpass filtering is applied to both TX and RX sections, the former having more stringent requirements due to emission masks, while the latter is more relaxed thanks to baseband filtering. Figure 3.12 shows a block scheme of a diplexer configuration.

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Figure 3.12  Basic branching: circulator and filters form a diplexer.

To improve the hop reliability, a hot-standby configuration can be implemented (Figure 3.13). Two transceivers work all the time at the same frequency pair, but only one TX is actually connected to the antenna, while the other one dissipates its output power on a dummy load. The RXs work in parallel, and the modem will select the best input signal. In case of failure of the main TX, a switch is used to commute the antenna on the other TX. Frequency diversity adopts instead two transceivers working in parallel at two different frequency pairs, and thus it does not require a switch on the TX path (Figure 3.14). Frequency diversity is effective as a countermeasure against Rayleigh fading, since it is highly probable that when a channel is affected by fading, the other is not, and this probability is higher for greater channel distance. The problem with frequency diversity is that it requires double-frequency allocation, meaning higher cost. Conversely, with respect to hot standby, it does not require a switch but selective filters, so the pros and cons must be carefully evaluated case by case.

Figure 3.13  Block scheme of a hot-standby branching configuration.

Figure 3.14  Block scheme of frequency diversity branching configuration.

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Figure 3.15  Block scheme of a spatial diversity branching configuration.

Spatial diversity is an alternative configuration to improve reliability in case of multipath fading. On every end, two receivers are connected to two antennas, while the TX is diplexed on one of the two antennas (Figure 3.15). The two antennas are mounted at a certain distance in the vertical direction, improving the probability to receive a sufficient signal also when fading causes a bending of the ray. The higher the distance between the antennas, the best the effectiveness of the branching. The higher reliability is paid in this case with the use of four antennas per link instead of two. Normally, the top-side antenna is shared with the TX, and it can be considered as the main antenna. The low-side antenna can be designed for smaller gain, since it intervenes rarely and the probability of simultaneous heavy rain and multipath fading is in fact very low. Protection and diversity techniques can be combined in hybrid branching configurations to obtain the needed reliability of the link with the lowest possible cost.

3.4 Capacity Improvement In microwave links, the link capacity can be seen as the product between the channel bandwidth (in hertz) and the spectral efficiency in bits/second/hertz (b/s/Hz). This clearly shows that, to increase capacity, either or both bandwidth and spectral efficiency must be improved. Increasing bandwidth appears as the simplest solution, but it has a direct cost in terms of licensed spectrum that has to be bought. Moreover, very large channel bandwidths also stress the electronics, in particular in the digital signal processing and digital-to-analog and analogto-digital-conversion. However, increasing the spectral efficiency also means to complicate the electronics, and it may lead to a reduction of fade margins, lowering the maximum hop distance. Links operating up to K-band usually employ high-spectral efficiency techniques to exploit the best from the relatively narrow channels that are available, while radios operating at higher frequencies, where a lot of spectrum is still available, make use of large bandwidth channels. Figure 3.16 shows how the spectral efficiency can be highly improved thanks to the techniques that will be described next.

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Figure 3.16  S  pectral efficiency increase using enhancement techniques (Source: Microwave capacity evolution. Ericsson Review 2011).

3.4.1 Modulation and Roll-Off Digital microwave radios make use of quadrature amplitude modulation (QAM) (details in Chapter 4). The modulation order M of an M-QAM modulation indicates how many different symbols compose the constellation, meaning that log2(M ) bits can be transmitted with each symbol. Modulation order up to 4,096 is nowadays adopted in radios up to K-band, while E-band radios use modulations up to 64/256-QAM. The spectral efficiency is related to the modulation order, but is also affected by the shaping applied when sampling the symbols. This filtering, normally of a squared-root cosine shape, can be very smooth, relaxing the time alignment constraints of the sampling, but leading to a low spectral efficiency. Conversely, if the pulse shape is very steep, spectral efficiency is improved, at the cost of more severe time alignment requirements.1 The roll-off factor is an indicator of the shaping of this filter: low roll-off means higher spectral efficiency. The spectral shaping in modern systems has improved the spectral efficiency almost 1.5 times with respect to first microwave backhaul radios. 3.4.2 Dual Polarization The orthogonality of E- and H-planes in propagating electromagnetic waves can be exploited in radio links to practically double the spectral efficiency; see Figure 3.17. This means that two radios will operate almost independently on the two 1. This also includes the cost of an increased peak-to-average ratio of an analog signal, which can affect the transmitter operation; see Chapter 6.

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Figure 3.17  Block scheme of a microwave link using dual polarization and XPIC.

planes, using a single antenna with double-polarization. However, due to the nonperfect isolation between the E- and H-planes introduced by antenna non­ ideality, nonperfect alignment of antennas, and propagation, the introduction of cross-polarization interference cancellation (XPIC) is necessary to improve the signal-to-noise ratio of every polarization, permitting to actually achieve the expected double improvement in spectral efficiency. 3.4.3 Line-of-Sight MIMO Multiple input, multiple output (MIMO) concept is based on the fact that isofrequency signals can become independent if they follow propagation paths with certain phase relationships. This allows to increase spectral efficiency and to perform spatial multiplexing. In the mobile access and wireless LANs, multi­path propagation due to reflection and diffraction is exploited to implement the technique. In microwave radio links, that use highly directive antennas the multipath is forced by using properly spaced antenna arrays with high directivity elements; see Figure 3.18. An N×N MIMO system allows for an N improvement of the spectral efficiency if the array elements are placed at the optimum distance. In a 2×2 MIMO system this means that the two transmitted signals will be received with a 90° phase difference at each receiver. In this condition, the two signals are orthogonal and the two streams can be separated thanks to DSP. For the scheme of Figure 3.18, the array design rule is d1d2 = D



c , 2f

where c is light speed in free space and f  is the carrier frequency.

Figure 3.18  Block scheme of a microwave link using a 2x2 line of sight MIMO.

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Figure 3.19  Adaptive modulation operation for varying rain conditions.

3.4.4 Adaptive Communication Techniques High-order modulations lead to high capacity links, but at the same time they reduce the fade margin of the link with respect to simpler schemes. This means that in order to reliably use high-order schemes the link hop must be lowered. The use of adaptive modulation partially solves this issue. In clear-sky conditions, the highest modulation order is adopted and full capacity is achieved; while propagation condition worsen, the modulation order is lowered and the available capacity is dedicated to priority traffic. Figure 3.19 shows the typical sketch about adaptive modulation operation. Adaptive coding operates similarly: in good propagation conditions, the coding overhead is minimized and the payload portion is maximized. When conditions worsen, heavier error correction becomes necessary, and the overhead is increased, lowering the effective capacity of the link, that is dedicated to priority traffic. Transmitter power is also adapted to minimize power consumption and dissipation and to minimize interference to other radios. In clear-sky conditions, the transmitted power is minimum, and it is increased when propagation attenuation worsen. All these kinds of techniques are effective at physical level if the above layers have ways of controlling the quality of the communication and they can prioritize traffic. As discussed in Chapter 1, IP based-backhaul can use different solutions for assigning priority to packet streams.

References [1]

Grayson, M., K. Shatzkamer, and S. Wainner, IP Design for Mobile Networks, Indianapolis, IN, Cisco Press, 2009.

[2]

Salmelin, J., and E. Metsala, Mobile Backhaul, New York, Wiley, 2012.

[3]

Ferrant, J. et al., Synchronous Ethernet and IEEE 1588 in Telecoms: Next Generation Synchronization Networks, New York, Wiley, 2013.

[4]

Manning, T., Microwave Radio Transmission Design Guide, 2nd ed., Norwood, MA, Artech House, 2009.

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4 Modem Maurizio Martina, Guido Masera, and Goran Biscevic 4.1 Modulations and Coding Wireless backhaul technology makes use of microwave point-to-point radio links with nonstandardized air interfaces. While wireless multiple access interfaces are developed according to precise international standards (and so the interoperability between different vendors’ products is guaranteed), backhaul technology is regulated by standards only regarding its performance, but no interoperability is required. Therefore, microwave and millimeter-wave backhaul equipment is vendor specific [1]. In particular, the single-carrier transmission with high-order quadrature amplitude modulation (QAM) and adaptive coding and modulation is routinely used for this application. In the microwave range, the modulation order can be as high as 4096-QAM, while the 256-QAM is typical choice for the millimeter-wave band [2]. Higher modulation orders are envisaged for the coming generations of links, in order to increase the spectral efficiency and channel occupation, which is today around 1 Gbps for the microwave band with 2048-QAM modulation, 112-MHz channelization and appropriate coding. In microwave systems, adaptive coding and modulation (ACM) is adopted to trade data rates for improved robustness during times of unstable signal quality, such as periods of heavy rainfall. This concept can also be extended to the signal bandwidth, in this case referred to as adaptive coding, modulation and bandwidth (ACMB).

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4.1.1 Radio Frame Generation The radio frame is generated through different steps, which can be grouped into two main blocks, referred to as modulation and coding. Indeed, modulation and coding are important parts of the modem architecture. They can be split into four building blocks, as shown in Figure 4.1, referred to as: 1. Symbol generator, devoted to formatting and shaping; 2. Carrier product; 3. Demodulation, detection, and synchronization, where filtering, carrier, and radio frame synchronization and equalization are performed in order to correctly demodulate the signal and detect symbols; 4. Forward error correction (FEC). Each block is made of several parts, which will be addressed next. 4.1.2 QAM Symbol Generation and Detection Nowadays, QAMs are among the most used schemes both in wired and wireless digital transmissions. QAM relies on the concurrent use of two orthogonal carrier signals at the same frequency [3]. If f c is the carrier frequency, then wc = 2 p f c is the angular frequency, and the corresponding QAM signal is:

s(t ) = I (t ) × cos(2π f c t ) - Q (t ) × sin(2π f c t ) (4.1)

where I ( t ) and Q ( t ) are the in-phase and quadrature signals related to QAM symbols. Depending on the actual radio frequency, f c can be either the real carrier frequency or an intermediate frequency. Let us assume, for the sake of simplicity, that I ( t ) and Q ( t ) are shaped via rectangular pulses pTs ( t ) with duration Ts. As a consequence, I ( t ) and Q ( t ) are obtained associating QAM symbols to pulses: I (t ) =



Ns

å βk × pT (t - k ×Ts ) (4.2)

k =0

s

Figure 4.1  General modem block scheme.

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Figure 4.2  Square 16-QAM constellation with subsets definition.

Q (t ) =



Ns

å α k × pT (t - k ×Ts ) (4.3)

k =0

s

where Ns is the number of symbols and bk and ak are the real and imaginary parts of the kth symbol. As an example, Figure 4.2 shows the constellation of a square 16-QAM, where subset mapping is used [4]. Nonsquare QAM modulations have been proposed in the literature as well as square ones [5–7]. Square QAM modulations permit separate processing on in-phase and quadrature rails. However, 2n-QAM constellations with n odd are also used (e.g., 32-QAM, 128-QAM, 512-QAM, 2048-QAM) and are referred to as cross-­ constellations. Thanks to their geometrical properties, cross-constellations have lower peak to average power ratio (PAPR) and have thus best performance in presence of nonlinear distortion. Nonsquare QAM modulations with n even have also been proposed to reduce PAPR. All the nonsquare constellations require joint processing of in-phase and quadrature rails, so performance and complexity must be evaluated case by case. As an example, nonsquare QAM modulations, such as the 32-QAM one shown in Figure 4.3, are often used in microwave systems. The four-subset division, which was used in trellis coded modulation (TCM)-based systems, has been replaced with the 16-subset division in modern systems, where iterative decoding FEC and high-order QAM are employed. 4.1.2.1 Symbol Generation

In order to map bits to an integer number of symbols, they must be formatted and padded with zeros. Then the in-phase and quadrature signals (4.1) are

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Figure 4.3  32-QAM constellation.

routinely generated by the means of digital circuits followed by two digital-toanalog converters (DACs). In order to avoid intersymbol interference [3], pulses associated with I ( t ) and Q ( t ) are not rectangular and pulse-shaping filters are used instead. One of the most used shaping filters, which meets the Nyquist criterion for zero intersymbol interference, is the raised cosine filter [3]. Thus, to achieve the matched filter condition, square root raised cosine (SRRC) pulses are used both at the transmitter and the receiver sides. These filters are usually implemented as finite impulse response (FIR) filters [8–10]. 4.1.2.2 Hardware Symbol Generation

The implementation of the SRRC shaping filter as an FIR filter can exploit several general techniques proposed in the literature. Some of these techniques deal with algorithmic-level transformations to achieve different representations of the filter [11], including direct or transpose data-flow [12]. Other techniques aim to exploit the value of filter taps to reduce the complexity of the multiplications, such as canonic sign digit (CSD) or reduced adder graph (RAG)

Figure 4.4  Symbol generation and shaping.

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representation [13, 14] and distributed arithmetic [15]. In [8], a direct-pipeline architecture, based on CSD representation, is proposed and implemented on an Altera field programmable gate array (FPGA). This architecture, which uses 14 bits for input/output data and 16 bits for internal processing, runs at about 200 MHz requiring 1,189 (46%) logic elements. The solution proposed in [9] relies on a polyphase structure [12] implemented via distributed arithmetic on a Xilinx FPGA, whereas in [10], a RAG-inspired direct form is presented and implemented on an Altera FPGA. The solution proposed in [10] features 5,161 and 2,192 combinational and register logic elements, respectively, for a 14-bit implementation. Thus, it requires less logic resources than other solutions, including the polyphase structure, at the expense of about 18 kbit of memory. 4.1.2.3 Symbol Detection

At the receiver side QAM symbols have to be detected for recovering the transmitted data. Usually, this is obtained by decomposing the received signal in two branches, each of which contains pulse amplitude modulation (PAM) symbols. Let 22·q be the size of the QAM modulation; then the size of each PAM modulation is 2q. Thus, the received in-phase and quadrature signals, which are the output of the two mixers controlled by a voltage controlled oscillator (VCO), are converted to digital samples by the analog-to-digital converters (ADCs). Then digital signal processing is applied to: apply IQ corrections and compensation, shape pulses with SRRC filters, equalize the data, and recover/correct the carrier frequency, as shown in Figure 4.5. For the sake of simplicity we assume that I and Q can be processed separately to detect in-phase and quadrature components of the transmitted symbols. Let c be the alphabet of each PAM constellation and xi∈ c be one element of the alphabet; then each detector computes di = |y–xi|2, the squared Euclidean distance between the received value (y) and xi for every xi∈ c [3]. Then the problem of choosing the best xi can be seen as an optimization problem. Several techniques have been proposed in literature [16] and, among them, maximum a posteriori (MAP) and maximum likelihood (ML) are the most used ones. The MAP estimation maximizes the a posteriori probability, namely:

Figure 4.5  QAM demodulator and digital processing.

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ì P[ y|xi ]× P[ xi ]ü xˆ = max{ P[ xi |y ]} = max í ý (4.4) P[ y ] xi ÎX xi ÎX î þ As it can be observed, the denominator in the right part of (4.4) is independent of xi; thus, it does not contribute to the maximization of the a posteriori probability. If symbols in c are equiprobable, then (4.4) simplifies to

xˆ = max{ P[ y|xi ]} (4.5)



xi ÎX

which is the ML estimation. Since the logarithm function is monotonic, the maximization problem can be reformulated as xˆ = max{log( P[ y|xi ])} (4.6)



xi ÎX

In particular, assuming an additive white gaussian noise (AWGN) channel, it can be shown that log( P[ y|xi ]) = K -



di (4.7) 2 ×σ 2

where K = - log( 2π × σ ) and s2 is the variance of the noise, and as a result, the ML criterion in (4.6) is equivalent to minimizing the Euclidean distance [3]. However, since modern FEC schemes rely on soft decoding, where probability expressions are used to quantify the likelihood of receiving a certain symbol [16], the detector has to output soft information about the reliability of each bit contained in a received symbol in the form of logarithmic likelihood ratios (LLRs). Let us assume, as an example, that each symbol x i corresponds to 3 bits (8-PAM) referred to as a, b, c, which is x i = (a, b, c). Let P[a] and P[ã] be the probability of bit a being 1 and 0 respectively, and æ P[ y|a ] ö λ a = log ç (4.8) è P[ y|a ]ø÷



the corresponding reliability in the form of an LLR. Both P[y|a] and P[y|ã] can be expressed as a function of log(P[y|xi]) through

P[ y|a ] =

å P[ y|xi ]

xi ÎA

P[ y|a ] =

å P[ y|xi ] (4.9)

xi ÎA

where A Ì X and A Ì X are the subsets of c containing the symbols with a = 1 or a = 0, respectively, namely A = {( a, b, c ), (a ,b, c ), (a ,b , c ), (a, b, c )} (4.10)



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A = {( a , b , c ),( a , b , c ),( a , b, c ),( a , b, c )} (4.11)



Thus, (4.8) can be rewritten as d æ - i2 2 × çS e σ λ a = log ç xi ÎA di 2 çS è xi ÎA e 2×σ



ö ÷ ÷ (4.12) ÷ ø

where each log(P[y|xi]) term is replaced by (4.7), and K , which is common to both numerator and denominator, has been neglected. A simple implementation of (4.12) is obtained resorting to Log-MAP and Max-Log-MAP approximations [17], which are routinely used in turbo and low density parity check (LDPC) code decoding [18]. According to [17], the twoinput log-sum-exp (lse) function can be written as

z = lse{u, υ} = log(e u + e υ ) = max{u, υ} + log(1+ e -|u -υ | ) (4.13)

where the right-end-side term represents the maximum between u and u plus a correction term. Thus, the lse function is often referred to as max*. The twoinput max* can be extended to recursively implement the n-input lse (max*) function as z = max*{u, u , w, . . . } = max*{max*{max*{ u , u}, w } , … } and applied to (4.12) to obtain

λa =

(

)

1 × max *xi ÎA { -di } - max *xi ÎA { -di } . (4.14) 2 2 ×σ

The correction term in (4.13), namely fc(|u – u|) = log(1+ e –|u­-u|), can be imple­ mented in several ways leading to different approximations [19]. The simplest approximation is obtained with fc(|u – u|) = 0, which is referred to as Max-Log-MAP. 4.1.2.4 Hardware Symbol Detection

The implementation of a symbol detector relies mainly on the computation of di, the squared Euclidean distance between the received value y and constellation symbols x i and the LLRs as in (4.14). The concurrent computation of all di, values, for a PAM constellation of 2q symbols, requires 2q multipliers to perform the square operation. However, the computation of the q LLRs is obtained with tree structures, where each node can be either a two-input max* or a two-input max function, depending on the approximation (Log-MAP, Max-Log-MAP). As an example, the architecture required to compute the three LLRs (la, lb, lc) for

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Figure 4.6  Example: LLR generation for the 8-PAM case.

the 8-PAM case is shown in Figure 4.6.1 The total number of nodes required to compute the q LLRs is 3 · 2q – 2 · q – 4. Several low-complexity detectors have been proposed in the literature, including [20–22]. In particular, in [22] it is shown that for different modulation schemes the complexity of the detector can be reduced from O(2q) to O ( g q ), with the same performance as the Max-Log-MAP detector, by exploiting the symmetry of binary-reflected Gray-labeled modulations [23]. Interesting FPGA implementations of soft-output detectors are described in [24, 25]. In [24] a flexible detector to support up to 64-QAM constellations is implemented on a Xilinx FPGA. The architecture relies on the Log-MAP approximation, where the two-input max function is implemented as a subtracter and a multiplexer, whereas the f c ( | u – u | ) correction term is precalculated and stored in an 8-location look-up table [17]. The complete demapper architecture requires 8 multipliers, 310 registers, and 577 logic elements, corresponding to about 327 equivalent kgates. The solution proposed in [25] aims to both simplify the computation of the Euclidean distance and to reduce the number of computed squared Euclidean distances, leading to approximated LLRs. The Euclidean distance is approximated as

ìmax{u, υ} u2 + υ 2 = í îmax{u, υ} + δ (u, υ )

if min{u, υ} £ max{u, υ}/4 (4.15) otherwise

where d ( u , u ) = (min{u, u } — max{u, u } / 4 ) / 2 . The number of computed squared Euclidean distances is reduced resorting to detection by subregion, 1.

−

The constant factor (2 · s 2) 1 in (4.14) is not shown in Figure 4.6.

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which means exploiting the Gray mapping to partition the constellation into subregions depending on the sign of the received value. Such a technique leads to a reduction of 60% and 69% in the number of computed squared Euclidean distances for 64-QAM and 256-QAM, respectively. The detector, implemented on a Xilinx FPGA, requires 791 registers and 4,667 logic elements. 4.1.3 Filtering, Timing, Carrier and Radio Frame Synchronization, Equalization As already discussed in Section 4.1.2, matched filtering at the receiver side is required to maximize the signal-to-noise ratio and SRRC FIR filters are routinely used to reduce intersymbol interference [3]. Moreover, in order to correctly receive the transmitted signal, the receiver has both to synchronize and track the received data and to perform equalization. Indeed, the description given in Section 4.1.2 assumes that at the receiver side the numerically controlled oscillator (NCO) can produce waveforms perfectly in-phase with the received signal. 4.1.3.1 Synchronization

There are three different types of synchronization that are performed in the demodulator. The first synchronism that needs to be acquired is symbol timing synchronism, then carrier synchronism and finally frame synchronism. Timing Synchronization

Time synchronism acquisition is usually blind [i.e., it does not depend on the status of the other synchronisms (carrier, radio frame)]. Symbol timing synchronism is briefly described in Chapter 5, since it is closely related to the received signal sampling process. Historically the term “clock recovery” was used, because the sampling clock signal used to actually be the symbol clock. Different algorithms can be used in the phase comparator within the loop, among which many of them are based on the classical Gardner algorithm [26]. Nowadays, symbol timing recovery is typically performed by using an all-digital control loop. The interpolator action and its control have been devised in [27, 28]. Carrier Synchronization

One of the most used techniques to implement carrier synchronization relies on a digital-phase-lock loop, which is a digital circuit able to synchronize the NCO with the received signal. This is achieved by the means of a phase detector, which measures the phase difference Df between the received signal and the one produced by the NCO. The phase difference is then used to change the frequency of the signal produced by the NCO. When the phase difference becomes zero, the receiver is synchronized. A possible technique to measure the phase difference is through sin approximation:

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Dφ » sin( Dφ ) =

x I × yQ - x Q × y I

(4.16)

( x I )2 + ( x Q )2 × ( y I )2 + ( yQ )2

Other techniques rely on the computation of the arctangent function. Digital architectures for the implementation of the arctangent function belong to the general class of architectures for trigonometric functions, such as those used for the implementation of the NCO. Mainly two approaches are used to implement the NCO and the products. The first one, referred to as direct digital synthesis (DDS), relies on ROM-based look-up tables (LUTs) where the signals are stored. Then a phase accumulator and little logic are used to access the content of the memories and to generate the complete waveforms. The second approach relies on the Coordinate Rotation Digital Computer (CORDIC) algorithm [29], a bitrecursive algorithm, which increases the accuracy of the results approximately by 1 bit per iteration. DDS is a known technique used in instrumentation and communication systems to generate a digitally controlled waveform, where different frequencies can be obtained from a reference frequency source. In the simplest form, a DDS unit is made of a LUT, which stores one or more integral number of cycles of a sine wave, an address counter, and a register. As the address counter steps through each memory location, the corresponding sample is read from the LUT and stored into the register. The main problem with this DDS system is that the final output frequency can be modified only by changing the reference clock frequency or by reprogramming the LUT. In order to increase flexibility, a better scheme can be used, known as NCO. In this solution, the content of a phase accumulator is updated at each clock cycle by adding a constant value M to the current value stored in the phase accumulator register. The output of the phase accumulator is used as the address to the LUT, where each location corresponds to a phase point on the sine waveform from 0 to 2π. Alternatively, to reduce the size of the LUT, only values for the range 0 to π/2 are stored and the adder is replaced with a properly controlled adder/subtracter. Let f C K be the clock frequency, and n the number of bits in the phase register, then the LUT has 2n locations and the frequency of the generated waveform is given by: f 0 = f CK



M (4.17) 2n

Thus, for a certain LUT size, the signal frequency can be increased by increasing M . A detailed scheme of the NCO-based generator is shown in Figure 4.7, where Q is the quantized version of the accumulated phase P, represented using nQ and np bits, respectively. As the LUT only stores the sine samples in the range [0,π/2], the waveform in the three remaining quadrants must be corrected. In the second and fourth quadrants, the LUT content must be read in the

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Figure 4.7  NCO-based sine wave generator.

opposite sequence, from the last location to the first one: this is obtained by taking the one complement of Q . To this purpose, MUX1 is allocated to generate the LUT address directly from Q for quadrants 1 and 3, while A is equal to one complement of Q for quadrants 2 and 4. The second most significant bit of Q discriminates between quadrants 1 or 3 and 2 or 4, and it is used to drive MUX1. Moreover, in quadrants 3 and 4, the LUT output must take the negative sign: in Figure 4.7, the subtracter changes the sign of the LUT output, and the multiplexer MUX2 is driven by the most significant bit of Q , which identifies quadrants 3 and 4. Thus, the frequency of the generated waveform is the same as the one given in (4.17) with n = nP, and the frequency resolution is equal to: Df = f CK



1 2

np

(4.18)

Starting from the specified f and Df, one can derive nP from (4.18): é f ù n p = ê log 2 CK ú (4.19) Df ú ê



and then compute from (4.17): M =2



np

f f CK

(4.20)

The number of bits to represent the quantized accumulated phase depends on the required spectral purity. As an example, in [30] ROM-based look-up tables are used to implement an NCO as discussed in previously paragraphs. Moreover, the representation of the waveform relies on coarse and fine ROM samples. Based on a trial-and-error approach, the authors in [30] showed that they could achieve a spectral purity of –84 dBc by using a 32-bit phase accumulation register, where the 14 most significant bits are used to address the ROMs. The coarse and fine ROMs contain both 256 samples represented using 9 and 3 bits, respectively. The complete design presented in [30] and fabricated in a 0.8 –μm technology, runs at 200 MHz, and requires about 16 mm2 of area.

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CORDIC-based implementations of the NCO, as the one proposed in [31], rely on rotating a vector in a plane. Indeed, a vector with the coordinate (I, Q ) can be rotated to (I´, Q ´ ) as follows:

I ¢ = I × cos(θ ) - Q × sin(θ ) (4.21)



Q ¢ = I × sin(θ ) + Q × cos(θ ) (4.22)

where it can be observed that (4.21) is very similar to (4.1). Thus, the implementation of (4.1) can be derived by rewriting (4.21) and (4.22) as:

I ¢ = cos(θ ) ×[ I - Q × tan(θ )] (4.23)



Q ¢ = cos(θ ) ×[Q + I × tan(θ )] (4.24)

If one restricts tan(θ) = ±2­-i then the computation of a one-bit approximation of (4.23) and (4.24) yields:

I i¢+1 = K i ×[ I i¢ - Qi¢ × g i × 2 -i ] (4.25)



Q i¢+1 = K i ×[Qi¢ + I i¢ × g i × 2 -i ] (4.26)

with Ki = cos(tan–1(2–1)) and gi = ±1. The value of gi depends on the value of the approximated angle θi¢+1 = θi¢ - g i × tan -1(2 -i ). In order to accommodate angles in the range ±π, the starting values for the iterations must be:

I 0¢ = g 0 × Q (4.27)



Q0¢ = - g 0 × I (4.28)



θ 0¢ = θ - g 0 × π (4.29)

where g0 = –1 if θ < 0 and g0 = 1 otherwise (see Figure 4.8). The CORDIC architecture proposed in [31] requires 1,029 (24%) logic elements on an Altera Flex EPF10K100A FPGA, and the complete single channel QAM modulator occupies about 4000 (~80%) logic elements, running at 86 MHz. Frame Synchronization

Carrier synchronization is a fundamental step to have reliable demodulation of the signal. However, carrier synchronization permits recovering symbols but is not sufficient to achieve correct reception of the transmitted message. Indeed,

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Figure 4.8  CORDIC architecture.

finding the correct alignment of the received information to rebuild the data frame is of paramount importance. This operation, referred to as frame synchronization, is obtained by the means of different techniques. A very common technique to synchronize frame reception is the use of sequences of symbols (pilot or synchronization sequence), known at the transmitter and receiver side. The receiver computes namely, qi =



L

å rn+i × wn (4.30)

n =0

which is the correlation between the received signal r = r0,r1,... and a local copy of the known sequence w = w0, w1,..., wl with length equal to L. Then the receiver detects the synchronization condition, by comparing the result of the correlation with a threshold, as shown in Figure 4.9. In order to achieve reliable synchronization these sequences must have pseudo noise (PN) characteristics, namely the autocorrelation function must exhibit a peak when the sequence is aligned and be as low as possible otherwise [32]. A lot of sequences with such characteristics have been proposed in the literature [32–35] for communications applications, including spread-spectrum communications. It is worth noting that PN sequences can be generated via linear feedback shift registers (LFSRs), which are shift registers made of D-type flip-flops with a proper feedback q

r

w Figure 4.9  Frame synchronization block scheme.

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Figure 4.10  Linear-feedback-shift-register block scheme.

network containing xor gates as shown in Figure 4.10. The position of the xor gates depends on the characteristics of the sequence to generate [33]. The computation of the correlation as in (4.30) requires several multiplications. Thus, reduced complexity techniques have been proposed. As an example [36] proposed to convert r and w to the frequency domain so that convolutions transform into products. 4.1.3.2 Equalization

Multipath reflections and distortions introduced by both the wireless channel and electronic equipment like amplifiers and filters generally cause significant degradation in the received signal and determine poor performance [37]. Unfortunately, the channel parameters are not known in advance, and moreover they tend to change with time. As a consequence, receivers must track the channel characteristics by means of an automatic channel equalizer that restores performance and improves reliability. An adaptive equalizer is basically a linear adaptive filter able to model the inverse function of the channel. The high-level view of a system using an adaptive equalizer is given in Figure 4.11, where t [ k ] is the transmitted symbol, x [ k ] is the channel output, y [ k ] is the equalized symbol, d [ k ] is a reference signal, e [ k ] is the equalization error, d [ k ] – y [ k ] , and h[i] are the tap coefficients. The adaptive filter is essentially a FIR filter with N adjustable tap coefficients h ( i ) (0 0 with | fB − fA|= B) delivers an available power

PN = k BTB



(5.4)

to a matched load. Such a simple formula can be applied to most stages, provided that a suitable definition of temperature, which may not coin­ cide with the physical one and is thus an equivalent one, is given. Every component introduces its own noise contribution.

•• Noise temperature T: an equivalent temperature associated to a point

of a chain of stages that allows to compute the noise power as kBTB. The noise temperature of a single stage can be evaluated at its input or output. •• Noise factor F: between an input and an output, it is the degradation of signal-to-noise ratio (SNR) between the two points, as F=



SNRin SNRout

(5.5)

The noise factor of a physical device is always larger than 1.

•• Noise figure: it is the decibel measurement of the noise factor:

NF = 10 log10 F



(5.6)

The noise figure of a passive attenuator is equal to its attenuation (in decibels).

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The Friis’ formula of noise for a chain of matched devices with available power gains GA,i [see definition in eq (5.12)] and individual noise factors Fi states that the overall noise factor F can be computed as [3]:

F = F1 +

F2 - 1 F3 - 1 F N -1 + + …+ G A,1 G A,1G A,2 G A,1G A,2 G A,N -1

(5.7)

The practical implication is that, as long as the gain of the first stage GA,1 is sufficiently large, the overall noise factor is dominated by the noise factor of the first stage, whereas successive contributions are made negligible by GA,1 at the denominator. In practice, therefore, a low noise amplifying (LNA) stage must be placed as early as possible in the RX chain to avoid degrading the received SNR. The LNA gain is usually in the order of 10 to 15 dB, which is not enough for the ADC (usually it requires voltage variations of at least a few tenths of volts, thus powers of about −20 dBm on 50 Ω), but sufficient to limit the effect of noise from the other stages of the chain. As explained later, an increased gain usually involves higher noise figures and an accurate trade-off between the two must hence be chosen. 5.2.2 Downconversion Once the LNA’s position as foremost component in the RX chain is justified, compatibly with hardware connection and other constraints, we need to decide how to choose the next stage. If carried out immediately after low noise amplification, channel selection filtering would suffer from the same limitations encountered for the diplexer: narrowband filtering close to a microwave carrier requires extreme selectivity (steepness), implying great filter complexity. Thus, narrowband filtering is more effectively achieved at a lower frequency. Similarly, sampling the microwave signal at this stage would require a huge sampling rate (at least twice the highest frequency, according to the Nyquist theorem) and is thus also deferred to a later point, where lower frequencies are involved. The power level, as already pointed out, is still too low for a precise quantization, and the required amplification can be carried out with more relaxed constraints on noise figure later in the chain. For these practical reasons, therefore, the downconverter is the block usually placed immediately after the LNA, as shown in Figure 5.2. The downconversion strategy plays a major role in terms of requirements and achievable performance with a given technology, and several options are available to the designer [4]: •• Superheterodyne: involvement of one or two lower intermediate frequen-

cies (IFs) before reaching the baseband;

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•• Direct conversion: direct mixing with the carrier frequency, thus obtain-

ing a baseband signal in a single step; •• Low IF: downconversion to a low nonzero intermediate frequency to avoid DC issues; •• Digital IF: downconversion to a low nonzero intermediate frequency and conversion to digital: channel filtering performed in the digital domain. Next, a brief description of these approaches is given, and they are compared along with considerations on the common choices for backhauling applications. 5.2.2.1 Superheterodyne

According to the literature, U.S. Army Major Edwin Armstrong invented in 1918 the superheterodyne receiver (SHR) in order to overcome the lack of highfrequency amplifiers by using a lower intermediate frequency. Independently from Armstrong, Schottky too proposed a similar method in the same period and the basic scheme is shown in Figure 5.4. In its modern form, a superheterodyne receiver basically consists of at least one mixer (followed by or including an image-rejection method) whose local oscillator (LO) input is driven by a nonzero frequency fLO different from the carrier fRF. The mixer acts by multiplying the local oscillator with the received signal and thus produces two copies, at | fRF ± fLO|, respectively. The information content of the two is identical and thus only one of them is retained, usually the one at lowest frequency, while the other is suppressed by means of lowpass filtering. Assuming fRF > fLO, the downconverted signal at fIF = fRF – fLO 1 and |detS | < 1, then the device is unconditionally stable. Otherwise, the test does not provide a definitive answer. An alternative criterion [41] is available based on a single parameter m, which guarantees unconditional stability if: 2

µ=



1 - S11 > 1 (5.19) S22 - DS11* + S12 S21

Otherwise, the stable regions must be checked as the test is not conclusive. Although unconditional stability is not strictly required, several techniques can be used to narrow the unstable regions. These typically involve placing additional resistive components, with the obvious result of increasing resistive losses and thus lowering the gain and introducing more noise. One often needs to accept a potential instability in order to strive for higher performances. In these cases, GL and GS must be chosen within their stable regions and at a sufficiently safe distance from the borders to account for manufacturing tolerances and model approximations.

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5.4.1.3 Constant Gain Circles

With the definition introduced in (5.13), one can obtain the following expression for the transducer gain: GT =



1 - S11

2

1 - S11G S

2

S21

2

1 - GL

2

1 - G out G L

2

(5.20)

The leftmost factor in (5.20) accounts for source mismatches, whereas the rightmost for load mismatches. The central factor S21 is instead the intrinsic amplification term due to the active stage. Active devices are, in general, nonreciprocal and often |S21| >> |S12|. Considering S12 to be 0 constitutes the unilateral approximation, which neglects the reverse gain from the output to the input but provides simpler expressions: Gin = S11 and Gout = S22 regardless of GS and GL. In general, and very often in high performance, the device is not unilateral, and the design procedure is slightly more involved. The maximum transducer gain can be found by analytically solving the simultaneous conjugate matching: G S = G *in and G L = G *out. The computation for the bilateral case entails two conditions, as Gin depends on GL and Gout depends on GS. The analytical solution for GS,opt and GL,opt is available, for example, in [38]. Under such optimal conjugate matching, the maximum transducer gain takes on the value: ì ïMax. Available Gain : ï í ï Max. Stable Gain : ïî

S21 (k - k 2 - 1) S12

GMAG

=

GMSG

S = 21 S12

if uncond. stable otherwise (5.21)

The two possible values take into account the fact that an unconditionally stable device suffers from excessive stabilization (k >> 1), which reduces its maximum available gain. Contrarily, a potentially unstable device can often exhibit a higher gain. One might nonetheless wish to attain a suboptimal gain, for example, to avoid input and/or output matching networks and this occurs by whenever GL and GS differ from the optimal ones in [38]. When both reflection coefficients are fixed, the resulting transducer gain comes as a consequence via (5.20), as the input and output reflection coefficients do. If only GL is instead fixed, the operating power gain in (5.11) can be computed and a proper choice of G S = G *in yields the same transducer gain. Dually, if only GS is known, the available power power gain in (5.12) is fixed and conjugate matching G L = G *out yields an equal transducer gain.

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The relation between operating power gain and GL in (5.11) [or between available power gain and GS, eq (5.12)] can be also used in order to find the set of values of reflection coefficient that yield a given value for the considered power gain. A known value for the left-side member of (5.11) [or (5.12)], in fact, defines a constant gain circle in the complex plane for GL (or GS). Expressions of its center and radius can be found on many textbooks (i.e., [38, 42]), as a function of the scattering parameters and the desired gain. A typical sketch of constant gain circles is shown in Figure 5.12. The constant gain circle collapses to a single point for the maximum attainable value. Drawing such circles on the Smith chart allows a graphic interpretation of the required locus of GL (or GS), so that a suitable termination can be chosen for the desired operating (or available) power gain. Once this step has been completed, the other termination, GS (or GL), can be chosen optimally for conjugate matching, which yields the transducer power gain equal to the operating (or available) power gain value selected. 5.4.1.4 Noise Figure

An arbitrary noisy two-port device can be traditionally modeled as a noiseless twoport with series noise sources on its ports. These may have a partial correlation.

Figure 5.12  Constant gain circles.

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In order to evaluate the noise figure and find its minimum with proper choice of source impedance, the noise source on the output port is converted to a shunt noise current In on the input by means of the ABCD matrix, where its own noise voltage source is already placed (Vn). A noisy source Is with its inter­ 1 nal reference impedance Z s = connected to the input completes the model Ys sketched in Figure 5.13. The noise factor can be evaluated from (5.5), but since the two-port device is now noiseless it does not alter the SNR and thus only the degradation introduced by the two equivalent sources must be taken into account. Since the source noise is uncorrelated from the internal noise sources, powers add up, and the noise factor is given by: F=



2 E é I n + YsVn ùû Ps + Pn = 1+ ë (5.22) 2 Ps E éë I s ùû

where Pn is the combined noise due to In and Vn, whereas the last equality exploits that all noise powers are referred to an arbitrary reference impedance whose value cancels out in the ratio. The correlation between In and Vn is introduced by splitting the former as an uncorrelated current Iu and a correlated one Ic = YcVn, where the correlation admittance includes a conductance Gc and susceptance Bc. Powers of Iu and Vn terms now add up. The three uncorrelated sources can also be expressed as equiva2 2 2 E[ I s ] E[ I u ] E [ Vn ] lent thermal resistances: Gs = , so that , Gu = , and Rn = 4kTB 4kTB 4kTB the following is obtained: 2

F = 1+



Gu2 + Ys + Yc Rn (5.23) Gs

Figure 5.13  M  odel for evaluation of noise figure of a two-port device with equivalent noise sources In and Vn connected to a noisy source Is, Ys. The current Isignal represents the useful signal coming from the source but it is not involved in the computation of noise figure.

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An optimum source impedance Ys,opt = Gs,opt + jBs,opt can be defined to give the minimum F:

Bs,opt = - Bc Gs,opt = Gc2 +



(5.24)

Gu (5.25) Rn

This result shows that, in general, the optimum source impedance for noise does not coincide with the maximum power transfer condition (conjugate matching of input). However, as the source reflection coefficient may be constrained or fixed (i.e., to avoid any input matching network), the optimum value is seldom practically obtainable and the noise factor can be computed as a function of the minimum noise factor (attained when the optimum load impedance is used) and the chosen source impedance Ys: F = Fmin +

Rn 2 Ys - Ys,opt Gs

æ G ö with Fmin = 1 + 2Rn ç Gc + Gc2 + u ÷ Rn ø è (5.26)

Most commercial datasheets provide Rn, Fmin, and Ys,opt, so that the noise figure corresponding to any other source termination can be computed. When GS is free to some extent, (5.26) can be used to draw the constant-noise figure circles, that is, the loci of values for the input reflection coefficient GS for which F takes on a specified value greater than Fmin. These circles are centered on Gs,opt and can be usually plotted alongside the gain circles to designate an appropriate trade-off between the two. Similarly to the constant gain circles, the analytic expressions of the center and radius of constant-noise figure in the GS plane are available in most microwave engineering textbooks, such as [38, 42]. For F = Fmin the constant noise circle collapses to a single point, which is GS = GS,opt corresponding to (5.25). For higher values of F , instead, the circles typically increase in radius similarly to the situation depicted in Figure 5.14. From such a visual representation, one may easily figure out the qualitative relationship between input termination and exhibited noise figure and thus select a proper GS for the required F. 5.4.1.5 Frequency Response

The previous theoretical description is valid at a single frequency. One is usually not interested in receiving a zero-bandwidth signal carrying no information whatsoever (completely predictable). The design must therefore take into account the behavior through the whole desired bandwidth and control stability factors as well as keep a flat gain. Gain, in turn, depends on the frequency variation of scattering coefficients of the active device and on the frequency response of matching networks, if any. The former

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Figure 5.14  Constant noise circles.

suffers from the typical 6 dB/octave rolloff of S21, whereas the latter exhibits an intrinsic limit described by the Bode-Fano criterion (see [43, 44]). Often broadband solutions are devised by: •• Matching sections to counteract the S21 decay but usually worsening the •• ••

•• ••

input and output reflections and complicating the networks; Resistive matching, at the expense of lower gain and higher noise, therefore not suited for LNAs; Gain flattening, by means of feedbacks which reduce the maximum gain and increase the noise figure, thus not compatible with LNA requirements; Balanced amplifiers, requiring two active stages and two 90° couplers; Distributed amplifiers, involving several active stages connected by transmission lines.

It is worth remarking that gain flatness is a constraint that can be partially relaxed by the baseband digital processing. A nonflat response, is evaluated by the digital equalizer as a further linear distortion of the received signal, similar to other channel nonidealities, and can, to some extent, be compensated with proper digital filtering. This is achieved by relying on the channel-estimation

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features of the equalizer (e.g., by means of known pilot signals) or by modeling the response of the RX chain in a tuning phase so to implement a nonreconfigurable equalization. 5.4.2 Topologies As described previously, the traditional approach to designing LNAs at microwave and millimeter-wave frequencies entails biasing the transistor in order to minimize its attainable minimum noise figure. The source termination is then chosen to match the optimum source noise termination, whereas the load termination is determined to achieve the required gain, involving ad hoc reactive matching networks on both sides. This approach, however, suffers from a severe drawback due to trading off power gain for a lower noise figure, heavily in favor of the latter: the optimum noise source termination is often very different from the expected input impedance, thus drastically limiting the attainable gain. The modern requirements on the RF front end place more constraints on the number of passive components, which must be reduced by removing, wherever possible, the passive reactive matching networks. The current trend is implementing an active matching, where the transistor itself is biased to close the gap between the real part of the optimum noise source termination and of the input impedance. This technique is moreover independent on both the transistor type (i.e., FET or bipolar) and the chosen semiconductor (Si, SiGe, GaAs, GaN, InP,...). All the schematics in this section, therefore, can be implemented with bipolar and FET transistors. The key factors to take into account for an effective design of low-noise amplifiers can be summarized as follows [45]: 1. Prefer active matching by acting on bias and topologies over the use of reactive matching networks. 2. Power consumption of the amplifying stage is determined by biasing current, which, in turn, is closely related to the desired optimum source termination. 3. By fixing the bias current density that flows in the transistor, other layout parameters such as gate width or emitter length, number of transistors in parallel, and number of fingers do not affect the minimum noise figure of the transistor. 4. The real part of the optimum noise termination can be tuned by using transistor dimensions, whereas the real part of the input impedance can be controlled by means of a reactive feedback element. The two aspects, therefore, are made independent on each other, thus simplifying the design procedure.

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These simple rules can be used to yield a very good trade-off between noise figure, power gain, and die area. Next, the two main topologies for LNAs are outlined and their common properties are presented: common emitter or common source and cascode. The cascode configuration is then discussed in greater detail in Section 5.4.3. 5.4.2.1 General Properties

Common emitter (or common source) and cascode topologies owe their popularity to the simple structure and to their possibility of achieving active matching by relatively easy means. The two structures are sketched in Figure 5.15. The cascode topology is actually the combination of a common emitter (or source) structure and a common base (or gate) amplifier, as shown in Figure 5.16. The latter typically exhibits a wider bandwidth than the former, although its input impedance is often too low for practical matching. As a consequence, a common emitter section is placed as buffer input stage for its moderately high input impedance. Its low gain is then improved by the common base stage. With respect to a single-stage amplifier section, the cascode solution entails two active components, but possibly exhibits: •• Higher output-to-input isolation; •• Higher input impedance; •• Higher output impedance;

Figure 5.15  C  ommon amplifier configurations with bipolar transistors. The structures can be implemented with an identical topology in MOS technology.

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Z

Figure 5.16  Equivalent model of cascode amplifier.

•• Higher gain; •• Wider bandwidth.

However, the voltage drop across the transistors is larger, which, in turn, affects the allowed voltage swing and may limit the linearity figures. Both structures in Figure 5.15 can be equivalently represented as seriesseries feedback networks with LB and LE as reactive feedback elements. Such an equivalent network can be used to transform the real part of the input impedance to the desired optimum noise termination and to also cancel out their imaginary parts. The expression of the input impedance of the inductive-feedback network can be derived at the operating frequency (ω = 2πf  ) from the simplified equivalent circuit and is found to be:

1 ö æ Z in » Rb + rE + ω T LE + j ç ω LE + ω LB è ωC in ÷ø

(5.27)

with C in =



g m,eff ωT

(5.28)

Equation (5.27) shows that the amplifier has a resonant input that behaves as a series RLC network, with a real part constant over frequency. As a further consideration, if the LNA needs to be connected to a 50 Ω source impedance, such as a typical antenna or filter, the real part of the input impedance must be matched to this value too, whereas the imaginary part should be canceled. The feedback inductance LE can be exploited to fulfill the first requirement, whereas the base inductance LB can be used to fulfill the second requirement as well. The value of the LE inductance depends only on the fT of the topology that depends on the bias current density of the transistor. It is independent on operating frequency, transistor dimension, and bias point.

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The value of the emitter inductance LE is typically small enough to neglect its associated series resistance. On the contrary, the base inductance is usually significantly larger and its parasitic series resistance should not be neglected. The design process should start with the determination of a suitable LE to match the real part of the input impedance to the given source: LE »



Z 0 - re - Rb 2π f T

(5.29)

and then proceed with the identification of Lb to minimize the imaginary residual at the chosen operating frequency: LB »



ωT - LE ω 2 g m,eff

(5.30)

When considering the output matching, instead, one often has to deal with a load provided by the input of another amplifying stage or mixer or phase shifter. The output of the LNA is intrinsically capacitive, and a reactive network can be implemented by an inductance toward the power supply and a capacitance between the transistor and the load. An advantage of this matching structure is that it also implements DC blocking towards the following stages, thus decoupling the bias networks. The inductance further allows to increase the voltage swing on the output terminals even beyond the supply voltage, thus providing higher overall linearity. Another remark concerning the linearity refers to high-voltage conditions: •• When the input voltage becomes large, the base-emitter junction intro-

duces a nonlinear behavior. •• High voltages may move the transistor outside the active region (saturation region for FET transistors); for a cascode topology, the common-base transistor always undergoes larger signal swings and, as a consequence, its collector-emitter voltage should be designed to be larger than the common-emitter transistor. To improve linearity performance, the biasing should be set to provide high current densities where the transconductance exhibits its peak. However, on the other side, the current density should also be set at an optimum value which determines the minimum attainable noise figure. A suitable trade-off between these two must be found. If the nonlinear junction behavior has strong impact on linearity, the optimum current density can be mantained while increasing the transistor size, thus increasing the overall bias current. This is also an expression of the usual trade-off between noise figure and DC power consumption, which always plays a role in the design of LNAs.

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5.4.3 Cascode Configurations Starting from a structure based on the cascode configuration in Figure 5.15(b), it is possible to apply some small modifications in order to optimize a given parameter or to reach more easily the best transistor performance. By taking into account the generic scheme in Figure 5.17, the most common modifications of the cascode configuration are: 1. LNA with an external resistance (Rex ) between collector and emitter in HBT or drain and source in MOSFET devices [46] (M2); 2. LNA with an inductance between the collector/drain of the transconductance amplifier (M1) and the emitter/source of the transistor in common base configuration (M2) [47]; 3. LNA with an inductance between the power supply and the base/gate of the transistor in a common base configuration (M2) [48]; 4. LNA with a Π -network in order to improve the input matching [49]. Each of these is detailed in the next paragraphs. These revisitations of the classical cascode structure constitute an example of how a mature structure can be modified in order to improve a specific parameter. 5.4.3.1 LNA with External Resistance

This technique helps to improve gain performance and is based on a small positive feedback. It is a common opinion that circuits with positive feedback should be avoided since the sensibility due to the environment (i.e., temperature) or to the inevitable process variations may bring the circuit to instability. This argument is true in a general debate on amplifiers, but a small positive feedback

Figure 5.17  Cascode configuration with input matching network.

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Figure 5.18  Cascode LNA configuration with feedback resistor.

1

2

2

2

Figure 5.19  E quivalent small-signal model of the LNA with external resistance (MOSFET version).

can be in practice controlled very well and exploited for a relevant performance boost. The feedback is implemented by simple integrated resistors (Rex) between collector and emitter in HBT or drain and source in MOSFET devices as shown in Figures 5.18 and 5.19. The value of Rex has to be high enough in order to allow the transistor M2 to work in saturation mode (or active mode in case of bipolar transistor). The high value of Rex provides a moderate amount of positive feedback, while still fulfilling the stability condition. It can be noted that the noise contribution of Rex on the overall noise figure of the circuit can be neglected since it is placed in the second stage of the amplifier. 5.4.3.2 LNA with Inductance between the Transistors

The traditional cascode structure of LNAs can be thought of as a two-stage amplifier, with an interstage matching network in between the two active elements. By inspection, one finds that a series inductor can be used as interstage matching network. It allows to improve the noise and gain performances of the LNA, as shown in the following.

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By considering Figure 5.17, an inductance Lm is inserted between the drain of the transistor M1 and the source of M2, as shown in Figure 5.20. As mentioned before, the cascode topology is a combination of a common source (CS) and a common gate stage (CG) as depicted in Figure 5.21. An interstage matching network can maximize the power transfer between the two stages and helps also to improve the signal-to-noise ratio due to the increase of the signal level at the output of the first stage. In order to show the gain improvement, the steps that bring to the choice of the interstage inductance will be explained in this section. In greater detail, it is necessary to: 1. Calculate the output impedance of the common source stage. 2. Calculate the input impedance of the common gate stage.

Figure 5.20  Cascode LNA configuration with interstage inductance.

Figure 5.21  Equivalent small-signal model of LNA with inductance between the transistors.

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In order to proceed with the first step, the output imped­ance of the CS stage can be computed by means of the small-signal model of MOSFET shown in Figure 5.22. Rg1 is the parasitic resistor at gate, Cgs1 and Cgd1 are gate-to-source capacitor and gate-to-drain capacitor, respectively, and RL is the load impedance. Since Cgd1 connects the input and output of MOSFET, the Miller effect should not be neglected. According to Miller theory and to the sketch in Figure 5.23, the impedance Z(s) connected between the input and output ports can be converted to the input impedance Z1(s) and output impedance Z2(s). Z1( s ) =



Z (s ) 1 - A( s )

Z 2 (s ) =

Z (s ) 1 1A( s )

(5.31)

where the transfer function is denoted by: V (s ) A( s ) = o (5.32) Vi ( s ) After applying Miller theory to the small-signal model of common source MOSFET the transfer function is found to be:



RL æ ç 1 ö æ ç sC gd1 ç1 è A( s ) ÷ø Vo ( s ) = - g m1Vi ( s ) ç 1 çR + ç L 1 ö æ sC gd1 ç1 ç è A( s ) ÷ø è

ö ÷ ÷ ÷ (5.33) ÷ ÷ ÷ ø

Figure 5.22  Small-signal model of MOSFET device.

Figure 5.23  Feedback network (left) and Miller equivalent (right).

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which yields A( s ) =



Vo ( s ) sRLC gd1 - g m1RL = Vi ( s ) 1 + sRLC gd1

and

Z out

g m1RL -1 sC gd1 = Z 2 (s ) = g m1RL + 1

155

(5.34)

(5.35)

This result clearly shows that the output reactance of the common source stage is capacitive. Concerning the second step, to calculate the input impedance of the CG stage, we consider again the small-signal model of the MOSFET, as shown in Figure 5.24. The gate-to-drain capacitor Cgd2 introduces a capacitance into the input impedance due to the Miller effect. At the same time, the gate-to-source capacitor Cgs2 also affects the input impedance of the common gate stage. By neglecting, in the interest of simplicity, the overlap capacitance Cgd2, it is possible to evaluate the input impedance contribution due to the gate-source capacitor Cgs2 only: 1 sC gs2 Vi ( s ) = Vx (5.36) 1 + Rg2 sC gs2



I x ( s ) + g m2Vi ( s ) =



Vx (5.37) 1 + Rg2 sC gs2

Figure 5.24  Equivalent circuit for computation of input impedance of the cascode configuration.

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Figure 5.25  Matching network for the cascode configuration.

thus obtaining:

Z in2 =

V x ( s ) sC gs2 Rg2 + 1 1 (5.38) = » I x ( s ) sC gs2 + g m2 sC gs2 + g m2

Zout and Zin2 in the previous equations show that the reactance of both the output impedance of M1 and the input impedance of M2 is capacitive. The equivalent interstage matching network is thus introduced as shown in Figure 5.25. C1 is the equivalent output capacitor of M1 and C2 is the equivalent input capacitor of M2. Based on impedance matching theory, in order to maximize power transfer from M1 to M2 the input impedance of M2 should have a value equal to the conjugate complex of output impedance of M1. The interstage matching network can be realized by inserting a series inductor between M1 and M2. This series inductor, together with the shunted C1 and C2, creates a matching network that gives a good impedance matching between the two MOSFETs and suppresses signals at undesired frequencies as well. 5.4.3.3 LNA with Inductance on Gate of Common Gate Stage

A way to increase the stability of the LNA is to add a small inductance Lz on the gate of the common-gate stage (M2). In this case too, a typical cascode CMOS LNA has been used as reference, as shown in Figure 5.26.

Figure 5.26  Cascode LNA configuration with inductance on gate of M2.

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Figure 5.27  Equivalent small-signal model of Figure 5.26.

The stability factor k of an amplifier is defined as in (5.l8). An LNA is unconditionally stable when k > 1 and the determinant of the scattering matrix is, in magnitude, lower than 1. The input and output are matched to source and load impedance, respectively. Therefore, |S11| and |S22| can be approximated as 0. Moreover, it is desirable to decrease S12 to improve stability of the LNA since when S12 decreases, |S12S21| decreases too. Considering the k factor and the fact that the LNA is matched at both ports, when S12 decreases but S21 is kept constant, the stability factor k increases also and the LNA is more deeply in the stable region. By putting the notch frequency of S12 at the center frequency of the LNA, it is possible to provide maximum stability at highest gain, unlike usual cases in which stability decreases when gain is increased. To decrease the reverse gain S12 as much as possible, it is desired to shape S12 to get more isolation between input and output in the desired band while maintaining high gain and low noise that are the main features of the circuit. This has been done by adding a small inductance on the gate of the cascode device as shown in Figure 5.27. This inductor, along with the inherent capacitances of the device, provides a low impedance path at the desired band from output to ground, preventing it to leak back to the input. In other words, this LC network provides a zero in the reverse transmission path to reduce the effect of output signal at the input of LNA. Another important advantage of this method is that since the value of Lz is low, it allows using high-Q integrated inductors (Q > 20) with low parasitic resistance. 5.4.3.4 LNA with Π Input Matching Network

This technique, sketched in Figure 5.28, can increase the performance in terms of input matching network and area consumption. It enables a fully integrated LNA by use of a Π-input matching network that involves a low-value inductor. This reduces input losses and allows the LNA to achieve a low noise factor even with the use of on-chip inductors, whose Q-factor is typically significantly lower than off-chip inductors or bond wires.

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Figure 5.28  Cascode LNA configuration with P input matching network.

5.5 Local Oscillators The local oscillators involved in the RX chain must provide the RF signals used for downconversion. Stability over time and temperature are key features that they must exhibit, since frequency drifts or phase wanders would introduce frequency offsets affecting the performance of the demodulator. The latter is typically designed in the digital domain to compensate for a limited amount of frequency errors due to the limitations of phase-locked loops (PLLs). Any unwanted frequency or phase variation which cannot be corrected by the feedback loop reaches the demodulator and can critically hamper the performance of the receiver. As a consequence, a huge amount of research effort is devoted to the careful design of oscillators and their control loops. Overall noise and phase noise, in particular, are also of equal and paramount importance, as they propagate through the downconversion to the baseband and thus concur directly in degrading the overall signal-to-noise ratio (SNR). 5.5.1 Fundamentals The traditional theory of feedback amplifiers based on a closed loop gain yields the oscillation condition to be achieved when the loop gain is unitary (Barkhausen’s condition), which allows a signal to sustain itself while circulating in the loop. Such an approach is practically employed mostly in the design of lowfrequency oscillators, where feedforward and feedback networks are more clearly identifiable. The negative resistance approach [50, 51] is instead much more used in high-frequency oscillators design, since it is based on the scattering matrix of a two-port device, which embodies an intrinsic feedback through S12, and where

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the source and load terminations can be seen to immediately affect the transfer characteristic. The basic theory for designing simple analog oscillators based on the negative resistance concept descends thus from the general model of the two-port device described for the low-noise amplifier (Section 5.4.1). However, unlike amplifiers, oscillators need to work in an unstable region, so to produce a non-null RF signal even in the absence of an RF input. 5.5.1.1 Design

In order to allow a self-sustained state on the input port, the reflection coefficients must satisfy the condition ΓinΓS = 1, that places the poles of the input loop exactly on the imaginary axis. A similar condition can be derived for the output port. However, for a nonunilateral device (S12 ≠ 0 ≠ S21), fulfilling the oscillation condition at one port also implies that it is verified at the other port. In practice, one designs to have |ΓinΓS| > 1 < |ΓoutΓL| so to enhance instability (poles on the right half-plane). In addition, the source termination is typically implemented as a reactive resonating circuit, and thus |ΓS| ≈ 1, whereas the load must have a non-negligible resistive part in order to accept real power from the oscillator: |ΓS| < 1. The following steps outline a simple practical design procedure based on the negative resistance concept: •• Choose a suitable magnitude for Γout greater than unity. •• Draw the corresponding circle for ΓS. •• Identify the intersections of this circle with the unitary circle (|ΓS| = 1);

if no intersections exist, a different magnitude for Γout must be chosen or the circuit must be made more unstable. •• Choose one of the intersections as value of ΓS to provide and design a resonating network which exhibits it, possibly introducing a suitable matching network. •• Compute the corresponding value of Γout, which, in turn, implies an output impedance Zout = Rout + jXout (or admittance Yout = Gout + jBout) with a negative real part (the more negative, the higher the initially chosen magnitude of Γout). •• Choose a load impedance ZL = RL + jXL (or admittance YL = GL + jBL) to fulfill the oscillation condition at the output loop (Γout ΓL = 1): GL =



1 (5.39) G out

In practice, however, one introduces a sufficient margin

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G L G out = α Î + (5.40)



with a a positive integer chosen to be slightly greater than 1, which typically entails reducing the resistive loading. •• Verify that with the chosen ΓL, the input satisfies the start-up condition

Rin + RS < 0 (or Gin + GS < 0): if not, a different choice of ZL is necessary. •• Design a suitable matching network towards the following stages. The input’s resonating behavior defines the range of frequencies which are capable of sustaining a steady-state, which influence the spectrum of the output signal. A very high-quality factor is thus required for these resonators, which, in turn, implies bulky components and usually off-chip solutions. 5.5.1.2 Startup Condition and Steady-State

Concerning the start-up conditions, additional external signals can be sometimes used to provide a pulse that triggers the oscillation: a pulse has a wideband spectrum and thus will also excite components at the oscillation frequency, which will build up on themselves during the start-up transient, whereas most other components will decay. However, an oscillator must often work without any trigger signal and should therefore self-initiate the oscillation. To this purpose, instability plays again a role, as it also affects the self-amplification of the signal: with poles located exactly on the imaginary axis, no self-amplification would occur. When the poles are instead located on the right half-plane a signal would increase its amplitude indefinitely. This phenomenon allows thermal noise to autonomously trigger the oscillation and then lets the amplitude grow to acceptable values. The nonlinear behavior of a practical circuit kicks in when the signals reach the physical limits (cutoff or saturation) by moving the poles towards the axis hence effectively preventing the amplitude to grow uncontrolled and in the end settling to a steady-state oscillation. 5.5.1.3 Harmonic Distortion

For downconversion purposes, a single-tone sinusoidal local signal is the ideal output of a local oscillator. However, the intrinsic nonlinear behavior, as for instance clipping phenomena and self-mixing, is the origin of harmonic and spurious signals and basically affects the spectrum of the oscillator’s output, as shown in Figure 5.29. Harmonics are necessary when a harmonic oscillator is being designed, but any spectral component fS different from the required fC contributes to two inconveniences:

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Figure 5.29  S  pectra of the output from an ideal oscillator at fc (a) and of a real one (b) with spurious tones and phase noise.

•• It potentially allows downconversion of undesired bands: the oscillator

output, when used in a downconversion stage, enters the mixer and all its components thus mix with the RF received signal. To see why this happens, suppose the mixer to be ideal and that its output filtering is also perfectly tuned to allow through only components at fIF = fRF,0 – fc , so that the desired component at fRF,0 is theoretically downconverted and sent to the output. When the real spectrum of the local oscillator is used as downconversion signal, all the RF frequencies satisfying | fRF – fS| = fIF will be downconverted to the expected IF and thus will simply go through the filter, affecting the output signal and representing an in-band interference that cannot be removed by further filtering. It is therefore often critical that the output of the oscillator is clean of spurious tones. Output filters are often introduced to suppress undesired components. •• It degrades the efficiency of the oscillator: all the DC (bias) power of the oscillator that is not converted to the desired frequency is substantially wasted, as the output filter will suppress the unwanted components. The efficiency of an oscillator measures the ratio of the RF power of the desired component with respect to the overall DC consumption, and will thus decrease when spurious components are generated by the oscillator. Harmonic distortion measures the quality of the generated signal as the ratio of the desired component with respect to the highest harmonic. In order to reduce harmonic distortion, several methods are available. The simplest, and probably the most used, is to suppress spurious components by means of an output filter. Alternatives involve instead circuital modifications: 1. Sizing the transistors following specific known rules, such as [52, 53]; 2. Adjusting oscillatory amplitude levels so that the tank voltage is comparable, in magnitude, to the breakdown voltage of the transistors without driving them into saturation regime;

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3. Other specific techniques strictly dependent on layout constraints, stage topology, and technological process. 5.5.1.4 Phase Noise

The subject of noise in oscillators involves aspects from different fields, ranging from quantum mechanics to statistical considerations. For an exten­sive treatment of this topic, therefore, specific textbooks and papers are available in the literature [54–57]. As shown in Figure 5.29(b), the spectrum of a real oscillator may contain spurious tones, but also exhibits a spreaded power around the fundamental component, which should instead ideally be confined at a single frequency [impulse in the frequency domain, as shown in Figure 5.29(a)]. This behavior is due to the random phase fluctuations (jitter) of the output signal. A sinusoidal voltage at a fundamental frequency fC with a phase noise component at a frequency fm can be represented as: V (t ) = A cos(2π f C t + φm sin(2 π f mt )) (5.41)



where fm represents phase noise amplitude. Developing (5.41) and assuming that fm C2, the phase noise is reduced. In fact, as explained in [45, 63], the phase noise in a Colpitts topology is a function of the two capacitances

Figure 5.31  (a,b) Colpitts oscillators.

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(C1 and C2), the oscillation amplitude (VOSC) on the output node and the total equivalent noise current of the active device (In). The phase noise spectrum [see (5.44)] of the Colpitts oscillator is given by: L( f m ) =



|I n |2 æ C ö (VOSC f mC1 ) × ç1 + 1 ÷ è C2 ø



(5.46)

2

From this formula it is clear that, in order to reduce the phase noise, the equivalent noise current of the active device have to be minimized whereas the capacitance C1 and the ratio of C1 and C2 have to be maximized. As mentioned before, single-ended architecture is used prevalently in III-V-based technologies, whereas in silicon technologies the differential one is preferred. Figure 5.31(b) shows an example of a Colpitts differential VCO. It is worth to note that the parasitic capacitances of the active devices must be taken into account since their value is a few hundreds of femtofarads and they are in parallel with the capacitance of the varicap diodes, thus affecting the resonance frequency. 5.5.3.2 Hartley Oscillator

Hartley oscillator is another type of selective-feedback LC VCO. This circuit was invented in 1915 by American engineer Ralph Hartley. The tuned tank circuit consists, in this case too, of an LC resonance circuit connected to a gain device (between collector and base of a bipolar transistor or between drain and gate of a MOSFET device), as shown in Figure 5.32(a), producing a sinusoidal output waveform. Hartley topology differs from the Colpitts one due to the design of the LC tank. The resonant tank circuit is made by a variable capacitance C and the series of two inductances (L1 and L2), or an inductance with a center tap, which determine the oscillation frequency according to:

Figure 5.32  (a,b) Simplified Hartley oscillator.

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f OSC =



1 2π LtotC

with Ltot = L1 + L2 + 2 M

(5.47)

Also for the Hartley oscillator, the single-ended architecture is used prevalently in III-V-based technologies, whereas in silicon technologies (BiCMOS or CMOS) the differential one is preferred. Figure 5.32(b) shows an example of Hartley differential VCO. 5.5.3.3 Clapp Oscillator

Clapp oscillator is a type of selective-feedback LC VCO. It was first published by James Kilton Clapp in 1948 [64], even if according to [65] this kind of oscillators was independently developed by several inventors. In this topology, the tuned tank circuit consists of an LC resonance circuit connected to a gain device (between collector and base of a bipolar transistor or between drain and gate of a MOSFET device), as depicted in Figure 5.33. The resonant tank circuit is made by the series of an inductance (L) and a variable capacitance (C1) and the series of two capacitances (C2 and C3) that determine the oscillation frequency:

f OSC

1 = 2π LCtot

-1

with Ctot

æ 1 1 1ö =ç + + ÷ è C1 C 2 C 3 ø

(5.48)

As mentioned before, the single-ended architecture is used prevalently in III-V-based technologies whereas in silicon technologies a differential version is preferred. In particular, for a single-ended architecture, the Clapp VCO is preferred to the Colpitts one. In fact, in Colpitts topology the oscillation frequency is changed by the two varactors, C1 and C2. Nevertheless, these capacitances fix also the voltage division ratio and modify the oscillation conditions. In the Clapp VCO, instead, the oscillation frequency is tuned by means of a diode

Figure 5.33  Simplified Clapp oscillator.

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C1 ratio is kept constant. FiC2 nally, in differential architecture the variable capacitance C1 is located in parallel with the inductance in order to minimize the effect of its parasitic capacitance. varactor C1 in series at the inductance L so that the

5.5.3.4 LC Cross-Coupled

The cross-coupled LC VCO is probably the most common VCO designed in the world, particularly for silicon technologies. It can be conceived using both HBT or MOSFET transistors. Figure 5.34 depicts the schematics of two configurations in bipolar and MOSFET devices. In the cross-coupled pair the negative resistance is generated by a differential common-source buffer in which the input is capacitively cross-coupled with its output. This negative resistance compensates the losses of the LC tank circuit that are mostly related to the parasitic series resistance of the inductance. The tunability feature is provided by the two variable capacitances that can be implemented by varactor diodes. By considering the schematic shown in Figure 5.34(a) and the small-signal equivalent circuit of the bipolar transistor (which contains the parasitic capacitances Cπ and Cμ), the oscillation frequency can be described by the following formula [45]:

f OSC =

1 2π LCtot

with Ctot = C + Cπ + CCS + 4C µ

(5.49)

In the cross-coupled VCO with PMOS active load [see Figure 5.34(b)] it is possible to control the resistance of the load by changing the gate bias voltage (VG). The main drawback of this solution is related to the limited transconductance that is provided, which does not allow sustaining of the self-oscillation when the quality factor of the inductors is poor (e.g., Q < 10). In case of designing VCO in Si RF-CMOS processes, this problem is addressed by employing the

Figure 5.34  (a,b) Cross-coupled oscillator configurations.

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Figure 5.35  (a,b) Complementary cross-coupled VCOs.

complementary cross-coupled architecture, shown in Figure 5.35(a). In this case the differential transconductance is doubled so that the circuit can sustain the oscillation even with lower quality factor inductors. The rapid growth of RF-CMOS process has besides enabled several architectural solutions whose implementation in other technologies is extremely difficult, if not possible at all. This is the case of the complementary cross coupled VCO with active LC tank shown in Figure 5.35(a) [65]. It is worth noting that the research of always more complicated solutions in RF-CMOS processes is related to the poor phase noise of pure silicon technologies, if compared with silicon germanium or III-V-based technologies. Further in-depth analyses of the circuital solutions shown here can be found in the large literature concerning this field [5, 45, 66]. 5.5.3.5 Push-Push Oscillator

All the VCO topologies analyzed in this section work at the fundamental frequency and are limited to the maximum frequency where the active device, either HBT or FET, has a unitary power gain (fMAX). If it is necessary to have an oscillation frequency greater than fMAX, the harmonic frequencies have to be taken into account and the power at these frequencies has to be enhanced. The push-push oscillator is used to generate and amplify the second harmonic (or the even harmonics in general) and cancel the unwanted frequencies (the fundamental one and the odd ones) [67]. This topology consists of two suboscillators at the same fundamental frequency but related in phase by a multiple of p.

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Ssub ,1 = S n¥=0 an sin(nω 0t + φn ) (5.50)



Ssub ,2 = S n¥=0 an sin(nω 0t + φn + Dφn ) (5.51)

where Dfn = np. The resulting signal is the sum of the two suboscillators outputs and can be found to be: Sout = Ssub ,1 + Ssub ,2 = S n¥= 2,4,… 2an sin(nω 0t + φn )



(5.52)

In this way, the fundamental oscillation frequency and its odd harmonics are cancelled out and the even harmonics are combined in phase and enhanced [67–69]. There are several advantages in the use of this topology: 1. It is possible to design oscillators with an oscillation frequency greater than the fT and fMAX of the transistors [70]. 2. This solution allows to reduce the die area if compared with other topologies (doubler or multipliers in general), reducing the costs [71]. 3. Temperature sensitivity is lower, compared with topologies that oscillate at the fundamental frequency [72–74]. 4. By choosing the right node, it is possible to have at the output both the fundamental frequency and its double (in case n = 2). 5. Push-push VCOs have a low phase noise (at least 3 dB better than the other VCOs that work at the same fundamental frequency) [59, 75–80]. The push-push oscillator’s main drawback is related to the complexity of the layout implementation. In fact, for the optimum suppression of the fundamental and the odd modes, the physical symmetry has to be guaranteed at layout level.

Figure 5.36  Push-push oscillator based on Colpitts configuration [53].

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Figure 5.37  Push-push oscillator based on Clapp configuration [81].

Figure 5.38  Push-push oscillator based on cross-coupled configuration [82, 83].

Figures 5.36 to 5.38 show circuital schemes of push-push oscillators based on various basic configurations.

5.6 Mixers The act of mixing two signals means obtaining the mathematical multiplication of the two, filtering out the undesired frequencies. This can only be achieved by using a nonlinear behavior, such as the quadratic I-V law of field-effect transistors (FETs) or the exponential relationship of PN junctions [including diodes and bipolar junction transistors (BJTs)] or modulating one signal with the other (such as with controlled switches). Nonlinear characteristics are also exhibited under large-signal condition by amplifiers, such as in the case of power

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amplifiers, but these are typically undesired phenomena that must be properly avoided or compensated for while ordinarily designing such devices. Mixers are instead explicitely conceived to exploit nonlinearities, for instance, via suitable alterations of the biasing point by one of the driving signals, whereas the other input passes through this variable-bias stage. Another method is instead passing the sum of the inputs through a nonlinear characteristic. In either case, among possibly several side products, the resulting output contains one component depending on the multiplication of the two inputs. Other components are filtered out whenever possible. In a receiving chain, mixers are typically used for downconversion of the received signal by means of a local oscillator (LO). For heterodyne downconversion, the local oscillator has a frequency slightly different from the desired signal to downconvert (fRF). For direct downconversion instead, the LO frequency (fLO) is equal to that of the desired signals. Requirements for RX mixers involve: •• Low conversion loss (for passive mixers); •• Very low coupling RF-to-LO and LO-to-RF (to avoid self-mixing); •• Low high-order products (n · fLO – n · fRF for n > 2).

These requirements are also critically exasperated when dealing with direct downconversion. Nonlinearities are also necessary for frequency multiplication of a lowfrequency local oscillator up to a desired RF band. The principle of frequency multipliers is not dissimilar to the basic concepts behind mixing. On-chip integration is usually possible only when FETs are used as mixing elements, since tuning diodes typically require different doping profile than those commonly available in a commercial chip. A vast amount of literature is devoted to this subject, for example, [50, 61, 84], and only the basic aspects are sketched in this section. 5.6.1 Fundamentals A nonlinear current-voltage (I-V) characteristic can be approximated by its polynomial Taylor expansion with coefficients ci. Most nonlinearities in practical components are acceptably approximated by the first few terms:

i (t ) » c0 + c1v(t ) + c 2v 2 (t ) + … + c N v N (t )

(5.53)

If the applied voltage v(t) contains the local oscillator and the RF signal,

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v(t ) = VLO cos(2 π f LOt ) + VRF cos(2 π f RF t + φ ) (5.54)

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then the current will contain: 2 VLO V2 + c 2 RF ; 2 2 2. The two input signals (at fLO and fRF); 3. Harmonics of the input signals at 2 fLO and 2 fRF; 4. The product of the inputs, which involves two components: fLO ± fRF, of which the lowest one is usually the one to be retained for downconversion.

1. Continuous component: c0 + c 2

Filtering around fLO – fRF typically yields the desired downconverted output. As a particular case, moreover, downconverting to the baseband requires a lowpass filter. By definition, this allows DC through and thus the continuous component may introduce an output offset which must be removed in other ways. In general, mixing efficiency is proportional to the conversion gain (inverse of the loss, for passive mixers), which measures the ratio of the desired downconverted component to the RF input power: Lconv =



2 2 C 22VLO VRF µ c 2 (5.55) 2 VRF

A strong quadratic nonlinearity (high value of c2) enhances the mixing efficiency, as the required product is made more significant. Increasing the local-oscillator power is also an obvious option to increase the output power, but the overall efficiency (depending on the total power entering the mixer) usually does not follow this trend. Furthermore, higher power levels may excite higher orders of non-linearity, similar to the behavior of power amplifiers (see Section 6.2). Higher Taylor coefficients (c3 and above) are actually useful when operating on higher harmonics, but also introduce further products, some of which occur at low frequency (such as 3fLO – 3fRF), which may fall into the band of the desired output signal and are not suppressible by filtering. Noise in intrinsically nonlinear components is a complicated subject, and the theory available so far is often restricted to specific cases and has limited practical applicability. 5.6.2 Basic Architectures Mixing circuits can be classified as passive (involving diodes or transistors but used as resistive elements) or active (using transistors to provide further amplification). Furthermore, based on the number of nonlinear components and on the input/output ports one normally deals with:

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•• Single-ended mixers; •• Balanced mixers; •• Double-balanced mixers.

Their most common topologies are described in the next paragraphs. These simple architectures serve as gradual complexity steps to reach the most widely used configuration for RF Si-based mixers, the Gilbert cell, described more extensively in Section 5.6.3. Image rejection, moreover, as explained in Section 5.2.2, can be introduced with specific architectures such as the two-branch Hartley or Weaver [5], although involving a higher number of components. 5.6.2.1 Single-Ended Mixer with Diode

The simplest mixer circuit is shown in Figure 5.39 and represents the direct implementation of the concept outlined in Section 5.6.1. The input filters work as a sum node to obtain a superposition of the RF and LO signals in the form of (5.54). The resulting input voltage is fed to the biased diode, which thus acts by applying its exponential I-V characteristic. The corresponding Taylor expansion is in the form of (5.53). The diode thus presents a nonlinear impedance creating a voltage divider together with the source equivalent resistance. As a consequence, the voltage on the diode depends nonlinearly on the instantaneous value of the input voltage and the resulting voltage contains therefore a number of mixing products. A tuned filter is then introduced to suppress all the unwanted terms and output only the desired one as vo. The DC block capacitor and the RF-choke inductors are standard methods to decouple the biasing network from the signal path.

Figure 5.39  Single-ended diode mixer circuit.

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When used for direct downconversion, a directional coupler must be used instead of the filters to provide some isolation between the two inputs. An input DC-block capacitor must also be introduced to decouple these RF signals from the bias, since the output DC block would alter the useful component too; as a consequence, this scheme is not suited for direct conversion. 5.6.2.2 Single-Ended Mixer with Transistor

Nonlinearities from FET or BJTs constitute an alternative to diode mixers. The same principle of Figure 5.39 is in fact exploited by the typical commonsource (or common-emitter) configuration, shown in Figure 5.40(a). The two input filters isolate LO and RF and provide their sum as gate voltage. The nonlinear transconductance of the transistor then acts, modulating the drain-source current, thus yielding the mixing products in the drain voltage, which is thus filtered around the desired component. Differently, the common gate architecture, in Figure 5.40(b), uses the LO to drive the FET, which therefore acts as variable resistor. At one end of this resistor, the (small) RF signal is provided; the drain voltage contains the mixing products to be filtered and delivered on the output. The intrinsic decoupling between transistor terminals, moreover, allows saving the input filters of Figure 5.40(a), while still providing a sufficient isolation. Similarly, the cascode or dual-gate structure [Figure 5.40(c)] has the LO and RF inputs applied both as gate voltages on their own transistor. The resulting current on the common channel depends on the series of the two transconductances, themselves nonlinearly related to the input voltages. The mixing products on the drain are picked up by the output filter to select and output the desired component.

Figure 5.40  (a–c) Single-ended mixer circuits with FETs.

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5.6.2.3 Balanced Mixer

A development of the single-ended mixer is represented by the balanced topology, which aims at improving the isolation between LO and RF inputs. The balanced configuration requires two non-linear components and a hybrid combiner, as shown in Figure 5.41. Assuming that the reflection coefficients exhibited by the diodes towards the input are equal, the 90° hybrid coupler cancels out the reflected waves from the nonlinear elements and thus exhibits, ideally, perfect matching at the LO and RF inputs. The mixing products appear as a current through the two diodes depending nonlinearly on the sum of the input voltages. The usual filtering is necessary to suppress the undesired components. The structure, furthermore, has intrinsic suppression of the even-mode products, which thus relaxes requirements on the output filter. A variation of the circuit in Figure 5.41 is obtained by replacing the 90° coupler with a 180° hybrid [85]. This structure exhibits a similar nonlinear behavior, whereas ideally provides perfect isolation between RF and LO inputs. 5.6.2.4 Double-Balanced Mixer

A further improvement of the balanced topology is represented by the double balanced configuration, shown in Figure 5.42. The version with diodes [Figure 5.42(a)] works in its switched form by bringing diodes in conduction in pairs. The LO signal can be thought to be responsible for this switching; when it

Figure 5.41  Balanced mixer with diodes.

Figure 5.42  (a,b) Balanced mixer circuits.

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occurs, the polarity of the RF signal reaching the output is reversed. This behavior therefore acts as a multiplication and the output filter removes the unwanted products. The transistor version, shown in Figure 5.42(b), works similarly. The LO driver brings pairs of opposite transistors in conduction at any given instant; the overall effect is again a change of polarity of the RF signal into the output. Isolation for double-balanced mixers is typically higher than with other structures, though the achieved inputs matching may not be that satisfactory. Even harmonics of RF and LO are also effectively suppressed, thus yielding typically a good conversion loss. These double-balanced structures are easily convertible into a form with differential ports and are the skeleton of the Gilbert cell, described in Section 5.6.3. 5.6.3 Gilbert Cell The Gilbert cell mixer (also referred to as “four-quadrant multiplier”) is a double balanced mixer circuit mostly used in silicon technologies. It is based on a linear time-varying circuit to achieve time-domain multiplication capability and consists of a differential transconductance amplifier (Q5–Q6) connected to a Gilbert cell (Q1–Q4), as shown in Figure 5.43. This cell consists of two differential amplifier stages formed by emitter-coupled transistor pairs whose outputs are

Figure 5.43  Gilbert cell with bipolar technology.

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connected with opposite phases [86]. The output of the Gilbert cell mixer is the multiplication of two input signals: 1. The signal coming from the local oscillator (LO) and the one coming from the radio-frequency input (RF) signals in case of downconversion mixer; 2. The LO signal and the low-frequency input signal (IF) in case of upconversion mixer. This double-balanced mixer provides high performance in terms of conversion gain and noise figure. Moreover, thanks to the differential circuital architecture and the possibility to design a symmetrical layout, the circuit is able to cancel the unwanted signal at its output node. The differential architecture and the high conversion gain make the Gilbert cell mixer the most widely used mixer topology for RF and millimeter-wave integrated circuits in silicon technologies (while on the other side, in GaAs, GaN, and other III-V-based technologies passive mixers are preferred). The main drawback of this topology is its limited linearity, especially in technology processes with low supply voltage (i.e., 1.8V). Since the Gilbert cell is the core of the mixer, a brief description is given here, in order to better understand its potential. Figure 5.44 shows the core architecture of the Gilbert cell. By taking into account the simplified schematic of the Gilbert cell, the expression of the voltage signal at the IF output can be calculated as follows:

æ v (t ) ö vIF (t ) = Z L I OUT = Z Lα F I S tanh ç LO ÷ (5.56) è 2VT ø

Figure 5.44  Simplified structure of the Gilbert cell.

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æ v (t ) ö = Z Lα FG MvRF (t )tanh ç LO ÷ (5.57) è 2VT ø

with:

I S = I S1 - I S2 = GMvRF (t ) and I OUT = I O1 - I O2



(5.58)

where ZL is the load impedance and GM is the transconductive gain of the RF stage. The voltage of the signal (VLO) is applied to a differential transistor pair and it has to be sufficiently high to bring the transistors fully in conduction mode or in off-mode, alternately. In this way the input signal (VRF) is multiplied by a square wave. By considering the RF input signal: vRF (t ) = VRFL cos(ω RFt ) (5.59)



the output voltage VIF can be approximated by the following expression:

vIF (t ) =

2 Z Lα FGMVRFL (cos((ω LO - ω RF )t ) + cos((ω LO + ω RF )t )) + … π (5.60)

The main output signal consists of a signal at frequency fLO + fRF and a signal at frequency fLO – fRF. Moreover, if the signal that drives the Gilbert cell (VLO) is sufficiently high, the transistors Q1–Q4 do not affect, in a significant way, the linearity performance of the mixer. For a deep analysis of the Gilbert cell and the Gilbert cell mixer in particular, it is possible to refer to several standard textbooks, that is, [45, 61].

5.7 Analog-to-Digital Conversion In the world of digital communications, highly sophisticated modulation schemes are normally involved for their high bit rate at no bandwidth increase, and the digital RX processor is responsible for their demodulation. Furthermore, as more and more of the traditionally-analog functions are moved into the digital domain in order to relax hardware requirements, an extreme level of accuracy is necessary for sampling and quantizing the received signal for further elaboration by the DSP. The analog-to-digital conversion (ADC) of a baseband signal is carried out by ADCs driven by a clock upon which the analog input is sampled, and its digital representation is made available in the output lines.

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5.7.1 Effective Number of Bits The number of bits Nb representing a sample is a physical feature of an ADC and it defines the minimum variation which can be measured. Quantization, inherently, is a source of equivalent noise with respect to the analog instantaneous value. Even by matching the analog signal excursion to the ADC’s limits (full scale), the quantization noise alone can be shown to give a signal-to-noise ratio approximately of:

SNR q ,dB = 6.02

dB N b + 1.76 dB - PAPR dB + 3 dB bit

(5.61)

where PAPR is the peak-to-average power ratio, which is 3 dB for a pure sinusoid and between 12 and 18 dB for complex QAM signals. The converter itself, furthermore, introduces additional perturbations and the overall ratio of signal noise and distortion (SINAD) can thus be evaluated after the ADC. As an alternative indicator of conversion quality, therefore, an equivalent number of bits (ENOB) is often used, so that by plugging it instead of Nb into (5.61), the formula returns the total SINADq available due to the conversion. The ENOB typically decreases as the sampling frequency increases, thus implying that the overall conversion is less accurate. A given modulation (typically M-QAM) has a symbol error probability PE,s depending on the overall SNR with which the signal reaches the decision stage. The symbol error probability can be converted to a bit error probability PE,b. Often, the mapping of bits to symbols is chosen as with Gray’s coding, so that PE ,s . A forward-erroradjacent symbols only differ by 1 bit, so that PE,b = log 2 M correction (FEC) code is also applied at the expense of slight overhead, in order to compensate for some random errors which may affect the decided bitstream. Based on the chosen FEC scheme [typically low-density parity check (LDPC)], a threshold SNR can thus be defined to formally be sufficient to have in practice an error-free communication. Such SNR value is the one used (possibly with sufficient margin) for dimensioning the link budget and thus allot noise figures of the analog front-end stages. Complex modulations demand very high SNRs and the analog circuitry is the one which is most strained to meet them. In order to keep the ADC noise negligible with respect to the SNR due to the thermal noise already affecting the downconverted signal, one typically selects the converter to have SINADq larger than the link SNR. As an example, a 1024-QAM radio link is designed to provide an SNR of about 45 dB in order to practically offer an error-free communication. The quantization SINAD, therefore, is kept below 55 dB, which, in turn, requires about 11 effective bits.

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Figure 5.45  Sampling clock driven by the recovered symbol clock.

5.7.2 Sampling Frequency By the Nyquist criterion, sampling must occur at least at twice the maximum frequency of the signal to sample in order to avoid aliasing.1 In case of baseband downconversion, the lowpass signal (either the in-phase or the quadrature component) has its highest component slightly above the symbol rate, due to the rolloff factor of the shaping filter and to guard bands to avoid interferences with adjacent channels. This signal is thus sampled at slightly more than the symbol rate and possibly oversampled at twice (or more) times this value in order to allow an accurate reconstruction. In the traditional architecture, the sampling clock was obtained by a voltagecontrolled oscillator driven by the symbol-clock signal recovered within the demodulator, as shown in Figure 5.45. This structure flexibly adjusts the sampling frequency by using the current symbol clock, closely proportional to the bandwidth. As drawback, however, the feedback loop is long and thus may hinder the overall performance. Systems with similar implementations have been very common until approximately 1995. A more modern approach employs instead a free-running local oscillator as fixed clock at twice the maximum channel bandwidth, regardless of actual smaller bandwidths and with an oversampling of 2. This signal is then later processed within the DSP to compensate for sampling errors by means of numerical interpolation, which implements a much shorter feedback loop including the recovered symbol clock. The functional scheme is shown in Figure 5.46. For wide channel bandwidths (such as the 500-MHz ones of the E-band), the sampling frequency may thus be very high and the ENOB offered by a cheap ADC may not be sufficient to support even relatively simple modulations.

1.

Self-interference due to loss of information.

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Figure 5.46  F ixed sampling clock with numerical adjustment. NCO denotes a numerical control oscillator, the digital equivalent of the analog voltage controlled oscillator (VCO).

Narrower channels, conversely, allow for lower sampling rates, at which the ENOB of commercial converters typically meets complex modulations’ requirements. When dealing instead with a digital-IF downconversion architecture, the trade-off between resolution (ENOB) and sampling speed plays a major role. Although undersampling schemes suited for bandpass (IF) signals can be devised, they are seldom used in practice. Sampling a signal with bandwidth B centered around an intermediate frequency fIF thus implies the sampling frequency to be, Bö æ without oversampling, at least 2 ç f IF + ÷ . Typical IFs are two or three times the è 2ø bandwidth, and thus the digital-IF sampling rate is higher than its baseband counterpart (2B without twofold oversampling). As a practical consequence, the ENOB of commercial ADCs when used for digital-IF receivers is smaller. IF sampling, however, still retains a sampled signal with unused bands, which remain so even through numerical downconversion and until decimation. This implies that quantization noise spreads uniformly and can be thus partially removed out by digital filtering. In the end, a few equivalent bits can be gained and used to compensate, at least in part, the reduced nominal ENOB. ADCs manufacturers, however, are improving the performance of their highspeed converters (≥ 1 Gsps), answering the demands of the expanding software-defined-radio systems [87]. Sampling IF signals with a bandwidth of about 100 MHz around an IF of 300 MHz is today a viable option even for 2048 QAM, as noise filtering recovers about 2 bits of resolution above the nominal ENOB. 5.7.3 Noise Sources 5.7.3.1 Quantization Error

The ADC involves two main operations: sampling in time and quantization into a finite number of discrete levels. Sampling, ideally, does not introduce noise, as

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long as the Nyquist sampling criterion is fulfilled, since a limited spectrum can be preserved. The ADC, however, is intrinsically a source of noise, as information loss occurs due to quantization. The ideal staircase input-output characteristic (shown in Figure 5.47) of the ADC maps a range of continuous input values to a given output combination, with no possibility of redistinguishing the original intermediate values from it. V For an input signal V with uniform distribution between ± m covering 2 the full scale of the Nb -bit quantizer, the quantization error Q is uniformly disV tributed in ± Nmb +1 and it follows linearly the input while “wrapping” it within 2 the quantization error bounds. The variance of a uniform input signal can be V2 found to be Ps = m . Whatever its distribution, its peak power always takes on 12 V2 the expression Ppk = m . 4 The quantization error, instead, can be computed by integrating its squared value in just one half of the triangles and multiplying by their number (2Nb+1). The resulting quantization error variance is given by Pq =



Ppk Vm2 (5.62) = 2 Nb 12 × 2 3 × 22 Nb

The overall signal-to-quantization noise ratio for a uniform input is thus

SNRq =

Ps P dB = s 32 2 Nb = 2 2 Nb Þ SNRq ,uniform,dB @ 6.02 N b (5.63) Pq Ppk bit

Figure 5.47  A  DC ideal input-output staircase characteristic (left) and quantization error as a function of input level (right).

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In order to generalize this result to signals with different distributions, a few considerations are necessary. Assuming that the number of bits is sufficiently high, the quantization error remains uniform within each quantization interval. The sum of contributions from each is instead modified by a different weight depending on the distribution, that is the probability of the input signal of being within each quantization interval. The weights sum up to 1 and, as a consequence, the overall quantization power is still given by (5.62). The signal power, instead, can be computed through its PAPR as Ppk = PAPRPs and the resulting SNRq can be computed from the first equality in (5.63) as: SNRq =

Ps P 32 2 Nb = s 32 2 Nb = Pq Ppk PAPR

Þ SNRq ,dB

(5.64)

dB = 4.76dB + 6.02 N b - PAPRdB bit

which coincides with (5.61) when PAPR = 0 dB. 5.7.3.2 Static Errors

Another source of errors is the alterations of the ideal quantization characteristic, often known as static errors. Such differences are normally attributed to manufacturing tolerances within the die that shift the conversion thresholds (see Figure 5.48). A different average slope of the characteristic is usually referred as a gain error, whereas horizontal shifts denote offset errors.

Figure 5.48  ADC input-output characteristic with static errors.

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Nonlinear distortions can also occur in practice, typically due to nonuniform quantization intervals. A differential nonlinearity (DNL) is related to the width of each interval, whereas an integral nonlinearity (INL) measures the error of the quantization threshold with respect to the ideal function. The interested reader can find the precise definitions and other relevant information in [88]. These nonlinear errors are smaller than the smallest quantization interval for most high-end ADCs, and their contribution can be neglected with respect to other noise sources. 5.7.3.3 Sampling Distortion

Since the quantization may require a nonnegligible time, often comparable with the sampling period, the value must be sampled and maintained for the necessary time. This task is accomplished by sample-and-hold or track-and-hold circuits. The latter is by far more common and its model and normal operation is shown in Figure 5.49. While tracking, the capacitor follows the input voltage through the small “on” resistance of the switch, usually implemented as a MOS transistor. When the switch is open, the last stored voltage remains constant at the output, neglect­ ing discharge decay. The “on” resistance depends on the instantaneous voltage via the MOS controlled-channel resistance and thus introduces a nonlinear behavior. More complicated structures than a single transistor are often needed for high-speed ADCs. Furthermore, the tracking bandwidth depends on this lowpass circuit and the “on” resistance must be kept to a minimum, typically using larger transistors.

Figure 5.49  ADC track-and-hold module (left) and time response (right).

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Figure 5.50  Equivalent noise model of sample-and-hold module.

This implies higher capacitances and these affect charge distribution with the transistor during the “on” state (introducing an offset on the sampled value) and while “opening” the switch (a further nonlinear distortion depending on the instantaneous value). A detailed description of these phenomena is available in [89]. 5.7.3.4 Thermal Noise

The track-and-hold scheme of Figure 5.49 is also source of thermal noise due to the nonzero “on” resistance of the switch, shown in Figure 5.50. This resistance also affects the transition frequency of the lowpass network. Increasing the resistance, in fact, introduces a higher-voltage noise but concurrently reduces the cutoff frequency. The integral of this noise lowpass noise can be shown to be kT . independent on the resistance and follow the well-known rule E[|vn |2 ] = Cs In order to keep this noise close to or lower than the quantization noise, the minimum sampling capacitance can be computed and a lower bound for power consumption is hence available, as described in [90]. 5.7.3.5 Sampling Jitter

The behavior of the sampler component is source of a frequency-dependent noise, which, in practice, heavily affects the overall performance (ENOB) of the ADC. A jitter in the sampling clock introduces a perturbation of the sampling instant with respect to the ideal one and thus samples a slightly different value. The difference between the sampled value subject to clock jitter and the theoretical value is strictly dependent on the time slope of the signal. As shown in Figure 5.51, a steeper signal introduces a larger error than a slowly varying one. The maximum steepness is related to the maximum frequency component in the input spectrum and the resulting maximum SNR can be expressed as [91]: which shows the

SNR jitter =

1 4π f 2t 2jitter 2

(5.65)

1 behavior, which is the dominant reason for the ENOB to f2

drop as the input frequency increases (an example is shown in Figure 5.52).

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Figure 5.51  S  ampling jitter affects the sampled value proportionally to the slope of the signal (dependent on its maximum frequency).

5.7.4 Architecture The different architectures of ADCs, seen from their interfaces, differ mainly for the latency between sampling and when the digital signal is available at the output. Although a higher latency does not imply a lower sampling rate, some of these stages may either require additional circuitry to store the intermediate results or to slow down sampling until the previous conversion is complete. The main architectures are briefly introduced here, whereas a more detailed overview is given in [92] along with a comparison of several proposed implementations available in the literature. Table 5.4 summarizes the main features of the considered solutions. CMOS-specific design techniques can instead be found in [93]. Table 5.4 Comparison of Various ADC Architectures Architecture

Speed

Number of Comparators

Issues

Flash

Very High

∝2Nb

“Bubbles”

Folding Pipelined Successive Approx. Time Interleaved

High Medium Low Very High

∝Nb ∝Nb 1 Depending on stages

Folding circuit Error propag. Error propag. Synchronization

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Figure 5.52  M  easured ENOB of a commercial 12-bit ADC (Texas Instruments ADC 12J1600) as a function of input frequency.

5.7.4.1 Flash ADCs

The flash converter is the one with the lowest latency, as it involves an all-parallel architecture made by 2 Nb comparators with increasingly high thresholds, as shown in Figure 5.53. In its simplest form, at any given sample the most significant comparator with high output is selected by an encoding circuit and its

Figure 5.53  Flash ADC structure.

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corresponding digital code is set on the output. A sample and hold module is not necessary, theoretically, as the conversion requires a single step, and is thus complete within one clock cycle. However, the comparators may set their outputs with different speeds and thus potentially create an unstable code before settling to the final value. The decoding circuit, moreover, can be designed to minimize errors of nonmonotonic outputs (“bubbles”) that typically occur close to the sampled value. The input impedance of this structure, if directly fed without sample and hold, is typically very low due to the large amount of comparators. Typical flash ADCs have a moderate number of bits (≤ 8) as the occupied chip area and consumption is proportional to the number of components, growing exponentially with the number of bits. 5.7.4.2 Folding ADCs

In order to improve resolution of flash ADCs while mantaining a low latency, folding ADCs have been introduced. The principle is increasing the number of comparators linearly with the number of bits. The functional scheme is reported in Figure 5.54: the coarse ADC provides the most significant bits of the conversion, whereas a folding circuit wraps the input value so that its output has an excursion equal to the least-significant bit (LSB) of the coarse converter. The fine ADC thus provides an additional set of bits corresponding to the underresolution of the coarse converter, which therefore constitute the least significant bits of the folding ADC. The folding circuit may introduce additional delay, and a corresponding one should be therefore inserted in front of the coarse ADC in order to match the sampling times of the two. 5.7.4.3 Pipelined ADCs

The pipelined architecture is a further step in reducing the number of components at the expense of the higher latency. The basic scheme is shown in Figure 5.55.

Figure 5.54  Folding ADC structure.

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Figure 5.55  Structure of a pipeline ADC.

Each stage operates with a 1-clock delay with respect to the previous one and receives as input the “remainder” after the previous conversion. The front stage thus works as a low-resolution ADC and creates the first few most significant bits. This value is converted back to analog via a digital-to-analog converter (DAC) and subtracted from the original value and passed on to the successive stage properly amplified. At the next clock, the successive stage stores the difference (through its own sample and hold) and converts it into a few bits more. In order to output a coherent word, an output digital buffer must properly delay the bits of the first stages until a sample reaches the end of the pipeline. Nonidealities in the DAC, however, can significantly affect performances, as their errors cumulate at each stage. 5.7.4.4 Successive Approximation ADCs

With a similar concept to the pipeline ADC but with a single stage sequentially performing the comparison, the successive-approximations ADC implements the conceptual scheme depicted in Figure 5.56. The successive-approximation converter is, in practice, often based on a single comparer, which outputs 1 bit at a time. The output bit is converted into the corresponding voltage and accumulated to the previous reference level, thus obtaining a new reference level fed back to the comparator.

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Figure 5.56  Structure of a successive-approximation ADC.

Figure 5.57  Successive-approximation ADC in exponential capacitors implementation.

One common implementation is also shown in Figure 5.57. The chargedistribution converter is based on a network of capacitors with exponential values which are initially all charged to provide the opposite of the sampled value and then charged or discharged based on the comparator output. In order to avoid the exponential capacitances, a C – 2C ladder network can be used instead [94]. Successive-approximation ADCs can drain very low power and are extremely compact in terms of chip area. The slow conversion process makes it unsuited for high-speed applications but turns out to be a good option as subcomponent of pipelined or time-interleaved ADCs. 5.7.4.5 Time-Interleaved ADCs

In order to speed up the conversion of slow ADCs, the interleaved solution basically provides very high sample rates by using several parallel ADCs driven by a sequential clock. In this way, each sample and hold module works at a slower rate, given by the high sampling frequency divided by the number of parallel

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Figure 5.58  Structure of a time-interleaved ADC.

ADCs. The digital outputs are then selected sequentially by a multiplexer, as shown in Figure 5.58. Interleaved ADCs often provide moderate resolution ( ROPT the voltage reaches knee or breakdown before current is at full swing. Both conditions correspond to a reduced power with respect to the ROPT case. The voltage and current waveform and the corresponding load line (i.e., the plot of current waveform versus voltage waveform curve) on the output characteristic for a class A device at full drive are shown in Figure 6.14. The maximum output power generated by a class A power amplifier is easily evaluated: POUT,max =



1 VDS,br - VDS,k IDSS (6.21) 2 2 2

The maximum DC power absorbed from the supply is instead given by the product of average current and the value of the DC voltage supply VDD = (VDS,br – VDS,k)/2: PDC,max =



VDS,br - VDS,k IDSS (6.22) 2 2

It has to be noticed that, in class A, the DC power supply PDC is independent of the power level. The maximum efficiency is then achieved at maximum output power, and, neglecting the knee voltage, its value is 50%. At the same

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Figure 6.14  C  lass A PA waveforms at full drive with ideal device. Output characteristics with dynamic load line (left) and time domain waveforms (right).

time, it is easy to derive the power dissipated by the device, that corresponds to PDC,max and PDC,max/2 at zero and maximum input drive, respectively. This behavior suggests a rather weird behavior of the class A device that is called to dissipate the maximum power when no input is applied, a state where it hence reaches its maximum temperature, while the coldest thermal regime coincides with the maximum output power conditions. It has to be noticed that the theoretical maximum efficiency of 50% is never achieved in practical implementations, and this is due to several factors. The knee voltage plays an important role, since it typically reduces the achievable voltage swing of 10%–20%. Moreover, the losses of the networks at the output of the PA reduce the output power, and the efficiency accordingly. As a result, realistic values of saturated efficiency in a class A PA are around 25%. For other PA classes, the same saturated efficiency scaling can be expected. When the bias current is reduced, the device works in class AB, B, or C: always referring to maximum input drive, that is, when vGS grazes zero, the iDS results as a truncated sinusoid. This situation is referred as a reduced conduction angle, where the iDS can be written as a function of conduction angle f, and of quiescent current IDD:

ìIDD + (IDSS - IDD )cos(θ ) iDS (θ ) = í î0

if – φ /2 £ θ £ φ /2 (6.23) otherwise

Table 6.3 reports the correspondence among f and different PA classes. The value of f can be related to the quiescent current:

æ IDD ö æ I /I φ = 2 arccos ç = 2 arccos ç DD DSS è IDD - IDSS ø÷ è IDD/IDSS - 1

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ö (6.24) ÷ø

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where is highlighted that IDD is commonly expressed relatively to IDSS. The cut iDS waveform contains harmonic components that are filtered by the output resonator of Figure 6.13 in order to provide a purely sinusoidal output. For this reason, these PA classes are referred to as tuned load class AB/B.3 It has to be noticed that also the vDS voltage becomes sinusoidal. Figure 6.15 shows the waveforms and dynamic load line in the class B case. In class B, fundamental components of drain current and voltage are identical to the class A case. As a result, both maximum output power and optimum load coincide for class A and tuned load class B. The power absorbed from the DC supply in class B is: PDC,max =



VDS,br - VDS,k IDSS (6.25) 2 π

Thus, the maximum efficiency in class B is p/4 ≈ 78%. It has to be noticed that in this case the dissipated power is null for zero input, meaning that its power management is improved with respect to the class A. Regarding instead the gain in class B, it is theoretically 6 dB lower than in class A, since the same output power is obtained for a doubled input drive. The evaluation of maximum output power, efficiency, and dissipated power in a class AB PA depends on the value of f. Figure 6.16 shows, for class AB bias points, the values at full input drive of the maximum efficiency and of the output power and absorbed power normalized to PDC,max in class A. It can be noticed that the efficiency monotonically decreases from class B to class A, as well as the DC absorbed power increases. The output power at fundamental is not constant, and has a maximum in class AB: this situation is verified selecting RL accordingly, to avoid early vDS saturation. It is also interesting to consider as a figure of merit the product of efficiency and normalized fundamental output power: Figure 6.16 reports, on the right graph, this value together with efficiency, and it can be noticed that there is a wide range of class AB bias points close to class B (i.e., deep class AB) sharing almost the same efficiency but with a quite flat product efficiency times normalized power. 6.2.3.1 Class AB Linearity

To have a rough estimation of the linearity of the aforementioned PA classes, the gain versus input drive behavior can be observed: however, some important considerations are mandatory. In class A and B PAs the circulation angle is independent of the input drive level, respectively, f = 360° and f = 180°. In this condition the PA gain (i.e., the ratio between the output and input power at fundamental), is constant. The situation is different for class AB PAs where the circulation angle depends on the input level. For example, f = 360° when vGS is 3. Other methods to reject harmonics are available, as the push-pull PA. Its usage is limited at high frequency by many factors: a description of push-pull pros and cons can be found in [1].

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Figure 6.15  C  lass B PA waveforms at full drive with the ideal device. Output characteristics with dynamic load line (left) and time-domain waveforms (right).

Figure 6.16  V  ariation of class AB point: impact on performance at full input drive. Top axis: conduction angle. Bottom axis: IDD/IDSS. Efficiency, DC absorbed power, and output power (left). Efficiency and product of efficiency and output power (right).

small enough that the device never is pushed below the pinch-off, and then it decreases if this condition is no more verified. This is enough to conclude that the amplitude of the fundamental component of the drain current that is related to the conduction angle depends on the input power level; in other words, the gain cannot be constant and decreases at the increase of the input driving level. Figure 6.17 reports on the left the constant gain obtained for class A and B amplifier as

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Figure 6.17  G  ain versus input drive for different bias points. Ideal model with constant piecewise trans-conductance (left). Realistic model (right).

a function of the normalized driving level (IBO) together with the gain of a class AB that instead exhibits strong gain variation. Figure 6.17 shows, on the right, a typical gain versus IBO behavior for different bias point for a microwave device, not approximated by an ideal trans-conductance. In class A compression starts for high IBO, while class B has a large gain expansion before starting its compression. Among the different bias points, in this case, the class AB with IDD = 10% IDSS has the most linear gain profile. This fact, combined with the good efficiency and power behavior, makes the class AB the preferred choice for quasi-linear PAs. While in base stations deep class AB biases are widely exploited to take advantages from IMD sweet spots, for backhaul application, higher bias current is generally adopted [8]. The reason relies on the more stable linearity figures across different devices this choice ensure, since the possibility to finely tune bias and power for linearity sweet spots’ exploitation becomes unfeasible in a relatively simple system as a microwave radio. Usually, a fixed drain current bias control is inserted in these systems to reduce the unacceptable dispersion adopting fixed gate voltage setting. Finally, if necessary, digital countermeasures can be adopted for bias optimization. 6.2.3.2 Limitations of Back-Off Efficiency

Figure 6.18 (right) shows efficiency versus IBO for class A and B in an ideal device, neglecting knee voltage effects. To understand the different behavior in the two cases, it has to be considered that, in both, output power is proportional to input power, while the PDC differs. PDC is given by the product between DC components of drain current, whose plot versus input drive is represented in Figure 6.18 (right) and voltage. The average drain current, corresponding to the DC from the power supply, is constant at IDSS/2 for the class A, while for the class B increases proportionally to the input drive. Regarding power supply voltage, in both cases it is constant at VDD. Let’s try to understand the reason of the slower efficiency degradation with back-off of class B with respect to class A. The

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Figure 6.18  C  lass A versus class B. Efficiency versus IBO (left). Average drain current IBO (right).

versus

cause of efficiency drop with back-off is related to the fact that when the input driving level decreases the output power is decreasing faster than the DC from the battery. In class A DC power is straight constant, therefore the efficiency linearly follows the inverse of the input level. In class B the DC follows the behavior of the output fundamental current that relieves the efficiency drop with backoff with respect to class A, resulting in an efficiency drop proportional to the square root of the input driving level. If PAs existed with DC and voltage both modulated by the input driving level, they would have constant efficiency independently of the power level, that would be perfect to handle high PAPR modulations. A further insight is given observing Figure 6.19, that reports dynamic load lines and waveforms at full drive and back-off for a class B. The product

Figure 6.19  W  aveforms (left) and dynamic load lines (right) for a class B ideal device with neglected knee voltage, at full input drive and half input drive.

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between drain voltage and current, which represents the instantaneous dissipated power on the device, is also shown, and it is proportionally much higher at backoff than at full drive. It can be noticed that drain voltage not reaching zero at back-off is the cause for dissipated power increasing and efficiency degradation. This problem is well known by the PA community, which has explored several solutions to mitigate the efficiency degradation at back-off. In this text, the Doherty PA and the envelope tracking are briefly introduced. They both try to minimize the dissipated power at back-off by allowing the drain voltage to reach zero at its minimum, but they apply this countermeasure through radically different approaches. While Doherty strategy can usually be seen as a microwave, analog technique, that permits a direct class AB replacement, the envelope tracking requires a system level modification of the TX system. Their block schemes are represented in Figure 6.20. The Doherty PA, introduced in [9], uses the load modulation to increase the voltage swing at back-off. The basic Doherty is composed by two devices, the main and the auxiliary, which drive current into a common node. Typically, maximum efficiency is obtained at saturated power (POUT,SAT), and at 6 dB of OBO (POUT,SAT/4). For output power lower than POUT,SAT/4 (i.e., half of maximum input voltage), only the main device, biased in class B, is active, and it is loaded with 2ROPT. This permits, at POUT,SAT/4, to maximize the voltage swing at half maximum current, see Figure 6.21. For POUT,SAT/4 < POUT < POUT,SAT, the auxiliary device is on and injects current into the common node. This increases the total output power of the Doherty, and it gradually modulates, from 2ROPT to ROPT, the load seen by the main device. The correct load modulation is guaranteed by proper dimensioning of the common load impedance and impedance inverter. Input splitter and phase delay line provide the correct phase and amplitude relationship of currents at the common node. The accurate turning on of the auxiliary stage is obtained by some sort of gain control or, more commonly, by biasing the auxiliary device in class C. Several modifications to the basic Doherty structure have been proposed in the years, all aimed to optimize some of its features and to overcome its inherent limitations. A more ample discussion on Doherty PAs can

Figure 6.20  Block schemes of Doherty PA (left) and envelope tracking architecture (right).

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Figure 6.21  W  aveforms (left) and dynamic load lines (right) for the main stage of an ideal Doherty with neglected knee voltage, at full input drive and half input drive.

be found in [1, 2, 10, 11]. Specific examples of Doherty PAs for backhaul applications are shown in [12–15], and some of them are presented in details later in the chapter. These prototypes are still at research level, since their introduction in commercial products is still limited by the intrinsic bandwidth and linearity issues of the Doherty technique, as pointed out in [4, 16, 17]. The envelope tracking approach uses a linear amplifier stage as PA (e.g., a class AB), whose drain dynamic bias follows the envelope of the input signal (see Figure 6.22) to minimize the dissipated power. For example, to reach maximum efficiency when the input envelope is half of its maximum, the drain bias voltage can be reduced from VDD to VDD/2. The amount of drain bias reduction, the selection between a continuous or stepped control, and the technique for envelope amplifier realization are determined by the application, which fixes power levels, signal bandwidth, and linearity constraints. In general, there is a trade-off, for a given power level, between the bandwidth of the envelope amplifier and its efficiency.

Figure 6.22  W  aveforms (left) and dynamic load lines (right) for the microwave device of an ideal envelope tracking with neglected knee voltage, at full input drive and half input drive.

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The comparison between Doherty and envelope tracking can be carried out by reminding the main pros and cons of the two techniques. Figure 6.23 (left) compares the ideal efficiency of class B and Doherty PA, in its standard configuration (two-way Doherty), and with some solutions for first efficiency peak extension to higher IBO. Figure 6.23 (right) compares the efficiency profile of an envelope tracking PA with continuous voltage control and a PA with stepped voltage control. The latter is easier to implement because it requires a smaller bandwidth at the envelope amplifier side, but exhibits lower average efficiency. Regarding the possibility of extending the IBO at which the efficiency is maximized in both architectures some words must be spent. Doherty PAs for back-off extension have been explored, employing multistage or multiway topologies [11]. However, in MMIC implementations, a first efficiency peak at OBO higher than 7–8 dB is practically unfeasible, since that would require too complex PAs with inevitable degradation of the other figures of merit. In envelope tracking the high-efficiency region depends on the modulation depth achievable by the envelope amplifier: this has to be compromised with bandwidth and absolute efficiency level. Turning to the microwave bandwidth, it is one of the main issues when designing a PA. Among others, a clear band-limiting factor in Doherty PAs is the output impedance inverter that is commonly realized by means of a quarterwavelength transmission line, a resonant component by nature. However, as discussed in [18], the adoption of GaN devices could mitigate the problem with respect to GaAs, thanks to a ROPT closer to 50 Ω and reduced capacitive effects. The great advantage of envelope tracking is that the microwave bandwidth is, at least in principle, not a concern, since the envelope amplifier is theoretically independent of the center frequency at which the microwave amplifier works. Thus, a wideband class AB can be adopted as microwave PA.

Figure 6.23  C  omparison of ideal efficiency (left) versus IBO for class B, Doherty, and envelope tracking PAs.

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However, at least in a first approximation, the Doherty PA does not have particular problems in managing wideband baseband signals, while in envelope tracking the main design trade-off is between signal bandwidth, output power, and efficiency. Finally, regarding linearity, both systems require particular attention. In the Doherty several factors could severely degrade linearity, such as load modulation, auxiliary device onset, and early main device saturation. Advanced bias control and additional predistortion become necessary to keep linearity under control. In envelope tracking, the drain voltage modulation introduces variations of the microwave device characteristics: the drain voltage modulation profile can be optimized for linearity, however, a high-order predistorter becomes necessary. Since backhaul PAs are asked to manage very wideband signals for medium power levels, the adoption of the cited techniques in this framework is still at the research level, since their efficiency advantages with respect to class AB PAs are soon canceled by the need of too-complex predistorters. However, the possibility of adopting these architectures is a challenging research subject that will meet the interest of the microwave backhaul industry, since power consumption is a great driving requirement and, thanks to the improvement of semiconductor technology, some of their limitations will be overcome. 6.2.4 PA Design Strategies PA designers, once identified a suitable device family and a proper class of operation, have still to face some big challenges. First, a real microwave device cannot be simply modeled as a nonlinear current source. As a consequence, an apparently straightforward operation like the identification of the optimum load becomes not immediate. Second, once the optimum load has been identified, it hardly coincides with the system impedance of 50 Ω. For this reason, matching networks have to be designed in order to provide the optimum load to the device, with low enough losses not to compromise power and efficiency. Third, since device peripheries cannot usually be chosen too big, more transistors have to be combined to achieve the needed output power. Similarly, to provide the needed gain, more stages have to be cascaded, requiring the design of interstage matching networks. Furthermore, big efforts must be spent in order to ensure and, if necessary, enforce broadband stability in large single conditions, to realize bias-tee networks that do not compromise linearity and to provide good input matching of the stage. 6.2.4.1 Devices for Microwave Backhaul PAs

To understand device models requirements to ensure the needed accuracy of the final PA designs, it is necessary to have a brief look to the main transistor technologies adopted for microwave backhaul. It is out of the scope of this text

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to analyze the solid-state technology of microwave devices in details. Good references can be found in [19, 20]. The introductory part of [21] gives an essential overview of solid-state technology for PAs. In this text, the main electrical characteristics and the most diffused nonlinear models of the transistor families that are, or will potentially be, used for backhaul PAs are summarized. The most adopted semiconductor technology for microwave backhaul PAs is GaAs. In particular, GaAs pseudomorphic high electron mobility transistors (PHEMTs) are at present market leaders for the realization of PAs. Briefly, a HEMT is a field effect device where the channel is a quantum-well (i.e., a depletion region inside an undoped material created by an heterojunction with a highly doped materials). The channel carrier mobility is not affected by the carrier scattering with the dopant and is hence extremely high. This channel, also referred to as two-dimensional electron gas (2-DEG), has very low resistivity if compared to a MOSFET’s one. The term “pseudomorphic” indicates that the doped and undoped layers have slight lattice mismatch that can be absorbed by some lattice strain. They are preferred to standard HEMTs since they allow larger bandgap and better performance. The PHEMT electrical behavior can be modeled by adopting the circuit topology of Figure 6.24 (or a similar one). The external elements (i.e., the parasitics), model the gate, drain, and source connections to the active region. This portion of the device is modelled as a voltage controlled current source (VCCS), where the drain current is a function of both gate-source and drain-source voltage, and by nonlinear capacitances that bridge the three intrinsic terminals and account for the nonunilaterality of the device. While the topology illustrated in Figure 6.24 is almost universally accepted to model high electron mobility transistors (HEMTs) or, even more generally, field effect transistors (FETs), the constitutive equations describing the nonlinear drain current and capacitances as a function of the gate and drain intrinsic voltages can be really different according to the adopted modeling approach. The modeling approach is also important for

Figure 6.24  Typical large-signal model for HEMT devices.

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the identification of the model parameters from the experimental characterization data. In fact, the choice of the model must be carried out considering that its accuracy, which is the expression of its potential capabilities, becomes effective only when combined with a well-defined and reliable model parameter extraction procedure. The Angelov model (see [22]) is one of the most adopted empirical models for HEMTs. It is based on the concept of charge conservation, which allows obtaining a very stable model from the numerical simulation point of view. It also includes self-heating effects through temperature dependence of some model parameters. The EEHEMT model is also widely employed, and it offers a larger set of options with respect to the Angelov model. Model extraction is a procedure that requires great experience and a large set of skills and instrumentation. Normally, multibias scattering parameters are adopted to characterize the parasitics and to obtain a first guess of capacitance variation versus voltage. DC characteristics are measured to model the drain current dependance on gate and drain voltage. Since HEMTs are characterized by a strong influence of temperature and by trap effects [23], pulsed DC measurements are also used to implement more accurate models. In general, a second refinement is carried out on load pull data or on prematched stages, in well-defined bias and excitation conditions. For example, one-tone input is adopted for maximum accuracy to predict the saturated output power; instead, two tones is adopted if a model with accurate IMD prediction is necessary. A final remark has to be spent on the importance for the designers to know what the model is able to predict, and what instead cannot be trusted with due confidence. If the model is recognized not to provide the required accuracy for a certain target, then either new information through measurements have to be acquired, or the design has to be accomplished with large enough safe margins. For MMICs, foundries typically provide scalable models accounting for device gate peripheries, adapting the model parameters to the device size inside certain given limits defined by the foundry rules, a solution that presents advantages in terms of management of models and simplicity, but inevitably leads to accuracy issues. Regarding other semiconductor technologies, in the last two decades, a great effort has been given to the development and improvement of GaN-based transistors. For microwave applications, this rather exotic material offers great advantages if compared to other technologies. From an electrical point of view, GaN has larger bandgap than GaAs, that leads to higher breakdown and power density HEMTs. The comparable maximum mobility allows, at least in perspective, to cover the same bands of GaAs. The very high power density of GaN brings however thermal problems, that can be mitigated adopting SiC substrates with astonishing thermal properties. GaN on SiC HEMTs are widely used at research level to demonstrate state-of-the-art PAs [24], while

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military and space applications drove the initial development of GaN PAs. In the telecommunications field, GaN HEMTs as laterally diffused metal oxide semiconductor (LDMOS) replacement represents a possible solution for basestations. Cost and reliability are still an issue of GaN devices. In the last few years, GaN on SiC 0.15-μm processes have been made available by foundries to third parties, disclosing the use of GaN for K-band applications. Some examples of GaN on SiC MMIC specifically thought for backhaul applications have been proposed in recent literature, mainly based on 0.25 mm technology [4, 13–15, 25, 26]. Qorvo has proposed a Doherty PA in K-band using their 0.15 μm process [12]. Despite being at the research level, the proposed solutions already offer outstanding performance with respect to GaAs counterparts. Modeling of GaN devices is even more difficult than GaAs, since thermal and trap effects need to be carefully addressed [27, 28]. However, since GaN and GaAs HEMT structure is rather similar, modifications of the well-established GaAs HEMT models are usually adopted [29]. Another available solution is the adoption of GaN HEMT based on Si substrate. If compared to SiC devices, the lower power density is balanced by the reduced fabrication cost, extremely important for mass production, and GaN on Si HEMTs for millimeter-wave applications have already been demonstrated [30]. For increasing frequencies and decreasing power levels, also silicon technology, in particular SiGe, could become interesting [31]. The great driving force is in this case the possibility of integration, which would eventually reduce cost. Working with a SiGe process is quite different with respect to MMIC design. Normally, also the computer aided design (CAD) environment changes, and the possibility to parameterize the design and to easily insert digital controls in the chip discloses new, even if different, design possibilities. Some examples of millimeter-wave microwave circuits and PAs in SiGe technology can be found in [32–34]. Regarding devices’ models, the millimeter-wave range choice clearly affects the characterization phase [35], and the adopted topologies are different with respect to compound semiconductor. In this case, scalability is a mandatory feature of models, since a big part of the design phase is spent in choosing the proper size and device’s ratio between the several stages. An overview of Si and SiGe modeling process for optimal use in CAD is given, for example, by IBM in [36]. 6.2.4.2 Optimum Load in Real Devices

Assuming that a device is well represented by its nonlinear model, a search for the optimum load can be carried out at the simulation level by using the scheme of Figure 6.25, where a CW single-tone test is performed. The intrinsic drain voltage and current (vDS,i and iDS,i, respectively) can be measured to plot the

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dynamic load line at the current generator plane. The maximum output power will be obtained when the curve is a straight line as shown in Figure 6.15. If the device is loaded with a purely resistive impedance Ropt, then the obtained load line would look as the dashed line in Figure 6.26. As discussed before, the dominant reactive part of the device impedance is capacitive. By tuning the imaginary part of the load admittance (BL), in particular moving it to an inductive load, the dynamic load line can be narrowed and brought very close to the target straight line. It has to be noticed that if an accurate model is employed, the curve cannot be perfectly straight. In fact, due to series parasitic effects, the output harmonic stopper resonator cannot be placed at the intrinsic device planes, and a small amount of voltage at harmonics will always be present at the intrinsic device terminal. This effect becomes more evident increasing the frequency. Finally, if

Figure 6.25  Schematic of a load pull setup.

Figure 6.26  Intrinsic dynamic toad lines in a class B load pull simulation. Effect of load susceptance (BL) tuning. Left: nonoptimum load; right: optimum load for maximum output power.

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the effect of series inductance is not negligible in band, a small retuning of the load resistive part could also be necessary. Since from previous considerations it is clear that the optimum load cannot be identified just observing DC or even pulsed small signal characteristics, two options can be followed for its identification: •• Through a set of measurements, a large signal model can be extracted

and, by the aim of CAD simulations, the optimum load can be identified.

•• Through load-pull measurements, the optimum load can be directly

identified. The load pull is, in general, the operation of varying the load condition of the device under test, monitoring its behavior and performance to associate (map) them to the corresponding value of the terminations. After that, load pull maps on the Smith chart can be defined, and contour plots can be used to identify the load, or a region of loads, that optimize the desired performance. Same considerations apply for the source-pull, where the microwave generator impedance is tuned to optimize PA response. Load pull data can also be obtained at the simulation level, a strategy that strongly relies on the adopted model accuracy, in the current operative conditions. Typically, when performing harmonic balance (HB) simulations, load pull is a quite simple operation to be set and performed, and microwave CAD softwares usually provide templates embedding load pull mapping facilities. At the measurement level, the principle of load pull remains the same, but all the issues of high-frequency measurements are present. A very extensive description of high frequency-measurements, and of load pull in particular, is given in [37–40]. The optimum load ROPT seen in the previous sections refers to optimum load for maximum output power. However, since the design target can be something different than output power maximization (e.g., a certain level of linearity or a best compromise between gain and output power), the term optimum load can be used in general to define the load condition at which the device provides the best desired performance. In both simulated or measured load pull, the figure of merit that defines the optimum load must be evaluated, and the appropriate input signals, bias points and environment conditions must be set during this evaluation. This includes the fact that it could be necessary a proper setting of harmonic terminations through harmonic load pull, since harmonic loads affect output power, efficiency and linearity. Some interesting efficiency enhancement techniques have been proposed by the PA community, such as class F, second-harmonic tuning, and class J [1, 2]. Class F and second-harmonic tuning can significantly improve the saturated efficiency. However, it can be seen that also back-off efficiency benefits from harmonic control. Class J is a very interesting solution since it allows, by exploiting

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the class B/J continuum, an easier wideband matching, relaxing the requirement of a purely real intrinsic optimum load. It is important to notice that, while class F can be in principle adopted with every device technology, second-harmonic tuning and class J are more keen to GaN HEMT devices, since they require drain voltage overshoot that would destroy a low breakdown voltage device. As a solution, the drain bias can be reduced, but this leads to a power reduction as well. The main problem in using techniques based on harmonic load control is that measured harmonic load pull, especially for frequencies above 10–20 GHz, requires quite advanced and rather costly setups. Concerning HB load-pull simulations, care must be taken when optimizing harmonic loads since normally nonlinear models accuracy for varying harmonic loads has not been experimentally. Designers have two options. The impedance at harmonics (generally second and third) can be kept low if compared to fundamental load impedance. In this way, even if the harmonic current generation is not well predicted by the model, the drain harmonic voltage can be minimized. As an alternative, designers can verify that the chosen harmonic loads do not degrade performance. In GaN MMIC, for C-band applications, some interesting tricks have been illustrated in [4] to avoid the need of very low-harmonic impedances. Some words must also be spent on the input matching of the devices. The classic design roadmap leaves the input matching network design after that the output has been well set. A solution that could be erroneously interpreted as indicating that input matching is not important, something that is instead completely wrong especially for wideband applications. PA gain is very important since a high gain reduces the demand in term of driver efficiency. Finally, a low reflection at the PA input, which represents the driver load, is necessary to ensure that the driver amplifier could behave as expected. This is particularly important when the driver is already available and cannot be optimized together with the PA. The input impedance of a microwave device is actually very critical to match to 50 Ω, because it has very low resistive and very large capacitive parts. For this reason, high-order filters are in general adopted at the device’s input rather than at its output. Moreover, harmonic control could also be applied at the device input to improve linearity, considering that the gate-to-source nonlinear capacitance contributes significantly to the PA distortion [1]. Conversely, to correctly apply techniques like second harmonic tuning, accurate input harmonic control is needed to provide the correct phase relationship between fundamental and higher-order harmonic [2]. 6.2.4.3 Input and Output Matching Networks

Considering narrowband conditions, matching networks could be designed by directly observing the intrinsic simulated waveforms, trying to maximize the output power. However, if load pull data only are available, and in general if a certain bandwidth must be achieved, other procedures are preferred.

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Designers prefer to support their CAD activities through linear simulations that, with respect to HB simulations, are faster and more keen to optimization algorithms and allow for accounting the process deviation statistics provided by the foundries. For different frequencies in the considered bandwidth, optimum loads can be derived from large-signal simulations as a first design step. Then the synthesis of these terminations across the band is carried out through linear optimization of passive structures able to transform the external 50 Ω in the required optimum loads. Figure 6.27 shows a sketch of this design approach. Following a similar approach, also the harmonic terminations can be considered, simply adding them during the optimization phase. Alternatively, in integrated PA designs, it is reasonably easy to extract a very compact equivalent circuit approximating the frequency behavior of the optimum load (see Figure 6.28). In fact, the complex conjugate of optimum load is in general well approximated by an R-C shunt, where the resistance ROUT is close to the intrinsic ROPT and the capacitance COUT accounts for the capacitive effects of the device. For high-frequency operation, a small series inductive effect Lout can be included to provide a better agreement. Even if this approach is based on an equivalent circuit that has no physical meaning, it represents a powerful tool for PA matching network design. In fact, the optimization of the matching network can now target the maximum power transfer between ROPT and the external load.4 Another advantage of this strategy is the possibility of its direct application to filter design and impedance matching techniques, giving a solid theoretical base to the design. Once the design bench is set, the choice of the matching network topology and elements can follow many directions. The designer experience is crucial, because every constraint influences the design choices. What can be said is that, since small-area occupation is always an important goal, semi-lumped approaches are preferred. For medium/high power PAs, the adoption of inductors in the final output matching is usually avoided, since their losses are quite high. Regarding capacitances, the usage of small value metal insulator metal (MIM) tuning capacitors is pursued if the process can guarantee good accuracy and repeatability in the insulator layer thickness. Transmission lines, especially for increasing frequency, can be used in a range of characteristic impedance between 60 and 85 Ω, that leads to reasonable small widths compatible with line bending for size optimization. Compensation of device’s output capacitance through a drain bias stub placed right at the device drain is apparently a convenient solution, but it must 4. This simulation setup does not give a formerly correct result on the output power degradation when ZA is not ZOPT, since power matching is different from maximum power transfer matching [41]. However, the approximation is very good and fine adjustments can be done in a nonlinear simulation.

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Figure 6.27  A  pproach for matching network synthesis. The design target is to obtain GA = GOPT for all the interested frequencies.

Figure 6.28  A  lternative approach for matching network synthesis. The design target is to maximize S21 of the cascade between equivalent output circuit and matching network. The equivalent output circuit reproduces the frequency behavior of G*OPT.

be carefully evaluated. Sometimes this approach does not guarantee the needed bandwidth, and the presence of a resonance could negatively impact on the sensitivity to process variations. As a practical, empirical rule, we can assume that if the shape ratio of the stub is unfavorable (i.e., the length is similar to or smaller than its width) or its phase length is only few degrees, probably it will give problems. Another viable solution is to absorb the device’s output capacitance in the matching network, using a longer drain stub that does not completely compensate the drain capacitive effects. Practically, this will lead to better bandwidth and, consequently, higher robustness. At limit, a quarter-wavelength stub terminated by a bypass capacitor can be adopted to supply the drain bias and to provide positive effect on second-harmonic rejection. In this case, attention must

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be given, since when the drain bias stub is too long or located too far from the device, stability issues can occur. Regarding the input matching, it has to be considered that the level of PIN is reasonably at least 6 dB lower than POUT. This value is low enough to enable using integrated inductors, and input matching networks can then be realized with lumped elements to save area. The input matching includes the gate bias network, which does not carry DC, except at very high compression, a situation avoided to preserve linearity. Input matching networks usually enforce also low frequency stabilization through large bypass capacitors (tens of picofarads). High-frequency stabilization networks instead adopt integrated precision resistors, in shunt or series configuration to the device gate. Particular care is given to the prevention of parametric oscillations that can occur at subharmonics [21], with the most critical frequency represented by half the fundamental. To this purposes, the gate bias networks are designed so that gates are loaded with impedances that are not too inductive (or at least with enough large real part) at these frequencies. This is in order to avoid the resonance, in a large-signal condition, of this impedance with the strongly nonlinear input capacitance. Since only one octave separates the fundamental and its half, and considering that at fundamental an inductive impedance is required to compensate the device gate capacitive behavior, a very accurate design of the bias filter is required. 6.2.5 Power Combining, Drivers, and Stability The output power is never too low: this is what system designers used to tell MMIC designers.5 However, GaAs power density is reasonably below 1 W/mm at microwave frequencies. To achieve saturated output power of few watts, a quite large periphery is needed; however a single device cannot be made too large. In fact, gate width is limited by process and thermal issues, and gate and drain series parasitics must be kept well under control. However, too many fingers on a single gate would lead to a device’s transversal dimension that is too large, with the risk of generation of spurious modes and oscillations. To overcome this issue, an array of transistors is deployed with some splitting/combining structures at input/ output so to add the power they are handling to reach the wanted total power level. For symmetry reasons transistors are normally combined in a number of the power of 2. Power splitting/combination is usually carried out by microstrip bifurcations, sometimes using Wilkinson dividers or at least resistors bridging gate pairs that do not intervene in nominal condition (with a slight change in the fundamental matching), but that help suppressing odd-mode oscillations caused by asymmetries between the several branches. 5.

Provided that nobody would exceed regulations limits that are in general quite optimistic.

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The device array is usually accurately optimized and, according to the particular design and designer’s experience, single devices can be prematched or paired directly. An alternative to this solution is to combine devices in a balanced configuration, using 90° couplers at input instead of bifurcations. Figure 6.29 clarifies the difference between the combined and balanced configurations. The input coupler splits the signal in equal magnitude between the two branches, which contain identical stages, but with a phase difference of 90°. The output coupler is flipped with respect to the input one, thus allowing in-phase recombination of signals from the two branches. The first and most recognized advantage of balanced PAs is the possibility of obtaining flat in-band gain with good input matching. The overall input matching, intrinsically ensured by the 90° balanced structure, relaxes the matching of the two combined PAs with a strong benefit in terms of gain flatness and bandwidth. Gain flatness can be achieved introducing lossy input matching or accepting some input mismatch at the lower frequencies of the working band. It is important to realize that, when the single PAs are badly matched, the combined one can remain well matched, even if the reflected power is dissipated by the termination required at the fourth port of the 90° coupler. Practically, Lange is the preferred 90° coupler in MMIC design up to 70– 80 GHz, where the process accuracy is good enough to obtain the needed lines’ width and spacing. Referring to GaAs process, 4-mil substrates are more keen to Lange realization than 2-mil ones, since they allow more reasonable width and spacing. Lange couplers provide a very good phase response versus frequency, and with accurate design, the classic “eye” can be obtained observing the transmission at the two ports. There are some drawbacks in the use of balanced PAs with Lange couplers. First, the two branches are fed with the same input power only at the frequencies where the coupler equally splits the power at its input. Therefore, a small unbalancing between the branches is always present. This creates some issues in compression, since one device will saturate earlier with respect to the other. From a layout point of view, Lange couplers can be quite space-consuming, and

Figure 6.29  Combined (left) versus balanced (right) PA approaches.

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in general require a determined positioning of the access ports.6 Moreover, they cannot be used for DC bias connection of the two branches, complicating the routing. Practically, most of MMIC designs are either completely based on bifurcations, or they have input and output Lange couplers that combine two PAs based on a multistage combined solution. Turning to the need of higher gain PAs, the solution is to add driver stages. The insertion of drivers in the PA integrated circuit requires the design of interstage matching, which is probably one of the most demanding phases. In fact, the very low input impedance of the power stage must be matched to the relatively high impedance at the output of a small device, thus requiring high-order filters to achieve the needed impedance transformation ratio. A very interesting approach is proposed in [42], where starting from Chebychev lowpass prototypes, some tips for achieving bandpass matching network for arbitrary impedance ratio are proposed. Another solution of interstage matching consists of insulating every stage with Lange couplers, thus reducing the impedance ratio and improving bandwidth. The main drawback in this case is the large area occupation. Power combination and driver insertion require a very careful stability analysis, because cross-talk could affect the active devices in the several branches yielding stability issues. In this case analysis considering a single device is obviously not exhaustive. Odd-mode oscillations are usually kept under control by using insulated couplers and combiners that help suppression. Nonlinear stability analysis is very difficult to be carried out, and thus usually linear stability analysis is adopted instead, and large enough safe margins are enforced. As stated before, parametric oscillations are prevented by careful gate bias network design at subharmonics. A last remark concerning the bias voltage distribution is that it has to be kept as uniform as possible along the device array to prevent unbalancing. 6.2.6 Backhaul PA Examples In this section, some examples of MMIC PAs for backhaul applications are shown and commented upon. Some of these solutions come from the academic world, and more details about their design and performance are available. However, for the commercial solution, the declared performance are presented while, observing the provided chip photograph, the most evident and characterizing design choices are discussed.

6. Their form factor can be changed introducing some curves, but this solution reduces bandwidth and inevitably increases the longitudinal dimension.

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6.2.6.1 GaN MMIC PAs for 7-GHz Backhaul

Figure 6.30 shows the microscope picture of a MMIC Doherty PA optimized for maximum efficiency at 7-dB OBO, with 5 W of saturated output power, and realized on 0.25-μm GaN on SiC Qorvo Inc. process (formerly TriQuint Semiconductor) [13]. For this PA, large-signal measurements, averaged over 16 MMICs, exhibit power gain of 10 dB, saturated output power in excess of 37 dBm (3-dB gain compression), together with drain efficiency of 47%, at 7dB of output backoff, and higher than 35% in a 350-MHz band. Figure 6.31 (left) shows the gain and efficiency versus output power for a 7-GHz single tone test: the Doherty efficiency profile is clearly identifiable. Figure 6.31 (right) refers to a modulated signal test performed with a 256 quadrature amplitude modulation (QAM) signal with a 7-MHz channel bandwidth and shows the spectra measured at input and output of the MMIC. With the application of digital predistortion (DPD), the PA respects the spectral emission mask provided by ETSI, and the resulting

Figure 6.30  Microscope picture of 7-GHz Doherty PA on GaN MMIC. Chip size is 4.6 × 4.7 mm2.

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Figure 6.31  T he 7-GHz Doherty PA on GaN MMIC. CW measured efficiency and gain versus output power (left). Measured spectra with and without predistortion in a modulated signal test (right).

average power is remarkably high. The design exploits the possibility, in MMIC implementation, of choosing an optimized device periphery ratio between main and auxiliary stages in order to set the first efficiency peak at 7-dB OBO. In particular, also accounting for some other nonidealities of Doherty PA, the chosen pair of devices is 4 × 100 μm and 10 × 100 μm, for the main and auxiliary, respectively. Second harmonic is controlled on the drain of the main device to enhance efficiency at back-off, while it is shorted at both device’s gate to enhance linearity. A Wilkinson splitter is adopted as a power divider. Another interesting feature is the use of series capacitors in the output section to create a single capacitances. This is due to the fact that this MMIC process uses passive components coming from a GaAs process, and MIM capacitors cannot handle the 30-V bias voltage of GaN. Figure 6.32 reports the complete circuit schematic of the Doherty PA. Figures 6.33 and 6.34 show the microscope pictures of two combined linear power amplifiers for 7-GHz microwave backhaul, realized in 0.25-μm GaN on SiC monolithic technology from Qorvo [4]. Both modules are based on a combined class AB structure conceived for maximum back-off efficiency and reduced phase distortion, very important requirements in backhaul systems to guarantee the needed linearity with low-order predistortion. Different secondharmonic loads are exploited in the two power amplifiers, leading to different performance in terms of output power, bandwidth, and efficiency. The first PA, referred to as tuned load PA, has the second harmonic shorted at drain, while the second PA uses second-harmonic tuning. The former is a more conservative approach that guarantees low distortion also if the nonlinear model is not able to accurately predict nonlinearity, while the latter removes the resonant behavior of second-harmonic stub enabling a larger bandwidth and higher efficiency. Theoretical analysis and simulations have been used to verify that the second-harmonic load has little influence on the linearity of the pro-

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Figure 6.32  A 7-GHz Doherty PA on GaN MMIC. Complete electrical schematic; transmission line electrical lengths refer to 7 GHz.

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Figure 6.33  M  icroscope picture of the 7-GHz class AB PA on GaN MMIC, with tuned load. Chip size is 2.8 × 1.9 mm2.

Figure 6.34  M  icroscope picture of the 7-GHz class AB PA on GaN MMIC, with second-harmonic tuning. Chip size is 2.8 × 2.8 mm2.

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posed amplifiers, and experimental results confirm this fact. Figures 6.35 and 6.36 show, on the left, the single-tone performance of the MMICs. The two stages exhibit a saturated output power in excess of 35 dBm and 36 dBm on 16% and 26% fractional bandwidth, respectively. The same figures show, on the right, the measured spectra of a modulated signal test, with 256-QAM modulation and 7-MHz channel bandwidth (PAPR of 7.4 dB). Compliance with the spectrum emission mask defined for the targeted application has been achieved through low-order polynomial digital predistortion, thus demonstrating the high linearity of the stages. The measured average efficiency in the presence of modulated signals is 18% and 24%, respectively. The comparison with the previously presented Doherty PA, realized in same technology and for the same application, shows that the two proposed stages need a simpler predistorter to achieve the

Figure 6.35  T he 7-GHz class AB PA on GaN MMIC, with tuned load. CW measured efficiency and gain versus output power (left). Measured spectra with and without predistortion in a modulated signal test (right).

Figure 6.36  The 7-GHz class AB PA on GaN MMIC, with second harmonic tuning. CW measured efficiency and gain versus output power (left). Measured spectra with and without predistortion in a modulated signal test (right).

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linearity required by standard specifications, and can work on a larger bandwidth. However, the advantages of the Doherty solution in terms of efficiency are remarkable. Figures 6.37 and 6.38 show the circuit schematic of the two PAs (transmission line electrical lengths refer to 7 GHz). The input signal is split into two 8 × 75 mm devices using a semilumped, lowpass Wilkinson power divider. The input sections are electrically identical, except of a small asymmetry introduced in the tuned-load PA to cancel possible high-frequency oscillations. The output section differs between the two stages: the tuned load PA realizes the second harmonic short at the drain by means of a loaded transmission line that also provides drain bias, while the second-harmonic tuned PA uses the combiner length and symmetrical stubs to synthesize the needed fundamental and harmonic loads. As in the Doherty GaN MMIC, for voltage-handling issues, two series capacitors are used to realize a single capacitance in the output section. 6.2.6.2 GaAs Linear PA for Ku-Band

In Figure 6.39 the microscope picture of the TGA2501 Power Amplifier for X-Ku point-to-point microwave radio applications is shown. The power amplifier is fabricated on Qorvo’s 0.25-mm Ku PHEMT process. Operating at 8 V of drain bias from 6 to 18 GHz, it achieves 34 dBm of saturated output power, 30% PAE, and 24-dB small signal gain. The PA is composed of seven equal devices, of which one is used as a predriver, two for the second stage and the remaining four for the last power stage. The structure is almost symmetric with bias pads on both sides of the chip, excepting for the stabilization network of the predriver and the output combiner/matching. The rather complex routing is probably exploited to assure flexibility in the device mounting. 6.2.6.3 GaAs Doherty PA for K-Band Backhaul

Figure 6.40 shows the microscope picture of a Doherty power amplifier for Kband point-to-point microwave radio, developed in Qorvo GaAs 0.15-mm PWR PHEMT monolithic technology [25]. Figure 6.41 reports the measured results on the Doherty PA. It exhibits, at 24 GHz in CW conditions, a saturated output power of 30.5 dBm, with an efficiency of 45% at saturation and 24% at 6 dB of output power back-off, together with a gain of 12.5 dB (Figure 6.41, left). System-level characterization at 24 GHz, with a 28-MHz channel, 7.5-dB peakto-average ratio modulated signal, showed full compliance with the standard emission mask, adopting a simple predistorter, with average output power of 23 dBm and average efficiency above 14% (Figure 6.41, right). The full electrical scheme of the Doherty PA is reported in Figure 6.42. Highly efficient driver stages on both the main and auxiliary branches have been designed to boost gain. This solution allows for low impact on PAE, if compared to a Doherty singlestage final amplifier preceded by an external driver. Matching network structures have been designed, according to simple equivalent circuit approaches, to obtain

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Figure 6.37  Circuit schematic of the 7-GHz class AB PA on GaN MMIC, with tuned load.

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Figure 6.38  Circuit schematic of the 7-GHz class AB PA on GaN MMIC, with second-harmonic tuning.

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Figure 6.39  Picture of the TGA2501. Size is 4.3 × 2.9 mm2. (Courtesy of Qorvo Inc.)

Figure 6.40  Microscope picture of K-band Doherty PA on GaAs MMIC. Chip size is 3 × 1.43 mm2.

the desired 10% fractional bandwidth. In particular, the output Doherty combiner embeds the devices’ output capacitance in the impedance inverter, while the interstage matching is based on a fourth-order bandpass filter. A branch-line hybrid is adopted as input splitter, providing the input delay line needed on the auxiliary chain. 6.2.6.4 GaAs Linear PAs for Ka-Band

A commercial example is the HMC693 by Hittite Microwave, a two-stage GaAs PHEMT MMIC 1-W power amplifier, which operates between 27 and 34 GHz. The HMC693 provides 17.5 dB of gain and 30 dBm of saturated output power at 23% PAE from a 5-V supply. By observing the microscope picture provided in the datasheet, it can be seen that the basic structure of the module consists of a balanced PA, with Lange couplers at input and output. Notice that the output insulator resistor is bigger than the input one: the reason is intuitive, that is, higher power must be dissipated at PA output for phase mismatch than at input for internal reflection. The total device count is 12: each balanced branch has

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Figure 6.41  K  -band Doherty PA on GaAs MMIC. CW measured efficiency and gain versus output power (left). Measured spectrum with and without predistortion in a modulated signal test (right).

three stages composed combining two devices. The periphery increases towards the output. Open circuit stubs are extensively adopted, probably instead of MIM capacitors that would result too small to be accurate and repeatable. There is a sort of mirror symmetry in the layout. For example, the input and output open stubs are dangerously close to the Lange couplers. However, symmetry is maintained, meaning that possible coupling between the Lange and the stub does not significantly affect the phase delay between Lange ports. Bias is given on both sides, due to the fact that Lange couplers are DC decoupled between direct and coupled port. Each balanced branch has not a total internal symmetry, since the bias stubs are placed only on one device output. Figure 6.43 shows the microscope picture of the TGA2575, a wideband power amplifier fabricated on Qorvo’s production-released 0.15-mm power PHEMT process. Operating from 32 to 38 GHz, it achieves 35.5-dBm saturated output power, 22% PAE, and 19-dB small-signal gain over most of the band. The PA operates at drain bias of 6 V and 2.1 A. Observing the MMIC from the input (left) to the output (right), it can be said that a balanced architecture is adopted. The power exiting every Lange port is split between four devices, each of them feeding two devices in the second stage, which contains eight devices. Each pair of the second stage feeds four devices in the final stage. The total device count is [2 (4 + 8 + 16)] = 56. The bias routing is at least complex: a guess is that large metal metallizations have been used to minimize the DC losses and guarantee a good drain bias uniformity across the MMIC. 6.2.6.5 E-Band PAs

Figure 6.44 shows the picture of a fully integrated 71–76-GHz power amplifier (PA) fabricated in a 0.12-mm SiGe Bipolar Complementary Metal Oxide Semiconductor (BiCMOS) technology [43]. The PA employs five common-emitter

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Figure 6.42  K-band Doherty PA on GaAs MMIC. Complete electrical schematic. Transmission line electrical lengths refer to 22.4 GHz.

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Figure 6.43  Picture of the TGA2575. Size is 4.1×5.4 mm2. (Courtesy of Qorvo Inc.)

Figure 6.44  P  icture of the SiGe BiCMOS E-band PA. Area: 2.1 mm2. (Courtesy of International Business Machines Corporation, © 2012 International Business Machines Corporation.)

stages with on-chip power combining to achieve power gain of 21 dB, 17.6-dBm output power at 1-dB compression, and saturated power of 20.1 dBm. The modified Wilkinson combiner that is used shows 0.5 dB of insertion loss. Small signal characteristics of the amplifier show peak gain at 72 GHz with 3-dB bandwidth of 17 GHz (24%). The PA’s bias is applied using digitally adjustable bias circuits with temperature compensation, and it consumes quiescent currents of 140 mA from a 2-V supply and 280 mA at 1-dB compression. The MMIC PA was fabricated in IBM’s 0.12-μm SiGe BiCMOS8HP process with cutoff frequencies of fT = 200 GHz and fMAX = 250 GHz for high-performance heterojunction bipolar transistors (HBTs). The five metal levels back end of line (BEOL) stack offers two thick 1.25μm and 4-μm aluminum top layers for low loss interconnects. The PA consists of two parallel paths unit cells, driven in-phase by a Wilkinson power divider at

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the input and summed by a similar combiner at their output. To our knowledge, a maximum output power of 20 dBm can be considered as the present limit for Si-based solutions in the E-band. The HMC-AUH320 from Hittite Microwave is a high dynamic range, fourstage GaAs HEMT MMIC medium power amplifier, which operates between 71 and 86 GHz. The HMC-AUH320 provides 16 dB of gain at 74 GHz, and an output power of 15 dBm at l-dB compression from a 4-V supply voltage (current consumption is 130 mA). The PA layout, available in the datasheet, looks simpler than the one presented before, but we can guess that, given the very high center frequency, hard graft must have been spent on careful optimization through EM simulations. The first three driver stages are single-ended HEMTs with increasing size, while the final stage combines two transistors. The driver part of the layout is nonsymmetrical, while the combined final stage is symmetrical. The power level is in this case similar to the one achievable with SiGe implementations. To further improve output power, more complicated structures must be adopted, however reducing the bandwidth. For example, Figure 6.45 shows a microscope picture of the Northrop Grumman APH668, a GaAs PA for the 71–76-GHz band. It provides a saturated output power of 28 dBm, with a typical linear gain of 19 dB. The drain bias is 4 V and 630 mA. By observing the layout, it can be noticed that a balanced configuration is adopted, but given the high frequency and relatively small

Figure 6.45  M  icroscope picture of the APH668. Chip size is 3.79 × 2.92mm2. (Courtesy of Northrop Grumman.)

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fractional bandwidth, a branch-line coupler is preferred as 90° power splitter. Every branch contains a single-stage predriver that feeds two HEMTs in the driver. Then the power is combined, and split into four HEMTs that compose the final stage.

6.3 Upconversion In the TX chain, the two streams of digital data (I and Q) that contain the information to be transmitted, together with all the necessary coding introduced by the modem and the baseband unit, are processed by a cascade of blocks that permit, at last, to prepare a signal suitable for transmission through the antenna. In particular, as resumed in Figure 6.1, the I/Q digital data are converted to analog signals and then upconverted in quadrature to the working RF frequency. The resulting microwave signal is then filtered, power amplified, and finally filtered again by branching filters before being fed to the antenna. In this section, the quadrature upconversion operation is analyzed, identifying the most important requirements of the main building blocks and their most common circuital implementations. Similar to what happens for the RX chain (see Chapter 5), the frequency conversion can be carried out in a single step (direct conversion or homodyne), or first to a convenient intermediate frequency (IF) and then, in one or more steps, to RF (heterodyne or multistep conversion). In the following, the two approaches will be shortly revised highlighting their general pros and cons, focusing then to the most adopted solutions for backhaul. 6.3.1 Direct Conversion (Homodyne) A general narrowband signal modulated both in amplitude and phase can be represented as: x(t ) = Ac (t )cos(ω c t + φ(t )) (6.26)



and conveniently rearranged in the form:

x(t ) = Ac (t )cos(ω ct )cos(φ(t )) - Ac (t )sin(ωc t )sin(φ(t )) (6.27)

where it is possible to identify the two quadrature baseband in-phase I and quadrature Q components as: I (t ) = Ac(t )cos(φ(t ))

(6.28) Q (t ) = Ac(t )sin(φ(t ))



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The amplitude Ac and phase f components of x(t) are directly given by: Ac (t ) = I 2 + Q 2

φ(t ) = arctan

æQ ö èIø

(6.29)

By controlling the I(t) and Q(t) amplitudes, amplitude and phase of the x(t) signal can be changed to implement simple amplitude or phase modulations or more complex modulation schemes. In digital transmitters, the analog I and Q signals are obtained from the corresponding digital streams through proper digital-to-analog conversion. In the homodyne approach, after the digital-to-analog conversion, I and Q are mixed with the cosine and sine components of a local oscillator (LO). The LO synthesized signal is normally realized with a voltage controlled oscillator (VCO) driven by a phase locked loop (PLL) locked to a reference.7 The LO is split to generate the in-phase and quadrature components (e.g., with a 90° hybrid coupler). After mixing, the translated baseband signals are added to generate the output RF signal. In Figure 6.46 the scheme of principle of the quadrature (or vector) modulator is shown. Notice that, at least in principle, in this approach the LO frequency is actually the desired RF carrier frequency and no intermediate frequencies are used. This technique is very popular thanks to its inherent simplicity, compactness, and eventually cost. Furthermore, the output spectrum contains only the wanted signal in the bands around the carrier and its harmonics, but no intermodulation components are present. However the approach presents some issues like I/Q mismatch, DC offsets, and oscillator pulling, not always of simple solution. I/Q mismatch results from the imbalance of the mixer pair and quadrature skew. The gain difference between the two signal paths of Figure 6.46 produces phase-to-amplitude modulation. Instead, deviation from the 90° ideal phase difference between the two LO components (Figure 6.46) introduces phase errors in the upconverted signal with nonexactly orthogonal I and Q upconverted components, and eventual distortion in the signal constellation. The mismatch can be reduced precharacterizing the quadrature upconverter and predistorting accordingly the input of the vector modulator in the digital signal processing unit. Quadrature

7. Modern transceivers are synthesized, meaning that a well stable reference oscillator is used to lock an RF VCO. Using this approach the transceiver frequencies can be selected by software across a broad range, changing the division factor within the phase loop chain. More details on this part of the system are given in Section 6.3.5.

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Figure 6.46  Direct conversion. Scheme of principle of the quadrature (or vector) modulator.

mismatch is more significant in direct conversion than in heterodyne converters (see Section 6.3.2) because of the larger effect of any impairment due to the higherfrequency translation associated to the single-step conversion. DC offset, also referred as carrier leakage, occurs when the analog baseband I and Q in the scheme of Figure 6.46 are affected by DC offset. The effect of this leakage is the addition to the upconverted signal of a fraction of the unmodulated carrier, leading to distortion of the signal constellation. It can be reduced increasing the baseband signal swing, but this negatively affects the upconversion mixer linearity. Some carrier leakage reduction systems are available in literature (see [44] page 234), based on detection and baseband correction of offset effects through additional DACs providing the required feedback. Oscillator pulling represents a peculiar phenomenon of the direct up conversion topology and probably its most important and studied issue. The output power of the PA at the end of the TX chain can reach large swings.8 A fraction of this output signal (not negligible) can couple by substrate, package, or printed circuit board (PCB) with the other parts of the module, including the LO. Since the center frequency of the PA is equal to the LO frequency, a periodical modulation of the LO output phase occurs. The impact of this unwanted effect on the scheme depends on several factors, including voltage and current swings of the LO, Q factor of the oscillator tank, and topology of the PA (single-ended or differential). Injection pulling can be avoided if the ω LO differs reasonably (more than 20%) from the ωRF. This is in open contrast with the main principle of the direct-conversion where ωLO = ωRF, unless some modifications are introduced in the classical direct-conversion scheme. To this end, a common solution is the adoption of a local oscillator frequency at 2ωRF followed by a frequency divider by 2. In this way the injection 8.

20 VPP for POUT = 1 W in a 50 Ω system.

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pulling is strongly reduced, even if residual effects can be present due the secondharmonic output of the PA. In this regard, the solution shown in Figure 6.47 further reduces pulling issues. The local oscillator is mixed with the results of a division of itself. Among the two mixing products the component at 3/2 ωLO is selected through filtering and used for the upconversion. The PA harmonics, at n3/2ωLO, cannot yield any pulling problems. With respect the previous solution, where the quadrature components for upconversion are obtained as by side product of frequency division, they have to be expressly generated in this case. The same idea of deriving the carrier frequency manipulating the LO can be carried out mixing two oscillators at frequency ωLO1, and ωLO2 or, more often, mixing the LO signal with a replica of it at different frequency (÷ 2, 4...). Despite these schemes completely eliminate pulling issues, elimination of the unwanted image component has to be addressed. 6.3.2 Heterodyne In heterodyne (or IF upconversion) systems, the upconversion is performed in more than one step, adopting intermediate frequencies (IFs) located at convenient frequencies. Baseband I and Q data streams are digitally upconverted to a first IF. After that, the analog IF obtained by a single DAC is mixed one (singlestage mixing) or more times (multistage mixing) to transfer the baseband modulation around the final RF. A key feature of this approach is the absence of injection pulling thanks to the distance in frequency between LO and RF frequencies. A schematic of a heterodyne upconverter is shown in Figure 6.48. In this scheme the mixed signal contains two frequency bands, that is, the desired RF located at ωLO + ωIF along with its image frequency band centered at ωLO –

Figure 6.47  D  irect conversion scheme for reduction of injection pulling based on doubled frequency of the local oscillator.

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Figure 6.48  Single-stage double sideband heterodyne upconversion scheme of principle.

ωIF that needs to be filtered out by bandpass filters to not interfere with the RF band. This method is often referred as filtering method. Digital generation of I and Q at IF strongly reduces gain imbalance and quadrature skews. However, the DAC requires higher bandwidth and is hence subject to errors such as harmonic distortion and passband ripple. Harmonic distortion arises from the mixing of the harmonics of the two (or more) local oscillators used for the upconversions. The spurious must be sufficiently suppressed by RF bandpass filters to ensure the required signal to noise ratio (SNR). In any case some IF filtering is necessary before final upconversion to RF. Bandpass ripple can be an issue, especially at IF, because the highpass filter needed to reject the image is close to the IF. To this purpose, multiple pole filters are required, which introduce high bandpass ripple (Chebyshev), or low ripple but poor selectivity (Butterworth). 6.3.3 Single Side-Band Converters A viable alternative to these approaches is the single sideband (SSB) mixing, in which both the injection pulling (homodyne) and image band (heterodyne) issues are overcome. Considering the trigonometric relationship

cos[(ω LO + ω IF )t ] = cos(ω LOt )cos(ω IFt ) - sin(ω LOt )sin(ω IFt ) (6.30)

the baseband I and Q signals are digitally translated at IF (IIF and QIF), converted to analog sine and cosine components, and then applied to a vector modulator (direct quadrature modulator), whose output pair are added to yield the final output RF. The translated I and Q components at frequency (ωLO – ωIF) are in phase opposition and hence cancel each other, leaving the single sideband (ωLO + ωIF) at the output of the vector modulator. Practical implementations of the SSB mixer are actually more complex than what could appear from the scheme of principle of Figure 6.49. In fact, in real implementations of this architecture, gain

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Figure 6.49  Scheme of principle of single-sideband mixer.

imbalance and quadrature skew prevent the complete cancellation of the image sideband at (ωlo – ωIF). The amplitude ratio between desired and unwanted single sidebands is often used as a FoM of the upconverter. Thanks to the absence of injection pulling and to strong reduction of the image band issue, many variants of this solution have been developed and are available in modern systems. 6.3.4 Mixers The mixer is the basic component of upconverters and downconverters. It is a device with three ports, the LO, the IF (that very often can start from 0 Hz), and the RF. In upconversion, the IF input mixes with the LO input, and its replica is provided at RF frequency. In principle, frequency mixing occurs not only in nonlinear systems, but also in linear, time-variant systems (e.g., switched circuits), but they won’t be considered here. Diode mixers, FET resistive mixers and active mixers are the three main mixer families: Table 6.4 briefly compares them in terms of the main FoMs. Conversion gain, noise, and linearity are the more important mixer FoMs that relate the RF output to the IF input. Conversion gain is the ratio between the RF output power and the IF input power. Noise is usually expressed in terms of noise figure, while linearity is given by means of OIP3 or CIMR. The indicated LO power level has to be supplied for Table 6.4 Comparison between Main Mixer Families Diode

FET Resistive FET Active

Conversion Gain Bad

Good

Very good

Noise

Very Good

Good

Bad

Linearity

Good

Very Good

Bad

LO level

Very High

Moderate

Low

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the mixer to provide the expected FoMs. The IF diode and FET resistive mixers are characterized by a positive conversion loss, or negative conversion gain, while active mixers have a positive conversion gain. However, passive mixers have superior behavior regarding noise and linearity. In this section, FET resistive mixers are analyzed in detail, since they represent a very common implementation of integrated mixers. Their performance is superior to that of active mixers in terms of linearity and noise, but also with respect to diode mixers some pros can be found, since, for example, FET resistive mixers achieve optimum behavior for a relatively lower LO signal level. Moreover, MMIC processes are optimized for FETs, while diodes are often provided with suboptimal layouts. Figure 6.50 shows the basic cell of a FET resistive mixer, the mixer core. The LO is applied to the gate of the transistor that has zero-drain bias (coldFET). The FET channel resistance varies at the LO rate, and can be exploited for frequency mixing. The gate is biased to optimize the cold-FET behavior, and the optimum bias varies according to the LO power level. The IF is applied at the drain side through a frequency diplexer that also serves to extract the mixed RF signal. Diplexer design can be carried out considering it as two separate filters, one centered at IF and the other at RF. These two frequencies fIF and fRF are normally far, so that the diplexer design is in principle not too complicated. When a mixer is properly operating, the amplitude of the RF signal is proportional to the amplitude of the IF signal only, with the LO just providing the dynamic bias. The mixer is usually hard-driven by the LO pump, a condition that guarantees the best performance in terms of linearity and conversion gain. To provide sufficient amount of LO signal to the device gate, an input matching at the LO frequency is usually adopted. For narrowband operation a series inductor on the gate is usually sufficient to compensate for the FET input capacitance. Then, a real-to-real impedance transformation is performed to reach the LO source impedance level. However, given the high Q of the gate impedance, a high-order filter is usually adopted even in narrowband mixers, to guarantee matching also accounting for the significant statistical deviations of the FET input impedance. On the IF port, the return loss is minimized through the matching provided by the diplexer. The matching network for the RF output is optimized to minimize the conversion loss.

Figure 6.50  Basic cell of a FET resistive mixer.

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To prevent leakage between ports, different frequency traps are placed in the mixer circuit. Great attention is given to the minimization of the LO leakage at the RF port. In the cold-FET condition, the gate-to-drain capacitance is higher than in an amplifying FET. If RF and LO frequencies are far enough in all the possible operative conditions, a notch filter (i.e., a short circuit) at the LO frequency can be placed at the FET drain. But if the LO cannot be discriminated in the frequency domain from the RF through a fixed notch filter, then other techniques must be adopted. To this aim, a balanced mixer topology is chosen, and for high frequency a feedback network between the FET’s drain and gate is adopted to compensate the intrinsic feedback of the device. In a balanced mixer, represented in Figure 6.51 in two possible configurations, two mixer cores are used together providing the LO pump and the IF in a smart way, in order to improve the LO rejection and preserve the other figures of merit. The LO is canceled since it combines in phase opposition at the output. The IF input is differential: off-chip transformers are usually employed, given the relatively low frequency. Figure 6.52 shows a complete scheme of an I/Q upconverter realized with balanced FET resistive mixer. The LO enters a Lange coupler, which is responsible for providing the 90° phase shift for the quadrature mixing. Two identical balanced mixers are fed by the LO ports and upconvert the I and Q IF signals to RF, then added by the output combiner. The residual LO leakage is probably the most critical feature of the I/Q upconverter, and perhaps one of the most influenced by the design choices. In fact, noise, linearity, and LO pump level are almost given once the technology and the FET size are determined. Observing Figure 6.52, it can be seen that the LO leakage can be determined by some factors. Leaving completely the LO cancellation duty to the balanced structure will not bring good results: a feedback network must be employed for high-frequency circuits. For broadband mixers, it is not always possible to compensate all frequencies; hence, priority must be given to suppress the LO in the frequency bands where the balancing results are more critical. The balanced mixer must be carefully paired since every phase or magnitude mismatch of the balun and combiner will lead to an increase of LO leakage.

Figure 6.51  T wo possible configurations of a balanced mixer. Left: input balun + output Wilkinson combiner. Right: input and output Lange couplers.

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Figure 6.52  Complete scheme of an I/Q upconverter based on balanced FET resistive mixers.

For high-frequency mixers, coupling and interactions between EM structures can cause an unbalance or even a direct cross-talk between the input and output LO ports. Very important is also the effect of packaging and external routing, which, for increasing frequencies, can be very important. In any case, the LO is rarely canceled enough to meet the system requirements: some countermeasure can be applied at the modem level to improve this FoM (see Chapter 4). The microwave designer must guarantee a reasonable level in order to allow the successful intervention of the modem. By analyzing the mixer design flow, it is clear that the design of an MMIC FET resisitive mixer requires an FET model that is accurate for zero-drain bias. The extraction of such a model can be a tough task, but once the model is available, the mixer design is quite simple. The scheme of Figure 6.50 can be used as a simulation bench to identify the ports’ impedance. The FET must be represented through a nonlinear model, in order to identify the optimum gate bias and LO level. An experimental approach can also be followed, by creating a measurement setup that tries to re-create this ideal simulation bench, thus determining the impedance levels without the need of a model. As a second step, “real” feedback and matching networks can be inserted one by one in simulation, retuning the overall mixer in the process to maintain the performance as much as possible. The combining structures, as baluns, splitters, or Lange couplers, can be preliminarily designed in a separate form. After that, especially for increasing frequency, they must be retuned to correct cross-talk effects between the different mixer cores, together with mismatch and coupling between physical structures. This overall optimization is often more difficult than the basic mixer design, since it requires a bit of experience for a fast identification of the critical structures and interconnections.

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6.3.5 Voltage Controlled Oscillators The microwave link is based on the communication between a TX and an RX by means of a microwave signal centered around a carrier frequency. As discussed previously, the mixer takes care, in the upconverter, to transfer the information from baseband to the carrier frequency, which is usually referred to as RF frequency. Independently from the usage of a homodyne or heterodyne up converter, this RF frequency is related to an LO frequency f0 that must be generated by the microwave radio. The generated LO frequency signal must comply with some specifications that are introduced in this subsection with the most common architectures for LO generation. The core component in the LO generation is an oscillator, in particular a voltage controlled oscillator (VCO) (Figure 6.53). The VCO is inserted in a phase locked loop (PLL) that enables a stable control of the oscillating frequency, thanks to the possibility of locking it to an external low-frequency reference. The latter is usually determined by a low-frequency quartz oscillator (tens of megahertz), with superior stability performance. The VCO usually provides to the PLL an auxiliary output at an nth fraction of f0. The most important FoM of a microwave VCO is the phase noise, which can be measured observing the spectral density of the generated LO. Phase noise is defined as the spectral power in a 1-Hz bandwidth at a certain frequency offset from the nominal carrier frequency. The maximum phase noise is not provided by ETSI, but is a system design parameter that is usually determined according to the link budget and the capability of modem and baseband section to maintain the synchronicity in presence of noise. Commercial VCOs present phase noise at 100-kHz offset ranging between –l55 dBc/Hz for a quality VCO up to 12–15 GHz, to –90 dBc/Hz for a lower-quality VCO. The resonator strongly attenuates the broadband noise, and the main noise source in a VCO is the active device; and particular care has to be given to its 1/f noise. For this reason, since bipolar devices have significantly lower 1/f noise than HEMTs, they are preferred for VCO design. The spectral purity of the VCO is also important, in order to avoid spurious frequency that act as interferers: harmonics are in general at least 10 dB lower than fundamental, and a filter can be employed to improve the rejection.

Figure 6.53  Local oscillator generation with PLL including a microwave VCO.

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The frequency stability of the VCO, related to temperature change in both transistor and resonator, is another important FoM, and it is expressed in ppm/°C or total ppm. ETSI typically indicates a value of radio frequency tolerance in the range 10–50 ppm, according to frequency band and equipment class: the VCO is designed to be well inside this tolerance in all the temperature range of operation. The frequency stability values provided by ETSI are needed to avoid the transmitter interfering with other radios. These values do not guarantee a correct communication between TX and RX: on the RX side, a PLL is used to lock the LO at the transmit frequency and phase. Moreover, at the modem level, some systems for phase noise reduction can also be implemented (see Chapter 4). A very stable resonator is necessary but not sufficient to ensure lowfrequency deviation with temperature: it must also present a high Q to avoid the drift of loading impedance shown by the transistors generating a frequency shift. Not to be forgotten is also the LO power level: in fact, to obtain good linearity and LO rejection from the upconverting mixers, it is important to pump them with sufficient LO power. If necessary, the VCO output must be amplified to achieve the needed level, but this will cost in terms of complexity and power consumption. In general, an output buffer is however inserted in the VCO to decrease its sensitivity to load conditions. A VCO can be seen as a frequency-selective feedback system, with a loop gain AF, typically composed by an amplifying stage and a resonator. The resonator includes some electronically tunable elements, a varactor in most of the cases, to enable the voltage control of the center frequency. By changing the controlling voltage, the oscillation frequency, at which the system has a unitary loop, is selected. An example of VCO is shown in Figure 6.54, where a common-base Colpitts scheme is adopted. In a first approximation, the circuit can be analyzed with a small signal equivalent circuit, using a Giacoletto circuit for the bipolar, including the input admittance Yi = Gi + jωCi. The amplification is the transconductance A = gm. The feedback can be described by a transfer function that results in F = –Z21, where Z21 is the impedance matrix parameter of the network of Figure 6.54 (right). By equating –gmZ21 = 1, the oscillation frequency can be determined. Actually, to ensure the oscillation start-up, a loop gain slightly higher than 1 is normally set: thanks to this, the noise present in the circuit is selectively amplified by the feedback system. When the amplitude of the oscillating wave increases, the device tends to saturate, lowering its gain and stabilizing the oscillation amplitude.

Figure 6.54  S  implified schematic of a Colpitis oscillator (left) and the feedback network arranged in a convenient form (right).

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A more widely adopted approach for microwave oscillators design is the negative-resistance method, proposed by [45]. A linear analysis that approximates the oscillation frequency relies on the identification of the impedance ZIN presented by an active device with feedback, which will show a negative resistance where it is not stable; see Figure 6.55. By loading this device with an impedance ZL = ZIN the oscillation condition is met. Actually, the condition | ZL | < | ZIN| is necessary to start up the oscillation, and once a certain power level is achieved, saturation will reduce the device negative resistance, stabilizing the oscillation. The two described approaches can lead to a good approximation of the oscillating frequency: in hybrid realization this could be enough if some fine post-tuning can be applied. However, for a more accurate oscillating frequency determination and to evaluate the other FoMs, a nonlinear analysis is necessary. In fact, the active device, when driven in large signal, will also change the imaginary part of the impedance it presents to the resonator, yielding a frequency shift with respect to the small signal condition analysis. A harmonic balance simulator can be effectively adopted for VCO design purposes, and further details can be found in [46]. The frequency limitation of a VCO design is usually dictated by the technology. Every technology has a frequency range where the VCO is optimized, and this is mainly determined by the quality of the passive structures. In fact, LC resonators have a well-defined frequency range where their Q is optimum. This interval changes with substrate technology and geometry of the components. In general, commercially available VCOs can be found up to around 15 GHz. For higher frequency, the availability of good HBT or complementary metal oxide semiconductor (CMOS) process is lacking, and the realization with HEMTs would not guarantee the needed phase noise. 6.3.6 Frequency Multipliers If a VCO that guarantees the needed phase noise cannot be found for the frequency band of interest, then frequency multipliers can be adopted. As the name suggests, these components provide an output signal at a frequency f0 that is nth

Figure 6.55  Simplified schematic of a negative resistance oscillator.

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multiple of the input signal frequency f0 /n. Frequency multipliers are generally integrated in the upconverter chip, in order to distribute the LO reference at lower frequency, and then amplifying it close to where it is needed. This reduces losses and the probability that the LO could be probed by some component and leaked to the output. The main FoMs of these components are output power and unwanted frequency rejection, and the design is normally based on harmonic balance simulations. More details can be found in [46]. For a better integration with other components such as mixers and amplifiers, FET-based multipliers are preferred. They are substantially saturated amplifiers where the FET input is matched at f0 /n, and the output is matched at f0; see Figure 6.56. All the output frequency components at f0 /n and multiples different from n are short-circuited. Feedback is normally adopted to improve stability and f0 /n isolation, while self-biased FET are employed to provide a stable bias minimizing the number of different bias pads. 6.3.7 Backhaul Upconverter Examples The commercially available solutions for backhaul applications are almost all based on dedicated chipsets that cover the different point-to-point radio frequency bands. Of course, the radio designers are free to select the different blocks from different providers, according to performance and cost considerations. In heterodyne transmitters, the first upconversion from digital to IF is well integrated at modem level. An example of a commercially available chip for IF upconversion is the TRF2443 from Texas Intruments, which implements fullduplex transceiver functionalities. It is conceived for the indoor unit, and its IF input and output are connected via coaxial cables to the outdoor unit. It permits a full remote control by the baseband unit and provides auxiliary outputs/input for advanced functions, as cross-polar interference cancellation (XPIC); see Chapter 4. Similar products can be found also from other suppliers. The subsequent upconversion steps usually employ compound semiconductor integrated circuits, which have less integrated functionalities but guarantee

Figure 6.56  Simplified schematic of an active FET multiplier.

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higher microwave performance. They are physically placed close to the power amplifier, typically on the same board. In the following, some commercial examples of MMICs for local oscillator generation and upconversion are described. 6.3.7.1 Upconverter MMICs

Several solutions are available on the market for MMIC upconverters. While in some cases only the mixer functions are integrated, in other cases the MMIC can include LO buffer or multiplier, gain chain, or variable gain amplifiers. Table 6.5 resumes the main characteristics of the described products. The Qorvo (former RFMD) RFUV1002 is a 9-GHz to 14-GHz GaAs HEMT upconverter, incorporating an integrated LO buffer amplifier, a balanced single-sideband (image rejection) mixer, followed by a variable gain amplifier. RFUV1002 is packaged in a 5 mm × 5 mm quad flat nonlead (QFN) package, and it is biased at 5 V. The IF I/Q inputs are single ended and DC-coupled. Two control pins allow to vary the conversion gain up to 23 dB, with a dynamic of 38 dB. The Qorvo RFUV1003 contains the same features of the RFUV1002, but it covers a higher-frequency band. The Qorvo TGC4402 is an upconverting FET-resistive mixer, working in K-band. Its conversion loss is limited at 9 dB in the working band, and a single gate control at –0.9 V must be used. In the same band, the Qorvo TGC4405 (microscope picture in Figure 6.57) also includes an LO frequency doubler (bottom of the picture) and a buffer amplifier (top left of the picture), reaching a conversion gain of 13 dB. The mixer balun can be seen on the right of the picture, feeding the two mixer cores. The IF is provided with a long transmission line that crosses the MMIC from left to right and is then split into the two mixer cores through a lumped element divider. In the Ka-band, the Qorvo TGC4546-SM contains a LO buffer and quadrupler, and amplifying stages, reaching 11 dB of conversion gain. The I/Q IF pads are DC-coupled and differential. Table 6.5 Main Characteristics of the Reported Commercial Upconverters Part Name RFUV1002 RFUV1003 TGC4402 TGC4405 TGC4546-SM HMC-MDB377

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RF Range (GHz)

LO Range (GHz)

IF Range (GHz)

Conversion Gain (dB)

9–14

5–18

0–4

–15÷+23

12–16 17–27 17–27 36–45 70–90

8–20 14–28 8–13 8.1–10.4 70–90

0–4 0.5–3 0.5–3 0–3.5 0–18

–10÷+23 –9 +13 +11 –12

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Figure 6.57  Microscope picture of TCG4405. (Courtesy of Qorvo Inc.)

In E-band, the Hittite HMC-MDB277 is an example of HBT diode mixer, with a conversion loss of 12 dB and a wide IF range between 0 and 18 GHz. Being a diode mixer, the LO and RF input work on the same frequency band. 6.3.7.2 HBT VCOs

Many integrated VCOs are available commercially. Table 6.6 resumes the main characteristics of the here reported examples. The Hittite HMC358MS8G is a GaAs InGaP HBT MMIC VCO. It integrates resonators, negative resistance devices, varactor diodes, and buffer amplifiers. The phase noise is of –100 dBc/Hz at 100-kHz offset, while an output power output of 11 dBm is provided thanks to the output buffer amplifier. In this case, the VCO is packaged in a low-cost, surface-mount, 8-lead MSOP package with an exposed base. The power consumption is 300 mW, while the output frequency is tuned by varying the controlled voltage in the range 0–10 V. Table 6.6 Main Characteristics of the Reported Commercial VCOs

Part Name HMC358MS8G HMC507LP5 RFVC1831 TGV2539-SM HMC533LP4

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Range (GHz) 5.8–6.8 6.65–7.65 7.3–8.2 10.7–11.5 23.8–24.8

270

Output Power (dBm)

Power Consumption (mW)

Phase Noise @100 kHz (dBc/Hz)

11

300

–110

13.5 10 10 12

1150 1150/850 930 1100

–115 –115 –111 –95

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Hittite portfolio includes also the HMC507LP5, a GaAs InGaP HBT MMIC VCO for the 7-GHz band. In this case, the device also features a halffrequency output, used in the PLL. The output power is 13.5 dBm, and the phase noise is –115 dBc/Hz at 100-kHz offset. In this case, the VCO is packaged in a leadless QFN 5 × 5 mm2 surface mount package. Power consumption is 1.15 W, while the output frequency can be tuned in the range 6.65–7.65 GHz with a 2–13-V control voltage. For the 8-GHz band, an example is the RFMD RFVC1831, 5V InGaP MMIC VCO with an integrated frequency divider providing f /2 and f /4 outputs. Output power is 10 dBm, flat across the tuning voltage range of 1.5V to 14.5V (7.3–8.2 GHz). Phase noise is typically –115 dBc/Hz at 100-kHz offset. The device operates from a supply current of 230 mA, which can be reduced to 170 mA by disabling the divider functions. The RFVC1831 is available in a low cost 5 × 5 mm2 surface mount plastic overmolded QFN outline. In X-band, the Triquint TGV2539-SM is fabricated on GaAs InGaP HBT, and typically provides 10 dBm output power with a –111-dBc/Hz phase noise at 100-kHz offset. A divide-by-2 output is provided for the PLL function. Its power consumption is 0.93 W, while a 1.5–13-V control voltage allows the frequency tuning in the range 10.7–11.5 GHz. Hittite also provides a K-band VCO, the HMC533LP4, based on GaAs InGaP HBT MMIC. The VCO features a divide-by-16 output, and shows a –95-dBc/Hz phase noise at 100 kHz. Output power is 12 dBm for a 1.1-W power consumption. Prescaler function can be disabled to conserve current. The voltage controlled oscillator is packaged in a leadless QFN 4 × 4 mm2 surface mount package. This product can be a viable alternative to a VCO + multiplier chain for the K-band, in particular if area and chip count are main requirements. 6.3.7.3 Frequency Multipliers

Table 6.7 summarizes the main characteristics of the examples reported here. The Hittite HMC577LC4B is a ×2 active broadband frequency multiplier based on GaAs PHEMT technology. For an input power of 5 dBm, the multiplier provides 20-dBm typical output power from 27 to 31 GHz. The f0/2 and 3f0/2 isolations are larger than 55 dBc at 29 GHz. The additive phase noise is of –128 dBc/Hz at 100-kHz offset. The multiplier is composed by self-polarizing transistors, since it is biased with a single 5-V supply, with around 1-W power consumption. The Hittite HMC579 is similar to the previously described ×2 multiplier, but works at higher frequency. It is based on GaAs PHEMT technology, and when driven by a 3-dBm input, it provides 13 dBm of output power from 32 to 46 GHz. The f0/ 2 isolation is higher than 25 dBc at 38 GHz. The additive phase noise is –127 dBc/Hz at 100-kHz offset. The HMC579 works at 5-V bias voltage, with a consumption of 70 mA.

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Table 6.7 Main Characteristics of the Reported Commercial Frequency Multipliers

Part Name

Factor

Output Range (GHz)

Output Power (dBm)

Added Phase Noise @100 kHz (dBc/Hz)

HMC577LC4B

x2

27–31

20

–128

HMC579 HMC1110 TGC4704-FC

x2 x6 x2

32–46 71–86 76–77

13 13 14

–127 n.d. n.d.

Figure 6.58  Picture of the TGC4704-FC. Size: 3.38×1.37 mm2. (Courtesy of Qorvo Inc.)

For the E-band, the HMC1110 from Hittite can be used as a ×6 broadband frequency multiplier. It is based on GaAs PHEMT technology, and provides 13 dBm of output power from 71 to 86 GHz if driven with 4-dBm input. The 5 f0/6 and 7 f0/6 harmonic isolations with respect to the output signal level are 25 dBc and 40 dBc, respectively. Always for the E-band, the TriQuint TGC4704-FC is a flip-chip frequency doubler; see layout in Figure 6.58. It consists of a frequency doubler followed by a medium-power buffer amplifier. It has been designed on a 0.13-μm PHEMT process and uses front-side Cu/Sn pillar technology for simplified assembly and low interconnect inductance. The TGC4704-FC typically provides 14 dBm of saturated output power with 5-dB conversion gain. The design is based on coplanar semi-lumped matching networks and fillers.

6.4 Digital-to-Analog Conversion 6.4.1 Basis The digital-to-analog converters (DACs) are the interface between the digital, numerical world of the modem, and the analog world of the microwave front

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end. Being the digital input discrete in both dynamic (y-axis) and time (x-taxis), DACs can be thought as composed by two blocks; see Figure 6.59. The first block translates the numerical code (i.e., a binary code) into a voltage or current; hence, it operates on the y-axis (see Figure 6.60). The output of this block is stepped, since the input code is not continuous, and it does not give any information about how two points are connected. However, if Nyquist criteria are met, the sampled signal contains all the information needed for the signal reconstruction, and the quantization error can be seen as noise. The second block smooths the stepped output in the time domain (x-axis), that is, it works as a reconstruction filter. It has to be noticed that the trend between discrete points can be determined only knowing the story of the signal, since it is obtained through a filter. Figure 6.61 compares the output of a DAC without and with the reconstruction filter: in the former case, the output voltage is maintained constant until the following sample is read at the input, while in the latter case the samples are smoothed in a continuous response. In reality, the two functions cannot be clearly distinguished in a DAC, since a stepped response is not physical.

Figure 6.59  Block scheme of a DAC.

Figure 6.60  Working principle of a 3-bit DAC: static transfer function.

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Figure 6.61  DAC output voltage versus time without (left) and with (right) reconstruction filter.

6.4.2 Characteristics and Figures of Merit The resolution of a DAC is related to the number N of bits at which it is expressed the input code, and defines how many different states the DAC can reproduce. It also determines the minimum voltage step that the DAC can resolve, usually defined as VLSB = Vref , where Vref is a reference voltage. However, some

2N

DAC architectures automatically reduce the actual resolution when used at high speed. Then the defined VLSB figure can be considered as a best-case resolution. The speed of the DAC is defined as maximum sample rate, which is the maximum clock frequency at which the output is still the correct one. Characteristics of great importance are the total harmonic distortion and noise that characterize the unwanted spectral components at the output of the DAC and the dynamic range, defined as the ratio between the maximum and minimum signal (in magnitude) that the DAC can generate. Typical static (i.e., characterized with very slow-varying input) DAC figures of merit are: •• Differential nonlinearity, the error calculated as DV - VLSB where DV

VLSB is the difference between the measured output voltage at two successive codes; •• Integral nonlinearity, the error between an ideal straight line connecting the output voltage points and the measured transfer function; •• Offset. Also, dynamic figures of merit are usually defined as: •• Spurious-free dynamic range, the difference between the output power

in the channel and the highest spurious frequency component; •• Signal-to-noise and distortion ratio, the difference between the power of the useful signal and the total of noise and spurious frequencies;

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•• Harmonic distortion, the power of a single harmonic (or the total of the

harmonics) generated at the output; •• ACPR or CIMR3, defined similarly as in PAs and mixers, particularly useful for system-level simulations to dimension the upconverting chain. 6.4.3 DACs for Microwave Radio In a microwave radio, the DAC can be placed at baseband or at IF, according to the adopted upconversion scheme. However, working with quadrature modulations, a pair of DACs is usually adopted (except in a pure heterodyne scheme), and commercial products (in the form of ICs) usually already include two DACs. This integration helps the reduction of impairments between the I/Q paths. A detailed analysis of DAC schemes is out of the scope of this book: good references can be found in [47, 48]. However, it is important to notice that the main trade-off in DAC design is between accuracy and speed. Very fast converters, such as binary-weighted DACs, have low resolution (generally below 8 bits). Very accurate converters, such as the R-2R ladder, successive approximation, or sigma-delta, are generally slow. But digital radios require both high speed operation (sampling frequency between some megahertz for low-frequency applications, and hundreds of megahertz for E-band radio) and high accuracy (to avoid spectral regrowth, due to very stringent masks). A solution is represented by the thermometer-coded DAC, which can achieve high resolution with high speed. However, since it is based on a voltage or current contributor for each possible state,9 it is very expensive. Because the cost, at the end of the day, is a major constraint, hybrid DAC solutions are preferred. For example, the segmented DAC is a clever architecture that uses a thermometer-coded DAC for the most significant bits and a binaryweighted DAC for the least significant bits. The most common DACs for backhaul radio applications generally present current driven differential outputs. The typical resolution is between 10 and 14 bits, and according to the radio working band, the sampling rate can range between 20Msps and 1 Gsps. As a first example, the Texas Instruments DAC5662 is a monolithic, dualchannel, 12-bit, high-speed, digital-to-analog converter (DAC) with on-chip voltage reference. Operating with update rates of up to 275 Msps, it is suitable in either I/Q baseband or direct IF communication applications. Each DAC has a high-impedance differential current output suitable for single-ended or differential analog-output configurations. The DAC5662 has two 12-bit parallel input ports with separate clocks and data latches. It is characterized by a 9.

While binary-weighted DACs have a contributor for every bit.

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spurious-free dynamic range (SFDR) of 85 dBc at 5 MHz, and by a CIMR3 of 78 dBc at 15.1 MHz. The optimal performance is obtained with output voltage in the range –0.5 V ÷ +0.5 V. Another example is the AD9785 from Analog Devices, which has a maximum sampling rate of 800 Msps and a resolution of 12 bits. Interpolation filtering can be enabled at the input, to improve the SFDR from 80 to 85 dBc, at the cost of power consumption passing from 0.4 to 1W. This DAC also contains a numerically controlled oscillator (NCO) (see Chapter 4) that permits I/Q modulation.

6.5 Linearization Techniques 6.5.1 Basis The TX chain introduces distortion, and the main contributor is usually the PA. Due to the very stringent emission masks and the EVM requirements, the insertion of a linearizer is necessary to guarantee the needed linearity. Practically, in microwave radios, predistorters are usually employed: the name derives from the fact that they are placed before the element that should be linearized; see Figure 6.62. In general, if the distorting block (or the cascade of blocks) has a descriptive function: y(t ) = f ( x (t ), t ) (6.31)



the predistorter will need to have a function: xDPD (t ) = g ( x (t ), t ) (6.32)

that gives:

yDPD = f ( xDPD (t ), t ) = f ( g ( x(t )), t ) = Gx (t ) (6.33)

where G is a constant system gain. In other words, g (•) is the inverse function of f (•), apart from the gain G. Nonlinear systems with memory are widely studied in mathematics and control theory; however, they remain very complex to be treated. For this reason, the nonlinearity introduced by the TX chain is usually considered static,10 and the TX design is carried out in order to obtain an actual behavior as close as possible to the static approximation. For example, the bias circuitry and the thermal 10. Meaning that nonlinear contribution can be clearly separated from the filtering effects, compensated by the equalizers.

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Figure 6.62  L eft: nonlinear function representing the TX. Right: introduction of a predistortion function for linearization.

management of the PA play a crucial role in reducing the memory effects. If the nonlinearity to be compensated is static, then a static predistorter will be sufficient to linearize the system. It has to be noticed that the considered static nonlinear functions present changes in their gain and phase response when the amplitude of the input changes, while the response do not change with the phase. In other words, two inputs with same amplitude but different phase will produce the same outputs, apart from a phase rotation. This means that the core of the predistortion function will be affected by the amplitude of the signal, while the input phase will only add to the contribution due to AM/PM. The function g (•) can be implemented by means of electronics at different levels, microwave, IF, or baseband, and theoretically it can achieve the same performance in the three cases. However, according to the radio architecture, the frequency band of operation and the modulation order, the three solutions present pros and cons that can direct the choice to one of them. In the following, a brief description of predistortion techniques is given. In Chapter 4, a more focused discussion on actual predistorter for microwave radios is provided. 6.5.2 Analog Microwave Predistortion Analog predistortion implies to place the predistortion function, in the form of a microwave circuit with a gain expansion response, after the last upconversion (see Figure 6.63). Usually, the expansion function is obtained by subtracting a compressive function (more typical of actual devices as diodes or transistors) from a linear response. The strong point of analog RF predistortion is the bandwidth, since it can be designed to be the same of the PA. However, the design of the predistorter will face the same issues that arise in the design of the microwave circuits in general, in particular in terms of repeatability and predictability of performance. For this reason, some postrealization tuning is almost always necessary to obtain a good distortion compensation. For example, by changing the bias point of the diode, some correction can be

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Figure 6.63  Block scheme of TX with analog predistortion at microwave frequency.

performed. Another important limitation is the difficulty to make this kind of predistorter adaptive, since it would require a very complex analog circuitry that elaborates a feedback from the PA output. A solution could be to elaborate the adaptation algorithm in digital, but at this point a fully digital predistorter would probably be preferred. Finally, the compensation of memory effects is practically unfeasible with analog microwave predistortion. 6.5.3 Analog IF Predistortion Another form of analog predistortion that has been suggested throughout the years is IF predistortion. This can be performed in a double-sideband heterodyne system (see Figure 6.64), where the aggregate information of I and Q is available.11 This solution still has the advantage of showing a very good bandwidth, since it is realized in analog form, but, working at a lower frequency than the microwave predistortion case, it simplifies the design and tuning of the predistorter. However, some limitations about the possibility to realize adaptive predistortion and to compensate memory effect still exist. Moreover, the double-sideband upconverter is rarely used in microwave radios, where SSB upconverters are preferred. 6.5.4 Digital Predistortion In digital predistortion (DPD), the predistorter is integrated in the modem, as depicted in Figure 6.65. This solution has taken a big advantage from the advances in DSP circuits and nowadays is probably the most widely adopted linearization technique. Its main advantages are the flexibility, the possibility to make it adaptive and to change the algorithm reprogramming an FPGA, and the capability to compensate for memory effects. However, the DPD works on the baseband signal that will become the complex envelope of the microwave signal. In order to be effective, the baseband signal must contain also the out-of-band frequency components generated by the predistortion function, making even tougher the sampling rate requirements of the modem and the DACs.

11. Otherwise, in an SSB, some added circuitry that calculates amplitude and phase of the signal would be required, and the function would have effects on both channels.

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Figure 6.64  Block scheme of TX with analog predistortion at IF.

Figure 6.65  Block scheme of TX with baseband digital predistortion (DPD).

The design of a DPD function is in general based on a behavioral model of the TX (in this case, we talk of direct method of DPD extraction), or on system level characterization of the TX (indirect method). Then an appropriate DPD function is selected, and its coefficients are determined. The latter can be updated during the operation with an adaptation algorithm that compares the samples derived through a feedback channel from the PA output with the expected ones. Typical functions are polynomials, memory polynomials, or neural networks. In particular, the memory polynomial [49] (see Figure 6.66) is based on a parallel Hammerstein scheme, where more branches are added at the output. Every branch processes the input sample through a static nonlinearity (an odd monomial form) followed by a filter. The polynomial contains odd terms only, up to the 2P + 1 order, that are sufficient for a narrowband system [50]. The filter is implemented through a finite impulse response (FIR) filter, with m memory taps. The most important feature of this model is that it is linear in its parameters, so the parameters extraction can be

Figure 6.66  Block scheme of memory polynomial DPD.

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made by solving a matrix pseudo-inverse. A static polynomial can be obtained by simply deleting the FIRs from the scheme, which is equivalent to m = 0. While very powerful DPDs are usually employed in base stations for linearizing the PAs, in backhaul radios it is not possible to perform too complex DSP functions. In fact, the power level and the cost of microwave radios PAs are at least one order of magnitude lower than base station PAs. For this reason, simpler predistortion functions are preferred, as discussed in detail in Chapter 4.

References [1]

Cripps, S., RF Power Amplifiers for Wireless Communications, Norwood, MA; Artech House, 2006.

[2]

Colantonio, P., F. Giannini, and E. Limiti, High Efficiency RF and Microwave Solid State Power Amplifiers, New York: John Wiley & Sons, 2009.

[3]

Kenington, P., High Linearity RF Amplifier Design, Norwood MA; Areech House, 2000.

[4]

Quaglia, R. et al., “Linear GaN MMIC Combined Power Amplifiers for 7-GHz Microwave Backhaul,” IEEE Trans. Microw. Theory Techn., Vol. 62, No. 11, November 2014, pp. 2700–2710.

[5]

“ETSI EN 302 217-2-2. Fixed Radio Systems; Characteristics and Requirements for Pointto-Point Equipment and Antennas; Part 2-2: Digital Systems Operating in Frequency Bands Where Frequency Co-ordination Is Applied,” 2012.

[6]

“ETSI EN 302 217-2-1. Fixed Radio Systems; Characteristics and Requirements for Pointto-Point Equipment and Antennas; Part 2-2: System Dependent Requirements for Digital Systems Operating in Frequency Bands Where Frequency Co-ordination is Applied,” 2012.

[7]

“ETSI EN 302 217-3. Fixed Radio Systems; Characteristics and Requirements for Pointto-Point Equipment and Antennas; Part 3: Equipment Operating in Frequency Bands Where Both Frequency Co-ordinated or Uncoordinated Deployment Might Be Applied; Harmonized EN Covering the Essential Requirements of Article 3.2 Of The R&TTE Directive,” 2007.

[8]

Cripps, S., Advanced Techniques in RF Power Amplifier Design, Norwood, MA; Artech House, 2002.

[9]

Doherty, W., “A New High Efficiency Power Amplifier for Modulated Waves,” Proc. IRE, Vol. 24, No. 9, September 1936, pp. 1163–1182.

[10] Grebennikov, A. and S. Bulja, “High-Efficiency Doherty Power Amplifiers: Historical Aspect and Modern Trends,” Proc. IEEE, Vol. 100, No. 12, December 2012, pp. 3190–32l9. [11] Camarchia, V. et al., “The Doherty Power Amplifier: Review of Recent Solutions and Trends,” IEEE Trans. Microw. Theory Techn., Vol. PP, No. 99, 2015 pp. 1–13. [12] Campbell, C. et al., “A K-Band 5W Doherty Amplifier MMIC Utilizing 0.15µm GaN on SiC HEMT Technology,” Compound Semiconductor Integrated Circuit Symposium (CSICS), 2012 IEEE, October 2012, pp. 1–4.

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[13] Camarchia, V. et al., “7 GHz MMIC GaN Doherty Power Amplifier with 47% Efficiency at 7 dB Output Back-Off,” IEEE Microw. Wireless Compon. Lett., Vol. 23, No. 1, January 2013, pp. 34–36. [14] Gustafsson, D. et al., “A Wideband and Compact GaN MMIC Doherty Amplifier for Microwave Link Applications,” IEEE Trans. Microw. Theory Techn., Vol. 61, No. 2, February 2013, pp. 922–930. [15] Piazzon, L. et al., “15% Bandwidth 7 GHz GaN-MMIC Doherty Amplifier with Enhanced Auxiliary Chain,” Microwave and Optical Technology Letters, Vol. 56, No. 2, February 2014, pp. 502–504. [16] Piazzon, L. et al., “Effect of Load Modulation on Phase Distortion in Doherty Power Amplifiers,” IEEE Microw. Wireless Compon. Lett., Vol. 24, No. 7, July 2014, pp. 505–507. [17] Quaglia, R. et al., “Experimental Investigation of Bias Current and Load Modulation Effects in Phase Distortion of GAN HEMTS,” Electronics Letters, Vol. 50, No. 10, May 2014, pp. 773–775. [18] Camarchia, V. et al., “High-Efficiency 7 GHz Doherty GaN MMIC Power Amplifiers for Microwave Backhaul Radio Links,” IEEE Trans. Electron Devices, Vol. 60, No. 10, October 2013, pp. 3592–3595. [19] Golio, M., Microwave MESFETs and HEMTs, Norwood, MA; Artech House, 1991. [20] Golio, M., RF and Microwave Semiconductor Device Handbook, Boca Raton, FL; CRC Press, 2002. [21] Sechi, F. and M. Bujatti, Solid-Slate State Microwave High-Power Amplifiers, Norwood, MA; Artech House, 2009. [22] Angelov, I., H. Zirath, and N. Rosman, “A New Empirical Nonlinear Model for HEMT and MESFET Devices,” IEEE Trans. Microw. Theory Techn., Vol. 40, No. 12, December 1992, pp. 2258–2266. [23] Filicori, F. et al., “Empirical Modeling of Low-Frequency Dispersive Effects Due to Traps and Thermal Phenomena in III-V FET’s,” IEEE Trans. Microw. Theory Techn., Vol. 43, No. 12, December 1995, pp. 2972–2981. [24] Pengelly, R. S. et al., “A Review of GaN on SiC High Electron-Mobility Power Transistors and MMICs,” IEEE Trans. Microw. Theory Techn., Vol. 60, No. 6, June 2012, pp. 1764–1783. [25] Quaglia, R. et al., “K-Band Gaas MMIC Doherty Power Amplifier for Microwave Radio with Optimized Driver,” IEEE Trans. Microw. theory Techn., Vol. 62, No. 11, November 2014, pp. 2518–2525. [26] Gustafsson, D. et al., “A GAN MMIC Modified Doherty PA with Large Bandwidth and Reconfigurable Efficiency,” IEEE Trans. Microw. Theory Techn., Vol. 62, No. 12, December 2014, pp. 3006–3016. [27] Binari, S. et al., “Trapping Effects and Microwave Power Performance in AlGaN/GaN HEMTs,” IEEE Trans. Electron Devices, Vol. 48, No. 3, March 2001, pp. 465–471. [28] Camarchia, V. et al., “Self-Consistent Electrothermal Modeling of Class A, AB, and B Power GaN HEMTs Under Modulated RF Excitation,” IEEE Trans. Microw. Theory Techn., Vol. 55, No. 9, September 2007 pp. 1824–1831.

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[29] Jardel, O. et al., “An Electrothermal Model for AlGaN/GaN Power HEMTs Including Trapping Effects to Improve Large-Signal Simulation Results on High VSWR,” IEEE Trans. Microw. Theory Techn., Vol. 55, No. 12, December 2007, pp. 2660–2669. [30] Medjdoub, F. et al., “Towards Highly Scaled AIN/GaN-on-Silicon Devices for Millimeter Wave Applications,” 2012 7th European, Microwave Integrated Circuits Conference (EuMIC), October 2012, pp. 321–324. [31] Avenier, G. et al., “0.13 µm SIGE BICMOS Technology Fully Dedicated to MM-Wave Applications,” IEEE J. Solid-State Circuits, Vol. 44, No. 9, September 2009, pp. 2312–2321. [32] Heydari, B. et al., “Millimeter-Wave Devices and Circuit Blocks up to 104 GHz in 90 nm CMOS,” IEEE J. Solid-State Circuits, Vol. 42, No. 12, December 2007, pp. 2893–2903. [33] Tai, W., L. Carley, and D. Ricketts, “A 0.7W Fully Integrated 42GHz Power Amplifier with 10% PAE in 0.13 µm SiGe BiCMOS,” 2013 IEEE International, Solid-State Circuits Conference Digest of Technical Papers (ISSCC), February 2013, pp. 142–143. [34] Pfeiffer, U. and D. Goren, “A 20 dBm Fully-Integrated 60 GHz-SIGE Power Amplifier with Automatic Level Control,” IEEE J. Solid-State Circuits, Vol. 42, No. 7, July 2007, pp. 1455–1463. [35] Pourchon, F. et al., “From Measurement to Intrinsic Device Characteristics: Test Structures and Parasitic Determination,” IEEE Bipolar/BiCMOS Circuits and Technology Meeting, 2008. BCTM 2008, October 2008, pp. 232–239. [36] Harame, D. et al., “Design Automation Methodology and RF/Analog Modeling For RF CMOS and SIGE BICMOS Technologies,” IBM Journal of Research and Development, Vol. 47, No. 2.3, March 2003, pp. 139–175. [37] Pirola, M., V. Teppati, and V. Camarchia, “Microwave Measurements Part I: Linear Measurements,” IEEE lustrum. Meas. Mag., Vol. 10, No. 2, April 2007, pp. 14–19. [38] Camarchia, V. et al., “Microwave Measurements-Part II Non-linear Measurements,” IEEE Instrum. Meas. Mag., Vol. 10, No. 3, June 2007, pp. 34–39. [39] Teppati, V. et al., “Microwave measurements-Part III: Advanced Non-Linear Measurements,” IEEE Inslrum. Meas. Mag., Vol. 11, No. 6, December 2008, pp. 17–22. [40] Teppati, V., A. Ferrero, and M. Sayed, Modern RF and Microwave Measurement Techniques, Cambridge U.K., Cambridge University Press, 2015. [41] Cripps, S., “A Theory for the Prediction of GaAs FET Load-Pull Power Contours,” 1983 IEEE MTT-S International, Microwave Symposium Digest, May 1983, pp. 221–223. [42] Dawson, D. E., “Closed-Form Solutions for the Design of Optimum Matching Networks,” IEEE Trans, Microwave Theory and Techniques, Vol. 57, No. 1, January 2009, pp. 121–129. [43] Yishay, R. et al., “A 20dBm E-Band Power Amplifier in SiGe BiCMOS Technology,” 2012 42nd European, Microwave Conference (EuMC), October 2012, pp. 1079–1082. [44] Razavi, B., RF Microelectronics, Upper Saddle River, NJ; Prentice Hall, 2012. [45] Kurokawa, K., “Some Basic Characteristics of Broadband Negative Resistance Oscillator Circuits,” Bell Sys. Tech. J., Vol. 48, No. 1, 1969, p. 1937.

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[46] Maas, S. A., Nonlinear Microwave and RF Circuits, 2nd ed., Norwood, MA; Artech House, 2003. [47] Razavi, B., Principles of Data Conversion System Design, IEEE, 1994. [48] Kester, W., Data Conversion Handbook, New York, Elsevier, 2004. [49] Quaglia, R. et al., “Real-Time FPGA-Based Baseband Predistortion of W-CDMA 3GPP High-Efficiency Power Amplifiers: Comparing GaN HEMT and Si LDMOS Predistorted PA Performances,” European, Microwave Conference, 2009. EuMC 2009, October 2009, pp. 342–345. [50] Benedetto, S., E. Biglieri, and R. Daffara, “Modeling and Performance Evaluation of Nonlinear Satellite Links-A Volterra Series Approach,” IEEE Transactions on Aerospace and Electronic Systems, Vol. AES-15, No. 4, July 1979, pp. 494–507.

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7 Antenna Matteo Oldoni, James Watts, Roberto Quaglia, and Vittorio Camarchia 7.1 Basic Concepts The antenna is the transducing element which converts electromagnetic energy between its guided form (in an electronic circuit) and free radiation (in the environment). As a transmission element, the upconverted modulated signal reaches the antenna port, which then allows power to propagate in the surrounding medium (e.g., air). As a receiver component, instead, the antenna works as a sensor by converting the impinging electromagnetic waves into electric signals for the receiver’s front end. For simplicity, we will mainly refer to transmission, except when some parameters are specifically defined at the receiver end. How does an antenna work? An alternating current flowing in a conductor generates a magnetic field H that excites an electric field E. The E-field superposes a voltage on the conductor counteracting the flowing current [1, 2]. For short-length conductors, this field interaction practically leads to heat dissipation. However, if the conductor length increases and becomes comparable to the wavelength associated with the oscillation frequency, a significant part of the energy will radiate instead of being dissipated. The radiated fields will propagate in the surrounding medium with propagation characteristics dictated by the oscillation frequency, the conductor geometry and medium electromagnetic (EM) parameters (see Figure 7.1). An antenna is a particular radiating structure for which the energy portion that is radiated is dominant and whose propagated fields are controlled to achieve a certain pattern (e.g., to concentrate the radiated 285

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Figure 7.1  Representation of basic radiation from a dipole (left) or from an aperture (right).

energy towards a certain direction). Another way of envisioning an antenna is as a waveguide matching structure where the fields pass from guided propagation to free-space propagation (see Figure 7.1). Fields adjust themselves gradually, transitioning from waveguide modes to unguided propagation, and at a certain distance from the discontinuity the fields will be governed by the medium characteristics. These types of structures are called aperture antennas, due to the fact that the propagating wave characteristics can be studied by knowing the field distribution at the physical aperture of the waveguide. According to the antenna type, it can be more convenient to assimilate it as a current-carrying conductor (e.g., in wire antennas) or as a waveguide discontinuity (e.g., for horn antennas). Most antenna equations are valid in the far field, meaning that the observation point is located at a sufficient distance from the antenna that allows approximating the radiated fields as a plane wave. A planar wave supports a transverse electromagnetic (TEM) mode, meaning that both the E and H fields are perpendicular to the direction of propagation. In reality, in a uniform medium, and in the far field, the wavefront is spherical. The E and H fields are perpendicular to the propagation direction, meaning that they are tangent to the sphere surface centered on the antenna (more precisely onto its center of phase) (see Figure 7.2). For this reason, the power density is expressed for unit area (surface power density) and is the absolute value of the Poynting vector S [3]. S is the external product of E and H*, where * denotes the complex conjugate, S = E × H*, and it is directed along the radial versor, which coincides with the propagation direction. In a lossless medium, for energy conservation, the power density will decrease with the square of the distance from the antenna (i.e., the center of the sphere) since the sphere surface increases with the square of its radius. This is the cause for the squared distance in the free-space attenuation equation. The planar approximation, even if wrong from an energy conservation point of view, allows simplifying the analysis with negligible accuracy loss. The wave polarization refers to the direction of the time-varying electric field vector: if it remains constant, the wave is linearly polarized, while if it rotates

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Figure 7.2  S  pherical wavefront propagation, with Cartesian and polar coordinate systems (left). Planar wave approximation (right).

along the propagation axes, the polarization is generically elliptical. The linear polarization can be: vertical, if it remains perpendicular to the earth surface; horizontal, if it remains parallel; or cross-polarized, if it maintains a different angle. Most antenna characteristics descend directly from the propagating wave that it generates. These and other antenna properties are described next. 7.1.1 Characteristics As previously depicted, the far-field wavefront is spherical. However, the wavefront is defined by the phase behavior only, meaning that the power density is not constrained to be uniformly distributed on the sphere surface. The power density distribution is dictated by the antenna geometry, and it is called the antenna, or radiation, pattern [2, 4]. An antenna whose power density is uniformly distributed over a whole spherical surface is called an isotropic radiator. The isotropic radiator is neither practically interesting nor even physically realizable, as it would violate the “hairy ball theorem” [5]. However, it is often used as a reference to define one of the most important antenna figures of merit (FoM), the directivity. The isotropic radiator, again for energy conservation, must exhibit a l/r 2 power density fall-off as any other antenna; this implies that its power density can be expressed as: Si (r ,θ , φ ) =



Pi (7.1) 4πr 2

with Pi being a power quantity expressing the overall amount of power radiated by this ideal source in a lossless medium. A generic antenna in a lossless medium radiates an overall power P given by the surface integral of the radiated power density over any closed surface enclosing the antenna. When such surface is

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spherical and centered on the antenna, this computation is also consistent with the power density defined for the isotropic radiator. The directivity of a generic antenna is defined as the ratio of its radiated power density S(r, q, f) to the radiated power density of the isotropic radiator when both radiate the same overall power P. Furthermore, since both power densities undergo a decay with r 2, their ratio can be shown to be independent on the distance r. This yields the standard definition of directivity depending only on the pair of angles (q,f) defining a direction: D(θ , φ ) =



S(r ,θ , φ ) (7.2) Si (r ,θ , φ )

for whatever choice of r in the far field. The angle f is called azimuth, while q represents the co-elevation, according to Figure 7.2. The co-elevation is the angle measured starting from the z-axis to the plane where f is defined; sometimes the elevation is used, that is the angle starting from the plane where f is defined to the z-axis. The directivity pattern is usually given in dBi: æ Sö DdBi = 10log 10 ç ÷ (7.3) è Si ø



meaning that the directivity of an isotropic antenna is 0 dBi in each direction. The antenna directivity D is the maximum of D(q, f). It is now clear that the antenna directive gain is to be intended not as an amplification of a signal, as in amplifying circuits, but rather as the capability of the antenna to concentrate the radiated power in a certain direction. The directivity pattern is a three-dimensional surface: for representing it on papers or datasheets, and in general for a better readability, radiation pattern is usually given as two-dimensional, slicing the 3-D representation along a certain plane that passes through the antenna. If a Cartesian representation is used, the x-axis is in degrees and reports either f or q. Otherwise, a polar plot can be used. The zero of the x-axis usually corresponds to the direction of maximum radiation, and the antenna pattern is usually normalized to this maximum directivity. The antenna gain (in dBi) is derived as the difference between the directivity (in dBi) and the antenna losses (in decibels), and it represents the figure usually adopted for the link dimensioning. Gain is defined according to the power accepted by the antenna at the input port, while the directivity is defined relative to the total power radiated by the antenna. Figure 7.3 shows a typical antenna pattern for the horizontal and vertical planes, respectively. In Figure 7.3 other antenna characteristics are also evident. The radiated power cannot be concentrated totally in a single direction, but it will form lobes, usually related to a sinc function. This derives from the fact that the antenna

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Figure 7.3  U  niform aperture antenna radiation pattern for the two main planes (f = 0, f = 90°). Aperture with x-dimension of 6l, y-dimension of 3l.

pattern, in the case of an aperture antenna, is the spatial Fourier transformation of the field distribution at the aperture. If the field is uniformly distributed, the gain is maximum. The main lobe is the lobe around the direction of maximum propagation. The 3-dB aperture, or beamwidth, is the angle at which the antenna pattern drops 3 dB below the maximum. The other lobes are called secondary lobes or sidelobes. The sidelobe rejection is the ratio between the maximum of the main lobe and the maximum of the sidelobes. A particular sidelobe is the one opposite to the maximum gain direction, which describes the backside radiation. A uniform field distribution leads to high sidelobes. Then there is usually a trade-off between gain and side-lobe suppression in aperture antennas. The antenna field can be linearly or circularly polarized. A linearly polarized antenna will generate a vertical, horizontal, or cross-polarized wave according to its mounting position. A circularly polarized antenna can be clockwise or counterclockwise polarized. A dual polarized antenna can transmit two separate signals on two orthogonal polarizations (i.e., horizontal and vertical or circular clock- and counterclockwise). The overall radiated power is related to the electrical power at the antenna input by the efficiency: η=



P (7.4) Pin

that is influenced by the losses of the antenna. Losses can be minimized by using quality materials for the conducting surfaces and by accurate antenna fabrication. The antenna will be connected to the electrical generator, which has an internal impedance. The maximum power transfer between the generator and the antenna is guaranteed in case of conjugate matching. The antenna mismatch,

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referred to a reference impedance that is usually 50 Ω for a coaxial feed, is an important antenna FoM. The antenna bandwidth is the frequency range over which certain antenna FoMs are above a defined threshold. For example, the bandwidth can be the frequency range over which the sidelobe suppression is higher than 16 dB and gain is higher than 30 dB. Maximum power handling and noise figure are other FoMs that link designers must consider when selecting an antenna. 7.1.2 Requirements for Microwave Backhaul In a point-to-point link antenna, the most important FoM is the gain, which has a direct influence on the fade margin for a given hop. There are however drawbacks in selecting high gain antennas. The first is given by the difficulty to correctly align the antennas: higher gain means a very narrow main beam, and thus even a small angle deviation from the peak can lead to significant gain and fade margin reduction. Another issue is that, in particular at low-frequency bands, high gain corresponds to a large antenna, which in turn typically implies bulkiness and weight, features that make it unpractical to handle, mount, and maintain. Moreover, the tower must be selected or designed accordingly, both to comply with the antenna weight and wind load. Environmental considerations are gaining relevance too, as persivaseness of wireless networks and increased traffic require a huge number of antennas to be placed, especially in urban environments. As a consequence, the visual impact must be factored in the choice too, the antenna often being the most visible component of the backhaul equipment. Typical values of gain are around 30 dBi for microwave frequencies, and up to 50 dBi for E-band. Of particular importance to allow frequency reuse is the front-to-back ratio (i.e., the ratio between the main lobe and the backside lobe): backhaul antennas are expected to have this figure in the order of 60–70 dB. Sidelobe suppression is a key parameter, and it is usually provided not as a single figure, but with pattern masks, where all lobes can be included. The first sidelobe figure is often around 15 dB below the maximum directivity. One further figure to be taken into account when dealing with backhaul equipment is the 90° radiation. This is related to the side-toside placement of antennas, either for different links in a common site or when one antenna is used for TX and one is used for RX. Such issues become of paramount importance, in particular when full-duplex techniques are used (transmission and reception at the same carrier frequency), since the transmitted power is several orders of magnitude higher than the expected signal to be received. If the TX signal is allowed to leak into the RX section it is likely to saturate the sensitive RX front end and to overpower the tiny useful signal. As a possible countermeasure, one tries to keep the TX and RX paths as physically separate as possible. At the limit this

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Figure 7.4  Mask for K-band directive antenna. Left: FCC [6], category A; right: ETSI [7], class 2.

evolves into two separate antennas. These are placed side-to-side and their spacing cannot be pushed too far, as connections become unpractical. Therefore, the side-to-side rejection plays a dominant role in the characterization of the radiating element and also affects the whole architecture of the transceiver (e.g., additional digital or analog canceling techniques are needed to suppress residual leakages). To successfully adopt the dual-polarization technique for capacity improvement, it is necessary that the native cross-polar discrimination of the antenna is high enough. Typical values are around 25–30 dB. The antenna reflection coefficient is usually lower than –15 dB on the operational frequency band. The antenna must comply with regulations, like [6] from FCC, that define the minimum gain, the minimum 3-dB beamwidth, and masks for side­lobes normalized to maximum gain, or [7] from ETSI, that defines a gain mask. Figure 7.4 shows an example of mask from both standards for a K-band antenna.

7.2 Antennas for Microwave Backhaul Parabolic antennas are the most adopted in microwave backhaul. At a low frequency Yagi-Uda antennas are sometimes used, while horn antennas can have some specific advantages. This section is dedicated to a more detailed overview of parabolic reflectors and feeders, followed by a brief description of horn antennas, Yagi-Uda, and flat antennas. 7.2.1 Parabolic Antenna: Basics The parabolic antenna is composed by a feeder and a reflecting surface shaped as a paraboloid. The feeder is placed at the focus point of the parabola, and can be approximated to a source of spherical wavefront. The reflector reflects the wave back, thus shaping the wavefront into a plane, ideally. This behavior is directly

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related to a property of the parabolic curve: the path from the focus to any point of the parabola and, following its reflected ray, to a plate orthogonal to the parabola axis is constant in length, independently from the incident point on the reflector. This ray-tracing interpretation is usually accurate enough to grasp the basic behavior of the parabolic antenna, although a more precise characterization is in practice needed for the design of the reflecting surface (e.g., diffraction effects and sharp edges need extensive electromagnetic simulations). The parabolic dish (see Figure 7.5) is drawn according to:



x 2 = 4Fz

(7.5)

D2 , where D is the diameter of the parabola and F is the focal dis16 F tance. To achieve a given directivity, one can approximately define the area of the aperture and thus the diameter D. The antenna gain improves when increasing the frequency. The choice of the focal point then affects: for 0 £ z £

•• The curvature of the surface, with impact on the suitable fabrication

techniques and wind load; •• How the feeder is “enveloped” within the reflector, affecting the requirement on the radiation pattern of the feeder; •• The overall amount of material required for the reflector, affecting the overall weight. In polar coordinates, the dish is calculated as:



ρ=

2F 1 + cos ϕ

(7.6)

Figure 7.5  Parabolic antenna basic diagram.

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Figure 7.6  L eft: standard parabolic configuration, with feeder in the focus point. Right: Cassegrain configuration, with feeder at the dish center and reflector at focus point.

In practice, at least two configurations are possible using parabolic reflectors (see Figure 7.6): the classic parabolic dish, where the feeder is at the focus; or the Cassegrain configuration, where the feeder is at the center of the dish and illuminates a hyperbolic subreflector that acts by reflecting a plane wave from the feeder into a spherical wave centered in the focus of the paraboloid. One disadvantage of the classical parabola is scattering from the feed horn, support structure, and feed waveguide. This causes increased sidelobes and backlobes and reduces the cross-polar performance of the antenna. This effect can be attenuated by using an offset feed where the feed and feed structure are held out of the radiated beam. Cassegrain configuration is widely used in point-topoint microwave link antennas, with the subreflector usually mounted on a plastic support, transparent at the working frequency. This solution improves the gain and sidelobe rejection of the antenna with respect to a classical parabolic antenna. Moreover, the connection of the feeder to the transceiver is much simpler and reduces the needs of curves and bends of the waveguide that are particularly critical when a circular or square waveguide for dual polarization is adopted. Another variation is the focal plane antenna, where the dish edge extends to the focal plane. Not all of the reflector is illuminated, sacrificing part of the potential gain, but reducing significantly the sidelobes and the back radiation. To further reduce the signal scattered into sidelobes and backlobes, again at the expense of lower gain and higher antenna cost, the reflector border and the feeder can be covered with absorptive material. 7.2.2 Feeder The feeder is usually a small horn antenna (see a better explanation of horns in Section 7.2.4), with low gain, connected through a coaxial cable or a waveguide

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[8]. Coaxial cables are more lossy, but since they carry a TEM mode, they are more flexible than waveguides and usually smaller. Waveguides are difficult to bend, since the transverse electric (TE) or transverse magnetic (TM) modes can easily degenerate if curves are too narrow. Particular care must be adopted for circular or square waveguides carrying dual polarization, since in this case a large deterioration of cross polar isolation can occur in presence of narrow curves. The feeder must be designed in order to trade off uniform illumination and spillover losses. The former affects the directivity of the antenna and dictates that higher gains are attained with uniform aperture fields. As a consequence, due to the typical radiation pattern of a feeder, its main lobe must impinge the whole parabolic surface with a negligible fall-off towards the edges. On the contrary, the larger the main lobe with respect to the illuminated reflector, the higher the fraction of power that is wasted backwards, denoting spillover losses that affect the overall efficiency of the antenna. Careful design is needed for an accurate control of the two phenomena. It is important to consider that, while the reflector is nonresonant, the feeder typically determines the center frequency and the bandwidth of the antenna. For this reason, reflectors are categorized according to their diameter, and the same reflector can serve to house antennas for different frequency bands. This is a feasible option if the mechanical quality of the reflector is compatible with the working frequency; see more details in Section 7.2.3. The choice of polarization heavily affects the selection of the appropriate feeder. In fact, rectangular horns and waveguides (typically designed to have only one propagating mode through the whole operating band of the transceiver) can only sustain one polarization, while circular or square horns are used to provide dual-polarization. The two signals are separated by an ortho mode transducer (OMT), which is usually a special waveguide combiner with two connections to rectangular waveguides (see a drawing in Figure 7.7), each transporting a single mode, to the circular or square waveguide that terminates with the horn antenna. One rectangular waveguide directly matches to the circular waveguide, maintaining its polarization, while the orthogonal polarization is provided from the other rectangular waveguide coupling through the side wall of the circular waveguide via a matched aperture or a probe antenna. More complex OMTs may combine waveguides with different polarizations and operating bands into one feed horn. The OMT performs the necessary mechanical transformations to merge the two signals into a circular or square waveguide connected to the antenna. The main challenge in the design of OMTs is the accurate control of leakage of one mode into the supposedly isolated polarization (cross-polar isolation). For singularly polarized parabolic antenna at lower bands, the feeder can also be a wire antenna, usually of smaller size with respect to a horn, with the advantage of lower cost and impact on the aperture efficiency.

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Figure 7.7  Drawing example of ortho-mode transducer (OMT).

7.2.3 Parabolic Antennas in Backhaul The parabolic reflector is made of a conductive material (metal). A grid reflector can be used instead of a solid body reflector if the polarization is single and if the backside radiation is not critical. In fact, if the grid spacing is small enough compared to wavelength, it appears to the EM wave as a continuous conductor. However, back-radiation due to diffraction still appears. Moreover, the direction of the grid affects the polarization. The advantages of grid antennas are the lower weight and wind load with respect to solid body antennas [9], but this means that they may be fragile and easily damaged or punctured, for example, by the sharp talons of perching birds of prey. Solid body antennas can be fabricated using pressed sheet reflectors or precision casting metal. The former is a cheap method to form light reflectors, and is used in the vast majority of commercial applications, while the latter is more expensive and leads to heavy reflectors but guaranteeing very good accuracy in the reflector surface. Pressed sheet reflectors are the de facto standard for microwave frequencies, while casting could be required for millimeter-wave reflectors. It has to be considered that a quality antenna should present a root mean square (RMS) surface deviation typically below l/64; thus, the accuracy needed in E-band is near to 0.05 mm, hardly achievable with pressed sheet fabrication. To protect the antenna reflector and the feeder, dielectric covers called radomes can be adopted. They also permit lowering the wind load up to three times, and avoiding the accumulation of dirt, snow, and ice on the reflecting surface. Their shape and material are optimized to minimize the impact on the EM wave [9]. Another important accessory for parabolic antennas is the pressurizer, which is needed with air-dielectric feeders. It avoids the formation of moisture in the waveguide that could prevent the correct operation of the antenna [9]. All the connectors are usually vulcanized, to avoid moisture penetration, and ultraviolet (UV)-resistant materials are selected.

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Antennas are usually mechanically fixed to the tower (or to the outdoor unit of the radio, see Figure 7.8) using brackets and flanges that allow a tuning also after the installation. A skew system allows to accurately pointing the antenna along the three axes (usually fine adjustment is on the order of 10°). Initially, the antenna is pointed roughly towards the remote site and then is slightly adjusted. The horizontal alignment between antennas is more critical, and it is very important to avoid the exploitation of sidelobes, see Figure 7.9. Several practical installations involve an initial rough aiming achieved visually or by other means while the antenna is secured to its mast, followed by a fine pointing phase accomplished by adjusting some ad hoc screws. During the latter, the operator is required to act on the aiming screws in order to maximize a real-time readout voltage, proportional but not immediately translatable into a received power. As a consequence, it is rather easy to lock onto a local maximum due to a sidelobe. In clear-sky conditions, furthermore, even the lower gain from such a lobe may be sufficient to guarantee error-free communication. However, if this happens, the overall fade margin would be inevitably compromised or severely reduced, yielding to unexpected link outages. To avoid these phenomena, a large sweep on the horizontal plane is usually initially performed to guarantee that the main lobe is the one pointing toward the other antenna [9]. Commercial antennas are available with different nominal diameters, usually multiple of 1 ft (or 30 cm). Large antennas with 2.4 m of diameter are used for lower bands, and their gain can be around 40–43 dB at 7 GHz. A 30-cm antenna can have a gain that ranges from 35 dB at K-band to 45 dB in E-band. A 2.4-m antenna in E-band would practically be impossible to point, as the gain would be on the order of 60 dB and the 3 dB angle about 0.1°.

Figure 7.8  T wo pictures of microwave outdoor unites with integrated antennas. (Courtesy of SIAE Microelettronica.)

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Figure 7.9  Antenna alignment: left, incorrect; right, correct.

7.2.4 Nonparabolic Antennas 7.2.4.1 Horn Antenna

This antenna is based on an open-ended waveguide and takes the name from its horn-like (metallic) shape. The development of horn antennas started in 1900 but their wide adoption was mainly related to radars, where they were used as feeders (called feed horns) during World War II. Today horns are still employed as feeders for larger antennas like parabolic dishes and in a stand-alone feature as calibration antennas or as directive antennas, although horn antennas are still used in some backhaul applications. Figure 7.10 shows a picture of a tower with different backhaul antennas, including horns and parabolic dishes, protected by radomes. A horn antenna behaves as a gradual impedance transformer between the guided modes in the waveguide (TE and TM) and the outer space, inducing a smooth impedance transition and reducing reflections at the interface between a waveguide and free space. Without this junction, the small aperture of waveguides, lower than a wavelength in free space, can widen the radiation pattern compromising directivity. Conversely, the flared end of the waveguide in the shape of a horn reduces significantly reflections at the interface. It has also to be considered that gain of aperture antennas is proportional to their area, so the horn is a way to taper the waveguide to a larger aperture. The main difference between feeder horns and horn antennas is their aperture and consequently gain, much bigger in the latter case. The horn shape clearly influences the beam profile, and different features can be optimized according to the chosen design. Typical shapes are metallic rectangular or cylindrical tubes with a short electrical length (with respect to the working frequency) that ends into an open-ended pyramidal or conical horn; see Figure 7.11. An important parameter for the design of these antennas are the expansion curve that can be exponential (for minimum internal reflection and almost constant impedance), elliptic, hyperbolic, and so forth and involving both E-field and H-field directions or only one of the two. Mechanically, flare or taper optimization improves the impedance matching to free space and increases gain. However, the flare must be very long to exhibit good matching over a wide bandwidth. Corrugated horns (see a drawing in Figure 7.12) are popular solutions when larger bandwidths are needed (i.e., as reference antennas). They also exhibit

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Figure 7.10  Picture of a tower with backhaul horns and parabolic dishes.

Figure 7.11  Drawing of horn antennas. Left: rectangular; right: circular.

Figure 7.12  Section drawing of a corrugated horn.

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lower sidelobes with respect to standard horns, thus simplifying measurements in reflective environments. The horn is fed with a coaxial cable to form a monopole antenna or by a metallic waveguide and the guided signal is radiated by the horn. Similarly to parabolic dishes, a protective cover, transparent to microwaves, is often used to protect the aperture of the horn from moisture for outdoor placement. Horn antennas are simple to manufacture and lightweight, with bandwidth limits typically of the order of 10:1, thanks to the absence of resonating elements. The low voltage standing wave ratio is another positive feature of these antennas. Typical gains are of the order of 10 to 20 dBi (this gain may be determined analytically from the antenna dimensions [10]), making them widely used as standard gain horns, and for the characterization of other antennas. However, aperture phase errors, resulting from the different path length (and consequently different phase) of the spherical waves to reach the aperture of the antenna (see Figure 7.13), are an issue that needs to be carefully considered. This error earns relevance by increasing the flare angle and the antenna dimensions, reducing gain and widening the beamwidth. To solve this issue, lens horn antennas have been proposed: in this variant the aperture phase error is corrected by dielectric lens; see Figure 7.14. Lens horn antennas have a naturally aerodynamic shape and may reduce the wind loading. As a drawback, for low-frequency horns the length of the horn and the volume of the lens may be impractical for backhaul use.

Figure 7.13  Representation of wrong phase alignment at the horn aperture.

Figure 7.14  Representation of lens effect on phase alignment at the horn aperture.

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Figure 7.15  Picture of a lens antenna (courtesy of Flann Microwaves).

Finally, going to millimeter-wave domain, lens horns of smaller size can be adopted, reducing mass but presenting very good aperture efficiency, usually of paramount importance at these frequencies. Furthermore, these antennas are normally up to 20% smaller than equivalent gain reflectors. Lens-machining tolerances have relatively little effect on performance, despite the very small wavelength, making these antennas suitable to work up to 170 GHz or even higher frequencies. Figure 7.15 shows a picture of a lens horn antenna. 7.2.4.2 Yagi-Uda Antenna

The Yagi-Uda antenna takes its name from the two Japanese inventors Shintaro Uda and Hidetsugu Yagi from the Tohoku Imperial University of Japan [11]. It is often referred simply as a Yagi antenna, and basically is a directional antenna consisting of an array of parallel dipole elements. Only one dipole, called the driven element, is connected to the transceiver through a transmission line, while all the others can be interpreted as additional parasitic elements: a reflector and one or more directors, with the reflector longer than the driven dipole, whereas the directors are a little shorter; see Figure 7.16. The elements are spaced by a quarter-wavelength, in order to sum in phase the director’s signals to the one of the driver dipoles, increasing the antenna directivity and gain with respect to a simple dipole. The resulting radiation pattern is almost unidirectional, with the main lobe along the axis perpendicular to the elements. The gain of the antenna depends on the number of elements used, increasing roughly by a factor 1.66 for each added director, with typical values around 15–l7 dBi. Due to the highly resonant behavior, gain and bandwidth are conflicting characteristics of Yagi antennas, and a trade-off between the two must be achieved, because

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Figure 7.16  Yagi-Uda antenna drawing.

the bandwidth narrows as the number of elements increases. As a result, typical bandwidths are limited to a few percent of the center frequency. Examples of Yagi antennas designed to operate on multiple bands are available, based on the addition of LC electrical traps along each element. Thanks to its lightweight, simple design and low production cost, it is widely adopted as a high-gain antenna for home television (HF, VHF, and UHF bands) but it is also employed for point-to-point fixed communication links, especially for the low frequencies, thanks to the lower tower loading but sufficient gain if compared to parabolic dishes antennas. Furthermore, the presence of a zero RF voltage at the centre of the dipole parasitic elements allows for anchor it to a conductive metal support without insulating it or disturbing the electrical operations. 7.2.5 Printed Antennas Printed antennas are composed of microstrip resonating patches; see an example in Figure 7.17. They are arranged into arrays [12] to increase the intrinsic low gain of the single patch. Their main usage is for frequencies up to 6 GHz; however, they are becoming popular also at higher frequencies. Their main limitations are bandwidth and losses in the microstrip feed network. They do present however important advantages: they are cheap and lightweight and their low profile makes them suitable for discrete or hidden mounting on buildings.

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Figure 7.17  Example of drawing of a printed antenna.

Moreover, by properly feeding the array, a fixed beam steering can be applied, further enabling the possibility of flexible placement and installation. The radiation pattern is the product of the single element pattern and the array pattern. The array pattern can initially be designed through a theoretical approach. The single patch pattern, as well as the final array configuration, is instead usually EM simulated, in order to accurately assess the feeding network impact and the coupling between adjacent elements. Printed antennas at millimeter wave need to be fabricated through highprecision PCB manufacturing. Examples of arrays for E-band [13] and up to 100 GHz have been produced. The main limitation is the loss in the substrate, which decreases significantly the gain. Thus, increasing the area of the antenna also increases significantly the losses in the feed network due to a longer path (see Figure 7.18), and this might mean that although an antenna with increased area has a higher directivity, it has a lower gain. 7.2.6 Slotted Waveguide Antennas Slotted waveguide array antennas are a well-established technique for realizing antennas for all microwave and millimeter wave frequencies and find application in telecommunication and radar antennas. The basic idea is to draw an aperture (slot) on the conductor surface of a waveguide. The EM field will then radiate outside of the waveguide with a pattern depending on the shape, position, and dimension of the slot. The simplest slot array is linear: along a piece of waveguide, more slots are spaced to increase the directivity. This kind of antenna is common for radar and sector antennas. For point-to-point links, planar slot arrays are

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Figure 7.18  Printed antenna drawing, highlighting the feeding path.

usually of higher practical interest, as they are able to increase the directivity along both axis. One example is the flat-panel array antenna [14]. They already find application for E-band radio links, where they show sufficient gain (higher than 40 dB) and are comparable to parabolic antennas in terms of cost and weight, but advantageous in terms of shape, being thinner. These arrays can be realized on the back-metal of a microstrip board (usually for low frequencies) or by a more complex design using air waveguides (for higher frequencies). Figure 7.19 shows an example of waveguide feeding of a slotted array. An interesting application is, for millimeter-waves, the possibility to design antennas on the back of an integrated circuit [15]. However, the main difficulty in designing these arrays is, as for printed antennas, the realization of a good feeding network that must maintain the correct phase relation between the array elements and minimize losses.

Figure 7.19  Example of waveguide feeding of a panel slotted array.

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If compared to printed antennas, slotted antennas have reduced losses, particularly when based on air dielectric, also in the feeding network [16], but they are also more expensive, due to the precision required to realize the slots and to the complex lost-wax casting fabrication. Some manufacturers have proposed plastic molding [17] as a technique for realizing slotted arrays, which should sensibly decrease the fabrication costs and the weight of such structures, enabling their use on the large scale. 7.2.7 Future While the realm of microwave frequencies antennas seems to be well dominated by parabolic dishes, millimeter-wave antennas are probably a more interesting subject for the development of novel ideas and techniques. There are several aspects that can drive the research in this field. Above all is the market interest: the growth of the number of micro- and nano-cells will demand a high number of radio links to be deployed, with short high-capacity hops. Millimeter-wave links seem then to be the right choice. However, the cost of the single radio equipment needs to be lowered, and in this case antennas represent an important portion of the budget burden. In fact, the present solutions rely on expensive reflectors or slotted arrays, where the need of fabrication accuracy dominates the cost. The antenna profile is also important, given the increasing attention paid to the environmental impact of the mobile infrastructure. Crucial research fields are focused on the fabrication methods, like plastic molding or 3-D printing, and on the material technology. New materials and fabrication methods could also enable the realization of PCB printed antenna on very low-loss substrates.

Figure 7.20  E xamples of lens profiles for millimeter-wave applications. Left: simple semiconvex, right: optimized profile for minimum weight and thickness.

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Figure 7.21  P  icture of a passive repeater formed connecting two horn lens antennas (courtesy of Flann Microwave).

Millimeter waves are interesting because they can be treated as quasioptical waves, especially for higher bands that will play an important role in the future (as 150 GHz). The integration of antennas for millimeter-wave application is not a new subject [18], and the interest on this topic will surely grow for 5G applications. At these frequencies, lens systems can be adopted for beam collimation and other promising techniques [19]. For example, by proper design of the shape and material of the lens, the antenna profile can be optimized and the weight can be minimized (see some examples of lenses in Figure 7.20). The need to achieve micro-cells in urban environment will require overcoming obstacles, considering that millimeter-wave communication requires line-of-sight links. Passive repeaters (Figure 7.21) are a very established technique used to reach hidden locations, and the application to millimeter-wave links can be convenient given the high gain of antennas at that frequency. Finally, the possibility of steering the antenna beam after the installation is a topic of great interest. In fact, the discussed difficulty to align high gain antennas and the high risk of losing fade margin due to suboptimal alignment could be drastically reduced by automatic aiming. By employing phased array antennas, it is possible to steer the direction of the main lobe electronically through the actuation of variable phase shifters. This solution, already adopted in radar and satellite systems, still faces cost issues due to its complexity, but a careful addressing of design issues, together with the advancing of microwave technology, will open new possibilities for its employment in mass markets.

References [1]

Cheng, D., Field and Wave Electromagnetics, Reading, MA; Addison-Wesley, 1984.

[2]

Balanis, C., Antenna Theory: Analysis and Design, 3rd ed., New York, Wiley, 2005.

[3]

Grant, I. and W. Phillips, Electromagnetism, New York, Wiley, 1991.

[4]

Stutzman, W. and G. Thiele, Antenna Theory and Design, New York, Wiley, 2012.

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[5]

Eisenberg, M. and R. Guy, “A Proof of the Hairy Ball Theorem,” The American Mathematical Monthly, Vol. 86, No. 7, July 1979, pp. 571–574.

[6]

“FCC 47 CFR 101.115. Directional Antennas,” 2010.

[7]

“ETSI EN 302 217-4-2. Fixed Radio Systems; Characteristics and Requirements for Pointto-Point Equipment and Antennas; Part 4-2: Antennas; Harmonized EN Covering the Essential Requirements of Article 3.2 of R&TTE Directive,” 2008.

[8]

Olver, A. et al., Microwave Horns and Feeds, New York, IEE, 1994.

[9]

Manning, T., Microwave Radio Transmission Design Guide, 2nd ed., Norwood, MA; Artech House, 2009.

[10] Slayton, W., “Design and Calibration of Microwave Antenna Gain Standards,” NRL Report 4433, November 1954. [11] Yagi, H. and S. Uda, “Projector of the Sharpest Beam of Electric Waves,” Proc. of the Imperial Academy of Japan (Imperial Academy), February 1926, pp. 49–52. [12] Hansen, R. C., Phased Array Antennas, New York, John Wiley & Sons, 2009. [13] Ghassemi, N. and K. Wu, “High-Efficient Patch Antenna Array for E-Band Gigabyte Point-to-Point Wireless Services,” IEEE, Antennas and Wireless Propagation Letters, Vol. 11, November 2012, pp. 1261–1264. [14] Thomson, A., C. Biancotto, and C. Hills, “Flat Panel Array Antenna,” U.S. Patent 8,558,746, October 15, 2013 http://www.google.com/patents/US8558746. [15] Yngvesson, K. et al., “The Tapered Slot Antenna-A New Integrated Element for MillimeterWave Applications,” IEEE Transactions on Microwave Theory and Techniques, Vol. 37, No. 2, February 1989 pp. 365–374. [16] Gueye, M. et al., “Antenna Array for Point-to-Point Communication in E-Band Frequency Range,” 201I IEEE International Symposium on Antennas and Propagation (APSURSI), July 2011, pp. 2077–2079. [17] Klebe, D., “Molded Plastic Microwave Antenna,” U.S. Patent 5,495,262, February 27, 1996 http://www.google.co.uk/patents/US5495262. [18] Rebeiz, G., “Millimeter-Wave and Terahertz Integrated Circuit Antennas,” Proceedings of the IEEE, Vol. 80, No. 11, Nov 1992, pp. 1748–1770. [19] Godi, G., R. Sauleau, and D. Thouroude, “Performance of Reduced Size Substrate Lens Antennas for Millimeter-Wave Communications,” IEEE Transactions on Antennas and Propagation, Vol. 53 No. 4, April 2005, pp. 1278–1286.

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Acronyms 2-DEG

2-dimensional electron gas

2G

Second generation of mobile

3G

Third generation of mobile

4G

Fourth generation of mobile

5G

Fifth generation of mobile

ACM

Adaptive coding and modulation

ACMB

Adaptive coding modulation bandwidth

ACPR

Adjacent channel power ratio

ADC

Analog-to-digital converter

ADPD

Adaptive digital predistortion

ADSL

Asymmetric digital subscriber line

AM

Amplitude modulation

AM/AM

Amplitude to amplitude modulation distortion

AM/PM

Amplitude to phase modulation conversion

ASIC

Application-specific integrated circuit

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ASIP

Application-specific instruction set processor

ATM

Asynchronous transfer mode

AWGN

Additive white gaussian noise

BCH

Bose Chaudhuri Hocquenghem

BCJR

Bahl Cocke Jelinek Raviv

BEOL

Back end of line

BER

Bit error rate

BICM

Bit interleaved coded modulation

BiCMOS

Bipolar complementary metal oxide semiconductor

BJT

Bipolar junction transistor

BMU

Branch metric unit

CAD

Computer-aided design

CC

Convolutional code

CCDP

Co-channel dual polarization

CDMA

Code division multiple access

CG

Common gate

CIMR

Carrier-to-intermodulation ratio

CMOS

Complementary metal oxide semiconductor

CORDIC

Coordinate rotation digital computer

CS

Common source

CSD

Canonic sign digit

CW

Continuous wave

DAC

Digital-to-analog converter

DDS

Direct digital synthesis

DNL

Differential nonlinearity

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DPD

Digital predistortion

DRRS

Digital radio relay system

DSP

Digital signal processor

EDGE

Enhanced Data rates for GSM Evolution

309

EM Electromagnetic ENOB

Equivalent number of bits

ETSI

European Telecommunications Standards Institute

EVC

Ethernet virtual connection

EVM

Error vector magnitude

FCC

Federal Communications Commission

FDD

Frequency division duplexing

FDM

Frequency division multiplexing

FEC

Forward error correction

FET

Field effect transistor

FIR

Finite impulse response

FLL

Frequency locked loop

FM

Frequency modulation

FoM

Figure of merit

FPGA

Field programmable gate array

FSL

Free-space loss

FSO

Free-space optical

GPRS

General Packet Radio Service

GPS

Global Positioning System

GPU

Graphics Processor Unit

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GSM

Global System for Mobile Communications

GTE

General Telephone and Electronics

HB

Harmonic balance

HBT

Heterojunction bipolar transistor

HEMT

High electron mobility transistor

HSDPA

High-speed downlink packet access

I/Q In-phase/quadrature IBO

Input back-off

IF

Intermediate frequency

IIP3

Third-order input intercept point

IMD

Intermodulation distortion

INL

Integral Nonlinearity

ITT

International Telephone and Telegraph

ITU

International Telecommunication Union

LDMOS

Laterally diffused metal oxide semiconductor

LDPC

Low density parity check

LFSR

Linear feedback shift registers

LIFO

Last in first out

LLR

Logarithmic likelihood ratios

LMS

Least mean square

LNA

Low noise amplifier

LO

Local oscillator

LOS

Line of sight

LPF

Lowpass filter or filtering

LTE

Long-term evolution

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LUT

Look-up table

MAP

Maximum a posteriori

MIC

Microwave integrated circuit

MIM

Metal insulator metal

MIMO

Multiple input multiple output

ML

Maximum likelihood

MLSE

Maximum likelihood sequence estimation

MMIC

Microwave monolithic integrated circuit

MMSE

Minimum mean squared error

MOSFET

Metal oxide semiconductor field effect transistor

NCO

Numerically controlled oscillator

NoC

Network on chip

OBO

Output back-off

OFDM

Orthogonal frequency-division multiplexing

OIP3

Third-order output intercept point

OMT

Ortho mode transducer

PA

Power amplifier

PAE

Power-added efficiency

PAM

Pulse amplitude modulation

PAPR

Peak to average power ratio

PCB

Printed circuit board

PCM

Pulse code modulation

PDD

Polarization division duplexing

PDF

Probability density function

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PDH

Plesiochronous digital hierarchy

PHEMT

Pseudomorphic high electron mobility transistor

PLL

Phase lock loop

PN

Pseudo noise

PoE

Power over Ethernet

PPM

Pulse position modulation

PRC

Primary reference clock

PtP Point-to-point

QAM

Quadrature amplitude modulation

QFN

Quad flat no leads

QPSK

Quadrature phase shift keying

RAG

Reduced adder graph

RF

Radio frequency

RFIC

Radio frequency integrated circuit

RLS

Recursive least square

RMS

Root mean square

RS

Reed Solomon

RSL

Receiver sensitivity level

RX Receiver

SAW

Surface acoustic wave

SDH

Synchronous digital hierarchy

SES

Severely errored second

SFDR

Spurius-free dynamic range

SIM

Subscriber identity module

SINAD

Signal noise and distortion

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SiP

System in package

SISO

Soft in soft out

SMS

Short messaging service

SNR

Signal-to-noise ratio

SoC

System on Chip

SONET

Synchronous optical networking

SRRC

Square root raised cosine

SSB

Single side band

TCM

Trellis coded modulation

TCO

Total cost of ownership

TCP/IP

Transmission Control Protocol/Internet Protocol

TDD

Time division duplexing

TDM

Time division multiplexing

TE

Transverse electric

TEM

Transverse electromagnetic

TM

Transverse magnetic

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TR Transceiver TWT

Traveling wave tube

TX Transmitter

UMTS

Universal Mobile Telecommunications System

UV Ultraviolet UWB Ultrawideband VCCS

Voltage controlled current source

VCO

Voltage controlled oscillator

VOIP

Voice over Internet Protocol

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VoIP

Voice over IP

VPN

Virtual private network

WCDMA

Wideband code division multiple access

WiMAX

Worldwide interoperability for microwave access

XPD

Cross-polar discrimination

XPIC

Cross-polarization interference cancellation

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About the Editors Vittorio Camarchia received the Laurea degree in electronic engineering and a Ph.D. in electronic and communications engineering from the Politecnico di Torino, Turin, Italy, in 2000 and 2003, respectively. In 2001, 2002, and 2003, he was a visiting researcher with the Electrical and Computer Engineering Department, Boston University, Boston, Massachusetts. He is currently an associate professor of electronics with the Electronics and Telecommunication Department, Politecnico di Torino, Turin, Italy. His research is focused on the design and experimental characterization of hybrid and integrated circuits and systems for microwave RF applications both linear and non-linear (scattering, source/ load-pull technique, system-level characterization) and large signal circuit and system level modeling of microwave power devices. He is author or co-author of around 120 technical publications in journal papers and conference proceedings as well as two book chapters and many technical reports. Roberto Quaglia received the Laurea degree and a Ph.D. from Politecnico di Torino, Italy, in 2008 and 2012, respectively. He has been a visiting Ph.D. student at Chalmers University, Sweden, working on reconfigurable microwave circuits. From 2012 to 2014, he was a post doctoral fellow at the Department of Electronics and Telecommunications at Politecnico di Torino, where he carried out his research on microwave power amplifiers theory, design, and characterization. In 2015 he worked for Huawei Technologies in Milano, Italy, as an MMIC designer for millimeter-wave radios. He is currently a researcher at the Centre for High Frequency Engineering at Cardiff University, working on the research project Power Amplifier Design for Wideband Communications (PADWIC) on high-efficiency broadband power amplifiers, supported by the European Union’s 315

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Horizon 2020 research and innovation program under a Marie SkłodowskaCurie grant agreement. Marco Pirola received the Laurea degree in electronic engineering and a Ph.D. degree from Politecnico di Torino, Italy, in 1987 and 1992, respectively. In 1992 and 1994, he was a Visiting Researcher at the Hewlett Packard Microwave ­Technology Division, Santa Rosa, California. Since 1992, he has been with the Electronics and Telecommunication Department, Politecnico di Torino, first as a researcher and, since 2000, as an associate professor of electronics, where his research concerns the simulation, modeling, and measurements of microwave devices and systems. He is author or coauthor of around 160 technical publications in journal papers and conference proceedings as well as some book chapters and many technical reports.

Residential addresses of the Authors: Vittorio Camarchia: Via della Bocciofila 5/B 10053 Bussoleno (TO) ITALY [email protected] Roberto Quaglia 13 Pen Bryn Hendy Pontyclun CF72 8QX Mid Glamorgan Wales UK [email protected] Marco Pirola Via Stretta 10 28021 Borgomanero (NO) ITALY [email protected]

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Index ideal input-output staircase characteristic, 185 input-output characteristics, 186 introduction to, 181 noise sources, 184–89 pipelined, 191–92 sample-and-hold module, 188 sampling frequency, 183–84 successive approximation, 192–93 symbol detection, 71 time-interleaved, 193–94 track-and-hold module, 187 See also Receivers Angelov model, 233 Antennas, 285–305 bandwidth, 290 basic concepts, 285–91 in block scheme, 55 characteristics of, 287–90 circularly polarized, 289 dipole radiation, 286 directivity, 288 dual polarized, 289, 291 feeder, 293–95 figures of merit, 290 future, 304–5 gain, 288 high gain, 23 horn, 297–300

Active mixers, 175 Adaptive coding, modulation and bandwidth (ACMB), 67 Adaptive coding and modulation (ACM), 67 Adaptive communication techniques, 66 Adaptive digital predistortion, 97 Add-drop multiplexers (ADMs), 14 Additive white gaussian noise (AWGN) ­channel, 72 Adjacent channel power ratio (ACPR), 215–16 AlfoPlus80 HD transceiver baseband analog processing, 197–98 connection to single antenna, 195 illustrated, 194 RF front end, 195, 196 RF system in package, 197 Amp-ADC-LPF, 131 Amplitude imbalance correction, 101 Amp-LPF-ADC, 131 Analog IF predistortion, 278 Analog microwave predistortion, 277–78 Analog-to-digital converters (ADCs), 181–94 architecture, 189–94 baseband analog processing, 129 effective number of bits (ENOB), 182 flash, 190–91 folding, 191

317

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isotropic radiator, 287–88 linearly polarized, 289 for microwave backhaul, 291–305 mismatch, 289–90 nonparabolic, 297–301 overview, 285–87 parabolic, 291–93, 295–97 printed, 301–2, 303 radiated power, 288–89 requirements for microwave backhaul, 290–91 sidelobes, 289 slotted waveguide, 302–4 uniform aperture, radiation pattern, 289 Yagi-Uda, 300–301 Architecture, 49–66 analog-to-digital converters (ADCs), 189–94 block scheme, 49, 50 branching, 61–63 building blocks, 53–61 capacity improvement, 63–66 check-node-centric, 86 full indoor, 50 full outdoor, 52–53 local oscillators, 164–66 mixers, 175–79 options, 49–53 receiver, 118–31 split mount, 50–51 Asynchronous transfer mode (ATM), 15–16 defined, 15 efficiency, 15 synchronization and, 18 AT&T microwave links, 38–39 Available gain, 139 Backhaul block scheme, 49, 50 as cost driver, 32 defined, 5–6 Ethernet-based, 7 functions, 6–7 line-of-sight links, 90 link budget, 118 microwave, 8–11 mobile evolution influence, 7–8 operation example, 6 PA requirements, 216–20

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qualitative costs versus capacity comparison, 31 upconverter examples, 268–72 See also Microwave backhaul Bahl, Cocke, Jelinek, and Raviv (BCJR) algorithm, 83 Balanced mixer, 178, 263 Balanced PAs, 241 Baluns, 264 Barkhausen-Kurz tube, 34–35 Baseband analog processing AlfoPlus80 HD transceiver, 197–98 Amp-ADC-LPF, 131 Amp-LPF-ADC, 131 LPF-Amp-ADC, 129–31 overview, 129 Baseband unit, 61 Base stations, data capacity capabilities, 44 Biasing, 150 Bipolar junction transistors (BJTs), 174 Bit interleaved coded modulation (BICM), 89, 90 Block scheme antenna, 55 baseband unit, 59 branching, 54, 61–63 IF board and modem, 59 illustrated, 50 microwave module, 55–59 modem, 68 power supply, 59 role in present system, 53–61 split mount configuration, 53 Branching, 61–63 connection, 54 defined, 61 frequency diversity, 62 hot-standby, 62 illustrated, 62 spatial diversity, 63 Branch metric units (BMUs), 84 Buffered oscillator, 164 Canonic sign digit (CSD), 70–71 Capacity improvement adaptive communication techniques, 66 dual polarization, 64–65 line-of-sight MIMO, 65 modulation and roll-off, 64

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Index Carrier recovery, 105 Carrier to intermodulation ratio (CIMR), 213–14 Cascode configurations common modifications, 151 input impedance, 155 LNA with external resistance, 151–52 LNA with inductance between transistors, 152–56 LNA with inductance on gate, 156–57 LNA with Õ-input matching network, 157–58 matching network for, 156 overview, 151 Cassegrain configuration, 293 CDMA, 3 CDMA2000, 4 Check-node-centric architecture, 86 Circularly polarized antennas, 289 Clapp oscillator, 169–70, 173 Class AB PAs average power level, 220 circulation angle, 224–25 gain versus input drive, 226 impact on performance at full input drive, 225 limitations on back-off efficiency, 226–31 linearity, 224–26 tuned load, 220–31 See also Power amplifiers (PAs) Class A PAs circulation angle, 224 compression, 226 DC power, 222, 227 efficiency versus IBO, 226–27 output characteristic, 222 waveforms, 223 Class B PAs circulation angle, 224 DC power, 227 efficiency versus IBO, 226–27 gain expansion, 226 intrinsic dynamic load lines, 235 maximum efficiency, 224 waveforms, 225 Co-channel dual polarization (CCDP), 88, 89 Colpitts oscillator, 167–68, 172 Constant gain circles, 142–43 Constant noise circles, 146

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Convolutional turbo codes, 83–85 Coordinate Rotation Digital Computer (CORDIC) architecture, 79 defined, 76 NCO implementations, 78 Cross-coupled oscillator complementary, 171 configurations, 170 defined, 170 push-push, 173 Cross polar discrimination (XPD), 88 Cross polarization interference cancellation (XPIC), 31, 268 DC offset, 103, 125 Differential nonlinearity (DNL), 187 Digital IF downconversion, 126–27 Digital predistorters (DPDs) block scheme of memory polynomial, 279 block scheme with TX, 279 coefficients, 96 defined, 278 illustrated, 96 use of, 279–80 Digital radio relay systems (DRRSs), 91, 95 Digital signal processor (DSP) equalizers, 41 Digital-to-analog converters (DACs), 272–76 basis, 272–74 block scheme, 273 characteristics of, 274 defined, 272–73 figures of merit, 274–75 hybrid solutions, 275 I/Q input signals to, 54 for microwave radio, 275–76 nonidealities in, 192 output voltage versus time, 273, 274 symbol generation, 70 working principle, 273 Direct conversion AM detection and, 125–26 DC offset and, 125 defined, 124 flicker noise and, 125 image frequencies and, 124 I/Q imbalance and, 125 LO radiation and, 126 LO-to-RF leakage and, 125

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Electronics for Microwave Backhaul

Direct conversion (homodyne) defined, 256–57 injection pulling, 258–59 local oscillators, 257–59 oscillator pulling, 258 scheme illustration, 258 Direct digital synthesis (DDS), 76 Directivity, antenna, 288 Doherty PA block scheme, 228 dynamic load lines, 229 envelope tracking comparison, 230 linearity, 231 total output power, 228 waveforms, 229 Double-balanced mixer, 178–79 Double-IF superheterodyne downconversion, 123 Downconversion digital IF, 126–27 direct, 124–26 low IF, 126 options, 121–22 strategies comparison, 127–29 strategy, 121 superheterodyne, 122–24 See also Receivers Drivers, 242 Dual polarization block scheme, 65 defined, 64–65 Dual polarized antennas, 289, 291 Duplexing techniques, 115–17 Earth curvature, 21, 22 E-band PAs, 252–56 Effective number of bits (ENOB), 182 Envelope tracking PAs, 229–30 Equalization, 80–82 Error performance, 19 Error vector magnitude (EVM), 216 Ethernet protocol, 16–17 Ethernet routers, 54 Ethernet Virtual Connection (EVC), 17 ETIS emission masks, 219 Fading multipath, 24, 25 precipitation and, 23–24

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Feedback network, 154 Feeder antennas, 293–95 FET multiplier, 268 FET resistive mixer, 262 Field-effect transistors (FETs), 174, 175, 232 Field programmable gate array (FPGA), 71, 74 Flash ADCs, 190–91 Flicker noise, 125, 162–63 Folding ADCs, 191 Forward error correction analog-to-digital converters (ADCs), 182 convolutional turbo codes, 83–85 hardware, 87–88 LDPC codes, 85–86 schemes, 83 Fourth generation (4G) mobile, 2 Free space loss (FSL), 20, 21 Free-space optical (FSO) networks, 41 Frequency bands backhaul, 10 mobile networks, 3 Frequency diversity branching, 62 Frequency division duplexing (FDD), 116 Frequency division multiplexing (FDM), 40 Frequency-locked loops (FLLs), 165, 166 Frequency multipliers characteristics of, 272 FET-based, 268 upconversion, 267–68 upconverter examples, 271–72 Frequency response, LNAs, 145–47 Frequency-selective I/Q transfer function, 105 Friis’ formula, 121 Full indoor configuration, 50 Full outdoor configuration architecture, 52–53 illustrated, 52 microwave board in, 57 sketch drawing, 52 GaAs Doherty PA (K-band) defined, 248 electrical scheme, 253 measured results, 252 microscopic picture, 251 GaAs linear PA (Ka-band), 251–52, 254 GaAs linear PA (Ku-band), 248, 251 GaAs technology, 132

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321

Gain, antenna, 288 GaN HEMTs, 233–34, 237 GaN MMIC PA (7-GHz) circuit schematic, 245, 249, 250 gain and efficiency versus output power, 243, 244 illustrated, 243 microscopic pictures, 244, 246 single tone performance, 247 GaN technology, 134–35 Gate-to-drain capacitor, 155 Gilbert cell bipolar technology, 179 deep analysis of, 181 defined, 179 output, 180 simplified structure, 180 Global System for Mobile Communications (GSM), 1–2

defined, 297 drawing, 297 lens, 300 tower picture, 297 Hot-standby branching, 62

Hardware FEC, 87–88 Hardware synchronization, 82–83 Harmonic balance (HB) simulations, 236 Harmonic distortion local oscillators, 160–62 measurement, 161 reduction of, 161–62 Hartley oscillator, 168–69 HBT VCOs, 270–71 Hertz, Heinrich R., 33–34 Heterodyne upconversion defined, 259 harmonic distortion, 260 single-stage double sideband, 260 Heterojunction bipolar transistors (HBTs), 132, 136, 254 High electron mobility transistors (HEMTs) Angelov model, 233 GaN, 233–34, 237 large-signal model, 232 modeling of, 232 temperature and, 233 trap effects and, 233 High gain antennas, 23 Horn antennas aperture, 298 behavior of, 297 characteristics of, 298 corrugated, 297

Lange couplers, 264 Laterally diffused metal oxide semiconductor (LDMOS), 234 LDPC codes, 85–86 decoding, 73 defined, 83, 85 multistandard decoders for, 88 WiFi, 88 WiMAX, 88 Least mean square (LMS) algorithm, 81, 82–83 Lens horn antenna, 300 Lens profiles, 304 Linear feedback shift registers (LFSRs), 79, 80 Linearity class AB PAs, 224–26 Doherty PA, 231 high-voltage conditions and, 150 power amplifiers (PAs), 213 Linearization techniques analog IF predistortion, 278 analog microwave predistortion, 277–78 basis, 276–77 digital predistortion, 278–80 predistortion function, 277 Linearly polarized antennas, 289 Line-of-sight (LOS) backhaul links, 90 Line-of-sight MIMO, 65 Link budget, 118–19

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IF board, 59 Integral nonlinearity (INL), 187 Interference cancellation in case of frequency reuse, 88–91 in microwave radio networks, 25 Intermodulation distortion (IMDs), 213–14 Inter-Working Function (IWF), 16 Inverse multiplexing of ATM (IMA), 15 I/O delay imbalance compensation, 103 I/Q imbalance, 125 I/Q upconverter, 263–64 Isotropic radiator, 287–88

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Electronics for Microwave Backhaul

Link dimensioning, 18–26 fading, 23–25 interference, 25 point-to-point propagation, 20–23 reliability objectives, 18–20 LMS-based adaptive equalizer, 81–82 Load pull setup, 235 Local oscillators, 158–73 architectures, 164–66 buffered oscillator, 164 Clapp oscillator, 169–70, 173 Colpitts oscillator, 167–68, 172 cross-coupled oscillator, 170–71 design, 159–60 direct conversion (homodyne), 257–59 fundamentals, 158–63 harmonic distortion, 160–62 Hartley oscillator, 168–69 leakage correction, 102 phase noise, 162–63 push-push oscillator, 171–73 residual leakage, 263 startup condition, 160 steady state, 160 synthesized and locked oscillator, 165–66 topologies, 166–73 voltage controlled oscillator (VCO), 164–65 See also Receivers Locked oscillators, 166 Logarithmic likelihood ratios (LLRs), 72, 73–74 Log-MAP, 73, 74 LO-to-RF leakage, 125 Low IF downconversion, 126 Low noise amplifiers (LNAs), 138–58 biasing, 150 branching, 54 cascode amplifier equivalent model, 149 cascode configurations, 151–58 configurations with bipolar transistors, 148 configuration with feedback resistor, 152 constant gain circles, 142–43 definitions, 139–41 effective design factors, 147–48 with external resistance, 151–52 frequency response, 145–47 functions of, 138–39 fundamentals, 139–47 gain, 121

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general properties, 148–50 with inductance between transistors, 152–56 with inductance on gate, 156–57 linearity, 150 noise figure, 143–45 with Õ-input matching network, 157–58 output, 150 power gain, 139 stability, 141 topologies, 147–50 LPF-Amp-ADC, 129–31 LTE, 2, 7, 8, 212 LTE-Advanced, 8 Magnetron, 37 Marconi, Guglielmo, 33–34, 35–36 Matching network synthesis, 239 Maximum a posteriori (MAP), 71 Maximum likelihood estimation sequence (MLSE), 82–83 Max-Log-MAP, 73, 74 Metal insulator metal (MIM) tuning capacitors, 238 Microwave backhaul antenna requirements for, 290–91 antennas for, 291–305 benefits, 8–9 frequency bands, 10 network implementation, 9 penetration, 43 point-to-point, block diagram, 116 PtP links, 9–10 trends and expected growth, 44–46 Microwave hops, 32–33 Microwave industry, 37–39 Microwave links AT&T, 38–39 analog to digital, 40–42 Barkhausen-Kurz tube, 34–35 defined, 30 historical evolution, 33–42 industry growth, 30 Telettra, 37–38, 39 today, 42–43 Microwave module, 55–59 Microwave radio applications, 42–43 benefits and disadvantages, 31–33

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Index DACs for, 275–76 in high-capacity trunk routes, 40 introduction, 29 line-of-sight requirement, 32 pioneers, 33–34 SDH and, 30–31 wired versus, 29–31 Microwave spectrum, 44–45 Miller theory, 154 Millimeter-wave technology, 45, 305 Minimum mean squared error (MMSE), 90 Mixers, 173–81 active, 175 architectures, 175–79 balanced, 178, 263 design flow, 264 double-balanced, 178–79 efficiency, 175 families comparison, 261 FET resistive, 262 fundamentals, 174–75 Gilbert cell, 179–81 introduction, 173–74 passive, 175 single-ended diode, 176–77 single-ended transistor, 177 single-sideband, 261 in upconverters/downconverters, 261 See also Receivers Mobile evolution backhaul and, 7–8 numbers, 3–5 services, 1–3 Mobile networks frequency bands, 3 infrastructure block scheme, 5 Mobile subscriptions, 4–5 Modem, 67–108 block scheme, 68 board, 59, 60 countermeasures against TX and RX chains, 91–108 modulations and coding, 67–91 Monolithic microwave integrated circuits (MMICs) advantages of, 218 blocks of, 58 GaN, 237 research, 46

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323

scalable models, 233 upconverter, 269–70 MOSFET devices, 154 Multipath propagation illustrated, 24 spectrum notch due to, 26 Multiple input multiple output (MIMO) 4G network use, 4 defined, 89 line-of-sight, 65 NCO-based sine wave generator, 77 Negative resistance oscillator, 267 Noise factor defined, 120 evaluation of, 144 Noise figure defined, 120 evaluation of, 144 low noise amplifiers (LNAs), 143–45 model for evaluation, 144 Noise sources quantization error, 184–86 sampling distortion, 187–88 sampling jitter, 188–89 static errors, 186–87 thermal noise, 188 See also Analog-to-digital converters (ADCs) Noise temperature, 120 Nonlinear distortion, 95–98 amplitude, 95 digital predistortion, 96–97 phase, 95 Nonlinearities, 174 Nonparabolic antennas, 297–301 Normalized driving level (IBO), 226 Numerically controlled oscillator (NCO), 75–78 N × STM-1 microwave system, 42 Nyquist criterion, 183 128-QAM, 106 Operating gain, 139 Organization, this book, 26–78 Ortho mode transducers (OMTs), 294–95 Parabolic antennas alignment, 297 in backhaul, 295–97

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Electronics for Microwave Backhaul

Cassegrain configuration, 293 components of, 291 diagram, 292 dish calculation, 292 focal point, 292 nominal diameters, 296 pressurizer, 295 reflector, 291–92 solid body, 295 standard configuration, 293 tower picture, 297 Passive mixers, 175 Peak to average power ratio (PAPR) defined, 212 normal operative condition, 212 power amplifiers (PAs), 211–13 reduction of, 69 Phase-locked loops (PLLs), 165, 166, 257, 265 Phase noise different levels of, 106 local oscillators, 162–63 power density, 163 sources, 162–63 SSB output, 163 as TX and RX impairment, 105–8 Pilot symbol phase estimator, 108 Pipelined ADCs, 191–92 Plesiochronous digital hierarchy (PDH), 11–12, 41 efficiency, 15 multiplexers, 54 multiplexing, 12 primary links, 18 scrambling, 12 Point-to-point (PtP) links, 9–10 Point-to-point communication high gain antennas, 23 line-of-sight, 20 microwave applications, 42–43 standards, 42 Point-to-point propagation, 20–23 Polarization division duplexing (FDD), 116–17 Power added efficiency (PAE), 209 Power amplifiers (PAs), 206–56 adjacent channel power ratio (ACPR), 215–16 backhaul examples, 242–56 backhaul requirements, 216–20

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in backhaul systems, 208 back-off, 211 balanced, 241 behavior of, 205 bias points, 221 branching, 54 carrier to intermodulation ratio (CIMR), 213–14 class A, 222–27 class AB, 220–31 class B, 224–27, 235 classifications, 220 compression, 210–11 DC to AC energy transfer efficiency, 207 defined, 206 descriptive function, 214–15 design strategies, 231–40 Doherty, 228–31 E-band, 252–56 efficiency, 209–10 envelope tracking, 229–30 error vector magnitude, 216 fundamentals, 207–16 GaAs Doherty (K-band), 248–51 GaAs linear (Ka-band), 251–52, 254 GaAs linear (Ku-band), 248, 251 GaN MMIC (7-GHz), 243–48 input and output matching networks, 237–40 input power, 208 integrated designs, 238 intermodulation distortion (IMDs), 213–14 introduction to, 206–7 linear behavior, 207 linearity, 213 microwave backhaul, devices for, 231–34 mission, 207 optimum load in real devices, 234–37 output power, 207 peak to average power ratio (PAPR), 211–13 power added efficiency (PAE), 209 power and gain, 207–9 power combining, drivers, and stability, 240–42 qualities related to operation, 208 saturated output power requirements, 217 scheme modeling active device, 222

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Index single-tone test, 208 See also Transmitters Power gain, 139 Power splitting/combination, 240 Power supply, 61 Predistortion analog IF, 278 analog microwave, 277–78 digital, 278–80 introduction for linearization, 277 Printed antennas, 301–2, 303 Probability density function (PDF), 212 Pseudomorphic high electron mobility transistors (PHEMTs), 132, 232 Pulse amplitude modulation (PAM), 71, 74 Push-push oscillator advantages of, 172 based on Clapp configuration, 173 based on Colpitts configuration, 172 based on cross-coupled configuration, 173 defined, 171–72 drawback of, 172 Quadrature amplitude modulation (QAM), 64, 67, 68–75 16-QAM, 69 32-QAM, 69, 70 64-QAM, 74–75, 107 128-QAM, 106 256-QAM, 67 1024-QAM, 107 demodulator and digital processing, 71 hardware symbol detection, 73–75 hardware symbol generation, 70–71 modulated signal representation, 99 nonsquare modulations, 69 symbol detection, 71–73 symbol generation, 69–70 symbol generation and shaping, 70 Quadrature error correction, 101 Quadrature modulation imperfections amplitude imbalance correction, 101 I/O delay imbalance compensation, 103 LO leakage correction, 102 quadrature error correction, 101 RX compensation, 104 TX compensation, 100 types of, 98–99 Quantization error, 184–86

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Radio frame generation, 68 Radio frequency integrated circuits (RFICs), 218 Radio interference capacity (RIC), 218 Rain intensity, 25 Rain zones, 24 Receivers, 115–98 analog-to-digital conversion (ADC), 181–94 architecture, 118–31 baseband analog processing, 129–31 block diagram, 118 downconversion, 121–29 duplexing techniques, 115–17 introduction, 115 local oscillators, 158–73 low noise amplifiers (LNAs), 138–58 mixers, 173–81 noise considerations, 118–21 noise figure, 119 real-world example, 194–98 signal-to-noise ratio (SNR), 45, 117 superheterodyne, 122 tasks, 117–18 Receiver sensitivity level (RSL), 22 Receiver technologies GaAs and III-V-based, 132 GaN, 134–35 manufacturing costs and capabilities, 137 for millimeter-wave applications, 135–37 overview, 131–32 practical considerations, 138 Si and SiGe BiCMOS, 133 Si RF-CMOS, 133–34 Recursive least square (RLS), 81 Reliability objectives, 18–20 Resource Reservation Protocol (RSVP), 17 Root mean square (RMS) surface, 295 RX compensation block diagram, 94 introduction and historical background, 91–95 nonlinear distortion, 95–98 other impairments, 108 phase noise, 105–8 quadrature modulation imperfections, 104

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Electronics for Microwave Backhaul

Sampling clock, 183, 184, 198 Sampling distortion, 187–88 Sampling frequency, 183–84 Sampling jitter, 188–89 Scattering matrix, 140 Second generation (2G) mobile, 2 Short Messaging Services (SMS), 2 Sidelobes, 289 SiGe BiCMOS technology, 133, 136, 252 Signal noise and distortion (SINAD), 182 Single-ended diode mixer, 176–77 Single-ended transistor mixer, 177 Single sideband converters, 260–61 Single-tone test, 208 Si RF-CMOS technology, 133–34 Si technology, 133 16-QAM, 69 64-QAM, 74–75, 107 Slotted waveguide antennas applications, 302–3 defined, 302 printed antennas versus, 304 waveguide feeding illustration, 303 Soft in soft out (SISO) block scheme, 84, 85 Spatial diversity branching, 63 Spectral efficiency classes, 219 Spectrum notch, 26 Spherical wavefront propagation, 287 Split mount configuration, 50–51 detailed block scheme, 53 direct mount, 51 illustrated, 51 modem board, 60 with multiple baseband units, 54 separate mount, 51 Splitters, 264 Square root raised cosine (SRRC), 70 Stability LNAs, 141 PAs, 240–42 VCOs, 266 Static errors, 186–87 Stuffing, 12 Successive approximation ADCs, 192–93 Superheterodyne downconversion, 122–24 Synchronization, 17–18 carrier, 75–78 Ethernet network, 18 frame, 78–80

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hardware, 82–83 timing, 75 types of, 75 Synchronous digital hierarchy (SDH), 12–15, 41 defined, 12–13 efficiency, 15 microwave radio and, 30–31 multiplexing, 13 network organization, 14 ring protection of, 19 SONET compatibility, 14 Synthesized oscillators, 165–66 System in Package (SiP), 58 Telettra microwave links, 37–38, 39 Thermal noise, 188 Thermal noise power, 120 Third generation (3G) mobile, 2 32-QAM, 69, 70 Time division duplexing (TDD), 117 Time division multiplexing (TDM), 4–5, 16, 41 Time-interleaved ADCs, 193–94 Transducer gain, 139 Transmitters, 205–80 digital-to-analog conversion, 272–76 introduction to, 205–6 linearization techniques, 276–80 power amplifier (PA), 206–56 upconversion, 256–72 Transport networks asynchronous transfer mode (ATM), 15–16 Ethernet protocol, 16–17 plesiochronous digital hierarchy (PDH), 11–12 synchronization, 17–18 synchronous digital hierarchy (SDH), 12–15 Transverse electromagnetic (TEM) mode, 286, 294 Traveling-wave tube (TWT), 36 256-QAM, 67 TX compensation block diagram, 93 introduction and historical background, 91–95 nonlinear distortion, 95–98

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Index other impairments, 108 phase noise, 105–8 quadrature modulation imperfections, 100 Uniform aperture antenna, 289 Upconversion backhaul examples, 268–72 direct conversion (homodyne), 256–59 frequency multipliers, 267–68 heterodyne, 259–60 introduction to, 256 mixers, 261–64 single sideband, 260–61 voltage controlled oscillators (VCOs), 265–67 Upconverters backhaul examples, 268–72 characteristics of, 269 frequency multipliers, 271–72 HBT VCOs, 270–71 MMICs, 269–70 Variable gain amplifier (VGA), 129 Virtual Private Network (VPN), 17 Voice over IP (VoIP), 2, 42

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327

Voltage controlled current source (VCCS), 232 Voltage controlled oscillators (VCOs) constant, 165 cross-coupled, 170 defined, 164 frequency limitation, 267 as frequency-selective feedback system, 266 frequency stability, 266 HBT, 270–71 LO frequency control, 122 in phase locked loop, 265 spectral purity, 265 symbol detection, 71 tuning mechanisms, 165 upconversion, 265–67 Wave polarization, 286 Wilkinson dividers, 240 WiMAX-A standard, 2 Yagi-Uda antenna, 300–301 Zero-forcing (ZF), 90

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Artech House Microwave Library Behavioral Modeling and Linearization of RF Power Amplifiers, John Wood Chipless RFID Reader Architecture, Nemai Chandra Karmakar, Prasanna Kalansuriya, Randika Koswatta, and Rubayet E-Azim Control Components Using Si, GaAs, and GaN Technologies, Inder J. Bahl Design of Linear RF Outphasing Power Amplifiers, Xuejun Zhang, Lawrence E. Larson, and Peter M. Asbeck Design Methodology for RF CMOS Phase Locked Loops, Carlos Quemada, Guillermo Bistué, and Iñigo Adin Design of CMOS Operational Amplifiers, Rasoul Dehghani Design of RF and Microwave Amplifiers and Oscillators, Second Edition, Pieter L. D. Abrie Digital Filter Design Solutions, Jolyon M. De Freitas Discrete Oscillator Design Linear, Nonlinear, Transient, and Noise Domains, Randall W. Rhea Distortion in RF Power Amplifiers, Joel Vuolevi and Timo Rahkonen Distributed Power Amplifiers for RF and Microwave Communications, Narendra Kumar and Andrei Grebennikov Electronics for Microwave Backhaul, Vittorio Camarchia, Roberto Quaglia, and Marco Pirola, editors EMPLAN: Electromagnetic Analysis of Printed Structures in Planarly Layered Media, Software and User’s Manual, Noyan Kinayman and M. I. Aksun An Engineer’s Guide to Automated Testing of High-Speed Interfaces, José Moreira and Hubert Werkmann

Envelope Tracking Power Amplifiers for Wireless Communications, Zhancang Wang Essentials of RF and Microwave Grounding, Eric Holzman FAST: Fast Amplifier Synthesis Tool—Software and User’s Guide, Dale D. Henkes Feedforward Linear Power Amplifiers, Nick Pothecary Filter Synthesis Using Genesys S/Filter, Randall W. Rhea Foundations of Oscillator Circuit Design, Guillermo Gonzalez Frequency Synthesizers: Concept to Product, Alexander Chenakin Fundamentals of Nonlinear Behavioral Modeling for RF and Microwave Design, John Wood and David E. Root, editors Generalized Filter Design by Computer Optimization, Djuradj Budimir Handbook of Dielectric and Thermal Properties of Materials at Microwave Frequencies, Vyacheslav V. Komarov Handbook of RF, Microwave, and Millimeter-Wave Components, Leonid A. Belov, Sergey M. Smolskiy, and Victor N. Kochemasov High-Linearity RF Amplifier Design, Peter B. Kenington High-Speed Circuit Board Signal Integrity, Stephen C. Thierauf Integrated Microwave Front-Ends with Avionics Applications, Leo G. Maloratsky Intermodulation Distortion in Microwave and Wireless Circuits, José Carlos Pedro and Nuno Borges Carvalho Introduction to Modeling HBTs, Matthias Rudolph Introduction to RF Design Using EM Simulators, Hiroaki Kogure, Yoshie Kogure, and James C. Rautio Introduction to RF and Microwave Passive Components, Richard Wallace and Krister Andreasson

Klystrons, Traveling Wave Tubes, Magnetrons, Crossed-Field Amplifiers, and Gyrotrons, A. S. Gilmour, Jr. Lumped Elements for RF and Microwave Circuits, Inder Bahl Lumped Element Quadrature Hybrids, David Andrews Microstrip Lines and Slotlines, Third Edition, Ramesh Garg, Inder Bahl, and Maurizio Bozzi Microwave Circuit Modeling Using Electromagnetic Field Simulation, Daniel G. Swanson, Jr. and Wolfgang J. R. Hoefer Microwave Component Mechanics, Harri Eskelinen and Pekka Eskelinen Microwave Differential Circuit Design Using Mixed-Mode S-Parameters, William R. Eisenstadt, Robert Stengel, and Bruce M. Thompson Microwave Engineers’ Handbook, Two Volumes, Theodore Saad, editor Microwave Filters, Impedance-Matching Networks, and Coupling Structures, George L. Matthaei, Leo Young, and E. M. T. Jones Microwave Materials and Fabrication Techniques, Second Edition, Thomas S. Laverghetta Microwave Materials for Wireless Applications, David B. Cruickshank Microwave Mixer Technology and Applications, Bert Henderson and Edmar Camargo Microwave Mixers, Second Edition, Stephen A. Maas Microwave Network Design Using the Scattering Matrix, Janusz A. Dobrowolski Microwave Radio Transmission Design Guide, Second Edition, Trevor Manning Microwave and RF Semiconductor Control Device Modeling, Robert H. Caverly

Microwave Transmission Line Circuits, William T. Joines, W. Devereux Palmer, and Jennifer T. Bernhard Microwaves and Wireless Simplified, Third Edition, Thomas S. Laverghetta Modern Microwave Circuits, Noyan Kinayman and M. I. Aksun Modern Microwave Measurements and Techniques, Second Edition, Thomas S. Laverghetta Neural Networks for RF and Microwave Design, Q. J. Zhang and K. C. Gupta Noise in Linear and Nonlinear Circuits, Stephen A. Maas Nonlinear Microwave and RF Circuits, Second Edition, Stephen A. Maas Q Factor Measurements Using MATLAB , Darko Kajfez QMATCH: Lumped-Element Impedance Matching, Software and User’s Guide, Pieter L. D. Abrie Passive RF Component Technology: Materials, Techniques, and Applications, Guoan Wang and Bo Pan, editors Practical Analog and Digital Filter Design, Les Thede Practical Microstrip Design and Applications, Günter Kompa Practical Microwave Circuits, Stephen Maas Practical RF Circuit Design for Modern Wireless Systems, Volume I: Passive Circuits and Systems, Les Besser and Rowan Gilmore Practical RF Circuit Design for Modern Wireless Systems, Volume II: Active Circuits and Systems, Rowan Gilmore and Les Besser Production Testing of RF and System-on-a-Chip Devices for Wireless Communications, Keith B. Schaub and Joe Kelly

Radio Frequency Integrated Circuit Design, Second Edition, John W. M. Rogers and Calvin Plett RF Bulk Acoustic Wave Filters for Communications, Ken-ya Hashimoto RF Design Guide: Systems, Circuits, and Equations, Peter Vizmuller RF Linear Accelerators for Medical and Industrial Applications, Samy Hanna RF Measurements of Die and Packages, Scott A. Wartenberg The RF and Microwave Circuit Design Handbook, Stephen A. Maas RF and Microwave Coupled-Line Circuits, Rajesh Mongia, Inder Bahl, and Prakash Bhartia RF and Microwave Oscillator Design, Michal Odyniec, editor RF Power Amplifiers for Wireless Communications, Second Edition, Steve C. Cripps RF Systems, Components, and Circuits Handbook, Ferril A. Losee The Six-Port Technique with Microwave and Wireless Applications, Fadhel M. Ghannouchi and Abbas Mohammadi Solid-State Microwave High-Power Amplifiers, Franco Sechi and Marina Bujatti Stability Analysis of Nonlinear Microwave Circuits, Almudena Suárez and Raymond Quéré Substrate Noise Coupling in Analog/RF Circuits, Stephane Bronckers, Geert Van der Plas, Gerd Vandersteen, and Yves Rolain System-in-Package RF Design and Applications, Michael P. Gaynor Terahertz Metrology, Mira Naftaly, editor TRAVIS 2.0: Transmission Line Visualization Software and User's Guide, Version 2.0, Robert G. Kaires and Barton T. Hickman

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