Compact Size Wireless Power Transfer Using Defected Ground Structures [1st ed.] 978-981-13-8046-4;978-981-13-8047-1
This book addresses the design challenges in near-field wireless power transfer (WPT) systems, such as high efficiency,
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Table of contents :
Front Matter ....Pages i-xxvi
Introduction (Sherif Hekal, Ahmed Allam, Adel B. Abdel-Rahman, Ramesh K. Pokharel)....Pages 1-8
Basics of Wireless Power Transfer (Sherif Hekal, Ahmed Allam, Adel B. Abdel-Rahman, Ramesh K. Pokharel)....Pages 9-31
Wireless Power Transfer Using DGSs (Sherif Hekal, Ahmed Allam, Adel B. Abdel-Rahman, Ramesh K. Pokharel)....Pages 33-72
Design Methods (Sherif Hekal, Ahmed Allam, Adel B. Abdel-Rahman, Ramesh K. Pokharel)....Pages 73-86
Future Directions (Sherif Hekal, Ahmed Allam, Adel B. Abdel-Rahman, Ramesh K. Pokharel)....Pages 87-91
Citation preview
Energy Systems in Electrical Engineering
Sherif Hekal Ahmed Allam Adel B. Abdel-Rahman Ramesh K. Pokharel
Compact Size Wireless Power Transfer Using Defected Ground Structures 123
Energy Systems in Electrical Engineering Series Editor Muhammad H. Rashid, Florida Polytechnic University, Lakeland, USA
More information about this series at http://www.springer.com/series/13509
Sherif Hekal Ahmed Allam Adel B. Abdel-Rahman Ramesh K. Pokharel •
•
•
Compact Size Wireless Power Transfer Using Defected Ground Structures
123
Sherif Hekal Faculty of Engineering at Shoubra Benha University Cairo, Egypt Adel B. Abdel-Rahman Egypt-Japan University of Science and Technology Alexandria, Egypt
Ahmed Allam School of Electronics, Communications, and Computer Engineering Egypt-Japan University of Science and Technology Alexandria, Egypt Ramesh K. Pokharel Kyushu University Fukuoka, Japan
ISSN 2199-8582 ISSN 2199-8590 (electronic) Energy Systems in Electrical Engineering ISBN 978-981-13-8046-4 ISBN 978-981-13-8047-1 (eBook) https://doi.org/10.1007/978-981-13-8047-1 © Springer Nature Singapore Pte Ltd. 2019 This work is subject to copyright. All rights are reserved by the Publisher, whether the whole or part of the material is concerned, specifically the rights of translation, reprinting, reuse of illustrations, recitation, broadcasting, reproduction on microfilms or in any other physical way, and transmission or information storage and retrieval, electronic adaptation, computer software, or by similar or dissimilar methodology now known or hereafter developed. The use of general descriptive names, registered names, trademarks, service marks, etc. in this publication does not imply, even in the absence of a specific statement, that such names are exempt from the relevant protective laws and regulations and therefore free for general use. The publisher, the authors and the editors are safe to assume that the advice and information in this book are believed to be true and accurate at the date of publication. Neither the publisher nor the authors or the editors give a warranty, expressed or implied, with respect to the material contained herein or for any errors or omissions that may have been made. The publisher remains neutral with regard to jurisdictional claims in published maps and institutional affiliations. This Springer imprint is published by the registered company Springer Nature Singapore Pte Ltd. The registered company address is: 152 Beach Road, #21-01/04 Gateway East, Singapore 189721, Singapore
To our families
Preface
The technology of wireless power transfer (WPT) has attracted considerable attention recently due to the increasing demand of wireless applications such as portable electronic devices, biomedical implants, and wireless buried sensors. WPT technology can be found also in contactless radio-frequency identification (RFID) and remote charging of electrical vehicles. It is beneficial to power electrical devices in cases where interconnecting wires are inconvenient, dangerous, or impossible. Wireless power transmission can be implemented by different methods that employ time-varying electric/magnetic (near-field) or electromagnetic (far-field) fields. Near-field (no-radiative) WPT systems have recently become popular as they are considered to be safe for health and provide high efficiency for short and mid-range applications. This book addresses the design challenges in the near-field WPT systems such as high efficiency, compact size, and long transmission range. Most of the near-field WPT systems depend upon magnetic resonant coupling (MRC) using 3D wire loops or helical antennas which are often bulky. This, in turn, poses technical difficulties for their use in small electronic devices and biomedical implants. Recently to get compact structures, the printed spiral coils (PSCs) have emerged as a candidate for low-profile WPT system. However, most of the MRC-WPT systems that use PSCs have limitations in the maximum achievable efficiency due to the feeding method. Inductive feeding constrains the geometric dimensions of the main transmitting (TX)/receiving (RX) resonators. The book presents new low-profile designs for the TX/RX structures using different shapes of defected ground structures (DGSs), such as H, semi-H, and spiral-strip DGS. The main advantage of the DGS WPT system is the feeding topology, where the power is transferred from/to the main TX/RX resonators by electrical coupling. We gain two advantages from this feeding topology: Firstly, the external quality factor can be easily optimized by an additional capacitor connected
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between the feed/load microstrip line and the DGS resonator. Secondly, no limitations exist on the optimization of the DGS resonator parameters to achieve maximum power transfer efficiency, unlike the case of conventional inductive feeding. Chapter 1 presents an overview of the technology of wireless power transfer (WPT) which can be utilized in many applications such as charging mobile devices and implanted biomedical devices as well as applications where interconnecting wires are inconvenient, dangerous, or impossible to implement, as in the cases of wireless buried sensors and sterilized rooms. Different implementation methods are discussed in brief by mentioning the current products that use these methods. The benefits of WPT are given by introducing the applications in our daily life and how they make our life hassle-free. This chapter also presents the challenges WPT systems face such as transfer efficiency, compact size, and transmission distance. After that, this chapter provides motivations to these vital topics that have attracted the attention of many researchers recently. Chapter 2 begins by discussing briefly the history of WPT over the last decades. The different techniques of transferring power wirelessly will be presented. Some commercial products and applications that use WPT are shown. A review of the current state of short-range WPT technology is given, and the trending research topics are noted. This chapter ends with a detailed explanation of the defected ground structures (DGSs) and their usage in microwave applications. Chapter 3 describes the principle of operation of using the defected ground structures (DGSs) as building blocks for WPT from circuit theory and microwave theories point of views. All design parameters and equivalent circuit elements that are associated with the proposed WPT systems will be defined. A more accurate circuit model is introduced to provide a better understanding of how the losses affect the efficiency of the WPT systems. This chapter also provides a detailed analysis of the design parameters that can realize the maximum achievable WPT efficiency. An asymmetric size WPT system with high efficiency is developed by the implementation of very compact size RX that can be embedded in the electronic consuming devices or biomedical implants to be charged wirelessly by larger-size TX. Chapter 4 reviews the different design methods that are currently being used in WPT systems. This chapter shows how the traditional design methods, which depend on iterative optimization, are not suitable due to a large amount of time needed to complete the design. This chapter provides an overview of the design methods that depend on circuit analysis using the impedance (Z-) parameters or the admittance (J-) inverters and discusses their principle of operation. A new design method is developed to represent the proposed WPT systems as a second-order Butterworth BPF using admittance inverters. A detailed mathematical analysis is performed to investigate the new design method and its effectiveness to reach the optimum design parameters and circuit elements accurately and fast. The novel design method is applicable to symmetric and asymmetric WPT systems. A design
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case is given detailing the design procedure and the experimental results to verify the new design method. Chapter 5 reviews the outcomes of the work presented in the book and concludes the book. Recommendations for future directions are also presented. Shoubra, Egypt Alexandria, Egypt Qena, Alexandria, Egypt Fukuoka, Japan
Sherif Hekal Ahmed Allam Adel B. Abdel-Rahman Ramesh K. Pokharel
Acknowledgements
Contributions from many colleagues in Kyushu University, Japan, and Egypt-Japan University of Science and Technology (E-JUST), Egypt, led to the completion of this book. Particularly, the authors would like to thank Professor Haruichi Kanaya, Associate Professor Hongting Jia, and Dr. Adel. Barakat. The authors would like to thank Professor Kuniaki Yoshitomi, Kyushu University, for his valuable cooperation and support in the designs fabrications and measurements. The authors would like to acknowledge that this work was supported in part by a Grant-in-Aid for Scientific Research (C) under Grant 16K06301, in part by the VLSI Design and Education Center (VDEC) at the University of Tokyo in collaboration with the Keysights Corporation, in part by the Egyptian Ministry of Higher Education and Scientific Research (MoHESR), Cairo, Egypt, and in part by Egypt-Japan University of Science and Technology (E-JUST), Alexandria, Egypt.
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2 Basics of Wireless Power Transfer . . . . . . . . . . . . . . . . 2.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2 History of Wireless Power Transfer . . . . . . . . . . . . . 2.3 Wireless Power Transfer Methods . . . . . . . . . . . . . . 2.3.1 Capacitive Coupling . . . . . . . . . . . . . . . . . . 2.3.2 Inductive Coupling . . . . . . . . . . . . . . . . . . . 2.3.3 Resonant Inductive Coupling . . . . . . . . . . . . 2.3.4 Strong Resonant Inductive Coupling . . . . . . . 2.3.5 Electromagnetic (EM) Radiation . . . . . . . . . . 2.4 Implementation of Near-Field WPT Systems . . . . . . 2.5 Implementation of Far-Field WPT Systems . . . . . . . 2.6 Frequency Selection . . . . . . . . . . . . . . . . . . . . . . . . 2.7 Overview of Commercial Products Supporting WPT References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
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3 Wireless Power Transfer Using DGSs . . . . . . . . . . . . . . 3.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.2 An Overview on Defected Ground Structures (DGS) 3.3 WPT Systems Using DGSs . . . . . . . . . . . . . . . . . . .
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1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.1 Overview of Wireless Power Transfer . . . . . 1.2 Applications of WPT . . . . . . . . . . . . . . . . . 1.3 Motivations of WPT . . . . . . . . . . . . . . . . . . 1.4 Challenges of WPT Systems Implementation 1.4.1 Non-radiative Systems . . . . . . . . . . . 1.4.2 Radiative Systems . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
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3.3.1 H-Shape DGS . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.3.2 Semi H-Shape . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.3.3 Spiral-Strips DGS . . . . . . . . . . . . . . . . . . . . . . . . 3.4 Design Method of the DGS-WPT Systems . . . . . . . . . . . . 3.5 Fabrication and Measurements . . . . . . . . . . . . . . . . . . . . . 3.6 Power Transmission Through the Human Body . . . . . . . . 3.7 Power Handling Capability of the Proposed WPT Systems References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
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4 Design Methods . . . . . . . . . . . . . . . . 4.1 Introduction . . . . . . . . . . . . . . . . 4.2 Design Method #1 . . . . . . . . . . . 4.3 Design Method #2 . . . . . . . . . . . 4.4 Verification of Design Method #2 4.4.1 Symmetric WPT System . 4.4.2 Asymmetric WPT System References . . . . . . . . . . . . . . . . . . . . .
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5 Future Directions . . . 5.1 Summary . . . . . . 5.2 Future Directions References . . . . . . . . .
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About the Authors
Sherif Hekal is currently an Assistant Professor at the Department of Electronics and Communications Engineering, Faculty of Engineering at Shoubra—Benha University, Cairo. He received his B.Sc. and M.Sc. degrees in Electrical Engineering from the same university in 2007 and 2012, respectively. He received his Ph.D. from the Egypt-Japan University of Science and Technology (E-JUST) in Electronics and Communications Engineering in 2016. As part of his Ph.D. program, he spent time at the Faculty of Information Science and Electrical Engineering, Kyushu University, Fukuoka, Japan. Dr. Hekal also worked as a communications engineer at Motorola Co. Ltd. and Nokia Siemens Networks in the field of 2G/3G RF optimization. His research interests include RF/microwave applications, antennas, wireless power transfer, and energy harvesting systems. Dr. Ahmed Allam is currently an Associate Professor at the Department of Electronics and Communications Engineering, Egypt-Japan University of Science and Technology, Alexandria, Egypt. He received his B.Sc. in Electrical Engineering from Alexandria University, Egypt, and his M.Eng. and Ph.D. from the University of Alberta, Canada. From April 1994 to January 1998, he worked as an instrument engineer with Schlumberger. From May 2000 to September 2001, he was with Murandi Communications Ltd., Calgary, Alberta, where he worked on RF transceivers design. From April 2007 to April 2008, he worked on RF CMOS transceivers design at Scanimetrics Inc., Edmonton, Alberta. His research interests include the design of RF circuits and systems. Adel B. Abdel-Rahman is currently a Professor at the Department of Electronics and Communications Engineering, Egypt-Japan University of Science and Technology, Alexandria, Egypt. He received his B.S. and M.S. in Electrical Engineering, Communication, and Electronics from Assiut University, Egypt, and his Dr.-Ing. degree in Communication Engineering from Otto von Guericke University, Germany in 2005. Since October 2006, he has been an Assistant Professor at the Electrical Engineering Department, South Valley University, Qena, Egypt. He has published more than 120 refereed journal and conference papers and xv
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has two patents. He was the Executive Director for Information and Communication Technology, South Valley University, from 2010–2012. Since October 2012, he joined the School of Electronics, Communications and Computer Engineering, Egypt-Japan University of Science and Technology (E-JUST), Alexandria, Egypt, and has been the Dean of the Faculty of Computers and Information, South Valley University from 2016–2018. His research interests include the design and analysis of antennas, filters, millimeter-wave devices, WPT, and metamaterials and their application in wireless communication, as well as optimization techniques with applications to microwave devices and antenna arrays. Ramesh K. Pokharel is a Professor in the Department of I&E Visionaries at Kyushu University. He received M.E. and PhD in Electrical Engineering from the University of Tokyo, Japan in 2000 and 2003, respectively. In April 2005, he joined the Graduate School of Information Science and Electrical Engineering, Kyushu University. He was the secretary of IEEE MTT-S Japan Society from Jan. 2012 to Dec. 2013 and the deputy-chair of the Education committee of IEEE-MTT-S Japan Society from Jan. 2014 to Dec. 2017 and has been serving as the chair of the same committee since 2017. His current research interests include low cost RFIC and analog circuits for microwave and millimeter wave wireless communications, and on-chip meta-materials in CMOS.
Abbreviations
2-D 3-D A4WP AC ADS BPF BSF CST DC DCP DGS EIRP EM FCC HFSS IMN IPT ISM JAXA LHCP LOS MCR MPE MRC NASA PCB PCE PMA PSC PV
Two Dimensional Three Dimensional Alliance for Wireless Power Alternating Current Advanced Design Systems Band Pass Filter Band Stop Filter Computer Simulation Technology Direct Current Dual Circularly Polarized Defected Ground Structure Effective Isotropic Radiated Power Electromagnetic Federal Communications Committee High Frequency Structures Simulator Impedance Matching Network Inductive Power Transfer Industrial Scientific Medical Japanese Space Agency Left hand circular polarization line-of-sight Magnetically Coupled Resonance Maximum Permissible Exposure Magnetic Resonant Coupling National Aeronautics and Space Administration Printed Circuit Board Power conversion efficiency Power Matters Alliance Printed Spiral Coil Photovoltaic
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RAMP Rectenna RF RFID RHCP RX SAE SAR SHARP SMD SPS TX UAV WBAN WPC WPT WRSN
Abbreviations
Raytheon Airborne Microwave Platform Rectifying Antenna Radio Frequency Radio Frequency Identification right hand circular polarization Receiver / Receiving Society of Automotive Engineers Specific Absorption Rate Stationary High-Altitude Relay Platform surface mounted Solar Power Satellite Transmitter / Transmitting Unmanned Aerial Vehicles wireless body area network Wireless Power Consortium Wireless Power Transfer wireless renewable sensor network
Nomenclature
Symbols L M k CP CS h D d u N
Self-inductance Mutual inductance Coupling coefficient Parallel capacitance Series capacitance Transmission distance Outer diameter of resonator Inner diameter of resonator Fill factor Number of turns of printed spiral
Units µ0 e0
Permeability of free space (4p 10−7 Henry/m) Permittivity of free space (8.85 10−12 Farad/m)
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List of Figures
Fig. 1.1 Fig. Fig. Fig. Fig.
1.2 1.3 1.4 2.1
Fig. 2.2 Fig. Fig. Fig. Fig. Fig.
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Fig. 2.8 Fig. 2.9 Fig. 2.10 Fig. 3.1
Fig. 3.2 Fig. 3.3 Fig. 3.4
Implementation of WPT systems using a near-field coupling, and b far-field radiations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Some of the most famous WPT applications [9] . . . . . . . . . . . Motivations of using wireless power transfer systems . . . . . . . Dream of wireless power society [10] . . . . . . . . . . . . . . . . . . . Tesla WPT experiments a Theory of operation. b Tesla’s lab and tower [7] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . First-ever laser-powered aircraft, designed and built by a team of NASA researchers [13] . . . . . . . . . . . . . . . . . . . WPT systems using capacitive coupling . . . . . . . . . . . . . . . . . WPT using inductive coupling . . . . . . . . . . . . . . . . . . . . . . . . WPT using resonant inductive coupling . . . . . . . . . . . . . . . . . WPT using strong resonant inductive coupling . . . . . . . . . . . . Schematic of the experimental setup of strongly coupled magnetic resonances implemented by the MIT team [26] . . . . Far-field wireless charging [30]. . . . . . . . . . . . . . . . . . . . . . . . ISM band . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Powercast wireless charging system a Transmitter. b Receiver. c Wireless rechargeable sensor system . . . . . . . . . Different shapes of DGSs. a Circular head dumbbell. b Triangular head dumbbell. c Square head dumbbell. d Spiral DGS. e Meander lines. f U-slot. g Square open-loop with a slot in middle section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Equivalent RLC circuit of DGS unit . . . . . . . . . . . . . . . . . . . . Conventional design and analysis method of DGS . . . . . . . . . Quasi-static modeling [38]. a Unit cell DGS. b Surface current on the ground plane . . . . . . . . . . . . . . . . . . . . . . . . . .
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List of Figures
Schematic equivalent current sheet (filament model) [38] . . . . Hekal et al. [1] a Proposed coupled H-shape DGS resonators WPT system. b H-shape DGS resonator as BSF at 300 MHz. c Simulated current distribution at phases (90°, and 270°) . . . Verification of the quasi-static model for H-shape DGS. a Equivalent circuit [1]. b Comparison between |S-parameters| of EM and circuit simulations [1] . . . . . . . . . . . . . . . . . . . . . . PCB layout of H-shape DGS resonator for the proposed WPT system [1] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . An equivalent circuit of the proposed WPT system using coupled H-shape DGS resonators [1] . . . . . . . . . . . . . . . . . . . Measurement setup of the fabricated WPT systems using H-shape DGS resonators [1] . . . . . . . . . . . . . . . . . . . . . . . . . . Comparison between the measured and the simulated |S-parameters| of the proposed WPT system using H-shape DGS resonators at 300 MHz and at transmission distance h = 13 mm [1] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3D schematic view of the proposed WPT system with the representation of the stub as a lumped capacitor [1] . . . . . . . . Representation of the stub as a lumped capacitor. a CST Simulated |S-parameters| [1]. b ADS Simulated |S-parameters| [1] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Misalignment studies for H-shape DGS-WPT system (20 20 mm2). a Schematic of misalignment due to horizontal shift and orientation [1]. b EM simulated WPT efficiency versus misalignment shifts [1]. c EM simulated WPT efficiency versus orientation angle [1] . . . . . . . . . . . . . . Schematic of semi H-shape DGS resonator [1] . . . . . . . . . . . . The proposed WPT system based on semi H-shape DGS resonators [1]. a PCB layout of a single resonator. b 3D schematic view. c equivalent circuit . . . . . . . . . . . . . . . . . . . . EM simulated magnetic field distribution of the coupled semi H-shape resonators WPT system at 300 MHz at plane X = 0 [1]. a U = 0°. b U = 90°. c U = 180°. d U = 270° . . . . . . . . Measurement setup of the fabricated WPT systems using semi H-shape DGS resonators [1] . . . . . . . . . . . . . . . . . . . . . . . . . . Comparison between the measured and the simulated |S-parameters| of the proposed WPT system using semi H-shape DGS resonators at 300 MHz and at a transmission distance h = 25 mm [1] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Measured WPT efficiency versus transmission distance (h) for H-shape and semi H-shape DGS resonators at 300 MHz [1] . Comparison between the measured power transfer efficiency versus misalignment [1] due a horizontal shift, and b different
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List of Figures
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orientation angles for H-shape (20 20 mm2) and semi H-shape (21 21 mm2) DGS-WPT systems . . . . . . . . . . . . . Comparison between three different shapes of DGS (H-shape, semi H-shape, spiral-strips DGS) [2]. a Current distribution. b Computed self-inductance . . . . . . . . . . . . . . . . . . . . . . . . . . Proposed spiral-strips DGS resonator as BSF [2]. a PCB layout. b, c EM and circuit simulated |S-parameters| embedded with equivalent circuit extracted by quasi-static modeling and analogy with one-pole Butterworth BSF response, respectively . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . a Model of the proposed spiral-strips DGS-WPT system, and b its EM simulated |S11 | & |S21 | . . . . . . . . . . . . . . . . . . . . a PCB layout of the realized TX/RX structure [2]. b The equivalent circuit of the proposed WPT system. c Analysis of the equivalent circuit using J-inverters [48] . . . . The proposed applications for wireless charging of mobile handsets [3] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Investigation of the computed U-factor of the coupled resonators versus the width Wt,i and separation si for the proposed symmetric WPT systems (50 50 mm2) [3] . . . . . Optimum WPT efficiency at different transmission distances for the symmetric WPT system (50 50 mm2) . . . . . . . . . . . Optimum WPT efficiency at different transmission distances for the asymmetric WPT system . . . . . . . . . . . . . . . . . . . . . . . Magnetic field distribution of the symmetric (50 50 mm2) WPT system at phases Ф = 0°, 45°, 90°, 135°, and 180° [3] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Magnetic field distribution of the asymmetric WPT system (TX 50 50 mm2 & RX 30 30 mm2) at phases Ф = 0°, 45°, 90°, 135°, and 180° [3] . . . . . . . . . . . . . . . . . . . . . . . . . . Measurement setup of the fabricated WPT systems (Symmetric 50 50 mm2, Symmetric 30 30 mm2, and Asymmetric TX = 50 50 mm2, RX = 30 30 mm2) [3] . . . . . . . . . . . . Comparison between the measured and the simulated |S-parameters|. a Symmetric 50 50 mm2 at h = 50 mm. b Symmetric 30 30 mm2 at h = 30 mm. c Asymmetric TX = 50 50 mm2, RX = 30 30 mm2 at h = 40 mm. d Measured WPT efficiency versus different transmission distances [3] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Fabricated designs and measurements of the proposed spiral-strips DGS-WPT system at f0 = 13.5 MHz [3] . . . . . . . Simulated and measured |S-parameters| for the proposed spiral-strips DGS-WPT system at h = 10 cm and f0 = 13.5 MHz [3] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
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List of Figures
Fig. 3.36 Fig. 3.37
Fig. Fig. Fig. Fig.
4.1 4.2 4.3 4.4
Fig. 4.5 Fig. 4.6 Fig. 4.7 Fig. 4.8 Fig. 4.9 Fig. 4.10
Fig. 5.1 Fig. 5.2
Fig. 5.3
Representing human life tissue effects on the efficiency of power transmission by insertion of a human hand . . . . . . . . . Comparison between the measured |S-parameters| of the WPT system in Fig. 3.36 with and without a human hand presence, where the transmission distance is 50 mm . . . . . . . . . . . . . . . A wireless power transfer systems using two coils [3] . . . . . . Equivalent circuits of admittance inverters in Fig. 4.1b [3] . . . Flowchart of design method #1. . . . . . . . . . . . . . . . . . . . . . . . a Proposed system block diagram. b Its equivalent circuit. c Equivalent circuit based on J-inverters . . . . . . . . . . . . . . . . . The analytical design procedure of the symmetric WPT system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The analytical design procedure of the asymmetric WPT system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The proposed WPT system a 3D view. b Planar view . . . . . . Comparison between the simulated |S-Parameters| of the symmetric WPT system using ADS and HFSS . . . . . . Measurement setup of the fabricated asymmetric WPT system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Comparison between the circuit (ADS), EM (HFSS) simulations, and the measured performance of the fabricated asymmetric WPT system. . . . . . . . . . . . . . . . . . . . . . . . . . . . . The proposed WPT system using dual-band adaptive near-field focusing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The difference in the received RF power density between the implementation of near-field focusing and far-field focusing using 8 8 array of single-band antennas [6] . . . . . Implementation of adaptive near-field focusing for a Single band. b Dual band . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
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Table 2.1 Table 2.2 Table 2.3 Table 2.4 Table 3.1 Table 3.2 Table 3.3
Table 3.4 Table 3.5
Table 3.6
Table 3.7
Table 3.8 Table 3.9
Table 4.1
Comparison between the different implementation methods of WPT systems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Dielectric characteristics of human body tissues at 50 MHz . Dielectric characteristics of human body tissues at 500 MHz . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Wireless power transfer standards . . . . . . . . . . . . . . . . . . . . . Design parameters of H-shape DGS resonator [1] . . . . . . . . . Design parameters and equivalent circuit elements of H-shape DGS resonator WPT system . . . . . . . . . . . . . . . . . . . . . . . . . Comparison between the optimum design parameters of H-shape and semi H-shape DGS resonators and their WPT efficiency at 300 MHz . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Comparative study with other compact WPT systems . . . . . . Optimized design parameters and equivalent circuit RLC values of the proposed WPT system using spiral-strips DGS [3] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Optimum design parameters ðCPTX ; CPRX ; CSTX ; and CSRX Þ to achieve gopt for the symmetric WPT system (50 50 mm2) at each transmission distance [3] . . . . . . . . . . . . . . . . . . . . . . Optimum design parameters (CPTX ; CPRX ; CSTX ; and CSRX ) to achieve gopt for the asymmetric WPT system at each transmission distance [3] . . . . . . . . . . . . . . . . . . . . . . . . . . . . Comparison of the proposed spiral-strips DGS-WPT system with the recent published planar WPT systems [3] . . . . . . . . Optimum design parameters and equivalent RLC values of the proposed spiral-strips DGS-WPT system (100 100 mm2) fabricated on FR4 substrate at f0 = 13.5 MHz . . . . . . . . . . . . Summary of the designed, simulated and optimized parameters and performance of the symmetric WPT system .
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Table 4.2
Table 4.3
List of Tables
Summary of the designed, simulated and optimized parameters and performance of the asymmetric WPT system. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Comparison between the different design methods for resonant inductive WPT systems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
84 85
Chapter 1
Introduction
1.1 Overview of Wireless Power Transfer Wireless power transmission (WPT) is defined as the transmission of electrical power from a power transmitter to one or more electrical loads, such as a network of wireless sensors or electronic devices, without the use of interconnecting cables or conductive wires. The wireless sensor networks hidden in bridges or buildings to track the effect of heavy loads and environmental changes on the structure strength [1–4], are an example of the importance of WPT. WPT is crucial for implanted biomedical devices in order to avoid performing surgical procedures to replace the battery [5]. WPT systems can be implemented by different power transmission techniques that employ time-varying magnetic, electric, or electromagnetic fields. In these systems, the power transmitter (resonator or antenna) is connected to a power source which transfers the field energy across an intermediate space to one or more receivers, where it is converted back to an electrical current and then exploited [6]. Wireless power transmission first emerged in the experiments of Nikola Tesla in the 1890s, wishing to transfer hundreds of Kilovolts of electricity through the air from Niagara Falls, and then feed it out to cities, factories, and private houses from the top of his tower without wires [7]. However, this technology has been made practical with touchable benefits to real-world applications in the past three decades due to advances in technology and better implementations of power transfer techniques. The implementation techniques of WPT can be divided into two main categories: non-radiative and radiative techniques. In the non-radiative power transmission, or what is the called near-field coupling techniques, power is transmitted by magnetic fields using magnetic inductive or resonant inductive coupling between 3D wire loops, helical antennas or printed spirals. Applications of this WPT technique include RFID tags, and chargers for implanted biomedical devices like pacemakers. Magnetic coupling is also utilized in inductive powering or charging of electric vehicles like cars, trains or buses [8]. Power may also be transferred by electric fields
© Springer Nature Singapore Pte Ltd. 2019 S. Hekal et al., Compact Size Wireless Power Transfer Using Defected Ground Structures, Energy Systems in Electrical Engineering, https://doi.org/10.1007/978-981-13-8047-1_1
1
2
1 Introduction
Fig. 1.1 Implementation of WPT systems using a near-field coupling, and b far-field radiations
using capacitive coupling between metal plates, which will be discussed in details in Chap. 2. Figure 1.1a illustrates the implementation of short-range WPT system using nearfield coupling, where the AC power is converted to DC in an AC/DC rectifier block, or alternatively using a DC supply. A high-efficiency switching amplifier converts the DC voltage into an RF signal that is used to drive the transmitting (TX) resonator. The impedance matching network (IMN) is used to efficiently couple the amplifier output to the TX resonator. The magnetic or electric field provided by the transmitting (TX) resonator is coupled to the receiving (RX) resonator, exciting the resonator which directs the energy efficiently to the RF/DC rectifier through an IMN. Finally, the rectified energy is coupled out to directly power a load or charge a battery. In radiative techniques, also called far-field techniques, power is transferred by electromagnetic radiation, like RF and microwave signals or laser beams. In these WPT systems, power is generated using RF/Microwave power sources like magnetron or klystron, and this generated power is passed to the TX antenna. As shown in Fig. 1.1b, the receiving unit consists of the RX antenna which captures the received RF/microwave power. The IMN matches the impedance of the antenna with that of the rectifier to transfer maximum captured power from the antenna to the rectifier circuit. This RX antenna along with the rectifier is known as the rectenna. Because of the safety considerations required for humans, the received power is in the range of mW or µW, so storage capacitors are needed to accumulate the received power.
1.1 Overview of Wireless Power Transfer
3
These techniques are used to transfer power for longer distances and can be found in some applications like solar power satellites, wireless powered drone aircraft, and wireless charging of portable handsets from cellular networks, or WiFi access points.
1.2 Applications of WPT The technology of Wireless Power Transfer (WPT) has attracted considerable attention recently due to its potential numerous applications such as wireless charging of portable electronic devices, biomedical implants, and wireless hidden sensors. This technology can support a wide range of applications, from low-power toothbrush to high-power electric vehicles because of its convenience. Nowadays, wireless charging is rapidly progressing from ideas and principles to standard features on commercial products, especially mobile phones and portable smart devices. It can be found also in contactless radio-frequency identification (RFID) for security applications and transportations, remote charging and powering of electrical vehicles. Figure 1.2 shows some of the most popular and vital applications of WPT that are found nowadays in our daily life or going to be current in the near future. With the growing demand for the variety and the large number of portable electronic devices (laptop, tablet, mobile phone, etc.), WPT offers feasibility of charging batteries without the annoyance of heavyweight cables, and the inconvenience of “plugging in”. Another usage of wireless power transfer can be found in biomedical applications, mainly medical implants. These fast-emerging applications can result in a major quality of life improvements and have significant life-extending consequences. Wireless power transmission can also be utilized in safety-critical
Sensor reader
Reading Display
Wall RFIDs
Wireless Buried Sensors
Biomedical Implants like Pacemakers Charging Electrical Vehicles
Fig. 1.2 Some of the most famous WPT applications [9]
Charging Portable handsets
4
1 Introduction
Complicated powering system Free and easy WPT
WIFI signal
Fig. 1.3 Motivations of using wireless power transfer systems
environments such as explosive or corrosive atmospheres, or any location where there is a safety risk when an electrical connection is made or broken.
1.3 Motivations of WPT The technology of WPT has replaced the complicated powering systems to a free wireless powering system as shown in Fig. 1.3. One can imagine how our life will be hassle free to power electrical devices, electronic consuming devices, and many other applications like wireless buried sensors or biomedical implants without interconnecting wires. Many organizations have adopted the idea of generalization of wireless power transfer in many applications, and also benefited from renewable energy sources to provide power through WPT systems like solar power satellites. Figure 1.4 depicts the dream of wireless power society supported by WiPoT (one of the organizations that are developing the standards of WPT systems) to use WPT systems in our life. This new system can support power in emergency cases where there are no available power sources. Solar power satellites will collect solar energy using photovoltaic cells. The collected energy will then be transferred to earth using microwave beams to power millions of devices on earth. As shown in Fig. 1.4, electrical vehicles can be charged wirelessly when they are running on roads using fixed power stations. The technology of WPT outperforms the wired power transfer systems in eliminating costs related to maintaining direct conductive cables, convenience for charging portable electronic devices, and safer power transfer to devices that require steriliza-
1.3 Motivations of WPT
5
Fig. 1.4 Dream of wireless power society [10]
tion. WPT can also provide robust power delivery to rotating and mobile industrial equipment like movable robots and robotic arms. It is very effective to transfer power to unreachable hidden sensors and critical systems working in dirty, wet, or disinfected environments.
1.4 Challenges of WPT Systems Implementation In spite of the advantages and the beneficial applications of wireless power transfer, there are some challenges that impede the implementation of WPT systems. Some of these challenges are market driven, while others are related to the methods of design and implementation of the WPT systems. Today the mobile gadget market is leading the development of WPT, thus setting many of its requirements and challenges.
1.4.1 Non-radiative Systems These requirements include high efficiency, particularly for the receiving devices due to the limited available power budgets, low physical profile, and robustness to all operating conditions. From the user point of view, WPT systems should not
6
1 Introduction
have limitations on the numbers, or sizes of the devices to be powered. The main requirements of WPT systems are summarized below: • High efficiency—limited power dissipation with high WPT efficiency of 80–90%. • Low profile—small design area (≤30 × 30 mm2 ) especially for the receiving (RX) resonator which is desirable for the portable market. • Robust to dynamic operating conditions—the WPT efficiency should keep its value within an acceptable range due to misalignment between the TX and RX resonators. • Defined response to foreign metal objects—recognition of the foreign objects around the system is of significant importance because of their ability to absorb energy from the wireless power supply field in the form of heat (parasitic heating) and possibly becoming a threat. • Compliance to commercial standards—e.g., Qi standard developed by Wireless Power Consortium (WPC), Power Matters Alliance (PMA), and Alliance for Wireless Power (A4WP).
1.4.2 Radiative Systems Working strictly with the agencies of national safety and health, the Federal Communications Committee (FCC) has approved restrictions for safe exposure to RF signals [11]. These restrictions are given in terms of (1) Maximum Transmit Output Power: Numerous FCC instructions govern the transmit power permitted in the Industrial, Scientific, and Medical (ISM) bands. A brief of these rules is listed below: (a) Maximum transmitter output power, fed into the antenna, is 30 dBm (1 W). (b) Maximum Effective Isotropic Radiated Power (EIRP) is 36 dBm (4 W). The EIRP is given by [E I R P]dBm = [Pt ]dBm + [G t ]dB − [L imp ]dB
(1.1)
where Pt is the output power level of the transmitter, G t is the gain of the transmitter antenna, L imp is the impedance mismatch of the Tx antenna. (2) Maximum Permissible Exposure (MPE) Limit: The equivalent FCC standard for uncontrollable exposure to an intentional transmitter operating at a frequency of 2.4 GHz is 10 W/m2 MPE [11]. [W f ]W/m2 =
EI RP W ≤ 10 2 4 × π × d2 m
(1.2)
1.4 Challenges of WPT Systems Implementation
7
(3) Specific Absorption Rate (SAR) Limit: The Specific Absorption Rate (SAR) is a measure of the amount of RF energy absorbed by the body when exposed to RF EM field. For example, the SAR limit as standardized by FCC for cell phones is 1.6 W/kg [12]. FCC requires mobile phones manufacturers to ensure that their handsets fulfill this limit for safe exposure. Any mobile phone at, or lower than this SAR level is a “safe” phone, as measured by this standard. SAR is a function of the induced E-field from the radiated energy (V/m), the electrical conductivity (S/m), and the mass density of the tissue (kg/m3 ). The SAR is calculated by averaging over a specific volume (typically a 1 g or 10-gram area) [13]: SAR = sample
σ (r )|E(r )|2 dr ρ(r )
(1.3)
(4) Focalized Temperature Limit: The absorbed power from an electromagnetic field can increase the temperature of body tissues. It is necessary that the temperature of the tissues nearby the implanted device do not increase more than 1–2 °C. To follow FCC regulations, the transmitted power level should be assigned carefully, and the permissible received power (Pr ) can be calculated using Friis radio link formula: Pr =
G t G r λ20 (1 − |S11 |2 )(1 − |S22 |2 )e P × Pt (4π d)2
(1.4)
where Pt is the transmitter output power, G t is the TX antenna gain, G r is the RX antenna gain, d is the distance between Tx and Rx, e p is the polarization mismatching between antennas. In real radio systems, there are many factors that can reduce the value of the received power given by Friis formula.
References 1. K. Shams, M. Ali, Wireless power transmission to a buried sensor in concrete. IEEE Sens. J. 7, 1573–1577 (2007) 2. S. Jiang, S.V. Georgakopoulos, Optimum power transmission of wireless sensors embedded in concrete, in 2010 IEEE International Conference on RFID (2010), pp. 237–244 3. O. Jonah, S.V. Georgakopoulos, Wireless power transmission to sensors embedded in concrete via magnetic resonance, in 2011 IEEE 12th Annual Wireless and Microwave Technology Conference (WAMICON) (2011), pp. 1–6 4. O. Jonah, S.V. Georgakopoulos, Wireless power transfer in concrete via strongly coupled magnetic resonance. IEEE Trans. Antennas Propag. 61(3), 1378–1384 (2013)
8
1 Introduction
5. B.M. Badr, R. Somogyi-Gsizmazia, N. Dechev, K.R. Delaney, Power transfer via magnetic resonant coupling for implantable mice telemetry device, in 2014 IEEE Wireless Power Transfer Conference (WPTC) (2014), pp. 259–264 6. J.I. Agbinya, Wireless Power Transfer (River Publishers, 2012) 7. http://www.teslasociety.com/tesla_tower.htm. Accessed 31 May 2016 8. T. Imura, H. Okabe, Y. Hori, Basic experimental study on helical antennas of wireless power transfer for electric vehicles by using magnetic resonant couplings, in Vehicle Power and Propulsion Conference, 2009 VPPC’09. (IEEE, 2009), pp. 936–940 9. J.I. Agbinya, Wireless Power Transfer, vol. 45 (River Publishers, 2015) 10. Wireless Power Transfer for Practical Application, http://www.wipot.jp/english/ 11. C. Liu, Y.-X. Guo, H. Sun, S. Xiao, Design and safety considerations of an implantable Rectenna for Far-Field wireless power transfer. IEEE Trans. Antennas Propag. 62(11), 5798–5806 (2014) 12. https://www.fcc.gov/general/specific-absorption-rate-sar-cellular-telephones 13. http://www.antenna-theory.com/definitions/sar.php
Chapter 2
Basics of Wireless Power Transfer
2.1 Introduction The usage of conductive cables is always preferred as the first choice to power electrical loads. It is appropriate and efficient especially for most of the stationary loads that support our daily applications, whether in our homes or in industry. On the other hand, with the rapid evolution of technology, products are becoming smaller and portable so depending on wired connection to gain energy may not be an applied solution for many applications. Therefore, having a direct cable connection may limit the movement freedom and in many cases may not be a safe choice [1]. In wireless power transmission (WPT), instead of using conductive transmission media, electrical power is converted to another form (electrical field, magnetic field, or electromagnetic radiation) that can be transmitted through a certain media (air, walls, tables, human body, etc.) without wires. A simple model of wireless energy transfer can be found in the use of radio waves to transfer information (voice, video, or data), which is possible now in broadcasting and cellular networks. The field of wireless power transfer is motivating, and the future of this field seems sunny. There is a growing interest nowadays in WPT applications especially in the field of wireless sensor networks [2], research in biology (e.g., animals and insects surveillance [3]), and implanted biomedical devices [4, 5]. There has been also an increasing demand from industry which was adopted by a number of organizations such as WiTricity, uBeam, Ossia, Artemis, Energous, and Proxi [6].
2.2 History of Wireless Power Transfer Nikola Tesla was the first person to demonstrate the concept of WPT in 1891 [7], Tesla illuminated fluorescent lamps 25 miles from the power source without wires. He achieved this result by means of static electric fields of the high frequency generated © Springer Nature Singapore Pte Ltd. 2019 S. Hekal et al., Compact Size Wireless Power Transfer Using Defected Ground Structures, Energy Systems in Electrical Engineering, https://doi.org/10.1007/978-981-13-8047-1_2
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2 Basics of Wireless Power Transfer
Fig. 2.1 Tesla WPT experiments a Theory of operation. b Tesla’s lab and tower [7]
from lightning sparks as illustrated in Fig. 2.1. He led further experiments in the Wardenclyffe Tower until it was destroyed in 1917 [7]. Despite the innovation of Tesla’s idea and his personal efforts to globalize WPT, he soon ran out of funding because it was cheaper to use copper than to build the system necessary to transmit power through radio waves. Research in WPT continued mainly through the development of telecommunications during the first few decades of the twentieth century. The next most important development in WPT was the utilization of microwaves to power distant objects by William C. Brown in the 1960s [8–11]. Brown invented the Rectenna (rectifying antenna) which converts microwaves to DC current, and in 1964 [8] he proved its ability to transfer power wirelessly by flying a small helicopter 60 feet away relying entirely on microwave power, as a part of a shared project with NASA [13 and 14]. The attention to transporting power wirelessly increased after the energy catastrophe in 1973. NASA proposed a novel concept for finding renewable energy resources and aimed to collect solar energy by satellites, convert it to microwaves, and beam it to earth, where it is reconverted into useful power [11]. Another line of research focused on powering unmanned aerial vehicles (UAV) wirelessly. In 1987, Canada’s Communication Centre successfully designed a small aircraft that could fly at a height of 21 km powered wirelessly by focusing a 2.45 GHz microwave beam to an
2.2 History of Wireless Power Transfer
11
Fig. 2.2 First-ever laser-powered aircraft, designed and built by a team of NASA researchers [13]
onboard microwave antenna from earth. This project was named SHARP (Stationary High Altitude Relay Platform) [12]. The use of laser for wireless power transmission has also been proposed since its invention in the 1950s. An electric current can be generated using a high-power laser beam focused on a photovoltaic (VP) cell. However, due to the low efficiencies of a photovoltaic cell, the use of laser to transfer energy wirelessly was not favored. Thus, microwave WPT was the pioneer for transferring power wirelessly over several kilometers during the second half of the twentieth century. The concept of using laser rose once more in the late twentieth century when the Japanese Space Agency (JAXA) led research programs in the field of space power stations to transfer Giga Watts of energy via laser beams using huge mirrors positioned in different orbits around the Earth. In 2003, a NASA research team designed a laser-powered airplane [13], where the full body of the model plane was covered with photovoltaic cells to convert power back from a ground-based infrared laser, shown in Fig. 2.2. Besides the mentioned far-field (radiative) WPT techniques, there were also substantial efforts done to develop the near-field techniques, which are built on either inductive or capacitive coupling. Inductive power transfer between neighboring coils appeared with the development of the electrical transformer in the 1800s. In 1892, M. Hutin and M. Leblanc patented a wireless method of powering railroad trains using inductively coupled resonant coils [14]. Inductive charging is used in small electronic appliances like the electric toothbrushes and the electric shavers. The first passive radio-frequency identification (RFID) devices were invented in the 1970s [15, 16], and applied widely in the industry (e.g., contactless smartcards) in the 1990s. In 2007, M. Soljacic with his team at MIT used strongly coupled resonant circuits made of a 60 cm resonant coil to transfer 60 W of power over a distance of 2 m achieving an efficiency of 40%.
12
2 Basics of Wireless Power Transfer
A variety of mobile devices such as smartphones, tablets, and laptops has led to launching Wireless Power Consortium in 2008. The target of this association is to establish international standards for wireless charging, which has been standardized in the form of the Qi standard [17]. The Qi standard has been launched in August 2009 to enable charging and powering of portable electronic devices of up to 5 W over distances of 4 cm.
2.3 Wireless Power Transfer Methods Several methods have been developed for transferring energy wirelessly between a power source and an electrical load. Any WPT system consists of two subsystems, which are the transmitter (TX) and the receiver (RX). The transmitter is located where energy from a power source is to be transmitted. The receiver is located where the electrical load is to be powered. There are two main implementation techniques for WPT depending on the required distance (h) between the TX and the RX. Near-field coupling (non-radiative) techniques: The near-field region is defined as the area within (λ/2π) of the antenna, where λ is the wavelength of the electrical signal. In this region, the electric (E) and magnetic (H) fields are separate (phases of E and H are near quadrature) [18], so the electric and the magnetic fields can exist independently of each other, and one type of field can dominate. These fields are non-radiative but the power can be transferred to another near antenna/resonator by capacitive or inductive coupling via electric or magnetic fields, respectively. The fields in this region decay with the distance by (1/h3 ), so the power transfer efficiency decays by (1/h6 ) [18]. The near-field techniques are a better choice for short-range and mid-range applications, because they provide safety and high-power transfer efficiency especially at lower frequencies. Other techniques, such as resonant inductive coupling and strong resonant coupling, have emerged to increase the coupling between the resonators, allowing higher efficiency at greater distances (mid-range) [19, 20]. Far-field (radiative) techniques: The far-field region is the area away one wavelength from the antenna, where the electric and magnetic fields are in phase and propagate as an electromagnetic (EM) radiative wave. The fields in this region decay with the distance by (1/h), so the power transfer efficiency decays by (1/h2 ) [18]. Far-field techniques are preferred in long-range applications, but they are limited due to lower power transfer efficiency and undefined safety. Far-field techniques using radio waves, microwaves, or laser beams are used in applications such as very-low-power devices or sensor networks, where efficiency is not of paramount importance. Also, they could be implemented
2.3 Wireless Power Transfer Methods
13
in high-power space, military, or industrial applications that are not sensitive to cost such as solar-powered satellites and drone aircraft. Near-field coupling methods will be discussed next because they are the most popular for applications such as RFIDs and wireless charging of portable electronic devices, biomedical implants, or wireless buried sensors.
2.3.1 Capacitive Coupling In capacitive coupling, power is transferred through the electric field between two parallel metal plates which induce what is called the displacement current. A capacitivecoupled system can be constructed as a pair of capacitors each consisting of two parallel plates separated by the transmission distance [21]. The transmitter is connected to the bottom plates of each capacitor and the receiver is connected to the top plates. If an AC voltage is applied to the TX plates, then a time-varying electric field will be induced across the two plates of both capacitors and a varying electric field will produce a displacement current. This current allows energy to be transferred across the medium between the plates of the capacitors. If an electrical load is connected between the top plates of the capacitors, then the time-varying displacement current will cause electric charges to be continuously moved forward and backward between the plates. Consequently, an electric current is formed in the receiver. Figure 2.3 illustrates the process of wireless power transfer using capacitive coupling [22, 23]. Capacitive coupling was one of the earliest approaches used to transfer electrical energy wirelessly as confirmed by Tesla. However, it was not an appropriate method to transfer power because of the requirement of high voltages (several K volts) and the need for large plates for long distances [21].
Power source
Oscillator
RecƟfier
TX
Fig. 2.3 WPT systems using capacitive coupling
RX
Load
14
2 Basics of Wireless Power Transfer
Advantages: • Simplicity; • Very short range (less than a few centimeters) with very high efficiency. Disadvantages: • The requirement of high voltages; • Efficient charging only on short distance.
2.3.2 Inductive Coupling Inductive coupling transfers energy between TX and RX through two coils located close to each other. When an alternating (AC) current passes in the transmitting (TX) coil, it generates a time-varying magnetic field which crosses the receiving (RX) one and induces a current. This system was the base for the first global wireless charging standard (Qi) produced by Wireless Power Consortium [11]. The first version of the Qi standard (for low-power inductive charging issued in 2009. The standard stipulates a technique for wireless power transfer over a small distance. The disadvantage of this technique is that the TX and RX coils should be located very near to each other and specifically aligned. That is the reason this technique is not the most suitable for Wireless Power Transfer Networks applications (which need distances longer than a few centimeters). Figure 2.4 illustrates the process of wireless power transfer using conventional inductive coupling. Inductive coupling has succeeded to be deployed in consumer electronics (e.g., mobile phones and toothbrushes) and electric vehicles. This technique will not be probably be used in other applications, as the power transfer using this technique is efficient only in close proximity.
RX RecƟfier Power source
TX Oscillator
Fig. 2.4 WPT using inductive coupling
Load
2.3 Wireless Power Transfer Methods
15
Advantages: • Simplicity; • Very short range (few centimeters) with high efficiency. Disadvantages: • Requires accurate orientation of the TX and RX coils; • Efficient charging only on short distance.
2.3.3 Resonant Inductive Coupling Resonant inductive coupling is used for mid-range applications. Inductive coupling is the most popular technique for high efficient WPT systems and is usually applied at lower frequencies for very short ranges. At higher frequencies, the resonant type becomes a good choice because resonant circuits focus power at specific frequency so that the efficiency of power transfer can be enhanced [5, 24, 25]. Figure 2.5 illustrates the process of wireless power transfer using resonant inductive coupling.
2.3.4 Strong Resonant Inductive Coupling Magnetic resonant coupling is a technique recognized by Kurs et al. [26] in 2007. This technique employs strongly coupled resonant coils to achieve very high efficiency over transmission distances up to 4 times the diameter of the coil. The authors of [26] were able to present an efficient (40% efficiency) power transfer to light a lamp within 2 m. The main idea behind this technique is to use two intermediate coils with a high quality factor (Q). In order to make those two coils resonant, capacitors are added to the system, creating resonant RC circuits. Figure 2.6 illustrates the process of wireless power transfer using strong resonant inductive coupling.
RX RecƟfier Power source
TX Oscillator
Fig. 2.5 WPT using resonant inductive coupling
Load
16
2 Basics of Wireless Power Transfer
RX RecƟfier
Power source
Load
TX Oscillator
Fig. 2.6 WPT using strong resonant inductive coupling
The developments presented in [26] were the basis of creating a WPT company |WiTricity| that was able to commercialize this technology. Advantages: • High efficiency over transmission distances equal to several times the coil diameter; • Unresponsive to weather conditions (unlike EM radiation); Disadvantages: • Requires alignment of the TX and RX coils; • Efficient charging only within a few meters.
2.3.5 Electromagnetic (EM) Radiation The far-field radiative techniques are able to transfer energy by the transmission of electromagnetic waves from a TX antenna connected to the power source to the RX antenna or what is called rectenna. These techniques using EM radiation realize longer transmission distances, where the distance is much greater than the diameter of the TX/RX device(s). The EM waves that are most frequently used in Wireless Power Transfer are microwaves (MHz–GHz), and visible light (frequencies approximately in THz). EM radiation is widely used for data transfer, but transferring power suffers from efficiency problems due to a very high attenuation of EM waves in space, where
2.3 Wireless Power Transfer Methods
17
the loss is usually inversely proportional to the square of the distance between TX and RX antennas [18]. Energy is transferred through the electric field of an electromagnetic wave, which is radiative. Because of the safety matters raised by RF exposures, radiative wireless charging generally operates in a low-power region [27]. For example, omnidirectional RF radiation is suitable only for sensor node applications with up to 10 mW power consumption [28, 29]. Systems based on light are mainly used directionally. Advantages: • Small receiver size; • Efficient power transfer over long distances specifically using directive radiations (microwave/laser). Disadvantages: • High losses of EM waves in the atmosphere; • For directional devices—requires complex tracking mechanisms and line of sight (LOS).
2.4 Implementation of Near-Field WPT Systems This section discusses the recent developments and contributions in the non-radiative WPT systems especially the inductive coupling, which is most popular due to their simplicity, high efficiency, and low cost. Inductive coupling systems can be mainly classified into two categories: magnetic induction and resonant inductive coupling. These systems, employing magnetic induction, transfer energy from a TX winding coil to an RX winding coil using an alternating magnetic field [79, 80]. This technology is usually suitable for short-range WPTs. On the other hand, mid-range WPT systems are usually implemented using electromagnetic resonant coupling technique. This technique uses resonance on both sending and receiving terminals and has received a great amount of attention in recent years after being first presented in 2007 by Kurs et al. [26]. This technique, developed by M. Soljacic and his team, utilized strongly coupled resonant circuits made of identical self-resonant helical coils of high quality factor to transfer power over a distance of 2 m achieving around 40% efficiency as illustrated in Fig. 2.7 [26]. Figure 2.7 shows the experimental setup of strongly coupled magnetic resonances implemented by the MIT team [26]. Because of the simplicity and low cost of implementation, most of the existing WPT applications have embraced inductive coupling, or what is called Inductive Power Transfer (IPT). IPT is capable of supporting high-power transfer above kilowatt level, so it is commonly used in industrial automation applications [81], automated underwater vehicles [82], electrical vehicles (EV) [83, 84], and high-speed trains [85, 86].
18
2 Basics of Wireless Power Transfer
Fig. 2.7 Schematic of the experimental setup of strongly coupled magnetic resonances implemented by the MIT team [26]
The medium-power near-field charging (operating from several watts to tens of watts) are used for charging of medical implants and in daily applications. Wireless charging based on magnetic resonance coupling exhibits more powerful penetration ability especially for biomedical implants [4, 5, 80, 87]. The authors in [88] have achieved, with a 3 cm TX coil and 2 cm RX coil, above 60% charging efficiency over 2 cm distance. The daily usage powering can be found in applications like portable electronic devices and everyday devices (inductive toothbrush [89], TV [90], and lighting [91]). Regarding portable devices, the authors in [31] have reported the available different standard compliant wireless chargers, such as Energizer Qi charger, Verizon Qi charging pad, RAV Power’s Qi charger, ZENS Qi charging pad, Airpulse charging pad, and Duracell Powermat, that have been commercialized to supply energy to portable devices [31]. Most of the WPT designs developed in the past decades were based on large wire wound coils, and several patterns of resonators that have been proposed for magnetically coupled resonance (MCR) WPT systems, such as helical and spiral resonators [92–94]. The disadvantages of these MCR-WPT structures are their threedimensional (3D) shape and their geometry (bulky) which are difficult to fabricate, and not suitable for charging small electronic devices like portable mobile phones or biomedical implants. In January 2014, M. Falavarjani and others reported a planar WPT system with square spiral resonators printed on the top and bottom layer of a single FR4 substrate, such that the printed spiral resonator is driven by the magnetic field of the coupling (feeding/load) loop [95]. The inductive coupled (feeding/load) loop is used to realize the input/output matching network. The proposed WPT system in [95] achieved a measured WPT efficiency of 43% using TX/RX size of 12 × 12 cm2 at a transmission distance of 10 cm.
2.4 Implementation of Near-Field WPT Systems
19
In August 2014, F. Jolani presented a magnetically coupled resonance WPT (MCR-WPT) system using printed spiral coil (PSC) resonators [96], where the TX/RX resonant coil and the feeding/load loop are printed on the same side of an FR4 substrate to create a fully planar TX/RX coil set. The proposed WPT design in [96] with a two-turn PSC resonator and optimized geometric parameters achieved a maximum WPT efficiency of 77.27% at a transmission distance of 10 cm; with the parallel paths created with auxiliary strips, the effective series resistance (ESR) of the TX/RX PSC is decreased, the quality factor of the PSC resonators is improved, and the maximum WPT efficiency increased to 81.68% [96]. In January 2015, F. Jolani presented another highly efficient planar MCR-WPT system using two additional resonators for the TX and RX sides to increase the quality factor of the resonators [97]. Two substrates are stacked, where two PSCs are printed at the opposite sides of one substrate, and the third PSC is printed on the bottom surface of the other substrate; and three PSC resonators are connected to each other at the ends through vias [97]. To further improve the transmission efficiency of the system, the outer edges of the multilayer resonator are connected together using shorting walls. The proposed planar MCR-WPT system in [97] with multilayer resonators offers higher transfer efficiency than the conventional planar MCR-WPT system, where a WPT efficiency of 84% has been achieved using the stacked substrates. By the end of 2014, a breakthrough technology, named magnetic MIMO (MagMIMO), has been introduced to perform multi-antenna beamforming based on magnetic waves [31]. This technology has opened an area for the magnetic-field beamforming research [31]. The authors in [98] presented Magnetic MIMO, a system that can charge portable devices at distances up to 40 cm irrespective of the device’s position and orientation, while commercial chargers are limited to less than 10 cm [98]. Magnetic MIMO achieved this performance by adapting the wireless concept of MIMO beamforming to concentrate the resonating magnetic field on the receiver device to maximize the power transfer efficiency. Magnetic MIMO does not require any modification to the phone and hence can be used with today’s phones by including the small receiver coil (and circuit) in an outer cover attached to the phone.
2.5 Implementation of Far-Field WPT Systems Figure 2.8 shows the building of an RF/microwave power transmission system. The power transmission starts with the AC-to-DC conversion, followed by a magnetron for DC-to-RF conversion at the transmitting side. After broadcasting through the air, the RF/microwave signal captured by the receiving rectenna is rectified into electricity again. The main factors that affect the efficiency of RF-to-DC rectification are the captured power density, the matching between the receiving antenna and the voltage multiplier, and the power efficiency of the voltage multiplier.
20
2 Basics of Wireless Power Transfer
Fig. 2.8 Far-field wireless charging [30]
Nikola Tesla was the first to perform experiments of wireless power transfer based on microwave technology. He focused on long-range wireless power transfer and realized the transfer of microwave signals over a distance of about 48 km in 1896 [32]. Another key step forward was accomplished in 1899 to transmit 100 MV of high-frequency electric power over a distance of 40 m to light 200 lamps [32]. However, the technology that Tesla applied was put off because of the potentially harmful effects of emitting high voltages in electric arcs. In 1964, W. Brown, the chief pioneer of practical wireless charging, used a rectenna to convert microwaves to electricity. Brown demonstrated the practicality of microwave power transfer by powering a model helicopter [8]. Furthermore, the solar-powered satellite (SPS), that was presented in 1968, is another motivating force for long-range microwave power transfer [34]. The idea is to position huge solar cells in geostationary earth orbit to collect sunlight energy. Electromagnetic microwave beam is used to transmit energy back to earth. NASA’s project on SPS prompted great developments in microwave power transfer during the 1970s and 1980s [31]. Far-field powering/charging systems can be implemented through either directive (beamforming) or non-directive radiation. Non-directive RF radiation techniques do not support line of sight, and they are less sensitive to the orientation or location relative to the transmitting antenna, so they are used in wireless power broadcasting. However, the resultant charging efficiency is relatively low. Low-power wireless systems, such as wireless renewable sensor networks (WRSNs) and RFID systems are the most embraced applications for non-directive charging [29, 31]. WRSNs with low power consumption can keep a continuous operation with RF power densities in the range of 20–200 μW/cm2 range [35]. The authors in [36] developed an ultra-low power sensor with far-field charging. The implemented transmitting and receiving sensor units consume only the power of 1.8 mW and 0.68 mW, respectively. Similar wireless charging system designs for sensors that work with intentional wireless charger have been reported in [37–39]. Furthermore, wireless charging systems based on ambient energy harvesting of RF/Microwave signals have also been developed. In [31], the authors reported the development of self-recharging sensors platform scavenging environmental RF signals from TV broadcast [40, 41], amplitude-modulated
2.5 Implementation of Far-Field WPT Systems
21
(AM) radio broadcast [42, 43], mobile communications bands (900 MHz/1800 MHz) [44–46], WiFi routers [47, 48], and satellite communications [49–51]. Wireless RF-powered sensors are also used in applications such as wireless body area networks (WBANs) [31]. WBANs are mainly categorized into implanted and wearable devices, and they have been reported in many articles [52, 53]. The power consumption of the wearable body sensors is tens of milliwatt, and the charging efficiency is very small (for example 1.2% in [54]). A charging efficiency of smaller than 0.1% was achieved for powering implanted sensors deeply inside body organs [55]. In [55–57], the authors have demonstrated biomedical implants that are powered away from tens of centimeters, with a microwatt-level RF power source. RF-powered sensors have also been used in the Internet of Things (IoT) applications [58, 59], and machine-to-machine (M2M) communication systems [60]. Directive RF beamforming is employed to provide larger power consumption for wireless charging of electronic devices. Delivering high power across long distance, through microwave beamforming techniques, has been offered in the 1960s and 1970s. In 1975 [61], William C. Brown presented the measured and calculated results for transporting electric power from one point to another via a wireless free space radiated microwave beam at 2.45 GHz. Far-field microwave beamforming has also driven the development of enormous wireless charging systems, such as SPS [62–64], microwave-driven unmanned vehicles [65, 66], Raytheon Airborne Microwave Platform (RAMP) [33], and Stationary High Altitude Relay Program (SHARP) [12, 67]. Throughout the last decade, directive RF beamforming has found its medium-power applications for charging electronic devices [31]. The commercialized Cota system [68] can offer power beam equal to 10 m without any directive transmission. Furthermore, the RF power station (beacon) has been developed to power mobile handsets through cellular networks [69, 70]. Concerning the recent contributions for the far-field (RF/microwave) energy transmission, we noticed that all researches have concentrated on three main aspects: increasing the power conversion efficiency (RF-to-DC) [35, 71, 72], human safety considerations [73, 74], and decreasing the size of the receiver (rectennas) [75]. In 2001 [76], Hagerty designed planar rectenna arrays using printed right-hand circular polarization (RHCP) and left-hand circular polarization (LHCP) spirals loaded with Schottky diodes to support fully right-and left-hand circular polarized rectennas that are able to capture all incident signals irrespective of their polarization. This design has achieved a conversion efficiency of 45% using linearly polarized wave, and with power densities of 1.5 mW/cm2 . In 2003 [77], B. Strassner, and K. Chang have developed a 5.8-GHz circularly polarized dual-loop traveling-wave rectifying antenna for low power-density wireless power transmission applications. The rectenna achieved 82% RF-to-DC conversion efficiency at an input power density of 2 mW/cm2 at 5.8 GHz using a low-profile band-reject filter to suppress the re-radiated second harmonic.
22
2 Basics of Wireless Power Transfer
Yong Park and others, in 2005 [78], designed a rectenna with a microstrip harmonic-rejecting circular sector antenna at 2.4 GHz. As compared to a conventional microstrip square patch antenna, the circular sector antenna exhibited high reflection coefficients at the second and the third harmonics generated by a diode. The rectenna integrated with circular sector antenna can remove the need for a low-pass filter (LPF) placed between the antenna and the diode, in addition to producing higher output power, with maximum power conversion efficiency of 77.8% at an input power of 10 dBm. in 2011 [78], Harouni and others proposed a 2.45-GHz rectenna using a compact dual circularly polarized (DCP) patch antenna with an RF/DC power conversion. The DCP antenna is joined to a microstrip line by an aperture in the ground plane and comprises a bandpass filter for harmonic rejections. The dual polarizations were achieved using two crossed slots etched on the ground plane. The maximum efficiency is 63% for a power density of 0.525 mW/cm2 . In 2015 [75], H. Visser has demonstrated a rectenna designed by neglecting the matching network and directly conjugate impedance matching the antenna to the rectifier. The power conversion efficiency (PCE) has been increased, and the rectenna size was reduced, as it was demonstrated with a prototype with a PCE of 55% for a −10 dBm RF input power. Table 2.1 summarizes the different implementation techniques of wireless power transmission using the near-field (inductive/capacitive) coupling techniques and the far-field (RF/microwaves/laserbeams) radiative techniques.
Table 2.1 Comparison between the different implementation methods of WPT systems Technology
Range
Frequency Antenna type
Applications
Inductive coupling
Short
Hz–MHz
charging electrical vehicles and electric tooth brush
Resonant inductive coupling
Mid
MHz–GHz Tuned wire coils, printed resonant spirals
Charging portable devices, biomedical implants, RFIDs, smartcards
Capacitive coupling
Short
KHz–MHz Electrodes
Charging portable devices, smartcards.
RF and Microwaves
Long
GHz
Rectennas, phased arrays
Solar power satellites, powering drone aircraft
Light waves
Long
THz
Laser beams, lenses, Photocells
Powering drone aircraft using photovoltaic cell panels
3D wire loops, helical antennas
2.6 Frequency Selection
23
2.6 Frequency Selection Figure 2.9 shows the possible frequencies that can be used for wireless power transfer which are called the industrial, scientific, and medical radio (ISM) band. The ISM band is a range of free license frequencies that are utilized in the scientific and industrial applications without interfering with other communications systems. Most WPT systems operate at lower frequencies to achieve high efficiency and reduce losses. Lower frequency is preferred especially for biomedical implants because body tissues are lossy media in high frequencies, also WPT systems operating at lower frequency are not affected by the surrounding environment. Tables 2.2 and 2.3 show the dielectric characteristics of biological tissues at frequencies 50 MHz and 500 MHz, respectively, to show the benefits of operating at lower frequencies [30].
915MHz
Under 135 KHz
1K
10K
13.56MHz
100K
2.45GHz
5.8GHz
433MHz
1M 10M Frequency (Hz)
100M
1G
10G
Fig. 2.9 ISM band Table 2.2 Dielectric characteristics of human body tissues at 50 MHz Tissue name
Conductivity [S/m]
Relative permittivity
Blood
1.1926
Fat
0.034677
Heart
0.65065
Muscle
0.67808
77.063
Bone cancellous
0.15505
33.258
Bone cortical
0.057124
17.744
Bone marrow
0.020102
94.205 6.8758 117.95
7.8271
Loss tangent
Wavelength [m]
Penetration depth [m]
4.5513
0.36722
0.07268
1.8131
1.8454
0.4974
1.9831
0.43503
0.11245
3.1634
0.46486
0.10098
1.676
0.85583
0.23988
1.1574
1.2657
0.44025
0.92333
1.9725
0.80276
24
2 Basics of Wireless Power Transfer
Table 2.3 Dielectric characteristics of human body tissues at 500 MHz Tissue name
Conductivity [S/m]
Relative permittivity
Loss tangent
Wavelength [m]
Penetration depth [m]
Blood
1.3834
63.257
0.78622
0.07073
0.032531
Fat
0.042793
0.27748
0.25227
0.29486
Heart
1.02
64.039
0.5726
0.072225
0.043208
Muscle
0.82245
56.445
0.52383
0.077352
0.050033
Bone cancellous
0.25397
21.95
0.41597
0.1254
0.099945
Bone cortical
0.10047
12.946
0.27901
0.16507
0.19192
Bone marrow
0.031217
5.619
0.19973
0.2517
0.4051
5.5444
2.7 Overview of Commercial Products Supporting WPT Applications using WPT technology have been most noticeable in the consumer electronics market, where wireless charging promises to deliver new levels of convenience for charging of everyday devices. In the 1990s, commercialized wireless charging products began to emerge due to the widespread usage of portable electronic devices [99]. Over the two past decades, many industrial organizations have initiated activities to develop standards and specifications correlated to the commercialization and the implementation of WPT. The Society of Automotive Engineers (SAE) has a committee developing recommendations and eventually a standard for wireless charging of electric vehicles. Furthermore, a number of industrial organizations have been established to develop specifications for WPT systems and their components (e.g., Power Matters Alliance (PMA), Wireless Power Consortium (WPC), and Alliance for Wireless Power (A4WP)) [20] as summarized in Table 2.4 [17, 100, 101]. Both far-field and near-field wireless charging approaches are experiencing developments. In 2007, a research team, led by Prof. Marin Soljacic, proposed WiTricity
Table 2.4 Wireless power transfer standards Standard
Frequency
6.78 MHz
~100–205 kHz
~201–315 kHz
Power
6.5 W
5W
5W
Coupling
Loose < 50 mm Resonance
Tight < 5 mm Inductive
Tight < 5 mm Inductive
In-band
In-band
Communications Bluetooth and power control
2.7 Overview of Commercial Products Supporting WPT
25
Above ceilings or inaccessible places
SensiƟve or hazardous areas Power TransmiƩer
Mobile devices Behind walls
Fig. 2.10 Powercast wireless charging system a Transmitter. b Receiver. c Wireless rechargeable sensor system
technology which verified through experimentations that mid-range non-radiative wireless charging is very efficient and practical. Furthermore, radiative wireless charging systems like Cota system [102], PRIMOVE [102], and Powercast wireless rechargeable sensor system [103] (shown in Fig. 2.10) have been commercialized. A survey regarding the current companies operating in the field of wireless charging, energy harvesting, and wireless power transfer, yielded the following companies: 1. Powermat This company was founded in 2006. Its first products were launched in 2009 [104]. The wireless charger uses a “ring” coil that can attach to a smartphone or tablet. Charging begins once the ringed device is placed on a mat (charging pad) [104]. 2. PowerByProxi This company was founded in 2007, and they specialized in wireless charging for smartphones and tablets, wearable devices, and industrial applications [105]. The company depends on transmitting electrical currents from charging platform to the receiving device wirelessly so that the electronic device stays charged without the need to use an outlet again. 3. WiTricity/Wireless Power Consortium (WPC) WiTricity was founded in 2007 to commercialize a new technology for wireless power transfer using strong resonant inductive coupling that was invented and patented by a team from Massachusetts Institute of Technology (MIT). WiTricity or what is called later Wireless Power Consortium is a very big organization that adopted standardization and commercialization of WPT technology. As a result, the Qi standard has emerged in 2009 [17, 106]. Qi provides wireless charging for over 300 products [17]. A Qi wireless charger can power any Qi device. It relies on the magnetic resonant
26
2 Basics of Wireless Power Transfer
coupling between the transmitting coil in the charging pad and the receiving coil in a Qi device. 4. Mojo Mobility It was founded in 2005 and they are interested in developing wireless charging technologies for products like wearable electronics, mobile phones, and even highpower electric vehicle charging [107]. One of the advantages of this product is that it can charge several devices simultaneously anyplace on the charging pad, contrasting other products that can charge only one device, and only if the device is located accurately. 5. WiPower/Qualcomm WiPower is the wireless solution being developed by Qualcomm through the Alliance for Wireless Power Standard [108]. Qualcomm is advertising the wireless charging technology to power your smartphone, lamps, wireless keyboards and mice, and digital cameras. In 2012, Qualcomm co-founded the Alliance for Wireless Power, supporting the evolution of wireless power technology, products, and services, in addition to launching a global standardization for wireless power transfer (Rezence brand) [108]. 6. Ossia It was founded in 2013, and they developed a new technology entitled “Cota” that deliver remote, targeted energy to devices safely and intelligently [68]. Cota wireless charger can charge a mobile device up to 12 m away without requiring a direct line of sight. The technology can even penetrate through walls and doors to power a device. 7. Energous Energous Corporation, founded in 2012, is the developer of WattUp wire-free charging system. WattUp is a revolutionary radio-frequency (RF)-based charging solution that delivers smart mountable power via radio bands, like WiFi access points [109].
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34. J.O. McSpadden, J.C. Mankins, Space solar power programs and microwave wireless power transmission technology. IEEE Microw. Mag. 3(4), 46–57 (2002) 35. Z. Popovi´c, E.A. Falkenstein, D. Costinett, R. Zane, Low-power far-field wireless powering for wireless sensors. Proc. IEEE 101(6), 1397–1409 (2013) 36. Y.-J. Hong, J. Kang, S.J. Kim, S.J. Kim, U.-K. Kwon, Ultra-low power sensor platform with wireless charging system, in Circuits and Systems (ISCAS), 2012 IEEE International Symposium on, (2012), pp. 978–981 37. S. Percy, C. Knight, F. Cooray, K. Smart, Supplying the power requirements to a sensor network using radio frequency power transfer. Sensors 12(12), 8571–8585 (2012) 38. C. Cato, S. Lim, UHF far-field wireless power transfer for remotely powering wireless sensors, in Antennas and Propagation Society International Symposium (APSURSI), 2014 IEEE, (2014), pp. 1337–1338 39. N. Shinohara, T. Ichihara, Coexistence of wireless power transfer via microwaves and wireless communication for battery-less ZigBee sensors, in Electromagnetic Compatibility, Tokyo (EMC’14/Tokyo), 2014 International Symposium on, (2014), pp. 445–448 40. H. Nishimoto, Y. Kawahara, T. Asami, Prototype implementation of ambient RF energy harvesting wireless sensor networks, in Sensors, 2010 IEEE, (2010), pp. 1282–1287 41. R. Vyas, B. Cook, Y. Kawahara, M. Tentzeris, A self-sustaining, autonomous, wireless-sensor beacon powered from long-range, ambient, RF energy, in Microwave Symposium Digest (IMS), 2013 IEEE MTT-S International, (2013), pp. 1–3 42. T. Sogorb, J.V. Llario, J. Pelegri, R. Lajara, J. Alberola, Studying the feasibility of energy harvesting from broadcast RF station for WSN, (2008), pp. 1360–1363 43. X. Wang, A. Mortazawi, High sensitivity RF energy harvesting from AM broadcasting stations for civilian infrastructure degradation monitoring, in Wireless Symposium (IWS), 2013 IEEE International, (2013), pp. 1–3 44. L.M. Borges et al., Design and evaluation of multi-band RF energy harvesting circuits and antennas for WSNs, in Telecommunications (ICT), 2014 21st International Conference on, (2014), pp. 308–312 45. T.B. Lim, N.M. Lee, B.K. Poh, Feasibility study on ambient RF energy harvesting for wireless sensor network, in Microwave Workshop Series on RF and Wireless Technologies for Biomedical and Healthcare Applications (IMWS-BIO), 2013 IEEE MTT-S International, (2013), pp. 1–3 46. M. Arrawatia, M.S. Baghini, G. Kumar, RF energy harvesting system from cell towers in 900 MHz band, in Communications (NCC), 2011 National Conference on, (2011), pp. 1–5 47. E.A. Kadir, A.P. Hu, M. Biglari-Abhari, K.C. Aw, Indoor WiFi energy harvester with multiple antenna for low-power wireless applications, in Industrial Electronics (ISIE), 2014 IEEE 23rd International Symposium on, (2014), pp. 526–530 48. F. Alneyadi, M. Alkaabi, S. Alketbi, S. Hajraf, R. Ramzan, 2.4 GHz WLAN RF energy harvester for passive indoor sensor nodes, in Semiconductor Electronics (ICSE), 2014 IEEE International Conference on, (2014), pp. 471–474 49. A. Takacs, H. Aubert, L. Despoisse, S. Fredon, Microwave energy harvesting for satellite applications. Electron. Lett. 49(11), 722–724 (2013) 50. A. Takacs, H. Aubert, L. Despoisse, S. Fredon, Design and implementation of a rectenna for satellite application, in Wireless Power Transfer (WPT), 2013 IEEE, (2013), pp. 183–186 51. A. Takacs, H. Aubert, S. Fredon, L. Despoisse, K-band energy harvesting circuits for satellite application, in Microwave Conference (EuMC), 2013 European, (2013), pp. 991–994 52. F. Zhang, S.A. Hackwoth, X. Liu, C. Li, M. Sun, Wireless power delivery for wearable sensors and implants in body sensor networks, in Engineering in Medicine and Biology Society (EMBC), 2010 Annual International Conference of the IEEE, (2010), pp. 692–695 53. W.Y. Toh, Y.K. Tan, W.S. Koh, L. Siek, Autonomous wearable sensor nodes with flexible energy harvesting. IEEE Sens. J. 14(7), 2299–2306 (2014) 54. N. Desai, J. Yoo, A.P. Chandrakasan, A scalable, 2.9 mW, 1 Mb/s e-textiles body area network transceiver with remotely-powered nodes and bi-directional data communication. IEEE J. Solid-State Circuits 49(9), 1995–2004 (2014)
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Chapter 3
Wireless Power Transfer Using DGSs
3.1 Introduction The recent rapid growth in wireless applications and the growing usage of consumer electronic devices have radically augmented the market prospective for wireless power transfer (WPT) technology [1–13]. The growing demand for WPT technology, especially near-field coupling techniques, is motivated by wide-ranging applications such as RFIDs [4], implanted medical devices [5–7], wireless buried sensors [8–11], and portable electronic devices [12, 13]. Moreover, near-field WPT is non-radiative and is considered to be safe for health. Non-radiative techniques are based on inductive or capacitive coupling for short-range applications [14], and resonant inductive coupling for mid-range applications [15]. Inductive coupling is the most widespread technique for highly efficient WPT systems and is usually utilized at low frequencies. At high-frequency ranges, the resonant type becomes a good preference. Resonant circuits concentrate more power at a definite frequency so that the power transfer efficiency can be enhanced. On the other hand, strong resonant coupling uses midway resonators with high Q-factors to increase the WPT efficiency [10, 16]. Most of the transmitters and the receivers for magnetic resonant coupling WPT systems were designed using 3D wired loops, spiral loops, or helical antennas [17–19]. Those coils are often massive in geometry and need accurate fabrication to keep high Q factor of the coil, which in sequence poses technical difficulties for the WPT systems to be used in small electronic devices and implanted biomedical devices. On the other hand, other WPT systems have utilized printed spirals with surface mounted (SMD) capacitors to get more compactness that are appropriate for biomedical implants and board-to-board applications [20, 21]. Small footprint WPT systems can be designed using the printed spiral coils (PSCs) [22]. Most of the reported designs, to the best of our information, employed the printed spirals for
© Springer Nature Singapore Pte Ltd. 2019 S. Hekal et al., Compact Size Wireless Power Transfer Using Defected Ground Structures, Energy Systems in Electrical Engineering, https://doi.org/10.1007/978-981-13-8047-1_3
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3 Wireless Power Transfer Using DGSs
strong resonant coupling. To reduce the size, strongly coupled printed spirals were offered for the transmitting (TX) and receiving (RX) terminals, where inductively coupled feeds were used on each terminal to realize input/output impedance matching [22–25]. A five turns self-resonant printed spiral was √ used to achieve a WPT efficiency of 43.5% at a maximum transmission distance/ bilateral ar ea (h/D) of 0.83 [23]. In [24], supplementary strips have been added to decrease the ohmic resistance which in turn increased the quality factor and realized, at h/D = 1, a power transfer efficiency of 81.7%. Furthermore, multilayer spirals with shorting walls were used to increase the Q factor and the mutual coupling [25], where an efficiency, at h/D = 1, of 84.4% was achieved. However, in [24, 25], the TX/RX spirals were designed on the same plane with the feed loops, and stacked substrates were stacked up to gather the inductance. In spite of the increased fabrication complexity in the stacked substrates, the available area for the inner loop is limited which limits further improvement of matching, mutual coupling and quality factor correspondingly. These systems, despite using high Q factor resonators, experience design complexity and restriction of transmission distance that does not exceed the maximum dimension of the resonators. Many microwave applications have employed the defected ground structures (DGSs) as quasi-lumped elements to implement low profiles of band-pass and bandstop filters (BSF) [26–29]. These compact structures have small sizes; which make them appropriate to low-profile applications like portable electronic appliances and biomedical implants. In the same manner, we can use capacitive loaded DGSs as building blocks for WPT systems. In [30], comparison between different shapes of DGSs, used to get the same band rejection response, has been performed and this study has proved that H-shaped DGS has the smallest size. The initial idea of using H-shaped DGSs for wireless power transmission has been illustrated in [31] by building H-shape resonators with an area of 25 × 25 mm2 to transfer power with transmission efficiency of 80% at a distance of 5 mm at a frequency of 1 GHz. This design has then been improved using strong resonant coupling approach in [32] to accomplish WPT efficiency of 70% at operational transmission distance of 9 mm at 1.5 GHz. We investigate here systematic and asymmetric approaches of WPT system designs using different DGS resonators, driven by capacitive coupling. Different shapes of DGS resonators (H, semi H, and spiral) are studied here with full-wave electromagnetic (EM) analysis and circuit simulations at different frequency bands. An estimated quasi-static model depending upon the current distribution is used to extract equivalent circuits for these WPT systems. In this work, full-wave EM simulator (High-Frequency Structures Simulator) for EM simulations, and Keysight’ ADS for circuit analysis are used.
3.2 An Overview on Defected Ground Structures (DGS)
(a)
(b)
(e)
35
(c)
(f)
(d)
(g)
Fig. 3.1 Different shapes of DGSs. a Circular head dumbbell. b Triangular head dumbbell. c Square head dumbbell. d Spiral DGS. e Meander lines. f U-slot. g Square open-loop with a slot in middle section
3.2 An Overview on Defected Ground Structures (DGS) In recent years, there has been a growing interest in several new concepts that can be applied to distributed (quasi-lumped) microwave circuits to meet the strict requirements of modern microwave communication systems like high performance, compact size, and low cost [33]. One of these techniques is defected ground structure (DGS), where the metal ground plane of a microstrip (or strip line, or coplanar waveguide) circuit is intentionally modified to enhance performance [34]. The basic element of DGS is etched periodic or nonperiodic cascaded resonant slots defected in the ground plane, placed directly under a transmission line, which perturb the shield current distribution in the ground plane [33, 34]. This perturbation change characteristics of the transmission line such as line capacitance and inductance. In other words, any defect etched in the ground plane of the microstrip can result in increasing the effective capacitance and inductance. Figure 3.1 shows some of the resonant DGSs that may be used. Each one varies in occupied area, equivalent RLC circuit, coupling coefficient, higher order responses, and other electrical parameters [34]. A user can choose the structure that operates better for a particular application. The design process, and the equivalent circuit extraction are challenging problems for the efficient use of DGS. The equivalent circuit aids in applying a DGS to a practical circuit design, and in knowing the critical dimensions that affect the frequency response of the DGS. The equivalent circuit elements (RLC) of DGS unit can be extracted using the following methods:
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3 Wireless Power Transfer Using DGSs
Fig. 3.2 Equivalent RLC circuit of DGS unit
Z0
L R
VS
C DGS unit
Z0
(a) Curve fitting of EM simulated S-parameters In order to extract the equivalent circuit parameters of a DGS unit at the reference plane, the scattering (S-) parameters versus frequency are calculated by EM simulation to get the cutoff and central pole frequencies of the DGS response. Then the circuit elements for the derived equivalent circuit can be extracted by fitting the EM simulated S-parameters response for the one-pole Butterworth-type low-pass response [35–37] using the formulas in [33]. The model shown in Fig. 3.2 comprising a parallel R, L, and C resonant circuit connected to transmission lines at its both sides can be used to model effectively a DGS unit. The resistance corresponds to the radiation, conductor and dielectric losses in the defect. A physical vision of the working theory of DGS can be given knowing the correspondence between the physical dimensions of DGS and the equivalent RLC parameters. The full-wave EM analysis do no assist in realizing this vision. The frequency response of DGS is not predictable until the optimized solutions are achieved through iterative optimization. The conventional design and analysis method of DGSs is described using the flow chart shown in Fig. 3.3. (b) Quasi-static modeling The limitations of the above-mentioned curve fitting method can be overcome using this modeling method, as an equivalent circuit model, can be derived from the physical dimensions of the DGS. As can be seen in Fig. 3.4a of the DGS-disturbed microstrip transmission line, the return path of the current is fully perturbed and this current is confined to the boundary of the perturbation, as shown in Fig. 3.4b, and returns below the microstrip line once the perturbation is over [38]. The width of the side filament arms, which contributes to the inductance of the DGS, is determined, based on the maximum concentration of the return current. The equivalent filament model of the DGS is shown in Fig. 3.5. Hence, the equivalent circuit model is derived using the quasi-static expressions for microstrip bends, gaps, and crosses as explained in details in [38]. The filament inductances for bends and straight lines are calculated using expressions found in [39–42]. This type of modeling clearly describes the physical operation of DGS including how the DGS generates band-pass and bandstop responses and which dimensions affect significantly the performance [38].
3.2 An Overview on Defected Ground Structures (DGS)
37
Start Select dielectric material (substrate), , thickness, metal thickness Guess dimensions of the DGS
Perform full-wave EM analysis
Extract the EM simulated Sparameters vs. frequency Change dimensions iteratively
Is the frequency response acceptable
No
Yes Extract the equivalent circuit elements ( ) Stop Fig. 3.3 Conventional design and analysis method of DGS
Fig. 3.4 Quasi-static modeling [38]. a Unit cell DGS. b Surface current on the ground plane
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3 Wireless Power Transfer Using DGSs
Fig. 3.5 Schematic equivalent current sheet (filament model) [38]
3.3 WPT Systems Using DGSs A new technique, based on coupled DGS resonators, for high-efficiency and compact size wireless power transfer (WPT) systems is proposed in this section [1–3]. Different shapes of DGSs (H, semi H, and spiral strips) are presented. Capacitive-fed resonant coupling is proposed instead of inductive-fed resonant coupling, in order to reduce the design complexity and to enhance the efficiency. The DGS resonator of the proposed systems is loaded by SMD (chip) capacitors for miniaturization. An equivalent circuit is extracted using approximated quasi-static modeling [1–3].
3.3.1 H-Shape DGS In [30], a comparison has been performed between different shapes of DGSs, used to get the same band rejection response. This comparison has verified that H-shape DGS has the smallest size and the highest Q-factor. The basic demonstration of using H-shape DGSs for wireless power transmission has been reported in [31] by designing H-shape resonators with an area of 25 × 25 mm2 to transfer power with efficiency of 80% at a transmission distance of 5 mm at 1 GHz, and has been improved using strong resonant coupling in [32] to achieve WPT efficiency of 70% at active transmission distance 9 mm at 1.5 GHz. In this part, the concept of using the H-shape DGS coupled resonators for WPT applications is confirmed. A DGS resonator located below a microstrip line perturbs the field and most of the power is coupled to the DGS slot [1]. Implementing a coupled configuration as shown in Fig. 3.6a will lead to transfer power between the two DGS slots as will be detailed in the next sections. At first, we designed the Hshape DGS resonator, shown in Fig. 3.6b, as BSF on Rogers (RO3003) substrate with permittivity εr = 3, thickness T sub = 0.762 mm, and metal thickness t = 18 μm [1]. The optimized design parameters are displayed in Table 3.1. After that, using quasistatic modeling [38], we extracted a circuit model for the H-shape DGS resonator. As can be interpreted from Fig. 3.6c, the extracted equivalent circuit includes two
3.3 WPT Systems Using DGSs
39
(a) 50 Port
h
CLS CLP 50 Port
(b)
(c) P2
P2
WS d
lsub
lH
g
Wf
H
CLP WH P1
Wsub
Top layer
W P1
Bottom layer
90°
270°
Fig. 3.6 Hekal et al. [1] a Proposed coupled H-shape DGS resonators WPT system. b H-shape DGS resonator as BSF at 300 MHz. c Simulated current distribution at phases (90°, and 270°)
parallel inductances L due to the two parallel rectangular loops of length H, width W, and thickness d. Moreover, the equivalent circuit includes two parallel capacitances: the first capacitance C g is because of the slot of length g and width W S , and the second capacitance C LP is due to the SMD chip capacitor. The equivalent circuit of H-shape DGS resonator is exported from its physical dimensions where the inductance L is calculated as (3.1) [42], the slot capacitance C g as (3.2), and the resistance as (3.3). By substituting with the calculated design parameters from Table 3.1 in (3.1)–(3.3), we get the following equivalent circuit parameters L = 30 nH, R = 0.12 , C g = 0.3 pF that lead to L P = 15 nH, C P = C g + C LP = 17.2 pF, Req = 0.06 . An equivalent circuit model is shown in Fig. 3.7a. As shown in Fig. 3.7b, we have good agreement between circuit and EM simulated scattering (S-) parameters of the DGS-BSF, which proves the offered equivalent circuit [1].
40
3 Wireless Power Transfer Using DGSs
Table 3.1 Design parameters of H-shape DGS resonator [1] Dimension
l sub
W sub
g
WS
lH
WH
H
W
d
Wf
Value (mm)
20
20
0.5
3
17
7
18.5
8.5
1.5
1.88
(a)
L
R
Cg L 50 Ω port
CLP R
LP Req CP 50 Ω port
(b)
Fig. 3.7 Verification of the quasi-static model for H-shape DGS. a Equivalent circuit [1]. b Comparison between |S-parameters| of EM and circuit simulations [1]
2W H W e − log W + W 2 + H 2 e L = e(W + H ) 0.0234 log t +d W+H
H log H + W 2 + H 2 e − W+H √
W2 + H2 t +d + 0.01 2 − 0.5 + 0.447 μ H, e = 39.37 (3.1) W+H W+H lH 2Ws (3.2) εo εr e f f cosh−1 Cg = π Ls − 21 Tsub εr + 1 εr − 1 εr e f f = 1 + 12 + 2 2 Wf
3.3 WPT Systems Using DGSs
41
CLS
Wst
Stub
WS
lst
d
lsub
Wf
g
lH H
CLP
lf
WH
Feed line
W
Wsub Top layer
Bo om layer
Fig. 3.8 PCB layout of H-shape DGS resonator for the proposed WPT system [1]
t R = Rdc −t / δ 1 − e δ 1 + dt
(3.3)
As shown in the PCB outline of the single resonator in Fig. 3.8, the top layer is a 50 microstrip line with length lf and width W f [1]. A stub loaded by capacitor is added for impedance matching, which is easily represented by a series capacitance C S calculated using (3.4). Where Z 0 is the characteristic impedance of the stub and is adjusted by the stub width (W st ), β is the phase shift constant, l S = lst + l L S is the total equivalent length of the stub loaded capacitor, lst is the stub length and l L S = β −1 tan−1 (ωZ 0 C L S ) is the additional stub length added by the loading capacitor (C LS ) [1]. CS =
1 tan(βl S ) ωZ 0
(3.4)
If the stub length offers sufficient capacitance, no loading capacitor will be required and the total length will reduce to lst . Most of the WPT structures, that use a strong resonant coupling, depend on inductive feed followed by resonant inductive coupling; here we use capacitive feed, realized by the stub-loaded capacitor, followed by resonant inductive coupling. The equivalent circuit model of the WPT system is shown in Fig. 3.9. The proposed WPT system operating at 300 MHz is fabricated using the designed parameters illustrated in Table 3.2 and Keysights’ vector network analyzer (PNA N5222A) was used in the measurements. In order to fix the distance between the two resonators, we insert a cuboid foam whose relative permittivity is 1.2. The measurement setups of the fabricated WPT systems using H-shape DGS resonators
42 Fig. 3.9 An equivalent circuit of the proposed WPT system using coupled H-shape DGS resonators [1]
3 Wireless Power Transfer Using DGSs
K Cst
CLS
50 Ω port
CP
CLS LP
LP
Req
Req
K
CS
50 Ω port
CP
CP
Cst
50 Ω port
CS
LP
LP
Req
Req
CP
50 Ω port
are shown in Fig. 3.10. The chip capacitors used are from Murata with package/case 0402 (1 mm × 0.5 mm). It can be found that the measured |S-parameters| of the H-shape DGS resonators WPT system are in good agreement with the circuit and EM simulation results as shown in Fig. 3.11. The fabricated design on a RO3003 substrate has a size of 20 × 20 mm2 . The achieved transmission distance of h = 13 mm indicates the critical coupling. The measured and simulated WPT systems are optimized for a very insignificant return loss (|S11 | < −20 dB); hence, the efficiency can be simply calculated from the following formula |S21 |2 ×100% and consequently a 68% measured WPT efficiency is achieved at 300 MHz. The following section intends to validate the proposed equivalent circuit, particularly representation of the stub, and its title role for impedance matching. The stub and its equivalent capacitance representation are shown in Fig. 3.12. From the comparison of circuit and EM simulation results shown in Fig. 3.13a, b, it can be observed that the stub is equivalent to a series capacitor, and it has a straight effect on the impedance matching and frequency adjustment. Misalignment Study This section investigates the dependability of the H-shape DGS-WPT system through analysis of its performance with the different WPT critical factors (horizontal shift and coaxial orientation misalignments) [1]. Horizontal shift misalignment The misalignment between the TX and RX resonators due to a horizontal shift in X and Y directions, as shown in the L.H.S of Fig. 3.14a, is studied. The vertical transmission distance is fixed at h = 13 mm. From the results shown in Fig. 3.14b, it can be noticed that the maximum power transfer efficiency is achieved at perfect alignment. A little decrease in the efficiency happens for the Y shift. However, a huge degradation in the efficiency occurs at a shift of 7 mm for the X and X = Y
W sub (mm)
20
Req ()
0.06
l sub (mm)
20
L P (nH)
15
0.2
C g (pF)
10
l f (mm)
17
C LP (pF)
1.9
Wf (mm)
17.2
C P (pF)
10
l st (mm)
0.7
C st (pF)
3
W st (mm)
1
C LS (pF)
0.5
g (mm)
1.7
C S (pF)
3
W S (mm)
0.025
k
16
l H (mm)
Table 3.2 Design parameters and equivalent circuit elements of H-shape DGS resonator WPT system
10
h (mm)
6
W H (mm)
73
η (%)
18.5
H (mm)
8
W (mm)
1.5
d (mm)
3.3 WPT Systems Using DGSs 43
44
3 Wireless Power Transfer Using DGSs
13 mm
Fig. 3.10 Measurement setup of the fabricated WPT systems using H-shape DGS resonators [1]
|S-parameters| (dB)
0 -5
|S21|
-10 -15
-1 -2 -3 -4 -5 295
|S11|
-20 -25 -30
Measured EM Sim. Circuit Sim.
-35
300
305
-40 285
290
295
300
305
310
315
Frequency (MHz) Fig. 3.11 Comparison between the measured and the simulated |S-parameters| of the proposed WPT system using H-shape DGS resonators at 300 MHz and at transmission distance h = 13 mm [1]
shifts where the position of the maximum magnetic field of RX resonator faces a minimum magnetic field of the TX resonator. Coaxial orientation misalignment We study here the coaxial orientation’s effect of the RX resonator at different angles with respect to the TX resonator as shown in the R.H.S of Fig. 3.14a. From the results shown in Fig. 3.14c, it can be found out that perfect orientation lead to the highest EM simulated WPT efficiency of 75% [1]. However, as the RX resonator rotates
3.3 WPT Systems Using DGSs
45
Fig. 3.12 3D schematic view of the proposed WPT system with the representation of the stub as a lumped capacitor [1]
around the z-axis, the WPT efficiency decayed to 50% at ±50° due to polarization misalignment [1]. Furthermore, the cross-polarization that takes place between the driving and the load resonators at angle ±90° leads to a destructive coupling.
3.3.2 Semi H-Shape In this section, a semi H-shape DGS resonator is presented as shown in Fig. 3.15. Considering the current distribution loops, the semi H-shape resonator consists of only one current loop which realizes a higher inductance than that of H-shape resonator that consequently increases the WPT efficiency, and increases the transmission distance [1]. The PCB layout of semi H-shape DGS resonator is shown in Fig. 3.16a. The problem with the H-shape DGS resonator emerges from the value of the equivalent inductance of the two parallel loops. The equivalent inductance is half of one of them. In addition, the opposite direction of the current flow in these loops as shown in Fig. 3.6c, is the reason for the degradation in the performance due to misalignments as explained earlier. The proposed semi H-shape DGS resonator with a half size of the H-shape doubles the inductance because of the existence of single loop [1]. The proposed structure for WPT is composed of two coupled semi H-shape DGS resonators set back to back as shown in Fig. 3.16b. The equivalent circuit of the proposed structure is shown in Fig. 3.16c. The mutual inductance between two coaxial identical square loops can be determined from the sum of the mutual inductances of the parallel wires [43]. For two parallel square loops of side length “a” and separated
46
3 Wireless Power Transfer Using DGSs
(a) 0
0
|S21| (dB)
-10
-4
-15 -6 -8
-10 285
-20
Cs=1.5pF Cs=1.9pF Cs=2.1pF Cs=2.5pF 290
295
|S11| (dB)
|S11| -5
-2
|S21| -25 300
305
310
-30 315
Frequency (MHz)
(b) 0
|S11| |S21| (dB)
-2
-5 -10
-4
-15 -6 -8 -10 285
-20
Cs=1.5pF Cs=1.9pF Cs=2.1pF Cs=2.5pF
290
295
|S21| 300
305
310
|S11| (dB)
0
-25
-30 315
Frequency (MHz) Fig. 3.13 Representation of the stub as a lumped capacitor. a CST Simulated |S-parameters| [1]. b ADS Simulated |S-parameters| [1]
by distance h, the mutual inductance, and the coupling coefficient can be calculated using (3.5), (3.6), respectively [20]. This design is implemented, approximately, with the same size of the H-shape DGS-WPT system. The same substrate is used for both designs. As a result, we get the following design parameters: lf = 10.5 mm, W f = 1.88 mm, lst = 7 mm, W st = 2.5 mm, A = 21 mm, d = 1.8 mm, g = 0.2 mm, and the circuit parameters: L P = 45 nH, C P = 5.2 pF, C S = 1.2 pF, L M = 1.1 nH, and k = 0.033.
√ a + a2 + h2 2μ0 LM = a ln − a2 + h2 + h π h
√ 2 + h2 2a a + + 2a 2 + h 2 − a 2 + h 2 − a ln (3.5) √ a2 + h2
3.3 WPT Systems Using DGSs
47
Fig. 3.14 Misalignment studies for H-shape DGS-WPT system (20 × 20 mm2 ). a Schematic of misalignment due to horizontal shift and orientation [1]. b EM simulated WPT efficiency versus misalignment shifts [1]. c EM simulated WPT efficiency versus orientation angle [1]
48
3 Wireless Power Transfer Using DGSs
H-shape
Semi H-shape
Fig. 3.15 Schematic of semi H-shape DGS resonator [1]
(a)
50 Ω port
lf
50 Ω port
CLS
d
Wf g
A lst
a
CLP
Wst
(b)
K (c)
CS 50 Ω port
CS
CP
LP
LP
Req
Req
CP
50 Ω port
Fig. 3.16 The proposed WPT system based on semi H-shape DGS resonators [1]. a PCB layout of a single resonator. b 3D schematic view. c equivalent circuit
3.3 WPT Systems Using DGSs
49
Table 3.3 Comparison between the optimum design parameters of H-shape and semi H-shape DGS resonators and their WPT efficiency at 300 MHz WPT system
Size (mm2 )
R (Ω)
LP (nH)
LM (nH)
C P (pF) C LP
C S (pF) Cg
C LS
C st
h (mm)
Meas. η (%)
H-shape
20 × 20
0.06
15
0.27
17
0.3
1.2
0.8
13
68
Semi H-shape
21 × 21
0.15
45
1.1
5
0.2
0.8
0.4
25
73
k=
LM LP
(3.6)
In [44, 45], a method to predict the WPT system maximum obtainable efficiency (ηopt ) by means of a factor named U has been discussed. The U-factor and the corresponding ηopt are calculated using (3.7) and (3.8), respectively. Table 3.3 shows a comparison between the optimum design parameters of the WPT systems using Hshape DGS and semi H-shape DGS resonators at 300 MHz. We discuss the distinction in the performance between the two structures in the following lines. As illustrated previously, the H-shape DGS is composed of two parallel loops. This configuration leads to lower losses than that of the semi H-shape DGS. However, the opposite flow of the current in the H-shape DGS loops leads to a poor coupling performance; hence, the mutual inductance is much lower than that of the semi H-shape DGS-WPT system. Consequently, the U-factor and ηopt of the H-shape DGS-WPT system have lower values than the same factors of the semi-H-shape DGS-WPT system with the same size and separation distance. ω0 L M R 2 = U 2/ 1 + 1 + U 2 U = kQ =
ηopt
(3.7) (3.8)
Table 3.3 shows a comparison between the optimum design parameters of the WPT systems using H-shape DGS and semi H-shape DGS resonators at 300 MHz. Figure 3.17 shows the magnetic field distribution of the proposed semi H-shape WPT system at different phases (0°, 90°, 180°, and 270°). Directly, the proposed WPT system using semi H- shape DGS resonators is fabricated with executing the optimum designed parameters described in Table 3.3. Keysights’ vector network analyzer PNA N5222A is used in the measurements. In order to fix the distance between the two resonators, we insert a cuboid foam with relative permittivity 1.2. The measurement setups of the fabricated WPT system using semi H-shape DGSs are presented in Fig. 3.18. The used chip capacitors are from Murata Electronics with package/case 0402 (1 mm × 0.5 mm). The parasitic effect of the used capacitor values (0.8 and 5 pF) is negligible at the desired operating frequency (300 MHz) [23]. It can be found that the measured |S-parameters| of the
50
3 Wireless Power Transfer Using DGSs
(a)
(b)
(c)
(d)
A=21 mm
h=25 mm
Fig. 3.17 EM simulated magnetic field distribution of the coupled semi H-shape resonators WPT system at 300 MHz at plane X = 0 [1]. a = 0°. b = 90°. c = 180°. d = 270°
Fig. 3.18 Measurement setup of the fabricated WPT systems using semi H-shape DGS resonators [1]
semi H-shape DGS resonators WPT system are in good agreement with the circuit and EM simulation results as shown in Fig. 3.19. According to the results shown in Fig. 3.20, the H-shape DGS-WPT system achieves maximum measured efficiency of 68% at a transmission distance of h = 13 mm, and the semi H-shape DGS type achieves a WPT efficiency of 73% at a transmission distance of h = 25 mm. Figure 3.21a, b present the measured misalignment performance comparison between the H-shape and semi H-shape DGS resonators WPT systems. The semi H-shape DGS resonators system shows no severe degradation due to the horizontal misalignment. In addition, the semi H-shape DGS resonator WPT system is not sensitive to the coaxial orientation
3.3 WPT Systems Using DGSs
51
|S-parameters| (dB)
0 -5 -10 -15 -20 -25 -30 -35 285
|S21|
|S11|
0 -1 -2 -3 -4 -5 290 295 300 305
Measured EM Sim. Circuit Sim.
290
295
300
305
310
315
Frequency (MHz) Fig. 3.19 Comparison between the measured and the simulated |S-parameters| of the proposed WPT system using semi H-shape DGS resonators at 300 MHz and at a transmission distance h = 25 mm [1]
Fig. 3.20 Measured WPT efficiency versus transmission distance (h) for H-shape and semi H-shape DGS resonators at 300 MHz [1]
misalignment. These features motivate the use of the semi H-shape DGS resonators WPT system for WPT applications. The semi H-shape DGS realizes larger inductance value, and this results in higher WPT efficiency. The proposed semi H-shape DGS-WPT system has a peak efficiency of 73% at a transmission distance of 25 mm. In turn, the figure-of-merit becomes the highest among the WPT systems proposed so far. The semi H-shape DGS resonators WPT system performance is summarized and compared with recently published WPT systems as shown in Table 3.4. A figure-of-merit (FoM) that can be computed
52
3 Wireless Power Transfer Using DGSs
Fig. 3.21 Comparison between the measured power transfer efficiency versus misalignment [1] due a horizontal shift, and b different orientation angles for H-shape (20 × 20 mm2 ) and semi H-shape (21 × 21 mm2 ) DGS-WPT systems
using (3.9) is used to compare the performance [32]. The proposed design shows the highest FoM. Besides, its WPT distance is about 1.2 times its resonators’ largest dimension. η×h FoM = √ Si ze
(3.9)
3.3 WPT Systems Using DGSs
53
Table 3.4 Comparative study with other compact WPT systems WPT system
Frequency (MHz)
Size (mm2 )
Efficiency η (%)
This work (H-shape)
300
20 × 20
68
13
0.44
This work (semi H-shape)
300
21 × 21
73
25
0.86
[23]
50
120 × 120
43.6
100
0.358
[24]
13.56
100 × 100
81.7
100
0.817
Distance h (mm)
FOM
3.3.3 Spiral-Strips DGS One of the public methods to design WPT systems is utilizing the strongly coupled printed spirals [23–25]. In strong resonant coupling WPT system, the inductive feeding was inserted between driving/loading loop and the transmitting (TX)/receiving (RX) resonator; where the driving/loading loop realizes the input/output impedance matching. On the other hand, the inductive feeding employed in [24] has the driving/load loop on the same plane to the TX/RX resonator, because it is integral to realize tight coupling to guarantee the maximum power transfer. Also, the inductive feeding needs a large size of the driving/load loops which may increase the resistive paths of the driving/loading loops and thus decrease the external quality (Q-) factor of the resonators. Moreover, within a given size of a WPT system, the driving/loading loop size limits the area where we cannot optimize the TX/RX resonators’ parameters. These parameters contain the track width (W t, i ), separation (si ), and the number of turns (N i ) to realize a high unloaded Q-factor. To avoid these problems [1], we have proposed quasi-lumped elements based on defected ground structures (DGS) for wireless power transfer. Etching a DGS on the opposite side of a microstrip feeding line realizes a band-stop filter (BSF) characteristics [27, 33, 30]. Also, the DGSs exhibit a band-pass filter (BPF) characteristics by introducing a discontinuity in the feeding microstrip line above the DGS’s slots and adding some stubs for matching [28, 29]. In the same manner, when two DGS resonators are coupled back to back, it builds a band-pass characteristic, and power is transferred from the source to load through the DGS resonators [31, 32]. In the proposed DGS-WPT system, we use capacitive coupling for feeding. As a result, the restrictions of the feeding loop is avoided and a higher achievable unloaded Q-factor is possible unlikely in the design offered in [23–25]. This section presents a new design for wireless power transfer (WPT) systems using symmetric [2] and asymmetric [3] structures for the transmitter (TX) and the receiver (RX). Both the TX and the RX are composed of high Q-factor spiral-strips DGS resonators. An analytic design procedure is used to derive the equivalent circuit of the proposed system. The quality factor of individual resonators, and the mutual
54
3 Wireless Power Transfer Using DGSs
(a)
2Di
W t,i
W t,i
CP,i
Di
CP,i
Self Inductance (nH)
vias CP,i
Di
Semi H-shape
H-shape
(b)
Di
Di
600
Di W t,i
si
Spiral strips DGS
H-shape Semi H-shape Spiral-strips DGS
500 400 300 200 100 0
20
30
40
50
60
70
Side length, Di (mm) Fig. 3.22 Comparison between three different shapes of DGS (H-shape, semi H-shape, spiral-strips DGS) [2]. a Current distribution. b Computed self-inductance
coupling are analyzed to achieve high power transfer efficiency. As discussed earlier at the beginning of this chapter, one of the first applications of DGS resonators was illustrated by the authors in [1], where two H-shape DGS resonators were coupled back to back to realize a WPT system, and power is transferred at a distance of 3.5 mm only. First, reasons for the low power transfer distance were investigated, and spiral-strips DGS was proposed to mitigate the problems in H-shape DGS resonators that encountered in [31, 32] and provides more enhancements rather than the semi H-shape DGS resonators. Figure 3.22a shows the H-shape DGS and its reformed versions (the semi Hshape, and the spiral-strips DGS) with the EM simulated current path, where there are two parallel current paths in H-shape DGS. This results in low self-inductance of the resonator which causes the lower efficiency at a larger distance [31, 32]. First, this problem is overcome by constructing a single current path only like in a Semi H-shape DGS resonator and the value of inductance is further improved as in Spiralstrips DGS resonator [2]. Figure 3.22b shows the calculated inductance of the three DGS resonators, where the elongated current path in the spiral-strips DGS provides twice the self-inductance that of the semi H- shape and four times that of the H-shape using only half the area of H-shape DGS. Therefore, the spiral-strips DGS delivers the uppermost Q-factor between the others, and they are employed in this work [2].
3.3 WPT Systems Using DGSs
(a)
55
P2
50
50
Microstrip line
Vias
Bridge
W f,i
CP,i
di Di W t,i
P1 Top layer
50
0
-10 LP,i Ri
-20 P1
-30 -40
Bottom layer
(c)
0
CP,i
P2
|S21| HFSS |S11| HFSS |S21| ADS |S11| ADS
-50 10 20 30 40 50 60 70 80 90 100
Frequency (MHz)
|S-parameters| (dB)
|S-parameters| (dB)
(b)
50
si
-10 -20 LP,i
|S21| HFSS
-30
|S11| HFSS
-40
Ri
P1
|S21| ADS
P2 CP,i
|S11| ADS
-50 10
20
30
40
50
60
70
80
90
100
Frequency (MHz)
Fig. 3.23 Proposed spiral-strips DGS resonator as BSF [2]. a PCB layout. b, c EM and circuit simulated |S-parameters| embedded with equivalent circuit extracted by quasi-static modeling and analogy with one-pole Butterworth BSF response, respectively
Figure 3.23a shows the PCB layout of the proposed spiral-strips DGS resonator operating as a band-stop filter (BSF). This DGS resonator has been designed on Rogers substrate (RO5880) with permittivity εr = 2.2, thickness T sub = 0.5 mm, and metal thickness t = 18 μm, where the top layer is 50 microstrip line of width W f,i . Using the quasi-static modeling proposed in [38], the bottom layer (DGS) is represented by the printed spiral inductor of self-inductance (L P,i ) with series resistance (Ri ), which are calculated by (3.10) and (3.11), [46, 47], respectively. Where N i is the number of turns (in this work N i = 2), μ0 is the permeability of free space = 4π × 10−7 H/m, Di and d i are the outer and the inner diameters, d i = Di − 2si − 4W t,i , and ϕ i is the fill factor, Rdc,i is the dc resistance, and δ is the skin depth. Figure 3.23b shows good agreement between the EM and circuit simulated magnitude of scattering parameters (|S-parameters|).
56
L P,i =
3 Wireless Power Transfer Using DGSs
1.27μ0 Ni2 Davg,i 2.07 + 0.18ϕi + 0.13ϕi2 Henry, i = 1, 2 (3.10) ln 2 ϕi Di + di Di − di Davg,i = , φi = 2 Di + di t , i = 1, 2 Ri = Rdc,i (3.11) δ 1 − e−t / δ 1 + Wtt,i
The corresponding RLC values of the suggested spiral-strips DGS, shown in Fig. 3.23a, are also extracted by (3.12)–(3.14) [33] from its EM band reject response. Where ω0 is the center angular frequency of the stopband, ωc is the 3 dB cutoff angular frequency taken from the S21 curve (shown in Fig. 3.23c), and Z0 = 50 . ωc 2 Farad 2Z 0 ω0 − ωC2 1 Henry LP,i = 2 ω0 C P 2Z 0 Ri = 2 1 1 − 2Z 0 (ωC P,i − ωL P,i ) − 1 |S11 (ω)|2 CP,i =
(3.12) (3.13) (3.14)
Figure 3.23b shows the electromagnetic (EM) simulated magnitude of the scattering parameters (|S-parameters|) performed using the high-frequency structure simulator (HFSS); this frequency response has been achieved using the design parameters (W f1 = 1.55 mm, D1 = 50 mm, d 1 = 34 mm, W t1 = 3.5 mm, and s1 = 1 mm) that give L P1 = 315 nH and R1 = 0.2 , and the DGS is loaded by the chip capacitor C P1 = 33 pF to get resonance at 50 MHz (ω20 = 1/L P1 C P1 ). The response in Fig. 3.23c has been achieved using the design parameters (N 1 = 2, W f1 = 1.55 mm, D1 = 50 mm, W t1 = 3.5 mm, and s1 = 1 mm) that give L P1 = 315 nH and R1 = 40 K, and C P1 = 33 pF. As earlier cited in the introduction, BPF features can be realized by introducing a discontinuity in the feeding microstrip line above the DGS’s slots, and adding some stubs for impedance matching [2]. Likewise, by adding an extra DGS [31, 32], the power is coupled between the two DGSs as can be interpreted from Fig. 3.24a, b. In this offered DGS-WPT system, capacitive coupling is used for feeding. Thus, the restrictions of the feeding/loading loops are avoided, and a higher achievable unloaded Q-factor (400) is possible not like the design proposed in [24] without/with auxiliary strips (274, and 328), respectively. Instantaneously, the DGS resonator is fed by a microstrip feeding line loaded by a further capacitor (C LS,i ). This capacitor value optimizes the external Q-factor for high-efficiency system. Figure 3.25a shows the 2D PCB layout of the planned TX/RX resonators for our proposed DGS-WPT system. The top layer is a 50 feeding line of length (L f,i ) and width (W f,i ) followed
3.3 WPT Systems Using DGSs
(a)
57 Jumper
Matching stub
Vias
CLS2
(b)
Spiral-strips DGS CP2
50 mm 50 mm
50 mm
0
|S11|&|S21| (dB)
50Ω feed line
-10 -20 -30 -40
|S21| |S11|
-50
CP1
46
48
50
52
54
Frequency (MHz) CLS1
Fig. 3.24 a Model of the proposed spiral-strips DGS-WPT system, and b its EM simulated |S11 | & |S21 |
by a stub of length (L st,i ) and width (W st,i ). The stub is represented by the parallel plate capacitance (C st,i ). The tuning capacitor (C P,i ) is an SMD chip capacitor to regulate the resonance and diminish the design area. The capacitances C LS,i and C st,i are used for impedance matching; C LS,i (IM cap) is a chip capacitor connected between the top and the bottom layers, parallel with the capacitance C st,i of the stub. Also, the equivalent circuit model of the optimized DGS-WPT system is shown in Fig. 3.25b, and its full analysis using admittance (J-) inverters is demonstrated in Fig. 3.25c to simplify the suggested design method from [48]. According to the Neumann’s formulations presented in [44, 45], the transmission distance (h) is a function of the diameter (Di ) and number of turns (N i ) of the TX/RX coils. Moreover, the maximum WPT efficiency is accomplished through an optimization of the Qfactor of individual resonators, the mutual coupling, and the impedance matching, where the power transfer efficiency can be maximized to give ηopt that is calculated by (3.8). The mutual inductance (M) and the coupling coefficient (k) are calculated by (3.15) and (3.16), respectively [49], where N 1 and N 2 are the numbers of turns of the TX and the RX resonators, respectively. 2 n=N 1 p=N 2 4 Mnp Henry π n=1 p=1 μ0 πan2 b2p 15 2 315 4 1 + γ γ = + 3 2 32 np 1024 np 2 a 2 + b2 + h 2 /
M= Mnp
n
p
D1 D2 Wt1 Wt2 an = − (n − 1)(Wt1 + s1 ) − , bp = − ( p − 1)(Wt2 + s2 ) − 2 2 2 2
58
3 Wireless Power Transfer Using DGSs
(a)
(b)
50Ω
Wf,i
50Ω
IM cap Cst1
Vias
Bridge
Lf,i
k CLS2
CLS1
Cst2
Stub Di Wst,i
50Ω port
Lst,i
CP,i
Wt,i
R1
LP1
LP2
R2
CP2
50Ω port
DGS
si
CLS,i Top layer
CP1
RX
TX
Bottom layer
(c) Lm
CS1
CS2
RS CP1
V
La
JS1
RL
JS2
Jm Lm
RS CS1 -Cse1
CP2
Lb
C1
L1
-Lm
CS2
-Lm
L2
C2
-Cse2
V
RL
CP1 = C1 - Cse1 CP2 = C2 – Cse2 L1 = La // Lm L2 = Lb // Lm
Fig. 3.25 a PCB layout of the realized TX/RX structure [2]. b The equivalent circuit of the proposed WPT system. c Analysis of the equivalent circuit using J-inverters [48]
2an b p γnp = 2 an + b2p + h 2
(3.15) k=√
M L P1 L P2
(3.16)
3.4 Design Method of the DGS-WPT Systems Figure 3.26 introduces the proposed applications using the asymmetric WPT system. For wireless charging of portable handsets, the TX is assembled in the charging pad to be unseen inside walls of transportation vehicles, or under a desk and the RX is fixed in the portable handset [3].
3.4 Design Method of the DGS-WPT Systems Desk
59
Portable handset
Vehicle wall
Charging pad Fixed handset holder
RX
D2 h = 40 mm
D2
h 40 mm
D1
RX
TX D1
TX
Fig. 3.26 The proposed applications for wireless charging of mobile handsets [3]
The frequency 50 MHz has been selected as the operating frequency to confirm the design procedure of the offered WPT systems. Bearing in mind the size constraints of the planned applications, we can encapsulate the design steps as the following: 1. Specify the available RX area and the needed transmission distance. 2. To achieve the maximum power transfer efficiency: a. In symmetric systems, we need D1 = D2 = h; where D1 and D2 are the outer diameters of both the TX and RX structures, respectively, and h is the transmission distance. √ b. In asymmetric systems, we need G M A = h, where√GMA is the geometric mean area of the TX and RX structures G M A = A T X × A R X . Hence, A T X = G M A2 /A R X → D1 = h2 /D2 . A T X and A R X are the areas of the transmitting and receiving resonators. 3. Extract the optimum design dimensions (W t,i , si , and √ N i ) for both TX and RX structures that realize the maximum U-factor (U = k Q 1 Q 2 ) using the study shown in Fig. 3.27, where k is determined by Eq. (3.5) and the unloaded Q-factor is calculated by Qi = Ri / 2π f 0 L P,i . 4. By substituting with the design parameters defined and calculated using the steps 1–3 in (3.12)–(3.16), we can catch the values of L P1 , L P2 , R1 , R2 , M, and k. 5. Apply the analytic design method in [48] (by substitution in Eqs. (3.2), (3.6), (3.5), (3.1), and (3.3), respectively, in [48]) to get the remaining circuit parameters C P1 and C P2 and the values of C S1 and C S2 that are realized in our circuit model by C S1 = C st1 + C LS1 , C S2 = C st2 + C LS2 . This design method depends on achieving the perfect impedance matching of the system through satisfying the condition J S1 J S2 R = J m (Eq. (3.5) [48]), where R = RS = RL = 50 . 6. Using L st,i and W st,i , the parallel plate capacitance C st,i can be simply calculated and consequently, the requisite chip capacitor can be calculated from C LS,i = C S,i − C st,i .
60
3 Wireless Power Transfer Using DGSs
Fig. 3.27 Investigation of the computed U-factor of the coupled resonators versus the width W t,i and separation si for the proposed symmetric WPT systems (50 × 50 mm2 ) [3] Table 3.5 Optimized design parameters and equivalent circuit RLC values of the proposed WPT system using spiral-strips DGS [3] L, M (nH), R (), and C (pF)
Design dimensions (mm) L f,i
W f,i
L st,i
W st,i
Di
di
W t,i
si
L P,i
Ri
C P,i
C st,i
TX, i=1
25
1.55
20
2.05
50
34
3.5
1
315
0.2
24
2
RX, i=2
15
1.55
11
2.15
30
18
2.5
1
150
0.15
55
1.5
C LS,i M 7
8.3
k 0.04
11
7. Lastly, the design parameters of the proposed symmetric and asymmetric WPT systems are fine-tuned using the EM simulator for further efficiency improvement and final optimization before fabrication. Figure 3.27 presents a study on the symmetric WPT system (50 × 50 mm2 ) that shows the influence of the strip width, W t1 , and the separation, s1 , variations on the values of the U-factor using N 1 = 2. We have executed this study also for N 1 = 3, 4, 5, etc. and found that the highest U-factor is reached at N 1 = 2. Similarly, a study has been performed on the symmetric WPT system (30 × 30 mm2 ) and the asymmetric system. These studies have determined the values of N 1 = N 2 = 2, W t1 = 3.5 mm, W t2 = 2.5 mm and s1 = s2 = 1 mm for the maximum value of the U-factor. Table 3.5 specifies the final design parameters and the equivalent RLC values for the proposed asymmetric WPT system.
3.4 Design Method of the DGS-WPT Systems
61
Optimum WPT efficiency, ηopt (%)
100 90 80 70 60 50 40 30 20 10 0 10
30
50
70
90
110
130
150
Transmission distance, h (mm) Fig. 3.28 Optimum WPT efficiency at different transmission distances for the symmetric WPT system (50 × 50 mm2 )
√ According to the formula in (3.8), ηopt = U 2 /(1 + 1 + U 2 )2 × 100%, we can predict the optimum WPT efficiency that can be achieved at different transmission distances (h) after selecting the optimum TX/RX dimensions (N i , W t,i , and si ) that result in the highest U-factor [3]. Figures 3.28 and 3.29 show the maximum predictable WPT efficiency of the symmetric system (50 × 50 mm2 ) and asymmetric system versus the transmission distance. As shown, ηopt decays with increase of h due to reduction of the mutual coupling as the distance between TX and RX increases. Each transmission distance results in a certain value for the mutual inductance (M), so different combinations of parallel and series capacitances C PT X , C PR X , C ST X , and C SR X ) are required to achieve ηopt with perfect impedance matching at the central resonant frequency (f 0 ). Table 3.6 and Table 3.7 display the optimum values of the parallel and series capacitances for the symmetric (50 × 50 mm2 ) and asymmetric WPT systems, respectively, at the different transmission distances. The values of parallel and series capacitances are calculated using the new design method, which is explained in detail in Sect. 4.3. Figures 3.30 and 3.31 show a glance of the EM simulations for the symmetric (50 × 50 mm2 ) and the asymmetric WPT systems, respectively, by showing the magnetic field distribution at different phases 0°–180° [3].
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3 Wireless Power Transfer Using DGSs
Optimum WPT efficiency (ηopt)
100 90 80 70 60 50 40 30 20 10 0
10
20
30
40
50
60
70
80
90
100
Transmission distance, h (mm) Fig. 3.29 Optimum WPT efficiency at different transmission distances for the asymmetric WPT system Table 3.6 Optimum design parameters (C PT X , C PR X , C ST X , and C SR X ) to achieve ηopt for the symmetric WPT system (50 × 50 mm2 ) at each transmission distance [3] h (mm)
M (nH)
k
C PT X = C PR X (pF)
C ST X = C SR X (pF)
10
139.22
0.450
12
40.5
20
72.4
0.234
13.4
24.3
30
37.88
0.123
17.6
16.6
40
21.53
0.069
21
12.3
50
13.17
0.0426
23.5
9.5
60
8.5
0.027
25.2
7.6
70
5.79
0.019
26.5
6.3
80
4.09
0.013
27.5
5.3
90
2.98
0.0096
28.2
4.5
100
2.24
0.007
29
4
110
1.72
0.0055
29.3
3.4
120
1.34
0.004
29.7
3
130
1.07
0.0035
30
2.7
140
0.866
0.003
30.3
2.4
150
0.71
0.0023
30.5
2.2
3.4 Design Method of the DGS-WPT Systems
63
Table 3.7 Optimum design parameters (C PT X , C PR X , C ST X , and C SR X ) to achieve η opt for the asymmetric WPT system at each transmission distance [3] h (mm)
M (nH)
k
C PT X (pF)
C ST X (pF)
C PR X (pF)
C SR X (pF)
10
64.77
0.30
20
30.6
0.142
12
28.8
44
46.7
16.7
18
46
30
15.17
27
0.07
21
12.4
51
18
40 50
8.27
0.039
24
9
55
13
4.89
0.023
26
7
58
10
60
3.08
0.014
27.3
5.5
60
8
70
2.06
0.009
28.3
4.5
61
6.5
80
1.43
0.007
29
3.7
62
5.4
90
1.03
0.005
29.5
3.2
63
4.5
100
0.76
0.004
30
2.7
63.7
4
50 mm
RX 50 mm
TX 50 mm
Ф = 0˚
Ф = 90˚
Ф = 135˚
Ф = 45˚
Ф = 180˚
Fig. 3.30 Magnetic field distribution of the symmetric (50 × 50 mm2 ) WPT system at phases F = 0°, 45°, 90°, 135°, and 180° [3]
64
3 Wireless Power Transfer Using DGSs
30 mm
RX 40 mm
TX 50 mm
Ф = 0˚
Ф = 90˚
Ф = 135˚
Ф = 45˚
Ф = 180˚
Fig. 3.31 Magnetic field distribution of the asymmetric WPT system (TX 50 × 50 mm2 & RX 30 × 30 mm2 ) at phases F = 0°, 45°, 90°, 135°, and 180° [3]
3.5 Fabrication and Measurements Figure 3.32 shows the measurement setup of the fabricated WPT systems (symmetric, and asymmetric, respectively) using the Vector Network Analyzer (Agilent N5222A). Figure 3.33a–c displays the measured and simulated S-parameters (|S 11 | & |S 21 |) for the offered WPT systems, also for the asymmetric WPT systems [3]. As shown, the symmetric and asymmetric WPT systems operate at 49.5 MHz, and the measured results are in good agreement with the simulated results. The frequency shift between the measured and the simulated |S-parameters| is due to the tolerance of the used SMD chip capacitors (C P,i and C LS,i ).
3.5 Fabrication and Measurements
65
Top layer
Bottom layer
RX h = 30 mm TX RX
RX
h = 40 mm
h = 50 mm
TX TX
Fig. 3.32 Measurement setup of the fabricated WPT systems (Symmetric 50 × 50 mm2 , Symmetric 30 × 30 mm2 , and Asymmetric TX = 50 × 50 mm2 , RX = 30 × 30 mm2 ) [3]
(b)
0
-10
-20 -30 -40
|S21| HFSS |S11| HFSS |S21| Meas. |S11| Meas. |S21| ADS |S11| ADS
0 -0.5 -1 -1.5 -2 -2.5 -3 48 49 50 51
|S-parameters| (dB)
|S-parameters| (dB)
(a)
-50 43 44 45 46 47 48 49 50 51 52 53 54 55
0 -5
-10 -15 |S21| HFSS |S11| HFSS |S21| Meas. |S11| Meas. |S21| ADS |S11| ADS
-20 -25 -30
Frequency (MHz)
(d) -5
-10
-15 -20 -25 -30 -35
|S21| HFSS |S11| HFSS |S22| HFSS |S21| Meas. |S11| Meas. |S22| Meas. |S21| ADS |S11| ADS |S22| ADS
0 -0.5 -1 -1.5 -2 -2.5 -3
48 49 50 51 -40 43 44 45 46 47 48 49 50 51 52 53 54 55
Frequency (MHz)
WPT efficiency, η (%)
0
|S-parameters| (dB)
48 49 50 51
-35 43 44 45 46 47 48 49 50 51 52 53 54 55
Frequency (MHz)
(c)
0 -0.5 -1 -1.5 -2 -2.5 -3
90 80 70 60 50 40 30 20 10 0
Symmetric 50x50
Asymmetric
Symmetric 30x30 20
Measured HFSS
30 40 50 60 70 Transmission distance, h (mm)
80
Fig. 3.33 Comparison between the measured and the simulated |S-parameters|. a Symmetric 50 × 50 mm2 at h = 50 mm. b Symmetric 30 × 30 mm2 at h = 30 mm. c Asymmetric TX = 50 × 50 mm2 , RX = 30 × 30 mm2 at h = 40 mm. d Measured WPT efficiency versus different transmission distances [3]
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3 Wireless Power Transfer Using DGSs
Table 3.8 Comparison of the proposed spiral-strips DGS-WPT system with the recent published planar WPT systems [3] Reference
f 0 (MHz)
TX area (mm2 )
RX area (mm2 )
Distance h (mm)
Efficiency η (%)
[24] one layer
13.5
100 × 100
100 × 100
100
77.27
[24] two layers
13.5
100 × 100
100 × 100
100
81.67
[25] one layer
13.5
100 × 100
100 × 100
100
77.27
[25] two layers
13.5
100 × 100
100 × 100
100
82
This work, symmetric
50
50 × 50
50 × 50
50
84
This work, asymmetric
50
50 × 50
30 × 30
40
78
Applying the formula of η to calculate the WPT efficiency, from Fig. 3.33d, the symmetric WPT system (TX = RX = 50 × 50 mm2 ) achieves a maximum efficiency of 84% at transmission distance h = 50 mm, which is comparable to the value recently achieved in [4], but without using complex stacked substrates. The asymmetric WPT system has achieved a measured WPT efficiency of 78% at h = 40 mm. The higher area of the TX offers high U-factor that allows the asymmetric WPT system to provide a superior efficiency at a longer transmission distance than the symmetric case (TX = RX = 30 × 30 mm2 ). Table 3.8 introduces a comparison of the offered WPT systems (symmetric and asymmetric) with the recent published planar WPT systems regarding measured WPT efficiency, the TX/RX areas, and the maximum transmission distance. This comparison verifies that the proposed WPT systems, using spiral-strips DGS resonators, can offer higher efficiencies than that of the conventional strongly coupled printed resonators. The proposed spiral-strips DGS resonators provide a new efficient and compact design for asymmetric WPT systems. The offered asymmetric WPT system achieves a TX efficiency of 78% at a distance of 40 mm. The TX and RX areas are 50 × 50 mm2 and 30 × 30 mm2 , respectively. This structure can be employed in wireless charging applications for the portable electronic devices (mobile phones, laptop, etc.) that need compact RX structure regardless of the TX’s size embedded in the external charging pad. In the same manner, the spiral-strips DGS-WPT system with the optimized design parameters in Table 3.9 was fabricated on FR4 substrates (TX = RX = 100 × 100 mm2 ) with permittivity εr = 4.4, and thickness = 0.8 mm to operate at the ISM band f 0 = 13.5 MHz. Figure 3.34 shows the measurement setup of the fabricated WPT system. Figure 3.35 shows the simulated and the measured |S-parameters| at transmission distance h = 10 cm. As shown, Fig. 3.35 records maximum power trans-
3.5 Fabrication and Measurements
67
Table 3.9 Optimum design parameters and equivalent RLC values of the proposed spiral-strips DGS-WPT system (100 × 100 mm2 ) fabricated on FR4 substrate at f 0 = 13.5 MHz Dimensions (mm) Equivalent circuit
Wf
W st
L st
D
Wt
s
1.5
2.4
40
100
8
2
L (nH)
R ()
C P (pF)
C st (pF)
C LS (pF)
k
550
0.15
205
7.7
42
0.04
10 cm
Fig. 3.34 Fabricated designs and measurements of the proposed spiral-strips DGS-WPT system at f 0 = 13.5 MHz [3]
fer efficiency at f 0 = 13.5 MHz with good agreement between the measured and the simulated results. The proposed WPT system is able to provide 81.7% measured efficiency using one layer due to higher Q-factor (339) compared to [23, 24] (efficiency 77.27%, and Q-actor = 274). In the future, we can apply the two layers (by printing auxiliary strips in the top layer to decrease the series resistance and so increase the unloaded Q-factor) that is expected to give improvement in WPT efficiency (2–3%).
68
3 Wireless Power Transfer Using DGSs 0
-0.5 -1.0 -1.5 -2.0 13
|S-parameters|, dB
-5 -10
-15
13.5
14
-20 -25
S11
-30
S21
Circuit sim. EM sim. Measured
-35 -40 -45 11
12
13
14
15
16
Frequency, MHz Fig. 3.35 Simulated and measured |S-parameters| for the proposed spiral-strips DGS-WPT system at h = 10 cm and f 0 = 13.5 MHz [3]
3.6 Power Transmission Through the Human Body The idea of using the electromagnetic waves as wireless power source dates back to the end of the nineteenth century when Nikola Tesla demonstrated the idea for the transmission of electrical energy over a certain distance without any electrical connections. Recently, due to the development of low-power-consuming electronic devices, and the emerging the Internet of Things (IoT) technology, Wireless Power Transfer (WPT), and Energy Harvesting (EV) techniques have become the supporting technology for execution of Energy Autonomous Systems (EASs), implemented to work without onboard batteries. Although the technology of WPT is widely used nowadays for charging of portable electronic devices (mobile phones, laptops, etc.), there are other devices that can benefit from this technique such as the implantable biomedical devices for acute diabetes and heart diseases. One of the major design challenges of these implantable devices is the need for using small size batteries. Because of the limited batteries lifetime, it is essential to operate on patients (like the operation of battery replacement every 2–3 years), with risks to the health and additional costs. A feasible solution to overcome this disadvantage is the use of wireless power transfer. WPT can be realized either by directly providing power to the implantable device, or by charging wirelessly rechargeable batteries. Therefore, it is extremely important to focus research efforts on efficient wireless powering of biomedical implants. This section tests the reliability of the symmetric (50 × 50 mm2 ) spiral-strips DGSWPT system in Fig. 3.32 to transfer the power through a human body. Figure 3.36 shows a sample representation of the existence of a human body tissue by insertion of one of the authors’ hand between the TX and the RX. Figure 3.37 displays a comparison of power transfer efficiency for the WPT system in Fig. 3.36 with and without insertion of the author’s hand. As shown in Fig. 3.37, there is 13% decrease
3.6 Power Transmission Through the Human Body
69
Fig. 3.36 Representing human life tissue effects on the efficiency of power transmission by insertion of a human hand 0
0
-1
|S21| Without Hand
|S21| (dB)
-10
|S21| with Hand
-3
|S11| Without Hand
-4
|S11| with Hand
-5
-20
-6
-7
|S11| (dB)
-2
-30
-8 -9
-10
-40
40
42
44
46
48
50
52
54
56
58
60
Frequency (MHz)
Fig. 3.37 Comparison between the measured |S-parameters| of the WPT system in Fig. 3.36 with and without a human hand presence, where the transmission distance is 50 mm
in the power transfer efficiency from η = 84% (without insertion of the author’s hand) to η = 73% (with the insertion of the author’s hand). This low value of decrease is expected to be 30% with the body tissues fully filling the area between TX and RX, which enables wireless charging through the human body.
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3 Wireless Power Transfer Using DGSs
3.7 Power Handling Capability of the Proposed WPT Systems The power handling capability can be defined as the maximum input or output power that a component, circuit, device, or system can provide or handle without damage. The power handling capability is limited by heating caused by conductor and dielectric losses. The used Rogers substrates, with a thermal conductivity of 0.5–0.7 and loss tangent of 0.001–0.0025, has been tested using the free software MWI, to calculate the temperature rise with RF power, and was found to be 0.18–0.25 Cº/W. The Average power handling capability (APHC) is calculated by (3.17) where Tmax is the maximum operating temperature of substrate, Tamp is the ambient temperature, and T is the rate of temperature rise (C°/W). According to the formula of APHC in (3.17), our proposed structure can be used for low power applications in the range of 0.1–1 W. Modifications for the structure like increasing metal thickness can be made in order to be suitable for higher power applications. AP HC =
Tmax − Tamb T
(3.17)
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Chapter 4
Design Methods
4.1 Introduction The equivalent circuit model of the inductive feed [1, 2] and capacitive feed [3] resonant inductive WPT systems have been analyzed to extract the S-parameters and the input impedance. Nevertheless, this method of analysis results in complicated design procedures. For example, complex equations needed to be solved in order to obtain the conditions for the perfect impedance matching and maximum power transfer for the WPT systems presented in [1, 2, 4]. Other design methods, like the one explained in Sect. 4.2 [3], have been developed to yield concise design equations using impedance and admittance inverters. These design methods result in simple design equations that can indicate rapidly, and accurately, the perfect impedance matching conditions.
4.2 Design Method #1 A conventional wireless power transfer (WPT) system composed of two coupled resonant coils is shown in Fig. 4.1a [3]. The coils are represented by inductors, L P1 and L P2 , and the parallel capacitors, C P1 and C P2 , respectively. The series capacitors C S1 and C S2 connect the transmitting coil to a power source, and the receiving coil to a load, respectively. An equivalent circuit based on admittance inverters can be used to provide concise analytic design equations to find the lumped element values shown in Fig. 4.1a. Figure 4.1b shows the equivalent circuit of Fig. 4.1a which is composed of two resonant coils. Parallel resonators are used to model the resonant coils, as the impedance of each resonant coil is maximum at the resonant frequency [5]. Admittance inverters with inverter values of J S1 and J S2 , connect the sending and receiving coils to the source and the load, respectively. An admittance inverter with an inverter value of J m connects the two coils. © Springer Nature Singapore Pte Ltd. 2019 S. Hekal et al., Compact Size Wireless Power Transfer Using Defected Ground Structures, Energy Systems in Electrical Engineering, https://doi.org/10.1007/978-981-13-8047-1_4
73
74 Fig. 4.1 A wireless power transfer systems using two coils [3]
4 Design Methods
(a)
M
C S1
C S2
R C P1
L P1
R
C P2
L P2
V
(b) Y S1
Y S2
Ym
J S1
J S2
Jm
R
C1
L1
L2
C2
R
V
The conditions under which the equivalences shown in Fig. 4.2 are valid can be given by [3]: C S1 =
JS1
ω0 1 − (JS1 R)2 C S1 C S1e = 1 + (ω0 C S1 R)2 JS2 C S2 = ω0 1 − (JS2 R)2 C S2 C S2e = 1 + (ω0 C S2 R)2
(4.1)
The ABCD matrix of the circuit in the right side of Fig. 4.2c, can be obtained, at the operating frequency as follows: 1 1 0 1 − 1 0 j Jm ABC Dr = 1 1 1 1 + j Jm 1 jω0 L 1 jω0 L 2 1 − j 1Jm ω0 L 2 Jm = (4.2) j Jm + jω2 L1 L J ω0 L11 Jm 0
1
2 m
4.2 Design Method #1
75
Fig. 4.2 Equivalent circuits of admittance inverters in Fig. 4.1b [3]
(a)
(b)
(c)
Similarly, the ABCD matrix of the circuit in the left side of Fig. 4.2c can be given by ABC D = l
L P1 M 1 jω0 M
P2 −M jω0 L P1 L M
L P2 M
2
(4.3)
The equivalency can be achieved when the two matrices ABC Dr and ABC Dl are identical to each other. By equating (4.2) and (4.3), the two circuits in Fig. 4.2c can be shown to be equivalent, when (4.4) is satisfied. L P1 = L P2 =
1−
L1 2 ω0 L 1 L 2 Jm2 L2 2 ω0 L 1 L 2 Jm2
1− ω0 L 1 L 2 Jm M= 1 − ω02 L 1 L 2 Jm2
(4.4)
With the knowledge of the design constraints for the TX and RX resonators, the equivalent circuit elements L P1 , L P2 , and M are calculated easily. By substitution of
76
4 Design Methods
L P1 , L P2 , and M in the system of equations (4.4), we can get the values of L 1 , L 2 , and J m. Continue applying the design method by substitution in (4.5) to find C 1 and C 2 to achieve resonance at ω0 [3]. C1 =
1 1 , C2 = 2 L 1 ω0 L 2 ω02
(4.5)
The impedance matching condition needed to achieve the maximum power transfer from the source to the load, can be used to obtain the input and output admittance inverters (J S1 and J S2 ). The input admittances looking into the input port of each inverter at the resonant frequency can be given by J2 2 Y S2 = S2 = JS2 R 1 R Ym = Y S1 =
Jm2 J2 = 2m Y S2 JS2 R
2 JS1 J2 J2 R = S1 2S2 Ym Jm
(4.6)
Perfect impedance matching is obtained when the input admittance seen looking into the input port of the first inverter, Y S1 , is identical to the source admittance, 1/R. Consequently, the circuit shown in Fig. 4.1b achieves the maximum power transfer from the source to the load provided that JS1 JS2 R = Jm
(4.7)
In the symmetric system, J S1 = J S2 . In the asymmetric system, we assume J S1 or J S2 to get the other. By substitution of J S1 and J S2 in (4.1), we can get the values of C S1 and C S2 that are realized in our circuit model by C S1 = C st1 + C LS1 , C S2 = C st2 + C LS2 [6]. Finally, we calculate C P1 and C P2 using (4.8) as follows: C P1 = C1 − C S1e , C P2 = C2 − C S2e
(4.8)
The following steps present the implementation of the design method #1 that can be applied for the proposed DGS WPT systems as follows [6]: 1. Define the required RX size (D2 × D2 ), then the necessary transmission distance (h) or the target WPT efficiency (η). 2. For the maximum WPT efficiency:
4.2 Design Method #1
77
In the symmetric systems, we need D1 = D2 = h; where D1 and D2 are the outer diameters of both the TX and RX structures, respectively, and h is the transmission distance. √ B. In the asymmetric systems, we need G M A = h, where GMA is the √ geometric mean area of the TX and RX structures, G2 M A = T X si ze × R X si ze. Hence, TX size should be equal to GMA /RX size → D1 = h2 /D2 . 3. Extract the optimum design dimensions (W t,i , si , and N √i ) for both the TX and RX structures that achieve the highest U-factor (U = k Q 1 Q 2 ). 4. By substitution with the design parameters defined and calculated using steps 1–3 in (3.10)–(3.16), we can get the values L P1 , L P2 , R1 , R2 , M, and K. 5. Applying the analytic design method discussed above and reported in details in [3] by substitution in (4.1), (4.4), (4.5), (4.7), and (4.8), respectively. 6. Apply the equivalent circuit elements on the circuit simulator (ADS) for verification and pre-optimization of the capacitor values C P1 , C P2 , C S1 , and C S2 .
Start Define the design constraints: RX area = D2 x D2 TX area = D1 x D1
Apply the impedance matching condition to get JS1and JS2
By using Matlab, compute the initial design parameters Wt1, Wt2, s1, s2 , N1, and N2 that give the optimum U-factor, U =
Is the WPT system symmetric?
Yes
Assume JS1 then calculate JS2
Calculate the equivalent circuit parameters LP1, R1, LP2, R2, M, and k Solve the following system of equations using matlab to get L1, L2, and Jm
No
Solve the following system of equations to get CS1 , CS2, CP1, and CP2
, , ,
Define the operating frequency C2 using
to get C1,
CS1 and CS2 are realized by CS1 = Cst1 + CLS1 and CS2 = Cst2 + CLS2 Apply the extracted design parameters on HFSS for fine tuning and final optimization End
Fig. 4.3 Flowchart of design method #1
78
4 Design Methods
7. Finally, apply the optimum design parameters of the proposed WPT system on the full-wave EM simulator (HFSS) for fine tuning and final optimization. The implementation steps of design method #1 are summarized in the flowchart shown in Fig. 4.3.
4.3 Design Method #2 We present a novel design method for coupled printed spiral resonators WPT systems. This design approach extends the concept of J-inverters and models the coupled resonators WPT system as a second-order Butterworth BPF. Figure 4.4a presents the proposed system block diagram. Each spiral inductor is loaded by a parallel capacitor to achieve resonance at the target resonance frequency. Also, a series capacitor is added to both of the TX/RX pairs to improve impedance matching. The equivalent circuit is shown in Fig. 4.4b. Figure 4.4c details the equivalent circuit showing the J-inverters. Equations (4.9) and (4.10) calculate the equivalent circuit elements of the pi model (L M , L a , and L b ) representing the magnetic coupled inductors (L 1 and L 2 ). The generalized Jinverters can be computed using (4.11) [7], and g0 , g1 , g2 and g3 are √the prototype filter’s (Butterworth) coefficients where g0 = g3 = 1 and g1 = g2 = 2. R S and R L are the source and load resistances, respectively, R S = R L = R0 = 50 . F BW is the system fractional bandwidth, and b1 and b2 are the susceptance slope parameters. The ith susceptance slope parameter (bi ) can be calculated using (4.12) [7]. Where, Bi is the ith susceptance and L ri is the ith resonating inductance. L1 L2 − M2 M L1 L2 − M2 La = L2 − M L1 L2 − M2 Lb = L1 − M
Lm =
(4.9)
L r 1 = L a //L m → L r 1 = L 1 − M 2 /L 2 L r 2 = L b //L m → L r 2 = L 2 − M 2 /L 1 J01 =
(4.10)
b1 F BW/R0 g0 g1 J12 = F BW b1 b2 /g1 g2 J23 = b2 F BW/R0 g2 g3
(4.11)
bi = (ω0 /2) × ∂ Bi /∂ω|ω=ω0 = 1/ω0 L ri
(4.12)
J12 = 1/ω0 L m
(4.13)
4.3 Design Method #2
79
D2
(a)
(b)
CS1
RX
CS2
M
RS
V
TX
CP1
L1
L2
RL
CP2
D1
Lm
CS1
CS2
RS CP1
V (c)
La
J01
RL
CP2
Lb
J23
J12 Lm
RS CS1 -Cse1
Cr1
Lr1
-Lm
-Lm
CS2 Lr2
Cr2
-Cse2
RL
V Fig. 4.4 a Proposed system block diagram. b Its equivalent circuit. c Equivalent circuit based on J-inverters
J01 ω0 1 − (J01 R0 )2 Cs1 Cse1 = 1 + (ω0 Cs1 R0 )2 J23 Cs2 = ω0 1 − (J23 R0 )2 Cs2 Cse2 = 1 + (ω0 Cs2 R0 )2 F BW = J12 g1 g2 /b1 b2 = (1/L m ) g1 g2 L r 1 L r 2 Cs1 =
J01 =
√ √ J12 g1 g2 J12 g1 g2 b1 /b2 = L r 2 /L r 1 R0 g0 g1 R0 g0 g1
(4.14) (4.15)
80
4 Design Methods
Specify the TX/RX sizes (DxD), the target transmission distance (h), and the operating frequency ( )
Determine the TX/RX self-inductance ( ), the resistance ( ), and the mutual inductance ( ) using (3.10), (3.11), and (3.15), respectively.
Substitute in (4.9)-(4.17) to calculate the values of the circuit
components (
and
)
Fig. 4.5 The analytical design procedure of the symmetric WPT system
J23 = Cr 1 =
√ √ J12 g1 g2 J12 g1 g2 b2 /b1 = L r 1 /Lr2 R0 g2 g3 R0 g2 g3 C p1 = Cr 1 − Cse1
1/ω02 L r 1
Cr 2 = 1/ω02 L r 2 Cr 2 = 1/ω02 L r 2
C p2 = Cr 2 − Cse2
L r 1 = L 1 − M 2 /L 2
(4.16)
(4.17)
L r 2 = L 2 − M 2 /L 1
The relations between the J-inverters and the circuit component can be derived easily. The center J-inverter ( J 12 ) is related to the mutual coupling between the two resonators as (4.13), where ω0 is the angular resonant frequency. Also, the input and output J-inverters are related to the coupling capacitors as (4.14) [7]. Using (4.11) and (4.12), we derive the F BW as (4.15). Also, using (4.11) and (4.15), the input and output J-inverters reduce to (4.16). Then, the resonating capacitors can be computed using (4.17). Finally, the design procedures for the symmetric and asymmetric structures are shown in Figs. 4.5 and 4.6.
4.4 Verification of Design Method #2 4.4.1 Symmetric WPT System In this section, we apply the proposed method on the symmetric WPT system at 50 MHz frequency. The size of the receiver is set to 30 × 30 mm2 . Applying the design procedures shown in Fig. 4.5, the optimum WPT separation distance is h = 30 mm. The design model is shown in Fig. 4.7.
4.4 Verification of Design Method #2
Specify the RX size (
81
), the target transmission distance (h), and the operating frequency ( )
Use
to determine the required TX size (
), the resistances
Determine the TX and RX self-inductances ( and (
and
)
), and the mutual inductance ( ) using (3.10), (3.11), and (3.15), respectively.
Apply and substitute in (4.9)-(4.17) calculate the values of the circuit component (
)
Fig. 4.6 The analytical design procedure of the asymmetric WPT system
(a)
(b)
via
hd
substrate
Fig. 4.7 The proposed WPT system a 3D view. b Planar view
A two-turn spiral inductor topology is selected. The used substrate is RO3003 (εr = 3, Tsub = 0.762 mm and t = 18 µm). The dimensions and the components of the WPT system and its performance in terms of WPT efficiency (ηW P T ) are summarized in Table 4.1. The spiral inductor parameters (trace width, trace separation, and number of turns) are selected to maximize the obtainable WPT efficiency [8].
82
4 Design Methods
Table 4.1 Summary of the designed, simulated and optimized parameters and performance of the symmetric WPT system Method
Analytical
Simulated (ADS)
Optimized (HFSS)
f 0 (MHz)
50
49.94
50
DoT X × DoT X (mm2 )
30 × 30
–
30 × 30
DoR X × DoR X (mm2 )
30 × 30
–
30 × 30
l FT X = l FR X (mm)
–
–
5
W FT X = W FR X (mm)
–
–
1
g FT X
=
g FR X
(mm)
h (mm)
–
–
2.5
30
–
30
N1 = N2
2
–
2
WT X = W R X (mm)
2.5
–
2.5
sT X = s R X (mm)
1
–
1
L T X = L R X (nH)
149
149
–
M (nH), k
6.76, 0.045
6.76, 0.045
–
RT X = R R X ()
0.3
0.3
–
= =
C PR X C SR X
(pF)
54.4
54.4
54
(pF)
14.4
14.4
13
–
75.2
75.5
ηWPT (%) Fig. 4.8 Comparison between the simulated |S-Parameters| of the symmetric WPT system using ADS and HFSS
0
|S-Parameters| (dB)
C PT X C ST X
|S11|
-10 -20 -30 -40
|S21| HFSS
-50 -60 40
ADS 45
50
55
60
Frequency (MHz)
We compute the WPT efficiency (ηW P T ) of a perfectly matched system as ηW P T = |S21 |2 . Where |S21 | is the magnitude of the transmission coefficient. The optimized capacitors values using HFSS agree with the calculated ones, which prove the accuracy of the proposed analytical design procedure. Also, the simulated performance using ADS and HFSS is compared in Fig. 4.8 showing good agreement.
4.4 Verification of Design Method #2
83
4.4.2 Asymmetric WPT System A major limitation of the symmetric resonant inductive coupling WPT system is the transmission distance (h). The WPT separation is limited to the square root of the spiral inductor area to achieve the critical coupling; hence, a maximum WPT efficiency [8]. So, the size of the spiral resonator must be increased to allow for a larger transmission distance. However, increasing the size is not desirable for many WPT applications such as medical implants. For that matter, we maintain the size of the receiver and increase the transmission distance by increasing the transmitter’s size. Hence, the transmission distance should be related to the effective geometrical area that result in the mutual coupling between the TX and the RX; hence, we calculate the transmission distance as A T X × A R X = DoT X × DoR X (4.18) h = Ae f f = 2 2 where A T X = DoT X and A R X = DoR X are the areas of the TX and RX, respectively. According to (4.18), by applying the design process in Fig. 4.6, we increase the transmission distance to 38 mm for a transmitter and receiver sizes of 50 × 50 mm2 and 30 × 30 mm2 , respectively. The performance is summarized in Table 4.2. Similar to the symmetric case, the design parameters of the TX/RX spirals are selected to maximize the obtainable efficiency [8]. Figure 4.9 shows the fabricated asymmetric WPT system and the measurement setup using VNA. The measured and simulated performances shown in Fig. 4.10 have a good agreement. The used capacitors values in the measurements are C PT X = 54 pF, C PR X = 23 pF, C ST X = 23 pF, and C SR X = 9 pF, which agree with the computed values in Table 4.2. The agreement between the analytical model, the circuit simulations, the EM simulations, and the measured performance is an additional proof to the proposed design procedure. In summary, a simple design method for the resonant inductive coupled symmetric and asymmetric WPT systems have been proposed. This technique models a WPT as a second-order bandpass filter (BPF). In this approach, mutual coupling between the TX and RX is used to compute the values of the J-inverters. After that, we extract the required circuit components from the J-inverters. An asymmetric WPT system is fabricated based on the proposed design method. The measured performance of the fabricated asymmetric WPT system is in good agreement with the simulated one. We achieve a WPT efficiency of 75% within a transmission distance of 38 mm for the asymmetric WPT system. The sizes of the TX and RX of this system are 50 × 50 mm2 and 30 × 30 mm2 , respectively. A comparison between the different design methods that have been presented in the chapter is shown in Table 4.3. The comparison distinguishes between the different methods in terms of the used technique, the consumed time, and finally the applicability for asymmetric systems.
84
4 Design Methods
Table 4.2 Summary of the designed, simulated and optimized parameters and performance of the asymmetric WPT system Method
Analytical
Simulations (ADS)
Optimized (HFSS)
f 0 (MHz)
50
49.94
50
DoT X × DoT X (mm2 )
50 × 50
–
30 × 30
DoR X × DoR X (mm2 )
30 × 30
–
30 × 30
l FT X = l FR X (mm)
–
–
5
W FT X = W FR X (mm)
–
–
1
g FT X
=
g FR X
(mm)
h (mm)
–
–
2.5
38
–
38
N1 = N2
2
–
2
WT X , W R X (mm)
3, 2.5
–
3, 2.5
sT X , s R X (mm)
1, 1
–
1, 1
L T X , L R X (nH)
339, 149
339, 149
–
M (nH), k
9.6, 0.043
9.6, 0.043
–
RT X , R R X ()
0.5, 0.3
0.5, 0.3
–
C PT X , C ST X ,
(pF)
23.5, 54.8
23.5, 54.8
23.5, 55
(pF)
9.6, 14
9.6, 14
9, 12.5
–
77.3%
77.4%
C PR X C SR X
ηWPT (%)
RX
TX
Bottom layers
TX
RX Top layers
Fig. 4.9 Measurement setup of the fabricated asymmetric WPT system
References
85 0 -5
|S-Parameters|
Fig. 4.10 Comparison between the circuit (ADS), EM (HFSS) simulations, and the measured performance of the fabricated asymmetric WPT system
-10
|S21|
-15
-1
-20
-2
ADS HFSS Measured
-25 -30 45
-1.5
|S11|
46
47
48
49
-2.5 49 50
51
52
50 53
51 54
55
Frequency (MHz)
Table 4.3 Comparison between the different design methods for resonant inductive WPT systems Traditional method
Design method #1 [3]
Design method #2 (proposed)
Technique
Depends on iterations to find the design values
Models the system using admittance (J-) inverters then find the design values by achieving the impedance matching
Models the system as a second-order BPF using admittance inverters then find the design values that achieve the filter equation
Time consumption
Very long
Short
Very short
Applicability for asymmetric systems
Very hard with a lot of assumptions
Simple, assumptions are required
Very simple and no assumptions are needed
References 1. A.P. Sample, D.T. Meyer, J.R. Smith, Analysis, experimental results, and range adaptation of magnetically coupled resonators for wireless power transfer. IEEE Trans. Ind. Electron. 58(2), 544–554 (2011) 2. S. Cheon, Y.-H. Kim, S.-Y. Kang, M.L. Lee, J.-M. Lee, T. Zyung, Circuit-model-based analysis of a wireless energy-transfer system via coupled magnetic resonances. IEEE Trans. Ind. Electron. 58(7), 2906–2914 (2011) 3. J. Lee, Y.-S. Lim, W.-J. Yang, S.-O. Lim, Wireless power transfer system adaptive to change in coil separation. IEEE Trans. Antennas Propag. 62(2), 889–897 (2014) 4. L. Chen, S. Liu, Y.C. Zhou, T.J. Cui, An optimizable circuit structure for high-efficiency wireless power transfer. IEEE Trans. Ind. Electron. 60(1), 339–349 (2013) 5. D.M. Pozar, Microwave Engineering (Wiley 2009) 6. S. Hekal, Adel B. Abdel-Rahman, A. Allam, H. Jia, A. Barakat, R.K. Pokharel, Asymmetric wireless power transfer systems using coupled DGS resonators. IEICE Electron. Exp. 13(21), 20160591 (2016)
86
4 Design Methods
7. G.L. Matthaei, L. Young, E.M.T. Jones, Microwave Filters, Impedance-Matching Networks, and Coupling Structure (Artech House, 1980) 8. U.-M. Jow, M. Ghovanloo, Design and optimization of printed spiral coils for efficient transcutaneous inductive power transmission. IEEE Trans. Biomed. Circuits Syst. 1(3), 193–202 (2007)
Chapter 5
Future Directions
5.1 Summary The world nowadays is dominated by portable electronic devices. Furthermore, researchers and industrial organizations are energetically exploring new portable devices, particularly devices that combine more functions, and devices (sensors) that monitor the world around us, such as the Internet of Things (IoT) technology. The essential demand for all electronic devices is the source of electrical power. With the ubiquity of portable electronic devices with low power needs, the power demands of each device through charging cables are becoming inevitable. However, the huge number of portable devices increases the number of charging wires leading to greater complexity, user inconvenience, increased costs, and decreased system sustainability. This book aims to resolve this issue by proposing new efficient and low-profile design for short-range WPT systems using quasi-lumped elements based on defected ground structures (DGS). Chapter 2 discussed the different techniques of WPT (near field and far field) using bulky lumped elements and printed resonators, then presented the theory behind using the defected ground structures (DGS) for low-profile bandpass Filters (BPF), and band-stop filters (BSF). DGSs exhibit BPF characteristics by introducing a discontinuity in the feeding microstrip line above the DGS’s slots and adding stubs for matching [1]. In the same manner, band-pass characteristics appear when two DGS resonators are coupled back to back, and power can be transferred from source to load through the DGS resonators [1]. As outlined in Chap. 3, different shapes of DGSs like (H, semi H, and spiral strips) have been analyzed and fabricated to design highly efficient and low-profile DGS-WPT systems. In the proposed DGS-WPT system, electrical coupling is used for feeding. Thus, the limitations of the feeding loop, used in inductive feeding, are avoided, and a higher achievable unloaded Q-factor is possible with compact sizes and higher power transfer efficiency [1]. The circuit model of the proposed WPT system is derived by extracting the equivalent circuit parameters (RLC) using quasi© Springer Nature Singapore Pte Ltd. 2019 S. Hekal et al., Compact Size Wireless Power Transfer Using Defected Ground Structures, Energy Systems in Electrical Engineering, https://doi.org/10.1007/978-981-13-8047-1_5
87
88
5 Future Directions
static modeling, and using another method by fitting the scattering (S-) parameters versus frequency response from full-wave EM simulator using Butterworth-type low-pass response. This design method developed a unique asymmetric size WPT system. The asymmetric system is implemented using very compact size RX that can be embedded in the portable devices or biomedical implants then wirelessly charged by large size TX. As detailed in Chap. 4, two design methods were applied and implemented to extract the optimum design parameters (geometric dimensions and values of lumped elements). The first method is the iterative optimization method which is very slow. The second one is the circuit analysis method using the admittance (J-) inverters which is fast, but applicable to symmetric WPT systems only. A novel design method was developed that is applicable for symmetric and asymmetric WPT systems. This method provided a general design approach by modeling the proposed WPT system as a second-order BPF.
5.2 Future Directions The book presented novel low-profile designs that are suitable for efficient shortrange WPT systems within 1–10 cm according to the resonator sizes. However, Chap. 1 presented some of the vital applications of wireless power transfer like biomedical implants, wireless sensor networks, portable electronic devices, etc., which require, more mobility and longer wireless charging distances. This vision can be achieved through the implementation of one of the following ideas: • Implementation of long-range WPT by concentrating on Wireless charging using scavenging of ambient signals like RF signals of mobile communications and WiFi signals (2.4 and 5 GHz). • Building novel long-range WPT systems embedded in mobile communication base stations and WiFi routers/access points. These systems are responsible for assigning power signal channels to the defined users taking into consideration the specific absorption rate (SAR) specifications to be safe for human health. • Implementation of mid-range WPT by building complete WPT systems that are composed of large transmitters embedded in the walls, the ceiling, and furniture, and compact receivers embedded in the portable electronic devices or the biomedical implants. The transmitters are 3D large wire coils stuck into fixed places that are one or two meters away from the multiple receivers like mobile phones, laptops, etc. • Mid-range WPT can also be implemented using Multi-Band Adaptive Near-Field Focusing by using, for example, dual-band (900 MHz and 2.45 GHz)-phased array antennas that exhibit adaptive near-field focusing for highly efficient wireless power transfer of portable electronic devices and biomedical implants. The proposed system allows multi-users to charge their appliances simultaneously by assignment power transfer via two frequency channels. Hence, we are using space-
5.2 Future Directions
89
Smart antennas system
Beacon signal
Beacon signal
User 1
Beacon signal
User 4
User 3
Beacon signal User 2
Fig. 5.1 The proposed WPT system using dual-band adaptive near-field focusing
Fig. 5.2 The difference in the received RF power density between the implementation of near-field focusing and far-field focusing using 8 × 8 array of single-band antennas [6]
division multiple access (SDMA) and frequency-division multiple access (FDMA) at the same time, as shown in Fig. 5.1. The proposed WPT system can be installed in the ceiling of homes, restaurants, supermarkets, etc. By employing NFF, more power will be transmitted before the effective isotropic radiated power (EIRP) limit is reached due to the inherent far-field diffusing [2–5]. As discussed in [2–5], and the studies performed in [6], we can get a narrower beam and an increase of 48% in the received power density to the target as shown in Fig. 5.2.
90
5 Future Directions Single band adaptive near-field focusing
(a) θ1
Weights of phase shifters W0 . . . . .
Power signal
W1
+
WM-1
Array Antenna θ1
DOA Estimation MUSIC algorithm
Beamformer MMSE Technique
The proposed Dual band adaptive near-field focusing
(b)
Weights of phase shifters
θ1 ∑ ∑ θ2
∑
Array of Dual band Antenna Elements
W0 W1
. . . . .
F1
+ WM-1
Power signal
Weights of phase shifters W0 W1
Incident beacon signals
MUSIC algorithm
F2 Power signal
WM-1
Directed power signals DOA Estimation
+
θ1
θ2
θ1
θ2 Beamformer MMSE Technique
Fig. 5.3 Implementation of adaptive near-field focusing for a Single band. b Dual band
Thanks to the usage of dual-band antennas as array elements, we expect the following results: 1. Multi-user WPT system. 2. High WPT efficiency (one user can receive more power via two channels). The schematic diagram of the proposed system is shown in Fig. 5.3b. Once the battery level decreases, the mobile will send a beacon signal to request power channel. The direction of arrival (DOA) and localization of targets is performed using MUSIC algorithm. The system will assign power channel slots (F1 , F2 , or both). The DSP platform will focus power at the targets using MMSE beamforming technique. Every two minutes, DOA estimation will be performed to locate the mobile handsets.
References
91
References 1. S. Hekal, Adel B. Abdel-Rahman, A. Allam, H. Jia, A. Barakat, R.K. Pokharel, Asymmetric wireless power transfer systems using coupled DGS resonators. IEICE Electron. Express 13(21), 20160591 (2016) 2. M. Bogosanovic, A.G. Williamson, Microstrip antenna array with a beam focused in the nearfield zone for application in noncontact microwave industrial inspection. IEEE Trans. Instrum. Meas. 56(6), 2186–2195 (2007) 3. Y.D. Huang, M. Barkat, Near-field multiple source localization by passive sensor array. IEEE Trans. Antennas Propag. 39, 968–975 (1991) 4. A. Buffi, P. Nepa, G. Manara, Design criteria for near-field focused planar arrays. IEEE Antennas Propag. Mag. 54(1), 40–50 (2012) 5. R. Van der Linden, H.J. Visser, Analysis, design and realization of a near-field focused RF power transfer system. J. Phys. Conf. Ser. 476(1), 012118 (2013). IOP Publishing 6. B.A. Mouris, T.A. Ali, I.A. Eshrah, A. Badawi, Adaptive near-field focusing for wireless power transfer applications. Master Thesis, Cairo University, Egypt (2016)